CN1893337A

CN1893337A - Emission diversity method for time-domain orthogonal frequency dividing duplexing system

Info

Publication number: CN1893337A
Application number: CNA2005100121280A
Authority: CN
Inventors: 杨知行; 王劲涛; 潘长勇; 宋健; 王军
Original assignee: Tsinghua University
Current assignee: Tsinghua University
Priority date: 2005-07-08
Filing date: 2005-07-08
Publication date: 2007-01-10
Anticipated expiration: 2025-07-08
Also published as: CN100553187C

Abstract

The invention belongs to the field of digital information transmission. The steps are: 1) Divide the frequency-domain input sequence into odd and even subsequences according to the subcarrier numbers, both of length N/2; 2) Perform inverse discrete Fourier transform on the above subsequences of two consecutive frames, and convert the 3) Calculate the buffered data to obtain the time-domain signals of the four transmission links; 4) Insert different PN sequences as frame headers in the TDS-OFDM guard intervals of the four transmission links, and combine the frame header and the upper The frame bodies obtained in one step form the complete signal frames of the respective transmission links; 5) The signal frames are subjected to shaping filtering, digital-to-analog conversion, and front-end processing, and are transmitted through four antennas in predetermined channel bandwidths. The invention is simple, fast and accurate, maintains the transmission efficiency of the system, is more suitable for time and frequency dual selective channels, supports "soft failure", increases system reliability, and is easy to transplant to other systems.

Description

Transmit Diversity Method for Time Domain Synchronous Orthogonal Frequency Division Multiplexing System

技术领域technical field

本发明属于数字信息传输技术领域，更具体地涉及一种时域同步正交频分复用(TimeDomain Synchronous OFDM，TDS-OFDM)系统中基于空时分组编码(Space-Time Block Code)的时频域联合发射分集方法。The invention belongs to the technical field of digital information transmission, and more specifically relates to a time-frequency signal based on a space-time block code (Space-Time Block Code) in a Time Domain Synchronous Orthogonal Frequency Division Multiplexing (TimeDomain Synchronous OFDM, TDS-OFDM) system. domain joint transmit diversity method.

背景技术Background technique

在复杂的无线环境中，周围的物体(如房屋、建筑物或树木等)对无线电波会起到反射的作用。这些障碍物会产生幅度衰减和相位延迟不同的反射波。如果发射一个调制信号，那么该发射信号的多个反射波就会从不同方向经过不同传播延迟到达接收天线。这些反射信号经过位于各处的接收机天线接收后，根据其随机相位的不同，对接收信号会起到加强或减弱的作用。由此会造成接收端信号的幅度变化，形成衰落。统计表明，在障碍物均匀的城市街道或森林环境中，信号包络起伏近似于满足Rayleigh分布，故多径快衰落又称为Rayleigh衰落。短期快衰落是由于收发信号双方的相对运动而产生：多径信号的存在造成时间扩散，从而引起传输信号的符号间干扰；而相对运动造成的多普勒效应会引起传输信号的相位迅速变化，在不同的测试环境下有不同的快衰落特性。In a complex wireless environment, surrounding objects (such as houses, buildings or trees, etc.) will reflect radio waves. These obstacles produce reflected waves with different amplitude attenuation and phase delay. If a modulated signal is transmitted, multiple reflections of the transmitted signal arrive at the receiving antenna from different directions with different propagation delays. After these reflected signals are received by receiver antennas located at various places, they will strengthen or weaken the received signal according to the difference of their random phases. This will cause the amplitude of the signal at the receiving end to change, forming fading. Statistics show that in urban streets or forest environments with uniform obstacles, the signal envelope fluctuations approximate to satisfy the Rayleigh distribution, so multipath fast fading is also called Rayleigh fading. The short-term fast fading is caused by the relative motion of the sending and receiving signals: the existence of multipath signals causes time diffusion, which causes inter-symbol interference of the transmitted signal; and the Doppler effect caused by the relative motion causes the phase of the transmitted signal to change rapidly. There are different fast fading characteristics in different test environments.

除了接收信号的瞬时值会出现快速Rayleigh衰落之外，场强中值也会出现缓慢变化。变化的原因主要有两个方面：一是由移动环境中的固定障碍物(如建筑物、山丘、森林等)的阴影效应引起的；二是由于气象条件的变化，导致大气相对介电常数的垂直梯度发生缓变，即电波折射系数随时间变化，从而多径传播到达固定接收点的信号时延也随之变化。这种由阴影效应和气象原因引起的信号变化，称为慢衰落。特别地，由气象原因引起的变化较小，通常忽略。慢衰落环境下的接收信号近似服从对数正态分布，变化幅度取决于障碍物状况、工作频率、变化速率、障碍物和接收机移动速度等。In addition to the fast Rayleigh fading of the instantaneous value of the received signal, there is also a slow change in the median value of the field strength. There are two main reasons for the change: one is caused by the shadow effect of fixed obstacles (such as buildings, hills, forests, etc.) in the mobile environment; the other is that due to changes in meteorological conditions, the atmospheric relative permittivity The vertical gradient of the signal changes slowly, that is, the refraction coefficient of the radio wave changes with time, so the signal delay of multipath propagation to the fixed receiving point also changes accordingly. This kind of signal change caused by shadow effect and meteorological reasons is called slow fading. In particular, changes due to meteorological causes are small and usually ignored. The received signal in a slow fading environment approximately obeys the logarithmic normal distribution, and the range of change depends on the obstacle status, operating frequency, rate of change, obstacle and receiver moving speed, etc.

在多径移动接收中，多径效应引起的时延扩展和多普勒效应引起的多普勒频率展宽同时存在，称为频率和时间选择性衰落信道。在这种信道下，接收信号的信噪比很不稳定，当信道处于深度衰落中时接收信噪比低，判决错误的概率就大，严重降低信号传输的可靠性。为了提高系统的抗衰落性能，可以采用各种信道均衡技术、正交频分复用(OFDM)多载波调制技术等。而分集技术是克服频率和时间选择性衰落的有效技术，它将相同信息经过几个不相关的衰落信道，然后对接收信号进行合成。因为几个信道同时处于深衰落的概率较低，因此可以达到平滑信道衰落，增加信噪比，改善接收机误码特性的目的。在数字电视地面广播网络中，由于分集技术降低了接收机的信噪比(SNR)门限要求，因此在同样的发射功率下，还可以扩展电视信号的覆盖范围。In multipath mobile reception, the time delay spread caused by multipath effect and the Doppler frequency broadening caused by Doppler effect exist at the same time, which is called frequency and time selective fading channel. Under this kind of channel, the signal-to-noise ratio of the received signal is very unstable. When the channel is in deep fading, the received signal-to-noise ratio is low, and the probability of judgment error is high, which seriously reduces the reliability of signal transmission. In order to improve the anti-fading performance of the system, various channel equalization techniques, Orthogonal Frequency Division Multiplexing (OFDM) multi-carrier modulation techniques, etc. can be used. Diversity technology is an effective technology to overcome frequency and time selective fading. It passes the same information through several uncorrelated fading channels, and then synthesizes the received signals. Because the probability that several channels are in deep fading at the same time is low, it can achieve the purpose of smooth channel fading, increase signal-to-noise ratio, and improve receiver error characteristics. In the digital TV terrestrial broadcasting network, because the diversity technology reduces the signal-to-noise ratio (SNR) threshold requirement of the receiver, it can also expand the coverage of TV signals under the same transmission power.

传统的信号分集技术如时间分集、频率分集等，它们的作用实际上相当于信道编码中的重复编码加交织技术，虽然能够改善系统的误码性能，但由于相同信息的重复传送，往往要牺牲较大的的传输效率。另一个常用分集方式是采用多重天线进行空间分集，这种技术在发射端或接收端都可以实现，分别称为发射分集和接收分集。Traditional signal diversity techniques, such as time diversity and frequency diversity, are actually equivalent to repetitive coding and interleaving techniques in channel coding. Although they can improve the bit error performance of the system, they often have to be sacrificed due to the repeated transmission of the same information. Greater transmission efficiency. Another commonly used diversity method is to use multiple antennas for space diversity. This technology can be implemented at either the transmitting end or the receiving end, and are called transmit diversity and receive diversity respectively.

其中，使用多个接收天线的接收分集是一种传统而有效的分集技术，它不需要牺牲传输效率，在接收端可以采用最大值切换、最大比例合并等简单方式完成多个接收信号的选择或合并，然后再按照常规方法进行译码和判决。2001年，法国Harris公司的研究人员进行了欧洲地面数字视频广播DVB-T的接收天线分集实验(Faria G.Mobile DVB-T using antennadiversity receivers.2001.Available： Http：∥www.broadcastpapers.com)，在各种复杂多径环境下，效果很好，测试样机的平均SNR门限下降约6dB，抗多普勒能力增加100％。该接收分集方案如图l所示，接收端采用两组独立的射频前端以及OFDM解调和信道估计模块，经过OFDM解调后，两路接收信号在第k个子载波上的样值分别为：Among them, receiving diversity using multiple receiving antennas is a traditional and effective diversity technology. It does not need to sacrifice transmission efficiency. At the receiving end, simple methods such as maximum switching and maximum ratio combining can be used to complete the selection or selection of multiple receiving signals. Merge, and then decode and judge according to the conventional method. In 2001, researchers from the French company Harris carried out the receiving antenna diversity experiment of European terrestrial digital video broadcasting DVB-T (Faria G.Mobile DVB-T using antenna diversity receivers.2001.Available: Http:∥www.broadcastpapers.com ), In various complex multipath environments, the effect is very good, the average SNR threshold of the test prototype is reduced by about 6dB, and the anti-Doppler ability is increased by 100%. The receiving diversity scheme is shown in Figure 1. The receiving end uses two sets of independent RF front-ends and OFDM demodulation and channel estimation modules. After OFDM demodulation, the sample values of the two received signals on the kth subcarrier are respectively:

$\{\begin{matrix} {R R}_{Rx Rx 11} ((k k)) = = {H h}_{Rx Rx 11} ((k k)) X x ((k k)) + + {N N}_{Rx Rx 11} ((k k)) \\ {R R}_{Rx Rx 22} ((k k)) = = {H h}_{Rx Rx 22} ((k k)) X x ((k k)) + + {N N}_{Rx Rx 22} ((k k)) \end{matrix} ((00 \leq \leq k k \leq \leq N N - - 11))$

其中，H_Rx1(k)和H_Rx2(k)分别为两条接收路径第k个子载波上的频率响应值，N_Rx1(k)和N_Rx2(k)则分别表示相应路径上的噪声。Among them, _HRx1 (k) and _HRx2 (k) are the frequency response values on the kth subcarrier of the two receiving paths, respectively, and _NRx1 (k) and _NRx2 (k) represent the noise on the corresponding path, respectively.

假设两路接收信号经历了互不相关的信道衰落且信道估计结果正确，那么最大比率合并是最好的信号合并方式。将两组接收信号分别乘以其子载波频率响应值的共轭再相加，得到：Assuming that the two received signals have experienced uncorrelated channel fading and the channel estimation result is correct, then the maximum ratio combination is the best signal combination method. Multiply the two groups of received signals by the conjugate of their subcarrier frequency response values and add them together to get:

$R R ((k k)) = = {H h}_{Rx Rx 11}^{* *} ((k k)) {R R}_{Rx Rx 11} ((k k)) + + {H h}_{Rx Rx 22}^{* *} ((k k)) {R R}_{Rx Rx 22} ((k k))$

$= = (({| | {H h}_{Rx Rx 11} ((k k)) | |}^{22} + + {| | {H h}_{Rx Rx 22} ((k k)) | |}^{22})) X x ((k k)) + + {H h}_{Rx Rx 11}^{* *} ((k k)) {N N}_{Rx Rx 11} ((k k)) + + {H h}_{Rx Rx 22}^{* *} ((k k)) {N N}_{Rx Rx 22} ((k k))$

其中，上式的合成信号与非分集系统得到的信号形式相同，可以直接用于译码和判决。显然，合并后的信号的信噪比高于两个支路信号的信噪比最大值，因此获得了分集增益。如果采用更多接收天线并按上述方式进行最大比率合并，还可以获得更大增益。Among them, the synthesized signal of the above formula is the same as the signal obtained by the non-diversity system, and can be directly used for decoding and judgment. Obviously, the signal-to-noise ratio of the combined signal is higher than the maximum value of the signal-to-noise ratio of the two branch signals, so a diversity gain is obtained. Even more gain can be obtained by using more receive antennas and performing maximum ratio combining as described above.

接收分集能够获得良好的效果，分集方法也很简单，但在应用于数字电视地面广播(DTTB)领域中时会受到一些限制。一是接收分集需要接收机有多套并行的射频前端处理，增加了接收机的成本和复杂度，这对于广播系统不合算；二是要使各路接收信号不相关，每两个接收天线的距离要为载波波长的10倍量级，在DTTB所处的VHF/UHF频段，此距离约为4～7m，这对于很多移动和便携式接收终端很难实现。相反地，对于发射机来说，上述限制都不成问题，因此，发射分集技术日益成为研究的热点。由于采用发射分集时，在多个发射天线间往往需要进行信号矢量的合理配置以尽量增加分集阶数(Diversity Order)，所以这项技术也被看成是放置在传统信道编码后的又一层“内码”，称为分集编码。其接收端的信号采样是多个发射信号的叠加，需要通过适当的处理方式进行分离和译码。Receive diversity can achieve good results, and the diversity method is also very simple, but it will be subject to some restrictions when applied to the field of digital television terrestrial broadcasting (DTTB). First, receive diversity requires multiple sets of parallel RF front-end processing in the receiver, which increases the cost and complexity of the receiver, which is not cost-effective for broadcast systems; second, to make the received signals of each channel uncorrelated, the The distance should be on the order of 10 times of the carrier wavelength. In the VHF/UHF frequency band where DTTB is located, the distance is about 4-7m, which is difficult to achieve for many mobile and portable receiving terminals. On the contrary, for the transmitter, the above-mentioned limitations are not a problem, therefore, transmit diversity technology has increasingly become a research hotspot. Since when transmit diversity is used, it is often necessary to reasonably configure signal vectors among multiple transmit antennas to maximize the diversity order (Diversity Order), so this technology is also regarded as another layer placed after traditional channel coding. "Inner code" is called diversity coding. The signal sampling at the receiving end is the superposition of multiple transmitted signals, which need to be separated and decoded through appropriate processing methods.

近年来已有很多发射分集的研究成果。在文献“Wittneben A.A new bandwidth efficienttransmit antenna modulation diversity scheme for linear digital modulation.in Proc.of IEEE ICC’93.Geneva，Switzerland：IEEE，1993.1630-1634”和“Winters J.The diversity gain of transmit diversityin wireless system with Rayleigh fading.in Proc.of IEEE ICC’94.New Orleans，LA：IEEE，1994.1121-1125”中采用的分集方式是将相同信号延时发射，形成一个“人为多径”并用类似Rake接收的方式加以合并。文献“Foschini G and Gans M.On limits of wireless communications in afading environment when using multiple antenna.Wireless Personal Communications，1998，6(3)：311-335”介绍了Blast系统中的空时分层码，它是将发射信号分成多路并分别进行传统的信道编码和交织。1998年，Tarokh在“Tarokh V，Seshadri N，and Calderbank A.Space-time codes forhigh data rate wireless communications：performance criterion and code construction.IEEE Trans.onInformation Theory，1998，44(2)：744-765”中介绍了空时格型编码(STTC)的概念，它将信道编码和天线分集联合进行设计，该方法在理论上可以获得最大限度的增益，但是需要改变整个发射系统的设计，而且即使在较少的发射天线数和低阶星座图的情况下，译码复杂度仍然很大。为了解决这个问题，Alamouti于1998年在其经典论文“Alamouti S.A simple transmitdiversity technique for wireless communications.IEEE Trans.on Select Areas in Communications，1998，16(8)：1451-1458”中提出了一种空时分组编码(STBC)方案，在两发射天线系统中应用，其编码构造和译码算法非常简单，同样可以获得分集增益。Tarokh等人在1999年将STBC推广至任意发射天线数的情况，对这种方案给出了理论分析和构造准则。上述这些STBC编码是基于正交(orthogonal)的结构提出的。在复数域上，假设具有k个符号(x₁，x₂，...，x_k)的p×n_T分组编码矩阵为G(x₁，x₂，...，x_k)，G中的元素满足：(1)每一项都是x_i、x_i ^*或其线性组合之一；(2)G^HG＝(|x₁|²+|x₂|²+...+|x_k|²)I_n，I_n为单位矩阵。此码字用于发射天线分集系统时，将连续k个输入符号按照G所示进行编码，矩阵的每一列都表示供一个发射天线发出的符号序列。由于编码矩阵G具有(2)所示的正交性质，使得接收端译码时可以将每个符号分开，对x_i分别进行译码，这样就大大降低了接收端的译码复杂度。In recent years, there have been many research results on transmit diversity. In the literature "Wittneben AA new bandwidth efficienttransmit antenna modulation diversity scheme for linear digital modulation.in Proc.of IEEE ICC'93.Geneva, Switzerland: IEEE, 1993.1630-1634" and "Winters J.The diversity gain of transmit diversity within wireless The diversity method used in Rayleigh fading.in Proc.of IEEE ICC'94.New Orleans, LA: IEEE, 1994.1121-1125" is to delay the transmission of the same signal to form an "artificial multipath" and use it in a way similar to Rake reception merge. The document "Foschini G and Gans M.On limits of wireless communications in afading environment when using multiple antenna.Wireless Personal Communications, 1998, 6(3): 311-335" introduced the space-time layered code in the Blast system, which is Divide the transmitted signal into multiple channels and perform traditional channel coding and interleaving respectively. 1998, Tarokh in "Tarokh V, Seshadri N, and Calderbank A. Space-time codes for high data rate wireless communications: performance criterion and code construction. IEEE Trans. on Information Theory, 1998, 44(2): 744-765" The concept of space-time trellis coding (STTC) is introduced, which combines channel coding and antenna diversity. This method can obtain the maximum gain in theory, but it needs to change the design of the entire transmitting system, and even in less In the case of the number of transmit antennas and the low-order constellation diagram, the decoding complexity is still very large. In order to solve this problem, in 1998, Alamouti proposed a space-time The block code (STBC) scheme is applied in a two-transmission antenna system. Its coding structure and decoding algorithm are very simple, and diversity gain can also be obtained. In 1999, Tarokh et al. extended STBC to the case of any number of transmitting antennas, and provided theoretical analysis and construction criteria for this scheme. The aforementioned STBC codes are proposed based on an orthogonal (orthogonal) structure. On the complex domain, suppose a p×n _T block coding matrix with k symbols (x ₁ , x ₂ , ..., x _k ) is G(x ₁ , x ₂ , ..., x _k ), G The elements in satisfies: (1) each item is one of x _i , x _i ^* or their linear combination; (2) G ^H G = (|x ₁ | ² +|x ₂ | ² +...+ |x _k | ² )I _n , where I _n is an identity matrix. When this code word is used in a transmitting antenna diversity system, the k consecutive input symbols are encoded as shown in G, and each column of the matrix represents a symbol sequence sent by a transmitting antenna. Since the encoding matrix G has the orthogonality shown in (2), the receiving end can separate each symbol when decoding, and decode _xi separately, which greatly reduces the decoding complexity of the receiving end.

定义STBC码字的编码效率为R＝k/p，其中，k为输入符号数，p为编码延时。对于Alamouti提出的2天线发射分集系统，R＝1。但是当发射天线数目(n_T)多于2时，在Tarokh等人的文献中已经证明，基于正交性质设计的STBC码的编码效率小于1，即p＞k，因而损失了有效传输码率。这就意味着要使采用发射分集后的系统仍保持原有的单发射机系统的传输码率，必须增大原有系统所占用的带宽，这对带宽固定的系统(如DTTB)来说是一个很大的缺点，因此引入了基于准正交(quasi-orthogonal)性质设计的STBC码(“Jafarkhani H，A quasi-orthogonalspace-time block code.IEEE Trans.on Communications，2001，49(1)：1-4”)。准正交STBC码放松了正交性条件的约束，虽然降低了一些分集增益，但可以使得编码效率达到1，即使在带宽固定的系统中也能有效应用。在复数域上，仍然假设分组编码矩阵为G，此时p＝k。以Jafarkhani在文献中提出的4发射天线情况为例，对于准正交设计G中的元素，上述性质(1)保留，(2)改为Define the coding efficiency of STBC code word as R=k/p, where k is the number of input symbols, and p is the coding delay. For the 2-antenna transmit diversity system proposed by Alamouti, R=1. However, when the number of transmitting antennas (n _T ) is more than 2, it has been proved in the literature of Tarokh et al. that the coding efficiency of STBC codes designed based on orthogonal properties is less than 1, that is, p>k, thus losing the effective transmission code rate . This means that in order to keep the transmission code rate of the original single-transmitter system in the system using transmit diversity, the bandwidth occupied by the original system must be increased, which is a problem for systems with fixed bandwidth (such as DTTB). Great shortcoming, therefore introduced the STBC code ("Jafarkhani H, A quasi-orthogonal space-time block code.IEEE Trans.on Communications, 2001,49(1): 1 based on quasi-orthogonal) property design -4"). The quasi-orthogonal STBC code relaxes the constraint of the orthogonality condition. Although it reduces some diversity gain, it can make the coding efficiency reach 1, and it can be effectively applied even in the system with fixed bandwidth. In the complex field, it is still assumed that the block coding matrix is G, and p=k at this time. Taking the case of 4 transmitting antennas proposed by Jafarkhani in the literature as an example, for the elements in the quasi-orthogonal design G, the above properties (1) are retained, and (2) is changed to

${G G}^{H h} G G = = [\begin{matrix} a a & 00 & 00 & b b \\ 00 & a a & - - b b & 00 \\ 00 & - - b b & a a & 00 \\ b b & 00 & 00 & a a \end{matrix}]$

其中， $a = Σ_{i = 1}^{4} {| x_{i} |}^{2},$ b＝2Re(x₁x₄ ^*-x₂x₃ ^*)。因此，对于矩阵中的各列向量(V_i，i＝1，2，3，4)来说，可以分为2组：(V₁，V₄)和(V₂，V₃)。每组内的向量之间不正交，但不同组中的向量之间是正交的。在接收端进行最大似然(ML)译码时，可以将判决公式按照组分为两部分，这样运算量仍然很小。in, $a = Σ_{i = 1}^{4} {| x_{i} |}^{2},$ b=2Re(x ₁ x ₄ ^* -x ₂ x ₃ ^* ). Therefore, for each column vector (V _i , i=1, 2, _{3, 4} ) in the matrix, it can be divided into two groups: (V ₁ , V 4 ) and (V ₂ , V ₃ ). Vectors within each group are not orthogonal to each other, but vectors in different groups are. When performing maximum likelihood (ML) decoding at the receiving end, the decision formula can be divided into two parts according to the group, so that the calculation amount is still very small.

由于STBC(正交设计或准正交设计)编码方式具有快速译码的优点，所以很快得到了广泛研究，并迅速由平衰落单载波信道扩展到频率选择性衰落的OFDM信道中，形成了基于OFDM系统的空时分组码。由于在OFDM调制技术中涉及到时域和频域两种信号，因此STBC编码可以分别在时域和频域中进行。Because the STBC (orthogonal design or quasi-orthogonal design) coding method has the advantages of fast decoding, it has been widely studied and quickly extended from flat fading single carrier channels to frequency selective fading OFDM channels, forming the Space-time block codes based on OFDM systems. Since two signals in the time domain and the frequency domain are involved in the OFDM modulation technique, STBC encoding can be performed in the time domain and the frequency domain respectively.

如果在时域输入符号中进行STBC编码(STC-OFDM)，即将连续的k个OFDM符号按照一定的编码格式G所示进行编码(“Lee K and Williams D.A space-time coded transmitter diversitytechnique for frequency selective fading channels.in Proc.IEEE Sensor Array and MultichannelSignal Processing Workshop.Cambridge，MA：IEEE，2000.149-152”)。此时，为了保持传输矩阵G正交(或准正交)的性质，必须假设信道是准静态的，即在相邻的连续k个OFDM符号时间内信道保持不变。这个假设在快衰落信道下会产生很大误差，因而像在DTTB的移动接收等情况下并不适用。If STBC coding (STC-OFDM) is performed in the time-domain input symbols, the continuous k OFDM symbols are coded according to a certain coding format G ("Lee K and Williams D.A space-time coded transmitter diversity technique for frequency selective fading channels.in Proc.IEEE Sensor Array and MultichannelSignal Processing Workshop.Cambridge, MA: IEEE, 2000.149-152"). At this time, in order to maintain the orthogonal (or quasi-orthogonal) property of the transmission matrix G, it must be assumed that the channel is quasi-static, that is, the channel remains unchanged within the time of k consecutive OFDM symbols. This assumption will produce large errors in fast fading channels, so it is not applicable in situations such as mobile reception of DTTB.

STBC编码也可以在频域中进行(SFC-OFDM)，将相邻的k个子载波上的数据编码(“LeeK and Williams D.A space-frequency transmitter diversity technique for OFDM systems.in Proc.IEEE GLOBECOM’00.San Francisco，CA：IEEE，2000，1473-1477”)。这样，为了保持传输矩阵G正交(或准正交)的性质，需要假设相邻的k个子载波上的频率响应相同。虽然SFC-OFDM可以适用于快衰落信道，但同样由于信道假设带来的误差，在频率选择性衰落信道中并不适用。而实际存在的信道环境大多是双选择性(时间和频率)的，因此STC-OFDM和SFC-OFDM系统在实际应用中都会带来较大误差。STBC encoding can also be performed in the frequency domain (SFC-OFDM), encoding data on adjacent k subcarriers ("LeeK and Williams D.A space-frequency transmitter diversity technique for OFDM systems.in Proc.IEEE GLOBECOM'00. San Francisco, CA: IEEE, 2000, 1473-1477"). In this way, in order to maintain the orthogonal (or quasi-orthogonal) property of the transmission matrix G, it is necessary to assume that the frequency responses on adjacent k subcarriers are the same. Although SFC-OFDM can be applied to fast-fading channels, it is not applicable to frequency-selective fading channels due to the error caused by channel assumptions. However, most of the actual channel environments are dual-selective (time and frequency), so STC-OFDM and SFC-OFDM systems will bring large errors in practical applications.

根据数字电视地面广播的工程特点，TDS-OFDM系统采用的发射分集方法应遵循以下设计原则：(1)不牺牲传输效率。分集方案应保证系统保持原有的信息吞吐能力，不能引入冗余，也就是说每个射频信道不会因为增加了分集编码而降低信息传输率，因而在设计多于2发射天线的分集系统时，要采用上述准正交的设计结构；(2)尽量不改动原有发射系统。使分集方案成为发射系统的一个可选“配件”，这样运营者可以根据本地信道特点和覆盖情况等客观实际决定是否需要采用分集，也最大限度降低了发射机成本；(3)接收机算法改动小，复杂度增加少。增加发射天线不可避免地会增加接收机的功能模块，如信道估计和信号合并器等；(4)支持“软失败(Soft Failure)”。所谓支持“软失败”，是指发射天线间的信号矢量配置应当保证，当一个接收路径因某种原因失效时，另一个接收路径仍能使系统正常接收，仅是牺牲了1/2的平均接收功率。因此，这种发射分集技术实际上还增加了系统的可靠性。According to the engineering characteristics of digital TV terrestrial broadcasting, the transmit diversity method adopted by TDS-OFDM system should follow the following design principles: (1) Do not sacrifice transmission efficiency. The diversity scheme should ensure that the system maintains the original information throughput capability and cannot introduce redundancy, that is to say, each radio frequency channel will not reduce the information transmission rate due to the addition of diversity coding, so when designing a diversity system with more than 2 transmit antennas , to adopt the above-mentioned quasi-orthogonal design structure; (2) try not to change the original launch system. Make the diversity scheme an optional "accessory" of the transmission system, so that the operator can decide whether to use diversity according to the objective reality of the local channel characteristics and coverage conditions, and also minimize the cost of the transmitter; (3) Algorithm change of the receiver Small, little increase in complexity. Increasing the transmit antenna will inevitably increase the functional modules of the receiver, such as channel estimation and signal combiner; (4) Support "soft failure (Soft Failure)". The so-called support for "soft failure" means that the signal vector configuration between the transmitting antennas should ensure that when one receiving path fails for some reason, the other receiving path can still enable the system to receive normally, only sacrificing 1/2 of the average receive power. Therefore, this transmit diversity technique actually increases the reliability of the system.

目前世界上数字电视地面广播传输标准主要有三种：美国的ATSC(高级电视系统委员会Advanced Television Systems Committee)、欧洲的DVB-T(地面数字视频地面广播Digital VideoTerrestrial Broadcasting-Terrestrial)和日本的ISDB-T(地面综合业务数字广播Integrated ServiceDigital Broadcasting-Terrestrial，ISDB-T)。我国自1994年起，也开始了高清晰度电视的研究工作。在此背景下，清华大学提出了地面数字多媒体广播(Digital Multimedia Broadcasting forTerrestrial，DMB-T)传输协议。At present, there are three main transmission standards for digital TV terrestrial broadcasting in the world: ATSC (Advanced Television Systems Committee) in the United States, DVB-T (Digital Video Terrestrial Broadcasting-Terrestrial) in Europe and ISDB-T in Japan (Integrated ServiceDigital Broadcasting-Terrestrial, ISDB-T). Since 1994, our country has also started the research work of high-definition television. In this context, Tsinghua University proposed the Digital Multimedia Broadcasting for Terrestrial (DMB-T) transmission protocol.

清华DMB-T中采用的时域同步正交频分复用(TDS-OFDM)调制属于多载波调制技术，但与欧洲DVB-T采用的编码正交频分复用(COFDM)技术不同，在TDS-OFDM系统中没有插入频域导频信号，而是在OFDM的保护间隔中以时域的方式插入了伪随机(PN)序列，用于帧同步、频率同步、定时同步、信道传输特性估计和跟踪相位噪声等。The Time Domain Synchronous Orthogonal Frequency Division Multiplexing (TDS-OFDM) modulation used in Tsinghua DMB-T is a multi-carrier modulation technology, but it is different from the Coded Orthogonal Frequency Division Multiplexing (COFDM) technology used in European DVB-T. In the TDS-OFDM system, the frequency domain pilot signal is not inserted, but a pseudo-random (PN) sequence is inserted in the time domain in the OFDM guard interval, which is used for frame synchronization, frequency synchronization, timing synchronization, and channel transmission characteristic estimation and tracking phase noise etc.

为了实现快速和稳定的同步，清华大学提出的TDS-OFDM传输系统采用了分级帧结构。帧结构的基本单元称为信号帧，如图2所示。200/225个信号帧定义为一个帧群，512个帧群定义为一个超帧。帧结构的顶层称为日帧，由超帧组成。帧群中的每一个信号帧有唯一的帧号，它被编码在帧头的PN序列中。In order to achieve fast and stable synchronization, the TDS-OFDM transmission system proposed by Tsinghua University adopts a hierarchical frame structure. The basic unit of the frame structure is called a signal frame, as shown in Figure 2. 200/225 signal frames are defined as a frame group, and 512 frame groups are defined as a super frame. The top level of the frame structure is called the day frame and consists of superframes. Each signal frame in the frame group has a unique frame number, which is encoded in the PN sequence of the frame header.

TDS-OFDM传输系统的信号帧使用时域同步的正交频分复用调制，或者称为以PN序列为保护间隔的正交频分复用调制。一个信号帧由帧同步和帧体两部分组成，它们具有相同的基带符号率7.56MS/s(1/T)。一个信号帧可以作为一个正交频分复用(OFDM)块。一个OFDM块进一步分成一个保护间隔和一个离散傅里叶逆变换(IDFT)块。对于TDS-OFDM来说，帧同步PN序列作为OFDM的保护间隔，而帧体作为IDFT块，如图3所示。The signal frame of the TDS-OFDM transmission system uses time-domain synchronous OFDM modulation, or is called OFDM with PN sequence as the guard interval. A signal frame is composed of frame synchronization and frame body, which have the same baseband symbol rate of 7.56MS/s (1/T). A signal frame can be regarded as an Orthogonal Frequency Division Multiplexing (OFDM) block. An OFDM block is further divided into a guard interval and an inverse discrete Fourier transform (IDFT) block. For TDS-OFDM, the frame synchronization PN sequence is used as the OFDM guard interval, and the frame body is used as an IDFT block, as shown in Figure 3.

关于DMB-T、TDS-OFDM的相关情况详见授权号为00123597.4名为“地面数字多媒体电视广播系统”、授权号为01115520.5名为“时域同步正交频分复用调制方法”、授权号为ZL01130659.9名为“地面数字多媒体电视广播系统中的帧同步产生方法”，以及授权号为01124144.6名为“正交频分复用调制系统中保护间隔的填充方法”等清华大学申请的中国发明专利。For details about DMB-T and TDS-OFDM, please refer to the authorization number 00123597.4 titled "Terrestrial Digital Multimedia Television Broadcasting System", the authorization number 01115520.5 titled "Time Domain Synchronous Orthogonal Frequency Division Multiplexing Modulation Method", and the authorization number Applied by Tsinghua University for ZL01130659.9 titled "Frame Synchronization Generation Method in Terrestrial Digital Multimedia Television Broadcasting System" and authorization number 01124144.6 titled "Guard Interval Filling Method in Orthogonal Frequency Division Multiplexing Modulation System" Patent.

为了在TDS-OFDM系统中实现发射天线分集，必须满足下列条件：In order to achieve transmit antenna diversity in a TDS-OFDM system, the following conditions must be met:

(1)发射天线之间保持足够的距离，以使到达接收机的各条传输信道统计独立；(1) Keep enough distance between the transmitting antennas so that the statistics of each transmission channel reaching the receiver are independent;

(2)接收端在进行信道估计时能够准确估计出当前时刻每个信道的信道信息；(2) The receiving end can accurately estimate the channel information of each channel at the current moment when performing channel estimation;

(3)通过合适的方法把接收到的多路信号分离出来，使其互不相关，然后将分离出的多路信号合并，获得最大的信噪比。(3) Separate the received multi-channel signals by a suitable method so that they are not correlated with each other, and then combine the separated multi-channel signals to obtain the maximum signal-to-noise ratio.

针对上述背景，本发明提出了一种针对TDS-OFDM系统的基于空时分组编码的时频域联合发射分集方法。Against the background above, the present invention proposes a time-frequency domain joint transmit diversity method based on space-time block coding for TDS-OFDM systems.

发明内容Contents of the invention

本发明的目的在于提供一种时域同步正交频分复用(Time Domain Synchronous OFDM，TDS-OFDM)系统中基于空时分组编码(Space Time Block Code，STBC)的一种时频域联合发射分集方法。The object of the present invention is to provide a time-frequency domain joint transmission based on space-time block coding (Space Time Block Code, STBC) in a time domain synchronous orthogonal frequency division multiplexing (Time Domain Synchronous OFDM, TDS-OFDM) system Diversity method.

本发明针对数字电视地面广播系统中的发射分集问题，提出了一种基于准正交结构STBC编码的时频域联合发射分集方法。本发明提出的发射分集方法框图如图4所示。Aiming at the transmit diversity problem in the digital TV terrestrial broadcasting system, the present invention proposes a time-frequency domain joint transmit diversity method based on quasi-orthogonal structure STBC coding. The block diagram of the transmit diversity method proposed by the present invention is shown in FIG. 4 .

本发明所述的时域同步正交频分复用，即TDS-OFDM，时频域联合发射分集方法，其特征在于，所述方法是一种基于空时分组编码的时频域联合发射分集方法，它在专用数字集成电路中是按照以下步骤依次实现的：The time domain synchronous orthogonal frequency division multiplexing described in the present invention, namely TDS-OFDM, time-frequency domain joint transmission diversity method, is characterized in that, described method is a kind of time-frequency domain joint transmission diversity based on space-time block coding method, which is implemented sequentially in the ASIC in the following steps:

步骤1.记频域输入序列为X(k，l)，其中k表示子载波序号，0≤k≤N-1，N为OFDM系统中的子载波数，l表示信号帧序号，将X(k，l)按照子载波序号分为奇数子序列X_o(k，l)和偶数子序列X_e(k，l)，它们的长度均为N/2；Step 1. Record the frequency domain input sequence as X(k, l), wherein k represents the subcarrier sequence number, 0≤k≤N-1, N is the subcarrier number in the OFDM system, and l represents the signal frame sequence number, X( k, l) is divided into an odd subsequence X _o (k, l) and an even subsequence X _e (k, l) according to the subcarrier number, and their lengths are both N/2;

步骤2.将X_o(k，l)和X_e(k，l)分别作N/2点反离散傅里叶变换，得到的时域序列为x_l ^to(n，l)和x_l ^te(n，l)；Step 2. Perform N/2-point inverse discrete Fourier transform on X _o (k, l) and X _e (k, l) respectively, and the obtained time domain sequences are x _l ^to (n, l) and x _l ^te (n,l);

步骤3.将连续两帧输入数据进行反离散傅里叶变换后的结果x₁ ^to(n，l)、x₁ ^te(n，l)、x₁ ^to(n，l+1)和x₁ ^te(n，l+1)存入到缓存中；Step 3. The results of inverse discrete Fourier transform of two consecutive frames of input data x ₁ ^to (n, l), x ₁ ^te (n, l), x ₁ ^to (n, l+1) and x ₁ ^te (n, l+1) is stored in the cache;

步骤4.然后将缓存中的数据按照下面所述四种情况进行不同运算，分别得到用于四个天线发射所需的时域信号：Step 4. Then perform different operations on the data in the cache according to the following four situations, and obtain the time domain signals required for the four antennas to transmit respectively:

(a)对于第一个发射天线Tx1，时域信号x_Tx1(n，l)、x_Tx1(n，l+1)为(a) For the first transmitting antenna Tx1, the time-domain signals x _Tx1 (n, l), x _Tx1 (n, l+1) are

$\{\begin{matrix} {x x}_{Tx Tx 11} ((n no,, l l)) = = [[{x x}_{11}^{te te} ((n no,, l l)) + + {x x}_{11}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 11} ((n no + + N N / / 22,, l l)) = = [[{x x}_{11}^{te te} ((n no,, l l)) - - {x x}_{11}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 11} ((n no,, l l + + 11)) = = [[{x x}_{11}^{te te} ((n no,, l l + + 11)) + + {x x}_{11}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 11} ((n no + + N N / / 22,, l l + + 11)) = = [[{x x}_{11}^{te te} ((n no,, l l + + 11)) - - {x x}_{11}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22,, \end{matrix} 00 \leq \leq n no \leq \leq N N / / 22 - - 11;;$

其中， $W_{N}^{- n} = e^{j \frac{2 π}{N} n},$ N为OFDM系统中的子载波数；in, $W_{N}^{- no} = e^{j \frac{2 π}{N} no},$ N is the number of subcarriers in the OFDM system;

(b)对于第二个发射天线Tx2，先将缓存中的数据经过空频编码后得到(b) For the second transmitting antenna Tx2, the data in the cache is firstly space-frequency coded to obtain

$\{\begin{matrix} {x x}_{22}^{te te} ((n no,, l l)) = = {x x}_{11}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l)),, \\ {x x}_{22}^{to to} ((n no,, l l)) = = - - {x x}_{11}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l)),, \\ {x x}_{22}^{te te} ((n no,, l l + + 11)) = = {x x}_{11}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)),, \\ {x x}_{22}^{to to} ((n no,, l l + + 11)) = = - - {x x}_{11}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)),, \end{matrix} 00 \leq \leq n no \leq \leq N N / / 22 - - 11;;$

其中，*表示复数共轭运算，(n)_N/2表示对n取模N/2运算，然后得到时域信号x_Tx2(n，l)、x_Tx2(n，l+1)为Among them, * represents the complex conjugate operation, (n) _N/2 represents the modulo N/2 operation to n, and then the time domain signals x _Tx2 (n, l), x _Tx2 (n, l+1) are obtained as

$\{\begin{matrix} {x x}_{Tx Tx 22} ((n no,, l l)) = = [[{x x}_{22}^{te te} ((n no,, l l)) + + {x x}_{22}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 22} ((n no + + N N / / 22,, l l)) = = [[{x x}_{22}^{te te} ((n no,, l l)) - - {x x}_{22}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 22} ((n no,, l l + + 11)) = = [[{x x}_{22}^{te te} ((n no,, l l + + 11)) + + {x x}_{22}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 22} ((n no + + N N / / 22,, l l + + 11)) = = [[{x x}_{22}^{te te} ((n no,, l l + + 11)) - - {x x}_{22}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22,, \end{matrix} 00 \leq \leq n no \leq \leq N N / / 22 - - 11;;$

(c)对于第三个发射天线Tx3，先将缓存中的数据经过空时编码后得到(c) For the third transmitting antenna Tx3, the data in the buffer is space-time coded first to obtain

$\{\begin{matrix} {x x}_{33}^{te te} ((n no,, l l)) = = {x x}_{11}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)),, \\ {x x}_{33}^{to to} ((n no,, l l)) = = {x x}_{11}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)),, \\ {x x}_{33}^{te te} ((n no,, l l + + 11)) = = {- - {x x}_{11}^{te te}}^{* *} (({((- - n no))}_{N N / / 22},, l l)),, \\ {x x}_{33}^{to to} ((n no,, l l + + 11)) = = - - {x x}_{11}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l)),, \end{matrix} 00 \leq \leq n no \leq \leq N N / / 22 - - 11;;$

然后得到时域信号x_Tx3(n，l)、x_Tx3(n，l+1)为Then get the time domain signal x _Tx3 (n, l), x _Tx3 (n, l+1) as

$\{\begin{matrix} {x x}_{Tx Tx 33} ((n no,, l l)) = = [[{x x}_{33}^{te te} ((n no,, l l)) + + {x x}_{33}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 33} ((n no + + N N / / 22,, l l)) = = [[{x x}_{33}^{te te} ((n no,, l l)) - - {x x}_{33}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 33} ((n no,, l l + + 11)) = = [[{x x}_{33}^{te te} ((n no,, l l + + 11)) + + {x x}_{33}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 33} ((n no + + N N / / 22,, l l + + 11)) = = [[{x x}_{33}^{te te} ((n no,, l l + + 11)) - - {x x}_{33}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22,, \end{matrix} 00 \leq \leq n no \leq \leq N N / / 22 - - 11;;$

(d)对于第四个发射天线Tx4，缓存中的数据先经过空时编码，得到如(c)中所示的结果x₃ ^to(n，l)、x₃ ^te(n，l)、x₃ ^to(n，l+1)和x₃ ^te(n，l+1)，然后再经过空频编码得到(d) For the fourth transmit antenna Tx4, the data in the cache is first space-time coded to obtain the results shown in (c) x ₃ ^to (n, l), x ₃ ^te (n, l), x ₃ ^to (n, l+1) and x ₃ ^te (n, l+1), and then through space-frequency coding to obtain

$\{\begin{matrix} {x x}_{44}^{te te} ((n no,, l l)) = = {x x}_{33}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l)),, \\ {x x}_{44}^{to to} ((n no,, l l)) = = - - {x x}_{33}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l)),, \\ {x x}_{44}^{te te} ((n no,, l l + + 11)) = = {x x}_{33}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)),, \\ {x x}_{44}^{to to} ((n no,, l l + + 11)) = = - - {x x}_{33}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)),, \end{matrix} 00 \leq \leq n no \leq \leq N N / / 22 - - 11;;$

最后得到时域信号x_Tx4(n，l)、x_Tx4(n，l+1)为Finally, the time domain signals x _Tx4 (n, l), x _Tx4 (n, l+1) are obtained as

$\{\begin{matrix} {x x}_{Tx Tx 44} ((n no,, l l)) = = [[{x x}_{44}^{te te} ((n no,, l l)) + + {x x}_{44}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 44} ((n no + + N N / / 22,, l l)) = = [[{x x}_{44}^{te te} ((n no,, l l)) - - {x x}_{44}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 44} ((n no,, l l + + 11)) = = [[{x x}_{44}^{te te} ((n no,, l l + + 11)) + + {x x}_{44}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22,, \\ {x x}_{Tx Tx 44} ((n no + + N N / / 22,, l l + + 11)) = = [[{x x}_{44}^{te te} ((n no,, l l + + 11)) - - {x x}_{44}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22,, \end{matrix} 00 \leq \leq n no \leq \leq N N / / 22 - - 11;;$

步骤5.根据时域同步正交频分复用系统信号帧帧头的长度420或945，生成四个不同的相应长度的PN序列；Step 5. Generate four PN sequences of different corresponding lengths according to the length 420 or 945 of the signal frame header of the time-domain synchronous OFDM system;

步骤6.按照时域同步正交频分复用系统的信道帧结构，在发射天线Tx1、Tx2、Tx3和Tx4四个链路的TDS-OFDM保护间隔内分别插入上述不同的PN序列作为帧头，将帧头PN序列和步骤(4)得到的帧体x_Tx1(n，l)、x_Tx2(n，l)、x_Tx3(n，l)、x_Tx4(n，l)分别组成四个发射链路各自完整的信号帧；Step 6. According to the channel frame structure of the time-domain synchronous OFDM system, insert the above-mentioned different PN sequences respectively in the TDS-OFDM guard intervals of the four links of the transmitting antenna Tx1, Tx2, Tx3 and Tx4 as the frame header , the frame header PN sequence and the frame body x _Tx1 (n, l), x _Tx2 (n, l), x _Tx3 (n, l), x _Tx4 (n, l) obtained in step (4) are respectively formed into four The respective complete signal frames of the transmission chains;

步骤7.将上述完整的TDS-OFDM信号进行成形滤波和数模变换处理，然后经过包含频率上变换和功放在内的前端处理，最后分别通过天线Tx1、Tx2、Tx3和Tx4在预定的频道带宽中发射出去，完成发射天线分集。Step 7. The above-mentioned complete TDS-OFDM signal is subjected to shaping filtering and digital-to-analog conversion processing, and then undergoes front-end processing including frequency up-conversion and power amplifier, and finally passes antennas Tx1, Tx2, Tx3 and Tx4 respectively in predetermined channel bandwidths Transmitting out in the center to complete transmit antenna diversity.

本发明所述的时域同步正交频分复用，即TDS-OFDM，时频域联合发射分集方法，其特征在于，所述的时频域联合编码的等效空时分组编码结构用矩阵G表示为：The time-domain synchronous orthogonal frequency division multiplexing of the present invention, namely TDS-OFDM, time-frequency domain joint transmit diversity method, is characterized in that, the equivalent space-time block coding structure matrix of described time-frequency domain joint coding G is expressed as:

$G G = = [\begin{matrix} X x ((22 k k,, l l)) & {X x}^{* *} ((22 k k + + 11,, l l)) & {X x}^{* *} ((22 k k,, l l + + 11)) & X x ((22 k k + + 11,, l l + + 11)) \\ X x ((22 k k + + 11,, l l)) & - - {X x}^{* *} ((22 k k,, l l)) & {X x}^{* *} ((22 k k + + 11,, l l + + 11)) & - - X x ((22 k k,, l l + + 11)) \\ X x ((22 k k,, l l + + 11)) & {X x}^{* *} ((22 k k + + 11,, l l + + 11)) & - - {X x}^{* *} ((22 k k,, l l)) & - - X x ((22 k k + + 11,, l l)) \\ X x ((22 k k + + 11,, l l + + 11)) & - - {X x}^{* *} ((22 k k,, l l + + 11)) & - - {X x}^{* *} ((22 k k + + 11,, l l)) & X x ((22 k k,, l l)) \end{matrix}],, ((00 \leq \leq k k \leq \leq N N / / 22 - - 11));;$

其中，X(k，l)为频域输入序列，k表示子载波序号，l表示OFDM帧序号，这是一个准正交(quosi-orthogonal)设计的空时分组编码(STBC)结构。Among them, X(k, l) is the frequency domain input sequence, k represents the subcarrier sequence number, and l represents the OFDM frame sequence number, which is a space-time block coding (STBC) structure of quasi-orthogonal design.

本发明所述的时域同步正交频分复用，即TDS-OFDM，时频域联合发射分集方法，其特征在于，可以保持原有的单发射机链路(Tx1)基本不变，而只是在其中加入了缓存器，使得网络构造灵活。其他三个发射链路(Tx2、Tx3、Tx4)的时域信号只须对缓存中的数据进行简单处理即可得到，因此，在传输一帧OFDM信号的时间内只须做一次IDFT运算，运算复杂度很小。The time-domain synchronous orthogonal frequency division multiplexing of the present invention, namely TDS-OFDM, time-frequency domain joint transmit diversity method, is characterized in that, can keep original single transmitter link (Tx1) substantially unchanged, and Only a buffer is added to make the network structure flexible. The time-domain signals of the other three transmission links (Tx2, Tx3, Tx4) can be obtained by simply processing the data in the buffer. Therefore, only one IDFT operation is required during the time of transmitting a frame of OFDM signal. The complexity is minimal.

本发明所述的时域同步正交频分复用，即TDS-OFDM，时频域联合发射分集方法，其特征在于，分集系统工作的鲁棒性强，即如果部分传输链路工作不正常，不用对原有系统进行任何修改，接收端仍然可以正常译码，并且误码性能至少不低于单发射机系统下的情况。在本发明中，如果任一个发射链路出现故障，那么其余三个发射链路仍可以组成准正交STBC结构，从而获得分集增益。当只有两个发射链路工作正常时，可以分为三种情况：1)Tx1和Tx3(或Tx2和Tx4)，可以组成一个2天线的正交STC-OFDM结构；2)Tx1和Tx2(或Tx3和Tx4)，可以组成一个2天线的正交SFC-OFDM结构；3)Tx1和Tx4(或Tx2和Tx3)。在1)和2)情况下，系统仍能获得分集增益。The time-domain synchronous orthogonal frequency division multiplexing described in the present invention, that is TDS-OFDM, time-frequency domain joint transmission diversity method, is characterized in that the robustness of the diversity system work is strong, that is, if part of the transmission link is not working properly , without any modification to the original system, the receiving end can still decode normally, and the bit error performance is at least not lower than that of the single-transmitter system. In the present invention, if any transmission link fails, the remaining three transmission links can still form a quasi-orthogonal STBC structure, thereby obtaining diversity gain. When only two transmission links work normally, it can be divided into three cases: 1) Tx1 and Tx3 (or Tx2 and Tx4), which can form a 2-antenna orthogonal STC-OFDM structure; 2) Tx1 and Tx2 (or Tx3 and Tx4), can form a 2-antenna orthogonal SFC-OFDM structure; 3) Tx1 and Tx4 (or Tx2 and Tx3). In the case of 1) and 2), the system can still obtain diversity gain.

同时，本发明所提出的时频域联合发射分集方法不失一般性，可以很方便地移植到其他多载波DTTB系统。本发明所述的发射分集方案并不排斥接收分集，在本发明中可以引入多个接收天线进行接收分集。At the same time, the time-frequency domain joint transmit diversity method proposed by the present invention does not lose generality, and can be easily transplanted to other multi-carrier DTTB systems. The transmit diversity scheme described in the present invention does not exclude receive diversity, and multiple receive antennas can be introduced in the present invention for receive diversity.

下面我们对本发明中提出的时频域联合发射分集方法的原理和性能进行分析。发射分集方法的应用系统结构框图如图5所示。Next, we analyze the principle and performance of the time-frequency domain joint transmit diversity method proposed in the present invention. The structural block diagram of the application system of the transmit diversity method is shown in Fig. 5 .

假设频域输入信号序列为X(k，l)，其中k表示子载波序号(0≤k≤N-1，N为OFDM系统中的子载波数)，l表示信号帧序号。在OFDM系统中，经过反离散Fourier变换(IDFT)，得到的时域信号x_Tx1(n，l)为Assume that the input signal sequence in the frequency domain is X(k, l), where k represents the subcarrier number (0≤k≤N-1, N is the number of subcarriers in the OFDM system), and l represents the signal frame number. In the OFDM system, after inverse discrete Fourier transform (IDFT), the obtained time domain signal x _Tx1 (n, l) is

${x x}_{Tx Tx 11} ((n no,, l l)) = = \frac{11}{N N} {Σ Σ}_{k k = = 00}^{N N - - 11} X x ((k k,, l l)) {W W}_{N N}^{- - nk nk},, ((00 \leq \leq n no \leq \leq N N - - 11))$

其中， $W_{N}^{k} = e^{- j \frac{2 π}{N} k} .$ in, $W_{N}^{k} = e^{- j \frac{2 π}{N} k} .$

将X(k，l)按照子载波序号分为奇数子序列X_o(k，l)和偶数子序列X_e(k，l)，它们的长度均为N/2，若记X_o(k，l)和X_e(k，l)做N/2点IDFT变换的结果为x₁ ^to(n，l)和x₁ ^te(n，l)，则时域信号x_Tx1(n，l)可以改变形式表示为：Divide X(k, l) into odd sub-sequence X _o (k, l) and even sub-sequence X _e (k, l) according to subcarrier numbers, and their lengths are N/2. If write X _o (k , l) and X _e (k, l) do N/2-point IDFT transformation results as x ₁ ^to (n, l) and x ₁ ^te (n, l), then the time domain signal x _Tx1 (n, l) It can be expressed as:

$\{\begin{matrix} {x x}_{Tx Tx 11} ((n no,, l l)) = = [[{x x}_{11}^{te te} ((n no,, l l)) + + {x x}_{11}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 11} ((n no + + N N / / 22,, l l)) = = [[{x x}_{11}^{te te} ((n no,, l l)) - - {x x}_{11}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \end{matrix} ((00 \leq \leq n no \leq \leq N N / / 22 - - 11))$

这个结果x_Tx1(n，l)就是用于第一个发射天线Tx1的时域信号。同时将x_Tx1(n，l)、x_Tx2(n，l)、x_Tx3(n，l)、x_Tx4(n，l)保存到缓存中。This result x _Tx1 (n,l) is the time domain signal for the first transmit antenna Tx1. Simultaneously save x _Tx1 (n, l), x _Tx2 (n, l), x _Tx3 (n, l), x _Tx4 (n, l) into the cache.

同样地，用于第二个发射天线Tx2的时域信号x_Tx2(n，l)和x_Tx2(n，l+1)可以表示为：Likewise, the time-domain signals x _Tx2 (n, l) and x _Tx2 (n, l+1) for the second transmit antenna Tx2 can be expressed as:

$\{\begin{matrix} {x x}_{Tx Tx 22} ((n no,, l l)) = = [[{x x}_{22}^{te te} ((n no,, l l)) + + {x x}_{22}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 22} ((n no + + N N / / 22,, l l)) = = [[{x x}_{22}^{te te} ((n no,, l l)) - - {x x}_{22}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 22} ((n no,, l l + + 11)) = = [[{x x}_{22}^{te te} ((n no,, l l + + 11)) + + {x x}_{22}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 22} ((n no + + N N / / 22,, l l + + 11)) = = [[{x x}_{22}^{te te} ((n no,, l l + + 11)) - - {x x}_{22}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22 \end{matrix} ((00 \leq \leq n no \leq \leq N N / / 22 - - 11))$

公式中的x₂ ^te(n，l)，x₂ ^to(n，l)，x₂ ^te(n，l+1)和x₂ ^to(n，l+1)是由缓存中的信号经过空频编码(SFC)后得到的：The x ₂ ^te (n, l), x ₂ ^to (n, l), x ₂ ^te (n, l+1) and x ₂ ^to (n, l+1) in the formula are obtained by passing the empty Obtained after frequency coding (SFC):

$\{\begin{matrix} {x x}_{22}^{te te} ((n no,, l l)) = = {x x}_{11}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l)) \\ {x x}_{22}^{to to} ((n no,, l l)) = = - - {x x}_{11}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l)) \\ {x x}_{22}^{te te} ((n no,, l l + + 11)) = = {x x}_{11}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)) \\ {x x}_{22}^{to to} ((n no,, l l + + 11)) = = - - {x x}_{11}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)) \end{matrix} ((00 \leq \leq n no \leq \leq N N / / 22 - - 11))$

其中，*表示复数共轭运算，(n)_N/2表示对n取模N/2运算。根据离散Fourier变换(DFT)的运算性质，可得x₂ ^te(n，l)，x₂ ^to(n，l)，x₂ ^te(n，l+1)和x₂ ^to(n，l+1)的N/2点DFT变换结果可以写为：Among them, * represents complex conjugate operation, and (n) _N/2 represents modulo N/2 operation on n. According to the operation properties of discrete Fourier transform (DFT), x ₂ ^te (n, l), x ₂ ^to (n, l), x ₂ ^te (n, l+1) and x ₂ ^to (n, l+ 1) The N/2-point DFT transformation result can be written as:

$\{\begin{matrix} {x x}_{22}^{te te} ((n no,, l l)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {X x}_{o o}^{* *} ((k k,, l l)) \\ {x x}_{22}^{to to} ((n no,, l l)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} - - {X x}_{e e}^{* *} ((k k,, l l)) \\ {x x}_{22}^{te te} ((n no,, l l + + 11)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {X x}_{o o}^{* *} ((k k,, l l + + 11)) \\ {x x}_{22}^{to to} ((n no,, l l + + 11)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} - - {X x}_{e e}^{* *} ((k k,, l l + + 11)) \end{matrix} ((00 \leq \leq n no,, k k \leq \leq N N / / 22 - - 11))$

因此，对于第二个发射链路，其等效频域输入信号为Therefore, for the second transmit chain, its equivalent frequency-domain input signal is

X_Tx2＝[X^*(1)，-X^*(0)…X^*(2k+1)，-X^*(2k)…X^*(N-1)，-X^*(N-2)](0≤k≤N/2-1)X _Tx2 ＝[X ^* (1),-X ^* (0)...X ^* (2k+1),-X ^* (2k)...X ^* (N-1),-X ^* (N-2)]( 0≤k≤N/2-1)

上式所示的性质与OFDM帧号无关，所以在括号中省略了第二项l。The nature shown in the above formula has nothing to do with the OFDM frame number, so the second item l is omitted in the brackets.

对于第三个发射链路Tx3，要先将缓存中的信号经过空时编码(STC)，可得For the third transmission link Tx3, the signal in the buffer should first be subjected to space-time coding (STC), which can be obtained

$\{\begin{matrix} {x x}_{33}^{te te} ((n no,, l l)) = = {x x}_{11}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)) \\ {x x}_{33}^{to to} ((n no,, l l)) = = {x x}_{11}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)) \\ {x x}_{33}^{te te} ((n no,, l l + + 11)) = = {- - {x x}_{11}^{te te}}^{* *} (({((- - n no))}_{N N / / 22},, l l)) \\ {x x}_{33}^{to to} ((n no,, l l + + 11)) = = - - {x x}_{11}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l)) \end{matrix} ((00 \leq \leq n no \leq \leq N N / / 22 - - 11))$

同理，应用DFT变换的运算性质，x₃ ^te(n，l)、x₃ ^to(n，l)、x₃ ^te(n，l+1)和x₃ ^to(n，l+1)的N/2点DFT变换的结果可以表示为In the same way, applying the operation properties of DFT transformation, x ₃ ^te (n, l), x ₃ ^to (n, l), x ₃ ^te (n, l+1) and x ₃ ^to (n, l+1) The result of N/2 point DFT transformation can be expressed as

$\{\begin{matrix} {x x}_{33}^{te te} ((n no,, l l)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {X x}_{e e}^{* *} ((k k,, l l + + 11)) \\ {x x}_{33}^{to to} ((n no,, l l)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {X x}_{o o}^{* *} ((k k,, l l + + 11)) \\ {x x}_{33}^{te te} ((n no,, l l + + 11)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {- - X x}_{e e}^{* *} ((k k,, l l)) \\ {x x}_{33}^{to to} ((n no,, l l + + 11)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} - - {X x}_{o o}^{* *} ((k k,, l l)) \end{matrix} ((00 \leq \leq n no,, k k \leq \leq N N / / 22 - - 11))$

因此，第三个发射链路的等效频域输入信号为Therefore, the equivalent frequency-domain input signal of the third transmit chain is

$\{\begin{matrix} {X x}_{Tx Tx 33} ((k k,, l l)) = = {X x}^{* *} ((k k,, l l + + 11)) \\ {X x}_{Tx Tx 33} ((k k,, l l + + 11)) = = - - {X x}^{* *} ((k k,, l l)) \end{matrix} ((00 \leq \leq k k \leq \leq N N - - 11))$

进一步，可得时域信号x_Tx3(n，l)、x_Tx3(n，l+1)为Further, the available time domain signals x _Tx3 (n, l), x _Tx3 (n, l+1) are

$\{\begin{matrix} {x x}_{Tx Tx 33} ((n no,, l l)) = = [[{x x}_{33}^{te te} ((n no,, l l)) + + {x x}_{33}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 33} ((n no + + N N / / 22,, l l)) = = [[{x x}_{33}^{te te} ((n no,, l l)) - - {x x}_{33}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 33} ((n no,, l l + + 11)) = = [[{x x}_{33}^{te te} ((n no,, l l + + 11)) + + {x x}_{33}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 33} ((n no + + N N / / 22,, l l + + 11)) = = [[{x x}_{33}^{te te} ((n no,, l l + + 11)) - - {x x}_{33}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22 \end{matrix} ((00 \leq \leq n no \leq \leq N N / / 22 - - 11))$

然后，考虑第四个发射链路，将经过空时编码后得到的数据x₃ ^te(n，l)、x₃ ^to(n，l)、x₃ ^te(n，l+1)和x₃ ^to(n，l+1)再经过空频编码可得Then, considering the fourth transmission chain, the data x ₃ ^te (n, l), x ₃ ^to (n, l), x ₃ ^te (n, l+1) and x ₃ te (n, l+1) obtained after space-time coding ^to (n, l+1) can be obtained by space-frequency coding

$\{\begin{matrix} {x x}_{44}^{te te} ((n no,, l l)) = = {x x}_{33}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l)) \\ {x x}_{44}^{to to} ((n no,, l l)) = = - - {x x}_{33}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l)) \\ {x x}_{44}^{te te} ((n no,, l l + + 11)) = = {x x}_{33}^{to to}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)) \\ {x x}_{44}^{to to} ((n no,, l l + + 11)) = = - - {x x}_{33}^{te te}^{* *} (({((- - n no))}_{N N / / 22},, l l + + 11)) \end{matrix} ((00 \leq \leq n no \leq \leq N N / / 22 - - 11))$

同样根据DFT变换运算的性质，x₄ ^te(n，l)、x₄ ^to(n，l)、x₄ ^te(n，l+1)和x₄ ^to(n，l+1)的N/2点DFT变换的结果可以表示为Also ^according _to _the nature of ^the ^DFT transformation operation, ^the _N _/ The result of 2-point DFT transformation can be expressed as

$\{\begin{matrix} {x x}_{44}^{te te} ((n no,, l l)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {X x}_{Tx Tx 33}^{o o}^{* *} ((k k,, l l)) = = {X x}_{o o} ((k k,, l l + + 11)) \\ {x x}_{22}^{to to} ((n no,, l l)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {- - X x}_{Tx Tx 33}^{e e}^{* *} ((k k,, l l)) = = - - {X x}_{e e} ((k k,, l l + + 11)) \\ {x x}_{22}^{te te} ((n no,, l l + + 11)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {X x}_{Tx Tx 33}^{o o}^{* *} ((k k,, l l + + 11)) = = - - {X x}_{o o} ((k k . . l l)) \\ {x x}_{22}^{to to} ((n no,, l l + + 11)) \overset{DFT DFT ((N N / / 22))}{&RightArrow; &Right Arrow;} {- - X x}_{Tx Tx 33}^{e e}^{* *} ((k k,, l l + + 11)) = = {X x}_{e e} ((k k . . l l)) \end{matrix} ((00 \leq \leq n no,, k k \leq \leq N N / / 22 - - 11))$

式中，X_Tx3 ^o(k，l)、X_Tx3 ^e(k，l)分别为x₃ ^to(n，l)、x₃ ^te(n，l)的N/2点DFT变换的结果。上式即为第四个发射链路的等效频域输入信号。In the formula, X _Tx3 ^o (k, l), X _Tx3 ^e (k, l) are the N/2-point DFT transformation results of x ₃ ^to (n, l), x ₃ ^te (n, l), respectively. The above formula is the equivalent frequency domain input signal of the fourth transmit link.

最后，得到时域信号x_Tx4(n，l)、x_Tx4(n，l+1)为Finally, the time domain signals x _Tx4 (n, l), x _Tx4 (n, l+1) are obtained as

$\{\begin{matrix} {x x}_{Tx Tx 44} ((n no,, l l)) = = [[{x x}_{44}^{te te} ((n no,, l l)) + + {x x}_{44}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 44} ((n no + + N N / / 22,, l l)) = = [[{x x}_{44}^{te te} ((n no,, l l)) - - {x x}_{44}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 44} ((n no,, l l + + 11)) = = [[{x x}_{44}^{te te} ((n no,, l l + + 11)) + + {x x}_{44}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 44} ((n no + + N N / / 22,, l l + + 11)) = = [[{x x}_{44}^{te te} ((n no,, l l + + 11)) - - {x x}_{44}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22 \end{matrix} ((00 \leq \leq n no \leq \leq N N / / 22 - - 11))$

从上面的分析中我们可以得到四个发射链路的等效频域输入编码矩阵为From the above analysis, we can get the equivalent frequency-domain input coding matrix of the four transmission chains as

这是一个准正交(quosi-orthogonal)设计的STBC结构。This is a quasi-orthogonal (quosi-orthogonal) designed STBC structure.

不失一般性，假设在接收端只有一个接收机，以使得分析简单。为了在接收端分别得到四个发射链路的信道估计结果，在TDS-OFDM系统的保护间隔内要分别插入不同的PN序列作为帧头。在下面的分析中，假设接收机可以得到每条发射链路的准确信道信息，各个发射天线的信号经过的信道是不相关的，并且是加性信道，用一个接收天线收到的信号是各个发射信号的叠加。经过OFDM解调后，接收信号在第2k、2k+1个子载波上的样值可以表示为：Without loss of generality, it is assumed that there is only one receiver at the receiving end to make the analysis simple. In order to obtain the channel estimation results of the four transmission links at the receiving end, different PN sequences are inserted as frame headers in the guard interval of the TDS-OFDM system. In the following analysis, it is assumed that the receiver can obtain accurate channel information of each transmit link, the channels passed by the signals of each transmit antenna are irrelevant and additive channels, and the signals received by one receive antenna are each Superposition of transmitted signals. After OFDM demodulation, the samples of the received signal on the 2k and 2k+1 subcarriers can be expressed as:

$\{\begin{matrix} {R R}_{l l} ((22 k k)) = = {H h}_{11,, l l} ((22 k k)) X x ((22 k k,, l l)) + + {H h}_{22,, l l} ((22 k k)) {X x}^{* *} ((22 k k + + 11,, l l)) + + \\ {H h}_{33,, l l} ((22 k k)) {X x}^{* *} ((22 k k,, l l + + 11)) + + {H h}_{44,, l l} ((22 k k)) X x ((22 k k + + 11,, l l + + 11)) + + {η η}_{l l} ((22 k k)) \\ {R R}_{l l} ((22 k k + + 11)) = = {H h}_{11,, l l} ((22 k k + + 11)) X x ((22 k k + + 11,, l l)) - - {H h}_{22,, l l} ((22 k k + + 11)) {X x}^{* *} ((22 k k,, l l)) + + \\ {H h}_{33,, l l} ((22 k k + + 11)) {X x}^{* *} ((22 k k + + 11,, l l + + 11)) - - {H h}_{44,, l l} ((22 k k + + 11)) X x ((22 k k,, l l + + 11)) + + {η η}_{l l} ((22 k k + + 11)) \\ {R R}_{l l + + 11} ((22 k k)) = = {H h}_{11,, l l + + 11} ((22 k k)) X x ((22 k k,, l l + + 11)) + + {H h}_{22,, l l + + 11} ((22 k k)) {X x}^{* *} ((22 k k + + 11,, l l + + 11)) - - \\ {H h}_{33,, l l + + 11} ((22 k k)) {X x}^{* *} ((22 k k,, l l)) - - {H h}_{44,, l l + + 11} ((22 k k)) X x ((22 k k + + 11,, l l)) + + {η η}_{l l + + 11} ((22 k k)) \\ {R R}_{l l + + 11} ((22 k k + + 11)) = = {H h}_{11,, l l + + 11} ((22 k k + + 11)) X x ((22 k k + + 11,, l l + + 11)) - - {H h}_{22,, l l + + 11} ((22 k k + + 11)) {X x}^{* *} ((22 k k,, l l + + 11)) - - \\ {H h}_{33,, l l + + 11} ((22 k k + + 11)) {X x}^{* *} ((22 k k + + 11,, l l)) + + {H h}_{44,, l l + + 11} ((22 k k + + 11)) X x ((22 k k + + l l)) + + {η η}_{l l + + 11} ((22 k k + + 11)) \end{matrix},, ((00 \leq \leq k k \leq \leq N N / / 22 - - 11))$

式中，H_i，l表示在第l个时间段内(一个时间段表示传输一个完整的OFDM帧所用的时间，下同)第i个传输链路的复值信号响应向量，η_l表示在第l个时间段内复值加性白高斯噪声(AWGN)向量。In the formula, H _{i, l} represent the complex-valued signal response vector of the i-th transmission link within the lth time period (a time period represents the time used to transmit a complete OFDM frame, the same below), and η _l represents the Complex-valued additive white Gaussian noise (AWGN) vector for the lth time period.

假设在相邻的两个时间段以及相邻的两个子载波之间的信道响应近似相同，即Assuming that the channel response between two adjacent time periods and adjacent two subcarriers is approximately the same, that is

H_i(k)＝H_i，l(2k)≈H_i，l(2k+1)≈H_i，l+1(2k)≈H_i，l+1(2k+1)H _i (k)=H _i,l (2k)≈H _i,l (2k+1)≈H _i,l+1 (2k)≈H _i,l+1 (2k+1)

(i＝1，2，3，4 0≤k≤N/2-1) (i=1, 2, 3, 4 0≤k≤N/2-1)

则接收信号可以简化表示为Then the received signal can be simplified as

$[\begin{matrix} {R R}_{l l} ((22 k k)) \\ {R R}_{l l}^{* *} ((22 k k + + 11)) \\ {R R}_{l l + + 11}^{* *} ((22 k k)) \\ {R R}_{l l + + 11} ((22 k k + + 11)) \end{matrix}] = = [\begin{matrix} {H h}_{11} ((k k)) & {H h}_{22} ((k k)) & {H h}_{33} ((k k)) & {H h}_{44} ((k k)) \\ - - {H h}_{22}^{* *} ((k k)) & {H h}_{11}^{* *} ((k k)) & - - {H h}_{44}^{* *} ((k k)) & {H h}_{33}^{* *} ((k k)) \\ - - {H h}_{33}^{* *} ((k k)) & - - {H h}_{44}^{* *} ((k k)) & {H h}_{11}^{* *} ((k k)) & {H h}_{22}^{* *} ((k k)) \\ {H h}_{44} ((k k)) & - - {H h}_{33} ((k k)) & - - {H h}_{22} ((k k)) & {H h}_{11} ((k k)) \end{matrix}] [\begin{matrix} X x ((22 k k,, l l)) \\ {X x}^{* *} ((22 k k + + 11,, l l)) \\ {X x}^{* *} ((22 k k,, l l + + 11)) \\ X x ((22 k k + + 11,, l l + + 11)) \end{matrix}] + + [\begin{matrix} {η η}_{l l} ((22 k k)) \\ {η η}_{l l}^{* *} ((22 k k + + 11)) \\ {η η}_{l l + + 11}^{* *} ((22 k k)) \\ {η η}_{l l + + 11} ((22 k k + + 11)) \end{matrix}],, ((00 \leq \leq k k \leq \leq N N / / 22 - - 11))$

记信道响应矩阵为H(k)，即Note that the channel response matrix is H(k), namely

$H h ((k k)) = = [\begin{matrix} {H h}_{11} ((k k)) & {H h}_{22} ((k k)) & {H h}_{33} ((k k)) & {H h}_{44} ((k k)) \\ - - {H h}_{22}^{* *} ((k k)) & {H h}_{11}^{* *} ((k k)) & - - {H h}_{44}^{* *} ((k k)) & {H h}_{33}^{* *} ((k k)) \\ - - {H h}_{33}^{* *} ((k k)) & - - {H h}_{44}^{* *} ((k k)) & {H h}_{11}^{* *} ((k k)) & {H h}_{22}^{* *} ((k k)) \\ {H h}_{44} ((k k)) & - - {H h}_{33} ((k k)) & - - {H h}_{22} ((k k)) & {H h}_{11} ((k k)) \end{matrix}] ((00 \leq \leq k k \leq \leq N N / / 22 - - 11))$

将接收信号表达式的两端均左乘H(k)的Hermit变换矩阵H^H(k)，可得最终的信号估计值X′(2k，l)、X′(2k+1，l)、X′(2k，l+1)和X′(2k+1，l+1)为：Multiply both ends of the received signal expression to the left by the Hermit transformation matrix H ^H (k) of H (k), and the final signal estimation values X′(2k,l), X′(2k+1,l), X'(2k, l+1) and X'(2k+1, l+1) are:

$[\begin{matrix} {X x}^{' '} ((22 k k,, l l)) \\ {X x}^{' ' * *} ((22 k k + + 11,, l l)) \\ {X x}^{' ' * *} ((22 k k,, l l + + 11)) \\ {X x}^{' '} ((22 k k + + 11,, l l + + 11)) \end{matrix}] = = [\begin{matrix} a a ((k k)) & 00 & 00 & b b ((k k)) \\ 00 & a a ((k k)) & - - b b ((k k)) & 00 \\ 00 & - - b b ((k k)) & a a ((k k)) & 00 \\ b b ((k k)) & 00 & 00 & a a ((k k)) \end{matrix}] [\begin{matrix} X x ((22 k k,, l l)) \\ {X x}^{* *} ((22 k k + + 11,, l l)) \\ {X x}^{* *} ((22 k k,, l l + + 11)) \\ X x ((22 k k + + 11,, l l + + 11)) \end{matrix}] + + [\begin{matrix} {η η}_{l l}^{' '} ((22 k k)) \\ {η η}_{l l}^{' ' * *} ((22 k k + + 11)) \\ {η η}_{l l + + 11}^{' '}^{* *} ((22 k k)) \\ {η η}_{l l + + 11}^{' '} ((22 k k + + 11)) \end{matrix}] ((* *)),, ((00 \leq \leq k k \leq \leq N N / / 22 - - 11))$

其中， $a (k) = Σ_{i = 1}^{4} {| H_{i} (k) |}^{2},$ b(k)＝2Re(H₁(k)H₄ ^*(k)-H₂(k)H₃ ^*(k))，η_l′仍然是复值加性白高斯噪声(AWGN)向量。in, $a (k) = Σ_{i = 1}^{4} {| h_{i} (k) |}^{2},$ b(k)=2Re(H ₁ (k)H ₄ ^* (k)−H ₂ (k)H ₃ ^* (k)), η _l ' is still a complex-valued additive white Gaussian noise (AWGN) vector.

可见，在接收端进行最大似然(ML)译码时，可以将判决式分为两部分：(X(2k，l)，X(2k+1，l+1))和(X(2k+1，l)，X(2k，l+1))，这样可以使得译码的运算量大大减小。从上面的分析中还可以看到，采用本发明所提出的发射分集结构虽然分集增益会有所减小，但保持编码效率为1。It can be seen that when performing maximum likelihood (ML) decoding at the receiving end, the decision formula can be divided into two parts: (X(2k, l), X(2k+1, l+1)) and (X(2k+ 1, l), X(2k, l+1)), which can greatly reduce the computational complexity of decoding. It can also be seen from the above analysis that although the diversity gain will be reduced by adopting the transmit diversity structure proposed by the present invention, the coding efficiency is kept at 1.

根据本发明提出的时频域联合发射分集方法结构(图4)，利用DFT变换的运算性质，在上面的分析中可以看到，四个发射链路的时域信号x_Tx1(n，l)、x_Tx2(n，l)、x_Tx3(n，l)、x_Tx4(n，l)都可以由缓存中的数据x₁ ^to(n，l)、x₁ ^te(n，l)、x₁ ^to(n，l+1)和x₁ ^te(n，l+1)通过简单的复数乘法和加法计算出来，也即在一个OFDM帧的时间段内平均做一次N点DFT运算以及3N/2次复数乘法和3N次复数加法。而在传统的STC-OFDM(“Lee K and Williams D.A space-time coded transmitterdiversity technique for frequency selective fading channels.in Proc.IEEE Sensor Array andMultichannel Signal Processing Workshop.Cambridge，MA：IEEE，2000.149-152”)和SFC-OFDM(“Lee K and Williams D.A space-frequency transmitter diversity technique for OFDM systems.inProc.IEEE GLOBECOM’00.SanFrancisco，CA：IEEE，2000，1473-1477”)系统中，在一个OFDM帧的时间段内平均做四次N点DFT运算。因此，当OFDM系统的子载波数N足够大时，本发明提出的方法的运算量约为原有方法的1/4。特别地，以TDS-OFDM系统为例，N＝3780，应用此方法运算量可以减少约70％。According to the time-frequency domain joint transmit diversity method structure (Fig. 4) that the present invention proposes, utilize the computing character of DFT transformation, can see in the above analysis, the time-domain signal x _Tx1 (n, l) of four transmission chains , x _Tx2 (n, l), x _Tx3 (n, l), x _Tx4 (n, l) can all be obtained from the data in the cache x ₁ ^to (n, l), x ₁ ^te (n, l), x ₁ ^to (n, l+1) and x ₁ ^te (n, l+1) are calculated by simple complex multiplication and addition, that is, an N-point DFT operation and 3N/ 2 complex multiplications and 3N complex additions. In the traditional STC-OFDM ("Lee K and Williams DA space-time coded transmitter diversity technique for frequency selective fading channels.in Proc.IEEE Sensor Array and Multichannel Signal Processing Workshop.Cambridge, MA: IEEE, 2000.149-152") and SFC -OFDM ("Lee K and Williams DA space-frequency transmitter diversity technique for OFDM systems.inProc.IEEE GLOBECOM'00.SanFrancisco, CA: IEEE, 2000, 1473-1477") system, within the time period of one OFDM frame Do four N-point DFT operations on average. Therefore, when the number N of sub-carriers of the OFDM system is large enough, the calculation amount of the method proposed by the present invention is about 1/4 of the original method. In particular, taking the TDS-OFDM system as an example, where N=3780, the calculation amount can be reduced by about 70% by applying this method.

一般地，在传统的STC-OFDM系统中，即将连续的k个OFDM符号按照一定的编码格式G所示进行编码。此时，为了保持传输矩阵G正交(或准正交)的性质，必须假设信道是准静态的，即在相邻的连续k个OFDM符号时间内信道保持不变。这个假设在快衰落信道下会产生很大误差，因而像在DTTB的移动接收等情况下并不适用。而在传统的SFC-OFDM系统中，将相邻的k个子载波上的数据编码。这样，为了保持传输矩阵G正交(或准正交)的性质，需要假设相邻的k个子载波上的频率响应相同。同样由于信道假设带来的误差，在频率选择性衰落信道中并不适用。而实际存在的信道环境大多是双选择性(时间和频率)的，因此STC-OFDM和SFC-OFDM系统在实际应用中都会带来较大误差。本发明提出的基于STBC的时频域联合发射分集方法只须假设相邻的两个时间段以及相邻的两个子载波之间的信道响应近似相同，因此更加适用于实际的时间、频率双选择性信道。Generally, in a traditional STC-OFDM system, k continuous OFDM symbols are coded according to a certain coding format G as shown. At this time, in order to maintain the orthogonal (or quasi-orthogonal) property of the transmission matrix G, it must be assumed that the channel is quasi-static, that is, the channel remains unchanged within the time of k consecutive OFDM symbols. This assumption will produce large errors in fast fading channels, so it is not applicable in situations such as mobile reception of DTTB. However, in a traditional SFC-OFDM system, data on adjacent k subcarriers are coded. In this way, in order to maintain the orthogonal (or quasi-orthogonal) property of the transmission matrix G, it is necessary to assume that the frequency responses on adjacent k subcarriers are the same. Also due to the error caused by the channel assumption, it is not applicable in the frequency selective fading channel. However, most of the actual channel environments are dual-selective (time and frequency), so STC-OFDM and SFC-OFDM systems will bring large errors in practical applications. The STBC-based time-frequency domain joint transmit diversity method proposed by the present invention only needs to assume that the channel responses between two adjacent time periods and adjacent two subcarriers are approximately the same, so it is more suitable for actual time and frequency dual selection sexual channel.

基于上述分析，对本发明所提出的TDS-OFDM系统中基于空时分组编码(STBC)的一种时频域联合发射分集方法进行了计算机仿真，仿真的系统结构与图5相同。在仿真中我们采用表1和表2所示的两种信道模型A和B。其中模型A的多径延时较短，而且回波强度较弱；模型B中引入了具有长延时的强多径(多径6)。Based on the above analysis, a computer simulation of a time-frequency domain joint transmit diversity method based on space-time block coding (STBC) in the TDS-OFDM system proposed by the present invention is carried out. The simulated system structure is the same as that in FIG. 5 . We adopt two kinds of channel models A and B shown in Table 1 and Table 2 in the simulation. Among them, model A has short multipath delay and weak echo intensity; model B introduces strong multipath (multipath 6) with long delay.

表1分集传输信道仿真模型A 模型A 多径1 多径2 多径3 多径4 多径5 多径6 类型幅度(dB)延时(us) Rayleigh00 Rayleigh-120.3 Rayleigh-43.5 Rayleigh-74.4 Rayleigh-159.5 Rayleigh-2212.7 Table 1 Diversity transmission channel simulation model A Model A multipath 1 Multipath 2 Multipath 3 multipath 4 Multipath 5 Multipath 6 Type amplitude (dB) delay (us) Rayleigh00 Rayleigh - 120.3 Rayleigh - 43.5 Rayleigh - 74.4 Rayleigh - 159.5 Rayleigh-2212.7

表2分集传输信道仿真模型B 模型B 多径1 多径2 多径3 多径4 多径5 多径6 类型幅度(dB)延时(us) Rayleigh00 Rayleigh-120.3 Rayleigh-43.5 Rayleigh-74.4 Rayleigh-159.5 Rayleigh030 Table 2 Diversity transmission channel simulation model B Model B multipath 1 Multipath 2 Multipath 3 multipath 4 Multipath 5 Multipath 6 Type amplitude (dB) delay (us) Rayleigh00 Rayleigh - 120.3 Rayleigh - 43.5 Rayleigh - 74.4 Rayleigh - 159.5 Rayleigh030

仿真中采用QPSK星座图和保护间隔为数据长度1/9的3780点TDS-OFDM系统，信道编码采用2/3码率的卷积码。并且假设接收机可以得到每条发射链路的准确信道信息，各个发射天线的信号经过的信道不相关。图6～9给出了不分集、传统的STC-OFDM、SFC-OFDM和本发明所提出的分集方法在不同信道情况下的误比特率(BER)仿真结果。为了使仿真结果具有可比性，图中的信噪比(Signal to Noise Ratio，SNR)是以接收天线为准的，也就是说，发射分集方法中每个发射天线的功率仅为单发射天线方法(不分集)中发射功率的1/4。In the simulation, a QPSK constellation diagram and a 3780-point TDS-OFDM system with a guard interval of 1/9 of the data length are used, and a convolutional code with a code rate of 2/3 is used for channel coding. And it is assumed that the receiver can obtain accurate channel information of each transmit link, and the channels through which the signals of each transmit antenna pass are irrelevant. Figures 6 to 9 show the bit error rate (BER) simulation results under different channel conditions of non-diversity, traditional STC-OFDM, SFC-OFDM and the diversity method proposed by the present invention. In order to make the simulation results comparable, the Signal to Noise Ratio (SNR) in the figure is based on the receiving antenna, that is to say, the power of each transmitting antenna in the transmit diversity method is only (without diversity) 1/4 of the transmit power.

图6为短时延慢衰落信道，采用信道模型A，不加多普勒频移效应。三种发射分集系统相比不分集系统都有很明显的增益，系统性能近似相同。Figure 6 shows a short-time-delay slow-fading channel, using channel model A without Doppler frequency shift effect. Compared with the non-diversity system, the three transmit diversity systems have obvious gains, and the system performance is approximately the same.

图7为短时延快衰落信道下，仍采用信道模型A，最大多普勒频移f_d＝50Hz。由于信道假设误差的影响，STC-OFDM系统已不能提供分集增益，它的性能甚至比不分级系统还要差。而SFC-OFDM系统和本发明所提的分集方案表现出对多普勒频移的不敏感，仍然保持了较高增益。Fig. 7 shows that channel model A is still used under the short-time-delay fast-fading channel, and the maximum Doppler frequency shift f _d =50 Hz. Due to the influence of the channel assumption error, the STC-OFDM system can no longer provide diversity gain, and its performance is even worse than that of the non-hierarchical system. However, the SFC-OFDM system and the diversity scheme proposed by the present invention show insensitivity to Doppler frequency shift, and still maintain a relatively high gain.

图8为长时延慢衰落信道，采用信道模型B，不加多普勒频移效应。可以看出，由于信道假设误差的影响，SFC-OFDM系统已不能提供分集增益。而SFC-OFDM系统和本发明所提的分集方案仍可以提供较高增益。Figure 8 shows a long-time-delay and slow-fading channel, using channel model B without Doppler frequency shift effect. It can be seen that due to the influence of the channel assumption error, the SFC-OFDM system can no longer provide diversity gain. However, the SFC-OFDM system and the diversity scheme proposed by the present invention can still provide relatively high gain.

图9所示信道既有长延时和强多径，又有快衰落，采用信道模型B，最大多普勒频移f_d＝50Hz。同样由于信道假设误差的影响，STC-OFDM和SFC-OFDM系统都已无法提供分集增益，而本发明所提的分集方案仍可以获得较高增益。The channel shown in Fig. 9 has both long delay and strong multipath, and fast fading. Channel model B is adopted, and the maximum Doppler frequency shift f _d =50 Hz. Also due to the influence of channel assumption errors, both STC-OFDM and SFC-OFDM systems are unable to provide diversity gain, but the diversity scheme proposed by the present invention can still obtain relatively high gain.

本发明所提出的TDS-OFDM系统中基于空时分组编码(STBC)的时频域联合发射分集方法分集系统工作的鲁棒性强，支持“软失败(Soft Failure)”，即如果部分传输链路工作不正常，不用对原有系统进行任何修改，接收端仍然可以正常译码，并且误码性能至少不低于单发射机系统下的情况。这是由于，如果某个发射链路无法正常工作，那么在最后的判决公式(*)中，只须将相应的信道响应矩阵中的元素置零即可，仍可以正常进行ML译码。在本发明中，如果任一个发射链路出现故障，那么其余三个发射链路仍可以组成准正交STBC结构，从而获得分集增益。当只有两个发射链路工作正常时，可以分为三种情况：1)Tx1和Tx3(或Tx2和Tx4)，可以组成一个2天线的正交STC-OFDM结构；2)Tx1和Tx2(或Tx3和Tx4)，可以组成一个2天线的正交SFC-OFDM结构；3)Tx1和Tx4(或Tx2和Tx3)。在1)和2)情况下，系统仍能获得分集增益。图10给出了系统中存在不同数目的正常工作的发射机的情况下的计算机仿真结果，仿真时采用信道模型A，不加多普勒频移效应。仿真结果很好的验证了上述分析结果。In the TDS-OFDM system proposed by the present invention, the time-frequency domain joint transmit diversity method based on space-time block coding (STBC) has strong robustness in diversity system work, and supports "soft failure (Soft Failure)", that is, if part of the transmission chain If the channel is not working properly, the receiving end can still decode normally without any modification to the original system, and the bit error performance is at least not lower than that of the single-transmitter system. This is because, if a transmission link fails to work normally, then in the final decision formula (*), it is only necessary to set the elements in the corresponding channel response matrix to zero, and ML decoding can still be performed normally. In the present invention, if any transmission link fails, the remaining three transmission links can still form a quasi-orthogonal STBC structure, thereby obtaining diversity gain. When only two transmission links work normally, it can be divided into three cases: 1) Tx1 and Tx3 (or Tx2 and Tx4), which can form a 2-antenna orthogonal STC-OFDM structure; 2) Tx1 and Tx2 (or Tx3 and Tx4), can form a 2-antenna orthogonal SFC-OFDM structure; 3) Tx1 and Tx4 (or Tx2 and Tx3). In the case of 1) and 2), the system can still obtain diversity gain. Figure 10 shows the computer simulation results under the condition that there are different numbers of normally working transmitters in the system. The channel model A is used in the simulation without Doppler frequency shift effect. The simulation results have well verified the above analysis results.

附图说明Description of drawings

图1接收分集方案框图Figure 1 Receive diversity scheme block diagram

图2为TDS-OFDM系统分级帧结构。Figure 2 shows the hierarchical frame structure of the TDS-OFDM system.

图3为TDS-OFDM系统信号帧结构图。FIG. 3 is a structural diagram of a TDS-OFDM system signal frame.

图4为本发明提出的时频域联合发射分集方法框图。Fig. 4 is a block diagram of the time-frequency domain joint transmit diversity method proposed by the present invention.

图5为本发明提出的发射分集方法的应用系统结构框图。Fig. 5 is a structural block diagram of the application system of the transmit diversity method proposed by the present invention.

图6为本发明对信道模型A的对比仿真结果(f_d＝0Hz)。Fig. 6 is the comparative simulation result of the present invention on channel model A (f _d =0 Hz).

图7为本发明对信道模型A的对比仿真结果(f_d＝50Hz)。Fig. 7 is the comparative simulation result of the present invention on channel model A (f _d =50 Hz).

图8为本发明对信道模型B的对比仿真结果(f_d＝0Hz)。Fig. 8 is the comparative simulation result of the present invention on channel model B (f _d =0 Hz).

图9为本发明对信道模型B的对比仿真结果(f_d＝50Hz)。Fig. 9 is the comparative simulation result of the present invention on channel model B (f _d =50 Hz).

图10为本发明提出的分集系统中存在不同数目的正常工作的发射机的情况下的对比仿真结果。Fig. 10 is a comparative simulation result under the condition that there are different numbers of normally working transmitters in the diversity system proposed by the present invention.

具体实施方式Detailed ways

见图4。本系统中的频域输入序列是经过调制映射(星座图映射)后的复数信号，其可以是非编码信号，也可以是经过编码后的信号。频域输入序列首先按照子载波序号分为奇数子序列和偶数子序列，它们的长度均为N/2。然后将连续两帧输入数据的共四个N/2长的子序列分别作N/2点反离散Fourier变换，其结果x₁ ^to(n，l)、x₁ ^te(n，l)、x₁ ^to(n，l+1)和x₁ ^te(n，l+1)存入到缓存中存入缓存中。利用反离散Fourier变换的运算性质，分下列四种情况对缓存中的数据进行简单的复数乘法和加法计算，分别得到四个发射链路的时域信号：See Figure 4. The frequency-domain input sequence in this system is a complex signal after modulation mapping (constellation map mapping), which can be a non-coded signal or a coded signal. The input sequence in the frequency domain is first divided into odd subsequences and even subsequences according to subcarrier numbers, and their lengths are both N/2. Then, a total of four N/2 long subsequences of two consecutive frames of input data are respectively subjected to N/2 point inverse discrete Fourier transform, and the results are x ₁ ^to (n, l), x ₁ ^te (n, l), x ₁ ^to (n, l+1) and x ₁ ^te (n, l+1) are stored in the cache and stored in the cache. Using the operational properties of the inverse discrete Fourier transform, simple complex multiplication and addition calculations are performed on the data in the cache in the following four cases to obtain the time-domain signals of the four transmission links:

1)对于天线Tx1：1) For antenna Tx1:

$\{\begin{matrix} {x x}_{Tx Tx 11} ((n no,, l l)) = = [[{x x}_{11}^{te te} ((n no,, l l)) + + {x x}_{11}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 11} ((n no + + N N / / 22,, l l)) = = [[{x x}_{11}^{te te} ((n no,, l l)) - - {x x}_{11}^{to to} ((n no,, l l)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 11} ((n no,, l l + + 11)) = = [[{x x}_{11}^{te te} ((n no,, l l + + 11)) + + {x x}_{11}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22 \\ {x x}_{Tx Tx 11} ((n no + + N N / / 22,, l l + + 11)) = = [[{x x}_{11}^{te te} ((n no,, l l + + 11)) - - {x x}_{11}^{to to} ((n no,, l l + + 11)) {W W}_{N N}^{- - n no}]] / / 22 \end{matrix} ((00 \leq \leq n no \leq \leq N N / / 22 - - 11))$

2)对于天线Tx2，先经过空频编码：2) For the antenna Tx2, first undergo space frequency coding:

然后计算得到时域信号：Then calculate the time domain signal:

3)对于天线Tx3，先经过空时编码：3) For the antenna Tx3, first pass the space-time coding:

然后计算得到时域信号：Then calculate the time domain signal:

4)对于天线Tx4，将经过空时编码后的数据再经过空频编码：4) For the antenna Tx4, the space-time coded data is then space-frequency coded:

然后计算得到时域信号：Then calculate the time domain signal:

按照TDS-OFDM系统的信道帧结构，在四个发射链路的TDS-OFDM保护间隔内分别插入不同的PN序列作为帧头，将帧头PN序列和上述步骤中得到的帧体分别组成四个发射链路各自完整的信号帧。然后再将完整的TDS-OFDM信号进行成形滤波和数模变换处理，然后经过频率上变换和功放等前端处理，最后分别通过四个天线在预定的频道带宽中发射出去，完成发射天线分集。According to the channel frame structure of the TDS-OFDM system, different PN sequences are inserted in the TDS-OFDM guard intervals of the four transmission links as frame headers, and the frame header PN sequences and the frame bodies obtained in the above steps are respectively composed of four Transmit the chain's respective complete signal frame. Then the complete TDS-OFDM signal is subjected to shaping filtering and digital-to-analog conversion processing, and then undergoes front-end processing such as frequency up-conversion and power amplifier, and finally transmits it through four antennas in the predetermined channel bandwidth to complete the transmit antenna diversity.

在接收端，假设在相邻的两个时间段以及相邻的两个子载波之间的信道响应近似相同，经过OFDM解调后，接收信号在第2k、2k+1个子载波上的样值可以表示为：At the receiving end, assuming that the channel response between two adjacent time periods and adjacent two subcarriers is approximately the same, after OFDM demodulation, the samples of the received signal on the 2k and 2k+1 subcarriers can be Expressed as:

$[\begin{matrix} {X x}^{' '} ((22 k k,, l l)) \\ {X x}^{' ' * *} ((22 k k + + 11,, l l)) \\ {X x}^{' ' * *} ((22 k k,, l l + + 11)) \\ {X x}^{' '} ((22 k k + + 11,, l l + + 11)) \end{matrix}] = = [\begin{matrix} a a ((k k)) & 00 & 00 & b b ((k k)) \\ 00 & a a ((k k)) & - - b b ((k k)) & 00 \\ 00 & - - b b ((k k)) & a a ((k k)) & 00 \\ b b ((k k)) & 00 & 00 & a a ((k k)) \end{matrix}] [\begin{matrix} X x ((22 k k,, l l)) \\ {X x}^{* *} ((22 k k + + 11,, l l)) \\ {X x}^{* *} ((22 k k,, l l + + 11)) \\ X x ((22 k k + + 11,, l l + + 11)) \end{matrix}] + + [\begin{matrix} {η η}_{l l}^{' '} ((22 k k)) \\ {η η}_{l l}^{' ' * *} ((22 k k + + 11)) \\ {η η}_{l l + + 11}^{' '}^{* *} ((22 k k)) \\ {η η}_{l l + + 11}^{' '} ((22 k k + + 11)) \end{matrix}],, ((00 \leq \leq k k \leq \leq N N / / 22 - - 11))$

其中， $a (k) = Σ_{i = 1}^{4} {| H_{i} (k) |}^{2},$ b(k)＝2Re(H₁(k)H₄ ^*(k)-H₂(k)H₃ ^*(k))，η_l′仍然是复值加性白高斯噪声(AWGN)向量。在进行最大似然(ML)译码时，将判决式分为两部分：(X(2k，l)，X(2k+1，l+1))和(X(2k+1，l)，X(2k，l+1))，这样可以使得译码的运算量大大减小。in, $a (k) = Σ_{i = 1}^{4} {| h_{i} (k) |}^{2},$ b(k)=2Re(H ₁ (k)H ₄ ^* (k)−H ₂ (k)H ₃ ^* (k)), η _l ' is still a complex-valued additive white Gaussian noise (AWGN) vector. When performing maximum likelihood (ML) decoding, the decision formula is divided into two parts: (X(2k, l), X(2k+1, l+1)) and (X(2k+1, l), X(2k, l+1)), which can greatly reduce the computational complexity of decoding.

上面本发明的一般性具体实施例进行了说明，但本发明并不限制于上述实施例，在不脱离本申请的权利要求的精神和范围情况下，本领域的技术人员可作出各种修改或改型。The general specific embodiment of the present invention above has been described, but the present invention is not limited to above-mentioned embodiment, without departing from the spirit and scope of the claims of the application, those skilled in the art can make various modifications or retrofit.

Claims

1, the emission diversity method of time-domain synchronous orthogonal frequency-division multiplexing system is characterized in that, described method is a kind of time domain and frequency domain combined emission diversity method based on space-time block code, and it is realized successively according to following steps in the special digital integrated circuit:

Step 1. note frequency domain list entries be X (k, l), wherein k represents the subcarrier sequence number, 0≤k≤N-1, N are the sub-carrier number in the ofdm system, l represents the signal frame sequence number, (k l) is divided into odd number subsequence X according to the subcarrier sequence number with X _o(k is l) with even number subsequence X for x _e(k, l), their length is N/2;

Step 2. is with X _o(k, l) and X _e(k l) makes N/2 point Inverse Discrete Fourier Transform respectively, and the time domain sequences that obtains is x ₁ ^To(n, l) and x ₁ ^Te(n, l);

Step 3. is imported x as a result after data are carried out Inverse Discrete Fourier Transform with two continuous frames ₁ ^To(n, l), x ₁ ^Te(n, l), x ₁ ^To(n, l+1) and x ₁ ^Te(n l+1) is deposited in the buffer memory;

Step 4. is carried out nonidentity operation with the data in the buffer memory according to following described four kinds of situations then, obtains being used for four required time-domain signals of antenna emission respectively:

(a) for first transmitting antenna Tx1, time-domain signal x _Tx1(n, l), x _Tx1(n l+1) is

\{\begin{matrix} x_{Tx 1} (n, l) = [x_{1}^{te} (n, l) + x_{1}^{to} (n, l) W_{N}^{- n}] / 2, \\ x_{Tx 1} (n + N / 2, l) = [x_{1}^{te} (n, l) - x_{1}^{to} (n, l) W_{N}^{- n}] / 2, \\ x_{Tx 1} (n, l + 1) = [x_{1}^{te} (n, l + 1) + x_{1}^{to} (n, l + 1) W_{N}^{- n}] / 2, \\ x_{Tx 1} (n + N / 2, l + 1) = [x_{1}^{te} (n, l + 1) - x_{1}^{to} (n, l + 1) W_{N}^{- n}] / 2, \end{matrix} 0 \leq n \leq N / 2 - 1;

Wherein,

W_{N}^{- n} = e^{j \frac{2 π}{N} n},

N is the sub-carrier number in the ofdm system;

(b), earlier the data in the buffer memory are obtained through behind the space-frequency coding for second transmitting antenna Tx2

\{\begin{matrix} x_{2}^{te} (n, l) = x_{1}^{{to}^{*}} ({(- n)}_{N / 2}, l), \\ x_{2}^{to} (n, l) = - x_{1}^{t e^{*}} ({(- n)}_{N / 2}, l), \\ x_{2}^{te} (n, l + 1) = x_{1}^{{to}^{*}} ({(- n)}_{N / 2}, l + 1), \\ x_{2}^{to} (n, l + 1) = - x_{1}^{{te}^{*}} ({(- n)}_{N / 2}, l + 1), \end{matrix} 0 \leq n \leq N / 2 - 1;

Wherein, * represents the complex conjugate computing, (n) _N/2Expression obtains time-domain signal x then to n delivery N/2 computing _Tx2(n, l), x _Tx2(n l+1) is

\{\begin{matrix} x_{Tx 2} (n, l) = [x_{2}^{te} (n, l) + x_{2}^{to} (n, l) W_{N}^{- n}] / 2, \\ x_{Tx 2} (n + N / 2, l) = [x_{2}^{te} (n, l) - x_{2}^{to} (n, l) W_{N}^{- n}] / 2, \\ x_{Tx 2} (n, l + 1) = [x_{2}^{te} (n, l + 1) + x_{2}^{to} (n, l + 1) W_{N}^{- n}] / 2, \\ x_{Tx 2} (n + N / 2, l + 1) = [x_{2}^{te} (n, l + 1) - x_{2}^{to} (n, l + 1) W_{N}^{- n}] / 2, \end{matrix} 0 \leq n \leq N / 2 - 1;

(c), earlier the data in the buffer memory are obtained through behind the Space Time Coding for the 3rd transmitting antenna Tx3

\{\begin{matrix} x_{3}^{te} (n, l) = x_{1}^{{te}^{*}} ({(- n)}_{N / 2}, l + 1), \\ x_{3}^{to} (n, l) = x_{1}^{t o^{*}} ({(- n)}_{N / 2}, l + 1), \\ x_{3}^{te} (n, l + 1) = - x_{1}^{{te}^{*}} ({(- n)}_{N / 2}, l), \\ x_{3}^{to} (n, l + 1) = - x_{1}^{{to}^{*}} ({(- n)}_{N / 2}, l), \end{matrix} 0 \leq n \leq N / 2 - 1;

Obtain time-domain signal x then _Tx3(n, l), x _Tx3(n l+1) is

\{\begin{matrix} x_{Tx 3} (n, l) = [x_{3}^{te} (n, l) + x_{3}^{to} (n, l) W_{N}^{- n}] / 2, \\ x_{Tx 3} (n + N / 2, l) = [x_{3}^{te} (n, l) - x_{3}^{to} (n, l) W_{N}^{- n}] / 2, \\ x_{Tx 3} (n, l + 1) = [x_{3}^{te} (n, l + 1) + x_{3}^{to} (n, l + 1) W_{N}^{- n}] / 2, \\ x_{Tx 3} (n + N / 2, l + 1) = [x_{3}^{te} (n, l + 1) - x_{3}^{to} (n, l + 1) W_{N}^{- n}] / 2, \end{matrix} 0 \leq n \leq N / 2 - 1;

(d) for the 4th transmitting antenna Tx4, the data in the buffer memory through Space Time Coding, obtain the x as a result as shown in (c) earlier ₃ ^To(n, l), x ₃ ^Te(n, l), x ₃ ^To(n, l+1) and x ₃ ^Te(n, l+1), and then the process space-frequency coding obtains

\{\begin{matrix} x_{4}^{te} (n, l) = x_{3}^{{to}^{*}} ({(- n)}_{N / 2}, l), \\ x_{4}^{to} (n, l) = {- x}_{3}^{t e^{*}} ({(- n)}_{N / 2}, l), \\ x_{4}^{te} (n, l + 1) = x_{3}^{{to}^{*}} ({(- n)}_{N / 2}, l + 1), \\ x_{4}^{to} (n, l + 1) = - x_{1}^{{te}^{*}} ({(- n)}_{N / 2}, l + 1), \end{matrix} 0 \leq n \leq N / 2 - 1;

Obtain time-domain signal x at last _Tx4(n, l), x _Tx4(n l+1) is

\{\begin{matrix} x_{Tx 4} (n, l) = [x_{4}^{te} (n, l) + x_{4}^{to} (n, l) W_{N}^{- n}] / 2, \\ x_{Tx 4} (n + N / 2, l) = [x_{4}^{te} (n, l) - x_{4}^{to} (n, l) W_{N}^{- n}] / 2, \\ x_{Tx 4} (n, l + 1) = [x_{4}^{te} (n, l + 1) + x_{4}^{to} (n, l + 1) W_{N}^{- n}] / 2, \\ x_{Tx 4} (n + N / 2, l + 1) = [x_{4}^{te} (n, l + 1) - x_{4}^{to} (n, l + 1) W_{N}^{- n}] / 2, \end{matrix} 0 \leq n \leq N / 2 - 1;

Step 5. generates the PN sequence of four different respective length according to the length 420 or 945 of time-domain synchronous orthogonal frequency-division multiplexing system signal frame frame head;

Step 6. is according to the channel frame structure of time-domain synchronous orthogonal frequency-division multiplexing system; in the TDS-OFDM of transmitting antenna Tx1, Tx2, four links of Tx3 and Tx4 protection at interval, insert above-mentioned different PN sequence respectively, the frame x that frame head PN sequence and step (4) are obtained as frame head _Tx1(n, l), x _Tx2(n, l), x _Tx3(n, l), x _Tx4(n l) forms four signal frames that transmitting chain is complete separately respectively;

Above-mentioned complete TDS-OFDM signal is formed filtering to step 7. and digital to analog conversion is handled, then through comprising frequency up-converted and power amplifier in interior front-end processing, launch in predetermined channel bandwidth by antenna Tx1, Tx2, Tx3 and Tx4 respectively at last, finish transmission antenna diversity.

2, the emission diversity method of time-domain synchronous orthogonal frequency-division multiplexing system according to claim 1 is characterized in that, the equivalent space-time block code structure of described time domain and frequency domain combined coding is expressed as with matrix G:

G = [\begin{matrix} X (2 k, l) & X^{*} (2 k + 1, l) & X^{*} (2 k, l + 1) & X (2 k + 1, l + 1) \\ X (2 k + 1, l) & {- X}^{*} (2 k, l) & X^{*} (2 k + 1, l + 1) & - X (2 k, l + 1) \\ X (2 k, l + 1) & X^{*} (2 k + 1, l + 1) & - X^{*} (2 k, l) & - X (2 k + 1, l) \\ X (2 k + 1, l + 1) & {- X}^{*} (2 k, l + 1) & {- X}^{*} (2 k + 1, l) & X (2 k, l) \end{matrix}], 0 \leq k \leq N / 2 - 1;

Wherein, (k l) is the frequency domain list entries to X, and k represents the subcarrier sequence number, and l represents the OFDM frame number, and this is the space-time block code structure of an accurate orthogonal design.

3, the emission diversity method of time-domain synchronous orthogonal frequency-division multiplexing system according to claim 1, it is characterized in that, when the link of any one transmitting antenna broke down, the link of its excess-three transmitting antenna was still formed the space-time block code structure of accurate orthogonal design.

4, the emission diversity method of time-domain synchronous orthogonal frequency-division multiplexing system according to claim 1 is characterized in that, when the link that has only two transmitting antennas is working properly, then has:

(1) Tx1 and Tx3, perhaps Tx2 and Tx4, coded OFDM structure when forming the orthogonal space of one 2 antenna respectively;

(2) Tx1 and Tx2, perhaps Tx3 and Tx4, the orthogonal space of forming one 2 antenna respectively is the coded OFDM structure frequently.