JP2531008B2 - Resonant power supply circuit - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a resonance type power supply circuit in which a voltage pulse is generated by a resonance operation of a primary side of a transformer, the voltage pulse is boosted by the transformer and output from a secondary side of the transformer. .

[0002]

2. Description of the Related Art A resonance type power supply circuit used in a television receiver or a display device incorporates a stabilizing circuit for stabilizing a high voltage output voltage applied from a flyback transformer to a cathode ray tube.

This type of stabilizing circuit is of a type that controls the power supply voltage of the drive power source on the primary side of the flyback transformer, and the correction voltage on the secondary side of the flyback transformer for the amount of drop in the high voltage output voltage. There is a secondary side control system that adds

The method for controlling the power supply voltage on the primary side is as follows:
There is a problem that the control response is extremely poor, and the secondary side control system has a problem that the correction width is narrow and the circuit becomes complicated.

Therefore, recently, in order to solve such a problem as much as possible, a method of directly controlling the primary side current of the flyback transformer is disclosed in Japanese Patent Laid-Open No. 2-2.
No. 22374 is proposed. However, this proposed example has an advantage that the control range of the high-voltage output voltage can be widened, but it requires a plurality of switching elements for controlling the primary side current, which increases the number of parts and complicates the circuit configuration. There is a problem of doing.

[0006]

The present inventor has focused on a basic circuit of a resonance type power supply circuit as shown in FIG. 13 as a circuit of a system for directly controlling the primary side current, and has improved its circuit characteristics. We are working.

In the figure, the positive side of the drive power source 3 is connected to one end side of the primary coil 2 of the flyback transformer 1, and the negative side of the drive power source 3 is connected to the ground.
A transistor 4 as a switching element is connected in series to the other end of the primary coil 2, and a damper diode 5 and a resonance capacitor 6 are connected in parallel to the transistor 4.

The high voltage end of the secondary coil 7 of the flyback transformer 1 is connected to the anode of the cathode ray tube 10 via a high voltage rectifying diode 8.

In this type of circuit, during the ON period of the transistor 4 (transistor period), the current shown in FIG. 14B flows from the drive power source 3 toward the transistor 4 through the primary coil 2 and the primary coil 2 Electromagnetic energy is applied to. Next, when the transistor 4 is turned off, the primary coil 2
And the resonance capacitor 6 start series resonance, the electromagnetic energy stored on the primary coil 2 side is converted into electrostatic energy of the resonance capacitor 6, and as shown in (a) of FIG. 14 , the flyback pulse (voltage Pulse) is generated. The flyback pulse has a peak when all the electromagnetic energy of the primary coil 2 is converted into the electrostatic energy of the resonance capacitor 6.

When the flyback pulse reaches its peak,
This time, the electrostatic energy of the resonance capacitor 6 is converted back into the electromagnetic energy of the primary coil 2, and as a result, the voltage of the flyback pulse gradually decreases. Then, when the pulse voltage is zero, that is, when the voltage at the point A in the circuit of FIG. 13 becomes zero, the damper diode 5 turns on and a reverse current flows from the ground side to the primary coil 2 side, and the voltage at the point A becomes Drive power supply 3
When the power supply voltage is restored, the damper diode 5
Turns off. Then, when the transistor 4 is turned on again, the first circuit operation state is set, and by repeating this operation, the circuit operation is continued. The flyback pulse generated on the primary coil 2 side is boosted by the flyback transformer 1 and applied to the anode of the cathode ray tube 10 via the high voltage rectifying diode 8.

By the way, the pulse voltage V _c of the flyback pulse generated on the primary side of the flyback transformer 1 is
It is represented by the formula of V _c = E _B + r _a sin (ωt−φ _a ).
However, t is time, E _B is the supply voltage of the drive power source 3, and

R _a = √ {E _B ² + (I ₀ / Cω) ² }

Ω = (LC) ^-1/2

Φ _a = tan ⁻¹ (E _B Cω / I ₀ ).

Here, I ₀ is the current flowing through the primary coil 2, C is the capacitance of the resonance capacitor 6, and L is the inductance of the primary coil 2.

As is clear from the above equation of V _c , the voltage peak of the flyback pulse is when ωt−φ _a = π / 2, and at this time, the peak voltage is V _c = E _B + √ {E _B
A _{^{2 + (I 0 / Cω)}} 2}.

The current I ₀ flowing in the primary coil is I ₀
= (E _B / L) t _on + I ₁ . Here, t _on is the time when the transistor 4 is on, and I ₁ is zero, so that I ₀ is proportional to the on period of the transistor 4. Therefore, the flyback pulse crest value can be controlled by controlling the ON period of the transistor 4, and thus the high voltage output voltage output from the secondary coil 7 can be controlled.

In this way, when the ON period of the transistor 4 is controlled to stabilize the high-voltage output voltage, the damper circuit (the ON period of the damper diode 5) is used for the damper from the ground side in the circuit of FIG. 13 as it is. When a reverse current flows through the diode 5 to the primary coil 2 side and the voltage at the point A is restored to the power supply voltage due to the flow of the reverse current and the damper diode 5 is turned off, the transistor 4 is already turned off. A current passing from the power supply 3 side through the primary coil 2 starts to flow through the resonance capacitor 6 to the ground side,
Series resonance occurs between the primary coil 2 and the resonance capacitor 6, and as shown in FIG. 14 (a), an unnecessary pulse P is generated in the interval from the end of the damper period to the next turning on of the transistor 4. _W occurs.

This P _W causes noise and adversely affects the circuit operation. Therefore, in this type of circuit, in order to prevent current from flowing from the primary coil 2 side to the resonance capacitor 6 at the end of the damper period, the drive power supply 3 passes through the primary coil 2 and the resonance capacitor 6 or the transistor 4 to the ground. A switch element for cutting off the current is provided in the path leading to, or the start side of the ON period of the transistor 4 is overlapped with the end side of the damper period as shown by the broken line in FIG. 14 (c). The current from the driving power supply 3 side was made to flow to the transistor 4 side.

However, since the ON period of the transistor 4 overlaps the damper period, the output of the transistor 4 cannot be controlled by the ON timing. In addition, from the drive power source 3 to the primary coil 2 and the resonance capacitor 6 or the transistor 4
Even in the case where a switch element for cutting off a current is provided in the path extending through the path to the ground, a switch element separate from the transistor 4 is provided in order to block the current, and the operation timing of the switch element is performed. Since a control circuit is required, there is a problem that the number of parts is increased, the circuit configuration is complicated, and the circuit cost is also high.

The present invention has been made to solve the above problems, and an object thereof is to control the magnitude of the voltage pulse generated on the primary side by controlling only one switch element, and further, the damper period. An object of the present invention is to provide a simple resonance type power supply circuit having a circuit configuration with a small number of components, which can prevent generation of unnecessary pulses from the end of the above to the ON period of the transistor.

[0022]

In order to achieve the above object, the present invention is constructed as follows. In other words, the
According to the first aspect of the invention, a drive power source is connected to a primary coil of a transformer, and a switching element for controlling on / off of a current flowing through the primary coil on the primary coil side, and a series connection of the primary coil when the switching element is off. A resonance capacitor that generates a voltage pulse by resonance and a damper diode that causes a current that flows in the direction opposite to the direction of the current flowing through the switch element to flow through the primary coil within the damper period are connected , and the current that flows through the primary coil is turned on / off. A resonant power supply circuit in which a switch element to be controlled is composed of one switch element, and the peak value of a voltage pulse generated is controlled by controlling the ON period of the one switch element.
A clamp circuit for clamping the voltage at both ends of the resonant capacitor to the voltage of the driving power source is provided from the end of the damper period to the time when the switch element is turned on. A second aspect of the invention is characterized in that the switch element is connected in series with a diode that reduces the apparent parasitic capacitance of the switch element.

[0023]

In the present invention having the above structure, when the switching element is turned on, a current flows from the drive power source side through the primary coil and the switching element, and electromagnetic energy is stored in the primary coil. When the switch element is turned off, a voltage pulse is generated by the series resonance of the resonance capacitor and the primary coil. After this voltage pulse is generated, the damper period starts, a reverse current flows from the damper diode side to the primary coil side, and after the damper diode turns off at the end point of the damper period, the voltage across the resonant capacitor is The clamp circuit clamps the power supply voltage of the driving power supply. Due to this clamping action, the drive power supply and the both ends of the resonant capacitor have the same potential, so that no current flows from the drive power supply side to the resonant capacitor side through the primary coil, and the next switch from the end of the damper period. By the time the device is turned on, there is no circuit loss due to unnecessary return current, and there is no longer the generation of unnecessary noise-causing voltage pulses.
When the clamp circuit is in the operating state, the switch element is turned on, so that a current flows from the drive power source side to the switch element side through the primary coil, and the first circuit operation state is reached, and the circuit operation is continued.

[0024]

Embodiments of the present invention will be described below with reference to the drawings. In the following description of the embodiments, the same circuit parts as those of the basic circuit described above are designated by the same reference numerals, and the duplicate description thereof will be omitted. FIG. 1 shows the circuit configuration of a first embodiment of a resonant power supply circuit according to the present invention. In the figure, the drive power source 3 is connected to one end side (eg, winding start end side) of the primary coil 2 of the flyback transformer 1, and the other end side (winding end end side) of the primary coil 2 serves as a switching element. MOS FET (field effect transistor) 11
Of the MOS FET 11 is connected to the ground, and the source of the MOS FET 11 is connected to the ground. And MOS FET
A damper diode 5 having a direction opposite to the direction of the current of the MOS FET 11 is connected in parallel with 11. The damper diode 5 may be an external diode connected to an electronic component, but a MOS FE is used as a switch element.
When T11 is used, the MOS FET 11 itself has the diode characteristic in the reverse direction, so that the external diode component can be omitted and the diode characteristic of the MOSFET 11 can function as the damper diode 5.

Further, one end side of the resonance capacitor 6 is connected to the winding end end side of the primary coil 2, and the cathode side of the diode 12 is connected to the other end side of the resonance capacitor 6,
The anode side of the diode 12 is connected to the ground. The anode side of the diode 13 is connected to the connecting portion between the diode 12 and the resonance capacitor 6, and the cathode side of the diode 13 is connected to the connecting portion between the primary coil 2 and the driving power supply 3. The diodes 12 and 13 form a characteristic clamp circuit 14 of this embodiment.

One end of a series circuit of voltage dividing resistors 15 and 16 is connected to the high voltage end side of the secondary coil 7 of the flyback transformer 1, and the voltage dividing resistors 15 and 16 are resistance-divided.
High voltage output voltage is detected. Then, this detection voltage is applied to the non-inverting input terminal of the operational amplifier 17. A reference voltage is applied from the reference power supply 18 to the inverting input terminal side of the operational amplifier 17, and the operational amplifier 17 compares the detected voltage of the high voltage output voltage with the reference voltage and compares the signal corresponding to the drop amount of the high voltage output voltage with the comparator. Add to inverting input terminal of 20.
On the other hand, the signal from the waveform shaping circuit 21 is applied to the non-inverting input terminal of the comparator 20.

The waveform shaping circuit 21 is shown in a horizontal deflection output circuit integrates the horizontal drive signal shown in FIG. 2 that synchronization (not shown) (a) (HD signal) shown in FIG. 2 (b) A ramp waveform is created and the signal of this ramp waveform is applied to the non-inverting input terminal of the comparator 20. The comparator 20 compares the ramp waveform signal with the signal from the operational amplifier 17, and as shown in FIGS. 2B and 2C, rises at the intersection of the output of the operational amplifier and the ramp waveform, and falls of the ramp waveform. That is, a drive signal that falls at the fall of the HD signal is produced. When the drop amount of the high voltage output voltage increases, the output level of the operational amplifier also decreases, and as a result, the pulse width of the drive signal increases. The comparator 20 produces a drive signal having a wider pulse width as the amount of drop of the high voltage output voltage increases, and applies the drive signal to the drive circuit 22. The drive circuit 22 drives the switch of the MOS FET 11 according to the ON pulse width of the drive signal.

This embodiment is constructed as described above, and its operation will now be described with reference to the circuit of FIG. 1 and the time chart of FIG. First, at t ₀ , MOS FE
When T11 is turned on, a current flows from the drive power source 3 side to the primary coil 2 and further to the ground side through the MOS FET 11. The current flowing through the primary coil 2 increases with time as shown in FIG. 2 (e), and electromagnetic current is stored in the primary coil 2 due to the current flow.

Next, when the MOS FET 11 is turned off at t ₁ , the resonance coil 6 and the diode 13 are transferred from the primary coil 2.
A current flows along the route passing through, the LC series resonance between the inductance of the primary coil 2 and the capacitance of the resonance capacitor 6 is started, and a flyback pulse (voltage pulse) is generated.
This flyback pulse becomes maximum when all the electromagnetic energy on the primary coil 2 side is converted into the electrostatic energy of the resonance capacitor 6. After all the electromagnetic energy of the primary coil 2 is transferred to the resonance capacitor 6, this time the diode 12,
A reverse current flows along the route of passing through the resonance capacitor 6 and the primary coil 2 to the drive power supply 3, and the electrostatic energy of the resonance capacitor 6 is inversely converted into the electromagnetic energy of the primary coil 2.

At t ₂ when the flyback pulse is completed, the voltage at the point A in the circuit of FIG. 1 becomes zero. At this time, the damper diode 5 is turned on, and the primary side passes through the damper diode 5 from the ground side. A current flows on the coil 2 side. This reverse current flow causes the voltage at point A to rise and t
_When the potential becomes equal to the power source voltage E _B of the driving power source 3 at 3, the damper diode 5 is turned off. At this time, MOS F
Since ET11 is off, current tends to flow from the drive power source 3 side to the resonance capacitor 6 side, but in the present embodiment, the clamp circuit 14 of the diodes 12 and 13 is provided,
Voltage at both ends of resonant capacitor 6 (voltage at points A and B)
It is clamped to the power supply voltage E _B of both the drive power source 3 E _B
Since it is held at the same potential as the current, no current flows from the primary coil 2 side to the resonance capacitor 6 side, and
The unnecessary pulse voltage P _W that causes noise as shown in FIG. 14A is not generated.

Next, at time t ₄ , when the MOS FET 11 is turned on, the point A is grounded, and the drive power source 3
The current flowing from the primary coil 2 to the ground side through the MOS FET 11 coincides with the initial state of t ₀ . The circuit operation is continued by repeating the operations from t ₀ to t ₄ .

In the present embodiment, as the high-voltage output voltage drops, the on-time of the MOS FET 11 becomes longer, which increases the electromagnetic energy stored in the primary coil 2 and also increases the peak value of the flyback pulse generated. Therefore, the high voltage output voltage is effectively stabilized. Moreover, since the switch element for controlling the high-voltage output voltage is composed of one MOS FET element, the number of parts is very small and the circuit configuration is simple.

From the end of the damper period, the MOS
During the period when the FET 11 is turned on, the voltage across the resonance capacitor 6 is clamped to the same potential as the power supply voltage of the drive power supply 3, so that a current may flow from the drive power supply 3 to the resonance capacitor 6 through the primary coil 2. Therefore, it is possible to prevent the generation of unnecessary pulse voltage P _W that causes noise during this period.

Moreover, the clamp circuit 14 for preventing the voltage pulse P _W has a very simple circuit structure using only the two diodes 12 and 13, and has a switch element for blocking the flow of current, It is not necessary to provide a complicated circuit for controlling the switch element.

Further, in this embodiment, the unnecessary voltage pulse P
For _W the prevention of the occurrence of is performed by the clamp circuit 14, as shown by the broken line in (c) of FIG. 14, MOS
There is no restriction that the ON period of FET11 overlaps the damper period.
The pulse width of the drive signal for switch operation can be expanded to the maximum of the pulse width of the horizontal drive signal, and the voltage can be controlled over an extremely wide range.

Further, in the circuit of this embodiment, the deflection cycle is 1
Since it is a resonance type circuit that charges and discharges every cycle, the response of stabilization of the high voltage output voltage is extremely good, and the control performance of the high voltage stabilization can be significantly improved.

FIG. 3 shows a second embodiment of the present invention. In this embodiment, a series circuit of a deflection yoke D _Y and an S-shaped correction capacitor C _S is connected in parallel with the resonance capacitor 6 to form an integrated type circuit configuration for high voltage generation and deflection drive, and also a comparator 20 and a drive circuit. A pulse width limiter 23 is provided between the circuit 22 and the circuit 22, and the other configuration is the same as that of the first embodiment.

Generally, in a multi-scan type circuit capable of driving deflection over a wide range from low frequency to high frequency, the upper limit voltage of the crest value of the flyback pulse is set on the side where the deflection frequency becomes high frequency. There is. In the circuit of this embodiment, the pulse width of the drive signal can be widened up to the width of the maximum horizontal drive signal. Therefore, when performing multi-scan drive, the pulse width of the drive signal reaches the HD signal at the low frequency drive. When spread, MOS
The ON period of the FET 11 is much longer than that in the high frequency drive, and the current flowing through the primary coil 2 is also large. As a result, the peak value of the generated flyback pulse is much larger than that in the high frequency drive. There is a problem that the peak value of the flyback pulse exceeds the upper limit voltage, that is, the designed upper limit voltage. In this embodiment, in order to avoid this, a pulse width limiter 23 is provided, and the pulse width of the drive signal is limited by limiting the pulse width of the drive signal so as not to exceed the upper limit voltage set on the basis of the high frequency drive even for the low frequency deflection drive. The multi-scan drive in a wide frequency range from high frequency to high frequency can be performed without any trouble.

As a circuit for this multi-scan drive,
As shown in FIG. 4, the resonance capacitor may be configured by a series circuit of 6a and 6b, and the switch 19 may switch the resonance capacitance between low frequency driving and high frequency driving.

FIG. 5 shows a third embodiment of the present invention. This embodiment is characterized in that a transistor 24 of a switch element is connected in parallel to the diode 12 of the clamp circuit 14, and the other structure is the same as that of the first embodiment. When the waveforms of the flyback pulse in the first and second embodiments are exaggeratedly drawn, the result is as shown in FIG. In the first and second embodiments, when the MOS FET 11 is turned off and a current of LC series resonance flows from the drive power source 3 side to the resonance capacitor 6 through the primary coil 2, the diode 12 is in the opposite direction. Since it does not fall to the ground side, when the flyback pulse reaches a peak, it becomes a waveform that is raised by the power supply voltage E _B of the driving power supply 3 more than the peak value when the diode 12 is not present, and the pulse waveform is asymmetrical. Becomes In this embodiment, in order to avoid this, a transistor 24 is provided in parallel with the diode 12 so that the transistor 24 is turned on during the period for creating the flyback pulse pulse. When the transistor 24 is turned on, a current passing from the primary coil 2 side through the resonance capacitor 6 flows through the transistor 24 to the ground side, whereby the peak voltage of the flyback pulse is the power source voltage E _{B of the} driving power source 3.
It is not lifted up by the amount, and as shown in FIG. 6B, a symmetrical flyback pulse waveform can be obtained.

FIG. 7 shows a fourth embodiment of the present invention, and FIG. 8 shows a fifth embodiment of the present invention. Fourth
In the embodiment of FIG.
The diode 26 for apparently reducing the parasitic capacitance of the FET 11 is connected, and in the fifth embodiment, the diode 26 for apparently reducing the parasitic capacitance of the MOS FET 11 is connected to the source side. The other structure is the same as that of the first embodiment. In addition,
The diode 26 can be provided on the drain side or the source side of the MOS FET 11 also in the circuits of the second and third embodiments.

In the circuit operation of each of the above-described embodiments, the current which is about to flow from the driving power supply 3 through the primary coil 2 while the next MOSFET 11 is turned on from the end of the damper period is clamped by the clamp circuit 14. Since there is no escape area for the current flow due to the parasitic capacitance or the like built in the MOS FET 11, noise of the vibration component is generated at the point A of the circuit in the section of t ₃ to t ₄ as shown in FIG. . This noise hardly causes damage when driving the cathode ray tube when driving at a low frequency, but may cause damage when driving at a high frequency. In the fourth and fifth embodiments, MOS FE
By connecting the diode 26 in series to the drain side or the source side of T11, the parasitic capacitance of the MOS FET 11 can be apparently reduced, and thus the oscillation component between t _{3 and} t ₄ can be significantly reduced. .

As a circuit configuration for removing the noise of the vibration component between t _{3 and} t ₄ , as shown in FIG. 10 (a), for example, as shown in FIG. 25, or (b) in Figure 10
As shown in, the snubber circuit 27 may be connected in parallel with the diode 12 to remove the noise of the vibration component between t _{3 and} t ₄ .

[0044] and in time, in the above embodiment, the diode 12
And the clamp circuit 14 of the MOS FET 11 are provided, and the diode 26 is provided in series with the MOS FET 11, but by providing these diodes 12, 13, 26, particularly in high frequency driving, the switch-on period of the MOS FET 11 is increased. That is, by variably controlling the on-pulse width of the MOS FET 11, the secondary side output H _V of the flyback transformer 1
The characteristic effect of being able to control linearly is obtained.

FIG. 11 is a graph confirming this effect by experiments. As can be seen from this graph, in the circuit in which the diode is only the damper diode 5 as in the basic circuit shown in FIG. 13 , the secondary side output H _V rises sharply in an extremely narrow section of the switch-on period. It becomes difficult to finely control the switch-on period in such a narrow section, and on the left and right sides of this steep rising position, even if the switch-on period of the MOS FET 11 is changed, the secondary side output is almost constant. As shown, the secondary output cannot be linearly variably controlled. Therefore, in the case of the basic circuit shown in FIG. 13, MOS of this example
And control the on period of the transistor 4 is a main switch corresponding to the FET 11, can not be performed variably control the secondary output, therefore, in the basic circuit as shown in prior art FIG. 13, It was unexpected to control the on-pulse width of the transistor 4 to control the secondary output H _V.

On the other hand, according to the circuit of the first embodiment in which the diodes 12 and 13 of the clamp circuit are provided as in each of the above embodiments, the secondary side output H _V changes over a considerably wide switch-on period. and which, in the et, in the circuit of the fourth and fifth embodiments in which a diode 26 and a clamp circuit of a diode 12 and 13 together, most secondary output H over a wide range of switch-on time _V can be changed linearly, and as is clear from these experimental results, the diodes 12 and 13 of the clamp circuit or the MOS
By providing the diode 26 connected in series FET 11, is revolutionary effect is obtained that can be controlled only at the secondary side output H _V controls the ON period of the one switch element MOSFET11 linear .

This epoch-making effect is brought about by the following reasons. In the basic circuit including only the damper diode 5 shown in FIG. 13 , when the transistor 4 is turned on, a current flows from the driving power supply 3 to the coil 2 and electromagnetic energy is stored in the coil 2, and when the transistor 4 is turned off, electromagnetic energy is generated. Although free oscillation starts due to LC series resonance flowing to the resonance capacitor 6, the counter electromotive force of the coil 2 causes the back electromotive force of the coil 2 to be the power source voltage E _B
In the above case, the damper current flows, and the pulse voltage shown in FIG. 14A does not drop below the ground level. On the other hand, the peak value of the flyback pulse V _C is proportional to the coil current I _O of the coil 2 when the transistor 4 is turned off.

[0048] I ₀ when the coil current I ₀ is as described above, the initial current I ₁ at the time when the transistor 4 is turned on
= (E _B / L) t _on + I ₁ . L is the inductance of the coil 2 and t _on is the on time of the transistor 4.
Considering only this, the secondary side output H _V can be linearly controlled by controlling the on period of the transistor 4, but in reality, even if the transistor 4 is turned on during the damper period, no transistor current flows. Until the damper period ends, the secondary side output remains constant and uncontrollable. As shown in FIG. 11 , in the case of the circuit including only the damper diode 5, the level of the free vibration is large and the damper period exists at a plurality of places. As a result, the section in which the secondary output is variable is a narrow range. It becomes difficult to linearly control the secondary side output depending on the ON period.

On the other hand, in the circuit of this embodiment in which the clamp circuit 14 is provided by the diodes 12 and 13, the resonance capacitor 6
The voltage across both ends is clamped at the power supply voltage E _B by the clamp circuit 14, so that free oscillation is prevented and the unnecessary pulse P _W
Will never occur. Therefore, MOS FE
By controlling the switch-on period of T11, the secondary side output can be freely changed.

[0050] However, in practice, the parasitic capacitance of the distributed capacitance and MOS FET 11 of the flyback transformer 1 is the same function as the resonant capacitor 6, the free oscillation caused by the series resonance between the inductance of the coils 2, drive When it is reverse-biased to the power source voltage E _B or more of the power source 3 and another damper period is generated during the scanning period as shown in FIG. 12, when the second damper period and the ON period of the MOS FET 11 overlap, this The section is a section where the secondary side output H _V cannot be controlled.

Further, at the position of the peak of the pulse during the scanning period, M
When the OS FET 11 is turned on, the current I ₀ flowing through the coil 2 changes, and the damper current equivalent thereto changes, so that the off point of the damper changes, causing an oscillation state. Therefore, it is necessary to reduce the parasitic capacitance and the distributed capacitance of the flyback transformer 1 so as not to generate an unnecessary oscillating current. In this respect, the distributed capacitance of the flyback transformer 1 is determined by the structure of the same transformer 1. Therefore, in this embodiment, by inserting the diode 26 in series with the MOSFET 11, the MOS FET 11
Reduce the apparent parasitic capacitance, coupled with the clamping effect of the clamp circuit 14 by diodes 12 and 13, as shown in FIG. 11, the control of the switch-on period of the MOS FET 11, substantially linear secondary side output H _V It became possible to control to. Incidentally, the effect of preventing the decrease in free vibration pulse apparent parasitic capacitance of the MOS FET 11 is that Do remarkable its effect higher frequencies.

The present invention is not limited to the above-mentioned embodiments, but various embodiments can be adopted. For example, in the circuits of the above-described respective embodiments, a smoothing capacitor 9 is provided on the high voltage end side of the secondary coil 7 as shown by the chain line in FIG. 1, or a speed-up capacitor for improving the response of high voltage stabilization.
It may be provided with 28. Further, since the circuits of the respective embodiments have a wide control range for stabilizing the high voltage and have good responsiveness, generally, a choke coil for improving regulation is connected in parallel to the primary coil of the flyback transformer 1. Of course, this choke coil may be connected in parallel to the primary coil 2, although it is not necessary to do so.

Further, in each of the above embodiments, the series circuit of the voltage dividing resistors 15 and 16 is connected to one end of the secondary coil 7 in order to take out the high voltage output voltage. Since a resistance circuit for extracting the focus voltage and the screen voltage is connected to the secondary coil side, the high voltage output voltage may be detected using this resistance circuit .

[0054] in is found, the above embodiment is to configure the switch element by MOS FET 11, this switching element may be constructed using bipolar transistors or the like, the other switching elements. Further, in the third embodiment, the switch element connected in parallel to the diode 12 is constituted by the transistor 24, but it can be constituted by using another switch element such as MOS FET.

Further, in each of the above embodiments, the clamp circuit 14
Is composed of two diodes 12 and 13. The clamp circuit 14 can clamp the voltage across the resonance capacitor 6 to the power supply voltage of the driving power supply 3 from the end of the damper period to the next switch-on period. Any circuit element other than the diodes 12 and 13 may be used.

Further, although the above embodiments have been described with reference to the resonance type power supply circuit for generating high voltage, the circuit of the present invention is also applicable as a resonance type power supply circuit for low voltage.

[0057]

According to the present invention, since the crest value of the voltage pulse generated on the primary side can be controlled by controlling the ON period of only one switch element, that is, the secondary side voltage can be stabilized, the circuit of the present invention can be implemented. When used as a power supply circuit for generating high voltage, the number of parts of the high voltage stabilizing circuit can be reduced and the circuit configuration can be greatly simplified. Moreover, it is possible to prevent unnecessary voltage pulses that are to be generated between the end of the damper period and the next turning on of the switch element by simply providing a clamp circuit with a simple circuit configuration. An independent switch element and a complicated control circuit for driving and controlling this switch element are not required, which reduces the number of parts and makes the circuit configuration extremely simple.
It is possible to significantly reduce the circuit cost.

Further, since the circuit of the present invention is a resonance type in which the resonance capacitor is charged and discharged in each deflection cycle, the high-voltage stabilization response is excellent.

Further, since the circuit of the present invention has a structure in which the generation of the unnecessary voltage pulse is prevented by using the clamp circuit, there is no restriction that the ON period of the switch element must overlap the damper period. ,
The ON pulse width of the switch element can be widened from zero to the end of the damper period, and the voltage control width can be remarkably widened as compared with the conventional one, and it is also optimal as a high voltage power supply for multi-scan.

Further, the current flowing through the primary coil is turned on.
In the configuration in which a diode is connected in series to the switch element that is turned off, the parasitic capacitance of that switch element can be apparently reduced, which allows the switch element to turn on after the end of the damper period. It is possible to suppress the vibration component noise that is about to occur, and the epoch-making effect that the secondary side output can be linearly variably controlled by variably controlling the switch-on period accordingly. .

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a first embodiment of a resonant power supply circuit according to the present invention.

FIG. 2 is a time chart showing the circuit operation of the embodiment.

FIG. 3 is a circuit diagram showing a second embodiment of the present invention.

FIG. 4 is an explanatory diagram of a circuit provided with a resonance capacitance switching unit as a modified example of the embodiment.

FIG. 5 is a circuit diagram showing a third embodiment of the present invention.

FIG. 6 is a comparative explanatory diagram of a waveform of a flyback pulse by the circuit of the same embodiment and a waveform of a flyback pulse of each of the first and second embodiments.

FIG. 7 is a circuit diagram showing a fourth embodiment of the present invention.

FIG. 8 is a circuit diagram showing a fifth embodiment of the present invention.

FIG. 9 is an explanatory diagram of vibration component noise generated from the end of the damper period to the next switch-on period.

FIG. 10 is an explanatory diagram of various circuit examples for removing vibration component noise.

FIG. 11 is an effect of a diode provided in this embodiment .
It is a graph of a fruit characteristic.

FIG. 12 shows the state of occurrence of a re-damper period due to free vibration
FIG.

FIG. 13 shows a resonance-type power supply circuit that has inspired the present invention .
It is a basic circuit diagram.

14 is a time chart showing the operation of the circuit of FIG .
It

[Explanation of symbols]

 1 Flyback transformer 2 Primary coil 3 Drive power supply 4 Transistor 5 Damper diode 6, 6a, 6b Resonant capacitor 11 MOS FET 12, 13 Diode 14 Clamp circuit

Claims (2)

(57) [Claims] 1. A drive power source is connected to a primary coil of a transformer, and a switch element for controlling on / off of a current flowing through the primary coil on the primary coil side, and a series connection of the primary coil when the switch element is off. A resonance capacitor that generates a voltage pulse by resonance and a damper diode that causes a current that flows in the direction opposite to the direction of the current flowing through the switch element to flow through the primary coil within the damper period are connected , and the current that flows through the primary coil is turned on / off. A resonance type power supply in which a switching element to be controlled is composed of one switching element, and the crest value of a generated voltage pulse is controlled by controlling the ON period of the one switching element.
Circuit, which is a switch element from the end of the damper period
The voltage across the resonant capacitor is
A clamp circuit that clamps to the voltage of the drive power supply is provided.
Resonant power supply circuit, characterized by that. 2. The on / off control of the current flowing through the primary coil
The switch element to be controlled is an apparent offset of the switch element.
2. The resonance type power supply circuit according to claim 1, wherein diodes for reducing raw capacity are connected in series .