JP2596643B2 - Coded modulation apparatus for mobile communication and method thereof - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

FIELD OF THE INVENTION This invention relates to coded modulation techniques, and more particularly to the use of such techniques in channel applications, such as fading in cellular mobile communications.

[0002]

BACKGROUND OF THE INVENTION During the past decade, trellis coded modulation has been demonstrated to be a practical power and bandwidth efficient modulation technique for channels with additive white Gaussian noise (AWGN) . . This technology is now widely used in commercial telephone line modems, increasing the line rate of these modems to 19.2 Kbit / s.

More recently, the applicability of the trellis coded modulation technique to a further class of channels, especially fast fading channels, ie, detection of signal amplitudes that change drastically in a short period of time and thereby accurate detection The applicability to channels where it is impractical to recover the transmitted information is being investigated. In fact, the signal amplitude is so weak that accurate data recovery is not possible even if the signal amplitude is detected.

[0004] Various types of mobile communication channels fall into this category. The use of trellis coding on such channels, as in telephone line modem applications, results in a so-called "coding gain" of signal power (compared to so-called "uncoded" modulation methods). The ultimate result is enhanced ability to accurately recover information without requiring additional signal bandwidth.

Unfortunately, the improvement in error rate performance achieved with a given amount of coding gain has been found to be much lower in fast fading channels than in telephone line channels. For example, a 3 dB coding gain improves the error rate by three orders of magnitude over the telephone line channel, but only by one third over the fast fading channel. This imbalance arises primarily from the very nature of the fast fading channel, the fading nature.

[0006]

The prior art uses a combination of a specific trellis code (a) exhibiting so-called "built-in time diversity" and an interleaving / deinterleaving technique (b) for rearranging transmitted signal points. It has been recognized that the use can take into account the occurrence of fast fading in the channel, and thereby provide increased coding gain. At the same time, however, such prior art solutions are generally not entirely satisfactory for certain applications. Digital cellular mobile communications (hereinafter simply referred to as "mobile communications") is a famous example.

[0007] In particular, to achieve the potential coding gain of such a method, an interleaver / deinterleaver having the property that a sufficient amount of transmission delay is introduced into both the transmitter and the receiver. The use of a lever is required. For example, the real-time nature of mobile communication systems means that such delays have a significant negative impact on system characteristics.

[0008] Furthermore, the realization of a possible coding gain is:
Systems using time division multiple access (TDMA), now the North American standard for next generation mobile communications, require even greater delays. (This effect is
In a TDMA system, the signal points originating from one particular source result from the fact that they are closer together than the other. In addition, due to the fact that the trellis decoder must wait until receiving a large number of subsequent signal points to output any particular signal point, some additional delay is introduced and / or Waste some of the channel capacity in that particular application. Such applications include systems that include a speech coder that encodes block by block.

[0009]

SUMMARY OF THE INVENTION In accordance with the present invention, the use of interleaved block coded modulation with built-in time diversity has many advantages and is comparable or superior to the prior art described above. Achieve coding gain. The effect is that the execution is less complex,
Interleaver / Deinterleaver and encoder with less delay, higher bandwidth efficiency for some codes available, achieving system design with desired trade-offs in complexity, power efficiency and bandwidth efficiency Increased flexibility and is less of a problem as a general system.

In a preferred embodiment, a constant amplitude signal constellation is used to compensate for the fast changes in signal amplitude that are characteristic of mobile communications and other fast fading channels. Further, the use of non-coherent differential detection is preferred because of the fast changes in carrier phase that occur in such channels. Both of these criteria are conveniently met by using M-point differential phase shift keying, or M-DPSK.

In accordance with the present invention, the particular manner in which signal points are rearranged by the interleaver is matched to the particular block code used, thereby increasing the effective but impractical size of the interleaver. This has the effect of reducing the interleaver / deinterleaver delay previously described.

[0012]

DESCRIPTION OF THE PREFERRED EMBODIMENTS In the transmission system of FIG.
The above input data is applied to the 2N dimensional block encoder / mapper at a rate of m bits per T second signal interval, where m is an integer or fraction. The block coder / mapper 13 integrates a block of input data consisting of bits corresponding to N signal intervals, and converts the selected 2N-dimensional block code into the integrated N × m bits into (m + r) codes. Used to encode into N groups of coded bits. These groups are subsequently provided on leads 16. Here, the parameter r is the average number of redundant bits per signal interval introduced by the block encoder / mapper 13.
Each allowable bit pattern of the (m + r) bit group is 2
Dimension (2D) M (≦ 2 ^{m + r} ) —Relates to a particular signal point in the PSK array. Where M is a selected integer. Especially,
The signal point on the lead 16 during the nth signal interval is denoted as _Pn . The block code is "2N-dimensional" because each signal point is two-dimensional and each "codeword" output by block encoder / mapper 13 is represented by N signal points.
Called.

To ensure an understanding of the terms and concepts used herein, reference is made to FIG. The 2N dimensional block encoder / mapper generates 2N dimensional "codewords". Each signal word consists of a sequence of N "signal points". Each signal point is
One of a predetermined two-dimensional "array", shown diagrammatically in FIG. 2 as a phase shift keying array having eight signal points, ie, 8-PSK.

In the 2N-dimensional code word, one signal point is sent for each signal interval during N "signal intervals". The set of all 2N-dimensional codewords is called a "2N-dimensional array" where each codeword is an "element" of a 2N-dimensional array. 2N-dimensional arrays are also called codebooks or writing systems.
In the following description, each of the 2N-dimensional codewords is often treated as a concatenation of two N-dimensional components in a constituent N-dimensional array. Here, the configured N-dimensional array is 2N
Arrives like a dimensional array. This view is N / 2, N /
Iterated over elements and arrays of dimension 4, etc.

As shown in FIG. 1, encoder / mapper 1 on lead 16 corresponds to each group of N × m input bits.
N consecutive two-dimensional signal points output by 3 are applied to the interleaver 21. The function of the interleaver is to rearrange the signal points _Pn such that the signal points belonging to a particular codeword are separated from one another in time during transmission. This method reduces the likelihood that channel fading will affect one or more of the N signal points of the codeword.

In the preferred embodiment, the block codes used in block encoder / mapper 13 have built-in time diversity, as described in detail below,
And, and also as described below, this time separation of signal points greatly enhances the ability to accurately recover the transmitted data of such a code. Further in accordance with the present invention, block codes and interleaver reordering algorithms are jointly selected to further enhance this capability.

Finally, the reordered signal points Q _n output by the interleaver on lead 24 are applied to a modulator 25 whose output is in turn applied to a fast fading channel 30. Modulator 25 is described in more detail below.

In the receiver, the demodulator 41 and the deinterleaver 44 reverse the functions of the modulator 25 and the interleaver 21, respectively. Accordingly, the output of the deinterleaver on lead 45 corresponds to the received but channel impaired signal point P corresponding to the sequence of signal points appearing on lead 16 at the output of encoder / mapper 13. _n
Is a continuation of These are block decoders 51 which restore the original transmitted input data on lead 53 and supply it.
Is added to

In accordance with the present invention, as will be described in more detail below, the block decoder 51 performs a so-called "soft decision", similar to the maximum likelihood decoder conventionally used for trellis coded signals in an AWGN environment. Work based on.

At this point, it is instructive to explain the concept of time diversity coding using the description of the first block code modulation scheme, referred to as code I, implemented in the embodiment shown in FIG. Would.

In particular, an 8-PSK signal constellation as shown in FIG. 3 is used to implement a four-dimensional (4D) code. This means that each codeword generated by the code is 8-P
It means that it consists of two two-dimensional points of the SK array. These points are transmitted during each signal interval. The eight points in the array are labeled with numbers from 0 to 7. In this case, the parameters m and r are both equal to 1.5, and of course N = 2. Thus, block encoder / mapper 13 generates a 3-bit word at each of two consecutive signal intervals. Each such word identifies a particular one of signal points 0-7 by its bit value.

Lead 11 in a codeword on lead 16
The input to the upper block encoder / mapper 13 (N ×
The encoding / mapping of m) is shown in the table of FIG. Here, each of the 3-bit patterns in parentheses is (m + r
=) Indicates the value of the three coded bits and is simply the binary value of the decimal label of the associated signal point. 2 ³
= 8 bit patterns and therefore 8 code words. Notationally, each codeword is referred to below as x and y
Are represented in the form of (x, y) as a first signal point and a second signal point constituting a codeword, respectively.

It is important that each of the codewords of code I is different from any other codeword in both of its constituent two-dimensional points. Thus, for example, the first or second signal point of codeword (0,0) is not equal to the first or second signal point of any other codeword. The importance of this property can be seen in the case where the amplitude of one of the two constituent signal points is significantly attenuated by channel fading, resulting in a complete loss of the information carried. As long as the other constituent signal points of the codeword are correctly recovered, it is nevertheless possible to recover that information.

In particular, if the first signal point of the restored codeword is "3" and the second signal point is lost, none of the other codewords has "3" as the first signal point. So
The transmitted codeword is nevertheless determined to be (3,7). (This analysis is oversimplified on how well the decoding process is preferably performed, but is useful for explanation.)

Thus, this code provides a built-in high immunity to transmission errors introduced by fading via a mechanism of time diversity. That is, information about each input data bit appears redundantly in the time domain in the encoded signal. Thus, for example,
Information about each of the three bits of the input bit pattern 010 appears in both the first signal point "3" and the second signal point "7" of the corresponding codeword (3,7).

Generally speaking, if X is an integer greater than 1, then each codeword consisting of a sequence of constellation points is:
A code is said to have X times time diversity if it differs by at least X signal point locations for each of the other codewords. Therefore, the code I is evaluated to have twice the built-in time diversity. There are in fact many ways to combine the signal points of FIG. 3 into a codebook of four-dimensional codewords exhibiting twice the built-in time diversity. Advantageously, however, the one shown in FIG. 4 has the further advantage of maximizing the least square Euclidean distance between codewords (given the requirement for double time diversity). It further enhances the error immunity of the overall coding method. The distance of this code is “4”.

It is important to emphasize here that what is being described is a coded modulation method. Thereby, a) the signal constellation is increased to more than 2 ^m signal points to reduce the signal bandwidth requirements caused by the introduction of redundant bits, and b) the block coding and constellation mapping are interdependent. There is a meaning. This is a)
The introduction of redundant bits is given by the extension of the signal bandwidth,
b) In contrast to conventional block coding methods, block coding and constellation mapping have nothing to do with each other.

More specifically, a) two-dimensional M
The code of the present invention using the signal points from the PSK array
b) Block coding the input bits using, for example, a Reed-Solomon code or other block code, each signal point being a resulting code as a sequence of signal points respectively taken from the same two-dimensional M-PSK array. It is appropriate to compare with the method of transmitting coded bits.

In such a case, the resulting set of transmitted signal points has some of the characteristics of the code of the present invention, such as X time diversity and some coding gain. However, such a method, unlike in the case of the present invention, is irrelevant to the minimum Euclidean distance between successive signal points. As a result, the least square Euclidean distance between successive signal points is as small as the distance defined therein as the "absolute least square Euclidean distance" associated with the script in question. The distance is a two-dimensional M-PSK multiplied by the diversity parameter "X".
It is given as the least square Euclidean distance of every pair of constellation points in the array.

Thus, for example, the absolute least squares Euclidean distance of a four-dimensional code having twice the time diversity and using the two-dimensional M-PSK array of FIG. 3 is represented by the code words (0, 0), ( (1,1), (2,2), ...,
1.17, which is the least square Euclidean distance between the code words of the four-dimensional code composed of (7, 7).

The code of the present invention, by contrast, has a least square Euclidean distance between successive signal points or codewords that is greater than the absolute least square Euclidean distance described above. For example, for code I, which also has twice the diversity and also uses the arrangement of FIG. 3, the least square Euclidean distance between codewords is four.

As a further difference, conventional block coding schemes generally use the same arrangement as used for non-coding schemes. Thus, for example, to transmit m (m is an integer) information bits per signal interval in an uncoded system, the constellation used has 2 ^m modulated signal points.
Even though r redundant bits per signal interval are introduced by conventional block codes, the same arrangement is still used and the baud rate is multiplied by (m + r) / m to accommodate these redundant bits. In contrast, a coded modulation scheme such as the present invention accommodates at least some of the redundant bits by expanding the size of the constellation to have 2 ^m or more modulated signal points.

In addition, there are many ways to assign an input bit pattern to the various codewords of FIG. Advantageously, however, the particular assignment method shown in FIG. 3 has the additional advantage of reducing the bit error rate that occurs when a transmitted codeword is incorrectly decoded. In particular, a Gray code type method is used to assign input bit patterns to codewords.

Assume that the second signal point of the codeword is lost and that the information bits are restored based solely on the first received signal point. In FIG. 4, codewords whose first signal points are closest to each other in Euclidean distance are 1
Notice how only one bit position is assigned to a different bit pattern. The underlying concept is that signal points that are closest to each other are the signal points that are most likely to be confused with each other. As this is the case, employing the aforementioned Gray coding type method for bit pattern assignment ensures that the minimum number of bit errors is associated with such most likely decoding errors.

Thus, since signal points 0 and 1 have the minimum distance between them, bit patterns "000" and "001" are each assigned to a signal point that differs by one bit position. (It is not possible for this particular code to simultaneously provide such increased error correction capability for the assumed fading of the first signal point. However, some general advantages are nonetheless only in one case. Is achieved by dealing with.)

Finally, this concept of Gray coding is useful in any of the other codes described herein, at any time and without any special mention of it. Be careful.

If a single signal point is lost due to fading, it is recognized that signal points that are close in time are also likely to be lost. Therefore, according to principles known to those skilled in the art, the error immunity provided by the built-in time diversity of the code is enhanced by temporally separating the two signal points of each codeword, so that the codeword 2
It is less likely that the points will fade simultaneously. The signal point P _n on the lead 16 of FIG.
Is added for this purpose.

Specifically, the interleaver 21
Take and store frames of J codewords. The locations where the codewords are stored in each row of the storage matrix held in the interleaver are schematically depicted in FIG. At the point depicted in the figure, the signal points P ₁ and P ₂ of the first codeword are stored in the first row, and the signal points P ₃ and P ₄ are stored in the second row.

In the most directly executable form, the interleaver waits until all J codewords have been read. It then lists the signal points that make up the codeword column by column, ie, initially odd numbered signal points P ₁ , P ₃ ,
…, P _2J-1 and then the even-numbered signal points P ₂ ,
P _4, ..., read the P _2J. (In a more efficient implementation, as long as enough codewords have been read in to ensure a synchronized flow of signal points on lead 24, odd numbers may be read before all J codewords are read. It is possible for the interleaver to start reading the signal points of the numbers.)

Note that the signal points are grouped together at the location of the corresponding signal point of the codeword in the frame. That is, assuming that i = 1, 2,..., N, the i-th signal point of the codeword of the frame is arranged in each group. Thus, as desired, the two signal points of each codeword appear in the reordered sequence on the interleaver output lead 24, but are now advantageously far apart in time, specifically They are separated by the J signal interval. This process obviously repeats for successive frames of J codewords.

Ideally, the effectiveness of the interleaver is maximized when the parameter J is greater than or equal to 1/4 the minimum speed of the medium of interest multiplied by the signal rate divided by the carrier wavelength. . (This formula, as in the case where frequency division multiple access (FDMA) is used,
Based on the assumption that there is only one user per mobile communication channel. Considerations for the case of the so-called TDMA scheme, where several users are time-multiplexed on one channel, are addressed at the following more convenient points. )

However, in certain applications, a value of J smaller than this optimum must be used to reduce the transmission delay introduced by the interleaver / deinterleaver. (This is necessary to guarantee the desired level of data throughput, ie, to avoid unnatural conversation delays that would otherwise be introduced into the conversation.)

Finally, the lead 2 is interleaved.
The rearranged signal points Q _n output on 4 are applied to a π / M shift M-DPSK modulator 25. The phase of the carrier is shifted from the phase during the previous signal interval by 2πQ _n / M, increased by a constant π / M radian. In accordance with the present invention, the use of such a π / M shift modulator helps reduce the ratio of peak power to average power, and ameliorates the potential timing recovery problem of the receiver.

Furthermore, according to the present invention, the fact that the transmitted signal points are interleaved advantageously removes the correlation between noise samples introduced by the M-DSPK demodulation process at the decoder.

The code I has a bandwidth efficiency of 1.5 bits / signal interval, and assuming a bit error rate of 10 ^-3 , for a mobile channel with a medium speed of 20 (60) miles per hour, It has a coding gain of 8.9 (14.3) dB, which is a so-called uncoded 4-DPSK method with the same information bit rate. These coding gains are determined by consideration of human factors, especially the delay of conversation, at 37 ms.
, Which is given by the size of the interleaving frame, which in this case is 2J, divided by the signal rate. These coding gains assumed a carrier frequency of about 900 MHz. From now on, all coding gains in other codes will be noted under these conditions.

Here are some alternative block coded modulation schemes suitable for use in a fading channel.
These codes achieve varying degrees of greater bandwidth efficiency and / or coding gain compared to the first code by accepting some higher decoder complexity. (For this purpose, the complexity of the decoder is understood to be the number of additions and / or comparisons per information bit that need to be performed when performing the decoding process described below.)

FIGS. 6 and 7 schematically show a second code called code II. This is an eight-dimensional (8
D) code, which is first of the type 2 shown in FIG.
This is achieved by defining a constitutive 4-dimensional 8-PSK array formed by selecting specific elements from a concatenated pair of dimensional 8-PSK arrays. The desired eight-dimensional array is then similarly formed by selecting particular elements from the connected pairs of the four-dimensional array. Each element of the eight-dimensional code is used as a code word of the eight-dimensional code.

In particular, when included in the four-dimensional array, 2
Select half of ³ × 2 ³ = 64 possible 4D elements.
They are elements of the form (a, b). Here, “a” and “b” are the two-dimensional 8-PSK shown in FIG.
Both of the arrays have even two-dimensional signal points or both have odd two-dimensional signal points.
One of the dimensional signal points. Thus, for example, (0,
(0) and (5,7) are signal points of a four-dimensional array, but (2,
5) and (7,0) are not. As shown in FIG.
The 32 elements of the four-dimensional array are divided into four subsets S ₀ ,
..., it is divided into S _3.

Finally, as to be included in the eight-dimensional array,
Select a quarter of the 2 ⁵ × 2 ⁵ = 1024 possible 8-dimensional elements. These 256 elements are the elements of the four 4-dimensional subset pairs shown in the codebook of FIG. Specifically, a specific pair of four-dimensional elements is connected only when the pair of four-dimensional subsets to which the first and second four-dimensional elements belong are each one of the four patterns shown in FIG. Is an element of an 8-dimensional array.

Thus, for example, (0, 0, 1, 5) becomes
a) (0,0) and (1,5) are both elements of the four-dimensional subset S ₀ , and b) the pattern of (S ₀ , S ₀ ) is the four allowed four-dimensional parts Since it is one of the patterns of a set pair, it is an element of an eight-dimensional array. On the other hand, (0,
2,0,6) is a) (0,2) and (0,6) are elements of subsets S ₀ and S ₃ , and b) The pattern of (S ₀ , S ₃ ) has 4 Since it is not one of the patterns of the four-dimensional subset pair, it is not an element of the eight-dimensional array.

The above method of generating a particular code is generally defined as follows. (A) Select a particular concatenation of signal points in the first array to be the first set of elements. (B) Group the set of elements defined immediately before into a subset. (C) selecting at least some of the selected connected elements of the subset of step (b);
Each such element is a succession of elements each taken from a respective subset of the concatenation. (D) Repeat steps (b), (c) and (d) until 2N-dimensional elements are formed. Thereby, those elements become 2N-dimensional codewords.

The eight-dimensional array thus formed has 2 ⁸ =
Having 256 code words, this code can communicate 8 information bits for each 8 dimensional code word. Generally, a method of assigning a bit pattern to a specific codeword is used. However, to reduce implementation complexity (as well as taking advantage of the possibility of using Gray codes as described above), the encoding / mapping process is split into two stages. As shown in FIG. 7, the two bits are
Used to select a particular four-dimensional subset pair, while three bits are one of the eight elements of each of the two selected four-dimensional subsets, as shown in FIG. Used to select. At this stage, a total of 6 bits are used, and of course, 8 bits are used when all are combined.

As with the first code, the second code exhibits twice the built-in time diversity as seen below. First of all, each of the four-dimensional subsets shown in FIG. 6 has itself twice the built-in time diversity. (In fact, it turns out that the four-dimensional subset S ₀ itself is identical to the code I described above. Further, each of the other three subsets can be used instead.)
Any pair of four-dimensional subsets is guaranteed to exhibit twice the built-in time diversity.

In particular, if two eight-dimensional elements belong to the same pair of four-dimensional subsets, that is to say from the description on the same line in FIG. It can easily be ascertained that two and in some cases all four are different. In addition, if two eight-dimensional elements belong to different four-dimensional subset pairs, ie, from different row descriptions in FIG. 7, then those two eight-dimensional elements are:
At least one two-dimensional signal point location in the first configuration four-dimensional element and at least one in the second configuration four-dimensional element.
It is also easily ascertained that the two two-dimensional signal point positions are different. This results from the fact that each four-dimensional subset appears only once in each of the two columns of the codebook of FIG. Thus, as described above, every two 8
It is guaranteed that the two-dimensional signal points of the two-dimensional codewords are different from each other.

This code interleaving is performed using the interleaving method described above, where the codewords are read into the interleaver in rows, one codeword per row, and then the first of all codewords. This is performed using the interleaving method described above, conceptualized as one two-dimensional element, then the second two-dimensional element of the entire codeword, and so on. Instead, FIG.
And the interleaving scheme described below in connection with reference III in FIG. 9 may be used.

In the description of the type of code just described, there are many ways in which double built-in time diversity can be provided. For example, a) select a four-dimensional array, b) select a specific number of subsets into which the selected four-dimensional array will be divided, c) subdivide the selected four-dimensional array into parts of the aforementioned number. Split into sets,
Or d) a specific 8-dimensional array for the codebook,
That is, there are many possible ways to select a particular four-dimensional subset pair.

For example, a 32-element four-dimensional array is composed of 32 elements not selected in the previous example. For example, (S
_{0, S 0), (S} 1, S 1), (S 2, S 3), and (S _3,
Four 4 that meet the criteria discussed above, such as S ₂ )
Different pairs of dimensional subsets can be used as codebooks. On the other hand, another method of providing double built-in time diversity shows significant differences in coding gain, bandwidth efficiency, and complexity (as described below).

The symbol II in FIGS. 6 and 7 is 2.0 bits /
It has bandwidth efficiency of signal spacing and 8.4 (14.5) dB coding gain at a media speed of 20 (60) miles per hour. Thus, it can be seen that this code provides greater bandwidth efficiency while achieving essentially the same coding gain as code I in FIG. However, this "improvement"
This was achieved at the expense of some complexity compared to Code I and some inferior performance at speeds below 20 mph (eg, Code 10 is 7.2 dB for Code I versus 5.4 dB for Code II).

FIGS. 8 and 9 give the same bandwidth efficiency (1.5 bits / signal spacing) as Code I, but again at the expense of some added complexity to achieve a sufficiently large coding gain. Another 8 with 2x built-in time diversity
A symbol III, which is a dimensional code, is drawn.

As shown in FIG. 8, reference numeral III denotes four subsets S of the 32-element 4-dimensional 8-PSK array shown in FIG.
Each of ₀ , S ₁ , S _2, and S ₃ is divided into four finer subsets, each consisting of two elements. The resulting finer subset is T ₀
Or it is displayed T _15. As shown in the codebook of FIG. 9, the concatenation of a particular pair of four-dimensional elements is the first and second
The four-dimensional subset pairs to which the four-dimensional elements of
Is an element of an eight-dimensional array only if it is one of the 16 patterns shown in FIG.

There are a total of 64 code words in an eight-dimensional array.
As further shown in FIG. 9, four bits are used to select a particular four-dimensional subset pair, while, as shown in FIG.
Two bits are used to select one of the two elements of each four-dimensional subset, for a total of six bits.

Conveniently, the code is 20
It has a code gain of 10.1 (17.1) dB for the mobile channel at a media speed of (60) miles. Thus, it can be seen that this code provides a greater coding gain than either of the two codes described above.

To fully understand how these larger coding gains occur, it is instructive to define parameters that are: a) easy to calculate, and b) provide an indication of channel fading performance. . In this, this parameter called "least squared Euclidean distance with X time diversity", that is, MDX,
The least squares Euclidean distance between any two codewords that is useful for the design of a code with X times time diversity for a fading channel and that differs by exactly X signal point locations from each other.

Generally, the greater the distance, the greater the coding gain. For example, media speed, interleaving length, number of code dimensions, and other factors, including the total minimum Euclidean distance between any two codewords, can also affect the coding gain, so that two with the same MDX A single code exhibits somewhat different coding gains. However, in general, there is a high correlation between MDX and coding gain, as long as the interleaving length is sufficient to ensure that the signal points of a particular codeword fade independently.

Given an assumed interleaving length of 37 ms and a media speed of 20 miles per hour, such independent fading is actually guaranteed, for example, for the four and eight dimensional codes described so far. . Thus, for each of these three codes,
And 8 MDX lead to substantially the same coding gain for the first two codes and a sufficiently large coding gain for the third code. (Throughout this description, all quoted values for MDX have been calculated assuming a 1-radius M-PSK array.)

Referring to the codes of FIGS. 6 and 7, the interleaving of code III is performed using the above-described interleaving scheme. However, according to the present invention, a particular way in which the signal points are rearranged by the interleaver is a)
A) increase the effectiveness of the interleaver, ie the contribution of the interleaving process to the coding gain, for a given interleaving distance where the interleaver is forced by consideration of the system as described above, or b) where It is used to allow for a shorter interleaving length for a given level of coding gain, thereby possibly meeting the needs of an otherwise unmet system, and is matched to a unique block code.

Referring to the codes of FIGS. 8 and 9, all codewords differing only by two signal points have different first and second signal point positions or third and fourth signal point positions. Is observed. As this is so, according to the logic set up previously for the interleaver of FIG. 5, in particular, as shown in FIG. 10, first the first two-dimensional signal points of all codewords (FIG. 10) Column A), then the third two-dimensional signal point of all codewords (column C), then the second two-dimensional signal point (column B), and the fourth two-dimensional signal point (column D) In addition, it is effective to rearrange the order in which the sequence of signal points is read from the interleaver. This method has the effect of separating the first and second signal points of every codeword at 2J signal intervals instead of at J signal intervals. The same applies to the third signal point and the fourth signal point.

In the more general case, to maximize the separation of signal points of codewords that differ by X signal point locations, and thus provide the desired X built-in time diversity,
It is understood that various rearrangement methods may be used within the interleaver, depending on the structure of the codeword. More formally speaking, given a particular codeword of a script that differs by exactly X signal point locations and has a codeword distance MDX, two of those X signal point locations are designated k-th. Indicated as the q-th signal point position, the interleaver calculates the code of each frame such that the signal points at the k-th and q-th signal point positions are separated by J signal points or more in the rearranged sequence. Operate to produce a rearranged sequence of word signal points. Given this general principle, it is unnecessary to further describe the structure of the interleaver that is advantageously used in connection with the various other codes described below.

In FIGS. 11-13, the code is a 16-dimensional code with twice the built-in time diversity, having the same MDX (ie, 4) as codes I and II.
IV is drawn. However, the 16-dimensional code has significantly higher bandwidth efficiency (2 + 3/8 bits / signal spacing) compared to the codes described to date at the expense of lower coding gain (at the same interleaving length). give. The complexity of the decoder is also somewhat higher than for the codes already described.

First, a 4-dimensional 8-PSK composed of 64 elements formed from a connected pair of the 2-dimensional 8-PSK array in FIG.
Define the PSK sequence. The four-dimensional array is then
As shown in, it is classified into eight subsets of S ₀ to S _7. (Note that subsets S ₀ through S ₃ are identical to the four subsets shown in FIG. 6.) Then, the constituent eight-dimensional array can be used to identify a particular element from a connected pair of four-dimensional arrays. It is formed by selection. In particular, 32 were selected from 64 possible subset pairs to construct an 8-dimensional array. 8-dimensional array is composed of 2 ^eleven elements are classified into each eight subsets consisting of four four-dimensional subset pairs, as shown in FIG. 12. 8 dimensional subset of ^{_{eight, S '0, S' 1}} , ..., indicated as S ^_'7.

Finally, to include in the 16-dimensional array, 2 ¹¹
× 2 Select 1/8 of the ¹¹ possible 16-dimensional arrays.
These two ¹⁹ element is an element of eight 8-dimensional subset pairs shown in the codebook of FIG. Specifically, the concatenation of a particular pair of eight-dimensional elements is performed if the pair of eight-dimensional subsets to which the first and second constituent eight-dimensional elements belong is one of the eight patterns shown in FIG. If
Only in that case, it is a 16-dimensional array element.

As further shown in FIGS. 11-13, three bits are used to select a particular eight-dimensional subset pair. Here, two bits are used to select a particular four-dimensional subset pair from each of the two selected eight-dimensional subset pairs, ie, a total of four bits are used at this level, and a further four bits are used. Three bits are used to select a particular pair of two-dimensional points from each of the selected four-dimensional subsets, i.e., a total of 12 bits are used at this level, and eventually a total of 19 bits are used. Since each codeword extends over eight signal intervals, the bandwidth efficiency is 19/8 = 2 + 3/8 bits / signal interval.

This code is 6.8 (13.9) d in a mobile channel at a medium speed of 20 (60) miles per hour.
B has a coding gain of B.

So far, various codes have been described using a quasi-graphical method. Instead, the block encoder /
The codes can each be described using a number of simple Boolean expressions that define the operation of many of the logic elements that make up the mapper, for example, block encoder / mapper 13 of FIG. This latter method is now used to describe the next code, since the graphical method for the next code is complex to represent. It is understood that any prior art reference has also been described using this latter method. vice versa,
The Boolean expressions that proceed below can be used to construct quasi-graphical representations made for other codes.

In particular, this next code is another 16-dimensional 8-PS with double built-in time diversity.
K code, and this code is the code of FIGS.
It has the same MDX (ie 8) as III. This code is
It has neither the highest bandwidth efficiency nor the highest coding gain, but advantageously provides a useful combination of both of these parameters. Specifically, 8.5 (20 mph)
And a coding gain of 16.7 (60 mph) and 20
It has a bit / signal interval bandwidth efficiency. The complexity of the decoder is the largest of all the codes described.

As shown in FIG. 14, the lead wire 11 shown in FIG.
The 16 input bits accumulated over the above 8 signal intervals are:
The circuit is connected to leads 1341-2,1
The three bits on each of 351-2, 1361-2 and 1371-2 cause a particular 8 on lead 16 of FIG. 1 to constitute a 16-dimensional codeword associated with 16 input bits.
The two-dimensional points are finally specified.

In FIG. 14 (similar to FIGS. 15 and 21 described below), the number of bits shown as being carried on a particular set of leads, eg, lead 11, is N Note that this is the number of bits collected over the signal interval, for example, where the number of bits indicated as being carried by a particular set of leads is the average number of bits per signal interval. ( Contrast with FIG. 1.)

As in the case of the codes already described (graphically), the mapping is performed stepwise. In particular, the 16 bits on lead 11 have their output lead 131
In addition to a 16-dimensional to 8-dimensional array mapper converter that supplies 10 bits to each of 1 and 1312. These 10
Each of the bits is associated with a particular element of the constituent eight-dimensional array. Only certain combinations of these eight-dimensional elements appear on leads 1311 and 1312. As will become even more apparent below, this restriction corresponds, for example, to the restriction of the reference IV in FIGS. 11 to 13 that the 16-dimensional array consists only of a particular selected pair of 8-dimensional subsets.

In the next stage, the lead wires 1311 and 1311
The bits on 12 are applied to the 8-D to 4-D array mapper converters 132 and 133, respectively. For example, the converter 132 supplies 6 bits to each of its output leads 1321 and 1322. Each of these six bits pertains to a particular element of the constituent four-dimensional array, and again, corresponding to the manner in which the code IV is constructed, only a particular combination of these four-dimensional elements leads 1321 and Appears on 1322. The converter 133 is the same as the converter 132. Depending on the implementation, only one of these transducers is used.
The two groups of 10 bits, each on leads 1311 and 1312, are subsequently sent to this one converter.

In the last stage, the bits on leads 1321-2 and 1331-2 are applied to four-dimensional array mappers 134-137, respectively. For example, mapper 13
4 provides 3 bits on each of its output leads 1341 and 1342. Each of these three bits relates to a particular point in the 8-PSK constellation of FIG. 3, and all 2 ⁶ = 64 combinations of two two-dimensional points are used.
The mappers 134 to 137 are the same. As in the case of the converters 132 and 133, only one such mapper is used.

16-to-8-dimensional array mapper converter 13
Embodiments of the one, eight-dimensional to four-dimensional array mapper converter 132 and four-dimensional mapper 134 are shown in FIGS. 15 to 17, respectively.

Referring first to FIG. 15, the 16 bits on lead 11 are divided into three groups. The first group includes I0 _{n to} I3 _n added to the 8-dimensional subset pair selector 231.
4 bits. I4 _{n to} I9 _n (I10 _{n to} I
15 _n) second group consisting of 6-bit (third group) is simply the output bits X4 _n through X9 _n (X4 _{n + 4} through X9 _{n + 4)} passes. The output bits of the selector 231 are two 4 bits.
Groups of bits, ie, X0 _{n to} X3 _n and X0 _{n + 4 to} X3 _{n + 4}
It is.

The 10 bits with the subscript “n” specify the first constituent eight-dimensional element of a codeword consisting of four two-dimensional signal points P _{n to} P _{n +3} (see FIG. 1). Finally used for. Similarly, the subscript “n +
10 bits with “4” are four two-dimensional signal points P _{n + 4}
ＰP _{n + 7} are finally used to identify the second constituent 8-dimensional element of the codeword. In particular, bit X0 _n
Or X3 _n is to identify a particular one of the sixteen eight-dimensional subsets, whereas bits X4 _n through X9 _n identifies the final one particular element of that eight-dimensional subset. The same applies to 10 bits with the subscript “n + 4”.

The selector 231 operates according to the following Boolean expression.
Give output bit.

(Equation 1)

The 10 bits of X0 _{n to} X9 _n correspond to the converter 1
32. These bits are also divided into three groups, as shown in FIG. The first group consists of eight bits X0 _{n to} X7 _n applied to the four-dimensional subset pair selector 232. A second group (third group) consisting of one bit X8 _n (X9 _n ) simply passes through and outputs bit Y5 _n (Y
5 _{n + 2} ). The output bits of the selector 232 are 5
Two groups of bits, that is, Y0 _n through Y4 _n and Y0 _{n + 2} through Y4 _{n + 2.}

The six bits with the subscript “n” specify the first constituent four-dimensional element of the first constituent eight-dimensional element of the codeword consisting of two two-dimensional signal points P _n and P _{n + 1.} Finally used to. Similarly, 6 bits with the subscript “n + 2” indicate two two-dimensional signal points P
It is used to identify the second constituent four-dimensional element of the first constituent eight-dimensional element of the codeword, consisting of _{n + 2 and} P _{n + 3} .

[0087] Bit Y0 _n through Y4 _n is to identify a particular one of 32 four-dimensional subsets, whereas one bit Y5
_n specifies a specific element of the four-dimensional subset.
The same applies to the 6 bits with the subscript “n + 2”.

Selector 232 provides its output bits according to the following Boolean equation:

(Equation 2)

The six bits Y0 _{n to} Y5 _n are applied to the mapper 134 as shown in FIG. Mapper 134
Output bits are two groups of three bits each, ie, Z0 _n
To Z2 _n and Z0 _{n + 1 to} Z2 _{n + 1} . The three bits with the subscript "n" define a two-dimensional signal point _Pn . Similarly, the three bits with the subscript “n + 1” define a two-dimensional signal P _{n + 1} . Mapper 134 provides its output bits according to the following Boolean equation:

(Equation 3)

The graphic representation of the sixth code, code VI, is:
This is shown in FIGS. This means that 12 points (M = 1) for 8 points, for example, as compared with the code II in FIGS.
2) Double built-in time to achieve high bandwidth efficiency of 2 + 3/8 bits / signal spacing at the price of a larger M-PSK constellation size, slightly higher complexity and some reduction in coding gain. This is another eight-dimensional code having diversity.

As with the other eight-dimensional codes described above, the code VI
Is reached by first defining a four-dimensional array formed by selecting a particular element from a connected pair of two-dimensional arrays. However, as already noted, this two-dimensional array is a 12-point array, specifically the 12-point array shown in FIG.
It is a PSK sequence. The sign thus shows that the value of M is 2
It is unique in that it is not an integer power. The desired eight-dimensional array is then formed by similarly selecting particular elements from the connected pairs of the four-dimensional array. Each element of the eight-dimensional array is used as a code word of the eight-dimensional code.

In particular, we have included one for inclusion in a four-dimensional array.
Select half of 2 ² = 144 possible 4D elements.
6 and 7, they are:
Illustratively, elements of the form (a, b), where "a" and "b" are both two-dimensional signal points in the two-dimensional 12-PSK array shown in FIG. One of the two-dimensional signal points. As shown in FIG. 19, the 72 elements of the four-dimensional array are divided into six subsets S ₀ ,..., S ₅ each having 12 elements.

Finally, we include from the 72 ² = 5184 possible 8D elements for inclusion in the 8D array.
Select the pieces. The reason for using 768 is described below. These 768 elements are specific among the elements of the six four-dimensional subset pairs shown in FIG. Note the difference between this code and all the other codes previously described. In other eight-dimensional codes, all elements of each of the selected four-dimensional subset pairs are used, ie, codewords. However, in the case of FIG. 20, the method yields 144 codewords from each four-dimensional subset pair, for a total of 864 codewords. However, only 768 code words are desired. Therefore, in this case, FIG. 2 is selected as described below.
Only 128 elements of each 4-dimensional subset pair of 0 are used.

From the foregoing, it will be appreciated that 10 bits are required to specify a particular codeword. Specifically, three bits are used to select a particular four-dimensional subset pair (FIG. 20), and since each such pair has 2 ⁷ = 128 elements, Are required to select a particular codeword.

However, not all possible combinations of these 10 bits are allowed, ie
Note that only 768 codewords are available, but 024 combinations. In fact, even though all 144 elements of each four-dimensional subset pair are included in the code, there is still an insufficient number of codewords, ie, 144 × 6 = 864.
It is. (Using all 864 codewords results in slightly greater bandwidth efficiency, but requires additional processing and introduces additional processing delay.)

The task at this point is then to provide a simple implemented method of generating 768 10-bit word patterns every four signal intervals from the input bit stream. This is achieved according to the technique of processing 19 bits captured over eight signal intervals to generate two eight-dimensional codewords, whereby the previously described 19/8 = 2 + 3
Achieve bandwidth efficiency of / 8 bits / signal interval.

Specifically, FIG. 21 illustrates the input bit being taken into the input lead 201 and the signal point P _n ,
, P _{n + 3} on the output lead 219. Specifically, this circuit comprises a precoder 208, a parallel-serial group converter 209, and a block encoder / mapper 213 corresponding to the block encoder / mapper 13 of FIG.

The input bits on the lead 201 are divided into three groups of 8, 8, and 3 bits, respectively. 8-bit 2
The group simply passes through the precoder. The group of three bits is provided by a bit pattern selector 2 which provides two groups of two bits each on its output leads 2082 and 2083.
081. Two bits on the lead wire 2082 take one of three patterns “00”, “01”, and “10”, but do not take “11”. The same applies to the two bits on the lead wire 2083. In addition, pattern "10" cannot appear on leads 2082 and 2083 at the same time. Thus, there are 2 ³ = 8 possible different combinations of bit values on leads 2082 and 2083.

The output bits of the precoder 208 are 1
0 bits into two groups, each group consisting of two consecutive 8 bits.
Parallel-to-serial converter 2 in order to identify the two-dimensional codeword
09 is added to the encoder / mapper 213. To this end, each group of 10 bits has a lead 2082 or 2
083 and one bit of the other input bits of the pre-encoder are divided into a 3-bit group and a 7-bit group. A group of 3 bits can thus take 3 × 2 = 6 possible patterns. They are used to select a particular four-dimensional subset pair.
Possible patterns. The 7-bit group is used to select a particular codeword from the selected group.

More specifically, the 3-bit group and the 7-bit group correspond to the 8-dimensional-4 in the block encoder / mapper 213.
Added to the dimensional array mapper converter. The four-bit subsets provide, via leads 2121, on their output lids 2134 and 2135, respectively, three-bit subsets each supplying a three-bit word identifying one of the four-dimensional subsets of the selected pair. It is added to the selector 2133.

The remaining 7 bits are divided into three groups of 3, 2, and 2 bits, respectively. The two groups of 2 bits simply pass through the converter 2131. The group of three bits is applied to a bit pattern selector 2132, which is shown identical to the bit pattern selector 2081. Thus, there are three possible combinations of 2-bit values on output lead 2136, and so on on lead 2137.

Each output lead 2136 and 2137
The upper two bits are 2 bits on leads 2138 and 2139.
Grouped together with one of the two 2-bit groups, each 3 ×
Gives a 4-bit word estimated to be one of 4 = 12-bit patterns. Each such 12-bit pattern is used to select a particular element from a particular subset of the four-dimensional subset identified by the bits on leads 2134 and 2135.

Lead wires 2136, 2138 and 2134
The upper seven bits are applied to a four-dimensional array mapper 2141 that supplies on its output leads two two-dimensional signal points P _n and P _{n + 1} that make up the first constituent four-dimensional element of the eight-dimensional codeword. . Similarly, lead wires 2137, 2139 and 213
The 7 bits on 5 are the four-dimensional array mapper 2142 which supplies two two-dimensional signal points P _{n + 2} and P _{n + 3} constituting the second constituent four-dimensional element of the eight-dimensional codeword on its output lead. Is added to The four-dimensional array mappers 2141 and 2142 are identical. As with the code V, only one such mapper is used.

The assignment of bit patterns to the four-dimensional subset pairs shown in FIG. 20 and the assignment of bit patterns to the elements of each four-dimensional subset shown in FIG. 19 are arbitrary. However, this allocation method shown in FIGS. 19 and 20 guarantees that there is no DC energy of code origin in the transmitted signal.

In particular, we assume that 4 in a particular four-dimensional subset
We start by recognizing that not all of the dimension elements are used with the same probability. This corresponds to the input leads 2136 and 2136 of the four-dimensional array mapper 2141 (2142).
Bit patterns on 138 (2137 and 2139) arise from the fact that they do not occur with the same probability. Thus, in particular, the four bit patterns “1000”, “10”
10 ”,“ 1001 ”, and“ 1011 ”each have a probability of 1.
/ 16, but the other eight bit patterns each occur with a probability of 3/32.

This situation causes a lack of DC balance in the code if care is not taken when assigning particular bit patterns to particular elements of the four-dimensional subset. However, in the preferred embodiment, the sum of the first coordinates of the four-dimensional elements is zero, and the same is true for the second, third, and fourth coordinates, so that bit patterns with equal probabilities are shown in FIG. Care is taken to ensure that each four-dimensional element of the four-dimensional subset of is assigned.

For example, as described above, two bit patterns “100” each having a probability of 1/16 are used.
Consider "0" and "1010". Referring again to FIG. 18, further two-dimensional signal point is four-dimensional elements having a radius Assuming that on a circle "A", four-dimensional subset S ₄ which is allocated the bit pattern "1000" (0,0) Are (A, 0) and (A, 0). Similarly, the coordinates of the four-dimensional elements (6,6) of the same four-dimensional subsets S ₄ which is allocated the bit pattern "1010" is, (- A, 0) and (-A, 0) It can be seen that a . Applying the above criterion, the sum of the first coordinates is (A) + (−)
A) = 0, the sum of the second coordinates is (0) + (0) = 0, the sum of the third coordinates is (A) + (− A) = 0, and the sum of the fourth coordinates is Is (0) + (0) = 0.

The criteria described above guarantee long-term DC balance. Imposing further restrictions on the assignment of bit patterns to particular four-dimensional elements across various four-dimensional subsets as shown in FIG. 19 and the assignment of bit patterns to four-dimensional subset pairs as shown in FIG. Thus, there is an effect that a short-term DC balance is also achieved.

In FIG. 19, it should be noted that the six four-dimensional elements, all of which start with a particular two-dimensional element in different four-dimensional subsets, are assigned a unique one bit pattern. Therefore, for example, all of the four-dimensional elements assigned to the bit pattern “0110” have “8” as their first two-dimensional signal points.
Having.

Further, it should be noted that not all four-dimensional subsets are used with equal probability. This means that the four-dimensional subset pair selector 2133
The input lead 2121 does not occur with equal probability. Thus, in particular, "1
The two bit patterns "00" and "101" each occur with a probability of 1/8, while the other four bit patterns each occur with a probability of 3/16.

[0111] In FIG. 19, for all four-dimensional elements of the subset S ₄ having a specific first two-dimensional point, therewith having a second two-dimensional points first two-dimensional point and the equilibrium of the same 4-dimensional elements, it should be noted that assigned to the subset S _5. Thus, for example, since the subset S ₄ is includes a four-dimensional elements (1,7), the subset S ₅ is 4-dimensional elements (1, 1). As a result of the foregoing, the subset S ₅ shows a larger number of "DC balanced" by a subset S ₄ than the other subset.

Similarly, the four-dimensional subset S ₀ is the subset S ₃
And “DC balance”, and the four-dimensional subset S ₁ shows the subset S ₂ and “DC balance”. In the preferred embodiment shown in FIG. 20, the selection of six 4-dimensional subset pairs to form an 8-dimensional array and a new "DC balanced" 4
Dimensional subsets S ₄ and probabilities other four four-dimensional subset is equal with S ₅ is used with equal probability 1/8 3/1
Attention is paid to assigning the input bit pattern to the selected four-dimensional subset pair as used in 6.

This code is equivalent to 6.7 (12.12) for a mobile channel at a medium speed of 20 (60) miles per hour.
1) It has a coding gain of dB. The MDX of this code is 2
In fact, the coding gain of this code is the smallest of all the codes described here.

Now, referring to FIG. 1 again, a preferred structure of the block decoder 51 will be described. The same basic type structure can be used for all codes described in this document. The decoding process for the assumed channel and the type of modulation / demodulation, for example the mobile channel using M-DPSK with non-coherent detection, is similar to the decoding process used for AWGN channels. This method has poor mobile channel and non-coherent M
-DPSK demodulation is not optimal in that it introduces noise that does not fit the AWGN model. However, this method has the advantage of being simple to implement, and exhibits at least near-optimal performance.

Decoding proceeds by examining each of the two-dimensional points of the particular received signal corresponding to the transmitted 2N-dimensional codeword. In particular, the so-called "two-dimensional point distance"
For each 2D receiving point, it is calculated by measuring the squared Euclidean distance between the receiving point and all possible transmitting 2D points. In performing this calculation, a particular radius of the transmission array is assumed, and the receiver is provided with an output array whose long-term average radius is at least approximately equal to the radius assumed during decoding. An automatic gain control (not shown) is provided.

The performance of the decoder is not sensitive to the exact setting of the receiving array radius. Further, to reduce the contribution of the fading (and therefore potentially unreliable) signal points to the decoding process, each of the two-dimensional point distances is
The weight may be weighted by a factor proportional to the amplitude of the corresponding received two-dimensional point.

The received N 2 2 corresponding to the transmission codeword
The dimensional received signal points are sequentially classified into N / 2 four-dimensional elements. Each of the N / 2 four-dimensional elements is further processed as follows. For each 4-D subset, one of that subset "closest" to the receiving 4-D element being processed
Find the element. This is achieved by sequentially extracting each element of the four-dimensional subset and calculating the sum of two two-dimensional point distances corresponding to the element. 4 corresponding to the minimum of such a sum, hereinafter referred to as the "four-dimensional subset distance"
The dimensional element is identified as the “closest” four-dimensional element. Therefore, as a result so far, for each received 4D element,
In each of the four-dimensional subsets, a particular four-dimensional element and its associated four-dimensional subset distance are identified.

If the code is a four-dimensional code, there is only one four-dimensional subset. Thus, only one four-dimensional element is identified and that element is taken to be the transmission codeword. If the code is an eight-dimensional code or more, the above process is repeated for each of the received eight-dimensional elements.

In particular, the received N / 2 four-dimensional receiving elements are sequentially grouped into N / 4 eight-dimensional elements, each of which is then processed in a manner similar to the four-dimensional case. You. In particular, for each 8-D subset, find one 8-D element of that subset that is "closest" to the received 8-D element being processed. This is the 4th of the 8D subset
This is achieved by taking out the dimensional subset pairs in turn and summing the two four-dimensional subset distances corresponding to the pairs. The eight-dimensional element in the four-dimensional subset pair is one element of each four-dimensional subset obtained from the previous step.
A pair of dimensional elements, corresponding to the smallest such sum, and further referred to as an "eight-dimensional subset distance";
Specified as the “closest” eight-dimensional element.

If the code is an eight-dimensional code, one eight-dimensional element has now been specified. Otherwise, the process is repeated again for 16 dimensions and so on.

The identified codeword is then
The data bits are mapped to data bits based on the bit allocation method for codes as shown in FIGS.

The decoding process described above can be performed in a highly parallel manner, and is therefore more complex and therefore more efficient, practical implementation of block coding schemes and / or higher input data rates. This has the effect of reducing the time required for decoding that allows In particular, the two-dimensional element distance can be calculated simultaneously for each of the N two-dimensional points of the received 2N-dimensional signal. The same applies to the four-dimensional and eight-dimensional subset distances. Further, once multiple 2N-dimensional signals are available at the deinterleaver output, each of those signals is decoded in parallel independently of the others. This property of a block code is not possible with a trellis code, for example, where each two-dimensional point needs to be processed in series.

According to the present invention, this type of code can be combined with spatial diversity to further increase performance. In particular, a receiving station such as a car is provided with two or more receiving antennas separated by at least a quarter wavelength, so that the power received from each antenna fades independently of each other.

Then, the first step of the decoding process is performed on the signals received from each antenna. The resulting "preliminary" two-dimensional point distances associated with each point in the array are added to the two-dimensional point distances for points obtained from other antennas. The resulting sum is the "final" two-dimensional point distance that will be used as the two-dimensional point distance in the decoding steps that follow.

The foregoing merely illustrates the gist of the present invention. For example, while the invention has been described primarily with reference to digital cellular mobile communications, it is equally applicable to other fading signal environments. Further, while the above description has assumed that there is only one user per mobile communication channel,
If a frequency division multiple access (FDMA) scheme is used, the invention is equally applicable to the so-called TMDA scheme, where some users are time multiplexed on one channel. In such applications, the interleaver design and T
The formats of the MDA frames match each other with the largest possible size.

In particular, the interleaving length is set to TMDA
It is an integral multiple of the frame length and is large enough to guarantee a proper time interval between codeword signal points. This leads to the requirement that signal points from a codeword be spread in two or more TMDA frames. Instead, the number of time slots allocated to each user within a TDMA frame, which is divided as much as possible within the frame, is 1
It is increased to a larger number, thereby reducing interleaving length requirements and / or increasing performance. These considerations favor the use of coding schemes as low as possible that allow other system goals to be achieved.

Trellis coding schemes require larger interleaving lengths to introduce delay or generally require more time slots per TMDA frame and per user that do not match other system requirements. Thus, these same considerations make the trellis coding scheme less attractive than the block coded modulation scheme of the present invention.

A further advantage of the use of block coded modulation, for example when compared to trellis coded modulation, is that once a signal captured within an interleaving frame is received, the information, unlike trellis coded, Are all to be decoded immediately. This property of block coded modulation avoids delays that may otherwise occur with trellis coding schemes if the system has a speech decoder next to a channel decoder.

[0129] Other variations are possible. For example, M = 4
Various values of the parameter M, such as 6, 8, 12, 16, etc., can be selected. Any integer greater than 1 can be used as parameter N. In each of the M and N combinations, a variety of different codes are reached that provide various combinations of bandwidth efficiency, coding gain, and implementation complexity. X is 2
Codes with X times diversity other than are constructed similarly. Only a few of the possible codes have been given in this. The particular code used in a particular application chosen is a function of the particular needs and limitations of that application.

As an example, increasing the signal rate, or reciprocal of the signal interval, introduces and / or increases intersymbol interference.
Relatively high, even at the expense of a possible increase in complexity and / or a possible decrease in coding gain, to maintain a given data rate without encountering a modest amount of intersymbol interference Choose a code with bandwidth efficiency so that the signal rate is kept acceptably low.

In addition, the diversity built into the code described herein can be implemented in the time domain (by ensuring that the transmission of the signal points providing the diversity occurs in well-separated signal intervals). The diversity can also be used in other areas, for example, by transmitting those signal points on physically separated and independently fading channels. Thus, while the symbols disclosed herein have been described as having built-in time diversity, more generally, built-in time diversity may be described as an example, "built-in diversity".

The above description relates to one embodiment of the present invention, and those skilled in the art can consider various modifications of the present invention, but all of them are within the technical scope of the present invention. Is included. It should be noted that reference numerals in the claims are for the purpose of easy understanding of the invention and should not be construed as limiting the technical scope thereof.

As described above, according to the present invention, a block coded modulation technique with interleaving having built-in time diversity is applied to a fading channel. This achieves excellent coding gain with less complex implementation, less delay in the interleaver / deinterleaver and encoder, and high bandwidth efficiency.

[Brief description of the drawings]

FIG. 1 is a block diagram of a data communication system showing one embodiment of the present invention.

FIG. 2 is a diagram illustrating technical terms and concepts.

FIG. 3 illustrates an 8-PSK constellation that forms the basis of many of the block codes disclosed herein.

FIG. 4 is a diagram illustrating a codebook for a first block coded modulation method according to the present invention;

FIG. 5 is a diagram illustrating an operation of the interleaver shown in FIG. 1 with respect to a first block coded modulation method.

FIG. 6 is a diagram graphically illustrating a second block coded modulation method according to the present invention.

FIG. 7 is a diagram graphically illustrating a second block coded modulation method according to the present invention.

FIG. 8 is a diagram graphically illustrating a third block coded modulation method according to the present invention.

FIG. 9 is a diagram graphically illustrating a third block coded modulation method according to the present invention.

FIG. 10 is a diagram illustrating an operation of an interleaver for a block coded modulation method.

FIG. 11 is a diagram graphically illustrating a fourth block coded modulation method according to the present invention.

FIG. 12 is a diagram graphically illustrating a fourth block coded modulation method according to the present invention.

FIG. 13 is a diagram graphically illustrating a fourth block coded modulation method according to the present invention.

FIG. 14 is a block diagram of the encoder / mapper of FIG. 1 for a fifth block coded modulation method according to the present invention.

FIG. 15 is a detailed view showing various portions of the encoder / mapper of FIG.

FIG. 16 is a detailed view showing various portions of the encoder / mapper of FIG.

FIG. 17 is a detailed view showing various portions of the encoder / mapper of FIG.

FIG. 18 is a diagram illustrating a 12-PSK array forming a basis of a sixth block coded modulation method according to the present invention.

FIG. 19 is a diagram graphically illustrating a sixth block coded modulation method according to the present invention.

FIG. 20 is a diagram graphically illustrating a sixth block coded modulation method according to the present invention.

FIG. 21 is a circuit block diagram for executing a sixth block coded modulation method according to the present invention.

[Explanation of symbols]

 Reference Signs List 11 input data 13 2N-dimensional block encoder / mapper 21 interleaver 30 fading channel 44 deinterleaver 51 block decoder

Continuation of the front page (56) References JP-A-58-151740 (JP, A) Kenichi Konishi, Yoshiya Miyazaki, "Error Rate Characteristics of Block Coded Multi-Level Modulation / Hard Decision Scheme in Rayleigh Fading Channel," Journal of Child Information and Communication Society, B-2, Vol. J-B-2 No. 11 683-690. IEEE INTERNATIONAL CONFERENCE ON COMMUNICATIONS 1988, 12-15 JUN 1988, PHILADELP HIA, US; 2 OF 3, PAGES 762-767; LI FUNG CHANG; "BLOCK-CODE D-MODULATION FOR DIGITAL RADIO COMMUNICATIONS SYSTEM"

Claims (26)

(57) [Claims] 1. A receiving means for receiving a sequence of blocks of input data, defining a character set of codewords each of which is associated with one of the possible values of the input data block, and Definition and generation means for generating an associated one of the codewords corresponding to the blocks, wherein each of the codewords of the script system comprises at least two modulation signal points taken from a predetermined array of modulation signal points. Where X is an integer greater than one, each codeword is at least X signals long with other codewords.
Apparatus having built-in X-fold diversity that differs at point locations, wherein the least square Euclidean distance between the codewords is greater than the absolute least square Euclidean distance for the script. 2. The apparatus according to claim 1, further comprising transmission means for transmitting a signal representing the generated codeword. 3. The method of claim 2, wherein the input data block is received at a rate of m bits every T seconds, which is a predetermined signal interval, and the array includes 2 ^m modulated signal points. An apparatus according to claim 1. 4. The apparatus of claim 1, wherein the array of modulation signal points is an M-PSK array where M is a selected integer. 5. The definition and generation means encodes each of the input data blocks into a block of coded bits and a signal comprising a code word generated in accordance with each of the blocks of coded bits. By selecting points,
Apparatus according to claim 1, characterized in that a code word association is generated. 6. The apparatus of claim 1, wherein each of the codewords of the script differs from each of the other codewords of the script at least at a particular signal point location X. 7. Each of the codewords of the script differs from each of the other codewords of the script at least at the position X of a particular signal point, and the apparatus further comprises generating during each signal interval. The apparatus of claim 1, further comprising transmission means for transmitting a signal point including a codeword. 8. The generated codewords occur in consecutive frames of codewords, and the apparatus further comprises: the signal points at corresponding signal point locations of the codewords in each frame are grouped together. 2. The apparatus according to claim 1, further comprising rearrangement means for rearranging signal points of codewords of each frame. 9. The apparatus of claim 8, further comprising transmission means for transmitting a signal representing the rearranged signal points using a carrier phase difference. 10. The apparatus of claim 8, further comprising transmission means for transmitting a signal representing said rearranged signal points using a carrier phase difference increased by a constant value. 11. The generated codewords occur in successive frames of codewords, each of the codewords consisting of N signal points,
The apparatus rearranges the codeword signal points of each frame such that the i-th respective signal points of the codewords of the frame are arranged in respective groups for i = 1, 2,..., N. 7. The apparatus of claim 6, further comprising reordering means for performing the reordering. 12. The generated codewords occur in successive frames of codewords, and the apparatus further comprises a codeword for each frame such that the separation between signal points in each codeword in the reordered sequence is increased. 7. The apparatus of claim 6, further comprising: generating means for generating a rearranged sequence of signal points. 13. The generated codewords each occur within a continuous frame of J codewords, and certain of the codewords of the script are two of the kth and qth signal point locations. , And the apparatus described above rearranges the codeword signal points of each frame such that the signal points at the kth and qth signal point positions are separated by more than J signal points in succession. 7. The apparatus according to claim 6, further comprising generating means for generating the generated sequence. 14. Receiving a sequence of blocks of input data, and corresponding codewords of a codeword script corresponding to one of the possible values of the input data block corresponding to the block of input data. Generating each of the codewords of the alphabet comprises a combination of at least two modulation signal points taken from a predetermined array of modulation signal points, wherein the alphabet of the codeword represents X If each codeword is another integer and at least X signal points
A coded modulation method having a built-in X-fold diversity that differs at different locations, wherein the least square Euclidean distance between the codewords is greater than the absolute least squares Euclidean distance associated with the script. 15. The method of claim 1, further comprising transmitting a signal representing the generated codeword.
4. The method according to 4. 16. The block of input data is received at a rate of m bits every predetermined signal interval T seconds, and the array includes at least 2 ^m modulated signal points. The method according to claim 15, wherein 17. The method of claim 16, wherein the array of modulation signal points is an M-PSK array where M is a selected integer. 18. In the generating step, one of the codewords encodes each block of input data into a block of coded bits and one of the codewords corresponding to each block of coded bits. 18. The method according to claim 17, wherein the method is generated by selecting signal points. 19. The method of claim 14, wherein each of the codewords in the script differs from each other codeword in at least the particular signal point location X. 20. Each of the codewords in the script differs from each of the other codewords in the script at least at a particular signal point location X, and the method further includes a codeword generated in each signal interval. The method of claim 18, further comprising transmitting a signal point. 21. The generated codewords occur in consecutive frames of codewords, and the method further comprises grouping signal points at corresponding signal point locations of the codewords in each frame. 20. The method of claim 19, further comprising the step of rearranging signal points of codewords of each frame. 22. The method of claim 21, further comprising transmitting a signal representing said rearranged signal points using a carrier phase difference. 23. The method of claim 21, further comprising transmitting a signal representing the reordered signal points using a carrier phase difference increased by a constant. 24. The generated codewords occur in successive frames of codewords, each of the codewords consisting of N signal points, the method comprising: i = 1, 2,. 20. The method of claim 19, further comprising the step of rearranging the signal points of the codeword of each frame such that the respective ith respective signal points of the codewords of the frame are arranged in respective groups. Method. 25. The method of claim 25, wherein the generated codeword occurs in successive frames of the codeword, and the method comprises: increasing separation between signal points in each codeword in the reordered sequence.
20. The invention of claim 19, further comprising the step of generating a reordered sequence of codeword signal points for each frame. 26. Each of the generated code words is J
Occurring in successive frames of codewords, particular ones of the codewords of the script system differ in exactly two X signal point locations, two of which are the kth and qth signal point locations; Generating a rearranged sequence of codeword signal points for each frame such that the signal points at the qth and qth signal point locations are separated by more than J signal points in the sequence. The method according to claim 19.