JP2633598B2 - Magnetron drive - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] (Industrial application field) The present invention relates to a magnetron driving device used for a high-frequency heating device such as a microwave oven, and in particular, applies a stable heater voltage to a filament of a magnetron device. The present invention relates to a magnetron driving device supplied to stabilize the operation of a magnetron device.

(Prior Art) It is desirable that a high frequency heating device such as a microwave oven, which has been widely used at present, is small, lightweight and designed for a general user. In response to this demand, in recent years, power transformers for driving magnetrons have low overcurrent loss at high-frequency currents, and ferrite that has sufficient magnetic flux density as well as iron core transformers that use silicon steel plates as cores despite their small size. A high frequency transformer using a dust core such as a core for the core is used. Then, to the primary side of this high-frequency transformer, after rectifying the commercial electrode, high-frequency power whose frequency has been converted by an inverter device is supplied,
This high-frequency power is boosted and rectified by the voltage doubler rectifier circuit, and then applied to the anode of the magnetron. As described above, by using a small and lightweight high-frequency transformer for inputting high-frequency power from the inverter device to the power supply unit, a high-frequency heating device, for example, a microwave oven is further reduced in size and weight, and furthermore, design constraints are imposed. Will also loosen.

FIG. 9 is a circuit diagram of a conventional magnetron driving device using such an inverter power supply.

In the figure, a smoothing circuit 4A including a choke coil 3 and a smoothing capacitor 4 is connected to a rectifying stack 2 for inputting a commercial power supply 1 and performing full-wave rectification. One end of the wire 5a is connected, and the other end of the primary winding 5a is connected via a resonance capacitor 6 to the minus line of the smoothing circuit 4A. . The primary winding 5a and the resonance capacitor 6 form a series resonance circuit. A freewheeling diode 7 and a collector and an emitter of a transistor 8 are connected in parallel to both ends of the resonance capacitor 6, and the transistor 8 is turned on and off by a control circuit 9 connected to the base.
F controlled. The series resonance circuit and the transistor control circuit 9 constitute a quasi-E-class inverter.

On the other hand, the secondary winding 5b of the step-up transformer 5 has a voltage doubler rectifier circuit 11 comprising a rectifier diode 10 and a charging capacitor 11.
A is connected, the plus output of the voltage doubler rectifier circuit 11A is connected to the anode AD of the magnetron 12, and the minus side is connected to the filament F of the magnetron 12.
Further, an AC voltage as a heater voltage is supplied to the filament F from the commercial power supply 1 by the step-up transformer 5 and another heater transformer 100.

Next, the operation of the conventional magnetron driving device will be described according to the above configuration. First. Commercial power supply 1 is rectifying stack 2
After full-wave rectification, the full-wave rectified output is
Is input to a series resonance circuit including a primary winding 5a of the step-up transformer 5 and a resonance capacitor. At this time, when the transistor 8 connected in parallel with the resonance capacitor 6 is turned on by the control circuit 9 for a predetermined time, a primary current flows through the primary winding 5a, and when the transistor 8 is turned off, the primary winding 5a and the resonance capacitor 6 By entering this resonance state and repeating this state, a changing current flows through the primary winding 5a, and a high-frequency magnetic field is generated in the primary winding 5a. The high-frequency magnetic field induces a high-frequency electromotive force on the secondary winding 5b side. The high-frequency electromotive force is boosted to about 4000 V DC by the voltage doubler rectifier circuit 11A, and then applied to the anode AD of the magnetron 12. The filament F of the magnetron 12 has a heater transformer 100 different from the step-up transformer 5.
As a result, a heating voltage of about 4 V is supplied, so that thermoelectrons are emitted from the filament F toward the anode AD. This causes the magnetron 12 to oscillate and emit microwaves. Heating is performed by irradiating the object to be heated with the high frequency (microwave) emitted from the magnetron.

However, the conventional magnetron driving device as described above requires a heater transformer 100 for the filament in addition to the step-up transformer 5 for the anode voltage, and the heater transformer 100 is a silicon steel plate in the core because of the commercial power frequency. It is relatively large and important. Therefore,
Inverter power supply to reduce the size and weight of the entire device,
In addition, the advantage of using a high-frequency transformer such as a small and lightweight ferrite core is lost.

In the case of using the conventional magnetron driving device shown in FIG. 9, the primary winding 10 shown in FIG.
A voltage proportional to the input voltage Vin on the 0a side is supplied to the filament F of the magnetron 12 as a heater voltage Pf. That is, when the input voltage Vin on the primary winding 100a side changes within a range of plus 10% or minus 10%, the heater voltage Pf fluctuates according to the change of the input voltage Vin. There is a drawback that the fluctuation range becomes large.

Therefore, a magnetron driving device as shown in FIG. 11 has been proposed in order to solve the above-mentioned disadvantages.

In the conventional example shown in FIG. 11, a heater winding 5c is provided on the secondary side of the transformer 5, and the heater voltage obtained by the heater winding 5c is supplied to the filament F of the magnetron 12, so that the magnetron 12 By using a high-frequency transformer for the heater voltage supplied to the filament F, the size and weight of the entire apparatus can be reduced.

(Problems to be Solved by the Invention) However, in the conventional example shown in FIG.
The value of the heater voltage Pf of the heater winding 5c side according to variation of the high frequency power P ₀ of 5b side fluctuates. That is, when lowering the RF power supplied P ₀ on the anode AD side of the magnetron 12 as shown in FIG. 12, will be reduced even heater voltage Pf with a decrease of the high frequency power P _0, the filament Insufficient thermoelectrons are sent from F to the anode AD, causing so-called moding in which the oscillation operation of the magnetron 12 becomes unstable. Conversely, the anode AD of the magnetron 12
When the value of the RF power P ₀ is increased to be supplied to, also increases the value of the heater voltage Pf, there is a risk that the filament F resulting in burnout. Therefore, a problem that can be varied over a wide range to supply the high frequency power P ₀ to the anode AD becomes difficult occurs.

The present invention has been made in view of the above, and provides a magnetron driving device capable of stably operating a magnetron by reducing the size and weight of the entire device and stabilizing a heater voltage supplied to a filament. The purpose is to do.

[Configuration of the Invention] (Means for Solving the Problems) In order to achieve the above object, the present invention provides a commercial power
A first inverter device for converting the first high-frequency voltage into a high-frequency voltage, a first transformer for transforming the first high-frequency voltage and supplying the same to the anode side of the magnetron device, and conducting / non-conducting control of the commercial power supply by switching means. A magnetron drive comprising: a second inverter device for converting to a second high-frequency voltage; and a second transformer for transforming the second high-frequency voltage and supplying the transformed predetermined heater voltage to a filament of the magnetron device. A voltage detecting means for detecting a voltage change of the commercial power supply;
Of the switching means on the primary side of the transformer
A winding provided separately from the primary winding, the conduction current of which is controlled by the non-conduction control, and a winding which is used for the second inverter device and varies depending on the voltage state of the primary winding induced in the winding. Power control means for starting conduction of the switching means and controlling the conduction time of the switching means according to the voltage fluctuation of the commercial power supply detected by the voltage detection means.

(Operation) In the magnetron drive device of the present invention, when the first inverter device converts a commercial power supply to a first high-frequency voltage, the first transformer converts the first high-frequency voltage to a predetermined voltage, and the Supply to the anode side. When the second inverter device independent of the first inverter device converts the commercial power supply into a second high-frequency voltage, the second transformer transforms the second high-frequency voltage into a predetermined heater voltage. Then, the predetermined heater voltage is supplied to the filament of the magnetron device. Further, in the second inverter device, the power control means causes the switching means to start conducting according to the voltage state of the primary winding of the second transformer, and the conduction time of the switching means according to the voltage fluctuation of the commercial power supply. Is controlling.

(Example) Hereinafter, an example according to the present invention will be described in detail with reference to the drawings.

First, an overall configuration of a magnetron driving device to which the present invention is applied will be described with reference to FIG.

In FIG. 1, the circuit configuration for supplying high-frequency power to the anode AD of the magnetron 12 is the same as that of the conventional example shown in FIG. 9, and the same reference numerals are given and the detailed description is omitted.

A series resonance circuit formed by the primary winding 5a and the resonance capacitor 6, a free-wheeling diode 7, a transistor 8 and an output control circuit 9 for converting a commercial power supply to a first high-frequency voltage. One inverter device is configured. The step-up transformer 5 constitutes a first transformer for transforming the first high-frequency voltage to a predetermined voltage and supplying it to the anode AD side of the magnetron 12.

Next, a circuit configuration for supplying a predetermined heater voltage to the filament F of the magnetron 12 will be described.

The positive side of the primary winding 5a of the step-up transformer 5 is a diode
It is connected to the primary winding 23a of the high-frequency transformer 23 via 21. The cathode of the diode 21 is connected to one end of the smoothing capacitor C1 and each of the voltage detection circuits 25, and the other end of the smoothing capacitor C1 is connected to the minus side of the rectifying stack 2. Also, primary winding 23
parallel resonant capacitor C ₂ and a are connected. And the resonance capacitor C ₂ to form a parallel resonant circuit with the primary winding 23a. That is, by the resonance capacitor C ₂ is repeated charging and discharging, energy transfer between the resonant capacitor C ₂ and the primary winding 23a is performed, a high-frequency magnetic field is generated in the primary winding 23a. This high frequency magnetic field causes the secondary winding 23
A predetermined heater voltage is induced in b. Also, a resistor is provided between one line of the primary winding 23a and the base of the transistor Q1.
R1 is connected, and a base current is supplied to the transistor Q1 at the start of operation via the resistor R1. Also, primary winding
The other end of 23a is connected to the collector of transistor Q1 and the cathode of flywheel diode D1. The emitter of the transistor Q1 is connected to the current detection circuit 27. The power control circuit 29 has the base of the transistor Q1,
It is connected to each of the current detection circuit 27 and the voltage detection circuit 25. The power control circuit 29 is connected to the second secondary winding 23C of the transformer 23. The base current of the transistor Q1 is supplied from the second secondary winding. The transistor Q1 is connected to the primary winding 23a of the transformer 23. A switching means for interrupting the current flowing through the switch.
Voltage detection circuit 25 outputs a signal corresponding to the voltage fluctuation of commercial power supply 1. In the current detection circuit 27, the transistor Q1 is ON.
Sometimes it outputs a signal corresponding to the value of the current flowing through the primary winding 23a. Additionally the power control circuit 29 controls the operation of the transistor Q ₁ is a switching means on the basis of the output of the voltage detection circuit 25 and the current detection circuit 27. That is, the power control circuit
When it is determined that the power supply voltage of the commercial power supply is low based on the signal from the voltage detection circuit 25, 29 sets the conduction time of the transistor Q1 as the switching means to be long. Conversely, when the power control circuit 29 determines that the power supply voltage of the commercial power supply is high based on the signal from the voltage detection circuit 25, it sets the conduction time of the transistor Q1 short.

By controlling the conduction time of the transistor Q1 as the switching means in this manner, the secondary winding 23 of the transformer 23 is controlled.
A predetermined stabilized heater voltage induced from b is supplied to the filament F of the magnetron 12.

Further, the above-described parallel resonance circuit, the transistor Q1, the free wheeling diode D1, and the power control circuit 29 constitute a second inverter device, and convert a commercial power supply into a second high-frequency voltage. The transformer 23 constitutes a second transformer for supplying a predetermined heater voltage to the filament F of the magnetron 12.

Next, a circuit section for supplying a predetermined heater voltage to the filament F of the magnetron 12 will be described in more detail with reference to FIG.

 First, the internal configuration of the voltage detection circuit 25 will be described.

One line of the primary winding 23a of the transformer 23 is grounded via a resistor R17 and a resistor R18. The voltage divided by the resistors R17 and R18 is applied to the operational amplifier OP1 via the resistor R21.
To the inverting input terminal. A predetermined DC voltage Vcc supplied from a constant voltage circuit (not shown) is connected to the resistor R19 and the resistor R19.
The voltage is divided by R20, and the divided voltage is supplied to the non-inverting input terminal of the operational amplifier OP1 via the resistor R22. Accordingly, the voltage Vc ₂ is given corresponding to the voltage variation of the commercial power supply as shown in FIG. 3 (a) to the non-inverting input terminal of the operational amplifier OP1. Further, a constant DC voltage Vc ₁ is applied to the non-inverting input terminal of the operational amplifier OP1. Further, the inverting input terminal of the operational amplifier OP1 is connected to the output terminal via the resistor R23 for a period. The output terminal of the operational amplifier OP1 is connected to ground via R2 and R25, and a capacitor C4 is connected in parallel to the resistor R25. Thus Figure 3 (b) to the operational amplifier OP1 as indicated outputs a signal voltage that decreases in response to the input voltage Vc ₂ increases. The output voltage of such a calculation width unit OP1 is divided by resistors R24 and resistor R25, providing the divided voltage Vf ₁ as shown in FIG. 3 (c) to the inverting input terminal of the comparator CP2.

The non-inverting input terminal of comparator CP2 is transistor Q1
And connected to ground via a resistor R2. This resistor R2 forms the current detection circuit 27. That is, when the base current of the transistor Q1 is given, the transistor Q1 turns on, and the collector current becomes the resistance R2.
Flows through. Therefore, the resistor R2 comparator CP2 signal voltage corresponding to the current value of the collector current as the input voltage Vc ₂
To the non-inverting input terminal.

Next, the internal configuration of the power control circuit 29 will be described. The inverting input terminal of the comparator CP2 with a constant input voltage Vf ₁ as shown in FIGS. 4 and 5 is given, the collector current Ia as described above to the non-inverting input terminal of the comparator CP2
Signal voltage Vf ₂ corresponding to the current value of. Thus the comparator CP2 outputs a signal input voltage Vf ₂ of the non-inverting input terminal is only an H level during the time that exceeds the input voltage Vf ₁ of the inverting input terminal.

Here, the operation when the power supply voltage of the commercial power supply 1 fluctuates will be described with reference to FIG. 4 and FIG.

The power supply voltage of the commercial power supply 1 fluctuates. For example, as shown in FIG. 4, the input voltage Vin of the primary winding 23a of the high-frequency transformer 23 becomes α
When reduced by the value of the input voltage Vf ₁ of the inverting input terminal of the comparator CP2 rises. The input voltage Vf ₂ is applied to the non-inverting input terminal of the comparator CP2 simultaneously increased in accordance with the current value of the collector current Ia. Thus the timing of input voltage Vf ₂ of the non-inverting input terminal exceeds the input voltage Vf ₁ of the inverting input terminal is delayed. This allows the comparator
CP2 sends an H-level signal at a later timing than normal.

Next, referring to FIG.
The operation in the case where the value of the input voltage Vin on the a side has increased by plus α will be described.

When the value of the input voltage Vin of the primary winding 23a side is increased by plus alpha, the value of the input voltage Vf ₁ of the inverting input terminal of the comparator CP2 as shown in FIG. 5 (a) becomes lower. The input voltage Vf ₂ of the non-inverting input terminal of the comparator CP2 will vary according to the current value of the collector current Ia shown in FIG. 5 (b). Thus when the value of the input voltage Vin of the primary winding 5a side rises by plus α is timing when the value of the input voltage Vf ₂ exceeds the value of the input voltage Vf ₁ is accelerated. This allows
The comparator CP2 sends a H-level signal with a fast cycle.

The output terminal of the comparator CP2 is a flip-flop circuit F
Connected to terminal A of F2. A predetermined DC voltage Vcc is applied to a terminal A of the flip-flop circuit FF2 via a resistor R11. The terminal B of the flip-flop circuit FF2
Is supplied with a predetermined DC voltage Vcc. The terminal CD of the flip-flop circuit FF2 is connected to the output terminal of the flip-flop circuit FF1. Furthermore, flip-flop circuit FF
The capacitor C5 is connected between the terminals T1 and T2 of the second terminal. This terminal T2 is connected to a predetermined DC voltage V via a resistor R12.
cc is given. This resistor R12 has a diode D4
Are connected in parallel. Furthermore, flip-flop circuit FF
The output terminal of 2 is a predetermined DC voltage V via resistors R14 and R13.
cc is given. The connection point between the resistors R13 and R14 is connected to the base of the transistor Q3, and the collector of the transistor Q3 is connected to ground via the resistors R15 and R16. The connection point between the resistors R15 and R16 is connected to the base of the transistor Q2. The emitter of the transistor Q2 is connected to ground, and the collector of the transistor Q2 is connected to the diode D3 and the base of the transistor Q1.

Accordingly, when the H-level signal from the comparator CP2 as shown in FIG. 6 (g) is input to the terminal A, the flip-flop circuit FF2 inverts the flip-flop circuit FF2, as shown in FIG. 6 (h). An L level signal is output. The maximum value of the pulse width of the L-level pulse signal sent from the output terminal of the flip-flop circuit FF2 is a resistance value.
It is determined by the time constant of the time constant circuit formed by R12 and the capacitor C5. When an L-level signal as shown in FIG. 6 (h) is transmitted from the output terminal of the flip-flop circuit FF2, the transistors Q3 and Q2 are turned on and the transistor Q1 is turned off.

A resistor R connected between the specified DC power supply VCC and ground
5 and has given to the inverting input terminal of the comparator CP1 a divided voltage between the resistor R6 as the input voltage Vc _2. Moreover, this non-inverting input terminal of the comparator CP1, giving the connected partial voltage of the resistor R7 and a resistor R8 between the predetermined voltage Vcc and the ground as the input voltage Vc _1. The non-inverting input terminal of the comparator CP1 is connected to the second secondary winding 23c of the transformer 23 via the resistor R4. This second secondary winding 23c is connected to the base of the transistor Q1 via the capacitor C3 and the resistor R3. The capacitor C3 has a diode
D2 is connected in parallel.

Therefore, the input voltage V is applied to the inverting input terminal of the comparator CP1.
constant DC current is applied as shown as c ₂ in FIG. 6 (c). The signal voltage corresponding to the operation of the transistor Q1 as shown in FIG. 6 as the input voltage Vc ₁ (c) are considered given the comparator CP1 non-inverting input terminal. Therefore, the comparator CP1 is the value of the input voltage Vc ₁ is input voltage Vc ₂
The signal which becomes H level only when the value exceeds the value is transmitted from the output terminal. The output terminal of the comparator CP1 is connected to the resistor R9 and the input terminal A of the flip-flop circuit FF1. A predetermined DC voltage Vcc is applied to each of the input terminal B and the terminal CD of the flip-flop circuit FF1. Further, a capacitor C6 is connected between the terminal T1 and the terminal T2 of the flip-flop circuit FF1. Also this terminal T2
Is supplied with a predetermined DC voltage Vcc via a resistor R10. Further, a diode D5 is connected in parallel to the resistor R10.

Accordingly, as shown in FIG. 6 (e), the flip-flop circuit FF1 outputs a signal which is inverted when the rising output of the H level is obtained from the comparator CP1 and falls to the L level. This L level pulse width is determined by the time constant of the resistor R10 and the capacitor C6.

Next, the operation of the embodiment according to the present invention shown in FIG.
This will be described with reference to the drawings.

When a base current is supplied to the transistor Q1 via the resistor R1, the transistor Q1 conducts. This transistor
Due to the conduction of Q1, a collector current Ia as shown in FIG. 6 (a) flows through the resistor R2. Thereby, as shown in FIG. 6 (f), the input voltage Vf corresponding to the current value of the collector current Ia
₂ is supplied to the non-inverting input terminal of the comparator CP2.

On the other hand, in the voltage detection circuit 25, the operational amplifier OP1 is connected to the primary winding 2
3a monitors the variation of the commercial based on variations of the input voltage of the signal voltage corresponding to the variation of the commercial power supply, i.e., the input voltage Vf ₁ as shown in FIG. 6 (f) the comparator CP2
To the inverted input terminal. Thus the signal voltage as shown in FIG. 6 (g) from the output terminal of the comparator CP2, i.e. outputs a signal which becomes H level only during the time above the value of the input voltage Vf ₂ the value of the input voltage Vf _1. Comparator CP2
The H-level signal from the flip-flop circuit
When sent to the input terminal A of FF2, the flip-flop circuit FF2 is inverted and a predetermined pulse whose pulse width is determined by the time constant of the resistor R12 and the capacitor C5 as shown in FIG. An L level pulse signal having a width is output. When an L level pulse signal is output from the flip-flop circuit FF2, the transistors Q3 and Q2 are simultaneously turned on for a time corresponding to the pulse width of the L level pulse. When the transistors Q3 and Q2 conduct as described above, the transistors as shown in FIGS.
The transistor Q1 is turned off only during the time when the transistor Q3 and the transistor Q2 conduct. The non-inverting input terminal of the comparator CP1 has the ON-state of the transistor Q1 as shown in FIG.
OFF operation is given an input voltage Vc ₁ corresponding to the comparator CP1 is a transistor Q as shown in Figure No. 6 (d)
Outputs a signal corresponding to 1 ON-OFF operation. The flip-flop circuit FF1 sends a signal that falls to L level when the signal that rises to H level is input from the comparator CP1 to the terminal CD of the flip-flop circuit FF2. Therefore, the flip-flop circuit FF2 is restored when an L-level signal is input to the terminal CD, the transistor Q2 is turned off, and a base current flows through the transistor Q1 due to the voltage generated in the second secondary winding 23C, and the transistor Q1 is turned on. I do.

When the flip-flop circuit FF2 receives the H level signal of the next cycle from the comparator CP2, it inverts again and outputs an H level signal to make the transistors Q3 and Q2 conductive. Hereinafter, similarly, the flip-flop circuit FF2 controls the ON-OFF operation of the transistors Q3 and Q2,
The operation cycle of the ON-OFF operation of the transistor Q1 is controlled.

When the ON-OFF operation of the transistor Q1 is continued as described above, a high-frequency current flows through the primary winding 23a to generate a magnetic field,
This high-frequency magnetic field induces a predetermined heater voltage on the secondary winding 23b side. The predetermined heater voltage induced in the secondary winding 23b is supplied to the filament F of the magnetron 12.

Since the transistor Q1 is turned on and off based on the control operation of the power control circuit 29,
A stabilized predetermined heater voltage as shown in FIG. 7 can be supplied to the filament F.

Also, as shown in FIG. 7, the input voltage Vin on the primary winding 23a side is
Of the input voltage Vin of the primary winding 23a is monitored by the voltage detection circuit 25 even if the voltage of the input terminal fluctuates within the range of plus 10% or minus 10%. , The ON-OFF operation cycle of the transistor Q1 is controlled on the basis of the constant heater voltage P even when the input voltage Vin fluctuates.
f can be supplied to the filament F.

Further, in the example shown in FIG. 2, the circuit configuration can be simplified because a so-called self-excited inverter device is provided.

Further, since the transistor Q2 is provided on the base side of the transistor Q1, the ON-OFF operation of the transistor Q1 can be reliably controlled even if the characteristics of the transistor Q2 vary.

Further, since the predetermined heater voltage induced from the secondary winding 23b is directly supplied to the filament F, it is possible to easily perform insulation measures on the secondary side of the transformer 23.

Next, another embodiment according to the present invention will be described with reference to FIG.

FIG. 8 is a circuit diagram showing another configuration example of the voltage detection circuit 25 shown in FIG.

The example shown in FIG. 8 is characterized in that the voltage fluctuation of the commercial power supply 1 is directly monitored by the photocoupler PC1.

The photocoupler PC1 includes photodiodes PD1 and PD2 connected in parallel in opposite directions, and a phototransistor PT for receiving light emitted from the photodiodes PD1 and PD2. The photodiodes PD1 and PD2 connected in parallel to each other are connected to the commercial power supply 1 via the resistor R31. Therefore, the photodiodes PD1 and PD2 send out to the phototransistor PT an amount of emitted light corresponding to the fluctuation of the power supply voltage of the commercial power supply 1. When the phototransistor PT receives the emission light according to the fluctuation of the power supply voltage of the commercial power supply 1 from the photodiode PD1 or PD2, the phototransistor PT flows a current according to the intensity of the received light,
A signal voltage corresponding to the fluctuation of the power supply voltage of the commercial power supply 1 is sent to the inverting input terminal of the operational amplifier OP1.

Therefore, in the example shown in FIG. 8, the fluctuation of the power supply voltage of the commercial power supply 1 can be directly monitored, and the ON / OFF control of the transistor Q1 can be more closely performed, thereby enabling the filament F to be controlled.
The heater voltage supplied to the heater can be further stabilized.

Also, in the example shown in FIGS.
Monitors the voltage fluctuation of the commercial power supply, and turns on / off the transistor Q1 based on the detection signal of the voltage detection circuit 25.
Since the operation is controlled, the heater voltage can be stabilized without detecting a change in the voltage on the secondary winding 23b side, that is, the heater voltage.

[Effects of the Invention] As described above, according to the present invention, a second inverter device is provided independently of a circuit unit for supplying a high-frequency voltage to the anode side of the magnetron device, and the second inverter device is provided. The second high-frequency voltage converted by the device is transformed using a second transformer, and the transformed predetermined heater voltage is supplied to the filament of the magnetron device. Therefore, only a small and lightweight high-frequency transformer is used. This can contribute to the reduction in size and weight of the magnetron driving device.

Further, in this second inverter device, the power control means starts the conduction of the switching means according to the voltage state of the primary winding of the second transformer and the conduction time of the switching means according to the voltage fluctuation of the commercial power supply. Is controlled, it is possible to supply a stable heater voltage to the filament, which is not affected by fluctuations in the power supplied to the anode of the magnetron device or the fluctuation of the commercial power supply voltage, and to operate the magnetron device stably.

Further, since the heater voltage for the filament is stabilized, the life of the filament can be improved.

[Brief description of the drawings]

FIG. 1 is a circuit diagram showing an overall configuration of a magnetron driving device to which the present invention is applied, FIG. 2 is a circuit diagram showing a main part of FIG. 1, FIG. 3, FIG. FIG. 6 and FIG.
FIG. 7 is a characteristic waveform diagram of the heater voltage Pf supplied to the filament F shown in FIGS. 1 and 2; FIG. 8 is a diagram showing the voltage detection circuit shown in FIG. FIG. 9 is a circuit diagram showing a conventional example, and FIG.
FIG. 11 is an explanatory diagram showing the characteristics of the conventional example shown in FIG.
FIG. 12 is a circuit diagram showing another conventional example, and FIG. 12 is an explanatory diagram showing characteristics of the conventional example shown in FIG. 12 Magnetron 25 Voltage detection circuit 27 Current detection circuit 29 Power control circuit Q1 Transistor F Filament 23 Transformer

Claims (1)

(57) [Claims] A first inverter for converting commercial power to a first high-frequency voltage; a first transformer for transforming the first high-frequency voltage and supplying the first high-frequency voltage to an anode side of a magnetron device; A second inverter device for converting the second high-frequency voltage into a second high-frequency voltage by conducting / non-conducting control of a switching means, and supplying the transformed predetermined heater voltage to a filament of the magnetron device A magnetron driving device having a second transformer that performs a voltage detection operation for detecting a voltage change of the commercial power supply; and a conduction current on the primary side of the second transformer that is controlled by conduction / non-conduction control of the switching unit. A winding provided separately from the controlled primary winding; and a winding provided on the second inverter device, the primary winding being induced in the winding. Power control means for starting conduction of the switching means in accordance with the pressure state and controlling conduction time of the switching means in accordance with voltage fluctuations of the commercial power supply detected by voltage detection means. Drive.