JP2965503B2 - Array antenna control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a control device for an array antenna.

[0002]

2. Description of the Related Art Conventionally, a phased array antenna for satellite communication (hereinafter referred to as a first conventional example) mounted on a vehicle or the like and automatically tracking the direction of a geostationary satellite has been prototyped by the Communications Research Laboratory of the Ministry of Posts and Telecommunications. ing. The phased array antenna of the first conventional example is composed of 19 microstrip antenna elements, and one element is provided for each element except one element.
Eight microwave phase shifters are provided to electrically scan the direction of the transmission beam without mechanical drive. Here, as a sensor for controlling the directivity of the antenna and tracking the direction of the incoming beam, a magnetic sensor for detecting the direction of geomagnetism and calculating the direction of the geostationary satellite seen from the vehicle that is known in advance,
Further, an optical fiber gyro is provided for detecting the rotational angular velocity of the vehicle and accurately keeping the beam direction constant. By combining these two sensors, regardless of the presence or absence of an incoming beam, the antenna directs the transmission directivity in a certain direction, and keeps the same directivity in the same direction even when the vehicle moves. Have been. In addition, by providing the microwave phase shifter with frequency characteristics, directivity is formed in the same direction in both transmission and reception even when the transmission and reception frequencies are different.

[0003]

However, the phased array antenna of the first prior art can direct a transmission beam in the direction of a signal source if the direction of the signal source is known regardless of the presence or absence of an incoming signal. However, unless the direction of the signal source is unknown or the signal source itself moves such as a low orbit satellite, unless the movement is all predictable,
Tracking is impossible. In addition, even when the direction of the signal source is known, the magnetic sensor can know the absolute azimuth, but is easily affected by the magnetic field due to the surrounding metal, and the optical fiber gyro is affected by the magnetic field due to the surrounding metal. No,
Since the angular velocities are detected and integrated to determine the absolute azimuth, errors are easily accumulated. The method of performing accurate tracking while complementing each other by combining these features has a problem that the configuration is complicated and the performance is limited as described above.

In order to solve the above-mentioned problems, as a method of forming a transmission beam in an arrival signal direction without using a special sensor for knowing the direction of an arrival wave, a reception phase difference can be used as it is or a transmission beam can be transmitted. A method of converting this into a transmission phase difference in accordance with the reception frequency (hereinafter, referred to as a second conventional example) has been disclosed by the present applicant in Japanese Patent Application No. 7-1.
No. 17167 is proposed in the patent application. Second
The method of the prior art calculates a reception phase difference between each two antenna elements close to each other, and forms a plurality of equiphase first-order regression planes corresponding to all candidates of phase uncertainty which the reception phase difference has. The calculation is performed using the least squares method, and furthermore, the reception phase difference is corrected by specifying only one primary regression plane corresponding to the maximum reception wave, and the maximum reception wave is corrected based on the corrected reception phase difference. The transmission beam is formed only in the direction of. In the second conventional example, a single transmission main beam can be formed only in the direction of the maximum received wave, even in an environment where a plurality of multiplexed waves arrive, or when a phase uncertainty occurs in a received phase difference. It has an excellent feature.

However, in the case of the transmission beam forming method of the second prior art, when calculating a transmission phase difference between a predetermined reference antenna element and each antenna element, a plurality of equal phase 1s are calculated by the least squares method. It is necessary to calculate the next regression plane and specify the plane by conditional branching, and there is a problem that the calculation is complicated.

SUMMARY OF THE INVENTION An object of the present invention is to solve the above problems, and to achieve maximum reception without using an azimuth sensor or the like, and even in an environment where a plurality of multiplexed waves arrive, or in a case where the phase difference is uncertain in the reception phase difference. An object of the present invention is to provide an array antenna control device that can form a single transmission main beam only in the direction of a wave and that is simple in operation.

[0007]

According to a first aspect of the present invention, there is provided an array antenna control apparatus for controlling an array antenna including a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape. A converter for converting a plurality of received signals respectively received by the antenna elements of the array antenna into two orthogonal baseband signals orthogonal to each other using a common local oscillation signal; A receiving signal received by a predetermined reference antenna element of the plurality of antenna elements based on each of the two orthogonal baseband signals converted by the conversion means, and an arbitrary antenna of the plurality of antenna elements; Calculates the reception weight corresponding to the given antenna element, which is the complex conjugate product of the received signal received by the element and the conjugate complex number A reception weight calculating unit that, the reception weights corresponding to the antenna elements adjacent to the antenna elements of the reference, which is calculated by the reception weight calculating means,
Based on a first complex conjugate product, which is a complex conjugate product of a reception weight corresponding to the reference antenna element calculated by the reception weight calculation means and a conjugate complex number of the reception weight, an antenna element adjacent to the reference antenna element First phase difference calculating means for calculating a first phase difference between a received signal received and a received signal received by the reference antenna element; and each reception weight calculated by the reception weight calculating means Of the two corresponding to the antenna elements adjacent to each other
A second complex conjugate product, which is a complex conjugate product of one of the two reception weights and a conjugate complex number of the other reception weight, based on the two reception weights; Complex conjugate product calculating means for calculating a third complex conjugate product which is a complex conjugate product of the complex conjugate product and the conjugate complex number of the first complex conjugate product; and a third conjugate product calculated by the complex conjugate product calculating means. A second phase difference for calculating a second phase difference, which is a difference between the phase difference between the received signals received by the antenna elements adjacent to each other and the first phase difference, based on the complex conjugate product of Arbitrary antenna element based on arithmetic means, based on the first phase difference calculated by the first phase difference calculating means and the second phase difference calculated by the second phase difference calculating means Received by the reference antenna element Receiving phase difference calculating means for calculating a receiving phase difference between the received signal and the received phase difference calculated by the receiving phase difference calculating means; Using the least squares method, the phase difference of an arbitrary antenna element with respect to the antenna element is compared with the maximum received wave having the largest power among the received waves received by the plurality of antenna elements. A regression plane calculating means for calculating the primary regression plane by regressing to a primary regression plane of a phase; and any two adjacent antenna elements of the primary regression plane calculated by the regression plane calculation means. Transmission phase difference calculation means for calculating a transmission phase difference by multiplying the phase difference by a ratio of a predetermined transmission frequency to a predetermined reception frequency, wherein each of the two antennas calculated by the transmission phase difference calculation means is provided. In transmission phase difference between the antenna elements, by transmitting the transmission signals from the respective antenna elements, and forming a transmission main beam in the direction of the maximum reception wave.

According to a second aspect of the present invention, there is provided an array antenna control apparatus for controlling an array antenna including a plurality of antenna elements juxtaposed at equal intervals. A conversion unit that converts a plurality of reception signals received by each of the antenna elements of the array antenna into two orthogonal baseband signals orthogonal to each other using a common local oscillation signal; A plurality of beams corresponding to a plurality of predetermined beam directions are formed using a predetermined beam forming method based on each of the two orthogonal baseband signals, and a plurality of beams corresponding to the plurality of formed beams are formed. A beam forming means for generating a reception signal; Beam selecting means for selecting a beam receiving signal having the following and outputting at least one beam receiving signal; a reference beam receiving signal among the beam receiving signals selected by the beam selecting means; and selecting by the beam selecting means Receiving beam weight calculating means for calculating a receiving beam weight corresponding to a given beam, which is a complex conjugate product of a given beam received signal and a conjugate complex number of the given beam received signal, and the receiving beam weight calculating means Based on the calculated reception beam weight, reception weight calculation means for calculating each reception weight corresponding to each antenna element, and among the reception weights calculated by the reception weight calculation means, among the plurality of antenna elements A receiving weight corresponding to an antenna element adjacent to the predetermined reference antenna element, A reception signal received by an antenna element adjacent to the reference antenna element based on a first complex conjugate product that is a complex conjugate product of a reception weight corresponding to the quasi antenna element and the reference antenna; A first phase difference calculating means for calculating a first phase difference between the received signal received by the element and a receiving weight calculated by the receiving weight calculating means, corresponding to antenna elements adjacent to each other; Based on the two reception weights, a second complex conjugate product that is a complex conjugate product of one of the two reception weights and a conjugate complex number of the other reception weight is calculated,
Complex arithmetic product operation means for calculating a third complex conjugate product that is a complex conjugate product of the second complex conjugate product and a conjugate complex number of the first complex conjugate product; A second phase difference which is a difference between a phase difference between reception signals received by mutually adjacent antenna elements and the first phase difference based on the third complex conjugate product obtained. 2 based on the first phase difference calculated by the first phase difference calculating means and the second phase difference calculated by the second phase difference calculating means. A reception signal received by the antenna element, a reception phase difference calculation means for calculating a reception phase difference between the reception signal received by the reference antenna element, and each of the reception phase differences calculated by the reception phase difference calculation means Based on the reception phase difference, corresponding to the arrangement of each antenna element and Using the least squares method, the phase difference of any antenna element with respect to the reference antenna element is compared with the maximum received wave having the maximum power among the received waves received by the plurality of antenna elements. 1 in phase
A regression plane calculation means for calculating the primary regression plane by regressing to the next regression plane; and a phase difference between any two adjacent antenna elements of the primary regression plane calculated by the regression plane calculation means. Transmission phase difference calculating means for calculating a transmission phase difference by multiplying by a ratio of a predetermined transmission frequency to a predetermined reception frequency, and the transmission between the two antenna elements calculated by the transmission phase difference calculation means. A transmission main beam is formed in the direction of the maximum received wave by transmitting a transmission signal from each of the antenna elements with a phase difference.

The control device for an array antenna according to a third aspect of the present invention is the control device for an array antenna according to the first or second aspect, further comprising: Phase correction means for converting a phase difference between any two antenna elements into a phase difference having a value in a range from -π to + π, wherein the transmission phase difference calculation means is converted by the phase correction means. A transmission phase difference is calculated by multiplying a phase difference by a ratio of the transmission frequency to the reception frequency.

[0010]

Embodiments of the present invention will be described below with reference to the drawings. <First Embodiment> FIG. 1 is a block diagram of an automatic beam acquisition and tracking device for a communication array antenna according to a first embodiment of the present invention. The automatic beam capture and tracking device of the communication array antenna according to the first embodiment is 、 of the wavelength of the received signal, の of the wavelength of the transmitted signal, or the average of the wavelength of the received signal and the wavelength of the transmitted signal. The directivity of the array antenna 1 including a plurality of N antenna elements A1, A2,..., Ak,... Aim at high speed in the direction of the incoming beam of wireless signal waves such as unmodulated waves,
Perform that tracking. Here, in particular, the acquisition and tracking device of the first embodiment is characterized by including the maximum ratio combining circuit 4 and the transmission weight calculating circuit 30. Then, even when the azimuth of the partner station serving as the signal source is unknown, the arriving signal based on the baseband signal for each antenna element obtained from the arriving wave transmitted from the signal source without using an azimuth sensor or the like. A transmission beam is formed in the direction of the wave, and even in an environment where a plurality of multiplexed waves arrive, or in a case where the phase difference is uncertain in the reception phase difference, multiplexing can be performed simply by calculating one primary regression plane. Wave effects and phase uncertainties can be eliminated, forming a single transmit main beam only in the direction of the largest received wave.

As shown in FIG. 1, the array antenna 1
It includes N antenna elements A1 to AN and circulators CI-1 to CI-N as transmission / reception separators. The receiving modules RM-1 to RM-N are respectively
Using the low-noise amplifier 2 and the common first local oscillation signal output from the first local oscillator 11, a down-converter converts a received radio signal having a radio frequency into an intermediate frequency signal having a predetermined intermediate frequency. Converter (D / C) 3
And

The receiving unit of the acquisition and tracking device further uses N A / D converters AD-1 to AD-N and a common second local oscillation signal output from the second local oscillator 12, The intermediate frequency signal after the A / D conversion is quasi-synchronously detected and two baseband signals orthogonal to each other (hereinafter, these 2
One baseband signal is called a quadrature baseband signal. N) quasi-synchronous detection circuits QD-1 through QD
D-N and said converted reception weight W ₁ ^RX for each quadrature baseband signal as maximum ratio combining based on the quadrature baseband signal, W ₂ ^RX, ..., and calculates the W _N ^RX, each reception weight W ₁ ^RX computed relative quadrature baseband signal, W ₂ ^RX, ..., W _N and a maximum ratio combining circuit 4 ^{which RX} and by phase combination was multiplied outputs to the demodulator 5, the maximum ratio combining circuit 4 Weights W ₁ ^RX and W ₂ calculated by
^RX, ..., based on W _N ^RX, transmission weight by the method according to the present invention ^{_{W 1 TX, W 2 TX,}} ..., phase-by computing the W _N ^TX
Synchronous detection or delay detection is performed by a predetermined baseband demodulation process from the transmission weight calculation circuit 30 output to the amplitude correction units 13-1 to 13-N and the baseband signal output from the maximum ratio combining circuit 4 to obtain a desired signal. A demodulator 5 for extracting digital data and outputting it as received data.

In the receiving section, the antenna elements A1 to AN in the array antenna 1 to the maximum ratio combining circuit 4 are cascaded for each antenna element system. Since the signal processing for each system of each antenna element in the receiving unit is executed in the same manner, the radio signal received by the antenna element Ak (Ak is represented as one of the antenna elements A1 to AN). The processing for waves will be described.

The radio signal wave received by the antenna element Ak is transmitted to the circulator CI-k and the receiving module RM-
The signal is input to the down converter 3 via the low noise amplifier 2 within k. The down converter 3 in the receiving module RM-k is connected to the common first
Using the local oscillation signal, the input radio signal is frequency-converted into an intermediate frequency signal having a predetermined intermediate frequency.
The signal is output to the quasi-synchronous detection circuit QD-k via the D converter AD-k. The quasi-synchronous detection circuit QD-k is connected to the second local oscillator 1
2 is quasi-synchronous detected using the common second local oscillation signal output from the A / D-converted intermediate frequency signal and is converted into two orthogonal baseband signals I _k , Q _k. Output to the ratio combining circuit 4.

The maximum ratio combining circuit 4 receives the weights W ₁ ^RX ,
^{_{W 2 RX, ..., W N}} RX a calculates the reception weight W ₁ ^RX computed with respect to the respective quadrature baseband signal, W ₂ ^RX, ...,
After multiplying by W _N ^RX , in-phase synthesis is performed and output to the demodulator 5.
Further, the transmission weight operation circuit 30 receives the reception weights W ₁ ^RX , W ₂ ^RX,.
Based on W _N ^RX, by the method according to the present invention, a transmission beam is formed in the direction of the incoming wave, and in the case in an environment plurality of multiple wave arrives, or the phase ambiguity in the reception phase difference is generated Also, the transmission weights W ₁ ^TX , W ₂ ^TX , W ₂ ^TX , W ₂ ^TX , W
, W _N ^TX are calculated and the transmission weights W ₁ ^TX , W ₂ ^TX ,
, W _N ^TX are output to the phase / amplitude correction units 13-1 to 13-N, respectively. On the other hand, the demodulator 5 performs synchronous detection or delay detection from the baseband signal output from the maximum ratio combining circuit 4 by a predetermined baseband demodulation process, extracts desired digital data, and outputs it as received data. The circuit processing of the maximum ratio combining circuit 4 and the transmission weight calculation circuit 30 will be described later in detail.

Next, with reference to FIG. 1, the transmission section of the acquisition and tracking device will be described. The transmission unit includes N transmission modules TM-1 to TM-N, a first transmission local oscillator 14, a second transmission local oscillator 10, and N quadrature modulation circuits 6-1 to 6-N And the phase / amplitude correction unit 13-
1 to 13 -N and an in-phase distributor 9. here,
Each of the transmission modules TM-1 to TM-N mixes the input intermediate frequency signal and the first transmission local oscillation signal input from the first transmission local oscillator 14, and sets a predetermined transmission radio frequency. An up-converter (U / C) 7 for frequency-converting the transmission signal into a transmission signal and a transmission power amplifier 8 are provided. Here, based on the transmission weights W ₁ ^TX , W ₂ ^TX ,..., W _N ^TX input from the transmission weight calculation circuit 30, the phase / amplitude correction unit 13-k calculates the respective phases and amplitudes corresponding thereto. a way to correct the phase and amplitude of each baseband transmission signal S ^TX, and outputs the baseband transmit signal S _k ^TX corrected to quadrature modulation circuit 6-k.

The transmission baseband signal S which is transmission data
^{The TX} is input to the in-phase distributor 9 and then distributed in-phase. The distributed transmission baseband signals S ^TX are transmitted by the respective phase / amplitude correction units 13-1 to 13-N to transmit weights W _1.
^The amplitude and phase are corrected so as to have respective amplitudes and phases corresponding to ^TX , W ₂ ^TX ,..., W _N ^TX , and the corrected baseband transmission signal S _k ^TX is input to the quadrature modulation circuit 6-k. Is done. The quadrature modulation circuit 6-k includes a second transmission local oscillator 10
Is converted to the transmission baseband signal S _k ^TX input from the phase / amplitude correction unit 13-k.
After performing quadrature modulation such as QPSK, the intermediate frequency signal after quadrature modulation is transmitted as a transmission radio signal via the up-converter 7 and transmission power amplifier 8 in the transmission module TM-k, and Input to the circulator CI-k. Here, the quadrature modulator 6-k converts the input transmission baseband signal S _k ^TX into a transmission quadrature baseband signal by serial / parallel conversion,
The intermediate frequency signal is obtained by orthogonally modulating and combining second transmission local oscillation signals having a phase difference of 90 ° with each other according to the transmission quadrature baseband signal. And
The transmission radio signal is transmitted and radiated from the antenna element A1. Therefore, the transmission signals weighted by the transmission weights W ₁ ^TX , W ₂ ^TX ,..., W _N ^TX are radiated from the antenna elements A1 to AN. In the first embodiment, the transmission signals transmitted from the respective antenna elements Ak are weighted by transmission weights W ₁ ^TX , W ₂ ^TX ,..., W _N ^TX as described in detail later, and It is transmitted at a predetermined amplitude.

In the first embodiment, for example, N = 1
Six antenna elements A1 to A16 are juxtaposed on a straight line at equal intervals. As described above, the interval is set to a half wavelength of the transmission signal, a half wavelength of the reception signal, or a half wavelength of an average value thereof. The antenna elements A1 to AN are, for example, circular patch microstrip antennas.

FIG. 2 is a block diagram showing signal processing in the maximum ratio combining circuit 4. In the signal processing in the maximum ratio combining circuit 4 according to the first embodiment, I / O conversion and quasi-synchronous detection are performed for each of the antenna elements A1 to AN.
This is performed on a quadrature baseband signal composed of a component and a Q component. Here, the number of antenna elements of the array antenna 1 is N
Then, an antenna element Ar serving as a phase reference and an arbitrary antenna element Ak (1 ≦ r) including the antenna element Ar
≤N, 1≤k≤N) baseband signals S _r , S _k
Are expressed as complex numbers, respectively. Here, it referred to as the reference base band signal of the base band signal S _r, the base band signal S _k that the processing baseband signal. The antenna element serving as a phase reference (hereinafter referred to as antenna element Ar) is a predetermined one of the N antenna elements. The antenna element receiving the processed baseband signal S _k as processing antenna element Ak.

[0020]

S _r = I _r + j · Q _r = √ (I _r ² + Q _r ² ) exp (jφ ₀ )

S _k = I _k + j · Q _k = {(I _k ² + Q _k ² ) exp {j (φ ₀ −θ _k )}

Here, φ ₀ is the phase of the baseband signal of the reference antenna element Ar, and θ _k is the reference antenna element A.
This is the phase difference between the received signal of r and the received signal of antenna element Ak. The reception weight W _k ^RX for each antenna element Ak is based on each two orthogonal baseband signals,
And reference baseband signal S _r received by the antenna elements Ar criteria can be represented by the complex conjugate product of the complex conjugate of the baseband signal processing S _k received by the antenna element Ak. That is, the reception weight W _k ^RX for each antenna element Ak can be expressed as in the following Expression 3.

[0022]

W _k ^RX = S _r · S _k * = | S _r || S _k | exp [j {φ ₀ − (φ ₀ −θ _k )}] = | S _r || S _k | exp ( jθ _k )

Here, | · | in Equation 3 represents the amplitude of the signal, and * represents a conjugate complex number. Number 3
As is clear from the above, θ _k is the phase difference between the received signal of the reference antenna element Ar and the received signal of each antenna element Ak, and is the phase of the received weight W _k ^RX .
The reception weight W _k ^RX for each antenna element Ak given by Equation 3 is multiplied by the processed baseband signal S _k received by the antenna element Ak. Thus, the baseband signal processing S _k is in phase with the reference antenna element Ar reference baseband signal S _r received by.
This is calculated for all the antenna elements A1 to AN, and their sum (W ₁ ^RX S ₁ + W ₂ ^RX S ₂ +... + W
_N ^RX S _N ). That is, a baseband signal processing S _k, processing the baseband signal S _k of a size proportional to the amplitude, and the reference antenna element reception weight W _k to phase with the phase of the reference baseband signal S _r of Ar ^RX is multiplied, and the result of the multiplication is synthesized. In this way, the maximum ratio combining by the element space without the feedback loop is realized. In actual communication, in order to prevent the beam from becoming unstable due to the influence of noise or the like, the real part and the imaginary part of the reception weight W _k ^RX expressed by Expression 3 are converted to a narrow band low-pass filter 42. Use low-frequency filtering at -k.

Further, the calculated sum (W ₁ ^RX S ₁ +
W ₂ ^RX S ₂ +... + W _N ^RX S _N ) to all antenna elements A1
2 for the size of each reception weight W _k ^RX for AN
Th power of the sum ^{_{(│W 1 RX │ 2 + │W}} 2 RX │ 2 + ... + │W N RX │ 2)
Square root ^{_{{√ (│W 1 RX │ 2}} + │W 2 RX │ 2 + ... + │W N RX
| ² ) By dividing by｝, a standardized maximum ratio combined output signal S ^RX is obtained. That is, the standardized maximum ratio combined output signal S ^RX is expressed as in the following ^Expression 4.

[0025]

(Equation 4)

The standardized maximum ratio combined output signal S ^RX
Is demodulated by the demodulator 5.

The configuration and operation of the maximum ratio combining circuit 4 for executing the above described operation will be described with reference to FIG. In the following description, the reference baseband signal _Sr is the reference baseband signal Sr received by the antenna element A1.
_Set to ₁ . In maximum ratio combining circuit 4, the reference baseband signals S ₁ via the two input terminals of the complex conjugate product calculating unit 41-1 is input to the complex conjugate product calculating unit 41-1, and the complex conjugate product calculating section 41 -2 to 41-N. Processed baseband signal S _k (k = 2, 3,..., N)
Is input to the complex conjugate product operation unit 41-k, and is input to the multiplier 44-k via the delay circuit 43-k.

The complex conjugate product calculating unit 41-1, based on the two reference baseband signals S ₁ inputted, which is the product of the complex conjugate of the reference baseband signals S ₁ and the reference baseband signals S ₁ complex The conjugate product S ₁ · S ₁ * is calculated, and the reception weight W ₁ ^RX , which is the complex conjugate product S ₁ · S ₁ *, is calculated by the multiplier 44-1 via the low-pass filter 42-1. The signal is output to the section 46-1 and the transmission weight calculation circuit 30. Here, the low-pass filter 42-1 is configured by a digital filter such as an FIR filter or an IIR filter, and allows a signal having a frequency lower than a cutoff frequency to pass therethrough. It is possible to prevent an error in the weight of the maximum ratio combination from increasing in accordance with an in-phase error due to a component or band limitation or an amplitude fluctuation. Further, the delay circuit 43-1 calculates the operation time and the low-pass filter 4 in the complex conjugate product operation unit 41-1.
2-1 in consideration of by delay, so that the delay times of the two signals are equal input to the multiplier 44-1, delays the reference baseband signals S ₁ inputted, the multiplier 44
Output to -1. The multiplier 44-1 multiplies the received reception weight W ₁ ^RX and the reference baseband signal S ₁ by
The multiplication result W ₁ ^RX S ₁ is output to the adder 45.

The complex conjugate product operation unit 41-2 calculates a complex operation product S ₁ · S ₂ * which is a product of the input reference baseband signal S ₁ and the complex conjugate of the processed base band signal S _2. , The complex operation product S ₁ · S ₂ *, the reception weight W ₂ ^RX is passed through the low-pass filter 42-2 to the multiplier 44-2, the complex conjugate product operation unit 46-2, and the transmission weight operation circuit 30. Output. Here, the low-pass filter 42-2 is, like the low-pass filter 42-1, an FIR filter or an IIR filter.
It is composed of a digital filter such as an IR filter, and passes a signal having a frequency lower than a cutoff frequency, whereby a receiver noise, a modulation component,
It is possible to prevent the weight error of the maximum ratio combination from increasing due to the in-phase error due to the band limitation or the amplitude fluctuation. Further, the delay circuit 43-2 includes a complex conjugate product operation unit 41.
In consideration of the operation time at −2 and the delay caused by the low-pass filter 42-2, the processing baseband signal S ₂ input so that the delay times of the two signals input to the multiplier 44-2 become equal. Is output to the multiplier 44-2. The multiplier 44-2 outputs the received reception weight W
₂ ^RX is multiplied by the processing baseband signal S _2, and the multiplication result W ₂ ^RX S ₂ is output to the adder 45.

The complex conjugate product operation unit 41-k (k = 3, 4,
.., N) calculates a complex operation product S ₁ · S _k *, which is a product of the input reference baseband signal S ₁ and the complex conjugate of the processed base band signal S _k , to obtain a complex operation product S ₁ · S The reception weight W _k ^RX , which is S _k *, is output to the multiplier 44-k, the complex conjugate product operation unit 46-k, and the transmission weight operation circuit 30 via the low-pass filter 42-k. Here, the low-pass filter 42-k includes the low-pass filters 42-1, 42.
Like -2, it is composed of a digital filter such as an FIR filter or an IIR filter, and passes a signal having a frequency lower than the cut-off frequency. It is possible to prevent the error of the reception weight in the maximum ratio combining from becoming large according to the error of the conversion and the amplitude fluctuation. Also,
The delay circuit 43-k takes into account the operation time in the complex conjugate product operation unit 41-k and the delay caused by the low-pass filter 42-k, and the delay times of the two signals input to the multiplier 44-k are equal. Thus, the input processing baseband signal _Sk is delayed and output to the multiplier 44-k. The multiplier 44-k multiplies the input reception weight W _k ^RX by the processing baseband signal S _k and obtains a multiplication result W _k ^RX S _k
Is output to the adder 45.

The adder 45 adds the input N multiplication results W ₁ ^RX S _{1 to} W _N ^RX _SN and adds the result (W ₁ ^RX S ₁ +
^{_{W 2 RX S 2 + ... +}} W N RX S N) to output. The complex conjugate product operation unit 46-k (k = 1, 2,..., N) calculates the product of the reception weight W _k ^RX and the complex conjugate of the reception weight W _k ^RX based on the input reception weight W _k ^RX. and it calculates the receiving weight 2 multiplied │W _k ^RX │ ² to the adder 47 is. The adder 47 receives N received weight squared products | W _k ^RX
│ ² by adding the addition result ^{_{(│W 1 RX │ 2 + │W}} 2 RX │ 2 +
... + │W _N ^RX │ ² ) are output to the square root calculator 48. Square-root calculation unit 48 adds the input result (│W ₁ ^RX │ ² + │
Square root of W ₂ ^RX │ ² + ... + │W _N ^RX │ ² ) ｛√ (│W ₁ ^RX │ ^{2
_{+ │W 2 RX │ 2 + ...}} + │W N RX │ 2)} calculates and outputs to the divider 49. The divider 49 calculates the input addition result (W ₁ ^RX S ₁ + W ₂ ^RX S ₂ +... + W _N ^RX S _N ) by the square root
^{_{(│W 1 RX │ 2 + │W}} 2 RX │ 2 + ... + │W N RX │ 2) is divided by}, and outputs a maximum ratio combined output signal S ^RX is a division result to the demodulator 5.

Next, a method of forming a transmission beam by calculating a transmission weight W _k ^TX using the reception weight W _k ^RX for each antenna element Ak calculated by the maximum ratio combining circuit 4 will be described. For example, TDD (Time
When the frequency of the transmission signal is equal to the frequency of the reception signal as in the case of the Division Duplex method or the like, the reception weight for combining the reception signals received by the antenna element Ak at the maximum ratio is transmitted by using the reception weight as it is. In addition, a transmission beam is formed in the reception wave having the maximum reception power and in the multipath direction, thereby forming a diversity transmission system. However, when the frequency of the transmission signal is different from the frequency of the reception signal, the phase relationship between paths differs between transmission and reception, so that transmission in the multipath direction needs to be suppressed. Therefore, in the first embodiment, based on each reception weight W _k ^RX for the array antenna 1, a transmission weight W _k ^TX is calculated as follows, and the reception weight W _k ^TX is calculated in the direction of arrival of the reception wave having the maximum reception power. The main beam of a single transmission signal is formed.

First, the phase of the reception weight W _k ^RX for each antenna element Ak will be considered. Phase difference theta _k - [theta] ₁ between the reception weight W ₁ ^RX reception weight W _k ^RX and the reference antenna element A1 for each antenna element Ak is generally
It can be obtained by calculating the sum of the phase differences △ θ _i (= θ _{i + 1} −θ _i ) of the two reception weights between the mutually adjacent antenna elements A (i + 1) and Ai.
It is represented as

[0034]

(Equation 5)

Here, arg ｛｝ is a complex number W in ｛｝
_It represents the angle of _{i + 1} ^RX · (W _i ^RX ) * and is indicated in the range from −π to π. Thus, the influence of multipath, adjacent antenna elements A (i + 1), the phase difference △ theta _i between Ai is or in the case of _{△ θ i> + π △ θ} i <-π
In the case of, the phase difference Δθ _i is −
Since it is calculated as a value in the range from π to π, the phase difference θ _k −
In some cases, θ ₁ cannot be determined accurately. Therefore, in order to prevent this, in the first embodiment, the phase difference Δθ ₁ between the reference antenna element A1 and the adjacent antenna element A2 adjacent to the reference antenna element A1 (hereinafter, referred to as a reference phase difference). And (に 従 っ_て θ _i −
△ theta ₁₎ the sum calculated of, was to seek theta _k - [theta] ₁ by adding with the sum of the _{(k-1) △ θ 1} .

[0036]

(Equation 6)

That is, in the array antenna 1 of the first embodiment, since N antenna elements Ak are arranged at equal intervals, the adjacent antenna elements A (i + 1),
Phase difference △ theta _i between the reception weight ^{_{W i + 1 RX, W i}} RX respectively corresponding to Ai, even influenced by multipath, considered significantly different from and without reference phase difference △ theta ₁ Can be Therefore, (△ θ _i − △ θ ₁ ) in Equation 6 is +
It is considered that π> (△ θ _i − △ θ ₁ )> − π is always satisfied. Accordingly, by using Equation 6, the phase difference θ _k −θ ₁ of the reception weight between each antenna element Ak and the reference antenna element A1 can be calculated during the calculation for calculating the sum by −
With π or π, it is obtained uniquely without so-called aliasing. Therefore, in the first embodiment, the phase difference θ _k − of the reception weight W _k ^{RX with} respect to each antenna element Ak.
It was configured to calculate θ ₁ using Equation 6.

That is, in the first embodiment, (1) the reception weight W ₂ ^RX corresponding to the antenna element A2 adjacent to the reference antenna element A1 and the reception weight W ₁ ^RX corresponding to the reference antenna element A1 Based on the complex conjugate product {W ₂ ^RX · (W ₁ ^RX ) *} with the conjugate complex number, the processing baseband signal S ₂ received by the antenna element A2 adjacent to the reference antenna element A1 and the reference antenna The reference phase difference Δθ ₁ between the reference baseband signal S ₁ and the reference baseband signal S ₁ received by the element A1 is calculated. (2) Next, among the reception weight W _i ^RX, two reception weight W _i ^RX corresponding to the antenna elements adjacent to each other, based on W i _{+ 1} ^RX, the two receiving weight W
_i ^RX, and the reception weight W _{i + 1} ^RX one of the W i _{+ 1} ^RX,
Complex conjugate product with conjugate complex number of the other reception weight W _i ^RX
Calculate {W _{i + 1} ^RX · (W _i ^RX ) *}. (3) And the complex conjugate product {W _{i + 1} ^RX · (W _i ^RX ) *},
Complex conjugate product of the complex conjugate product {W ₂ ^RX · (W ₁ ^RX ) *} with the conjugate complex number [{ _{Wi + 1} ^RX · ( _Wi ^RX ) *} · {W ₂ ^RX · (W ₁ ^RX ) * } *] To calculate the complex conjugate product [{W _{i + 1} ^RX · (W _i ^RX ) *} · {W ₂ ^RX · (W ₁
^RX ) *} *], the antenna elements A adjacent to each other
i, the processed baseband signal S received at A (i + 1)
The difference (△ θ _i- △ θ ₁ ) between the phase difference Δθ _i between _i and S _{i + 1} and the reference phase difference Δθ ₁ is calculated. (4) Next, based on the reference phase difference Δθ ₁ and the difference (△ θ _i- △ θ ₁ ), the processing baseband signal S _k received by the antenna element Ak and the reception base band signal S _k received by the reference antenna element A 1 the phase difference between the reference baseband signals S ₁ and (theta _k -
θ ₁ ).

The thus obtained phase difference θ _k -θ ₁ =
When a multipath wave is incident, the phase distribution of δθ _k generally does not become a straight line as shown in FIG. Therefore, in order to form a main beam of a single transmission signal only in the direction of arrival of the received wave having the maximum received power, the phase distribution is returned to a straight line by the least squares method. That is, as shown in FIG. 5, an array antenna 1 composed of four antenna elements A1 to A4 will be described. Least square regression corresponding to the arrangement of each antenna element Ak and located on one straight line on the phase plane Phase difference δθ ₁
^LSR , δθ ₂ ^LSR , δθ ₃ ^LSR , δθ ₄ ^LSR are ^calculated by sum of squares δθ ² =
(Δθ ₁ −δθ ₁ ^LSR ) ² + (δθ ₂ −δθ ₂ ^LSR ) ² + (δθ
_The calculation is performed so that ₃ −δθ ₃ ^LSR ) ² + (δθ ₄ −δθ ₄ ^LSR ) ² becomes minimum. As a result, each phase difference θ _k −θ ₁ = δθ _k of the antenna element Ak is regressed to a first-order regression plane having the same phase with respect to the maximum reception wave received by each antenna element Ak to calculate a first-order regression plane. Then, the phase difference Δθ after the least squares regression between adjacent antenna elements from the primary regression plane
Calculate ^LSR (= δθ _{i + 1} ^LSR −δθ _i ^LSR ). here,
When a direct wave arrives, the direct wave generally becomes the maximum received wave. Therefore, in FIG. 5, a direct wave is used as the maximum received wave. However, in an environment where a direct wave does not arrive, a maximum received wave having the highest power among received signals is used instead of the direct wave. In FIG. 5, the array antenna 1 including the four antenna elements A1 to A4 has been described.
The same applies to four or more cases.

In this case, if the reference phase difference Δθ ₁ itself has been turned back at −π or π due to the influence of multipath or the like, the adjacent antenna calculated from the straight line regressed by the least square method is used. The phase difference △ θ ^LSR (= δθ _{i + 1} ^LSR −δθ _i ^LSR ) after the least-squares regression between the elements is also a value obtained by folding the phase. Therefore, the direction of arrival of the maximum received wave is considered to be within the range from -90 ° to 90 °, and the direction of the transmission beam is also obtained by the above regression so as to be within the range from -90 ° to 90 °. Minimum 2
A phase correction process for converting the phase difference Δθ ^LSR after the power regression into a range of −π <Δθ ≦ π is executed.

That is, as shown in the flowchart of FIG. 4, the phase difference Δθ ^LSR after the least squares regression in step S1.
It is determined whether or not has been input. If it has been input, the process proceeds to step S2, and if it has not been input, step S1 is repeated. In step S2, -π <Δθ ^LSR ≦
It is determined whether or not π. If −π <Δθ ^LSR ≦ π, the process proceeds to step S6. If −π <Δ ^LSR θ ≦ π, the process proceeds to step S3. In step S3, π <
It is determined whether the [Delta] [theta] ^LSR, when a [pi <[Delta] [theta] ^LSR proceeds to step S4, and when not, [pi <[Delta] [theta] ^LSR proceeds to step S5. In step S4, (Δθ ^LSR
-2π) is substituted for Δθa, and the process proceeds to step S7. In step S5, (Δθ ^LSR + 2π) is substituted for Δθa, and the process proceeds to step S7. In step S6, Δθ ^{LSR is set} to Δ
Substitute for θa and proceed to step S7. In step S7,
Δθa is output to a multiplier 303 described later.

The transmission weight W _k ^TX for each antenna element Ak for forming a transmission beam is the detected phase difference Δθ between adjacent antenna elements obtained by the above calculation.
It is given by the following equation 7 using a.

[0043]

W _k ^TX = _ak exp [j _k (f _T / f _R ) · (k
-1) {θa}]

Here, a _k represents an arbitrary excitation distribution, f _T represents a transmission frequency of a transmission signal, and f _R represents a reception frequency of a reception signal. The phase of the transmission weight W _k ^TX given by Expression 7 is the reference antenna element A1. However, the present invention is not limited to this, and may be at any position, for example, at the center of the array antenna 1.

In actual communication, the baseband signal distributed in phase by the phase / amplitude correction unit 13-k is phase corrected based on the transmission weight W _k ^TX for each antenna element Ak obtained as described above. The quadrature modulator 6-k performs quadrature modulation using the second transmission local oscillation signal input from the common second transmission local oscillator 10, and the radio frequency, which is the transmission frequency, is transmitted by the transmission module TM-k. , And transmitted from each antenna element Ak via a circulator CI-k which is a transmission / reception separator.

The configuration and operation of the transmission weight operation circuit 30 that executes the above-described operation will be described with reference to FIG. The transmission weight calculation circuit 30 includes complex conjugate product calculation units 31-1 to 31- (N-1) and 32-1 to 32.
− (N−2) and the phase difference calculation units 33-1 to 33- (N
-1) and adders 34-1 to 34- (N-2), 35
-1 to 35- (N-2) and the least square regression processing unit 30
1; a phase correction unit 302; multipliers 303, 36-1 to 36-N; 38-1 to 38-N;
7-1 to 37-N.

In the transmission weight operation circuit 30, the reception weight W ₁ ^RX for the antenna element A1 is input to the complex conjugate product operation unit 31-1, and the reception weight W ₂ ^RX for the antenna element A2 is input to the complex conjugate product operation unit 31-1. -1 and the complex conjugate product operation unit 31-2. The reception weight W ₃ ^RX for the antenna element A3 is input to the complex conjugate product operation unit 31-2 and the complex conjugate product operation unit 31-3, and similarly received for the antenna element Ak (k = 4, 5,..., N). The weight W _k ^RX is calculated by the complex conjugate product operation unit 31- (k−
1) and the complex conjugate product operation unit 31-k.

The complex conjugate product calculating unit 31-1, the receiving weight W ₁ ^RX input, based on the W ₂ ^RX, and multiplies the reception weight W ₂ ^RX and complex conjugate of the received weight W ₁ ^RX, The complex conjugate product W ₂ ^RX W ₁ ^RX *, which is the result of the multiplication, is used to calculate the phase difference
3-1 and the complex conjugate product operation units 32-1 to 32- (N-
Output to 2). Based on the input complex conjugate product W ₂ ^RX W ₁ ^RX *, the phase difference calculation unit 33-1 calculates a reception weight W ₁ ^RX corresponding to the reference antenna element A1 and an antenna adjacent to the reference antenna element A1. and calculates the reference phase difference [Delta] [theta] ₁ between the reception weight W ₂ ^RX corresponding to elements A2, adders 34-1 to 34- (N-2) and the adder 35-1 and the least squares regression processing unit Output to 301.

The complex conjugate product calculator 31-2 multiplies the reception weight W ₃ ^RX and the complex conjugate of the reception weight W ₂ ^RX based on the input reception weights W ₂ ^RX and W ₃ ^RX . The complex conjugate product W ₃ ^RX W ₂ ^RX * that is the result of the multiplication is output to the complex conjugate product operation unit 32-1. The complex conjugate product operation unit 32-1
The input complex conjugate product W ₂ ^RX W ₁ ^RX * and complex conjugate product W ₃ ^RX W
₂ ^RX *, the complex conjugate product W ₃ ^RX W ₂ ^RX * is multiplied by the complex conjugate of the complex conjugate product W ₂ ^RX W ₁ ^RX *, and the complex conjugate product W ₃ ^RX W ₂ which is the result of the multiplication is obtained. ^RX * · (W ₂ ^RX W ₁ ^RX *) * is output to the phase difference calculator 33-2. The phase difference calculation unit 33-2 includes:
Input complex conjugate product W ₃ ^RX W ₂ ^RX * ・ (W ₂ ^RX W ₁ ^RX *) *
, The reception weight W ₂ ^RX corresponding to the antenna element A2 and the antenna element A3 adjacent to the antenna element A2.
And a phase difference Δθ ₂ between the receiving weight W ₃ ^RX corresponding to
The difference (Δθ ₂ −Δθ ₁ ) from the reference phase difference Δθ ₁ is calculated, and
Output to the adder 34-1.

The adder 34-1 adds the input reference phase difference Δθ ₁ and the difference (Δθ ₂ −Δθ ₁ ), and adds the addition result {Δθ ₁ + (Δθ ₂ −Δθ ₁ )}. 35-1. The adder 35-1 adds the reference phase difference Δθ ₁ and the phase difference {Δθ ₁ + (Δθ ₂ −Δθ ₁ )}, and adds the addition result {2Δθ ₁ + (Δθ ₂ −Δθ ₁ )} to the minimum value of 2 It outputs to the power regression processing unit 301 and the adder 35-2. Here, the addition result {2Δθ ₁ + (Δθ ₂ −Δθ ₁ )} is a phase of the reception weight W ₃ ^RX corresponding to the antenna element A3 and a reception weight W corresponding to the reference antenna element A1 expressed by the equation (6). ₁
Is the phase difference δθ ₃ between the ^RX of the phase.

The complex conjugate product operation unit 31-3 multiplies the reception weight W ₄ ^RX and the complex conjugate of the reception weight W ₃ ^RX based on the input reception weights W ₃ ^RX and W ₄ ^RX . The complex conjugate product W ₄ ^RX W ₃ ^RX * that is the result of the multiplication is output to the complex conjugate product calculation unit 32-2. The complex conjugate product operation unit 32-2
Complex conjugate product W ₂ ^RX W ₁ ^RX * and complex conjugate product W ₄ ^RX W
₃ ^RX *, the complex conjugate product W ₄ ^RX W ₃ ^RX * is multiplied by the complex conjugate of the complex conjugate product W ₂ ^RX W ₁ ^RX *, and the complex conjugate product W ₄ ^RX W ₃ which is the multiplication result is obtained. ^RX * · (W ₂ ^RX W ₁ ^RX *) * is output to the phase difference calculator 33-3. The phase difference calculation unit 33-3 includes:
Input complex conjugate product W ₄ ^RX W ₃ ^RX * ・ (W ₂ ^RX W ₁ ^RX *) *
, The reception weight W ₃ ^RX corresponding to the antenna element A3 and the antenna element A4 adjacent to the antenna element A3.
Phase difference Δθ ₃ between the receiving weight W ₄ ^RX corresponding to
The difference (Δθ ₃ −Δθ ₁ ) from the reference phase difference Δθ ₁ is calculated, and
Output to the adder 34-2.

The adder 34-2 adds the input reference phase difference Δθ ₁ and the difference (Δθ ₃ −Δθ ₁ ), and adds the addition result {Δθ ₁ + (Δθ ₃ −Δθ ₁ )}. 35-2. The adder 35-2 outputs the addition result {2Δθ ₁ + (Δθ ₂ −Δθ ₁ )} input from the adder 35-1, and the adder 3
4-2, the addition result ｛Δθ ₁ + (Δθ ₃ −Δθ
₁ )} and the addition result {3Δθ ₁ + (Δθ ₂ −Δ
θ ₁ ) + (Δθ ₃ −Δθ ₁ )｝ is converted to the least squares regression processing unit 30
1 and to the adder 35-3. Here, the addition result {3Δθ ₁ + (Δθ ₂ −Δθ ₁ ) + (Δθ ₃ −Δθ ₁ )}
Is the phase difference δθ4 between the phase of the reception weight W ₄ ^RX corresponding to the antenna element A4 and the phase of the reception weight W ₁ ^RT corresponding to the reference antenna element A1 represented by the equation ( ₆ ).

Similarly, the complex conjugate product operation unit 31-k (k =
, 5,..., N−1) are input reception weights W _k
^{Based on RX} and W _{k + 1} ^RX , the reception weight W _{k + 1} ^RX is multiplied by the complex conjugate of the reception weight W _k ^RX , and a complex conjugate product W _{k + 1} ^RX W _k ^RX * resulting from the multiplication is _obtained . From the complex conjugate product operation unit 32-
(K-1). Complex conjugate product operation unit 32- (k-
1), based on the complex conjugate product W ₂ ^RX W ₁ ^RX input * and a complex conjugate product ^{_{W k + 1 RX W k RX}} *, a complex conjugate product W _{k + 1} ^RX W _k
^RX * is multiplied by the complex conjugate of the complex conjugate product W ₂ ^RX W ₁ ^RX *,
Complex conjugate product W _{k + 1} ^RX W _k ^RX * · (W ₂ ^RX W ₁
^RX *) * is output to the phase difference calculator 33-k.

The phase difference calculator 33-k calculates the complex conjugate product W _{k + 1} ^RX W _k ^RX * · (W ₂ ^RX W ₁ ^RX *) *
The phase difference Δθ _k between the reception weight W _k ^RX corresponding to the antenna element Ak and the reception weight W _{k + 1} ^RX corresponding to the antenna element A (k + 1) adjacent to the antenna element Ak, the reference phase difference Δθ ₁ , The difference (Δθ _k −Δθ ₁ ) is calculated and output to the adder 34- (k−1). The adder 34- (k-
1) is the difference between the input reference phase difference Δθ ₁ and the difference (Δθ _k −Δθ).
₁ ) and the phase difference ｛Δθ ₁ + (Δ
θ _k −Δθ ₁ )} is output to the adder 35- (k−1).

The adder 35- (k-1) (k = 4, 5,
.., N−2) is the addition result (k−2) Δθ ₁ + ｛Δθ ₁ + (Δθ ₂ −Δ) input from the adder 35− (k−2).
θ ₁ ) +... + (Δθ _k−1 −Δθ ₁ )} and the adder 34− (k
−1), the addition result ｛Δθ ₁ + (Δθ _k −Δθ)
₁ )｝ is added, and the addition result (k−1) Δθ ₁ + ｛Δθ
_{1 + (Δθ 2 -Δθ 1)} + ... + outputs the (Δθ _k -Δθ _1)} to the least squares regression processing unit 301 adders and 35-k.
The adder 35- (N-2) adds the addition result (N-3) input from the adder 35- (N-3) Δθ ₁ + 加 算 Δθ ₁ + (Δ
θ ₂ −Δθ ₁ ) +... + (Δθ _N−2 −Δθ ₁ )} and the adder 34
− (N−2), the addition result ｛Δθ ₁ + (Δθ
_N−1 −Δθ ₁ )｝, and the addition result (N−2) Δθ
₁ + ｛Δθ ₁ + (Δθ ₂ −Δθ ₁ ) + ... + (Δθ _N-1 −Δ
θ ₁ )｝ is output to the least squares regression processing unit 301.

Here, the adder 35- (k-1) (k =
, N−1), the addition result (k−
1) Δθ ₁ + ｛Δθ ₁ + (Δθ ₂ −Δθ ₁ ) + ... + (Δθ _k
−Δθ ₁ )} is the antenna element A (k +
This is the phase difference δθ _{k + 1} between the phase of the reception weight W _{k + 1} ^RX corresponding to 1) and the phase of the reception weight W ₁ ^RX corresponding to the reference antenna element A1.

The least squares regression processing unit 301 executes the above least squares regression processing based on the input N phase differences δθ _{1 to} δθ _N, and The phase difference Δθ ^LSR after the square regression is used as the phase difference correction unit 3
02 is output. As described above, the phase difference correction unit 302 ^calculates the phase difference △ θ ^LSR after the least squares regression as −π <△ θ ^LSR ≦ π
, And outputs the detected phase difference Δθa after the phase correction processing to the multiplier 303. The multiplier 303 multiplies the transmission and reception frequency ratio f _T / f _R of the transmission frequency f _T with respect to a reception frequency f _R that is input with the detected phase difference Δθa inputted, the multiplier multiplication result Δθaf _T / f _R 36-1 to 36-N. Here, the transmission / reception frequency ratio f _T / f _R is determined from the predetermined transmission frequency f _T and the predetermined reception frequency f _R, and is determined by the multiplier 30.
3 is input.

Multiplier 36-k (k = 1, 2,..., N)
Has a multiplier set to (k-1) in advance, and the multiplier 3
03 multiplication result Δθaf _T / f _R and the multiplier (k
-1) and the result of the multiplication (k-1) Δθaf
And outputs the _T / f _R to complex arithmetic section 37-k. Complex arithmetic unit 37-k on the basis of the multiplication result kΔθaf _T / f _R which is input from the multiplier 36-k, the complex exp [j {(f
_T / f _R ) · (k−1) {θa}], and the multiplier 3
Output to 8-k. The multiplier 38-k multiplies an excitation distribution a _k determined by a predetermined transmission pattern by a complex number exp [j {(f _T / f _R ) · (k−1) {θa}]. Result a _k exp [j ｛(f _T / f _R ) ·
(K−1) {θa}] is output to each antenna element Ak as a transmission weight W _k ^TX corresponding to the antenna element Ak.

The automatic beam capture and tracking apparatus configured as described above forms a transmission beam in the direction of an incoming wave based on the baseband signal S _k for each antenna element Ak obtained from the incoming wave. Even in an environment where multiple waves arrive, or when a phase uncertainty occurs in the reception phase difference, the influence and phase uncertainty of those multiple waves are removed, and a single transmission main beam only in the direction of the maximum received wave. Can be formed.

In the automatic beam capturing and tracking apparatus according to the first embodiment, the transmission weight calculation circuit 30 calculates the phase difference δθ _{k according} to the equation (6), and performs the transmission weight based on the phase difference δθ _k. I have. Thus, the transmission weight calculation circuit 30 can calculate the phase difference Δθ ^LSR by calculating one linear regression plane. It is possible to calculate a transmission weight W _k ^TX for forming a main beam of a single transmission signal only in the direction of the reception wave.

<Second Embodiment> FIG. 6 is a block diagram showing a configuration of an automatic beam capturing and tracking apparatus according to a second embodiment of the present invention. The automatic beam capturing and tracking device of the second embodiment of FIG. 6 is configured similarly to the automatic beam capturing and tracking device of the first embodiment of FIG. 1 except for the following (1) to (3). (1) A maximum ratio combining circuit 4a is provided instead of the maximum ratio combining circuit 4. (2) A digital beam forming circuit (hereinafter referred to as D) between the quasi-synchronous detectors QD-1 to QD-N and the maximum ratio combining circuit 4a.
It is called a BF circuit. ) 50 and a beam selection circuit 60. (3) Maximum ratio combining circuit 4a and transmission weight calculation circuit 30
And a weight decomposition operation circuit 70.

In the automatic beam acquisition and tracking device according to the second embodiment, the DBF circuit 50 converts the baseband signal S _k received by each antenna element Ak and input from the quasi-synchronous detector QD-k into an antenna. A fast Fourier transform (hereinafter, referred to as FFT) is performed based on the phase of the element A1 to simultaneously form a plurality of N beams, that is, a multi-beam, and each beam B represented by the following equation 8 is formed.
i (i = 1,2, ..., N) and outputs the baseband signal S _i corresponding to the beam selection circuit 60. In each beam thus formed, a null is always formed in the direction in which another beam is formed. In the second embodiment, a beam is formed in a predetermined direction using the FFT as described above. However, the present invention is not limited to this, and a plurality of signals are received by each antenna element Ak. A plurality of beams are simultaneously formed corresponding to a plurality of predetermined beam directions by using a phase shifter, and each received signal of the plurality of beams corresponding to the formed plurality of beams is converted into a digital signal. Alternatively, a baseband signal for each reception may be generated. In this case, the decomposition of the reception weight for each antenna element can be calculated based on the phase shift amount for each antenna element added to the phase shifter to form the plurality of beams and the reception weight for each beam. In the case of controlling the amplitude in the transmission using the configuration using the phase shifter,
For example, an attenuator or amplifier that can be digitally controlled is separately used. As described above, using another beam forming method, a plurality of beams may be formed corresponding to a plurality of predetermined beam directions, and a plurality of beam reception signals corresponding to the formed plurality of beams may be generated. Good. In the second embodiment, a discrete Fourier transform (DFT) can be used instead of the FFT. Here, in order to make the phase of the baseband signal of each beam Bi having the main lobe at the same incident angle the same, the baseband signal S of each beam Bi is set so that the physical center of the array antenna 1 becomes a phase reference. Phase shift _i . Exp {j (−3/4) πi} in Expression 8 is a coefficient for shifting the phase of the baseband signal S _i of each beam so that the physical center of the array antenna 1 becomes a reference for the phase. Thus, for example, when the number N of elements of the antenna is four, four beams Bi are formed, and each beam is 0 °, ±
The maximum power is obtained in the directions of 30 ° and 90 °.

[0063]

(Equation 8)

The beam selection circuit 60 generates N baseband signals S corresponding to the N beams Bi to be formed.
_i, a plurality of M baseband signals are selected in _{descending order} of the received power, and the selected M baseband signals SB _i (i = 1, 2,..., M) are sent to the maximum ratio combining circuit 4a. Output. Here, among the selected M baseband signals SB _i , baseband signal S having the maximum power
The B _i as the reference baseband signal, represented by SB _r. In an actual communication, the baseband signal Si of each beam Bi is low-pass filtered by a narrow-band low-pass filter in order to prevent the beam selected due to the influence of noise or the like from frequently switching. The signals after low-pass filtering are compared, and the above-described beam selection is performed.

The maximum ratio combining circuit 4a includes N antenna elements A in the maximum ratio combining circuit 4 as shown in FIG.
k (k = 1, 2,..., N), N complex conjugate product operation units 41-k, N low-pass filters 42-k, and N delay circuits 43-k And N multipliers 44-k and N complex conjugate product operation units 46-k, and M beams Bi (i = 1, 2,..., M) corresponding to M beams, respectively. Complex conjugate product operation units 41-i, M low-pass filters 42-i, and M delay circuits 43-i
i, M multipliers 44-i, and M complex conjugate operation units 46-i. Then, the maximum ratio combining circuit 4a
, Instead of the N baseband signals S _k, based on the M baseband signals SB _i, the N base band signals S _k the M baseband operations and similar operations were performed on This is performed on the signal SB _i to output a synthesized signal described later to the demodulator 5 and to receive a beam weight W _i described later.
^RB is output to the weight decomposition operation circuit 70. Here, the reception beam weight W _i ^RB includes a base band signal SB _r corresponding to the reference beam Br, can be represented by the complex conjugate product of the complex conjugate of the baseband signal SB _i corresponding to the beam B _i, It can be expressed by the following equation 9. here,
In the present invention, preferably, although a baseband signal SB _r of the reference baseband signal having the maximum power as described above, the present invention is not limited to this, other than the base band signal having the maximum power The baseband signal may be used as a reference.

[0066]

[Equation 9] ^{_{W i RB = SB r · SB}} i * = | SB r || SB i | exp {j (φ r -φ i)}

Here, φ _r and φ _i are the phases of the baseband signals of the reference beams Br and Bi, respectively, | · | is the amplitude of the signal, and * is a conjugate complex number. I have. The received beam weight W _i ^RB for each beam Bi given by Equation 9 is multiplied by the baseband signal S _i of that beam Bi, and this is calculated for all the beams selected by the beam selection circuit 60, and the sum of them is calculated. Calculate. As a result, the baseband signal SB _i
Is a magnitude proportional to the amplitude of the baseband signal SB _i, and the reference beam Br baseband signal SB _r
^Are multiplied by the reception beam weight W _i ^RB for in-phase with the above phase, and are combined, that is, the maximum ratio combining by the beam space without a feedback loop is realized. In the actual communication, in order to prevent the beam from becoming unstable due to the influence of noise or the like, the real part and the imaginary part of the reception beam weight W _i ^RB expressed by Expression 9 are reduced using a narrow-band low-pass filter. Low-pass filtering.

Further, by dividing the calculated sum by the square root of the sum of the squares of the sizes of the reception beam weights for all the beams selected by the beam selector 60, the standardized synthesized signal is obtained. SB _N is obtained. If you choose the M beams in the beam selector 60, the combined signal SB _N is expressed as Expression 10.

[0069]

(Equation 10)

The standardized maximum ratio combined output signal SB
_N is input to the demodulator 5 and demodulated by the demodulator 5.

[0071] Next, a method for forming a single transmit beam in the arrival direction of the received wave with the largest power with the received beam weight W _i ^RB corresponding to the beam Bi by the calculation. When transmission and reception frequencies are equal (TD
D system, etc.), and by using the reception weights used when the maximum ratio combining of the beam reception signals corresponding to the beams is transmitted as it is, a transmission beam is formed in the multipath direction with the maximum reception wave, thereby forming a diversity transmission system. can do. However, when the transmission and reception frequencies are different, the phase relationship between the paths differs between transmission and reception, so it is necessary to suppress transmission in the multipath direction.

First, in the weight decomposition operation circuit 70, the reception beam weight W _i ^RB for performing the maximum ratio combining obtained for each beam Bi of the reception signal is converted into the DFT (or FFT) executed by the DBF circuit 50. The coefficient exp {j (i /
2) Using πk} and a coefficient exp {j (−3/4) πi} for phase shifting the baseband signal S _i , decompose the reception weight W _k ^RX corresponding to each antenna element Ak. For example, when the number N of antenna elements is four, each antenna element A
The reception weight W _k ^RX of _k can be expressed as in Equation 11 using the reception weight W _i ^RB of each beam Bi.

[0073]

[Equation 11]

Here, each of the obtained antenna elements Ak
Of the reception weight W _k ^RX of each antenna element Ak with respect to the reference with respect to the center of the array antenna 1.
Is the phase difference of the received signal. Here, the antenna element Ak is used by using all the formed beams.
Although the reception weight W _k ^RX is calculated for each of them, it is also possible to calculate using only the M beams selected by the beam selection circuit 60.

Based on the reception weight W _k ^RX calculated as described above, the transmission weight W _k ^TX is calculated in the same manner as in the first embodiment, and the transmission weight W _k ^TX is used. The transmission signal is transmitted by forming a transmission beam in the arrival direction of the received wave having the maximum power.

The automatic beam capturing and tracking apparatus of the second embodiment configured as described above forms a beam in a predetermined direction, receives a received wave corresponding to the beam, and outputs received data. As well as having the same effects as the first embodiment.

<Modifications> Although the automatic beam capture and tracking devices of the first and second embodiments are configured using the array antennas 1 arranged on one straight line, the present invention is not limited to this. As shown in FIG. 8, the antenna elements are separated by a predetermined distance λ.
The antenna may be configured using an array antenna 100 that is two-dimensionally juxtaposed at / 2. Hereinafter, using an array antenna 100 in which antenna elements are two-dimensionally juxtaposed,
A method of calculating the reception weight and the transmission weight when the automatic beam capture and tracking device is configured will be described.

As shown in FIG. 8, the array antenna 100 includes sixteen antenna elements A arranged in the u and v directions at intervals of λ / 2 of a half wavelength of the reception frequency λ on the plane. (0,0) to A (3,3).
Here, each of the antenna elements A (0,0) to A (3,3)
The numbers in parentheses after A in the reference symbol are attached in correspondence with the coordinates (u, v) normalized by λ / 2 on the uv plane. In the following description, the reception weight corresponding to each antenna element A (u, v) is represented by W ^RX (u,
v), and the phase of the reception weight W ^RX (u, v) is θ
(U, v). Now, with reference to antenna element A (0, 0) at coordinates (0, 0), reference antenna element A
The phase difference of the reception weight between (0, 0) and the antenna element A (1, 0) at coordinates (1, 0) adjacent in the u direction is Δθ _u (0, 0), and the reference antenna element A (0,
0) and the antenna weight A (0, 1) at the coordinates (0, 1) adjacent to the antenna element A (0, 1) in the v direction.
θ _v (0,0).

First, when the coordinates (u, v) on the uv plane are used, the phase difference in the case of a one-dimensional (array antenna arrayed on a straight line) is obtained by transforming equation (6) to obtain the following equation (1).
It can be represented by 2.

[0080]

(Equation 12)

Here, u = 1, 2, 3 and
Are coordinates arranged on a straight line in the u direction at the position of v = 0
Each of the antenna elements A (1,0), A (2,
0), A (3, 0) for each antenna element A (u,
Reception weight corresponding to ^{0) W RX (u, 0} ) and the reference reception weight W corresponding to the antenna elements A (0,0) of
^RX (0,0) phase difference θ (u, 0) −θ (0,0)
0). Further, in Equation 12, Δθ
_u (u, 0) {= θ (u + 1,0) -θ (u, 0)}
Are antenna elements A (u + 1,
0), A (u, 0) and the reception weight W ^RX (u +
1,0), W ^RX (u, 0), and arg
() Is represented in the range from -π to π.

As described above, from Expression 12, the phase difference θ (u, 0) −θ (0,0) (u = 1) of the reception weights of the antenna elements arranged linearly in the u direction at the position of v = 0 ,
2,3) was found. Next, similarly, the antenna elements A arranged on a straight line in the v-axis direction, for example, each antenna element A of coordinates (0, v) arranged on a straight line in the v direction at u = 0
For (0, v), the phase difference Δθ _v (0, 0) of the reception weight between the reference antenna element A (0, 0) and the antenna element A (0, 1) adjacent thereto in the v direction is used. , The phase difference θ between each two receiving weights between each antenna element A (0, v) and the reference antenna element A (0,0)
(0, v) -θ (0,0) can be expressed by the following equation 13.

[0083]

(Equation 13)

Here, Δθ _v (0, v) (= θ (0, v
+1) -θ (0, v)) are the reception weights W ^RX (0, v + 1), W ^RX (0, v) is the phase difference.

Further, any u (u = 0, 1, 2, 3)
For each antenna element A (u, v) at coordinates (u, v) aligned on a straight line in the v direction
Using the phase difference Δθ _v (0,0) of the reception weight between (0,0) and the antenna element A (0,1) at the coordinates (0,1) adjacent in the v direction, each antenna element A
Antenna element A (u, v) at (u, v) and coordinates (u, 0)
0), the phase difference θ (u,
v) -θ (u, 0) can be expressed by the following equation 14.

[0086]

[Equation 14]

Here, Δθ _v (u, v) (= θ (u, v)
+1) −θ (u, v)) are reception weights W ^RX (u, v + 1), W ^RX ( u, v). From the above Expressions 12 and 14, the reference phase difference Δθ _u (0, 0) and the reference phase difference Δθ are obtained for each antenna element A (u, v) at an arbitrary coordinate (u, v).
_{Using v} (0,0), reception weight W ^RX (u, u) corresponding to each antenna element A (u, v) at each coordinate (u, v)
v) and the phase difference θ (u, v) −θ between the reception weight W ^RX (0,0) corresponding to the reference antenna element A (0,0).
(0,0) can be represented by Expression 15.

[0088]

(Equation 15)

From Equation 15, each antenna element A (u, v)
Reception weight W ^RX (u, v) of the reception weights corresponding to the reference antenna element A (0,0) W ^RX (0 for,
0), the phase distribution is regressed to a plane by the least squares method, the phase difference after the square regression is calculated, and the first and second phases are calculated based on the phase difference. Similarly to the embodiment, it is possible to calculate a transmission weight for forming a main beam of a single transmission signal only in the arrival direction of a reception wave having the maximum reception power.

Next, a description will be given of a two-dimensional fast Fourier transform method in the case where the antenna element is configured using the array antenna 100 arranged two-dimensionally.
Here, as described above, a plurality of N (N = 16 in this modification) antenna elements of the array antenna are arranged at regular intervals d.
Are arranged side by side in a two-dimensional matrix shape, and the input signal of each antenna element is Skm (θ) (k = 0, 1,...,
N−1; m = 0, 1,..., N−1), the result of the Fourier transform in the first dimension is expressed as in Expression 16. Here, k is a coordinate in the Fourier transform of the first dimension, and m
Is the coordinates of the second-dimensional Fourier transform. The combined beam output Bkm (k = 0, 1,..., N-1; m = 0, 1,.
N-1) is the result of the Fourier transform in the second dimension, and is expressed as in Expression 17 using the result of the Fourier transform in the first dimension.

[0091]

(Equation 16)

[0092]

[Equation 17]

Here, the relationship between the direction θkm of the multi-beam and the beam obtained by the Fourier transform is expressed by the following equation (18). In Equation 18, the multi-beam direction θkm is expressed in the form of (x, y), where x is an angle with respect to the Z axis in the XZ plane, and y is with respect to the Z axis in the YZ plane. Angle.

[0094]

Equation 18 θkm = [sin ⁻¹ {sin (−2π · k / N)}, sin
⁻¹ {sin (−2π · m / N)}], k = 0, 1, 2, 3,..., N−1; m = 0, 1, 2,
3, ..., N-1

As is apparent from the above equations (16) and (17), the antenna beam Bkm is equal to the signal output Skm of the receiver.
Is the discrete Fourier transform of. Thus, according to the present invention,
Using the principle of the automatic beam capturing and tracking device of the first and second embodiments, an automatic beam capturing and tracking device can be configured using the array antenna 100 in which antenna elements are juxtaposed two-dimensionally.

<Other Modifications> In the first embodiment described above, the antenna element A1 is used as a reference antenna element.
The present invention is not limited to this, and may be based on another antenna element. Even with the configuration described above, the same operation as in the first embodiment is performed and the same effect is obtained.

In the second embodiment described above, the signal having the largest power among the signals selected by the beam selection circuit 60 is used as the reference signal. However, the present invention is not limited to this, and other signals are used as reference signals. It may be a signal. Even with the configuration described above, the same operation as in the second embodiment is performed and the same effect is obtained.

[0098]

According to the first aspect of the present invention, there is provided an array antenna control device for controlling an array antenna including a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape. A converting means for converting a plurality of received signals respectively received by the antenna elements of the array antenna into two orthogonal baseband signals orthogonal to each other using a common local oscillation signal; Based on each of the two orthogonal baseband signals, a reception signal received by a predetermined reference antenna element of the plurality of antenna elements and a reception signal received by an arbitrary antenna element of the plurality of antenna elements Reception way for calculating a reception weight corresponding to the given antenna element, which is a complex conjugate product of the received signal and a conjugate complex number Calculation means, a reception weight corresponding to the antenna element adjacent to the reference antenna element calculated by the reception weight calculation means, and a reception weight corresponding to the reference antenna element calculated by the reception weight calculation means. Based on a first complex conjugate product, which is a complex conjugate product with a conjugate complex number, between a reception signal received by an antenna element adjacent to the reference antenna element and a reception signal received by the reference antenna element A first phase difference calculating means for calculating the first phase difference, and two reception weights corresponding to antenna elements adjacent to each other among the reception weights calculated by the reception weight calculation means. A second complex conjugate product that is a complex conjugate product of one of the two reception weights and a conjugate complex number of the other reception weight is given by Complex conjugate product calculating means for calculating a third complex conjugate product that is a complex conjugate product of the second complex conjugate product and a conjugate complex number of the first complex conjugate product; Based on the third complex conjugate product calculated by the calculating means, the phase difference between the received signals received by the antenna elements adjacent to each other is calculated based on the first complex conjugate product.
Second phase difference calculating means for calculating a second phase difference which is a difference from the phase difference between the first phase difference calculating means and the first phase difference calculated by the first phase difference calculating means. A receiving position for calculating a reception phase difference between a reception signal received by an arbitrary antenna element and a reception signal received by the reference antenna element based on the second phase difference calculated by the means. Phase difference calculation means, based on each of the reception phase differences calculated by the reception phase difference calculation means, corresponding to the arrangement of each antenna element and each phase difference of any antenna element with respect to the reference antenna element, the minimum By using the square method, of the received waves received by the plurality of antenna elements, the largest received wave having the largest power is regressed to an equiphase first-order regression plane to perform the first-order regression. Regression plane calculation means for calculating a plane, The transmission phase difference is calculated by multiplying the phase difference between any two adjacent antenna elements of the primary regression plane calculated by the return plane calculation means by the ratio of a predetermined transmission frequency to a predetermined reception frequency. Transmission phase difference calculating means. This allows the main beam to be transmitted in the direction of the received wave received at the maximum power without using the azimuth sensor and even in an environment where a plurality of multiplexed waves arrive, or even when the phase uncertainty occurs in the received phase difference. It is possible to provide an array antenna control device that can be formed and that can be easily operated.

According to a second aspect of the present invention, there is provided an array antenna control apparatus for controlling an array antenna including a plurality of antenna elements juxtaposed closely at equal intervals. Conversion means for converting a plurality of reception signals received by each antenna element of the antenna into two orthogonal baseband signals orthogonal to each other using a common local oscillation signal,
Based on each of the two orthogonal baseband signals converted by the conversion means, a predetermined beam forming method is used to form a plurality of beams corresponding to a plurality of predetermined beam directions. Beam forming means for generating a plurality of beam reception signals corresponding to the beams, and among the plurality of beam reception signals generated by the beam formation means, selecting a beam reception signal having power equal to or higher than a predetermined threshold value; Beam selection means for outputting at least one beam reception signal, a reference beam reception signal among the beam reception signals selected by the beam selection means, and an arbitrary one of the beam reception signals selected by the beam selection means Receiving beam weight calculation means for calculating a reception beam weight corresponding to the arbitrary beam, which is a complex conjugate product of a conjugate complex number of the beam reception signal of Based on the reception beam weights calculated by the reception beam weight calculation means, reception weight calculation means for calculating each reception weight corresponding to each of the antenna elements, and among the reception weights calculated by the reception weight calculation means, A first complex product that is a complex conjugate product of a reception weight corresponding to an antenna element adjacent to a predetermined reference antenna element of the plurality of antenna elements and a conjugate complex number of a reception weight corresponding to the reference antenna element A first phase calculating a first phase difference between a reception signal received by an antenna element adjacent to the reference antenna element and a reception signal received by the reference antenna element based on the conjugate product; The phase difference calculating means and the reception weights calculated by the reception weight calculating means correspond to antenna elements adjacent to each other. Based on the reception weight, and one of the reception weights of the two reception weight, and calculates a second complex conjugate product is the complex conjugate product of the complex conjugate of the other reception weight, the second
Complex conjugate product of the first complex conjugate product and a complex conjugate product of the first complex conjugate product, and a third complex conjugate product of the first complex conjugate product, A second phase difference which is a difference between the phase difference between the reception signals received by the antenna elements adjacent to each other and the first phase difference, based on the complex conjugate product of the third position. A phase difference calculating means, and an arbitrary antenna element based on the first phase difference calculated by the first phase difference calculating means and the second phase difference calculated by the second phase difference calculating means. Receiving phase difference calculating means for calculating a receiving phase difference between the received signal received and the received signal received by the reference antenna element; and each of the receiving phase differences calculated by the receiving phase difference calculating means Based on the above, it corresponds to the arrangement of each antenna element and Using the least squares method, the phase difference between an arbitrary antenna element and an antenna element is equalized in phase with respect to the largest received wave having the largest power among the received waves received by the plurality of antenna elements. Regression plane calculating means for calculating the first regression plane by regressing on the first regression plane; and a position between any two adjacent antenna elements of the first regression plane calculated by the regression plane calculation means. A transmission phase difference calculating means for calculating a transmission phase difference by multiplying the phase difference by a ratio of a predetermined transmission frequency to a predetermined reception frequency. Thereby, a plurality of beams are formed corresponding to a plurality of predetermined beam directions, a plurality of beam reception signals corresponding to the plurality of formed beams can be received, and without using the direction sensor, Further, even in an environment where a plurality of multiplexed waves arrive or a phase uncertainty occurs in a received phase difference, a transmission main beam can be formed in a direction of a received wave received with the maximum power, and the operation is simple. A simple array antenna control device can be provided.

The array antenna control device according to a third aspect of the present invention is the array antenna control device according to the first or second aspect, further comprising: Phase correction means for converting a phase difference between any two antenna elements into a phase difference having a value in a range from -π to + π, wherein the transmission phase difference calculation means is converted by the phase correction means. The transmission phase difference is calculated by multiplying the phase difference by the ratio of the transmission frequency to the reception frequency. Thereby, even when the phase difference between adjacent antenna elements is a value near π or −π, the phase uncertainty is determined without using the azimuth sensor and in the environment where a plurality of multiplexed waves arrive or the reception phase difference. Even when this occurs, a transmission main beam can be formed in the direction of the received wave received with the maximum power, and a control device for an array antenna that can be easily operated can be provided.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a configuration of an automatic beam acquisition and tracking device for a communication array antenna according to a first embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration of a maximum ratio combining circuit 4 of FIG.

FIG. 3 is a block diagram illustrating a configuration of a transmission weight calculation circuit 30 of FIG. 1;

FIG. 4 is a flowchart of a phase correction process executed by a phase difference correction unit 302 in FIG. 3;

FIG. 5 is an explanatory diagram illustrating a regression process of a reception phase to a primary plane by a least square method in a least square regression processing unit in FIG. 3;

FIG. 6 is a block diagram showing a configuration of an automatic beam acquisition and tracking device for a communication array antenna according to a second embodiment of the present invention.

FIG. 7 is a block diagram illustrating a configuration of a maximum ratio combining circuit 4a of FIG. 6;

FIG. 8 is a plan view showing the arrangement of antenna elements in an array antenna 100 of a communication array antenna automatic tracking and tracking apparatus according to a modification of the present invention.

[Explanation of symbols]

DESCRIPTION OF SYMBOLS 1 ... Array antenna, 2 ... Low noise amplifier, 3 ... Down converter, 4, 4a ... Maximum ratio combining circuit, 5 ... Demodulator, 6, 6-1 thru 6-N ... Quadrature modulator, 7 ... Up converter, 8 ... 9: in-phase distributor, 10: second local oscillator, 11: first local oscillator, 12: second local oscillator, 13-1 to 13-N: phase / amplitude corrector, 14 ... 1 transmission local oscillator, 31-1 to 31- (N-1), 32-1 to 32-
(N-2), 41-1 through 33-1 through 33- (N-
1)... Phase difference calculator, 34-1 to 34- (N-2), 35-1 to 35-
(N-2), 45, 47... Adders, 36-1 to 36-N, 38-1 to 38-N, 44-
1 to 44-N, 303: multiplier; 37-1 to 37-N: complex number operation unit; 41-N, 46-1 to 46-N: complex conjugate product operation unit; 42-1 to 42-N: low Band-pass filters, 43-1 to 43-N delay circuits, 48 square root calculation units, 49 dividers, 30 transmission weight calculation circuits, 50 DBF circuits, 60 beam selection circuits, 70 weight separation circuits, 301: least square regression processing unit; 302: phase difference correction unit; A1 to AN: antenna elements; CI-1 to CI-N: circulator; RM-1 to RM-N: receiving module; AD-1 to AD- N: A / D converter; QD-1 to QD-N: quasi-synchronous detection circuits; QM-1 to QM-N: quadrature modulation circuits; TM-1 to TM-N: transmission modules.

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁶ , DB name) H01Q 3/26-3/42 H01Q 25/04

Claims (3)

(57) [Claims] 1. An array antenna control device for controlling an array antenna including a plurality of antenna elements closely arranged in a predetermined arrangement shape, wherein a plurality of antenna elements received by each of the antenna elements of the array antenna are provided. A conversion unit for converting the received signal into two orthogonal baseband signals orthogonal to each other using a common local oscillation signal; and a plurality of the plurality of orthogonal baseband signals based on the two orthogonal baseband signals converted by the conversion unit. The received signal received by a predetermined reference antenna element of the antenna elements and the complex conjugate product of the conjugate complex number of the received signal received by any of the plurality of antenna elements Receiving weight calculating means for calculating a receiving weight corresponding to the antenna element; and A first complex conjugate that is a complex conjugate product of a reception weight corresponding to an antenna element adjacent to the reference antenna element and a conjugate complex number of the reception weight corresponding to the reference antenna element calculated by the reception weight calculation unit A first phase difference calculating a first phase difference between a received signal received by an antenna element adjacent to the reference antenna element and a received signal received by the reference antenna element based on the product; Calculating means, based on two reception weights corresponding to antenna elements adjacent to each other among the reception weights calculated by the reception weight calculation means, one reception weight of the two reception weights and the other reception weight. A second complex conjugate product, which is a complex conjugate product of the reception weight of the second complex conjugate and the conjugate complex number, is calculated, and the second complex conjugate product and the first complex conjugate product are calculated. Complex conjugate product operation means for calculating a third complex conjugate product that is a complex conjugate product with a complex conjugate product of A second phase difference calculating means for calculating a second phase difference which is a difference between the phase difference between the received signals received by the elements and the first phase difference; and the first phase difference calculating means. Based on the first phase difference calculated by the second phase difference calculating means and the second phase difference calculated by the second phase difference calculating means, a reception signal received by an arbitrary antenna element and a signal received by the reference antenna element A receiving phase difference calculating means for calculating a receiving phase difference between the received signal and the received signal, based on each of the receiving phase differences calculated by the receiving phase difference calculating means, corresponding to the arrangement of the antenna elements; And any antenna for the reference antenna element Using the least squares method, each phase difference of the elements is converted into a first-order regression plane having the same phase with respect to the maximum reception wave having the maximum power among the reception waves received by the plurality of antenna elements. Regression plane calculation means for performing regression to calculate the primary regression plane; and a phase difference between any two adjacent antenna elements of the primary regression plane calculated by the regression plane calculation means,
Transmission phase difference calculation means for calculating a transmission phase difference by multiplying a ratio of a predetermined transmission frequency to a predetermined reception frequency, and a transmission position between each of the two antenna elements calculated by the transmission phase difference calculation means. A controller for an array antenna, wherein a transmission main beam is formed in the direction of the maximum received wave by transmitting a transmission signal from each of the antenna elements with a phase difference. 2. An array antenna control device for controlling an array antenna including a plurality of antenna elements juxtaposed closely at equal intervals, comprising: a plurality of reception signals received by each antenna element of the array antenna; Means for converting the two orthogonal baseband signals into two orthogonal baseband signals orthogonal to each other using a common local oscillation signal, and a predetermined beamforming method based on each two orthogonal baseband signals converted by the conversion means. Beam forming means for forming a plurality of beams corresponding to a plurality of predetermined beam directions, and generating a plurality of beam reception signals corresponding to the formed plurality of beams, A beam reception signal having power equal to or higher than a predetermined threshold value is selected from the plurality of beam reception signals, and at least one beam reception signal is selected. Beam selecting means for outputting a received signal, a reference beam receiving signal among the beam receiving signals selected by the beam selecting means, and an arbitrary beam receiving signal among the beam receiving signals selected by the beam selecting means Receiving beam weight calculating means for calculating a receiving beam weight corresponding to the arbitrary beam, which is a complex conjugate product of a complex conjugate with a conjugate complex number of: Receiving weight calculating means for calculating each of the receiving weights corresponding to the elements; of the receiving weights calculated by the receiving weight calculating means, an antenna element adjacent to a predetermined reference antenna element among the plurality of antenna elements; Between the corresponding reception weight and the conjugate complex number of the reception weight corresponding to the reference antenna element. Based on a first complex conjugate product that is a prime conjugate product, a first signal between a received signal received by an antenna element adjacent to the reference antenna element and a received signal received by the reference antenna element is generated. A first phase difference calculating means for calculating a phase difference; and a receiving weight calculated by the receiving weight calculating means, based on two receiving weights corresponding to antenna elements adjacent to each other, based on the two receiving weights. A second complex conjugate product that is a complex conjugate product of one of the reception weights and a conjugate complex number of the other reception weight is calculated, and the second complex conjugate product and the first complex conjugate product are calculated. Complex arithmetic product operation means for calculating a third complex conjugate product that is a complex conjugate product with a conjugate complex number; antennas adjacent to each other based on the third complex conjugate product calculated by the complex arithmetic product operation means A second phase difference calculating means for calculating a second phase difference which is a difference between the phase difference between the received signals received by the elements and the first phase difference; and the first phase difference calculating means. Based on the first phase difference calculated by the second phase difference calculating means and the second phase difference calculated by the second phase difference calculating means, a reception signal received by an arbitrary antenna element and a signal received by the reference antenna element A receiving phase difference calculating means for calculating a receiving phase difference between the received signal and the received signal, based on each of the receiving phase differences calculated by the receiving phase difference calculating means, corresponding to the arrangement of the antenna elements; And, using the least squares method, the phase difference of each antenna element with respect to the reference antenna element is converted into the maximum received wave having the maximum power among the received waves received by the plurality of antenna elements. On the other hand, regression to the first-order linear regression plane A regression plane calculating means for calculating the linear regression plane, the phase difference between any two antenna elements adjacent to each other in the linear regression plane calculated by the regression plane calculating unit,
Transmission phase difference calculation means for calculating a transmission phase difference by multiplying a ratio of a predetermined transmission frequency to a predetermined reception frequency, and a transmission position between each of the two antenna elements calculated by the transmission phase difference calculation means. A controller for an array antenna, wherein a transmission main beam is formed in the direction of the maximum received wave by transmitting a transmission signal from each of the antenna elements with a phase difference. 3. The array antenna control device further includes a phase difference between any two adjacent antenna elements on the primary regression plane calculated by the regression plane calculation means, the phase difference being from -π to + π. Phase correction means for converting the phase difference into a value within a range, wherein the transmission phase difference calculation means multiplies the phase difference converted by the phase correction means by a ratio of the transmission frequency to the reception frequency to transmit a signal. 3. The control device for an array antenna according to claim 1, wherein a phase difference is calculated.