JP3133537B2 - PWM control device for AC motor - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a PWM controller for an AC motor, and more particularly, to a controller for calculating a three-phase voltage reference signal for reducing a deviation of a speed feedback signal from a speed reference signal to zero. The present invention relates to a PWM control device for an AC motor, comprising: a PWM control circuit for performing PWM control of an inverter based on a phase voltage reference signal, wherein the AC motor is variable-speed controlled by an output of the inverter.

[0002]

2. Description of the Related Art A voltage-type PWM inverter device can be controlled with high accuracy and high speed by a PWM inverter for both output voltage and output frequency, and is therefore widely used for driving an AC motor at a variable speed.

In this type of inverter device, when the output frequency of the inverter is continuously changed in a wide frequency range including direct current (0 Hz), the continuous energization time of a specific arm becomes longer in a low frequency output range. Losses are concentrated in the elements of the arm, and as a result, the current carrying capacity must be reduced and used as a whole.

FIG. 11 shows a configuration example of a control section 1A according to the prior art. The controller 20 included in the CPU unit 10 receives the speed reference signal ω _r ^* , the speed feedback signal ω _r and the current feedback signal I, performs well-known speed control and vector control calculations, and outputs a voltage reference signal V ^* . . The voltage reference signal V ^* is converted into a PWM pulse train by the carrier signal CR in a well-known PWM control circuit 33 based on the principle of a comparator, and outputs a gate pulse signal GP. The drive of the AC motor is controlled by a gate pulse signal GP via a voltage source inverter (not shown).

FIG. 12 shows a main circuit waveform in principle when the inverter is driven by such a PWM control device. Here, a description will be given of a typical case of driving in a low-frequency region, which is a problem of the related art, under a condition that a DC (0 Hz) output is output and a Z-arm current is maximum. 3-phase voltage reference signal _{^{V u *, V v *,}} V w * is the carrier signal C
A pulse train (gate pulse) V _{AK (U)} corresponding to R,
The _signals are modulated to V _{AK (V)} and V _{AK (W),} and the constituent elements of the inverter, for example, GTO elements (large power transistors) are turned on / off based on the gate pulses. In the case of DC (0 Hz) output, the induced voltage of the AC motor is almost zero, each voltage reference signal has a very small value corresponding only to the voltage drop, and the inverter main circuit 100 connected in a three-phase bridge form. U, V, W, X, Y,
Of the six arms Z, a so-called chopper operation is performed in which the power is fixed to the two arms U and V on the positive side and the one arm Z on the negative side. U, V, and Z arm voltages are V _{AK (U)} , V
_{AK (V)} , V _{AK (Z)} and the frequency of the carrier signal CR are represented by f
_c (therefore, the period of the carrier signal CR 1 / f _c)
FIG. 12 shows the waveforms of the current and voltage of each part defined as shown in FIG.

In FIG. 13, an inverter 100 has a GT
It is assumed that a three-phase bridge is formed by O elements, and a freewheeling diode is connected to each arm element in anti-parallel. The currents flowing through the GTO elements of the U, V and Z arms are respectively denoted by I _{GTO (U)} ,
_{IGTO (V)} and _{IGTO (Z),} and the currents flowing through the diodes of the W arm and the X and Y arms are represented by _{ID (W)} and _{ID (W)} , respectively.
_{ID (X)} and _{ID (Y),} and U and W of the AC motor 101
The _inter- phase voltage and the V- and W-phase voltages are V _uw and V _vw , respectively, and the W-phase line current is I _LW .

[0007] In the case of Figs.
Unlike the case in which the conduction current _{IGTO (Z) of the} TO element is the maximum and the AC output, it is not shared with the GTO element of the positive W arm every half cycle, and in the form of the waveform shown in the drawing. Note that it flows continuously.

FIG. 14 is a conceptual diagram of the energizing current I _am and the generated loss P _am of one arm (switching operation is regarded as an equivalent sine wave). In the case of AC output, it is sufficient to share only half a cycle. For example, when the frequency becomes 0 Hz at time t ₁ , a current of π times the average value in the case of AC output is continuously supplied, and the generated loss is reduced. P _am also has a continuous heating waveform substantially corresponding to the current.

From the viewpoint of generated loss, 0 Hz is the most severe condition. However, in the case of extremely low frequency which is close to this, a large current reduction is necessary from the viewpoint of temperature rise of the element as shown in FIGS. Become. FIG. 15 shows a heat transfer model of the element and the cooler, and is represented by a cascade network of generated loss P, thermal resistance R _thx, and heat capacity C _thx (X = 1, 2, 3). T _j is the junction temperature of the element, T _c is the element case temperature, T _f is the cooler temperature, T T
_a is the ambient air temperature. Actually, if the output frequency is sufficiently high with respect to the thermal time constant of the element (generally on the order of about 1 second), the loss P is averaged as shown in FIG. T _j and element case temperature T
Whereas appears as an increase of _c, FIG (B), (C)
As the output frequency becomes lower, the temperature rise fluctuates greatly following the loss P, and the ultimate point of the junction temperature _Tj of the element becomes higher.

FIG. 17 shows the generated loss P according to the output frequency f.
FIG. 5 is a diagram illustrating an example of _LOSS and a temperature rise _Tr of an element. The horizontal axis indicates the frequency f, and the vertical axis indicates the inverter output voltage V, the average conduction current I _{AV of} the element, and the element generation loss P in order from the top.
_It shows _LOSS , current limit value I _LIM , and element temperature rise _Tr .

The conduction ratio of the GTO element is approximately expressed by the following equation.

(Conduction rate) = 1− {1− (modulation rate) × (load power factor)} / 2 (1) Here, the modulation rate is substantially expressed by the following equation.

(Modulation rate) = K × f / f _B (2) where K: constant f: output frequency f _B : base frequency If the base frequency is f _B or less, the output frequency decreases and GT increases.
Although the conduction loss of the O element is slightly reduced, the temperature rise of the element rapidly increases at the frequency (f ₂ or less) at which the above-mentioned element starts to interfere with the thermal time constant of the element, and the current concentrates near 0 Hz (f ₁ or less). The temperature rise becomes significantly large.

[0014]

Based on the above study,
SUMMARY OF THE INVENTION An object of the present invention is to provide a PWM control device capable of alleviating a situation in which a generated loss is concentrated on a specific element of an inverter in a low frequency region and avoiding a reduction in a low frequency conduction capacity of the inverter.

[0015]

In order to achieve the above object, the present invention provides a common voltage reference having a polarity opposite to the polarity of the phase having the largest absolute value among the three-phase voltage reference signals in a low frequency output range. The present invention is characterized in that a voltage reference correcting means is provided, which corrects each of the three-phase voltage reference signals using the correction signal and uses the corrected three-phase voltage reference signals as new three-phase voltage reference signals.

[0016]

In the low-frequency output region, each phase voltage reference signal is corrected by a voltage reference correction signal having a polarity opposite to the polarity of the maximum amplitude phase among the voltage reference signals of each phase, so that a line voltage of an inverter output is obtained. Can be reduced and the loss can be dispersed by reducing the conduction ratio of the maximum current carrying arm, thereby improving the carrying ability in the low-frequency output region.

[0017]

FIG. 3 is an overall conceptual diagram showing a PWM control device according to the present invention together with a main circuit configuration. AC motor (M)
The control device 1 is provided for an inverter main circuit 100 that drives the variable speed 101 at a variable speed. The control device 1 controls speed or torque. Speed of the AC motor 101 is detected by the speed detector (SS) 102, is introduced to the control device 1 as a speed feedback signal omega _r. The speed feedback signal ω _r is compared with a separately provided speed reference signal ω _r ^*, and outputs a gate pulse signal GP for zeroing the deviation (speed deviation) to control the inverter 5. The motor current is detected by the current detector 7 and used as a current feedback signal I of a current control minor loop (not shown).

FIG. 1 shows a detailed configuration of the control device 1. The control device 1 includes a CPU 10 that outputs an operation amount (here, a voltage reference signal V ^** ) obtained as a result of the control operation, and a PWM that modulates the voltage reference signal V ^** into a gate pulse train GP using a carrier signal CR. It comprises a control circuit 33.
The CPU unit 10 is actually configured by software, and includes a voltage reference correction unit 30, a mode determination unit 31, and a function generation unit 32, in addition to the controller 20 that performs speed control and vector control calculations.

The controller 20 has a well-known configuration for performing calculations for speed control and vector control.
As shown in FIG. 2, a speed control unit 21 which receives the speed reference signal ω _r ^* and the speed feedback signal ω _r and outputs a torque current reference signal τ ^* , is connected to its output side, and receives a voltage reference signal V ^* And a vector control unit 22 for outputting a frequency reference signal F ^* . Vector control unit 22, current reference enter the speed feedback signal omega _r and the torque current reference signal tau ^* vector I _dq ^* and sliding speed reference signal omega _s
^* , And a voltage reference vector V _dq for _reducing the difference (current deviation) between the current feedback vector I _dq and the current reference vector I _dq ^* obtained by coordinate transformation from the current feedback signal I to zero. ^* Current control unit 24 for calculating ^*
And the voltage reference vector V _dq using the phase reference signal θ _o ^*
And a coordinate converter 25 for coordinate-converting ^* into a voltage reference signal V ^* and coordinate-converting the current feedback signal I into a current feedback vector I _dq . The vector control unit 22 includes a means for obtaining a primary angular frequency reference signal ω _o ^* by adding the slip velocity reference signal ω _s ^* to the velocity feedback signal ω _r , in addition to the above-described circuit parts. It includes means for integrating the frequency reference signal ω _o ^* to obtain the phase reference signal θ _o ^* , and means for obtaining the frequency reference signal F ^* from the primary angular frequency reference signal ω _o ^* .

Referring back to FIG. The three-phase voltage reference signal V output from the controller 20 having the above configuration
_{^{* (V * = (V u}} *, V v *, V w *)) is facilities in the voltage reference correction unit 30, for each energization mode, the correction as the current duty ratio of the maximum current supply arm is reduced And converted into a new voltage reference signal V ^** . That is, in the mode determining unit 31, switches the mode approximately every 60 ° in the voltage reference of phase, 3-phase, respectively the correction signal V _c of a polarity opposite the polarity of the amplitude maximum phase among the voltage reference signal V ^* of 3-phase , The conduction ratio of the maximum current carrying arm is reduced without changing the line voltage, and the generated loss is dispersed. Function generator 32 is intended to determine the value of the voltage reference correction signal V _C is added frequency range to function the working voltage reference correction signal V _C, for a voltage reference correction the frequency reference signal F ^* as an input Voltage reference correction signal V _C
Is calculated and output.

Next, based on the voltage reference signal V ^* output from the controller 20, the mode determining unit 31
The generation of the mode signal MD which is switched approximately every 60 ° by the logical procedure shown in FIG. Voltage reference signal V ^*
Voltage reference signal is actually three-phase _{^{V u *, V v *,}} V
_w ^* (see FIGS. 7 and 8).
By multiplying the phase voltage reference signal (step 50), a signal P _fl representing a sign that changes approximately every 60 ° in the voltage reference phase is obtained, and this signal P _fl is non-negative (P _fl ≧
0) and negative (P _fl <0) (step 51). In the case of P _fl <0, it is checked which of the three-phase voltage reference signals has the minimum value (the one with the maximum absolute value). If the U-phase, the mode signal MD is set; if MD = 1, MD is set; = 2, MD = 3 for W phase (step 5
2). Similarly, if P _fl ≧ 0, the maximum value of the three-phase voltage reference signals is checked, and if U-phase, MD = −
1, MD = −2 for V phase, MD = −3 for W phase (step 53).

As described above, since it is in the low frequency region that the continuous conduction time of one element becomes longer and the loss occurs, and the concentration of the temperature rise becomes harmful, the function generator 32 performs the following operations. 5 or 6 to form a voltage reference correction signal V _C as a function of the frequency reference signal F ^* obtained by a controller 20.

[0023] In Figure 5, the limit frequency -F _L, as F _L that are considered low-frequency ^{_{region, -F L ≦ F * ≦ F}} L
In the frequency range, F ^* ≦ -F _L or F ^* ≧ F _L
In this case, V _C = 0, but a voltage reference correction signal V _C that changes so as to increase substantially in the form of a linear function as F ^* = 0 is output.

In FIG. 6, the voltage reference correction signal V _C takes a constant value in a frequency range considered as a low frequency region, but from the viewpoint of control stability, the frequency reference signal F
^* When increase of the have to have a hysteresis characteristic take different elapsed between the time of reduction, F _L2 <as F _L1, when increasing frequency reference signal F ^* is the voltage reference correction by -F _L2 ≦ F ^* ≦ F _L1 The signal V _C takes a constant value, and when the frequency reference signal F ^* decreases, the voltage reference correction signal V _C takes a constant value with −F _L1 ≦ F ^* ≦ F _L2 . In other frequency regions, V _C = 0.

The correction limit frequency F shown in FIGS.
_L to F _L1 and F _L2 may be determined using the frequency f ₂ in FIG. 17 as a guide.

As a first embodiment, the voltage reference correction unit 30 corrects the initial voltage reference signal V ^* according to the following equation using the voltage reference correction signal V _C according to the sign of the signal P _fl , Generate signal V ^** . [P when the ^{_{fl ≧ 0] V u ** =}} V u * -V C ... (3) V v ** = V v * -V C ... (4) V w ** = V w * -V C ... (5) [when the ^{_{P fl <0] V u **}} = V u * + V C ... (6) V v ** = V v * + V C ... (7) V w ** = V w * + V C ... (8) The contents represented by the above equation are illustrated in FIG. The signal P _fl , which is obtained by multiplying the voltage reference signals V _u ^* , V _v ^* , V _w ^* of each phase before correction and which switches between positive and negative at every 60 °, is opposite to the signal P _fl polarity of the voltage reference correction signal V _C of the voltage reference signal _{^{V u *, V v *,}} V w
^* New voltage reference signal by adding each of the V _u ^**, V _v
^** and V _w ^** are generated. When such an operation is added, the phase voltages (signals _Vu ^** , _Vu ^** ,
V _v ^**, V _w ^**) is made discontinuous, as the line voltage is expressed as the difference between the phase voltage of two-phase voltage reference correction signal V
_C is offset original voltage reference signal _{^{V u *, V v *,}} V
A continuous sine wave output voltage having a waveform corresponding to _w ^* can be obtained.

By performing the above operation on the voltage reference signal V ^* , the conduction ratio of the maximum current carrying arm can be reduced as shown in FIG. FIG. 9 corresponds to FIGS. 12 and 13 shown as the prior art, and shows a DC (0 Hz) output, a Z arm current maximum, and P _fl <0, M
The principle main circuit waveform at D = 3 is shown. In addition,
Here, each signal is used according to FIGS. The voltage reference signal V ^* is modulated by being biased to the positive side by the voltage reference correction signal V _C.
Current I of the GTO element of the Z arm compared to the case of
It can be seen that the _{GTO (Z)} is almost identical in size and has a much shorter flow duration.

[0028] By doing as described above, as shown in FIG. 10, under the same output voltage V and 1 element average energizing current I _AV in the case of FIG. 17, the generation loss P _LOSS in a low frequency region Concentration is reduced, and the maximum value of the element temperature rise _Tr is suppressed low, so that the low-frequency current carrying capacity of the entire inverter device can be improved.

Next, a second embodiment of the present invention will be described.

The mode determining section 31 shown in FIG. 1 obtains the mode signal MD which switches at intervals of about 60 ° in the procedure shown in FIG. 4 in the same manner as described above. According to the mode signal MD obtained by the mode determination unit 31, the voltage reference correction unit 30 outputs the three-phase voltage reference signals _Vu ^* ,
V _v ^*, V _w ^* voltage reference absolute value of the opposite polarity to the polarity of the voltage reference signal for the maximum phase of the correction signal V _C
(Or -V _C ), and for other phases, a voltage reference correction signal V _{C having a} polarity opposite to the polarity of the voltage reference signal of that phase.
(Or -V _C), the absolute value of the voltage reference signal of that phase is corrected by adding a minus the voltage reference signal of the largest phase, the new voltage reference signal V _u ^**, V _v ^** ,
V _w ^** . For example, the U-phase voltage reference signal _Vu ^*
Is the maximum (MD = −1 or MD = 1
Is as follows. [In the case of ^{_{MD = -1] V u ** =}} -V C ... (9) V v ** = -V C + (V v * -V u *) ... (10) V w ** = -V C ^{_{+ (V w * -V u *}} ) ... (11) [ in the case of ^{_{MD = 1] V u ** =}} V C ... (12) V v ** = V C + (V v * -V u *) ... (13) in ^{_{V w ** = V C + (}} V w * -V u *) ... (14) the above equation, equation (10), (11), (13), the second term on the right-hand side of (14) Note that each of the terms expresses the line voltage using the phase voltage signal, and has a sinusoidal waveform. Therefore, this case also the phase voltage (signal V _u ^**, V _v
^**, V _w ^**) is made discontinuous, as the line voltage is expressed as the difference between the phase voltage of two-phase voltage reference correction signal V _C is canceled original voltage reference signal V _u ^*, V _v ^* , V _w ^*
Can be obtained.

FIG. 8 shows the contents of the above equation. Approximately 60 ° voltage reference signal V by cutting switched mode signal MD for each _{^{u *, V v *, V}} w * a new voltage reference signal V _u ^**, respectively, V _v ^**, converted to V _w ^** I have.
Although the output phase voltage (reference voltage signal) becomes discontinuous as shown in the figure even when operated in this manner, a continuous sinusoidal voltage can be obtained as the line voltage as described above.

As described above, the conduction ratio of the maximum current carrying arm is reduced in the same manner as in the first embodiment, and as shown in FIG.
_LOSS , and consequently, the concentration of the element temperature rise _Tr can be avoided, and the low-frequency current carrying capacity of the entire inverter device can be improved.

In the above description, for convenience, the inverter 5 is made of a GTO element. However, the present invention is applicable to any PWM inverter using a PWM inverter using a self-extinguishing type switching element. is there.

[0034]

As described above, according to the present invention, P
In a WM-controlled inverter, the duty ratio of the maximum current flow element in a low-frequency region is reduced without affecting the line voltage, that is, without affecting the control characteristics, thereby reducing the generated loss. Is significantly reduced, and the low-frequency current carrying capacity of the inverter device can be greatly improved.

[Brief description of the drawings]

FIG. 1 is a block diagram showing an embodiment of a PWM control device according to the present invention.

FIG. 2 is a block diagram showing a configuration example of a controller inside a control device according to the present invention.

FIG. 3 is a connection diagram showing a relationship between a control device according to the present invention and a main circuit.

FIG. 4 is a flowchart showing a mode determining procedure of a mode determining unit in FIG. 1;

FIG. 5 is a characteristic diagram showing a first function example of a function generating unit in FIG. 1;

FIG. 6 is a characteristic diagram showing a second function example of the function generator in FIG. 1;

FIG. 7 is an explanatory diagram for explaining the operation of the first embodiment of the voltage reference correction unit in FIG. 1;

FIG. 8 is an explanatory diagram for explaining the operation of a second embodiment of the voltage reference correction unit in FIG. 1;

FIG. 9 is an operation waveform diagram of each part of the main circuit for explaining the operation and effect of the present invention.

FIG. 10 is a diagram illustrating various characteristics with respect to an output frequency for explaining the operation and effect of the present invention.

FIG. 11 is a block diagram schematically showing a general PWM control device according to the related art.

FIG. 12 is an operation waveform diagram of each part of the main circuit corresponding to FIG. 9 for explaining the operation of the conventional device.

FIG. 13 is a main circuit connection diagram shown for explaining reference numerals in FIG. 12;

FIG. 14 is a conceptual waveform diagram for explaining a relationship between a current flow state of one arm and a generated loss.

FIG. 15 is an equivalent circuit diagram showing a heat transfer circuit model of the arm element.

FIG. 16 is an explanatory diagram showing a difference between a generation loss and a temperature rise of an element due to a difference in frequency.

FIG. 17 shows various characteristics with respect to the output frequency of the conventional device.
FIG.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 Control apparatus 10 CPU part 20 Controller 30 Voltage reference correction part 31 Mode determination part 32 Function generation part 33 PWM control circuit 100 Inverter main circuit 101 AC motor 102 Speed detector

──────────────────────────────────────────────────の Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H02P 5/408-5/412 H02P 7/628-7/632 H02P 21/00 H02M 7/42-7 / 98

Claims (7)

(57) [Claims] A controller for calculating a three-phase voltage reference signal for reducing a deviation of the speed feedback signal from the speed reference signal to zero;
A PWM control circuit for controlling the AC motor at a variable speed by the output of the inverter.
In the M control device, the three-phase voltage reference signal is corrected by using a common voltage reference correction signal having a polarity opposite to the polarity of the phase having the largest absolute value among the three-phase voltage reference signals in the low-frequency output region. A PWM control device for an AC motor, further comprising a voltage reference correcting means for generating a three-phase voltage reference signal. 2. The control device according to claim 1, wherein said voltage reference correction means includes a common voltage having a polarity opposite to the polarity of the phase having the largest absolute value among the three-phase voltage reference signals in a low frequency output region. A PWM controller for an AC motor, wherein a new three-phase voltage reference signal is obtained by adding a reference correction signal to each of the three-phase voltage reference signals. 3. The control device according to claim 1, wherein said voltage reference correcting means determines a polarity of the voltage reference signal for a phase having a maximum absolute value among the three-phase voltage reference signals in a low frequency output region. Is replaced with a voltage reference correction signal of the opposite polarity, and for the other phases, a voltage reference correction signal of the opposite polarity to the polarity of the voltage reference signal of that phase is used. A PWM control device for an AC motor, wherein a new three-phase voltage reference signal is obtained by adding a value obtained by subtracting a voltage reference signal. 4. The control device according to claim 1, wherein a three-phase voltage reference signal is input to said voltage reference correction means, and a phase having an absolute value which changes substantially every 60 ° is a maximum value. A PWM control device for an AC motor, further comprising mode determination means for determining the polarity and outputting a mode signal. 5. The control device according to claim 1, wherein the speed reference signal is input to the voltage reference correction means, and a voltage reference correction signal according to a predetermined function is provided in a low frequency output region. A PWM control device for an AC motor, further comprising a function generating means for outputting. 6. A PWM control apparatus for an AC motor according to claim 5, wherein said function generating means outputs a voltage reference correction signal which changes in a substantially linear function form in a low frequency output range. 7. The PW of an AC motor according to claim 5, wherein the function generating means outputs a voltage reference correction signal having a constant value in a low frequency output range.
M control device.