JP3206478B2 - High frequency heating equipment - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

The present invention relates to a high-frequency heating device using a magnetron, and more particularly to a power supply for driving a magnetron.

[0002]

2. Description of the Related Art As shown in FIG. 13, a conventional high-frequency heating apparatus uses a circuit configuration called a one-piece voltage resonance type circuit. 1 is a commercial power supply, 2 is a rectifier composed of full-wave rectification, which rectifies the commercial power supply to form a DC power supply. The choke coil 3 smoothes the fluctuation of the current, and the smoothing capacitor 4 smoothes the fluctuation of the voltage. These inductors and capacitors form a low-pass filter circuit. Energy is accumulated in the winding of the leakage type transformer 14 from the DC power supply at the point (a). The primary winding 13b of the leakage transformer 14 and the resonance capacitor 6 arranged in parallel with it constitute a tank circuit, and resonate with the energy stored in the winding of the leakage transformer 14. Switching means comprising a transistor 7 and a diode 8 arranged in parallel with the transistor 7 are connected in series to the tank circuit.

The secondary side of the leakage type transformer 14 is
A secondary winding 13a and a heater winding 13c are provided. The voltage induced in the secondary winding 13c is voltage-double rectified by a half-wave voltage rectifier circuit comprising a high voltage capacitor 10 and a high voltage diode 11, and the voltage of -4k is applied to the cathode of the magnetron 12.
It is configured to apply a DC high voltage of about V.

A current transformer 15 is a magnetron 12
The anode current flowing to the primary side is detected,
The next-side inverter control circuit 9 is electrically insulated and the information is fed back. The inverter control circuit 9 controls the on time of the transistor 7 so that the signal from the current transformer 15 becomes constant, and controls the power transmitted to the secondary side. In this control algorithm, since the anode current is controlled at a constant value, the duty due to the anode current, which causes a large damage to the magnetron if it increases excessively, can be controlled to an appropriate level, which is a very advantageous control algorithm in terms of magnetron quality and reliability. It has become.

Next, a mechanism for driving the transistor 7 and supplying power to the secondary side will be described with reference to FIG. The time chart in the upper part of FIG. 14A shows a signal relationship for generating a pulse signal (e) for driving the transistor 7 with a certain set high-frequency output, and the time chart in the lower part shows the pulse train.

The upper triangular waveform is sliced by the high frequency output reference signal (f). When the triangular wave (g) signal is lower than (f), a high signal is output, and conversely, a low signal is output. The respective times are t1 and t2, and when t1 is long, the energy stored in the primary winding 13b increases. The operating frequency f is 1 / (t1 + t2).
At the vertex of the triangular wave (g), the voltage at the point (a) crosses the voltage at the point (b) in FIG. 13, and when the voltage transitions from (a)> (b) to (b) <(b) A cross trigger occurs, and the slope turns down. (Period tB in the upper time chart) Then, when the signal crosses the inversion threshold value (C), the mode is naturally changed, and the slope changes to an ascending slope (period tA in the upper time chart). By repeating this, a triangular wave as shown in (g) is formed.

Further, the circuit configuration and operation for forming the above-described triangular wave (g) will be specifically described with reference to FIG.
Point (a) is a complete direct current when the circuit is not operating, but once operated and power is consumed on the secondary side, it has a waveform obtained by full-wave rectification of the sine shape of the power supply. When the effective voltage E is E = 100 V, the peak value is about 141 V when E = 100 V. On the other hand, point (b) is a resonance voltage waveform generated in the above-described tank circuit, which is the voltage waveform of the semiconductor switching means in FIG. 17D, and its peak voltage is about 500 V, which is much higher than that of point (a). Big. Referring back to FIG. 15, reference numeral 26 denotes a comparison timing circuit, and a voltage obtained by dividing the point (A) by the resistors 22 and 23 and a voltage obtained by dividing the point (B) by the resistor 2
The crossed trigger is detected by comparing the divided voltages in steps 4 and 25. The inversion threshold (C) is determined by dividing the circuit power supply voltage Vcc by the resistors 20 and 21. Now, when the transistor 31 is on, the electric charge of the capacitor 17 is discharged via the resistors 16 and 30, and the triangular wave (g) enters a falling slope state. Next, when the triangular wave (g) falls and crosses the inversion threshold (c), the output (d) of the comparison timing circuit 26 turns off the transistor 31 via the resistor 29, and the capacitor 17 is discharged by the resistor 16 to discharge the triangular wave. (G) turns to the rising slope. And it reverses again by the above-mentioned cross trigger and turns into a descending slope.

The triangular wave (g) is compared with a high-frequency output reference signal (f) of 28 by a comparator 27 to generate a pulse signal (e). The circuit configuration shown here exists inside the inverter control circuit 9 and the pulse signal drives the transistor 7 via an appropriate driver circuit.

As described above, the reason why the transistor 7 is driven by using the cross trigger and the comparison timing circuit 26 is as shown in FIG. 16 showing the relationship between the current and the voltage of the semiconductor switching means. (A) The rising slope of the voltage waveform rises relatively gently in a sinusoidal manner to reduce the switching loss generated as the product of the current and the voltage, and the diode current with a predetermined delay from the cross trigger. Is flowing, that is, (a) when the voltage waveform is 0 V, the transistor 7 is turned on to similarly reduce the switching loss. This is a major feature of the voltage resonance system.

FIG. 17 is a time chart of waveforms of main parts. (A) is a current waveform of the semiconductor switching means,
(B) shows the current waveform of the magnetron, (c) shows the current waveform of the leakage type transformer, and (d) shows the voltage waveform of the semiconductor switching means. A current flows through the transistor during the period (I), and a resonance voltage of the tank circuit occurs during the period (II). The diode current flows during the period (III), and the transistor is turned on for this period. In this manner, the switching loss of the transistor is minimized.

[0011]

However, since the conventional magnetron drive power supply uses a single-type voltage resonance type circuit, when power control is performed by the operation of the resonance circuit, the off period (resonance circuit) shown in FIG. T2) is almost fixed, so that when the ON period (t1) for changing the power is changed, the operating frequency changes.

For example, in the high-frequency output setting of FIG. 14A, the frequency f is f = 1 / (t1 + t2), whereas in the case of the higher power setting (b), the frequency f 'is f' = 1 /. (T1 ′ + t2 ′). Here, the resonance period is almost the same for both, and considering t2 = t2 ′, the frequency decreases in proportion to the increase in power.

FIG. 18A shows the relationship between the high frequency output and the operating frequency. Obviously, it has a high frequency output dependency. Further, as shown in FIG. 18B, the current flowing through the magnetron heater also has a dependency corresponding to the same high-frequency output. FIG. 19A is a diagram illustrating a main part of the high-voltage circuit. Heater winding 1 of transformer 14
As for the electric power supplied from 3c, choke coils for reducing unnecessary radiation from the working space of the magnetron are included in the normal lines. The capacitor 33 is a feed-through type line bypass capacitor, and is similar to a choke coil in terms of effect. Under the influence of the high frequency filter circuit formed inside the magnetron 12, the power supplied to the cathode 34 changes depending on the operating frequency of the transformer, that is, the frequency of the voltage supplied from the heater winding 13c. Therefore, it is obvious that the cathode temperature also changes.

FIG. 19B is a graph showing the characteristics. As the high-frequency output decreases, the heater current decreases and the power supply to the cathode decreases due to the high-frequency removing effect of the high-frequency filter circuit described above. Of course, the cathode temperature tends to decrease. The recommended cathode temperature of the magnetron is generally 1900K to 2100K, based on the material of the cathode, and as can be seen from the figure, the cathode temperature clearly deviates from the recommended range in the low-frequency output part. I understand.

At this time, insufficient emission of electrons from the cathode occurs, which is referred to as moding (a type of abnormal oscillation. In addition to the absence of microwave output, an increase in reflected waves causes an increase in the temperature of the magnetron). For magnetrons, a very damaging phenomenon occurs,
In the worst case, there was a problem that the magnetron was destroyed. This problem is inherent in the voltage resonance type inverter, and is a factor that limits the power control function, which is a feature of the high-frequency heating device including the inverter.

[0016]

SUMMARY OF THE INVENTION In order to solve the above-mentioned problems, the present invention provides a leakage type transformer connected to a DC power source and a primary side of the leakage type transformer connected to the DC power source.
A first semiconductor switching means connected in series with,
A first capacitor first semiconductor switching means resonates with energy stored in the leakage transformer during OFF, clamping all the voltage of the first capacitor
Ku and series connection of the second capacitor and the second semiconductor switching means provided, a power control circuit for driving the first semiconductor switching means and a second semiconductor switching means alternately, secondary leakage type transformer Rectifier circuit connected to the second side, and a magnetron connected to the rectifier circuit, wherein the second capacitor and the second semiconductor
One end of the series connection of the switching means is connected to the leakage
And the first semiconductor switching means.
Connect the other end to the + or-side of the DC power supply at the intermediate point
With the power control circuit has a structure to maintain the allowed operating frequency varies between conduction time of the first semiconductor switching means and the second semiconductor switching means for varying the input power constant.

According to the above invention, even if the power is controlled by extending or reducing the on-time of the first semiconductor switching means, the operating frequency itself can be controlled to be substantially constant by adjusting the second semiconductor switching means. No dependent heater current and cathode temperature drop occurs.

[0018]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention provides a leakage type transformer connected to a DC power supply, first semiconductor switching means connected in series with the DC power supply on the primary side of the leakage type transformer, Energy stored in the leakage transformer when the semiconductor switching means is off
A first capacitor that resonates with the first capacitor, a second capacitor provided to clamp the voltage of the first capacitor, and a second capacitor.
And a series connection of the second semiconductor switching means, a power control circuit for alternately driving the first semiconductor switching means and the second semiconductor switching means, and a rectifier circuit connected to the secondary side of the leakage type transformer. , And a magnetron connected to the rectifier circuit.

The second capacitor and the second capacitor
Series connection of the semiconductor switching means of
Leakage type transformer and first semiconductor switch
The other end is connected to the positive side or the negative side of the DC power
Since is connected to the side, the power control circuit is configured to maintain the first semiconductor switching means and the operating frequency is changed between conduction time of the second semiconductor switching means for varying the input power constant, the operating frequency itself And the stability of the heater current and the cathode temperature depending on the operating frequency can be improved.

Further, an anode current detecting means for detecting the anode current, provided the input current detecting means for detecting an input current, a first semiconductor switching means and the second semiconductor switching means Alternating drive to the anode current It has a power control circuit for stabilizing the output of the detection means, a rectifier circuit connected to the secondary side of the leakage type transformer, and a magnetron connected to the rectifier circuit.

[0021] Then, the power control circuit for changing between conduction time of the first semiconductor switching means and the second semiconductor switching means for the power to variable, can control the operating frequency itself almost constant. Furthermore, even if the magnetron temperature rises and the voltage between the anode and cathode drifts and the operating frequency drops, the drop in input current is detected and the operating frequency itself is controlled to be almost constant, so the heater current and cathode temperature stability Improves.

Further, the power supply apparatus has power supply voltage detecting means for detecting the power supply voltage. By doing so, the power control circuit can use the first semiconductor switching means to vary the power.
And second semiconductor switching means controlled between conduction time
The operating frequency is varied and unrelated to the power, and because it is a configuration that changes to best match with the operating frequency to the power supply voltage detecting means, a heater current supply voltage to complement the operating frequency to change the heater current fluctuates Is controlled so that the heater current and the cathode temperature depending on the power supply voltage can be improved.

Further, an anode current detecting means for detecting the anode current, provided the input current detecting means for detecting an input current, a first semiconductor switching means and the second semiconductor switching means Alternating drive type current It has a power control circuit for making the output of the detection means a constant value, a rectifier circuit connected to the secondary side of the leakage type transformer, and a magnetron connected to the rectifier circuit.

[0024] Then, the power control circuit for changing between conduction time of the first semiconductor switching means and the second semiconductor switching means for the power to variable, can control the operating frequency itself almost constant. Further, even if the magnetron temperature rises and ebm drifts to lower the operating frequency, the operating frequency itself can be controlled to be substantially constant by detecting the increase in the anode current, so that the stability of the heater current and the cathode temperature is improved.

Hereinafter, embodiments of the present invention will be described with reference to the drawings. Embodiment 1 FIG. 1 is a circuit diagram of a magnetron driving power supply according to an embodiment of the present invention.

In FIG. 1, 3 is a DC power supply, and 54 is a first power supply.
, 55 is a second capacitor, and 39 is a second semiconductor switching means, which has a parallel configuration of a transistor and a diode as shown in FIG. Reference numeral 38 denotes a first semiconductor switching means, which also has the same configuration as the second semiconductor switching means 39. 9 is an inverter control circuit having the function of a power control circuit, 14 is a leakage type transformer, 57 is high voltage diodes 60 and 61
And a high voltage condenser 58, 59, a full wave voltage doubler rectifier circuit, and 12 is a magnetron. This part is the main power circuit of the power supply for driving the magnetron. The same circuit configuration as shown in FIGS. 2A, 2B, and 2C can be considered.
, The current of the magnetron, the primary current of the leakage type transformer, and the voltage of the first semiconductor switching means all have the same waveform.

The inverter control circuit 9 has a first
An oscillation circuit function for generating drive signals for the semiconductor switching means 38 and the second semiconductor switching means 39 is built in. The operation will be described. The oscillation circuit is a self-excited oscillation circuit in which the timing of the waveform change is determined by using the magnitude relation of the internally generated voltage as a cross trigger, and the frequency changes depending on the internally generated voltage waveform as in the conventional example. is not. Therefore, the frequency of the drive signal is constant.

The lower threshold value (d) is a divided voltage value of the control circuit voltage Vcc of the resistors 47 and 48, the upper threshold value (e) is a divided value of the control circuit voltage Vcc of the resistors 45 and 46, and a triangular wave signal. (A) is a voltage across the capacitor 52,
The charge / discharge of the DC current I generated at 9 and 50 changes in a triangular waveform. The upper triangular waveform (a) in FIG. The upper threshold value (e) is a voltage at which the rising ramp waveform is charged by the DC current I and turned up. Then, the mode shifts to the mode of discharging with the DC current I, and the falling ramp waveform returns again at the lower threshold value (d), returns to the mode of the rising ramp waveform, and repeats this cycle.

This circuit operation mechanism can be understood by returning to FIG. 1 and describing the function of the comparison timing circuit 44 in detail. First, the comparison timing circuit 44 switches the switch 5
1 will be described from a state where it is opened under a command signal for opening. At that time, since the DC 2I of the current source 50 does not flow, the DC I of the current source 49 flows into the capacitor 52 and is charged to increase the voltage. Naturally, it is considered that the impedance of the comparison timing circuit 44 viewed from the capacitor 52 is almost infinite, and no current flows. At that time,
This is the mode of the rising ramp waveform described above. Therefore, the comparison timing circuit 44 compares the upper threshold value (e) with the voltage at the point (a) by a comparator, and when the voltage exceeds the upper threshold value (e), a command signal for closing the switch 51 is output and a short circuit occurs. Becomes Then, the DC current of I (∵2I
The capacitor 52 is discharged and flows so that −I) establishes the 2I current of the current source 50. This is the mode of the falling ramp waveform described above. Then, when it reaches the lower threshold value (d), a command signal for opening the switch 51 is generated, the cycle goes through one cycle, and a triangular wave (a) of self-excited oscillation is created.

This triangular wave (A) is supplied to a reference signal generation circuit 43.
Are compared by the comparator 42 with the reference signal (c) created in step (a), and the driving signals (f) and (g) as shown in FIG.
Is created. To elaborate further on this part,
The signal sent to the first semiconductor switching means 38 is a signal delayed for a predetermined short time by the second delay circuit 52. The drive signal of the second semiconductor switching means 39 is a signal which is logically inverted by the inverter 41 and similarly similarly delayed by the first delay circuit 40 for a predetermined short time. This is provided in order to secure a timing margin so that both semiconductor switching means are simultaneously turned on and are not damaged.

The above is the description of the operating principle and mechanism of driving the semiconductor switching means. Next, the operation of the circuit when the semiconductor switching means is driven will be described with reference to FIGS. First, the first
When the semiconductor switching means 38 is turned on, the equivalent circuit of the main circuit part after the DC power supply 56 is shown in FIG.
The collector current Ic of the first semiconductor switching means 38 is supplied from the smoothing capacitor 4 through the primary winding of the leakage type transformer 14 (FIG. 4A).
State m). At this time, the secondary side output of the leakage type transformer 14 starts charging the second high voltage capacitor 59 of the full wave voltage doubler rectifier circuit 57. Since the initial voltage V2 is stored in the first high-voltage capacitor 58, the magnetron is oscillated when the voltage V3 of the second high-voltage capacitor 59 satisfies the relationship of V2 + V3> Vcut (1) Vcut: cut-off voltage of the magnetron. As shown in FIGS. 4A and 4B, the current Ia
Begins to flow.

When the first semiconductor switch is turned off, the equivalent circuit becomes as shown in FIG. 3B, and the current flowing on the primary side of the leakage type transformer 14 is reduced by the first capacitor 54.
Start flowing towards. At this time, the secondary output of the leakage transformer 14 starts charging the capacitor 58 (state n). At this time, when Expression 1 is satisfied, the magnetron 12 starts oscillating, and the anode current naturally flows if the expression (1) is satisfied. The current on the primary side of the leakage type transformer 14 is as shown in FIG. The voltage of the first semiconductor switching means 38 is as shown in FIG. When this voltage reaches the initial voltage of the second capacitor 55, the diode constituting the second semiconductor switching means 39 is turned on, and the charging of the second capacitor 55 is started. The equivalent circuit at this time is as shown in FIG. Since the capacitance value of the second capacitor 55 is larger than that of the first capacitor 54, the slope of the voltage of the first semiconductor switching means 38 suddenly becomes gentle, and the state shown in FIG. Move to o. When the current flowing from the primary side of the leakage type transformer 14 toward the second capacitor 5 starts flowing from the second capacitor 5 toward the primary side, the state transits to the state p.
At this time, it is necessary to turn on the transistor constituting the second semiconductor switching means 39. When the transistor forming the second semiconductor switching means 39 is cut off during an arbitrary conduction period of T1, the state transits to a state q in which a current starts flowing from the first capacitor 4 toward the primary side of the leakage transformer. At this time, the slope of the voltage of the first semiconductor switching means 38 becomes steep,
Of the capacitor 4 to zero. When this voltage becomes zero, the state shifts to a state r in which current flows in the direction in which the diode of the first semiconductor switching means 38 conducts, and the forward voltage (0. 7
(Approximately V) is applied, and substantially zero voltage application is maintained. When the first semiconductor switching means 38 is driven again in this state, the same operation is repeated from the state m, and a switching operation for reducing switching loss can be realized. The above-mentioned initial voltage of the second capacitor is determined by turning on the second semiconductor switching means 39 for an arbitrary time T1 in the state p. That is, the longer the ON time of the second semiconductor switching means 39, the lower the initial voltage of the second capacitor 55, and as a result, the voltage of the first semiconductor switching means 38 can be reduced.

As described above, it cannot be realized by the conventional circuit configuration. The off time of the first semiconductor switching means 38, in other words, the second semiconductor switching means 39
On time can be set arbitrarily,
Further, the voltage of the first semiconductor switching means 38 is reduced (clamped) by setting the capacity of the second capacitor 5 to a value sufficiently larger than that of the first capacitor 4.
can do. Next, FIG. 5 shows that the operating frequency of the inverter does not change even when the power is controlled. Since FIG. 5A has been described above, the detailed operation thereof will not be described. FIG. 5A shows a state of a predetermined high-frequency output, and FIG. 5B shows a state of an even higher output. Expressed simply, since both slice the triangular wave (a) having the same oscillation frequency with the reference signal (c), the frequency in the state of (a) is 1 / (t1 + t
In (2), the frequency in the state (b) is 1 / (t1 + t2) and the duty ratio is different, and the frequency is constant. Therefore, in this embodiment, the frequency is constant even if the power is controlled. By doing so, the choke coil for reducing unnecessary radiation from the magnetron and the penetrating type line bypass capacitor eliminate the phenomenon that the heater power transmission amount fluctuates due to the change in the set power and the cathode temperature changes. .

FIG. 6 shows this . FIG.
Since the frequency does not depend on the high-frequency output as shown in FIG. 6A, the phenomenon in which the heater current occurs as the power is reduced as shown in FIG. 6B is greatly improved, and the cathode temperature is naturally stabilized. It has been achieved. However, it is not completely flat. The reason for this is that the heater winding of the leakage type transformer 14 that generates the heater current has a small amount of magnetic coupling to the secondary winding 13a, so that the voltage of the secondary winding 13a is reduced and the power is reduced. This is because, as the temperature increases, the heater current also decreases.

(Embodiment 2) In FIG. 7, the same reference numerals are used for the same elements as those in FIG. The operation is controlled so that the anode current detected by the anode current detecting means 62 is constant. The difference is that an input current detecting means 63 for detecting the input current is added.

Next, the operation of the power control circuit in the inverter control circuit 9 will be described in detail with reference to FIG. The anode current is detected by the anode current detecting means 62 and its load resistor 68, and the voltage across the load resistor corresponds to the anode current. Averaging is performed by a low-pass filter including a resistor 81 and a capacitor 80, and an error from an output reference signal 92 is inverted and amplified by an inverting amplifier including an operational amplifier 83 having an amplification factor determined by the resistors 79 and 65. The output is low and vice versa. As will be described later, it is finally fed back to the semiconductor switching means for controlling the output, and forms a so-called negative feedback control system, and the anode current becomes constant.

On the other hand, the input current is similarly input current detecting means 6.
3 and its load resistance 64, the voltage across the load resistance corresponds to the input current. In this system, the error from the inversion voltage 84 is inverted and amplified by an inverting amplifier composed of an operational amplifier 71 having an amplification factor determined by the resistors 69 and 70, and the output decreases as the input current increases and vice versa. ing. Then, the current I flowing through the resistor 72 changes. When the voltage is high, the voltage across the resistor 72 is small and the current is small. On the contrary, the current is large.

The first current mirror circuit formed by the transistors 73, 74 and 75 allows the transistor 7
4 and 75, a constant current source of the current I is formed. Further, the transistor first semiconductor switching means 3
A constant current source for the current I is formed in the eight transistors 77 and 78.

Thus, a circuit for putting the current I into and out of the capacitor 52 (push-pull) is formed.

The operation of the circuits after the comparison timing circuit 44 and the comparator 42 is the same as that of the first embodiment, and therefore will not be described. With such a configuration, the magnetron operates at a constant anode current, and the temperature of the magnetron rises due to aging. The generated voltage of the magnetron (which is proportional to the voltage of the secondary winding 13a of the leakage transformer 14) drops, and the input current decreases. Even if it decreases, the current I decreases and the operating frequency decreases. As a result, when the voltage of the secondary winding 13a decreases, a chain is formed in which the frequency is also reduced under the influence of the voltage, and the heater current is kept constant.

FIG. 9 shows this concretely.
FIG. 9A shows an initial state, and FIG. 9B shows a case where the generated voltage of the magnetron decreases due to aging.
As is noticeable in FIG. 9B, when the correction by the input current is applied, the constant current becomes small, so that the slope of the triangular wave (a) becomes gentle and the frequency decreases. That is, t1 + t2 <t
1 ′ + t2 ′ . FIG. 10 shows the characteristic, and when the generated voltage of the magnetron decreases, that is, when the high-frequency output decreases, the frequency also decreases, and the stability of the heater current is further improved from the characteristic of FIG. Understand.

(Embodiment 3) When the power supply voltage fluctuates, the heater current also fluctuates in proportion. For example, when the power supply voltage decreases, the secondary side output decreases when the step-up ratio of the leakage type transformer 14 is a physical fixed element. Then, the current value generated by the heater winding affected by the magnetic coupling also decreases. Thus, the power supply voltage affects the heater current.

FIG. 11 shows a main part of the inverter control circuit of this embodiment. The commercial power supply 1 is full-wave rectified by a diode bridge 91, divided into predetermined levels by resistors 89 and 90, and
6. DC is formed by a low-pass filter circuit including a capacitor 67. It is obvious that this voltage is proportional to the commercial power supply voltage. Therefore, 93 is a power supply voltage detection circuit. This voltage is different from the inversion voltage 84 by the difference of the resistance 69.
And an inverting amplifier comprising an operational amplifier 71 having an amplification factor determined by the resistor 70 and the output, the output decreases when the power supply voltage increases, and increases when the power supply voltage increases. Then, the current I flowing through the resistor 72 changes.

This is exactly the same operation as described in (Embodiment 2), and works in a direction to offset and stabilize the fluctuation of the heater current due to the power supply voltage. Therefore, even if the power supply voltage changes, the correction is made at the operating frequency and the heater current is stabilized.

(Embodiment 4) FIG. 12 shows a main part of an inverter control circuit for controlling an input current to be constant. The constant control of the input current is the same as the anode current constant control described with reference to FIG.

The characteristics when the input current is controlled to be constant are that the voltage generated by the magnetron decreases as the temperature of the magnetron increases due to aging, and the anode current increases. Here, naturally, a decrease in the heater current occurs as shown in (Example 2). Therefore, the circuit shown in FIG. 12 is used to compensate for the influence and realize a stable heater current characteristic which is not affected by the voltage generated by the magnetron.

First, the anode current is detected by the anode current detecting means 62 and its load resistor 64, and the voltage across the load resistor corresponds to the anode current. Averaging is performed by a low-pass filter including a resistor 66 and a capacitor 67, and an error from a bias voltage 88 is amplified by a non-inverting amplifier including an operational amplifier 87 having an amplification factor determined by the resistors 85 and 86. It is configured to be large and vice versa. As a result, when the magnetron is cold, the anode current is small, the output of the non-inverting amplifier is small, and the reference current I of the current mirror flowing through the resistor 72 is large. At this time, based on the concept of FIG. 9, the operating frequency is lowered, and the heater current tends to be hard to flow. On the other hand, when the aging advances and the magnetron warms up, the operating frequency increases and the heater current tends to flow easily. By optimizing the constants of the circuit series, it is possible to make the heater current stable and flat as shown in FIG.

[0048]

According to the present invention as described above, according to the present invention, a first semiconductor switching means and a second semiconductor switching means Alternating driven by the power control circuit, the first semiconductor switching means is turned on A storage period in which energy is stored on the primary side of the leakage type transformer, a resonance period in which energy flows independently to the first capacitor, and a second storage period.
The semiconductor switching means is turned on to provide a resonance period 2 in which energy flows to both the first capacitor and the second capacitor, so that the resonance periods 1 and 2 can be changed. Since the total period of the periods 1 and 2, that is, the operating frequency can be kept constant, the operating frequency is constant even when the power is controlled, and the power attenuation does not change due to the frequency characteristics of the filter circuit for preventing radio wave radiation of the magnetron. A heater current can flow. Therefore, even if the power control is performed, the heater current can be stably maintained in a region where the magnetron is not damaged, and the reliability and the life can be significantly improved.

Furthermore, in the case where the anode current detecting means controls the anode current to have a constant power, the operating frequency is changed based on the input current information obtained by the input current detecting means to complement the heater current and make it constant. The heater current is stabilized even during aging when the generated voltage changes, and the magnetron can be maintained in a region where the magnetron is not damaged, so that the reliability and the life are remarkably improved.

Further, when the same correction is applied to the fluctuation of the power supply voltage, even if the power supply voltage fluctuates and the generated voltage of the magnetron changes, the heater current is stabilized and the magnetron is not damaged. It can be held in the area, and the reliability and life are significantly improved.

Furthermore, in the case where the input current is controlled by the input current detecting means at a constant power, the operating frequency is changed based on the anode current information obtained by the anode current detecting means to complement the heater current and make it constant. The heater current can be stably maintained in a region that does not damage the magnetron even during aging when the generated voltage changes, thereby significantly improving the reliability and life.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a first embodiment of the present invention.

FIG. 2A is a circuit diagram showing an embodiment of a main power circuit of the magnetron driving power supply. FIG. 2B is a circuit diagram showing another embodiment of a main power circuit of the magnetron driving power supply. Circuit diagram showing another embodiment of the main power circuit of the driving power supply

FIG. 3A is a diagram showing an equivalent circuit of a main circuit portion of the magnetron driving power supply. FIG. 3B is a diagram showing an equivalent circuit of a main circuit portion of the magnetron driving power source. Diagram showing equivalent circuit

FIG. 4 is a waveform diagram of a main circuit part of the magnetron driving power supply.

FIG. 5A is a time chart of a voltage in a high-frequency output state of the power control circuit. FIG. 5B is a time chart of a voltage of the power control circuit in a high-frequency output state higher than FIG. 5A.

6A is a characteristic diagram of an operating frequency with respect to the high-frequency output. FIG. 6B is a characteristic diagram of a heater current with respect to the high-frequency output.

FIG. 7 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a second embodiment of the present invention.

FIG. 8 is a circuit diagram of a power control circuit of the power supply for driving the magnetron.

9A is a time chart of a voltage in a high-frequency output state of the power control circuit. FIG. 9B is a characteristic diagram of a heater current with respect to a high-frequency output in a high-frequency output state higher than FIG. 9A.

10A is a characteristic diagram of the same operating frequency with respect to a high-frequency output. FIG. 10B is a characteristic diagram of the same heater current with respect to a high-frequency output.

FIG. 11 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a third embodiment of the present invention.

FIG. 12 is a circuit diagram of a power supply for driving a magnetron of a high-frequency heating device according to a fourth embodiment of the present invention.

FIG. 13 is a circuit diagram of a power supply for driving a magnetron of a conventional high-frequency heating device.

FIG. 14A is a time chart of a voltage in a high-frequency output state of the power control circuit. FIG. 14B is a time chart of a voltage of the power control circuit.

FIG. 15 is a main part circuit diagram of the power control circuit.

FIG. 16 is a view showing a relationship between current and voltage of the semiconductor switching element.

FIG. 17 is a waveform diagram of a main circuit portion of a conventional magnetron driving power supply.

18A is a characteristic diagram of the same operating frequency with respect to a high-frequency output. FIG. 18B is a characteristic diagram of the same heater current with respect to a high-frequency output.

FIG. 19A is a circuit diagram of a main part of the high-voltage circuit. FIG. 19B is a characteristic diagram of the power control circuit.

[Explanation of symbols]

 9 Inverter Control Circuit (Power Control Circuit) 12 Magnetron 14 Leakage Transformer 38 First Semiconductor Switching Means 39 Second Semiconductor Switching Means 54 First Capacitor 55 Second Capacitor 56 DC Power Supply 57 Full-Wave Voltage Rectification Circuit ( Rectifier circuit) 62 anode current detecting means 63 input current detecting means 93 current / voltage detecting means

──────────────────────────────────────────────────の Continuing on the front page (72) Inventor Kenji Yasui 1006 Kadoma Kadoma, Osaka Prefecture Matsushita Electric Industrial Co., Ltd. (72) Inventor Yoshiaki Ishio 1006 Odakadoma Kadoma City, Osaka Matsushita Electric Industrial Co., Ltd. (56) References JP-A-5-199768 (JP, A) JP-A-63-66893 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H05B 6/66-6 / 68

Claims (4)

(57) [Claims] 1. A DC power supply, a leakage transformer connected to the DC power supply, and one of the leakage transformers.
A first semiconductor switching means connected in series with the DC power source to the next side, first to the first semiconductor switching means is resonant at <br/> energy stored in the leakage transformer during OFF , A second capacitor provided to clamp the voltage of the first capacitor, and a second semiconductor switching means connected in series.
A connection, and a power control circuit that drives and wherein the first semiconductor switching means the second semiconductor switching means alternately a rectifier circuit connected to the secondary side of the leakage transformer, connected to the rectifier circuit The second capacitor and the second semiconductor switch.
One end of the series connection of the
Between the type transformer and the first semiconductor switching means
And the other end is connected to the + or-side of the DC power supply.
Rutotomoni, wherein the power control circuit is a high frequency heating apparatus has a structure to keep the operating frequency is changed between conduction when the first semiconductor switching means and said second semiconductor switching means for varying the input power constant. 2. A DC power supply, a leakage transformer connected to the DC power supply, and one of the leakage transformers.
A first semiconductor switching means connected in series with the DC power source to the next side, first to the first semiconductor switching means is resonant at <br/> energy stored in the leakage transformer during OFF , A second capacitor provided to clamp the voltage of the first capacitor, and a second semiconductor switching means connected in series.
A connection, an anode current detecting means for detecting an anode current, an input current detecting means for detecting an input current, said first
A power control circuit for alternately driving the semiconductor switching means and the second semiconductor switching means to control the anode current constant, a rectifier circuit connected to the secondary side of the leakage transformer, and the rectifier circuit And a magnetron connected to the second capacitor and the second capacitor.
The series connection of the two semiconductor switching means has one end connected.
The leakage transformer and the first semiconductor switch
The other end at the midpoint of the
- is connected to the side, the power control circuit is Oyo first semiconductor switching means to vary the input power
Beauty and is not operating frequency varies between conduction time of the second semiconductor switching means and independent of the input power, and high-frequency heating apparatus has a structure for changing the operating frequency in response to said input current detecting means. 3. A DC power supply, a leakage transformer connected to the DC power supply, and one of the leakage transformers.
A first semiconductor switching means connected in series with the DC power source to the next side, first to the first semiconductor switching means is resonant at <br/> energy stored in the leakage transformer during OFF , A second capacitor provided to clamp the voltage of the first capacitor, and a second semiconductor switching means connected in series.
A connection, and the power supply voltage detecting means for detecting a power supply voltage, the first
The first semiconductor switching means and the second semiconductor switch.
A power control circuit for alternately driving the switching means;
Rectifier circuit connected to the secondary side of the cage type transformer,
And a magnetron connected to said rectifying circuit, said
Second capacitor and second semiconductor switching means
Is connected in series with one end of the leakage type transformer.
The other end is located at an intermediate point of the first semiconductor switching means.
Connected to the + or-side of the DC power supply,
The power control circuit controls the first power to vary input power.
The semiconductor switching means and said second operating frequency is changed between during conduction of the semiconductor switching means and independent of the input voltage <br/> power and operating frequency <br/> wavenumber depending on the power supply voltage detecting means A high-frequency heating device configured to change. 4. A DC power supply, a leakage transformer connected to the DC power supply, and one of the leakage transformers.
A first semiconductor switching means connected in series with the DC power source to the next side, first to the first semiconductor switching means is resonant at <br/> energy stored in the leakage transformer during OFF , A second capacitor provided to clamp the voltage of the first capacitor, and a second semiconductor switching means connected in series.
A connection, an anode current detecting means for detecting an anode current, an input current detecting means for detecting an input current, said first
A power control circuit for alternately driving the semiconductor switching means and the second semiconductor switching means to control the output of the input current detection means to be constant, and a rectifier circuit connected to a secondary side of the leakage type transformer When, and a magnetron connected to said rectifying circuit, said second con
Series connection of a capacitor and a second semiconductor switching means
Has one end of the leakage transformer and the first
Connect the other end to the midpoint of the semiconductor switching means.
Source of + side or - is connected to the side, the power control circuit inputs the operating frequency by varying the conduction time of the first semiconductor switching means and the second semiconductor switching means in order to vary the input power power Unrelated to
And a high-frequency heating device configured to change an operation frequency in accordance with an output of the anode current detecting means.