JP3339636B2 - Frequency modulation converter with series-parallel resonance - Google Patents

Abstract

PCT No. PCT/NO92/00133 Sec. 371 Date Feb. 28, 1994 Sec. 102(e) Date Apr. 26, 1994 PCT Filed Aug. 25, 1992 PCT Pub. No. WO93/04570 PCT Pub. Date Mar. 4, 1993A frequency-modulated converter with a series-parallel resonance, particularly for driving any ohmic or inductive load, including gas discharge tubes, wherein a commutating voltage switch in the form of a transistor is provided and is connected in series between a negative electrode of a direct current voltage source and a first terminal of an inductor, and a pulse generator circuit is provided between the voltage source and the transistor's control electrode and the inductor's second terminal is connected to the primary winding of a transformer. A first capacitor and rectifier diode are also provided in a first and second parallel branch respectively between the transistor's charge receiving and charge emitting electrodes, and a second capacitor is provided across the voltage source's electrodes, and a smoothing capacitance is provided for the voltage source, the second capacitor being connected in series with the inductor via the diode.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention is a resonant mode power converter for exciting an ohmic load ( _Rg ), including a gas discharge tube, comprising a negative electrode of a DC voltage power supply and an inductor (L). A changeover switch (Q) in the form of a transistor connected in series with a first terminal; a pulse generator circuit having an output connected to the control electrode of the transistor; and a second switch of the inductor (L). , And a transformer having a primary winding (P) connected between a positive terminal of the voltage power supply and a first and a second terminal between a charge discharging electrode and a charge receiving electrode of the transistor (Q). A first capacitor (C ₁ ) and a diode (D ₂ ) connected to each of the two parallel branches;
A second capacitor (C ₃ ) connected in series with the inductor (L) via the diode (D ₂ ) by being connected between the terminals of the voltage power supply, It relates to a power converter.

In recent years, the physical size of power converters has been dramatically reduced, which has been achieved by increasing the operating frequency. The upper limit of the operating range of a typical quasi-square pulse converter has now reached about 0.5 MHz. This greatly reduces the size of the most important passive power components, such as magnetic components and capacitors, compared to, for example, 20 kHz switches. But,
The larger partial pulse is a field transducer, where:
Efficiency is reduced due to large switching losses in power semiconductors, thus requiring more cooling, reducing the physical size of the converter and reducing its potential.

To enhance the power converter's ability to handle overloads or large load changes, for example, when used as a power source for gas discharge tubes, to enhance the performance of the power converter,
For example, as is apparent from PCT Application No. WO 90/01248 and GB-PS 1378465, it has been proposed to monitor or prevent saturation of the switching transistor by a special circuit design.

A more efficient way of converting power at higher frequencies is based on what is called "zero current switching". This uses a sinusoidal voltage that may be generated by an LC resonant tank circuit connected either in parallel or in series. Such a converter is a “resonant converter”
Called. The advantage of using a sinusoidal voltage is that the losses in power semiconductors are dramatically reduced since switching occurs mainly at zero crossings. A disadvantage of the resonant converter is that at a given power level, the peak current is many times greater than the peak current of the pulse width modulation converter. By using a low resistance semiconductor in conduction, it is possible to increase the operating frequency above 1 MHz. Therefore, power densities well above 1 W / cm ³ may be achieved.

Controllers that can be used in the form of integrated circuits in the range of over 1 MHz for use in such converters are now known, including Gennum Corporation, Burlington, Ontario, Cana.
It can be obtained from da under the name LD405. The use of this control circuit in the controller is described in Gennum Corpo
ration's `` Using LD405 in a 125 Wresonant mode powe
r supply ". For this purpose, the same company has launched the resonant circuit of the embodiment shown in principle in the attached FIG. This circuit has an inductance L, a capacitor C, a resistor R, and a load _RL . A changeover switch S is provided before the inductance L, for example in the form of a transistor. The purpose of this is to supply a direct current from the power supply V to the series resonant tank circuit LC. The resistance of the load _RL steals current from the tank circuit. As soon as the resonance process is over, the switch S
Is opened, and the power conversion from the power source S to the load _RL is interrupted.
After a predetermined time, switch S is closed again and the process is repeated. The switching frequency may change so that the average power consumed by the load _RL changes.

In a practical embodiment, this type of resonant converter operates using two changeover switches, each handling a respective half cycle of the resonance period. These switches are based on MOS field effect transistors, which are driven by respective MOSFET stages. The output stage of the illustrated embodiment is based on a Schottky rectifier diode.

However, in this embodiment of the conventional resonant converter, it is difficult to completely prevent harmonics of the resonance voltage, and it is also difficult to make the half cycles symmetric so that each half cycle has the same amount of energy. Finally, there will still be significant losses in power switches and Schottky output diodes. The RC Networks has a power switch on the RC Networks to attenuate voltage transients. These attenuation circuits cause additional losses. Thus, the efficiency will be reduced to at least 25%, and if an output stage is used, except for the rectifier diode, the loss will be about 16%.

In the converters described above and conventional devices of the same type, the capacitor is generally connected directly in parallel with the inductor and the switch is connected in series with the voltage supply. Additionally, the load will steal energy from the resonant circuit. This resonant circuit may be made as a transformer. Due to the limited amount of energy at resonance, this conventional device is generally difficult to calculate and implement. If excess energy is taken from the LC circuit, the frequency will change and complex electronic control means will be required to control the switching of the changeover switch so that the circuit remains resonant. If an overload occurs in such a circuit, the switching current of the transistor increases and becomes uncontrollable, and if the transistor is disconnected, a transient can occur that causes irreparable damage to the converter. The problem is that the transistor, ie the switch, is loaded abnormally, since the control means for protecting the transistor does not work immediately. As already mentioned, the efficiency of the exchanger is about 84% without the use of rectified output
There are still significant losses that do not reach higher.

Finally, US-PS No. 4613769 discloses a transistor oscillator, in which a capacitor is connected in parallel via the terminals of a secondary winding in the converter, the converter comprising a first waveform. It operates with a parallel sinusoidal resonance at a predetermined frequency, together with a capacitor as a shaping means. The second waveform shaping means is composed of another capacitor connected in parallel via the collector and emitter electrodes of the transistor oscillator, and operates together with the inductor in series resonance at twice the predetermined frequency.

It is an object of the present invention to provide a resonant circuit without the above or other disadvantages. This object is achieved by an embodiment of the present invention. In the present invention, in the converter comprising a first series resonance capacitor C ₁ and the rectifier diode D ₂ in the first and second parallel branch of the between the respective charge emitting and the charge receiving electrode of the transistor Q, the second the parallel resonance capacitor C ₃ is provided between the electrodes of the voltage source, further provided, the second capacitor C ₃ is connected to the inductor L in series via the diode D ₂ a smoothing capacitor for the power supply voltage Have been. In the present invention, the transistor Q is in a high resistance state in both the series and parallel resonance modes,
Diode D ₂ is conductively in the parallel resonance mode, the capacitor C ₃ higher charge than level voltage source, before the transistor Q is switched to the low resistance state, a new series when switched to the high resistance state - parallel Initiate resonance. Relationship between the capacitance of the inductor voltage U _L and the capacitor C ₁ determines the series resonance frequency of the first half cycle, the relationship between the capacitance of the inductor voltage U _L and the capacitor C ₃ is the parallel resonance frequency of the second half cycle And each half cycle of the resonance period is timed by the transistor Q in the high resistance state, and the transformer T, the inductor L, and the capacitors C ₁ , C ₃
Constitutes an RCL circuit operating in series-parallel to the voltage power supply in this manner, and the characteristic factor of the resonator is the relationship between the inductor voltage _UL or the capacitor voltages _UC1 and _UC3 and the voltage power supply U, respectively. Depends on Between the terminals of the _first secondary winding S1 of the transformer T so that in each half cycle of the resonance period a load _RG is connected in series to an inductor L consuming energy from both the inductor and the DC voltage supply. Is connected to the load _RG . Thus, the transistor operates in series with the voltage supply during the first half cycle and in parallel with the voltage supply during the second half cycle and is constantly carrying some of the total energy consumed by the load _RG . The present invention is claim 1
Are characterized in the manner shown in the characteristic part of Further features and advantages will be apparent from the appended dependent claims.

Hereinafter, the present invention will be described in detail with reference to the accompanying drawings.

FIG. 1 shows, as already mentioned, a basic circuit of a parallel resonator according to the prior art.

FIG. 2 shows a basic circuit of a converter with series-parallel resonance according to the present invention, which is used in a cold cathode gas discharge tube.

FIG. 3 shows a portion of a modification of the converter of FIG. 2 used in a hot cathode gas discharge tube.

4a-c show the voltage curves for the normal load between outputs, for the shorted output, and also for the cycle of the resonant voltage, measured between the inductors of the converter under different conduction and load conditions. ing.

FIG. 5 shows a practical embodiment applied for driving a hot cathode gas discharge tube in a converter according to the invention.

FIG. 6 shows details of an actual embodiment of a transformer as given in the circuit of FIG.

2, the first resonance capacitor C ₁ is provided in parallel between the charge emitting and the charge receiving electrode of the transistor Q which operates as a changeover switch. Load R _G
It is provided in series with an inductor L which is connected to the transistor Q and the capacitor C _1. Second resonant capacitor C ₃ is provided between the electrodes of the voltage source is connected to the inductor L via diode D _2. Diode D ₂ is between the charge emitting and the charge receiving electrode of the transistor Q, are further provided as a parallel branch.
Further, RCL is a characteristic factor determined by the relationship between the power supply voltage U and the inductor voltage _UL or the capacitor voltages U _C1 and U _C3 , and operates in series-parallel to the voltage power supply.
The resonator transformer T, the inductor L, and as the capacitor C ₁ and C ₂ form, the primary winding P of the transformer T is connected to the inductor L. Load R _G is transformer first is connected between the secondary winding S ₁ terminal T, then thus as described above, it is connected to the inductor L in series. The required size of the resonator determines the appropriate size for the apparent power that is 30% greater than the apparent power required at the selected operating frequency of the resonator. The required size of the resonator or resonant tank circuit is determined. As described above, the magnitude may be effectively determined based on the apparent power requirement. Transistor Q is controlled at a predetermined frequency that does not need to change with load.

The operation of the converter according to the invention will now be described in more detail. Transistor Q is controlled by a substantially rectangular pulse. When the transistors conduct, current flows through inductor L and transformer T such that they magnetize. The inductor L is made up of a coil and a core, an air gap ferrite. Transistor Q
There the finished conductive by reverse induction of the inductor L, a capacitor C ₁ and C ₃ are charged. But the capacitor
Volume of C ₃ is much larger than the capacitance of the capacitor C _1,
It will also be charged to the opposite polarity. Transformer T is currently being supplied with a current of the same polarity that it is receiving through transistor Q. When the voltage of the capacitor C ₁ reaches the maximum value, the direction of current is reversed, the capacitor C ₁ is discharged against inductor L and the transformer T. afterwards,
The direction of the current is reversed again, and the inductor L is connected to the diode D ₂
And discharging the energy of the transformer T to the capacitor C _3.
Transistor Q becomes conductive again, and this process is repeated.

This process can be described as having four steps. In the first stage, the transistor Q conducts, current through the transformer T flows in the I _A direction. In the second stage, the transistor ceases to conduct, but the current for the fact that the inductor L works as "tank circuit" is flowing yet I _A direction (Fig. 2) through the transformer T, while the capacitor C ₁ at the same time charges for reverse induction of the inductor L. In the third stage, the reverse induction from the inductor L stops,
The current capacitor C _3, the capacitor C ₁ so as to flow in the I _B direction (FIG. 2) through the transformer T is discharged, while the inductor L is "filled". In the fourth stage, the diode D2 as well as the transformer T until the transistor Q conducts again.
Via the capacitor C3, the inductor L is made "empty".

Each time the diode D ₂ is conducting, so that transistor Q each time becomes "0" current and voltage should be noted that it may be switched. Negative reverse induced voltage U _L of the inductor L 'is added to the supply voltage U, it is supplied to the primary winding P of the transformer T, while the capacitor C ₃ is U and U _L' is discharged by both .

Discharge of energy from the secondary winding S ₁ to the load R _G did not result in the same phase as in the primary winding P, thus only a portion of the resonant energy can be used. This allows
If the converter according to the invention is used in a gas discharge tube, a good relationship between current and voltage will generally be obtained.

Transistor Q is to operate only as a replenisher of energy imparted to the transformer T, and for phase shift, when the transistor Q is again switched on in the fourth step described above, the diode D ₂ is already transistor Q It is disconnected. Thus, the converter according to the invention achieves a very high efficiency. When the diode D ₂ is conducting, the transistor is switched on in the negative phase of the resonance, the switching losses are completely eliminated, the supply voltage when the transistor Q is disconnected is taken over by the capacitor C _1. Therefore, transistor Q operates only at the voltage necessary to maintain the nature of the induction curve of inductor L.

If the first secondary winding S ₁ is short-circuited, decreasing the impedance 0 of the transformer T, the phase of the transition between the inductor L and the transformer T is eliminated. All energy is then used to maintain resonance, and the energy consumption of the converter drops to zero. That is, the converter is in all respects safe against short circuits.

If the load R _{G on the} secondary winding S ₁ is removed, the impedance of the transformer increases and, at the wrong time, the transistor Q is switched on, so that the current consumption increases due to the lower frequency. To prevent this, a second
Secondary winding S ₂ are used for, is connected to the rectifier bridge in order to return a part of the energy to respectively the positive and negative electrodes of the voltage source. In this way, there is always some minimum impedance in the transformer T. The resonator operates within a predetermined frequency range, and energy is supplied through a rectifying bridge B to a power supply of a supply voltage and a secondary winding as shown in FIG.
Circulates between the S _2.

By determining the size of the secondary winding S ₂ fixes the voltage,
Free running loss can be minimized. A detector (not shown) may be provided to warn of a possible defect in the load _RG , for example a defective gas discharge tube, to disconnect the pulse generator circuit connected to the control electrode of the transistor Q. It is possible. Therefore, transistor Q stops replenishing the resonator.

This means that if a hot cathode gas discharge tube is used as a load on the secondary side of the transformer T, as shown in FIG.
The terminal winding S ₁ can be easily performed by connecting at least one capacitor C ₆ via the electrode K ₁ and K ₂ of the gas discharge tube.

As is known, gas discharge tubes using hot cathodes are:
In order to achieve sufficient ionization of the gas in the tube, it must be started by preheating the electrodes. So,
Discharge can occur. This is achieved by the secondary winding S ₁ and the capacitor C ₆ which are adapted to the resonance frequency of the transformer T with the cathodes K ₁ and K ₂ of heating state. Such a fit can be determined empirically or by the measured cathode thermal resistance and is added to the impedance. As long as the cathodes K ₁ and K ₂ are not sufficiently heated, the impedance is too low, most of the current from the secondary winding S ₁ is used to heat the cathode. Only when a condition for resonance exists does the voltage rise to a level that causes the electrodes to be ignited. When the discharge between the electrodes K ₁ and K ₂ occurs, the capacitor C ₆ is not operated as a longer resonance capacitor, due to the low tip of impedance than the frequency, provide a degree of glow voltage continue heating the electrodes. This is even more beneficial if dimming is used by reducing the supply voltage.

The converter according to the invention also has a power factor cos φ of up to 0.95, as required by the new European norm,
Also, pulsating DC current may be used without smoothing for direct driving of the gas discharge tube without using phase compensation. If the frequency is 60kHz, for example, capacitor C1
Is determined to be 0.005 μF and capacitor C ₃ is determined to be 0.22 μF, but at 100 kHz, capacitor C ₁ is determined to be 0.003 μF
, The capacitor C ₃ is chosen in 0.15MyuF. Considering the wavelength of the transport of the cathodic material between the electrodes, it is further suggested that an operating frequency of 30-35 kHz is optimal for the current length of the gas discharge tube.

The actual operation of a converter according to the present invention can be readily understood by considering FIGS. 4a-c.

Figure 4a and 4b shows the properties of the inductor voltage U _L measured at inductor L terminals. Voltage in Figures 4a and 4b, the average peak - shows the peak value U _L. Figures 4a and 4b
In, while the transistor conducts in t _2, the total period of the voltage is pulse t _1. In Figure 4a, 2 winding load of S ₁ is usually due to the outflow of energy voltage is low (here, 0.6 U _L), t ₁ is short (in this case, 0.15
t _1). In FIG. 4b, the secondary winding is shorted. Since the energy flow stops, the voltage increases (here 1.3U _L
), And t ₂ also increases.

FIG. 4c shows the period of the resonance voltage under different load conditions. The nature of the normal sine wave is indicated by F _1. Transformer T
Curve “F ₂ ” exists when the “turn off” energy, and curve F ₃ exists when the secondary winding S ₁ is short-circuited.
Always conducting in the free-running diode negative half cycle, i.e. the period T _D. Whatever the load condition, transistor Q conducts in period _tQ1 .
If energy is taken from transformer T, transistor Q will operate in period _tQ2 , since the system is self-regulating. If If the secondary winding S ₁ is shorted, the transistor Q conducts during a period T _Q2, energy is no longer transformers without being emitted toward the T, and in the negative half cycle, the inductor L is energy transmitting a majority of the capacitor C _3.

A practical embodiment of the converter according to the invention will now be described with reference to FIG. In this regard, it should be understood that FIGS. 2 and 3 show a rather basic embodiment of the converter according to the invention.

As seen in FIG. 5, rectifier bridge B ₂ is passing a DC voltage from the AC power supply, the voltage is smoothed by the capacitor C ₂ and C _3. And a bistable multivibrator in the form of a Schmitt trigger circuit having an inverting output of six gates A ₁ -A _6, the pulse generator circuit a diode D ₁₂ in Aru embodiment shown supplies. Voltage pulse generator circuit is regulated by the zener diode Z _1, it is smoothed by the capacitor C _4. In the embodiment shown, via resistors R _U , R ₆ and capacitor C ₅ , the fundamental frequency and pulse width are adjusted to the desired values,
Across the resistor R ₆ , the diode D ₁ , and the variable resistor R _U (R ₅ ), the pulse generator circuit provides an astable multivibrator circuit. The output of the gate A ₁ is to transmit the pulse is substantially rectangular, is controlled substantially in parallel to the input of the four gates A ₃ -A _6. The respective outputs from the same gate are also connected in parallel and are connected to the control electrode of a transistor Q used as a switch. When a normal bipolar transistor is used, the control input naturally becomes the base of the transistor. When a MOS field-effect transistor is used instead, the control electrode is naturally the same as the gate electrode. Therefore, transducers according to the present invention, as away of a rectifier diode D ₂ and the second parallel branch in Figure 2, it may be implemented as it is convenient using integrated free-running diode.

When the transistor Q conducts, the exciting current is transmitted to the inductor L and the primary winding P of the converter T, and a resonance process is started. Resonant frequency may be summed finely variable resistor R _U. Secondary winding S ₁ of the transformer T, as described in more detail FIG. 2, conveys the voltage and current to a given load. In FIG. 5, a third secondary winding in a transformer T
S ₃ is further provided. It is if the load is a gas discharge tube, the first terminal of the secondary winding S ₃ as is shown is connected to the electrodes of the gas discharge tube, since its second electrode is grounded, secondary winding S ₃ is used to increase the ionization voltage of the load for secure ignition at very low temperatures.

The transformer T is in an actual embodiment implemented as an E-core converter, as shown in detail in FIG. The core and windings may be made for high frequency, ie in the MHz range, for example in the form of a ferrite strip with a dielectric film on which the windings are arranged. However, very small configurations are possible in non-traditional applications, such as E-core transformers used at frequencies of 30-100 kHz. Further, as shown in the embodiment of FIG. 5, the inductor L is combined with the primary winding P of the transformer T.

Rectifier bridge B 2 winding S _2, which is connected to _1, the rectifier diode D ₇ -D ₁₀ in the DC voltage is so sized resulting in bridge B ₁ is determined. This voltage is lower than the voltage between C ₂ and C ₃ in the normal operation. Resistors R ₁₀ and R _11,
Constitute a voltage divider against the capacitor C _20. Capacitor C
_20, the pulse generator circuit is given a value that determines a desired time interval before disconnecting the astable multivibrator via the diode D _5. If If the signal is low A _1, the output of the gate A ₃ -A ₆ is also low. Period of disconnection is determined by the capacitor C ₂₀ through the resistor R ₇ and R _8. After a certain time, so multivibrator is started again, will also enter the low of the inverting amplifier A _2, the output is high. However, it is also possible to implement the safety function in other ways using the prior art, and the circuit shown here is intended only as an example of a practical embodiment of the converter according to the invention and has no meaning However, the purpose of the present invention should not be limited.

The point of the transducer in accordance with the present invention, the resonance capacitor C ₁ in the embodiment of FIG. 5 is nine capacitors C _1a -C _1i (not shown)
And operates only as a resonance capacitor during a half cycle of the frequency. Resonant tank circuit, i.e. the second half period of the resonant frequency used for refilling the inductor L ₁ is given by the discharge of C ₁ to via a capacitor C _3. In this regard, flyback oscillators tend to provide an asymmetric frequency characteristic with energy drawn from the resonant tank circuit already in the first flyback so that the next half cycle receives a lower amount of energy. It may be mentioned what is known in the art. Since the transformer T shall operate as a load to operate symmetrically, the resonance capacitor C ₁ of FIG. 2 must receive a greater charge by during the second half period. This already exists in the capacitor C ₃ connected in series with the inductor L via diode D _2, is achieved by a charge voltage level is identical in the first half period. The correct size determined and appropriate supply voltage U of the resonant capacitor C _3, the transformer is given the same amount of energy in both half-cycles. However, the energy in each half cycle of resonance becomes symmetric. This allows the primary winding P of the transformer to be coupled with the use of an air gap in the transformer T.
Without being biased by DC components, the resonator supplies a nearly perfect sinusoidal voltage.

Correct capacitance value of the capacitor C ₁ and C _3, and similarly to the appropriate supply voltage U, by choosing the value of the impedance of the inductance values and the transformer T of the inductor L correctly, switching losses are completely eliminated, inductive components Transistor Q due to the phase shift between current and voltage at
Can operate at very low currents in the circuit, so that very high efficiencies can be achieved. In effect, transistor Q turns the resonant circuit 0 for positive and negative cycles of resonance.
Voltage switch to be set to Therefore,
The transistor eliminates the tendency of the resonator to relax and maintains a predetermined frequency. On the other hand, transistor Q
Is not conducting, the current is mainly collected by the inductor L. This may also be achieved by an individual adaptation of the air gap used in the transformer T for a given load _RG characteristic. Therefore the air gap may be used actively for controlling the energy flow out of the inductor L and capacitor C _1. The determination of the correct size of the transformer T used and the air gap is determined by S ₁
Is completely short-circuited, thereby completely resonating the resonator in the frequency range determined by the transistor Q.

Finally, it should be noted that pulse generators can be properly implemented in other ways than as non-stable multivibrators, as it would be obvious to one skilled in the art, for example, that non-stable multivibrators could be replaced by digital frequency synthesizers. I have to put it. When using an asymmetric multivibrator, the frequency can only be controlled within the range of 10-15%. The digital frequency synthesizer is 10
It drives the converter according to the invention over a frequency range extending over 0 MHz, but the frequency generated can be easily controlled over an octave band. The converter can also be used in HF and VHF applications where a stable, symmetric, high resonance voltage is required. In addition, all components included in the pulse generator circuit, Schmitt trigger gates A ₁ -A ₆ ,
It is clear that the transistor Q and the transistor Q may be conveniently integrated into one chip. Using the converter according to the invention, the losses are limited to losses in the transformer, losses in the pulse generator circuit, losses in the energy loss in the resonant inductor and losses at the input of the rectifying bridge. Thus, practical embodiments of the converter according to the invention achieve efficiencies on the order of 97%, since the total loss is kept below 5%.

──────────────────────────────────────────────────続 き Continued on the front page (58) Field surveyed (Int.Cl. ⁷ , DB name) H05B 41/24 H02M 7/48 H02M 7/537

Claims (10)

(57) [Claims] A resonant mode power converter for exciting a resistive or inductive load ( _Rg ), including a gas discharge tube, comprising: a negative electrode of a DC voltage power supply and a first electrode of an inductor (L). A changeover switch (Q) in the form of a transistor connected in series between the terminal; a pulse generator circuit having an output connected to the control electrode of the transistor; a second terminal of the inductor (L); A transformer having a primary winding connected between the positive electrode of the voltage source and a first and a second parallel branch, respectively, across the charge emitting and charge receiving electrodes of the transistor (Q). The first capacitor (C ₁ ) and the diode (D ₂ ) are connected to each other, and the inductor (L) is connected in series with the inductor (L) through the diode (D ₂ ) by being connected between the terminals of the voltage power supply. Connected to the Has a capacitor (C _3), the respective components, the second capacitor (C ₃₎ is dimensioned to have a several times greater value of the first capacitor (C _1), the The pulse generator is characterized in that it is adapted to turn the transistor on and off for the following periods and intervals: In a first operating phase, the transistor (Q) is turned on and the current is Flows in series through the primary winding (P) of the transformer, the inductor (L) and the transistor (Q), and in a second phase of operation, the transistor is turned off and current flows through the transformer. By continuing to flow in series through the primary winding (P) and the inductor (L), the first
Charging a capacitor (C _1), said first capacitor (C ₁₎ and the inductor (L) acts as a resonant circuit to the voltage source in series, in a third phase of operation, the transistor is off As the first capacitor (C ₁ ) discharges, a current flows in the opposite direction through the inductor (L) and the primary winding (P) of the transformer. Charging the capacitor (C ₃ ), wherein the second capacitor (C ₁ ) and the inductor (L) act as a resonant circuit in parallel with the voltage power supply; After the capacitor (C ₁ ) discharges, the current continues to flow in the reverse direction through the diode (D ₂ ) through the primary winding of the transformer and the inductor, so that the second capacitor (C 1) ₃ ) is charged, this fourth operating phase In which the transistor (Q) is turned on again and current flows from the voltage source in a first direction and from the charged second capacitor (C ₃ ) to the primary winding of the transformer and the The first operating phase of the circuit is resumed by allowing it to flow through the inductor as well as the transistor, the load (R _g ) being connected to the secondary winding (S ₁ ) of the transformer (T). The resonance mode power converter connected between the terminals of. 2. The transistor according to claim 1, wherein said transistor (Q) is a MOS field-effect transistor with an integrated free-running diode, whereby the rectifier diode (D ₂ ) in said second parallel branch is omitted. The resonance mode power converter according to claim 1, wherein 3. The resonant mode power converter according to claim 1, wherein said pulse generator circuit includes an astable multivibrator. 4. The resonant mode power converter according to claim 1, wherein said pulse generator circuit includes a digital frequency synthesizer. 5. The impedance of the secondary winding (S ₁ ) of the transformer (T) is adapted to a characteristic factor of the resonator at a nominal load (R _G ). 3. The resonance mode power converter according to item 1. When the instantaneous value of wherein said load (R _G) is less than the nominal load, characterized in that the load (R _G) is balanced by the compensation load, the resonant modes of Claim 5 Power converter. 7. A rectifying bridge (B ₁ ) is provided between the terminals of a second secondary winding (S ₂ ) of said transformer to balance said load (R _G ). T) the primary winding (P)
Wherein the rectifier bridge between the electrodes of the voltage source so that the impedance of lower than a predetermined maximum value (B ₁₎ is connected, resonant mode power converter according to claim 6. 8. The method according to claim 1, wherein said capacitor (C ₁ ) limits a voltage of a counter-induction pulse generated by said inductor (L) through said transistor (Q).
The resonance mode power converter according to claim 1. 9. In order to drive a hot cathode gas discharge tube, a terminal of said first secondary winding (S ₁ ) is connected to a capacitor (C ₁ ) via electrodes (K ₁ , K ₂ ) of said gas discharge tube. ₆ ), the secondary winding (S ₁ ) and the capacitor (C ₆ ) are connected to the resonance frequency of the transformer (T) when the electrodes (K ₁ , K ₂ ) are heated. The resonant mode power converter according to claim 1, wherein the converter is adapted to: 10. An electrode operating as said cathode (K ₁ , K ₂ ) for driving a hot cathode gas discharge tube, further comprising:
The resonant mode power converter according to claim 9, characterized in that it is connected to one of the terminals of a _third secondary winding (S3) of the transformer (T).