JP3536809B2 - High frequency circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a low noise block of a direct broadcasting satellite receiver system, and more particularly to a gallium arsenide (GaAs) monolithic microwave integrated circuit (MM).
IC) using a single balanced frequency downconverter for low noise blocks.

[0002]

2. Description of the Related Art A frequency downconverter for a low-noise block of a direct broadcasting satellite receiver system has a function of converting a microwave signal into an intermediate frequency suitable for processing by an indoor tuner directly under an antenna. DBS) widely used in television receivers. The low-noise block is attached to an antenna or a parabolic reflector for satellite reception, a so-called parabolic antenna, and is housed in a weatherproof housing. The output is connected to an indoor tuner attached to a television or video cassette recorder (VCR) using a cable. The low noise block generally includes a frequency downconverter mounted on a printed circuit board, with auxiliary circuits, additional amplification stages, and filters to obtain increased amplitude and reduced front-end noise. The frequency downconverter for the low noise block of the direct broadcasting satellite receiver system is 11 to 12GH
1000 to 2000MHz microwave signal with frequency in z range
To an intermediate frequency (IF) signal with a frequency in the range With the rapid development of gallium arsenide monolithic microwave integrated circuit technology, commercial gallium arsenide monolithic microwave integrated circuit frequency downconverters have become very feasible for low noise block applications.

Currently, several companies offer devices made with gallium arsenide technology to participate in the development of a large consumer market for direct broadcast satellite receivers. However, for commercial gallium arsenide monolithic microwave integrated circuit frequency downconverters, cost reduction is important.
The cost can be reduced by reducing the dice area by reducing the number of circuits and mounting the device in a low-cost surface-mount type package by an effective circuit configuration. Gate-pumped active field-effect transistor (FET) mixers have the advantage of reducing die area by reducing circuitry,
Conventionally, a gate pump type active field effect transistor (FE
T) Mixers are widely used. Gate pump mixers have the advantage of a lower noise figure and higher conversion gain than other types of mixers in principle. Based on the low noise performance of this device, the use of a gate-pumped field-effect transistor mixer combined with an intermediate frequency amplification circuit allows the preamplifier and image integrated in the frequency downconverter of a gallium arsenide monolithic microwave integrated circuit. Comparable gain and noise figure are obtained without using a frequency rejection filter. In terms of mounting in a low-cost surface-mount package with an effective circuit configuration, a single balance mixer has been widely used. With a single balance mixer, the balanced microwave and intermediate frequency signals provide a virtual electrical ground on the surface of the die, thus reducing the parasitic ground inductance typically encountered in low cost surface mount plastic packages. On the contrary, it becomes very insensitive.

FIG. 7 shows a conventional frequency downconverter circuit having a gate pump type single balance mixer proposed in Japanese Patent Publication No. 2938392 (publication number: Japanese Patent Application Laid-Open No. 9-107243).

[0005] The microwave signal and the local oscillator signal are connected to the terminal 10 of the 180-degree hybrid coupler 100 on the printed circuit board, respectively.
When input to 1 and 102, a combined signal of the microwave signal and the local oscillator signal is output from terminals 103 and 104. Due to the characteristics of the 180-degree hybrid coupler 100, the microwave signal in the first synthesized signal output to the terminal 103 is
A microwave signal in the second composite signal output to 4;
The local oscillator signal in the first composite signal output to the terminal 103 has the same amplitude and the same amplitude as the local oscillator signal in the second composite signal It has a phase. The gate pump single balance mixer 200 includes a terminal 201 for receiving the first synthesized signal output from the terminal 103, a terminal 202 for receiving the second synthesized signal output to the terminal 104, and an intermediate frequency signal, that is, a microwave signal. 20 that outputs the frequency signal of the difference between
3 and 204, and have drain resistors 212 and 213. The DC bias between the gate and the source of the field effect transistor is adjusted by the gate resistors 214 and 215 and the source resistor 216, and the DC bias between the drain and the source of the field effect transistor is adjusted by the drain resistors 212 and 213 and the source resistor 216. Is done. Due to the signal configuration of the microwave signal and the local oscillator signal, the intermediate frequency signal output from the terminal 203 has the same amplitude and a 180 degree phase difference as the intermediate frequency signal output from the terminal 204. In order to suppress the leakage of the microwave signal and the local oscillator signal at the terminals 203 and 204, a low-pass filter including an inductor 206 and a capacitor 207 is connected to the drain 205 of the field-effect transistor of the gate pump single balance mixer. The drain 208 is connected to a low-pass filter including an inductor 209 and a capacitor 210. The intermediate frequency signal output from terminals 203 and 204 and the inverted intermediate frequency signal are input to terminals 301 and 302 of active balun 300 through DC blocking capacitors 10000 and 10001, respectively. The active balun 300 receives a balanced intermediate frequency signal including an intermediate frequency signal and an inverted intermediate frequency signal, and outputs an unbalanced intermediate frequency signal to a terminal 303. The active balun 300 is configured by connecting the source 305 of the field effect transistor 304 and the drain 307 of the field effect transistor 306. The DC bias of the field effect transistors 304 and 306 is adjusted by the gate resistors 309, 310 and 311 and the source resistor 312. The source resistor 312 is short-circuited by a bypass capacitor 313 at a high frequency. The unbalanced intermediate frequency signal output from the terminal 303 at a high frequency passes through a DC blocking capacitor 10002, a single-ended amplifier 400 having a source resistor 401 and a drain resistor 402, a source resistor 501, and a drain resistor 50.
The signal is amplified by a single-ended amplifier 500 having 2 and finally output from a terminal 503. As described above, the DC bias between the gate and the source of the field-effect transistor of the single-ended amplifier 400 is equal to the gate resistance 40.
The DC bias between the drain and the source is adjusted by the drain resistance 402 and the source resistance 401. The source resistance 401 is short-circuited by a bypass capacitor 404 at a high frequency. The capacitor 10003 is a DC blocking capacitor. Single-ended amplifier 500
Is the same as that of the single-ended amplifier 400.
DC power on the drain side of the field effect transistor of each circuit is supplied from a power supply 600.

The use of a frequency downconverter circuit having a gate-pumped single-balanced mixer as shown in FIG. 7 provides comparable gain and noise figure without using a radio frequency preamplifier and image rejection filter. Also, when processing high frequency signals, the performance of the frequency downconverter circuit becomes very insensitive to the parasitic ground inductance typically encountered in low cost surface mount plastic packages. The desensitization to the parasitic ground inductance can be explained as follows. Terminal corresponding to the gate input of the gate pump type single balance mixer
By inputting a microwave signal (balanced microwave signal) having the same amplitude and a phase difference of 180 degrees to the terminal 201 and the terminal 202, the balanced intermediate frequency signal is converted to the terminal
Output to 203 and terminal 204. The balanced microwave and intermediate frequency signals provide a virtual electrical ground to the surface of the semiconductor chip for a monolithic microwave integrated circuit, i.e., the junction 211 between the sources of the two field effect transistors of the single balance mixer. Therefore, the performance of the frequency downconverter circuit becomes very insensitive to the parasitic ground inductance generated in the package. Therefore, when a frequency downconverter circuit having a single balance mixer is used, a semiconductor chip for a monolithic microwave integrated circuit that processes a signal having a high frequency, such as a frequency downconverter for direct broadcast satellite reception, is incorporated in a low-cost surface-mount package. And an acceptable level of performance can be obtained. The use of a gate pump mixer as shown in FIG. 7 has the advantage that the noise figure is smaller and the conversion gain is larger than other types of mixers in principle.

[0007]

In spite of the above advantages, the conventional frequency downconverter having the gate pump mixer has the following problems to be solved.

(1) As described above, in order to drive a frequency down-converter circuit having a single balance mixer, it is necessary to input a balanced signal. Therefore, it is necessary to input microwave signals (balanced microwave signals) having the same amplitude and a phase difference of 180 degrees to the terminals 201 and 202 corresponding to the gate inputs to the single balance mixer as shown in FIG. There is. Also, terminal 203
And the balanced intermediate frequency signal is output to the active balun terminals 301 and 302, so that the two balanced microwave signals are combined with the local oscillator signal having the same amplitude and phase. Input to terminals 201 and 202. That is, the microwave signal in the first composite signal input to the terminal 201 has the same amplitude and a 180 degree phase difference as the microwave signal in the second composite signal input to the terminal 202, and The local oscillator signal in the first composite signal needs to have the same amplitude and the same phase as the local oscillator signal in the second composite signal input to the terminal 202. In addition, insulation is required between the terminal 101 for inputting a microwave signal and the terminal 102 for inputting a local oscillator signal so that signals do not leak from each other. Accordingly, a 180-degree hybrid coupler is required on the printed circuit board as shown in FIG. 7 due to the above-described signal synthesis and insulation between the microwave signal and the local oscillator signal. However, such a 180-degree hybrid coupler generally occupies a much larger area than a semiconductor chip for a monolithic microwave integrated circuit, and therefore cannot be mounted inside a semiconductor chip.
Therefore, since the semiconductor device is mounted on a printed circuit board on which a semiconductor chip for a monolithic microwave integrated circuit is mounted, the size of the printed circuit board is large, the layout on the printed circuit board is complicated, and the manufacturing cost is high. Become.

(2) Since balanced intermediate frequency signals are output to terminals 203 and 204, a balanced-unbalanced conversion circuit is required to convert these balanced signals into unbalanced signals. The above-mentioned 180-degree hybrid coupler on a printed circuit board is often used as a balanced-unbalanced conversion circuit. However, as described above, the size of the printed circuit board is increased, and the problem of layout complexity on the printed circuit board appears. . In particular, the intermediate frequency band hybrid coupler
As shown in FIG. 7, the size is significantly larger than the microwave-band hybrid coupler located at the input stage of the single balance mixer. Therefore, a balanced-unbalanced conversion circuit that can be integrated inside a semiconductor chip for a monolithic microwave integrated circuit, that is, an active balun is required. An active balun 300 in which a source 305 of a first field-effect transistor 304 and a drain 307 of a second field-effect transistor 306 are connected as shown in FIG.
It is. However, the active balun 300 as shown in FIG.
Are connected to the source 305 of the first field-effect transistor 304 and the second
The drain 307 of the field effect transistor 306 is connected,
Since the output is taken out from the connection point, there is no gain.
Therefore, additional single-ended amplifiers 400 and 500 are required as shown in FIG. Further, the conventional active balun 300 as shown in FIG.
Is connected to the drain 307 of the second field-effect transistor 306.
A virtual electric ground point such as a connection point 211 between the sources of the two field effect transistors of the single balance mixer does not occur. Therefore, the active balun 300 having the conventional circuit configuration is easily affected by the parasitic ground inductance. That is, it becomes sensitive to the parasitic ground inductance.

(3) In a high-frequency communication system such as direct broadcast satellite reception, one of the important factors that determines the performance of a system is distortion characteristics. As is well known, the distortion characteristic is determined by the current value flowing between the drain and source of the field effect transistor of the final stage amplifier of the frequency down converter circuit and the voltage value applied between the drain and source of the field effect transistor of the final stage amplifier. Strongly depends on. That is,
As the current flowing between the drain and source of the field effect transistor of the final stage amplifier increases, and the voltage applied between the drain and source of the field effect transistor of the final stage amplifier increases, the distortion increases. The characteristics are better. Conventionally, the value of the source resistance 501 of the final-stage single-ended amplifier 500 is reduced, and the amount of current flowing between the drain and the source of the field-effect transistor of the final-stage single-ended amplifier 500 is increased. Strain characteristics were obtained. However, according to this method, there is a problem that the current value flowing through the single-ended amplifier 500 at the final stage becomes large and the total current consumption value becomes large. In particular, in the receiving system, since the power supply is always on, a large current consumption is a problem. (1)
For details of the conventional circuit configuration having the problems (3) to (3), see Japanese Patent Publication No. 2938392 (Publication No. 9-107243).
It is described in.

For the above-mentioned reason (1), the input stage of the frequency downconverter circuit can be integrated inside the semiconductor chip for a monolithic microwave integrated circuit,
A new circuit configuration having the same function as a 180-degree hybrid coupler on a printed circuit board is required. For the reason (2) described above, an active balun having a new configuration that has a gain and is insensitive to parasitic ground inductance is required between the single balance mixer and the single-ended amplifier. In addition, for the reason (3) described above, a new circuit configuration is required for the highest-stage amplifier of the frequency down-converter circuit to obtain a good distortion characteristic with a low current consumption value.

Accordingly, the present invention provides a circuit item belonging to the frequency downconverter, in which the input stage is provided on the printed circuit board.
Functionally similar to a 180-degree hybrid coupler,
In addition, a single balance mixer with a new circuit configuration that can be integrated inside a semiconductor chip for a monolithic microwave integrated circuit, an active balun that has gain and is insensitive to parasitic ground inductance, and good distortion characteristics with low current consumption are obtained. It is intended to realize the highest stage amplifier that can be used.

[0013]

The high frequency circuit according to the present invention comprises:
An active balun formed by connecting a source of the first field-effect transistor and a source of the second field-effect transistor;
A first resistor connected between the rain and the voltage supply means;
The drain and the gate of the first field effect transistor
Between the second resistor and the second resistor in series.
A feedback circuit having a first capacitor is connected;
Drain of a second field effect transistor and said voltage supply
A third resistor connected in parallel with the third resistor between the means;
A second high-frequency input signal is input to the gate of the first field-effect transistor, and has the same amplitude as the first high-frequency input signal;
A second high frequency input signal having a phase difference of 0 degrees
Is input to the gate of the field effect transistor.

The present invention eliminates the need for a 180-degree hybrid coupler on a printed circuit board, thus providing the advantages of greatly reducing the area of the printed circuit board and simplifying the layout of the printed circuit board.

[0015]

FIG. 1 shows a frequency downconverter according to an embodiment of the present invention for solving the problems (1) to (3). First, three types of mixer circuits for solving the problem (1) will be described.
Then, two types of active baluns for solving the problem (2) and two types of amplifiers for solving the problem (3) will be described in detail. As shown in FIG. 1, a differential input amplifier 100a at the front stage of the single balance mixer and a gate pump type dual gate single balance mixer 20 are provided.
The problem (1) can be solved by using a mixer circuit composed of 0a. A terminal 10 corresponding to the gate input of the first field effect transistor 101a of the differential input amplifier 100a of FIG.
When a microwave signal is input to 2a and a terminal 104a corresponding to the gate input of the second field-effect transistor 103a is connected to the ground point, the characteristics of the first field-effect transistor 101a are reduced due to the characteristics of a typical differential input amplifier. Terminal 10 connected to the drain
Two microwave signals having the same amplitude and a phase difference of 180 degrees are output to 5a and a terminal 106a connected to the drain of the second field-effect transistor 103a, respectively. First field effect transistor 101a and second field effect transistor 10
DC bias of 3a is drain resistance 107a and 108a and source resistance
Adjusted by 109a. Preferably, the value of the resistor 107a needs to be the same as the value of 108a in order to increase the degree of balance of the output signal. The first field-effect transistor 101a and the second
Because of the DC bias of the field-effect transistor 103a,
Gate resistors may be inserted into 102a and 104a. Instead of the resistor 109a, a field-effect transistor or a current mirror including a field-effect transistor may be used. One of the microwave output signals is a DC blocking capacitor 1000.
0a through dual gate single balance mixer 20
0a is input to a terminal 202a corresponding to a first gate input of a first dual-gate field effect transistor 201a;
The other one of the microwave signals passes through a DC blocking capacitor 10001a and passes through a second dual gate field effect transistor of a dual gate single balance mixer 200a.
The signal is input to a terminal 204a corresponding to the first gate input of 203a. Also, two local oscillator signals having the same amplitude and phase are
The second of the first dual-gate field-effect transistor 201a of the dual-gate single balance mixer 200a, respectively.
And a terminal 206a corresponding to the second gate input of the second dual-gate field-effect transistor 203a of the dual-gate single balance mixer 200a. The DC bias of the first dual-gate field-effect transistor 201a and the second dual-gate field-effect transistor 203a is equal to the drain resistances 215a and 216a,
It is adjusted by gate resistances 217a and 218a and source resistance 219a. The first dual-gate field-effect transistor 201a and the second dual-gate field-effect transistor 203a
Each one operates on the principle of a typical dual gate mixer. Therefore, the first dual gate field effect transistor of the dual gate single balance mixer 200a
201a and second dual-gate field-effect transistor 203a
Output an intermediate frequency signal having a frequency equal to the difference between the frequency of the microwave signal and the frequency of the local oscillator signal to terminals 207a and 208a by the function of a typical dual gate mixer. Capacitors 10002a and 10003a are DC blocking capacitors. The basic operating principle of a mixer composed of field-effect transistors with dual gates is described in Step
"Microwave Mixers" by henA. Maas, 2nd edition, Artec
h House, Inc., 9.2, 1993, pages 328-333. The circuit including the inductor 209a, the capacitor 210a, and the ground 211a is a low-pass filter for blocking a high-frequency leak signal. That is, the microwave signal and the local oscillator signal leaking to the drain of the first dual-gate field-effect transistor 201a are cut off by the low-pass filter, and only the intermediate frequency signal is output to the terminal 207a. When a low-pass filter is manufactured on a printed circuit board, the area of the printed circuit board becomes large. Therefore, the low-pass filter should be integrated inside a semiconductor chip for a monolithic microwave integrated circuit as shown in FIG.

However, in order to suppress the deterioration of the performance of the low-pass filter due to the parasitic ground inductor, preferably, the ground 211a formed as a conductor ground pad on the semiconductor chip for a monolithic microwave integrated circuit is connected to the monolithic microwave integrated circuit. It should be connected to a conductor ground plane formed on the back surface of the monolithic microwave integrated circuit semiconductor chip by a conductor layer penetrating the circuit semiconductor chip from the front surface to the back surface. FIG. 2 is a sectional view showing the connection between the ground 211a and the conductor ground plane on the back surface of the semiconductor chip for a monolithic microwave integrated circuit. As shown in FIG. 2, a ground 211a formed as a conductor ground pad on the monolithic microwave integrated circuit semiconductor chip 2000 is connected to a conductor layer 2001 penetrating the monolithic microwave integrated circuit semiconductor chip 2000.
Is electrically connected to the conductor ground plane 2002, which is effective in suppressing the performance deterioration of the low-pass filter due to the parasitic ground inductor at a high frequency. The circuit composed of the inductor 212a, the capacitor 213a, and the ground 214a is also a low-pass filter having the same function as the above-described low-pass filter for blocking a high-frequency leak signal, and preferably, the ground 214a is also used for a monolithic microwave integrated circuit. The conductor layer should penetrate the semiconductor chip 2000 and be connected to the conductor ground plane 2002. The active balun 300a is a field effect transistor 30
An active balun formed by connecting the source of the field effect transistor 301a to the source of the field effect transistor 302a at a connection point 309a.
303a is connected, resistance 303a and field effect transistor
The drain of 302a is connected to the power supply 600a, a feedback circuit including a resistor 304a and a capacitor 305a is connected between the gate and the drain of the field effect transistor 301a, and the intermediate frequency signal output from the terminal 207a is supplied to the field effect transistor 301a. The intermediate frequency signal input to the terminal 306a corresponding to the gate input and output from the terminal 208a is applied to the field effect transistor 302a.
Input to a terminal 307a corresponding to the gate input of the terminal 308a connected to the drain of the field-effect transistor 301a.
This is an active balun that extracts an output signal from the balun. Active balun 300a and field effect transistors 301a and 302a
DC bias is applied to the drain resistor 303a and the gate resistors 310a and 31.
It is adjusted by 1a and source resistance 312a. Capacitor 1000
4a is a DC blocking capacitor. Terminal as described above
Since the intermediate frequency signal output from 207a and the intermediate frequency signal output from terminal 208a have the same amplitude and a phase difference of 180 degrees, active balun 300a is driven by the balanced intermediate frequency input signal. Amplifier 500a is an inductor
A distortion improvement circuit consisting of 501a and a resistor 502a is connected between the drain of the field effect transistor 503a and the power supply 600a,
Further, a resistor 504a connected in parallel with the distortion improving circuit is connected between the drain of the field effect transistor 503a and the power supply 600a, and the source of the field effect transistor 503a has a resistor 50a.
5a is an amplifier to which an output is extracted from a terminal 506a connected to the drain of the field effect transistor 503a. The DC bias of the field effect transistor 503a of the amplifier 500a is adjusted by the drain resistors 504a and 502a, the gate resistor 507a, and the source resistor 505a. The source resistor 505a is short-circuited at a high frequency by the bypass capacitor 508a. The capacitor 10005a is a DC blocking capacitor.

Due to the function of the differential input amplifier 100a, the microwave signal input to the terminal 202a corresponding to the first gate input of the first dual gate field effect transistor 201a of the dual gate single balance mixer 200a and the second
The microwave signals input to the terminal 204a corresponding to the first gate input of the dual-gate field-effect transistor 203a have the same amplitude and a different phase by 180 degrees. Therefore, the terminal 207a of the dual gate single balance mixer 200a
And the intermediate frequency signal output to the terminal 208a are balanced, that is, the amplitudes are equal and the phases are different by 180 degrees. Due to the characteristics of a typical dual gate mixer, the terminal 20 corresponding to the first gate input of the first dual gate field effect transistor 201a of the dual gate single balance mixer 200a
The microwave signal input to 2a and the local oscillator signal input to terminal 205a corresponding to the second gate input have insulation from each other. The microwave signal input to the terminal 204a corresponding to the first gate input of the second dual-gate field-effect transistor 203a of the dual-gate single balance mixer 200a and the microwave signal input to the terminal 206a corresponding to the second gate input. The local oscillator signals are isolated from each other. Therefore, as shown in FIG. 1, when a mixer circuit including the differential input amplifier 100a in the preceding stage of the single balance mixer and the gate pump type dual gate single balance mixer 200a is used, the 180-degree hybrid coupler on the printed circuit board is used. Without using, the signal configuration of the microwave and the local oscillator input signal at the gate input terminal of the dual gate single balance mixer 200a can be the same signal configuration as the 180-degree hybrid coupler, and the dual gate single balance mixer 20a
Insulation is maintained between the terminal 102a for inputting the microwave input signal 0a and the terminal 205a or 206a for inputting the local oscillator. Therefore, by using the new circuit configuration,
The 180-degree hybrid coupler occupying a large area on the printed circuit board is not required, and the problem (1) can be solved. That is, an attractive feature of the novel circuit configuration of FIG. 1 compared to the conventional circuit configuration of FIG. 7 is that a circuit having the same function using an active circuit without using a 180-degree hybrid coupler on a printed circuit board. And it can be integrated inside a semiconductor chip for a monolithic microwave integrated circuit. Therefore, the use of the novel circuit configuration can significantly reduce the area of the printed circuit board and simplify the layout of the printed circuit board. This leads directly to cost savings.

FIG. 3 shows a mixer circuit according to another embodiment of the present invention for solving the above-mentioned problem (1). As shown in FIG. 3, even if the dual-gate field-effect transistors 201a and 203a of the dual-gate single-balance mixer 200a of FIG. 1 are replaced with two single-gate field-effect transistors each having a source and a drain connected, in principle, It has the same function as the circuit of FIG. 1,
The above advantage is obtained, and the above-mentioned problem (1) can be solved.

Due to the function of the differential input amplifier 100b described above, one of the two microwave signals having the same amplitude and 180 degrees different in phase corresponds to the gate input of the first field effect transistor 201b of the single balance mixer 200b. Terminal
The other microwave signal input to the second field effect transistor 20b of the single balance mixer 200b
The signal is input to the terminal 204b corresponding to the gate input of 3b. or,
Two local oscillator signals having the same amplitude and phase are respectively supplied to the terminal 206b corresponding to the gate input of the third field-effect transistor 205b and the terminal 208b corresponding to the gate input of the fourth field-effect transistor 207b of the single balance mixer 200b.
Is input to The first field effect transistor 201b and the third field effect transistor 205b of the single balance mixer 200b are functionally identical to the first dual gate field effect transistor 201a of the dual gate single balance mixer 200a of FIG. Further, the second field-effect transistor 203b and the fourth field-effect transistor 207b
Is the dual gate single balance mixer 20 of FIG.
It is completely identical in function to the second dual-gate field-effect transistor 203a of FIG. Therefore, as in the dual gate single balance mixer of FIG. 1, an intermediate frequency signal having a frequency equal to the difference between the frequency of the microwave signal and the frequency of the local oscillator signal is output to terminals 209b and 210b. The basic operating principle of a mixer in which the source of one field-effect transistor is connected to the drain of another field-effect transistor as in the single balance mixer 200b of FIG. 3 is described in “Microwave Mixers” by Stephen A. Maas,
Edition, Artech House, Inc., Section 9.2 of 1993, page 328
-333. Also, the intermediate frequency signals output to the terminals 209b and 210b of the single balance mixer 200b are balanced, that is, have the same amplitude and a phase difference of 180 degrees. As in the dual gate single balance mixer of FIG. 1, the microwave signal input to the terminal 202b corresponding to the gate input of the first field effect transistor 201b of the single balance mixer 200b and the third field effect transistor
Local oscillator signals input to a terminal 206b corresponding to the gate input of 205b have insulation properties from each other. Also, the second field-effect transistor 203b of the single balance mixer 200b
The microwave signal input to the terminal 204b corresponding to the gate input of the fourth field-effect transistor 207b and the local oscillator signal input to the terminal 208b corresponding to the gate input of the fourth field-effect transistor 207b have insulating properties. Therefore, as shown in FIG. 3, when a mixer circuit including the differential input amplifier 100b at the preceding stage of the single balance mixer and the single balance mixer 200b is used, the single-stage mixer can be used without using the 180-degree hybrid coupler on the printed circuit board. Balance mixer 200b
The signal configuration of the microwave and local oscillator input signal at the gate input terminal of the same can be the same signal configuration as the 180-degree hybrid coupler, and input the microwave input signal of the single balance mixer 200b terminal 102b and the local oscillator Insulation is maintained between the terminals 206b and 208b. Therefore, by using the circuit configuration of FIG.
The 180-degree hybrid coupler occupying a large area on the printed circuit board is not required, and the problem (1) can be solved. That is, the attractive feature of the novel circuit configuration of FIG. 3 compared to the conventional circuit configuration of FIG. 7 is that it has the same function using an active circuit without using a 180-degree hybrid coupler on a printed circuit board. Configure the circuit,
This is advantageous in that it can be integrated inside a semiconductor chip for a monolithic microwave integrated circuit. Therefore, when the novel circuit configuration of FIG. 3 is used, the area of the printed circuit board can be significantly reduced, and the layout of the printed circuit board is simplified. This leads directly to cost savings.

FIG. 4 shows a mixer circuit according to another embodiment of the present invention for solving the above-mentioned problem (1). Further, even if a new circuit configuration as shown in FIG. 4 is used, the above advantage can be obtained, and the above-mentioned problem (1) can be solved. Due to the function of the differential input amplifier 100c described above, one of the two microwave signals having the same amplitude and 180 degrees different in phase is connected to the terminal 202c corresponding to the gate input of the first field effect transistor 201c of the single balance mixer 200c. The other input microwave signal is input to a terminal 204c corresponding to the gate input of the second field effect transistor 203c of the single balance mixer 200c. The local oscillator signal is input to a terminal 206c corresponding to the gate input of the third field effect transistor 205c of the single balance mixer 200c. First field effect transistor 201c and third field effect transistor of single balance mixer 200c
205c is the first dual gate field effect transistor 20 of the dual gate single balance mixer 200a of FIG.
1a, and the second field effect transistor 203c and the third field effect transistor 205c are functionally equivalent to FIG.
Of the dual gate single balance mixer 200a
Is functionally equivalent to the dual gate field effect transistor 203a of FIG. The difference is that in the single balance mixer 200c of FIG. 4, the local oscillator signal is commonly input to the terminal 206c corresponding to the gate input of the third field effect transistor 205c, and the first field effect transistor 201c and the second field effect transistor This is to drive the transistors 203c in common. The basic operating principle of the single balance mixer 200c is
n “Microwave Mixers” by A. Maas, 2nd edition, Artech
House, Inc., Sec. 10.3, 1993, pp. 354-367. Therefore, as in the dual gate single balance mixer 200a of FIG. 1, an intermediate frequency signal having a frequency equal to the difference between the frequency of the microwave signal and the frequency of the local oscillator signal is output to the terminals 207c and 208c. The intermediate frequency signals output to the terminals 207c and 208c of the single balance mixer 200c are balanced, that is, have the same amplitude and 180 degrees different phases. As in the dual gate single balance mixer 200a of FIG. 1, the microwave signal input to the terminal 202c corresponding to the gate input of the first field effect transistor 201c of the single balance mixer 200c and the third
The local oscillator signals input to the terminal 206c corresponding to the gate input of the field effect transistor 205c have insulating properties. A terminal 20 corresponding to the gate input of the second field effect transistor 203c of the single balance mixer 200c.
The microwave signal input to 4c and the local oscillator signal input to the terminal 206c corresponding to the gate input of the third field-effect transistor 205c have insulation from each other. Therefore,
As shown in FIG. 4, the differential input amplifier 100c at the preceding stage of the single balance mixer 200c and the single balance mixer 200c
Using a mixer circuit consisting of
Without using the 180-degree hybrid coupler, the signal configuration of the microwave and local oscillator input signal at the gate input terminal of the single balance mixer 200c can be the same as that of the 180-degree hybrid coupler, and the single balance mixer 200c Terminal 102c for inputting microwave input signal and terminal for inputting local oscillator
Insulation is maintained between 206c. Therefore, by using the circuit configuration of FIG. 4, the 180-degree hybrid coupler occupying a large area on the printed circuit board is not required, and the above-mentioned problem (1) can be solved. That is, an attractive feature of the novel circuit configuration of FIG. 4 compared to the conventional circuit configuration of FIG. 7 is that the active circuit has the same function without using the 180-degree hybrid coupler on the printed circuit board. An advantage is that a circuit can be formed and integrated inside a semiconductor chip for a monolithic microwave integrated circuit.
Therefore, when the novel circuit configuration of FIG. 4 is used, the area of the printed circuit board can be significantly reduced, and the layout of the printed circuit board is simplified. This leads directly to cost savings.

FIG. 1 shows an active balun according to an embodiment of the present invention for solving the above-mentioned problem (2). The problem (2) can be solved by using the active balun 300a in which the sources of two field effect transistors are connected to each other as shown in FIG. Active balun 3
00a is an active balun formed by connecting the source of the field-effect transistor 301a and the source of the field-effect transistor 302a at a connection point 309a, and a resistor 303a is connected between the drain of the field-effect transistor 301a and the power supply 600a. The drain of the field effect transistor 302a is connected to a power supply 600a, a feedback circuit including a resistor 304a and a capacitor 305a is connected between the gate and the drain of the field effect transistor 301a, and an intermediate frequency signal output from a terminal 207a. Is input to a terminal 306a corresponding to the gate input of the field effect transistor 301a, the intermediate frequency signal output from the terminal 208a is input to a terminal 307a corresponding to the gate input of the field effect transistor 302a,
Is an active balun for extracting an output signal from the terminal 308a connected to the drain of the balun. Terminal 20 as described above
Since the intermediate frequency signal output from the terminal 7a and the intermediate frequency signal output from the terminal 208a have the same amplitude and a phase difference of 180 degrees, the active balun 300a is driven by the balanced intermediate frequency input signal. In order to improve the frequency characteristics of the conversion gain, a feedback circuit including a resistor 304a and a capacitor 305a is connected between the gate and the drain of the field effect transistor 301a. The active balun 300a of FIG. 1 has a configuration in which the source of the field effect transistor 301a and the source of the field effect transistor 302a are connected, and is clearly different from the conventional active balun 300 of FIG. In order to take out the output from the drain of the field effect transistor 301a, a resistor 303a serving as a load is connected to the drain side of the field effect transistor 301a. However, if a resistor is connected to the drain side of the field effect transistor 302a that does not take out an output, a loss of gain occurs. Therefore, preferably, no resistor is connected to the drain side of the field effect transistor 302a.
Terminal corresponding to the gate input of field effect transistor 301a
306a has dual gate single balance mixer 200a
The intermediate frequency signal output from the terminal 207a of
The terminal 307a corresponding to the gate input of the field effect transistor 302a is connected to a dual gate single balance mixer.
The intermediate frequency signal output from terminal 208a of 200a is input. The two intermediate frequency signals output from the outputs 207a and 208a of the dual gate single balance mixer 200a are balanced, that is, have the same amplitude and a 180 degree phase difference. The sources of the field effect transistors 301a and 302a are connected to the connection point 30.
Connected at 9a, and two balanced intermediate frequency signals are input to terminals 306a and 307a corresponding to the gate input,
Active balun 300a operates on the principle of a typical differential input amplifier. Therefore, the active balun 300a has a gain, and a virtual electric ground point is generated at the connection point 309a. Since the active balun 300a of FIG. 1 has a gain,
As compared with the circuit configuration using the conventional active balun 300 of FIG. 7, an additional amplifier 400 is not required. Further, since a virtual electric ground point is supplied to the connection point 309a of the active balun 300a in FIG. 1, the performance of the active balun 300a together with the differential input amplifier 100a and the dual-gate single balance mixer 200a is reduced by the parasitic ground inductance. On the contrary, it becomes insensitive. Therefore, the conventional active balun 300 of FIG.
An attractive feature of the active balun 300a of the novel circuit configuration of FIG. 1 is that the sources of the two field-effect transistors are connected to each other and driven by a balanced input signal, thereby having a gain, Further, it is insensitive to parasitic ground inductance. Thereby, compared to the circuit configuration using the conventional active balun 300 of FIG.
The additional amplifier 400 is no longer needed, and the semiconductor chip for a monolithic microwave integrated circuit can be incorporated into a low-cost surface-mount plastic package to obtain an acceptable level of performance, thereby solving the above-mentioned problem (2). .

FIG. 5 shows an active balun according to another embodiment of the present invention for solving the above-mentioned problem (2). By using the active balun 300b of FIG. 5, the above-mentioned problem (2) is solved, and the performance is further improved as compared with the active balun 300a of FIG. The active balun 300b is an active balun formed by connecting the source of the field-effect transistor 301b and the source of the field-effect transistor 302b at a connection point 309b,
A resistor 303b is connected between the drain of the field effect transistor 301b and the power supply 600b, a resistor 310b and a capacitor 311b are connected between the drain of the field effect transistor 302b and the power supply 600b, and the gate and the drain of the field effect transistor 301b are connected. A feedback circuit consisting of a resistor 304b and a capacitor 305b is connected between them. The intermediate frequency signal output from the terminal 207a in FIG. 1 is input to a terminal 306b corresponding to the gate input of the field effect transistor 301b, and the terminal 208a in FIG. Is output to a terminal 307b corresponding to the gate input of the field effect transistor 302b.
This is an active balun that extracts an output signal from a terminal 308b connected to the drain of 1b. As described above, FIG.
Since the intermediate frequency signal output from the terminal 207a and the intermediate frequency signal output from the terminal 208a have the same amplitude and a phase difference of 180 degrees, the active balun 300b in FIG. Driven by As shown in FIG. 5, when the resistor 310b is connected to the drain of the field-effect transistor 302b and the capacitor 311b is connected in parallel with the resistor 310b, the degree of balance is higher than that of the active balun 300a of FIG. Become insensitive. It can be explained as follows. In order for the electric field effect transistor 301b and the electric field effect transistor 302b to operate symmetrically with respect to the balanced intermediate frequency input signal at a high frequency, a virtual electric ground point is generated at the connection point 309b. And the small-signal equivalent circuit of the field-effect transistor 302b at high frequencies need to be equal. The parameter value of the small-signal equivalent circuit of the field-effect transistor 301b at a high frequency, that is, the transfer conductance (gm) value, the capacitor (Cgs) value between the gate and the source, and the like are the parameters of the small-signal equivalent circuit of the field-effect transistor 302b. The closer the value, the stronger the virtual electrical ground that occurs at node 309b, and the less sensitive the circuit performance is to parasitic ground inductance. As is well known, the AC and high frequency small signal equivalent circuit of a field effect transistor is determined by the DC operating point. Therefore, from the viewpoint of the DC operating point, a resistor 310b having the same value as the resistor 303b should be connected to the drain of the field-effect transistor 302b. Then, the voltage value and the current value between the drain and the source of the DC of the field-effect transistor 301b are changed.
Since the voltage and current values between the drain and source at DC of 2b are equal, both field-effect transistors have equal small-signal equivalent circuits at high frequencies. However, since the output is taken out from the drain of the field effect transistor 301b, it is better not to connect the resistor 310b to the drain of the field effect transistor 302b at high frequencies in order to suppress loss of gain at high frequencies. Therefore, as shown in FIG. 5, a resistor 310b having the same value as the resistor 303b is connected in series to the drain of the field effect transistor 302b, and a capacitor is connected in parallel with the resistor 310b.
When 311b is connected, in DC, only the resistor 310b is connected to the drain side of the field effect transistor 302b, and the small signal equivalent circuit at high frequency of the field effect transistor 301b and the field effect transistor 302b becomes equal. A virtual electrical ground point 309b is generated which is stronger than the connection point 309a of one active balun 300a. And at high frequency, the capacitor 31
By the function of 1b, the effect of the resistor 310b is eliminated, and the loss of gain at high frequencies can be suppressed. Therefore, active balun 300b of FIG. 5 has a gain value equal to active balun 300a of FIG. 1 and is less sensitive to parasitic ground inductance.

FIG. 1 shows an amplifier according to an embodiment of the present invention for solving the above-mentioned problem (3). The amplifier 500a has a distortion improvement circuit composed of an inductor 501a and a resistor 502a, and the drain of the field-effect transistor 503a is connected to a power supply.
A resistor 504a connected between the drain of the field effect transistor 503a and the power supply 600a is connected between the capacitor 600a and the strain improving circuit.
This is an amplifier that is connected to a resistor 505a at its source and extracts an output from a terminal 506a connected to the drain of the field effect transistor 503a. As shown in FIG.
When a resistor 502a having a low resistance value and an inductor 501a are connected in parallel with a resistor 504a connected between the drain of the field effect transistor 503a and the power supply 600a, a good distortion characteristic can be obtained with low current consumption. ) Can be solved. Preferably, for high gain, resistor 504a is hundreds to thousands of ohms. To improve the distortion characteristics, the resistor 502a may be omitted, but a transient voltage at DC causes the field effect transistor 503a.
It is better to insert a resistor 502a in order to prevent the source and drain from being applied. In this case, preferably, the resistance 502a is several to several tens Ω. In order to suppress the influence of the resistor 502a at a high frequency, the inductor 501a is preferably several nH or more.
As is well known, typically the distortion characteristic is the current value flowing between the drain and source of the field effect transistor 503a of the final stage amplifier 500a of the frequency downconverter circuit of FIG.
Alternatively, the field-effect transistor 503a of the last-stage amplifier 500a
Strongly depends on the value of the voltage applied between the drain and the source. That is, the field-effect transistor of the final stage amplifier 500a
The higher the current value flowing between the drain and the source of the 503a, or the higher the voltage value between the drain and the source of the field effect transistor 503a of the final stage amplifier 500a, the better the distortion characteristics. In the conventional method, the distortion characteristic was improved by reducing the source resistance 505a of the amplifier 500a of FIG. 1 and increasing the value of the current flowing between the drain and the source. This is a problem in the receiving system in the state. One method of improving the distortion characteristics with low current consumption is to improve the distortion characteristics by increasing the voltage between the drain and the source of the field effect transistor 503a. Considering the above-mentioned relationship between the voltage value between the drain and the source of the field effect transistor 503a and the distortion characteristic, the drain of the field effect transistor 503a and the power supply 600a
, A resistor having a low resistance value should be connected to increase the voltage value between the drain and the source of the field effect transistor 503a. However, this method has the following problems. When taking output from the terminal 506a connected to the drain of the field-effect transistor 503a, a resistor 504a having a high resistance value is connected between the drain of the field-effect transistor 503a and the power supply 600a for high-frequency gain. Should. Therefore, if a resistor having a low resistance is connected between the field effect transistor 503a and the power supply 600a to improve the distortion characteristics, the gain at high frequencies is reduced. Therefore, there is a need for a method of improving distortion characteristics with low current consumption without lowering the gain at high frequencies. It is proposed as follows. 1, a resistor 504a for gaining a gain is connected between the drain of the field effect transistor 503a and the power supply 600a, and a resistor 502a having a low resistance value and a high inductance value are connected in parallel with the resistor 504a. When the inductor 501a is connected, good distortion characteristics can be obtained with low current consumption without reducing the gain at a high frequency. Preferably, for high gain, resistor 504a is hundreds to thousands of ohms. To improve the distortion characteristics, the resistor 502a may be omitted, but a transient voltage at DC may cause
It is better to insert a resistor 502a in order to prevent the connection between the drain and source of 3a. In this case, preferably the resistance
502a is several to several tens of ohms. In order to suppress the influence of the resistor 502a at a high frequency, the inductor 501a is preferably several nH or more. The effectiveness of this circuit configuration can be explained as follows. When the resistor 502a and the inductor 501a are connected in parallel with the resistor 504a connected between the drain of the field effect transistor 503a and the power supply 600a of the amplifier 500a in FIG.
Because it has a resistance value sufficiently lower than 504a, DC power supply 600a
From the field effect transistor 503a to the drain
Current flows only through 502a, and almost no current flows through resistor 504a. Typically, amplifier 500a is a field effect transistor
Since the 503a operates near the saturation region of the current value between the DC drain and the source, if the voltage value between the gate and the source of the field-effect transistor 503a is constant, the DC current value between the drain and the source Is constant without depending on the resistance value connected to the drain of the field effect transistor 503a. Therefore, when the resistor 502a has a sufficiently lower resistance value than the resistor 504a, a voltage value corresponding to the product of the constant current value and the resistor 502a is applied between the drain and the source of the field-effect transistor 503a. The voltage value between the drain of the field effect transistor 503a and the power supply 600a decreases, and the voltage value between the drain and the source of the field effect transistor 503a increases. In order to increase the voltage value between the drain and the source of the field effect transistor 503a, the resistor 502a may not be provided. It is better to insert a resistor 502a having a resistance value of several tens Ω. For the above reason, the voltage value applied between the drain and the source of the field effect transistor 503a increases, and the distortion characteristics can be improved. When an inductor 501a having a high inductance value is connected, the
Due to the action of 1a, only the resistor 504a is connected between the drain of the field effect transistor 503a and the power supply 600a, and the resistor 502a
The loss of gain at high frequencies due to the above is suppressed. Therefore, when the resistor 502a having a low resistance value and the inductor 501a are connected in parallel to the resistor 504a for gaining a gain between the drain of the field effect transistor 503a and the power supply 600a as in the amplifier 500a of FIG. The distortion characteristic is improved without increasing the current value, and the loss of gain due to the resistor 502a is suppressed at high frequencies by the operation of the inductor 501a. An attractive feature of the amplifier 500a of the novel circuit configuration shown in FIG. 1 is that the value of the resistor 505a connected to the source of the field-effect transistor 503a of the final stage amplifier 500a is lower in the conventional method than in the conventional distortion improvement method. Although the distortion characteristic was improved by increasing the current value between the drain and the source, the voltage between the drain and the source of the field effect transistor 503a was increased without increasing the current value in the amplifier 500a having the new circuit configuration. By doing so, the strain characteristics can be improved.

FIG. 6 shows an amplifier according to another embodiment of the present invention for solving the aforementioned problem (3). The amplifier 500b has a conductor line 501b having a length of a quarter wavelength with respect to the center frequency and a resistor 502b having a low resistance value.
Is connected between the drain of the field effect transistor 503b and the power supply 600c, a capacitor 507b is connected between the connection point of the conductor line 501b and the resistor 502b and the ground point, and between the drain of the field effect transistor 503b and the power supply 600c. The resistor 504b is connected, the resistor 505b is connected to the source of the field effect transistor 503b, and the conductor line 501b and the resistor 502b are formed on a semiconductor chip for a monolithic microwave integrated circuit, or a printed circuit board. This is an amplifier that extracts an output from a terminal 506b connected to the drain.
Preferably, for high gain, resistor 504b is in the hundreds to thousands of ohms.
It is. To improve the distortion characteristics, the resistor 502b may be omitted. However, in order to prevent a DC transient voltage from being applied between the drain and the source of the field effect transistor 503b, the resistor 502b is used.
It is better to put In this case, preferably, the resistance 502b is several to several tens Ω. In this amplifier 500b, as compared with the circuit configuration of the amplifier 500a of FIG. 1 described above, the inductor 501a has a frequency of 4% with respect to the center frequency connected to the capacitor 507b.
A conductor line 501b having a length of one-half wavelength was connected. Therefore, the capacitor 507b and the conductor line 501b having a quarter wavelength length with respect to the center frequency are functionally the same as the inductor 501a of FIG. That is, at high frequency, the conductor line 501b having a length of a quarter wavelength with respect to the center frequency is connected to the connection point 50b.
Open at 8b, only the resistor 504b is connected between the drain of the field effect transistor 503b and the power supply 600c, so that the loss of gain due to the resistor 502b having a low resistance value is suppressed. In DC, current flows only to the resistor 502b having a low resistance value between the drain of the field-effect transistor 503b and the power supply 600c. Therefore, for the above-described reason, the distortion characteristic can be improved by increasing the voltage value between the drain and the source of the field effect transistor 503b without increasing the current value even in the circuit configuration of FIG.

[0025]

From the above description, according to the present invention, the first
And the second differential transistor pair are provided on the same substrate, and the entire circuit can be integrated inside the semiconductor chip for a monolithic microwave integrated circuit. As a result, the circuit scale can be reduced, and it can be incorporated in a low-cost surface-mount plastic package due to insensitivity to parasitic grounding inductance, and a frequency downconverter having an advantage of obtaining good distortion characteristics with low current consumption can be realized. Accordingly, it can be seen that the use of the frequency downconverter having the novel circuit configuration of the present invention satisfies the direct requirement for the single-balanced frequency downconverter of the low-cost gallium arsenide monolithic microwave integrated circuit used for the low noise block. The novel circuit configuration of the present invention is expected to be effectively used not only in the direct broadcast satellite receiver system but also in other communication systems.

[Brief description of the drawings]

FIG. 1 is a diagram showing a high-frequency circuit according to an embodiment of the present invention.

FIG. 2 is a sectional view of a low-pass filter according to the embodiment.

FIG. 3 is a circuit diagram showing another mixer according to the embodiment.

FIG. 4 is a circuit diagram showing another mixer according to the embodiment.

FIG. 5 is a circuit diagram of an active balun according to the embodiment.

FIG. 6 is a circuit diagram showing an amplifier according to the embodiment;

FIG. 7 is a diagram showing a conventional high-frequency circuit.

[Explanation of symbols]

100 180 degree hybrid coupler 100a, 100b, 100c Differential input amplifier 101, 102, 103, 104, 201, 202, 203, 204, 301, 302,
303, 503, 102a, 104a, 105a, 106a, 202a, 204a, 205
a, 206a, 207a, 208a, 306a, 307a, 308a, 506a, 102
b, 202b, 204b, 206b, 208b, 209b, 210b, 102c, 202
c, 204c, 206c, 207c, 208c, 306b, 307b, 308b, 506b
Terminals 107a, 108a, 109a, 212, 213, 214, 215, 216, 215a, 2
16a, 217a, 218a, 219a, 309, 310, 311, 312, 303a, 3
10a, 311a, 312a, 304a, 303b, 304b, 310b, 401, 40
2, 403, 501, 502, 502a, 504a, 505a, 507a, 502b, 50
4b, 505b Resistance 200, 200b, 200c Single balance mixer 200a Dual gate single balance mixer 201a, 203a Dual gate field effect transistor 205, 208, 307 Drain 206, 209, 209a, 212a, 501a Inductor 207, 210, 210a, 213a, 305a, 305b, 311b, 313, 404,
508a, 507b, 10000, 10001, 10002, 10003, 10000a, 10
001a, 10002a, 10003a, 10004a, 10005a Capacitor 211, 308, 309a, 309b, 508b Connection point 211a, 214a Ground 300, 300a, 300b Active balun 304, 306, 101a, 103a, 301a, 302a, 503a, 201b, 203
b, 205b, 207b, 201c, 203c, 205c, 301b, 302b, 503b
Field-effect transistor 305 Source 400, 500, 500a, 500b Amplifier 501b Conductor line 600, 600a, 600b, 600c Power supply 2000 Monolithic microwave integrated circuit semiconductor chip 2001 Conductor layer 2002 Conductor ground plane

Continuation of the front page (56) References JP-A-2000-307349 (JP, A) JP-A-6-310943 (JP, A) JP-A-59-95705 (JP, A) JP-A-63-142908 (JP, A) A) JP-A-10-224152 (JP, A) JP-A-61-182305 (JP, A) JP-A-49-11547 (JP, A) JP-A-63-76510 (JP, A) −9332 (JP, U) (58) Fields investigated (Int. Cl. ⁷ , DB name) H03F 3/45 H03D 7/00 H03D 7/14

Claims (1)

(57) [Claims] 1. An active balun formed by connecting a source of a first field-effect transistor and a source of a second field-effect transistor, wherein a drain of the first field-effect transistor and a voltage supply means are connected. A feedback circuit having a first resistor connected between the drain and the gate of the first field-effect transistor, the feedback circuit having a second resistor and a first capacitor connected in series to the second resistor; Is connected, and a third resistor and a second capacitor connected in parallel with the third resistor are connected between a drain of the second field-effect transistor and the voltage supply means, and the first field-effect transistor is connected to the first field-effect transistor. A first high-frequency input signal is input to the gate of the transistor, and a second high-frequency input signal having the same amplitude as the first high-frequency input signal and having a phase difference of 180 degrees is applied to the second high-frequency input signal. A high-frequency circuit which is input to a gate of a field effect transistor.