JP3611235B2

JP3611235B2 - Active filter control method

Info

Publication number: JP3611235B2
Application number: JP26777898A
Authority: JP
Inventors: 中俊彦田; 曳繁之舩
Original assignee: ジェーピーイー株式会社
Priority date: 1998-09-22
Filing date: 1998-09-22
Publication date: 2005-01-19
Anticipated expiration: 2018-09-22
Also published as: JP2000102168A

Description

【０００１】
【発明の属する技術分野】
本発明は、電力系統に接続された負荷側で発生する高調波無効電力及び不平衡を打ち消すように補償するアクティブフィルタ制御方法に関する。
【０００２】
【従来の技術】
半導体応用機器の普及により電力系統の高調波が問題となっている。これに加えて、電力系統の不平衡が将来問題となる可能性があることが論文などで指摘され、不平衡と高調波を一括して補償する補償装置が、重電機メーカの発表する論文などで提案されるとともに系統安定用として実用化されつつある。これらの補償装置は、いわゆる無効電力補償装置ＳＶＣに不平衡と高調波の補償機能を付加したものと考えることができる。従来、この様な目的として、平成７年電気学会全国大会一般講演Ｎｏ．８１０及び平成５年電気学会全国大会一般講演Ｎｏ．６０４等で発表されているように、その制御方式は、表現方法は異なるものの、概ね三相ａ−ｂ−ｃ座標の電流を正相変換し、フィルタを用いて高調波分と基本波分を分離し、正相分の有効無効電流を検出する。他方、逆相変換を行って同様のことを行う。これらから、高調波電流及び逆相電流を検出し、これを指令値として不平衡と高調波の補償を行うものである。従って、これまでの方法は、少なくとも正相変換及び逆相変換の２系統の座標変換を行う必要があった。
【０００３】
【発明が解決しようとする課題】
本発明は、高調波および不平衡を一括補償する際に、２系統の座標変換が必要であったものを、１系統の座標変換だけを行って負荷電流の基本波正相分の有効電流だけを検出し、これと負荷電流の差を一括補償することにより、個別の補償対象成分を個々に検出することのない、簡易なアクティブフィルタ制御方法を提供することを目的とする。
【０００４】
【課題を解決するための手段】
本発明によれば、電力系統に接続された負荷側で発生する高調波、無効電力及び不平衡を打ち消すように補償するアクティブフィルタ制御方法において、受電端における該負荷に流入する電流の基本波正相分の有効電流を検出し、該電流の基本波正相分の有効電流以外の成分を一括補償する際に、前記受電端の電圧を実際の電気角を検出することなく電源の基本周波数と等しい鋸歯状波を用いてｄ−ｑ座標上へ変換し、ローパスフィルタを用いてｄ軸上の極低周波分を含む直流分を分離し、該直流分のｄ軸成分とｑ軸成分の比の逆正接を求め、該逆正接と該鋸歯状波の和を求めることにより該受電端電圧の基本波の電気角を求め、該電気角を用いて負荷電流をｄ−ｑ座標上に変換し、別のローパスフィルタを用いて極低周波を含む直流分を分離し、該電気角を用いてａ−ｂ−ｃ座標上に変換することにより負荷電流の基本波正相分の有効電流を検出し、該負荷に流入する電流の基本波正相分の有効電流と負荷電流との差を求め、アクティブフィルタの指令値とするようになっている。
【０００６】
【発明の実施の形態】
以下、図１〜３を参照して、本発明の実施例について説明する。
【０００７】
図１は、本実施例の高調波および不平衡の同時補償法を示すアクティブフィルタの構成図を示す。図において、１は平衡三相電源、２は電源インピーダンス、３は不平衡負荷、４はサイリスタ整流回路、５はサイリスタ整流回路の出力電流、６はフィルタリアクトル、７はアクティブフィルタを示す。該平衡三相電源１には、Ｒ−Ｌ回路から構成される該不平衡負荷３と高調波の発生源である該サイリスタ整流回路４が接続され、これらと並列に該アクティブフィルタ７が接続された構成になっている。
【０００８】
図２は、本実施例のアクティブフィルタ制御方式のブロック図を示す。以下に、図１、図２を参照して、制御方法について説明する。図１において、Ｖｓは電源電圧、Ｉｓは電源電流、Ｖｔは受電端電圧、Ｉｌは負荷側電流、Ｉｃはアクティブフィルタ７の出力電流（補償電流）、Ｖｄはアクティブフィルタ７の直流電圧、Ｚｌは負荷インピーダンス、Ｉｄｃはサイリスタ整流回路の出力電流を示す。図２において、Ｖｔａ、Ｖｔｂ、Ｖｔｃは各相の受電端電圧で、Ｉｌａ、Ｉｌｂ、Ｉｌｃは各相の負荷側電流である。これら受電端電圧を実際の電気角θｖを検出することなく周波数６０Ｈｚの理想的な鋸歯状波θｒを用いてｄ−ｑ座標へ変換する。このとき、受電端電圧のｄ軸成分Ｖｔｄおよびｑ軸成分Ｖｔｑは、下記の数１の式で与えられる。また、数２は、ａ−ｂ−ｃ座標からｄ−ｇ座標への変換行列Ｃ１の計算式を示す。
【０００９】

このとき、ｄ−ｑ座標のｄ軸成分Ｖｔｄおよびｑ軸成分Ｖｔｑは、電源周波数に比較し極低周波数を含む直流分と交流分に分解することが出来る。この極低周波分を含む直流分をＬＰＦ１（ローパスフィルタ）を用いて分離し、逆ｄ−ｑ変換することで、電源周波数が微妙に変動した場合でも基本波分を検出することが出来る。これと同様に、負荷電流についてもｄ−ｑ座標上へ変換することで、正相分を検出できる。しかしながら、受電端電圧の電気角θｖと鋸歯状波θｒに位相差が存在するため、受電端電圧の正相分と同相成分を検出することが出来ない。そこで、ｄ−ｑ座標上でＬＰＦを用いて検出したＶｔｄ及びＶｔｑに着目すると、受電端電圧の電気角θｖは下記の数３の式で与えられる。
【００１０】

数３において、Ｖｔｄ１は受電端電圧のｄ軸電圧の直流分で、Ｖｔｑ１は受電端電圧のｑ軸電圧の直流分を示す。数３を用いて負荷電流をｄ−ｑ座標上へ変換すると下記の数４の式となる。
【００１１】

数４で、負荷電流のｄ軸成分Ｉｌｄ、負荷電流のｑ軸成分ＩｌｑからＬＰＦ２を用いてｄ軸成分の直流分Ｉｌｄ１のみを抽出し、これらを再びａ−ｂ−ｃ座標上へ変換することで受電端電圧の正相分Ｖｔｐと同相の負荷電流の正相分Ｉｌａｐ、Ｉｌｂｐ、Ｉｌｃｐを検出できる。このとき、前記Ｉｌａｐ、Ｉｌｂｐ、Ｉｌｃｐは下記の数５の式で与えられる。数６はｄ−ｑ座標からａ−ｂ−ｃ座標への変換行列Ｃ２の計算式を示す。
【００１２】

今、負荷電流Ｉｌと検出した負荷電流の正相分Ｉｌｐとの差を求め、これをアクティブフィルタ７の指令値とする。これにより、高調波、無効電力および不平衡を一括して補償できる。このとき、サイリスタ整流回路の各相への指令値ＩＣａ、ＩＣｂ、ＩＣｃは、下記の数７の式に示す値となる。
【００１３】

本システムは、補償対象とする電気量を検出するのではなく電源側でどの様な電流波形が望ましいかということに着目している点に特色がある。
【００１４】
次に、本発明の制御方法の有効性を確認するために、計算機シミュレーションを行った。このとき、不平衡負荷の一例であるＲ−Ｌ負荷の定数を下記の表１に示す。
【００１５】

表１においてＺ１は不平衡負荷インピーダンス、Ｒａ〜Ｒｃは各相の負荷抵抗、Ｌａ〜Ｌｃは各相の負荷リアクトル、Ｉｄｃはサイリスタ整流回路の出力電流、αはサイリスタ整流回路の位相制御角である。アクティブフィルタ７の主回路は、ヒステリシスコンパレータ方式ＰＷＭインバータから構成されている。また、ｄ−ｑ座標上における直流分抽出のＬＰＦには２次バタワース形を使用し、カットオフ周波数を５Ｈｚとした。図３にシミュレーション結果を示す。図３において、添字”ｓ”は電源側、”ｔ”は受電端、”ｌ”は負荷側を表している。一番上はａ相、２番目はｂ相、そして３番目はｃ相の電圧および電流を示し、一番下はａ、ｂ、ｃ各相の負荷側電流を示し、いずれも横軸が経過時間である。図から解るように、不平衡Ｒ−Ｌ負荷と高調波発生源であるサイリスタ整流回路が接続されているために、線Ｉｌａ、ＩｌｂおよびＩｌｃは不平衡状態で高調波電流を含んでいる。一方、電源側において電源電流Ｉｓａ、ＩｓｂおよびＩｓｃは高調波電流が補償されており、且つ、平衡三相となっている。また、受電端電圧Ｖｔａ、Ｖｔｂ、Ｖｔｃに比較し、電源電流はそれぞれ同位相となっており、電源側で無効電流が補償されていることが確認できる。
【００１６】
【発明の効果】
本発明によれば、これまで必要とされた正相及び逆相変換のなかで正相変換を用いることのみで不平衡と高調波の補償の一括補償が可能であり、且つ、電源側には、正相分の無効電流も補償されることから、電力会社から見ると理想的な電流波形となる。一方、需用家から見ると受電設備には常に正弦波で、且つ有効電流のみが流れることにより受電設備の有効利用が可能となるという効果がある。
【図面の簡単な説明】
【図１】実施例のアクティブフィルタの構成図。
【図２】実施例のアクティブフィルタの制御方法のブロック図。
【図３】実施例のシミュレーション結果。
【符号の説明】
１・・・平衡三相電源
２・・・電源インピーダンス
３・・・不平衡負荷
４・・・サイリスタ整流回路
５・・・サイリスタ整流回路の出力電流
６・・・フィルタリアクトル
７・・・アクティブフィルタ
Ｖｓ・・・電源電圧
Ｉｓ・・・電源電流
Ｉｓａ・・・ａ相の電源電流
Ｉｓｂ・・・ｂ相の電源電流
Ｉｓｃ・・・ｃ相の電源電流
Ｖｔ・・・受電端電圧
Ｉｌ・・・負荷側電流
Ｉｃ・・・アクティブフィルタの出力電流（補償電流）
Ｖｄ・・・アクティブフィルタの直流電圧
Ｚｌ・・・負荷インピーダンス
Ｉｄｃ・・・サイリスタ整流回路の出力電流
Ｖｔａ・・・ａ相の受電端電圧
Ｖｔｂ・・・ｂ相の受電端電圧
Ｖｔｃ・・・ｃ相の受電端電圧
Ｉｌａ・・・ａ相の負荷側電流
Ｉｌｂ・・・ｂ相の負荷側電流
Ｉｌｃ・・・ｃ相の負荷側電流
Ｖｔｄ・・・受電端電圧のｄ軸成分
Ｖｔｑ・・・受電端電圧のｑ軸成分
Ｖｔｄ１・・・受電端電圧のｄ軸電圧の直流分
Ｖｔｑ１・・・受電端電圧のｑ軸電圧の直流分
Ｉｌｄ・・・負荷電流のｄ軸成分
Ｉｌｑ・・・負荷電流のｑ軸成分
Ｉｌｄ１・・・負荷電流のｄ軸成分の直流分
Ｖｔｐ・・・受電端電圧の正相分
Ｉｌｐ・・・負荷電流の正相分
Ｉｌａｐ・・・ａ相の負荷電流の正相分
Ｉｌｂｐ・・・ｂ相の負荷電流の正相分
Ｉｌｃｐ・・・ｃ相の負荷電流の正相分
ＬＰＦ１・・・ローパスフィルタ
ＬＰＦ２・・・ローパスフィルタ
Ｃ１・・・ａ−ｂ−ｃ座標からｄ−ｑ座標への変換行列
Ｃ２・・・ｄ−ｑ座標からａ−ｂ−ｃ座標への変換行列
Ｒａ・・・ａ相負荷抵抗
Ｒｂ・・・ｂ相負荷抵抗
Ｒｃ・・・ｃ相負荷抵抗
Ｌａ・・・ａ相負荷リアクトル
Ｌｂ・・・ｂ相負荷リアクトル
Ｌｃ・・・ｃ相負荷リアクトル
α・・・サイリスタ整流回路の位相制御角
ＩＣａ・・・サイリスタ整流回路のａ相指令値
ＩＣｂ・・・サイリスタ整流回路のｂ相指令値
ＩＣｃ・・・サイリスタ整流回路のｃ相指令値
θｖ・・・受電端の実際の電気角
θｒ・・・受電端の周波数６０Ｈｚの理想的な鋸歯状波[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an active filter control method for compensating so as to cancel harmonic reactive power and unbalance generated on a load side connected to a power system.
[0002]
[Prior art]
With the spread of semiconductor application equipment, harmonics in the power system have become a problem. In addition to this, it is pointed out in papers that unbalance in the power system may become a problem in the future, and a compensation device that compensates for unbalance and harmonics at once is a paper published by a heavy electrical machinery manufacturer. And is being put into practical use for system stabilization. These compensators can be thought of as a so-called reactive power compensator SVC with an unbalanced and harmonic compensation function. Conventionally, for this purpose, the 1995 IEEJ National Conference General Lecture No. No. 810 and 1993 National Congress of the Institute of Electrical Engineers of Japan As announced in 604 etc., although the control method is different in the expression method, the current of the three-phase abc coordinate is converted to normal phase, and the harmonic component and the fundamental component are converted using a filter. Separate and detect the effective reactive current for the positive phase. On the other hand, the reverse phase conversion is performed and the same thing is performed. From these, a harmonic current and a negative phase current are detected, and these are used as command values to compensate for unbalance and harmonics. Therefore, the conventional method needs to perform coordinate conversion of at least two systems of normal phase conversion and reverse phase conversion.
[0003]
[Problems to be solved by the invention]
In the present invention, when the harmonics and unbalance are collectively compensated, only the effective current corresponding to the positive phase of the fundamental wave of the load current is obtained by performing only the coordinate conversion of one system, which is necessary for the coordinate conversion of the two systems. It is an object of the present invention to provide a simple active filter control method in which individual compensation target components are not detected individually by detecting the difference between the current and the load current and collectively compensating for the difference between the current and the load current.
[0004]
[Means for Solving the Problems]
According to the present invention, in an active filter control method for compensating so as to cancel harmonics, reactive power and unbalance generated on the load side connected to the power system, the fundamental wave positive current of the current flowing into the load at the power receiving end is compensated. When detecting the effective current of the phase and collectively compensating components other than the active current of the fundamental wave positive phase of the current, the voltage at the power receiving end is set to the fundamental frequency of the power supply without detecting the actual electrical angle. Conversion to dq coordinates using equal sawtooth wave, separation of DC component including extremely low frequency component on d axis using low pass filter, and ratio of d component and q axis component of DC component Is obtained, the sum of the arc tangent and the sawtooth wave is obtained to obtain the electrical angle of the fundamental wave of the receiving end voltage, and the load current is converted into dq coordinates using the electrical angle. Using a separate low-pass filter, separate the DC component including extremely low frequencies. The effective current corresponding to the fundamental positive phase of the load current is detected by converting the coordinates into the abc coordinates using the electrical angle, and the effective current corresponding to the fundamental positive phase of the current flowing into the load is detected. Is obtained as a command value for the active filter.
[0006]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to FIGS.
[0007]
FIG. 1 is a block diagram of an active filter showing a simultaneous harmonic and unbalance compensation method of this embodiment. In the figure, 1 is a balanced three-phase power source, 2 is a power source impedance, 3 is an unbalanced load, 4 is a thyristor rectifier circuit, 5 is an output current of the thyristor rectifier circuit, 6 is a filter reactor, and 7 is an active filter. The balanced three-phase power source 1 is connected to the unbalanced load 3 composed of an R-L circuit and the thyristor rectifier circuit 4 that is a harmonic generation source, and the active filter 7 is connected in parallel thereto. It has a configuration.
[0008]
FIG. 2 shows a block diagram of the active filter control system of this embodiment. Hereinafter, a control method will be described with reference to FIGS. 1 and 2. In FIG. 1, Vs is a power supply voltage, Is is a power supply current, Vt is a receiving end voltage, Il is a load side current, Ic is an output current (compensation current) of the active filter 7, Vd is a DC voltage of the active filter 7, and Zl is The load impedance, Idc, indicates the output current of the thyristor rectifier circuit. In FIG. 2, Vta, Vtb, and Vtc are the receiving end voltages of the respective phases, and Ila, Ilb, and Ilc are the load-side currents of the respective phases. These receiving end voltages are converted into dq coordinates using an ideal sawtooth wave θr having a frequency of 60 Hz without detecting the actual electrical angle θv. At this time, the d-axis component Vtd and the q-axis component Vtq of the power receiving end voltage are given by the following equation (1). Equation 2 shows the calculation formula of the conversion matrix C1 from the abc coordinate to the dg coordinate.
[0009]

At this time, the d-axis component Vtd and the q-axis component Vtq of the dq coordinate can be decomposed into a direct current component and an alternating current component including an extremely low frequency compared to the power supply frequency. By separating the DC component including this extremely low frequency component using LPF1 (low pass filter) and performing inverse dq conversion, the fundamental component can be detected even when the power supply frequency fluctuates slightly. Similarly to this, the positive phase can be detected by converting the load current to the dq coordinate. However, since there is a phase difference between the electrical angle θv of the power receiving end voltage and the sawtooth wave θr, it is impossible to detect the same phase component as the positive phase of the power receiving end voltage. Therefore, paying attention to Vtd and Vtq detected using LPF on the dq coordinate, the electrical angle θv of the receiving end voltage is given by the following equation (3).
[0010]

In Equation 3, Vtd1 represents the direct current component of the d-axis voltage of the power receiving end voltage, and Vtq1 represents the direct current component of the q axis voltage of the power receiving end voltage. When the load current is converted onto the dq coordinates using Equation 3, the following Equation 4 is obtained.
[0011]

In Equation 4, only the DC component Ild1 of the d-axis component is extracted from the d-axis component Ild of the load current and the q-axis component Ilq of the load current by using LPF2, and these are converted again into abc coordinates. Thus, the positive phase components Ilap, Ilbp, and Ilcp of the load current in phase with the positive phase component Vtp of the receiving end voltage can be detected. At this time, the Ilap, Ilbp, and Ilcp are given by the following equation (5). Equation 6 shows the calculation formula of the conversion matrix C2 from the dq coordinates to the abc coordinates.
[0012]

Now, the difference between the load current Il and the positive phase component Ilp of the detected load current is obtained, and this is used as the command value of the active filter 7. Thereby, harmonics, reactive power, and unbalance can be compensated collectively. At this time, command values ICa, ICb, ICc to the respective phases of the thyristor rectifier circuit are values shown in the following equation (7).
[0013]

This system is characterized in that it focuses on what kind of current waveform is desirable on the power supply side, rather than detecting the amount of electricity to be compensated.
[0014]
Next, computer simulation was performed in order to confirm the effectiveness of the control method of the present invention. At this time, constants of the RL load which is an example of the unbalanced load are shown in Table 1 below.
[0015]

In Table 1, Z1 is an unbalanced load impedance, Ra to Rc are load resistances of each phase, La to Lc are load reactors of each phase, Idc is an output current of the thyristor rectifier circuit, and α is a phase control angle of the thyristor rectifier circuit. . The main circuit of the active filter 7 is composed of a hysteresis comparator type PWM inverter. Further, a secondary Butterworth shape was used for the LPF for DC component extraction on the dq coordinate, and the cut-off frequency was 5 Hz. FIG. 3 shows the simulation results. In FIG. 3, the suffix “s” represents the power supply side, “t” represents the power receiving end, and “l” represents the load side. The top is the a phase, the second is the b phase, and the third is the c phase voltage and current. The bottom is the load side current of each phase a, b, c. It's time. As can be seen from the figure, because the unbalanced RL load and the thyristor rectifier circuit that is the harmonic generation source are connected, the lines Ila, Ilb, and Ilc contain harmonic currents in an unbalanced state. On the other hand, the power source currents Isa, Isb, and Isc are compensated for harmonic currents and have a balanced three-phase on the power source side. Further, the power supply currents are in phase with each other as compared with the power receiving end voltages Vta, Vtb, and Vtc, and it can be confirmed that the reactive current is compensated on the power supply side.
[0016]
【The invention's effect】
According to the present invention, it is possible to perform collective compensation of unbalance and harmonic compensation only by using positive phase conversion among the normal phase and negative phase conversions required so far, and on the power supply side, Since the reactive current corresponding to the positive phase is also compensated, an ideal current waveform is obtained from the viewpoint of the electric power company. On the other hand, from the viewpoint of consumers, there is an effect that the power receiving facility can be effectively used by always receiving a sine wave and only an effective current flowing in the power receiving facility.
[Brief description of the drawings]
FIG. 1 is a configuration diagram of an active filter according to an embodiment.
FIG. 2 is a block diagram of an active filter control method according to the embodiment.
FIG. 3 shows a simulation result of the example.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 ... Balanced three-phase power supply 2 ... Power supply impedance 3 ... Unbalanced load 4 ... Thyristor rectifier circuit 5 ... Output current 6 of thyristor rectifier circuit ... Filter reactor 7 ... Active filter Vs ... Power supply voltage Is ... Power supply current Isa ... A phase power supply current Isb ... B phase power supply current Isc ... C phase power supply current Vt ... Receiving end voltage Il ... Load side current Ic ... Active filter output current (compensation current)
Vd ... DC voltage Zl of active filter ... Load impedance Idc ... Output current Vta of thyristor rectifier circuit ... a-phase receiving end voltage Vtb ... b-phase receiving end voltage Vtc ... c Phase receiving side voltage Ila ... a phase load side current Ilb ... b phase load side current Ilc ... c phase load side current Vtd ... d axis component Vtq of receiving end voltage ... Q-axis component Vtd1 of the receiving end voltage ... DC component Vtq1 of the d-axis voltage of the receiving end voltage ... DC component Ild of the q-axis voltage of the receiving end voltage ... d-axis component Ilq of the load current ... load Q-axis component Ild1 of the current: DC component Vtp of the d-axis component of the load current Vtp: positive phase component Ilp of the receiving end voltage ... positive component of the load current Ilap ... positive of the load current of the a phase Phase Ilbp ... Positive phase Ilcp of b-phase load current ... Phase load current of positive phase LPF1 ... low pass filter LPF2 ... low pass filter C1 ... transformation matrix C2 from abc coordinate to dq coordinate a ... from dq coordinate to a- Conversion matrix Ra to bc coordinates Ra ... a phase load resistance Rb ... b phase load resistance Rc ... c phase load resistance La ... a phase load reactor Lb ... b phase load reactor Lc. .. c-phase load reactor α ... phase control angle ICa of thyristor rectifier circuit ... a-phase command value ICb of thyristor rectifier circuit ... b-phase command value ICc of thyristor rectifier circuit ... c of thyristor rectifier circuit Phase command value θv: actual electrical angle at the receiving end θr: ideal sawtooth wave with a receiving end frequency of 60 Hz

Claims

In an active filter control method that compensates to cancel harmonics, reactive power and unbalance generated on the load side connected to the power system, the active current corresponding to the fundamental positive phase of the current flowing into the load at the receiving end is calculated. When a component other than the active current of the fundamental wave positive phase of the current is detected and collectively compensated, a sawtooth wave equal to the fundamental frequency of the power supply is used for the voltage at the power receiving end without detecting the actual electrical angle. Converting to dq coordinates, using a low-pass filter, separating a direct current component including an extremely low frequency component on the d axis, and obtaining an arctangent of a ratio of the d axis component and the q axis component of the direct current component, The electrical angle of the fundamental wave of the receiving end voltage is obtained by obtaining the sum of the arctangent and the sawtooth wave, and the load current is converted into dq coordinates using the electrical angle, and another low-pass filter is provided. Use to separate the DC component including extremely low frequencies and use the electrical angle The effective current corresponding to the fundamental positive phase of the load current is detected by converting the coordinates on the abc coordinate, and the difference between the effective current corresponding to the fundamental positive phase of the current flowing into the load and the load current is calculated. An active filter control method characterized by obtaining an active filter command value.