JP3632962B2 - Beat component generating circuit and broadcast wave receiving tuner including this circuit - Google Patents

Description

と表される。ただし、
Ｒ^２（ｔ）＝Ｂ_１ ^２（ｔ）＋Ｂ_２ ^２（ｔ）＋２Ｂ_１（ｔ）Ｂ_２（ｔ）ｃｏｓ（φ_１−φ_２）
Ｒ（ｔ）ｃｏｓβ（ｔ）＝Ｂ_１（ｔ）ｃｏｓφ_１＋Ｂ_２（ｔ）ｃｏｓφ_２
Ｒ（ｔ）ｓｉｎβ（ｔ）＝Ｂ_１（ｔ）ｓｉｎφ_１＋Ｂ_２（ｔ）ｓｉｎφ_２
とする。
【００２２】
ゴーストのない場合、β（ｔ）＝φ_１と定数であるが、ゴーストのある場合には、β（ｔ）は不要な放送波映像信号の振幅変化に応じて変動する時間関数となる。表記の煩雑さを避けるためにβ（ｔ）を単にβで表すと、不要な放送波映像信号が単一ゴーストを伴って入力される場合のＲＦ入力信号はＲＦ入力信号＝Ａ（ｔ）ｃｏｓ（ωｔ）＋Ｒ（ｔ）ｃｏｓ｛（ω＋δ）ｔ＋β｝と表せる。この式を（１）式とする。
【００２３】
つまり、この直交検波部３では、このＡ（ｔ）ｃｏｓ（ωｔ）＋Ｒ（ｔ）ｃｏｓ｛（ω＋δ）ｔ＋β｝で表記されるＲＦ入力信号を直交検波している。そして、希望波搬送波と同相の成分（同相成分）をＩ信号、希望波搬送波と直交の成分（直交成分）をＱ信号と呼称すると、Ｉ信号は、Ｉ信号＝Ａ（ｔ）＋Ｒ（ｔ）ｃｏｓ（δｔ＋β）と表記でき、Ｑ信号は、Ｑ信号＝Ｒ（ｔ）ｓｉｎ（δｔ＋β）と表記できる。なお、直交検波部３で発振器出力の位相をπ／２だけ進ませた信号により乗算を行うと、直交成分はマイナス符号が付くことになるが、以後、説明を簡略にするため、Ｑ信号を正（プラス）の信号として表記することにする。
【００２４】
Ｉ信号＝Ａ（ｔ）＋Ｒ（ｔ）ｃｏｓ（δｔ＋β）を（２）式とし、Ｑ信号＝Ｒ（ｔ）ｓｉｎ（δｔ＋β）を（３）式とする。
同相成分Ａ／Ｄ変換部５は、希望波搬送波と同相成分であるＩ信号をアナログデジタル変換するものである。
【００２５】
同相成分遅延部７（図中２ＨＤＬ）は、後記するビート成分生成部１７Ａの構成要素である同一構成の２系統の奇数タップトランスバーサルフィルタの中央タップの出力信号とＩ信号との相対的な時間関係を合わせるために、放送波受信チューナ１の直交検波部３で分離されているＩ信号を２Ｈ遅延（２水平走査期間分シフト）させるものである。これにより、トランスバーサルフィルタでＩ信号に対して相対的に２Ｈ進相したＱ信号まで取り扱うことができる。
【００２６】
高周波成分抽出遅延部９は、同相成分Ａ／Ｄ変換部５でアナログデジタル変換された信号の高周波成分を取り出して、この高周波成分の信号を、低周波成分抽出遅延部１１およびビート成分生成部１７Ａから出力される信号と同期するように遅延調整させるものである。この実施の形態では、一例として、高周波成分抽出遅延部９はハイパスフィルタ（ＨＰＦ）と遅延調整回路（図中ＤＬ調整）とによって構成されている。
【００２７】
低周波成分抽出遅延部１１は、同相成分Ａ／Ｄ変換部５でアナログデジタル変換された信号の低周波成分を取り出して、この低周波成分の信号を、高周波成分抽出遅延部９およびビート成分生成部１７Ａから出力される信号と同期するように遅延調整させるものである。この実施の形態では、一例として、低周波成分抽出遅延部１１はローパスフィルタ（ＬＰＦ）と遅延調整回路（図中ＤＬ調整）とによって構成されている。
【００２８】
なお、Ｉ信号には希望波映像信号と妨害波映像信号（Ｒ（ｔ））とによる映像ビート成分が混合している。このＩ信号に混合している映像ビート成分（ビート妨害）は、映像信号の低周波成分に関して除去できれば、視覚上十分に良好な希望波映像信号が得られることから、ビート成分を除去する除去対象は低周波成分に限定しても差し支えない。それゆえ、高周波成分抽出遅延部９と低周波成分抽出遅延部１１とによって、Ｉ信号が高周波成分と低周波成分とに分離されている。
【００２９】
直交成分Ａ／Ｄ変換部１３は、希望波搬送波と直交する成分であるＱ信号をアナログデジタル変換するものである。
低周波直交成分抽出部１５（図中ＬＰＦ）は、直交成分Ａ／Ｄ変換部１３でアナログデジタル変換された信号の直交成分の低周波成分を抽出するものである。この低周波直交成分抽出部１５は、この実施の形態では、一例として、単独のローパスフィルタ（ＬＰＦ）によって構成されている。
【００３０】
ビート成分生成部１７Ａは、低周波直交成分抽出部１５で抽出された直交成分の低周波成分から映像ビート成分を生成するものであり、詳細は後記する。
【００３１】
映像信号出力部１９は、低周波成分抽出遅延部１１によって取り出された低周波成分からビート成分生成部１７Ａで生成された映像ビート成分（詳細は後記する）を減算し、妨害波成分を除去し、その後で、高周波成分抽出遅延部９で抽出された高周波成分と合成することによって妨害波成分が十分低減された希望波映像信号をデジタル信号として出力するものである。
【００３２】
（ビート成分生成部［ビート成分生成回路］の構成）
次に、ビート成分生成部１７Ａについて、詳細に説明する。ビート成分生成部１７Ａは、乗算器２１ａと、乗算器２１ｂと、発振器２１ｃと、加算係数乗算器２３（２３ａ_１・・・、２３ｂ_１・・・）と、１水平走査期間の単位遅延素子２５（２５ａ_１・・・、２５ｂ_１・・・）と、加算集計器２７（２７ａ、２７ｂ）と、乗算器２９ａと、乗算器２９ｂと、減算器３１とを備えている。このビート成分生成部１７Ａでは、妨害波映像信号の瞬時位相を検出して妨害波映像信号の瞬時位相に追随した映像ビート成分を生成しているので、極めて大きなビート抑圧量を得ることができるように構成されている。ビート抑圧量とは、入力ビート成分に対する残差ビート成分の電力比のことである。
【００３３】
乗算器２１ａは、低周波直交成分抽出部１５で抽出されたアナログデジタル変換後のＱ信号の低周波成分に発振器２１ｃで発振されたオフセット周波数と同一周波数のｓｉｎ波を乗算して、第一直交成分検波出力信号を得るものである。
【００３４】
乗算器２１ｂは、低周波直交成分抽出部１５で抽出されたアナログデジタル変換後のＱ信号の低周波成分に発振器２１ｃで発振させたオフセット周波数と同一周波数のｓｉｎ波の位相をπ／２だけ進ませたｃｏｓ波を乗算して、第二直交成分検波出力信号を得るものである。ここで、乗算器２１ａおよび乗算器２１ｂでのＱ信号の演算処理は、Ｑ信号を直交検波することに等しい。
【００３５】
発振器２１ｃは、オフセット周波数と同一周波数の発振信号を生成するものであり、オフセット周波数とは、２つの放送局から送信される電波に同一の搬送波周波数が使用される場合に、混信が生じてもその妨害を視覚的に見えにくくするために、一方の放送局の搬送波周波数を１０ｋＨｚ程度ずらした周波数のことである。この発振器２１から出力されるｓｉｎ波をｓｉｎ（δｔ＋θ）と表記すれば、このｓｉｎ波の位相をπ／２だけ進ませたｃｏｓ波はｃｏｓ（δｔ＋θ）と表記される。
なお、乗算器２１ａと発振器２１ｃとが請求項に記載した第一直交成分検波出力手段に相当するものであり、乗算器２１ｂと発振器２１ｃとπ／２位相器（図中、π／２）とが請求項に記載した第二直交成分検波出力手段に相当するものである。
【００３６】
２系統あるトランスバーサルフィルタを構成する加算係数乗算器２３は、第一直交成分検波出力信号および第二直交成分検波出力信号に加算係数（Ｃ_２）を乗算すると共に、１水平走査期間の単位遅延素子２５の各出力信号に加算係数（Ｃ_０、Ｃ_１、Ｃ_２）を乗算するものである。この加算係数は、トランスバーサルフィルタの各タップ出力をベクトル表示した際に得られるトランスバーサルフィルタ出力信号の瞬時位相を検出するための拘束条件（後記する）によって制限を受けるものである。
【００３７】
２系統あるトランスバーサルフィルタを構成する１水平走査期間の単位遅延素子２５（図中１ＨＤＬ）は、第一直交成分検波出力信号および第二直交成分検波出力信号を１水平走査期間を単位として遅延させ、トランスバーサルフィルタの中央タップの出力信号に対して１水平走査期間（１Ｈ）前の信号（１Ｈ遅延信号（表示される画面では画面上方に該当する））、２Ｈ前の信号、１Ｈ先の信号（１Ｈ進相信号（表示される画面では画面下方に該当する））、２Ｈ先の信号のタップ出力を可能にさせるものである。なお、この実施の形態では、１水平走査期間の単位遅延素子としているが、遅延単位を変更することにより、トランスバーサルフィルタの周波数特性を変えることができる。ただし、タップ数および遅延単位を変更する場合には同期成分遅延部７の遅延時間もそれに合わせて変更することが必要である。
【００３８】
この図１に示したように、加算係数乗算器２３ａ_１が乗算器２１ａからの出力である第一直交成分検波出力信号（２Ｈ進相信号）に加算係数Ｃ_２を乗算しており、加算係数乗算器２３ａ_２が１水平走査期間の単位遅延素子２５ａ_１からの出力であるタップ出力信号（１Ｈ進相信号）に加算係数Ｃ_１を乗算しており、加算係数乗算器２３ａ_３が１水平走査期間の単位遅延素子２５ａ_２からの出力である中央タップ出力信号に加算係数Ｃ_０を乗算している。また、加算係数乗算器２３ａ_４が１水平走査期間の単位遅延素子２５ａ_３からの出力であるタップ出力信号（１Ｈ遅延信号）に加算係数Ｃ_１を乗算しており、加算係数乗算器２３ａ_５が１水平走査期間の単位遅延素子２５ａ_４からの出力であるタップ出力信号（２Ｈ遅延信号）に加算係数Ｃ_２を乗算している。なお、これら加算係数乗算器２３ａ_１〜２３ａ_５と１水平走査期間の単位遅延素子２５ａ_１〜２５ａ_４と加算集計器２７ａとからなるトランスバーサルフィルタが請求項に記載した第一瞬時位相検出手段に相当するものである。
【００３９】
同様に、加算係数乗算器２３ｂ_１が乗算器２１ｂからの出力である第二直交成分検波出力信号（２Ｈ進相信号）に加算係数Ｃ_２を乗算しており、加算係数乗算器２３ｂ_２が１水平走査期間の単位遅延素子２５ｂ_１からの出力であるタップ出力信号（１Ｈ進相信号）に加算係数Ｃ_１を乗算しており、加算係数乗算器２３ｂ_３が１水平走査期間の単位遅延素子２５ｂ_２からの出力である中央タップ出力信号に加算係数Ｃ_０を乗算している。また、加算係数乗算器２３ｂ_４が１水平走査期間の単位遅延素子２５ｂ_３からの出力であるタップ出力信号（１Ｈ遅延信号）に加算係数Ｃ_１を乗算しており、加算係数乗算器２３ｂ_５が１水平走査期間の単位遅延素子２５ｂ_４からの出力であるタップ出力信号（２Ｈ遅延信号）に加算係数Ｃ_２を乗算している。なお、これら加算係数乗算器２３ｂ_１〜２３ｂ_５と１水平走査期間の単位遅延素子２５ｂ_１〜２５ｂ_４と加算集計器２７ｂとからなるトランスバーサルフィルタが請求項に記載した第二瞬時位相検出手段に相当するものである。
【００４０】
加算集計器２７（図中２７ａ、２７ｂ［Σ：シグマ］）は、加算係数が乗算された第一および第二のトランスバーサルフィルタの各タップ出力信号を加算集計するものであり、加算集計器２７ａが第一直交成分検波出力信号を入力信号とする第一のトランスバーサルフィルタの各タップ出力信号を加算集計するものであり、加算集計器２７ｂが第二直交成分検波出力信号を入力信号とする第二のトランスバーサルフィルタの各タップ出力信号を加算集計するものである。後記するように加算係数Ｃ_０、Ｃ_１、Ｃ_２を一定の関係を満足するように設定すると、この図１に示したように、加算集計器２７ａの出力をＲ（ｔ）ｃｏｓ（β−θ）、加算集計器２７ｂの出力をＲ（ｔ）ｓｉｎ（β−θ）とすることができ、これによりＱ信号の瞬時位相が検出できる。
【００４１】
乗算器２９ａは、加算集計器２７ａからの出力Ｒ（ｔ）ｃｏｓ（β−θ）とｃｏｓ波（ｃｏｓ（δｔ＋θ））とを乗算するものである。
乗算器２９ｂは、加算集計器２７ｂからの出力Ｒ（ｔ）ｓｉｎ（β−θ）とｓｉｎ波（ｓｉｎ（δｔ＋θ））とを乗算するものである。
減算器３１は、乗算器２９ａの出力から乗算器２９ｂの出力を減算するものであり、この減算器３１の出力がＲ（ｔ）ｃｏｓ（δｔ＋β）、すなわち映像ビート成分となる。なお、図１の場合、乗算器２９ａ、２９ｂと減算器３１とが請求項に記載したビート成分出力手段に相当するものである。
【００４２】
（ビート成分生成部［ビート成分生成回路］における瞬時位相検出）
これより、図１に示したビート成分生成部１７Ａの瞬時位相検出について説明する。ここでは、まず、ビート成分生成部１７Ａよりも回路構成がより簡略な２系統の３タップトランスバーサルフィルタをその構成要素として備えるビート成分生成部１７Ｂ（図２参照）の瞬時位相の検出について説明してから、ビート成分生成部１７Ａの瞬時位相の検出について説明する。図２に示したように、ビート成分生成回路１７Ｂは、乗算器２１ａと、乗算器２１ｂと、発振器２１ｃと、加算係数乗算器２３と、１水平走査期間の単位遅延素子２５と、加算集計器２７と、乗算器２９ａと、乗算器２９ｂと、減算器３１とを備えている。なお、図１に示したビート成分生成部１７Ａと比較した場合、各構成において、差違がないものに関しては、同じ符号を付してその説明を省略する。
【００４３】
ところで、この図２に示したビート成分生成部１７Ｂと図１に示したビート成分生成部１７Ａとの違いは加算係数乗算器２３の数と１水平走査期間の単位遅延素子２５との数である。つまり、乗算器２１ａからの出力である第一直交成分検波出力信号の出力ラインに３個の加算係数乗算器２３ａ_１、２３ａ_２、２３ａ_３と、２個の１水平走査期間の単位遅延素子２５ａ_１、２５ａ_２とが備えられており、乗算器２１ｂからの出力である第二直交成分検波出力信号の出力ラインに３個の加算係数乗算器２３ｂ_１、２３ｂ_２、２３ｂ_３と、２個の１水平走査期間の単位遅延素子２５ｂ_１、２５ｂ_２とが備えられている。つまり、このビート成分生成部１７Ｂでは、第一直交成分検波出力信号および第二直交成分検波出力信号の出力ラインに、２個の１水平走査期間の単位遅延素子２５と３個の加算係数乗算器２３が備えられており、これによって、各タップ出力として、各３つのサンプル値（加算係数が乗算された数値）が得られる。また、ビート成分生成部１７Ａでは各５つのサンプル値が得られる。
【００４４】
（ビート成分生成部１７Ｂにおける瞬時位相検出）
このビート成分生成部１７Ｂでは、映像ビート成分を生成する際に妨害波映像信号の瞬時位相が検出される。ここで、Ｑ信号にオフセット周波数と同一周波数のｓｉｎ波を乗算したものをＸ、オフセット周波数と同一周波数のｃｏｓ波を乗算したものをＹとし、妨害波映像信号にはゴースト波が含まれているものとすると、Ｑは前記した（３）式、Ｒ（ｔ）ｓｉｎ（δｔ＋β）であり、

と表記でき、これらを（４）式とする。
【００４５】
この（４）式で表記できる信号を１Ｈ遅延線（図中、１ＨＤＬ）に通した信号をＸ_ｋ、Ｙ_ｋ（図２中、１ＨＤＬ２５ａ_１、２５ｂ_１の遅延線出力をｋ＝０とする）とし、１Ｈの遅延時間をτとし、（ｔ_ｋ＝ｔ−ｋτ）とすると、

と表記でき、同様にＹ_ｋも

と表記できる。
【００４６】
図３に（５）（６）式で表記される信号をベクトルとして図示する。各遅延線出力（Ｘ_ｋ、Ｙ_ｋ）は、（１／２Ｒｃｏｓ（β−θ）、１／２Ｒｓｉｎ（β−θ））を中心に半径１／２Ｒ、角速度２δで回転しているベクトルである。つまり、図３において、ｖ１はＰ１（Ｘ_０、Ｙ_０）を指すものであり、ｖ２はＰ２（Ｘ_１、Ｙ_１）を指すものであり、ｖ３はＰ３（Ｘ_−１、Ｙ_−１）を指すものである。ここで、Ｒ、βなどが各サンプル時刻（ｔ_ｋ）で同一の値を取るものと仮定すると、＋１Ｈ信号のベクトルと、−１Ｈ信号のベクトルと、０Ｈ信号の０．２８倍（＝−２ｃｏｓ（２δτ）＝−２ｃｏｓ（９８．０６度））のベクトルの和は、各ベクトルが時間とともに回転しても、一定点Ｐ４に留まる。このため、この点Ｐ４の座標位置から位相（β）と振幅（Ｒ）とを知ることができる。
【００４７】
図３を用いて行った説明を数式で表記すると、下記のようになる。

【００４８】
ここで、Ｒ（ｔ_１）＝Ｒ（ｔ_−１）＝Ｒ（ｔ_０）（＝Ｒ）、β（ｔ_１）＝β（ｔ_−１）＝β（ｔ_０）（＝β）と仮定しており、各遅延線出力を加算した信号Ｕおよび信号Ｖとすると、次式に示すようになる。

【００４９】
この（８）式において、係数Ｃ_１、Ｃ_２が次式を満たすような値を取ると、位相（β−θ）および振幅（Ｒ）を求めることができる。

【００５０】
すなわち、（９）式よりＣ_０＝−２Ｃ_１ｃｏｓ（２δτ）＝−２Ｃ_１ｃｏｓ（９８．０６度）＝０．２８Ｃ_１となることから、＋１Ｈ信号のベクトルと、−１Ｈ信号のベクトルと、０Ｈ信号の０．２８倍のベクトルの和は各ベクトルが時間とともに回転しても図３の一定点Ｐ４に留まり、位相（β−θ）が検出できる。
【００５１】
この（９）式を解いて
Ｃ_０＝−２ｃｏｓ（２δτ）／（１−ｃｏｓ（２δτ））＝０．２４５８６
Ｃ_１＝１／（１−ｃｏｓ（２δτ））＝０．８７７０７・・・（１０）式が得られ、この場合のβおよびＲは次式から求められる。
Ｕ＝Ｒｃｏｓ（β−θ）、Ｖ＝Ｒｓｉｎ（β−θ）、
【数１】

・・・（１１）式
【００５２】
なお、この実施の形態では、所望の放送波の搬送波周波数に対して不要な放送波の搬送波周波数がプラス側に１０．０１ｋＨｚだけずれている場合を想定しているが、不要な放送波の搬送波周波数がマイナス側に１０．０１ｋＨｚだけずれている場合には、Ｂ（ｔ）の搬送波はｃｏｓ｛（ω−δ）ｔ＋φ｝となるため、ＲＦ入力信号を表す（１）式はＲＦ入力信号＝Ａ（ｔ）ｃｏｓ（ωｔ）＋Ｒ（ｔ）ｃｏｓ｛（ω−δ）ｔ＋β｝となる。この場合には、Ｉ信号＝Ａ（ｔ）＋Ｒ（ｔ）ｃｏｓ（−δｔ＋β）、Ｑ信号＝Ｒ（ｔ）ｓｉｎ（−δｔ＋β）となり、（１１）式は、
Ｕ＝−Ｒｃｏｓ（β＋θ）、Ｖ＝Ｒｓｉｎ（β＋θ）、
【数１】

・・・（１１′）式となる。
【００５３】
以上の（４）式から（１１′）式では、βおよびＲが、各サンプル時刻（ｔ_ｋ）で同一の値をとるものと仮定したが、通常の映像信号では、大きな垂直相関を持っているので、この仮定は、高い精度で成立していると想定でき、ビート成分生成部１７Ｂでは、正確な映像ビート成分が生成できるといえる。
【００５４】
（ビート成分生成部１７Ａにおける瞬時位相検出）
次に、ビート成分生成部１７Ａにおける瞬時位相検出について説明する。
ビート成分生成部１７Ｂの瞬時位相検出の説明において記載した（１１）式で求められる瞬時位相β（ｔ）を用いて、従来の技術で説明したＰＬＬを用いたオフセットビートキャンセラーのＰＬＬ発振波の位相を瞬時瞬時に制御することにより、課題を解決することもできるが、妨害波映像信号の瞬時位相に追随した映像ビート成分はＰＬＬを用いなくても図１または図３に示す回路により直接生成することができる。次にこのことを説明する。
【００５５】
ビート成分生成部１７Ａで説明した加算集計器２７ａおよび加算集計器２７ｂの出力信号が、（１１）式で得られた信号に相当し、Ｕ信号にオフセット周波数と同一周波数のｃｏｓ波（ｃｏｓ（δｔ＋θ））を乗算器２９ａで乗算し、Ｖ信号にオフセット周波数と同一周波数のｓｉｎ波（ｓｉｎ（δｔ＋θ））を乗算器２９ｂで乗算し、減算器３１で減算すると、
Ｕ×ｃｏｓ（δｔ＋θ）−Ｖ×ｓｉｎ（δｔ＋θ）＝Ｒ（ｔ）ｃｏｓ｛（β−θ）＋（δｔ＋θ）｝＝Ｒ（ｔ）ｃｏｓ（δｔ＋β）・・・（１２）式となる。つまり、極めて簡単に映像ビート成分が生成できることになる。
【００５６】
ここで、所望の放送波の搬送波周波数に対し不要な放送波の搬送波周波数がマイナス側にずれている場合には、Ｖ×ｓｉｎ（δｔ＋θ）−Ｕ×ｃｏｓ（δｔ＋θ）＝Ｒ（ｔ）ｃｏｓ｛（β＋θ）−（δｔ＋θ）｝＝Ｒ（ｔ）ｃｏｓ（−δｔ＋β）・・・（１２′）式となって、極めて簡単に映像ビート成分が生成できる。つまり、所望の放送波の搬送波周波数に対して不要な放送波の搬送波周波数がプラス側にずれているかマイナス側にずれているかによって、減算器３１で乗算器２９ａの出力から乗算器２９ｂの出力を減算するか、乗算器２９ｂの出力から乗算器２９ａの出力を減算するかを決めてやれば、ビート成分生成部１７Ａから映像ビート成分を出力することができる。
【００５７】
この（１２）式および（１２′）式によると、発振器２１ｃの発振位相（θ）は最終的に生成される映像ビート成分に影響を及ぼしていないので、任意の値でよいことになる。従って、発振器２１ｃで発生する１０．０１ｋＨｚについては、従来のビート成分生成回路（図示せず）のＰＬＬ（Ｐｈａｓｅ ＬｏｃｋｅｄＬｏｏｐ）手段のように、入力波に位相ロックする必要がなくなり、フリーランの発振器でよいことになる。ＰＬＬのようなフィードバック系が不要であることは、ハードウエアの安定性や応答速度の観点からメリットが大きいと言える。
【００５８】
また、ビート成分生成部１７Ｂは、３サンプル加算（−１Ｈ、０Ｈ、＋１Ｈ）であるのに対し、ビート成分生成部１７Ａでは、５サンプル加算（−２Ｈ、−１Ｈ、０Ｈ、１Ｈ、２Ｈ）であり、各サンプル加算処理は、垂直周波数領域でのフィルタ処理であって、この特性は前記した加算係数で決定される。
【００５９】
ビート成分生成部１７Ｂの場合では、３サンプル加算であり、加算係数Ｃ_０、Ｃ_１は（９）式の拘束条件で一意的に決まってしまうので、図４（ａ）に示す周波数応答以外の特性をとることができない。しかし、ビート成分生成部１７Ａの場合では、５サンプル加算であり、拘束条件に対して加算係数の数が一つ多い。このため、各係数に自由度を持たせることができ、周波数特性を図４（ｂ）に示すように、ある程度自由に設定できる。なお、２Ｍ個の遅延線（１水平走査期間の単位遅延素子２５）を使用して２Ｍ＋１個のタップ出力の荷重和をトランスバーサルフィルタ出力とする場合の係数の拘束条件は、
【００６０】
【数２】

【００６１】
（１３）式となり、加算処理による雑音の増加は次式の（１４）式で与られる。
【００６２】
【数３】

（１４）式
なお、トランスバーサルフィルタの各加算係数は理論上、前記の拘束条件により制限を受けるが、現実的には総合的なビートキャンセル動作を見ながらの調整となるため、実用上は一定の範囲内に設定されていればよいという性質のものである。
【００６３】
（トランスバーサルフィルタを偶数タップで構成した場合について）
以上では、トランスバーサルフィルタを奇数タップで構成した場合を例にビート成分生成部１７Ａの動作を説明したが、トランスバーサルフィルタを偶数タップで構成しても奇数タップのトランスバーサルフィルタの場合と同様の効果を得ることが可能であることを次に補足的に説明する。
【００６４】
図５に偶数タップのトランスバーサルフィルタでビート成分生成部１７Ｃを構成した場合の放送波受信チューナ１Ａのブロック図を示す。この図５に示したように、ビート成分生成部１７Ｃは、乗算器２１ａと、乗算器２１ｂと、発振器２１ｃと、加算係数乗算器２３と、１水平走査期間の単位遅延素子２５（図５中、１ＨＤＬ）と、加算集計器２７と、乗算器２９ａと、乗算器２９ｂと、減算器３１とを備えている。加算係数乗算器２３は、第一直交成分検波出力信号および第二直交成分検波出力信号に加算係数（Ｃ_１．５）を乗算すると共に、１水平走査期間の単位遅延素子２５の各出力信号に加算係数（Ｃ_０．５、Ｃ_１．５）を乗算するものである。なお、図５に示した放送波受信チューナ１Ａのビート成分生成部１７Ｃと、図１に示した放送波受信チューナ１のビート成分生成部１７Ａとを比較した場合、各構成において、差違がないものに関しては、同じ符号を付してその説明を省略する。
【００６５】
図５に示したビート成分生成部１７Ｃと図１に示したビート成分生成部１７Ａとの違いは、加算係数乗算器２３の個数と１水平走査期間の単位遅延素子２５の個数である。つまり、乗算器２１ａからの出力である第一直交成分検波出力信号の出力ラインに４個の加算係数乗算器２３ａ_１、２３ａ_２、２３ａ_３、２３ａ_４と、３個の１水平走査期間の単位遅延素子２５ａ_１、２５ａ_２、２５ａ_３とが備えられており、乗算器２１ｂからの出力である第二直交成分検波出力信号の出力ラインに４個の加算係数乗算器２３ｂ_１、２３ｂ_２、２３ｂ_３、２３ｂ_４と、３個の１水平走査期間の単位遅延素子２５ｂ_１、２５ｂ_２、２５ｂ_３とが備えられている。つまり、このビート成分生成部１７Ｃでは、第一直交成分検波出力信号および第二直交成分検波出力信号の出力ラインに３個の１水平走査期間の単位遅延素子２５と４個の加算係数乗算器２３が備えられており、これによって、各タップ出力として、各４つのサンプル値（加算係数が乗算された数値）が得られる。
【００６６】
具体的には、図５に示したように、加算係数乗算器２３ａ_１が乗算器２１ａからの出力信号である第一直交成分検波出力信号（１．５Ｈ進相信号）に加算係数Ｃ_１．５を乗算しており、加算係数乗算器２３ａ_２が１水平走査期間の単位遅延素子２５ａ_１からの出力であるタップ出力信号（０．５Ｈ進相信号）に加算係数Ｃ_０．５を乗算しており、加算係数乗算器２３ａ_３が１水平走査期間の単位遅延素子２５ａ_２からの出力であるタップ出力信号（０．５Ｈ遅延信号）に加算係数Ｃ_０．５を乗算しており、加算係数乗算器２３ａ_４が１水平走査期間の単位遅延素子２５ａ_３からの出力であるタップ出力信号（１．５Ｈ遅延信号）に加算係数Ｃ_１．５を乗算している。
【００６７】
同様に加算係数乗算器２３ｂ_１が乗算器２１ｂからの出力信号である第二直交成分検波出力信号（１．５Ｈ進相信号）に加算係数Ｃ_１．５を乗算しており、加算係数乗算器２３ｂ_２が１水平走査期間の単位遅延素子２５ｂ_１からの出力であるタップ出力信号（０．５Ｈ進相信号）に加算係数Ｃ_０．５を乗算しており、加算係数乗算器２３ｂ_３が１水平走査期間の単位遅延素子２５ｂ_２からの出力であるタップ出力信号（０．５Ｈ遅延信号）に加算係数Ｃ_０．５を乗算しており、加算係数乗算器２３ｂ_４が１水平走査期間の単位遅延素子２５ｂ_３からの出力であるタップ出力信号（１．５Ｈ遅延信号）に加算係数Ｃ_１．５を乗算している。
【００６８】
ここで、図５のトランスバーサルフィルタの入力信号は、Ｑ信号にオフセット周波数と同一周波数のｓｉｎ波を乗算したＸおよびオフセット周波数と同一周波数のｃｏｓ波を乗算したＹであり、これらの信号は前記（４）式で表記される。この（４）式で表記される信号を１Ｈ遅延線（図５中、１ＨＤＬ）に通した信号Ｘ_ｋ、Ｙ_ｋもまた（５）式および（６）式の通りになる。
【００６９】
図５中、１ＨＤＬ２５ａ_１、２５ｂ_１の遅延線出力をそれぞれＸ_−０．５、Ｙ_−０．５と表し、１ＨＤＬ２５ａ_２、２５ｂ_２の遅延線出力をそれぞれＸ_０．５、Ｙ_０．５と表すと、各遅延線出力を加算した信号ＵおよびＶは次のように表される。

【００７０】
この（１５）式において、係数Ｃ_０．５、Ｃ_１．５が次式を満たすような数値をとると、位相（β−θ）および振幅（Ｒ）を求めることができる。

このようにトランスバーサルフィルタを偶数タップで構成した場合、トランスバーサルフィルタを奇数で構成した場合と比較して瞬時位相を検出するための各加算係数の拘束条件に差違は生じるが、その基本的な動作に差違が生じるものではない。
【００７１】
（放送波受信チューナの動作）
次に、図６に示すフローチャートを参照して、放送波受信チューナ１の動作を説明する（適宜図１を参照）。
まず、放送波受信チューナ１では、受信アンテナ（図示せず）から直交検波部３にＲＦ入力信号Ａ（ｔ）ｃｏｓ（ωｔ）＋Ｒ（ｔ）ｃｏｓ｛（ω＋δ）ｔ＋β｝が入力されたかどうか判断される（Ｓ１）。このＲＦ入力信号が入力されるまで（Ｎｏ）待機され、ＲＦ入力信号が入力されたと判断された場合（Ｙｅｓ）、直交検波部３でＲＦ入力信号が直交検波され、希望波搬送波と同相成分であるＩ信号、直交成分であるＱ信号とに分離される。直交検波部３で分離された同相成分であるＩ信号の場合（Ｓ２、Ｙｅｓ）、分離されたＩ信号は同相成分Ａ／Ｄ変換部５でアナログデジタル変換される（Ｓ３）。
【００７２】
そして、この同相成分Ａ／Ｄ変換部５でアナログデジタル変換されたＩ信号は、ビート成分生成部１７Ａの出力信号との相対的な時間関係を合わせるために同相成分遅延部７（図１中２ＨＤＬ）で２Ｈ分遅延される（Ｓ４）。この同相成分遅延部７で遅延された信号は分岐され、高周波成分である場合（Ｓ５、Ｙｅｓ）、高周波成分抽出遅延部９で当該高周波成分が抽出され、遅延調整される（Ｓ６）。また、低周波成分である場合（Ｓ５、Ｎｏ）、低周波成分抽出遅延部１１で当該低周波成分が抽出され、遅延調整される（Ｓ７）。
【００７３】
また、直交検波部３で分離された直交成分であるＱ信号の場合（Ｓ２、Ｎｏ）、分離されたＱ信号は、直交成分Ａ／Ｄ変換部１３でアナログデジタル変換される（Ｓ８）。そして、低周波直交成分抽出部１５（図１中ＬＰＦ）で直交成分の低周波成分が抽出される（Ｓ９）。その後、この直交成分の低周波成分がビート成分生成部１７Ａに入力され、このビート成分生成部１７Ａで映像ビート成分Ｒ（ｔ）ｃｏｓ（δｔ＋β）が生成される（Ｓ１０）。
【００７４】
そして、映像信号出力部１９で、低周波成分抽出遅延部１１にて遅延調整されている低周波成分からビート成分生成部１７Ａにて生成された映像ビート成分が減算される（Ｓ１１）。さらに、この映像信号出力部１９で、低周波成分から映像ビート成分が減算された信号と、高周波成分抽出遅延部９にて遅延調整されている高周波成分とが加算され、デジタル映像出力Ａ（ｔ）として出力される（Ｓ１２）。
【００７５】
（ビート成分生成部［ビート成分生成回路］の動作）
次に、図７に示すフローチャートを参照して、ビート成分生成部１７Ａにおいて映像ビート成分（オフセットビート成分）を生成する動作を説明する（適宜図１参照）。
まず、ビート成分生成部１７Ａでは、Ｑ信号Ｒ（ｔ）ｓｉｎ（δｔ＋β）の入力があるかどうかが判断される（Ｓ２１）。このＱ信号の入力があるまで待機（Ｎｏ）され、Ｑ信号の入力があると判断された場合（Ｙｅｓ）には、乗算器２１ａでオフセット周波数と同一周波数のｓｉｎ波が、乗算器２１ｂでオフセット周波数と同一周波数のｃｏｓ波がそれぞれＱ信号に乗算され、第一直交成分検波出力信号、第二直交成分検波出力信号とされる（Ｓ２２）。
【００７６】
これら第一直交成分検波出力信号、第二直交成分検波出力信号がそれぞれ加算係数乗算器２３ａ_１、２３ｂ_１と、１水平走査期間の単位遅延素子２５ａ_１、２５ｂ_１に入力される（Ｓ２３）。加算係数乗算器２３ａ_１、２３ｂ_１では、加算係数Ｃ_２が乗算され、１水平走査期間の単位遅延素子２５ａ_１、２５ｂ_１では、第一直交成分検波出力信号、第二直交成分検波出力信号が１水平走査期間だけ遅延される。
【００７７】
そして、これら１Ｈだけ遅延された第一直交成分検波出力信号、１Ｈだけ遅延された第二直交成分検波出力信号がそれぞれ加算係数乗算器２３ａ_２、２３ｂ_２と、１水平走査期間の単位遅延素子２５ａ_２、２５ｂ_２に入力される（Ｓ２４）。さらに、１水平走査期間の単位遅延素子２５ａ_２からの出力信号は、加算係数乗算器２３ａ_３、１水平走査期間の単位遅延素子２５ａ_３に、続いて、１水平走査期間の単位遅延素子２５ａ_３からの出力信号は、加算係数乗算器２３ａ_４、１水平走査期間の単位遅延素子２５ａ_４に入力され、１水平走査期間の単位遅延素子２５ａ_４からの出力信号は加算係数乗算器２３ａ_５に入力される。また、１水平走査期間の単位遅延素子２５ｂ_２からの出力信号は、加算係数乗算器２３ｂ_３、１水平走査期間の単位遅延素子２５ｂ_３に、続いて、１水平走査期間の単位遅延素子２５ｂ_３からの出力信号は、加算係数乗算器２３ｂ_４、１水平走査期間の単位遅延素子２５ｂ_４に入力され、１水平走査期間の単位遅延素子２５ｂ_４からの出力信号は加算係数乗算器２３ｂ_５に入力される。
【００７８】
その後、加算集計器２７ａにおいて、加算係数乗算器２３ａ_１、２３ａ_２、２３ａ_３、２３ａ_４、２３ａ_５で加算係数が乗算された各タップ出力信号が集計され、加算集計器２７ｂにおいて、加算係数乗算器２３ｂ_１、２３ｂ_２、２３ｂ_３、２３ｂ_４、２３ｂ_５で加算係数が乗算された各タップ出力信号が集計される（Ｓ２５）。さらに、乗算器２９ａにおいて加算集計器２７ａの出力信号である第一トランスバーサルフィルタ出力信号とオフセット周波数と同一周波数のｃｏｓ波とが乗算され、乗算器２９ｂにおいて加算集計器２７ｂの出力信号である第二トランスバーサルフィルタ出力信号とオフセット周波数と同一周波数のｓｉｎ波とが乗算される（Ｓ２６）。それらの結果である乗算器２９ａおよび乗算器２９ｂの出力信号が減算され、映像ビート成分として映像信号出力部１９に出力される（Ｓ２７）。
【００７９】
ここで、所望の放送波の搬送波周波数に対して不要な放送波の搬送波周波数がプラス側にずれている場合は乗算器２９ａの出力から乗算器２９ｂの出力が減算され、所望の放送波の搬送波周波数に対して不要な放送波の搬送波周波数がマイナス側にずれている場合は乗算器２９ｂの出力から乗算器２９ａの出力が減算される。なお、所望の放送波の搬送波周波数に対して不要な放送波の搬送波周波数がプラス側にずれているかマイナス側にずれているかは、これらの放送波を送信する放送局の設定により定まるものであるので、本発明の一実施の形態である放送波受信チューナ１、１Ａを設置しようとする受信点が決定すれば自ずと定まり、これらの関係を既知のものとして取り扱うことができる。
【００８０】
（ビート成分生成部における加算係数（サンプル加算）について）
ここで、ビート成分生成部１７Ａ、１７Ｂにおける加算係数乗算器２３で乗算される加算係数（サンプル加算）について補足説明しておく。（４）式に示した式（Ｑ信号にオフセット周波数と同一周波数のｓｉｎ波、オフセット周波数と同一周波数のｃｏｓ波を乗算したもの）は、直交検波の一般形式を与えている。つまり、入力信号であるＱ信号は周波数δの搬送波がＲ（ｔ）という信号で振幅変調され、さらにβ（ｔ）という信号で位相変調されたものであるとみなすことができる。そして、この変調された信号（Ｑ信号）に搬送波と同一周波数のキャリアを乗じたものが（４）式に示したＸおよびＹである。
【００８１】
通常の信号復調では、搬送波の２倍の周波数成分である（４）式の各式第２項がフィルタ（図示せず）で除去されて復調成分（第１項）が抽出されている。この動作を図３に示したベクトル図で説明すると以下の通りになる。すなわち、キャリア（オフセット周波数と同一周波数のｓｉｎ波、オフセット周波数と同一周波数のｃｏｓ波）を乗算した信号は、図３の０Ｈ信号で表される回転ベクトルであるが、これをフィルタ（図示せず）に通して第２項の高周波成分を除去することは回転ベクトルを１回転期間（２δの１周期）に亘って積分することと等価である。つまり、積分によりベクトルの回転成分が平均化されてゼロとなり、回転中心である第１項を復調出力として取り出すことができる。
【００８２】
これに対し、サンプル加算は、回転ベクトルを１周期に亘って（連続）積分する代わりに、ベクトルを定期的にサンプリングし、各サンプル値の重み付け和（ｗｅｉｇｈｔｅｄ ｓｕｍ）を求めるものである。１水平走査期間の単位遅延素子２５（図１参照）によりトランスバーサルフィルタを構成し、サンプリング周波数として水平走査周波数（ｆ_ｈ＝１５．７３ｋＨｚ）を用いているため、重み付け和の周波数特性は、ｆ_ｈで繰り返す櫛型フィルタ特性をとることになる。
【００８３】
一例として、±２Ｈ遅延線を用いた場合の特性を図８に示す。この櫛型フィルタにおいて、サンプル加算の拘束条件（１３）式が成立することは、オフセット周波数の２倍の周波数（高周波キャリア＝２０．０２ｋＨｚ）でのレスポンスがゼロ（図８中ほぼ中央の黒点、周波数２０ｋＨｚ付近）となることである。またこの時、櫛型フィルタの特性に基づいて、高周波キャリアからｆ_ｈの整数倍離れた周波数に対するレスポンスもゼロ（図８中、周波数２δ＋ｆ_ｈおよび２δ−ｆ_ｈ）となる。
【００８４】
ところで、垂直相関が１００％である映像信号とは、周波数成分がｆ_ｈの整数倍にのみ存在する信号であるが、妨害波映像信号がこのような垂直相関１００％の信号である場合、映像信号成分（（４）式第１項）は、ｆ_ｈの整数倍の周波数にのみ成分を持ち、高周波成分（（４）式第２項）については、高周波キャリアからｆ_ｈの整数倍離れた周波数にのみ成分を持つ信号となる。このような信号を図８に示した特性を有するフィルタに通過させると映像信号成分は１００％（無歪で）通過し、高周波成分は完全に遮断され、所望の映像信号成分のみを取り出すことができる。
【００８５】
また、逆に、垂直相関が１００％でない映像信号は、ｆ_ｈの整数倍以外の周波数にも成分を持っており、このような映像信号の場合、映像信号成分がフィルタの周波数特性の影響を受けて歪む（線形歪）と共に、高周波成分はフィルタの通過域に成分を持つようになる。このため、映像信号成分と高周波成分との混ざり合ったものがフィルタ出力となり、映像信号成分と高周波成分との分離が不十分なものとなってしまう。
【００８６】
つまり、映像信号成分が歪んだり、映像信号成分と高周波成分との分離が不十分であることは、妨害波映像信号の振幅および位相（Ｒとβ）の検出が不完全であることであり、従って、これらから生成される映像ビート成分には誤差が含まれることになり、放送波受信チューナ１では正確なビートキャンセル動作が行えなくなる。
【００８７】
そこで、映像信号成分の歪みや高周波成分分離の問題を緩和する１つの方法として、オフセット周波数の変更がある。このオフセット周波数を変更した例を図９に示す。この図９（ｂ）は、現行のオフセット周波数１０．０１ｋＨｚの場合であり、図９（ａ）は、オフセット周波数をｆ_ｈの３／４倍にした場合、すなわち、高周波成分がｆ_ｈの整数倍（この周波数に信号のエネルギーが集中している）の中間になるようにした場合である。図９（ｂ）と比較すると、明らかなように、信号の通過帯域が広く（信号歪みが少ない）、また、高周波キャリアがｆ_ｈの整数倍の周波数から最も離れているため、映像信号成分と高周波成分との分離も比較的優れている。なお、オフセット周波数を変更するには、既存設備の設定変更が必要であることは言うまでもない。
【００８８】
（フレーム相関を利用したビート成分生成部について）
これまでは、映像信号の垂直相関が高いという性質を利用したビート成分生成部１７Ａ、１７Ｂについて説明してきた。これより、映像信号のフレーム相関を利用したビート成分生成部（図示せず）について説明する。通常の映像信号はフレーム相関も高く、例えば、静止画ではフレーム相関は１００％となる。この静止画の場合、映像信号成分はフレーム周波数（２９．９７Ｈｚ：以下３０Ｈｚと略記する）の整数倍にのみ存在する。
【００８９】
このフレーム相関を利用する場合について、図１０を参照して説明する。フレーム相関を利用する場合は、垂直相関を利用する場合に用いた１水平走査期間の単位遅延素子の代わりに、１フィールドまたは１フレーム期間の単位遅延素子を利用する。具体的には図１に示したビート成分生成部１７Ａの「１ＨＤＬ」を「１フィールド遅延線（１ＶＤＬ）または１フレーム遅延線（１ＦＤＬ）」（図示せず）に置き換えたものである。なお、図１の放送波受信チューナ１でビートキャンセル動作を行わせるには、単位遅延素子の変更に合わせ同期成分遅延部７の「２ＨＤＬ」を「１ＦＤＬ」または「２ＦＤＬ」に変更することが必要である。図１０は、ビート成分生成部１７Ａの「１ＨＤＬ」を「１フィールド遅延線（１ＶＤＬ）」に置き換えた場合の動作説明図である。
【００９０】
現行のオフセット周波数１０．０１ｋＨｚは、フィールド周波数の１６７倍（またはフレーム周波数の３３４倍）であるため、フィールド遅延しても回転ベクトルは現信号ベクトルと同一になってしまう。その結果、これらの信号をどのように加算しても回転ベクトル成分を相殺することができない。つまり、現行オフセット周波数のままでは、フレーム相関を利用することができない。このため、オフセット周波数を７．５Ｈｚ（垂直周波数（ｆ_ｖ）の１／８）だけシフトさせる。このオフセット周波数の場合、図１０（ａ）に示すように、１フレーム遅延信号および１フレーム進相信号の回転ベクトルｖｒ２、ｖｒ３は現信号の回転ベクトルｖｒ１と逆相関係となり、これらを一定の加算係数で重み付け加算することにより回転ベクトルの相殺が可能になる。
【００９１】
すなわち、オフセット周波数を７．５Ｈｚだけシフトさせた場合には、１フィールド期間の単位遅延素子を構成要素とする５タップのトランスバーサルフィルタまたは１フレーム期間の単位遅延素子を構成要素とする３タップまたは５タップのトランスバーサルフィルタを用いれば、１水平走査期間の単位遅延素子を構成要素とするトランスバーサルフィルタを用いた場合と同様に映像ビート成分を生成することができる。同様に、オフセット周波数を１５Ｈｚだけシフトさせた場合には、１フィールド遅延信号および１フィールド進相信号の回転ベクトルは現信号の回転ベクトルと逆相関係になるため、１フィールド期間の単位遅延素子を構成要素とする３タップまたは５タップのトランスバーサルフィルタを用いれば、映像ビート成分を生成することができる。ただし、いずれの場合も既存設備の設定変更が必要である。
【００９２】
この実施の形態では以下の効果を奏す。
放送波受信チューナ１では、受信した放送波映像信号が直交検波部３によって直交検波され、同相成分と直交成分とに分離される。一方の同相成分は同相成分Ａ／Ｄ変換部５によってアナログデジタル変換され、同相成分遅延部７で遅延され、高周波成分抽出遅延部９によって高周波成分が取り出され、遅延される。また、低周波成分抽出遅延部１１によって低周波成分が取り出され、遅延される。他方の直交成分は直交成分Ａ／Ｄ変換部１３によってアナログデジタル変換され、低周波直交成分抽出部１５によって、直交成分の低周波成分が取り出される。そして、ビート成分生成部１７Ａによって直交成分の低周波成分から妨害波映像信号の瞬時位相の変化に追随した映像ビート成分が生成され、映像信号出力部１９によって、同相成分の低周波成分から映像ビート成分が減算され、さらに同相成分の高周波成分と加算されデジタル映像信号として出力される。このため、妨害波映像信号の瞬時位相が変化した場合でも映像ビート成分が除去された希望波映像信号が得られ、放送波受信チューナ１のユーザは放送局から放送される所望の放送番組を良好な状態で視聴することができる。
【００９３】
また、ビート成分生成部１７Ａ、１７Ｂ（ビート成分生成回路）では、乗算器２１ａと乗算器２１ｂと発振器２１ｃとによって得られた第一直交成分検波出力信号と第二直交成分検波出力信号とに、加算係数乗算器２３で加算係数が乗算されつつ、１水平走査期間の単位遅延素子２５で順次遅延され、他の加算係数が乗算され集計される。そして、これらの検出結果に基づいて加算集計器２７ａ、２７ｂと乗算器２９ａ、２９ｂと減算器３１とにより、妨害波映像信号の瞬時位相の変化に対応した映像ビート成分を出力することができる。
【００９４】
さらに、オフセット周波数を現行周波数から若干ずらすことを前提とした場合、トランスバーサルフィルタを構成する単位遅延素子を１水平走査期間の遅延素子から１フィールド期間または１フレーム期間の遅延素子に置き換えることにより、フレーム相関を利用した映像ビート成分の生成が可能となる。例えば、オフセット周波数を７．５Ｈｚ（垂直周波数（ｆ_ｖ）の１／８）だけシフトさせれば、１フレーム遅延信号および１フレーム進相信号を用いた瞬時位相検出が可能となり、特に静止画において完全なビートキャンセル動作を行わせることができる。
【００９５】
３個ずつの加算係数乗算器２３が備えられた場合（ビート成分生成部１７Ｂの場合）、乗算結果として加算される値（サンプル値）は３個とされ、回路をより簡略に構成することができる。５個ずつの加算係数乗算器２３が備えられた場合（ビート成分生成部１７Ａの場合）、乗算結果として加算される値（サンプル値）は５個とされ、加算係数に自由度を持たすことができる。なお、このサンプル値の増加に伴い、加算係数の自由度も増加する。
【００９６】
さらに、不要な放送波が複数存在する場合は、それぞれのオフセット周波数が同一とならないよう、各放送局のオフセット周波数をシフトさせておけば、本発明の一実施の形態であるビート成分生成部１７Ａ、１７Ｂ、１７Ｃ（ビート成分生成回路）を不要な放送波に対応して複数用意することにより、別々に映像ビート成分を生成することが可能になり、それらを線形演算することによって、ビートキャンセル動作を実行することができる。
【００９７】
以上、一実施形態に基づいて本発明を説明したが、本発明はこれに限定されるものではない。
ビート成分生成部１７Ａ、１７Ｂ、放送波受信チューナ１の各構成の処理を各過程（ステップ）とみなしたビート成分生成方法、放送波受信方法と捉えることも可能である。この場合、ビート成分生成部１７Ａ、１７Ｂ、放送波受信チューナ１のそれぞれの効果と同様の効果が得られる。また、前記したように、トランスバーサルフィルタを偶数タップで構成しても図８、図９、図１０（ｂ）と同等の周波数特性を得ることができるため、本発明のトランスバーサルフィルタは、この実施の形態で詳細に説明した奇数タップのトランスバーサルフィルタに限定されるものではない。
【００９８】
【発明の効果】
請求項１記載の発明によれば、第一直交成分検波出力手段と第二直交成分検波出力手段とによって得られた第一直交成分検波出力信号と第二直交成分検波出力信号とに、第一瞬時位相検出手段および第二瞬時位相検出手段でトランスバーサルフィルタによる信号処理がそれぞれ施されて妨害波映像信号の垂直相関を利用した瞬時位相検出が行われ、さらに、これらの検出結果に基づいてビート成分出力手段で映像ビート成分が出力される。このため、この映像ビート成分を同相成分の映像信号から減算すれば、妨害波映像信号の瞬時位相が変化した場合でも映像ビート成分が除去された、つまり妨害映像のない良好な希望波映像信号を得ることができる。
【００９９】
請求項２記載の発明によれば、オフセットキャリア方式におけるオフセット周波数を現行の１０．０１ｋＨｚから例えば、７．５Ｈｚ（垂直周波数（ｆ_ｖ）の１／８）だけシフトした場合に、妨害波映像信号のフレーム相関を利用した瞬時位相検出が可能となり、特に静止画において完全なビートキャンセル動作を行わせることができる。
【０１００】
請求項３記載の発明によれば、３タップのトランスバーサルフィルタとした場合には回路規模を小さくすることができ、５タップのトランスバーサルフィルタとした場合には瞬時位相を検出するための拘束条件によって制限を受ける各タップ出力に乗算される各加算係数に設定の自由度を持たせることができる。
【０１０１】
請求項４記載の発明によれば、直交検波手段によって分離された同相成分は、高周波成分抽出遅延手段によって高周波成分が取り出され、低周波成分抽出遅延手段によって低周波成分が取り出され、遅延される。他方の直交成分は直交成分低周波抽出手段によって、直交成分の低周波成分が取り出される。そして、ビート成分生成回路によって直交成分の低周波成分から映像ビート成分が生成され、映像信号出力手段によって、同相成分の低周波成分から映像ビート成分が減算され、さらに同相成分の高周波成分と加算され出力される。このため、放送局の放送区域内外における混信妨害を防ぐことができ、特に妨害波映像信号がゴースト波成分を伴っている場合に直接妨害波とゴースト妨害波との合成波である妨害波映像信号の瞬時位相が混信妨害を与える映像信号の振幅変動に伴い変化するオフセットビート（映像ビート成分）を除去することができ、放送波受信チューナのユーザは放送局から放送される所望の放送番組を良好な状態で視聴することができる。
【図面の簡単な説明】
【図１】本発明による一実施の形態である放送波受信チューナのブロック図である。
【図２】ビート成分生成部の他の構成を示すのブロック図である。
【図３】図２に示したビート成分生成部によるサンプル加算処理を説明するベクトル図である。
【図４】図１および図２に示したビート成分生成部の周波数特性を説明した図である。
【図５】図１に示したビート成分生成部を偶数タップのトランスバーサルフィルタにより構成した場合の放送波受信チューナのブロック図である。
【図６】図１に示した放送波受信チューナの動作を説明したフローチャートである。
【図７】図１に示したビート成分生成部の動作を説明したフローチャートである。
【図８】図２に示したビート成分生成部によるサンプル加算処理の周波数特性を説明した図である。
【図９】図２に示したビート成分生成部によるサンプル加算処理の周波数特性およびオフセット周波数をシフトした場合の周波数特性を説明した図である。
【図１０】（ａ） オフセット周波数を現行周波数から７．５Ｈｚシフトさせた上で、ビート成分生成部の５タップのトランスバーサルフィルタを１フィールド期間の単位遅延素子により構成した場合のベクトル図である。
（ｂ） オフセット周波数を現行周波数から７．５Ｈｚシフトさせた上で、ビート成分生成部の５タップのトランスバーサルフィルタを１フィールド期間の単位遅延素子により構成した場合の周波数特性を説明した図である。
【符号の説明】
１、１Ａ 放送波受信チューナ
３ 直交検波部（直交検波手段）
５ 同相成分Ａ／Ｄ変換部（同相成分Ａ／Ｄ変換手段）
７ 同相成分遅延部
９ 高周波成分抽出遅延部（高周波成分抽出遅延手段）
１１ 低周波成分抽出遅延部（低周波成分抽出遅延手段）
１３ 直交成分Ａ／Ｄ変換部（直交成分Ａ／Ｄ変換手段）
１５ 低周波直交成分抽出部（低周波直交成分抽出手段）
１７Ａ、１７Ｂ、１７Ｃ ビート成分生成部（ビート成分生成回路）
１９ 映像信号出力部
２１ａ、２１ｂ 乗算器（第一および第二直交成分検波出力手段）
２１ｃ 発振器（第一および第二直交成分検波出力手段）
２３ 加算係数乗算器（第一および第二瞬時位相検出手段）
２５ １水平走査期間の単位遅延素子
２７ａ、２７ｂ 加算集計器（第一および第二瞬時位相検出手段）
２９ａ、２９ｂ 乗算器（ビート成分出力手段）
３１ 減算器（ビート成分出力手段）[0001]
BACKGROUND OF THE INVENTION
The present invention includes a broadcast wave video signal to be received when there is a desired broadcast wave video signal and a disturbing broadcast wave video signal that is broadcast from a different broadcast station outside the broadcast area with the carrier frequency of the same channel slightly shifted. The present invention relates to a beat component generation circuit used for removing a video beat component generated by interference of a disturbing broadcast wave video signal with a broadcast wave video signal at the time of detection, and a broadcast wave receiving tuner including this circuit.
[0002]
[Prior art]
A conventional offset beat canceller (“beat component generation circuit and broadcast wave reception tuner including this circuit” Japanese Patent Application No. 2001-378206) generates a video beat component included in a broadcast wave video signal and generates the broadcast wave. It can be removed from the video signal and can prevent interference interference inside and outside the broadcasting area of the broadcasting station.
[0003]
[Problems to be solved by the invention]
However, in the conventional offset beat canceller, a large beat suppression effect can be obtained when the phase of the interfering wave video signal that causes the video beat component is constant, but the ghost wave component is included in the interfering wave video signal When the phase of the composite wave component of the orthogonal wave and the ghost wave changes instantaneously due to the amplitude fluctuation of the video signal that causes interference interference (when the instantaneous phase of the interference wave video signal including the ghost wave component changes) The conventional offset beat canceller PLL cannot follow this change. Therefore, the beat component generation circuit of the conventional offset beat canceller cannot generate a video beat component having the same phase as the video beat component generated by interfering with the broadcast wave video signal at the time of detecting the disturbing broadcast wave video signal. The video beat component cannot be removed. This left room for further improvement.
[0004]
Accordingly, an object of the present invention is to solve the above-described problems of the prior art, and includes a beat component generation circuit capable of generating a video beat component with high accuracy when the instantaneous phase of an interfering wave video signal changes, and this circuit. It is to provide a broadcast wave receiving tuner.
[0005]
[Means for Solving the Problems]
In order to achieve the above-described object, the present invention is configured as follows.
The beat component generation circuit according to claim 1 is a beat component generation circuit that generates a video beat component that is included in a received broadcast wave video signal and is generated by interference of the disturbing broadcast wave video signal with the broadcast wave video signal. The first quadrature component obtained by multiplying the quadrature component obtained by quadrature detection of the broadcast wave video signal by a sine wave having the same frequency as the offset frequency generated by the oscillator to obtain a first quadrature component detection output signal Detection output means, and second orthogonal component detection output means for multiplying the orthogonal component by a cosine wave having the same frequency as the offset frequency obtained by shifting the phase of the sine wave by π / 2 to obtain a second orthogonal component detection output signal A first transversal filter to which the first quadrature component detection output signal obtained by the first quadrature component detection output means is input, and a plurality of unit delay elements for one horizontal scanning period are provided. The first instantaneous phase detection means composed of a transversal filter configured to obtain the load sum of each tap output and the second quadrature component detection output signal obtained by the second quadrature component detection output means are input. A second transversal filter comprising a plurality of unit delay elements for one horizontal scanning period and configured to obtain a load sum of each tap output; Subtract the calculation output obtained by multiplying the second detection result obtained by the second instantaneous phase detection means by the sin wave from the calculation output obtained by multiplying the first detection result obtained by the time phase detection means by the cos wave, or the second detection result. A beat component output that subtracts a calculation output obtained by multiplying the first detection result by the cosine wave from a calculation output obtained by multiplying the sin wave by the beat output. And means.
[0006]
According to this configuration, the first instantaneous component detection output signal and the second orthogonal component detection output signal obtained by the first quadrature component detection output means and the second quadrature component detection output means, Signal processing by a transversal filter is performed by the phase detection means and the second instantaneous phase detection means to perform instantaneous phase detection using the vertical correlation of the interfering wave video signal, and further, beat component output based on these detection results The video beat component is output by the means.
[0007]
Here, the offset frequency is the difference in carrier frequency between the desired broadcast wave video signal and the disturbing broadcast wave video signal broadcast from a different broadcast station outside the broadcast area with the carrier frequency of the same channel slightly shifted. Since each carrier frequency shifted by ± 10.01 kHz with respect to the reference carrier frequency is currently defined as the same channel as the reference carrier frequency, the offset frequency described here is Can be in two ways: 10.1 kHz and 20.02 kHz. In addition, in the transversal filters that constitute the first instantaneous phase detection means and the second instantaneous phase detection means, each addition coefficient that is multiplied by each tap output to obtain the load sum of each tap output represents the instantaneous phase. It is restricted by the constraint conditions for detection.
[0008]
According to a second aspect of the present invention, there is provided a beat component generation circuit according to the first aspect, wherein the first instantaneous phase detection means and the second instantaneous phase detection means respectively constitute the first instantaneous phase detection means and the second instantaneous phase detection means. The unit delay element of the transversal filter is replaced by a delay element of one field period or one frame period instead of a delay element of one horizontal scanning period.
[0009]
According to this configuration, when the deviation from the reference carrier frequency in the offset carrier system is shifted from the current 10.01 kHz by, for example, 7.5 Hz (1/8 of the vertical frequency (fv)), the disturbance video signal Instantaneous phase detection using frame correlation can be performed, and a complete beat cancel operation can be performed particularly in a still image.
[0010]
A beat component generation circuit according to claim 3 is the beat component generation circuit according to claim 1 or 2, wherein the first instantaneous phase detection means and the second instantaneous phase detection means respectively constitute first and second beat phase generation means. The number of taps of the second transversal filter is 3 taps or 5 taps.
[0011]
According to this configuration, the circuit scale can be reduced when a 3-tap transversal filter is used, and when the 5-tap transversal filter is used, the circuit is limited by the constraint condition for detecting the instantaneous phase. Each addition coefficient multiplied by each tap output can have a degree of freedom of setting.
[0012]
The broadcast wave receiving tuner according to claim 4 receives a broadcast wave video signal, and includes a video beat component generated by interference of the disturbing broadcast wave video signal with the broadcast wave video signal included in the broadcast wave video signal. A broadcast wave receiving tuner for removing and outputting, wherein the broadcast wave video signal is received, quadrature detection is performed on the received signal and separated into an in-phase component and a quadrature component, and quadrature detection means performs quadrature detection. In-phase component A / D conversion means for analog-to-digital conversion of the detected in-phase component, and high-frequency component extraction delay means for taking out the high-frequency component of the video signal analog-digital converted by the in-phase component A / D conversion means and adjusting the delay And low frequency component extraction delay means for taking out a low frequency component of the video signal analog-digital converted by the in-phase component A / D conversion means and adjusting the delay An orthogonal component A / D conversion means for analog-to-digital conversion of the orthogonal component orthogonally detected by the orthogonal detection means, and a low frequency for extracting a low-frequency component of the video signal analog-to-digital converted by the orthogonal component A / D conversion means 2. An orthogonal component extracting unit and a video beat component generated by interference of a disturbing broadcast wave video signal with a broadcast wave video signal included in the orthogonal component extracted by the low frequency orthogonal component extracting unit. Subtracting the beat component generated by the beat component generation circuit from the low frequency component of the video signal extracted by the beat component generation circuit according to any one of item 3 and the low frequency component extraction delay means, Video output as a digital video output signal by adding to the high frequency component of the video signal extracted by the high frequency component extraction delay means Characterized by comprising Nos output means.
[0013]
According to this configuration, first, the received broadcast wave video signal is quadrature detected by the quadrature detection means, and separated into an in-phase component and a quadrature component. One in-phase component is converted from analog to digital by the in-phase component A / D conversion means, and the high frequency component is extracted and delayed by the high frequency component extraction delay means. Further, the low frequency component is extracted and delayed by the low frequency component extraction delay means. The other orthogonal component is analog-digital converted by the orthogonal component A / D conversion means, and the low frequency component of the orthogonal component is extracted by the low frequency orthogonal component extraction means. Then, the beat component generation circuit generates a video beat component from the low-frequency component of the quadrature component, and the video signal output means subtracts the video beat component from the low-frequency component of the in-phase component and further adds it to the high-frequency component of the in-phase component. Output as a digital video signal.
[0014]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, as an embodiment of the present invention, a case where a carrier frequency of an unnecessary broadcast wave is shifted to the plus side by 10.01 kHz with respect to a carrier frequency of a desired broadcast wave will be described in detail based on the drawings.
(Broadcast wave reception tuner configuration)
FIG. 1 shows a block diagram of a broadcast wave receiving tuner. As shown in FIG. 1, a broadcast wave receiving tuner 1 includes a quadrature detection unit 3, an in-phase component A / D conversion unit 5, an in-phase component delay unit 7, a high-frequency component extraction delay unit 9, and a low-frequency component extraction. A delay unit 11, an orthogonal component A / D conversion unit 13, a low-frequency orthogonal component extraction unit 15, a beat component generation unit 17A, and a video signal output unit 19 are provided.
[0015]
The broadcast wave receiving tuner 1 is a disturbing broadcast wave video signal included in the received broadcast wave video signal, in particular, an unnecessary broadcast wave video signal (interfering broadcast wave video signal) having a frequency approximate to the broadcast wave video signal. Is a receiving device that removes a video beat component generated by interference with a desired broadcast wave video signal and outputs a desired wave video signal that is a modulated signal component of the desired broadcast wave video signal. In particular, the broadcast wave receiving tuner 1 can sufficiently remove the video beat component when the unnecessary broadcast wave video signal and its ghost wave are included in the broadcast wave video signal received as the disturbing broadcast wave video signal.
[0016]
The quadrature detection unit 3 performs quadrature detection on the RF input signal (broadcast wave video signal). In an actual receiving circuit, an input signal from the antenna input terminal is amplified and then converted into an intermediate frequency signal by a mixing circuit, and this intermediate frequency signal is detected, but here the explanation is simplified. Therefore, the expression broadcast wave video signal including the signal to be detected is used. Note that a video signal of a channel that the user of the broadcast wave receiving tuner 1 desires to view is a desired wave video signal (modulated signal), this desired wave video signal is A (t), and the carrier wave of this A (t) is cos. (Ωt) (desired wave carrier wave). Further, an interference wave video signal (modulated signal) that is an unnecessary broadcast wave video signal mixed in the RF input signal is represented by B (t) (B is the amplitude of the interference wave video signal). Let the carrier wave be cos {(ω + δ) t + φ}. Here, δ corresponds to the offset frequency (+10.01 kHz).
[0017]
At this time, the RF input signal can be expressed as RF input signal = A (t) cos (ωt) + B (t) cos {(ω + δ) t + φ}. In order to avoid the complexity of the mathematical formula, D / U, which is the ratio between the desired wave and the disturbing wave, is set to D / U = 0 dB.
[0018]
Here, when there are a plurality of unnecessary broadcast wave video signals mixed in the RF input signal, these unnecessary broadcast wave video signals are designated as B._k(T), this B_kThe carrier wave of (t) is cos {(ω + δ_k) T + φ_k}, The RF input signal is RF = A (t) cos (ωt) + ΣB_k(T) cos {(ω + δ_k) T + φ_k}. As described above, even when there are a plurality of unnecessary broadcast wave video signals, it can be considered that one disturbing broadcast wave video signal obtained by synthesizing these signals is causing interference. However, the phase of the combined interference wave is determined by the video signal amplitude (B_k) And offset frequency (δ_k) And a time function that changes in response to.
[0019]
A case where an unnecessary broadcast wave video signal is input with a single ghost will be specifically assumed as the most practical case. In a general home reception environment, it is not considered that a receiving antenna is installed so as to reduce ghost even for unnecessary broadcast wave video signals, and in some cases, ghost D / U of disturbing broadcast waves is extremely high. This is because bad things are also assumed.
[0020]
In this case, since the offset frequency of the disturbing main wave and the ghost wave are the same, δ₁= Δ₂= Δ. Also, the ratio of disturbing main wave to ghost wave is (D / U)_gohst, Ghost delay time τ_gohstThen, the video signal amplitude B of the disturbing main wave₁(T) and the video signal amplitude B of the ghost wave₂The relationship with (t) is B₂(T) = (D / U)_gohst× B₁(T-τ_gohst) And the phase of the disturbing main wave φ₁And ghost wave phase φ₂The relationship between₂= Φ₁-(Ω + δ) τ_gohstIt becomes.
[0021]
That is, the RF input signal in this case is

It is expressed. However,
R²(T) = B₁ ²(T) + B₂ ²(T) + 2B₁(T) B₂(T) cos (φ₁−φ₂)
R (t) cosβ (t) = B₁(T) cosφ₁+ B₂(T) cosφ₂
R (t) sinβ (t) = B₁(T) sinφ₁+ B₂(T) sinφ₂
And
[0022]
If there is no ghost, β (t) = φ₁In the case where there is a ghost, β (t) is a time function that fluctuates in accordance with an amplitude change of an unnecessary broadcast wave video signal. If β (t) is simply represented by β to avoid the complexity of the notation, the RF input signal when an unnecessary broadcast wave video signal is input with a single ghost is RF input signal = A (t) cos. (Ωt) + R (t) cos {(ω + δ) t + β}. This equation is defined as equation (1).
[0023]
That is, the quadrature detection unit 3 performs quadrature detection on the RF input signal represented by A (t) cos (ωt) + R (t) cos {(ω + δ) t + β}. When a component in phase with the desired wave carrier wave (in-phase component) is called an I signal and a component orthogonal to the desired wave carrier wave (orthogonal component) is called a Q signal, the I signal is I signal = A (t) + R (t) cos (δt + β), and the Q signal can be expressed as Q signal = R (t) sin (δt + β). If the quadrature detection unit 3 performs multiplication by a signal obtained by advancing the phase of the oscillator output by π / 2, the quadrature component will be given a minus sign. It will be expressed as a positive signal.
[0024]
Let I signal = A (t) + R (t) cos (δt + β) be equation (2) and Q signal = R (t) sin (δt + β) be equation (3).
The in-phase component A / D converter 5 performs analog-to-digital conversion on the I signal that is in-phase with the desired wave carrier wave.
[0025]
The in-phase component delay unit 7 (2HDL in the figure) is a relative time between the output signal of the center tap and the I signal of two odd tap transversal filters having the same configuration, which is a component of the beat component generation unit 17A described later. In order to match the relationship, the I signal separated by the quadrature detection unit 3 of the broadcast wave receiving tuner 1 is delayed by 2H (shifted by two horizontal scanning periods). As a result, the transversal filter can handle up to a Q signal that is advanced by 2H relative to the I signal.
[0026]
The high frequency component extraction delay unit 9 extracts a high frequency component of the signal analog-digital converted by the in-phase component A / D conversion unit 5, and uses the high frequency component signal as a low frequency component extraction delay unit 11 and a beat component generation unit 17A. The delay is adjusted so as to synchronize with the signal output from. In this embodiment, as an example, the high-frequency component extraction delay unit 9 includes a high-pass filter (HPF) and a delay adjustment circuit (DL adjustment in the figure).
[0027]
The low frequency component extraction delay unit 11 takes out the low frequency component of the signal analog-digital converted by the in-phase component A / D conversion unit 5 and generates the low frequency component signal as a high frequency component extraction delay unit 9 and a beat component generator. The delay is adjusted so as to be synchronized with the signal output from the unit 17A. In this embodiment, as an example, the low-frequency component extraction delay unit 11 includes a low-pass filter (LPF) and a delay adjustment circuit (DL adjustment in the figure).
[0028]
The I signal is mixed with the video beat component of the desired wave video signal and the disturbing wave video signal (R (t)). If the video beat component (beat disturbance) mixed with this I signal can be removed with respect to the low-frequency component of the video signal, a desired wave video signal that is sufficiently good visually can be obtained. May be limited to low frequency components. Therefore, the I signal is separated into a high frequency component and a low frequency component by the high frequency component extraction delay unit 9 and the low frequency component extraction delay unit 11.
[0029]
The orthogonal component A / D converter 13 performs analog-to-digital conversion on the Q signal, which is a component orthogonal to the desired wave carrier wave.
The low-frequency quadrature component extraction unit 15 (LPF in the figure) extracts a low-frequency component of the quadrature component of the signal analog-digital converted by the quadrature component A / D conversion unit 13. In this embodiment, the low-frequency orthogonal component extraction unit 15 is configured by a single low-pass filter (LPF) as an example.
[0030]
The beat component generation unit 17A generates a video beat component from the low-frequency component of the orthogonal component extracted by the low-frequency orthogonal component extraction unit 15, and details will be described later.
[0031]
The video signal output unit 19 subtracts the video beat component (details will be described later) generated by the beat component generation unit 17A from the low frequency component extracted by the low frequency component extraction delay unit 11 to remove the interference wave component. Thereafter, the desired wave video signal in which the interference wave component is sufficiently reduced by combining with the high frequency component extracted by the high frequency component extraction delay unit 9 is output as a digital signal.
[0032]
(Configuration of beat component generation unit [beat component generation circuit])
Next, the beat component generation unit 17A will be described in detail. The beat component generation unit 17A includes a multiplier 21a, a multiplier 21b, an oscillator 21c, and an addition coefficient multiplier 23 (23a₁..., 23b₁...) and the unit delay element 25 (25a) in one horizontal scanning period.₁..., 25b₁.., A summation counter 27 (27a, 27b), a multiplier 29a, a multiplier 29b, and a subtractor 31. The beat component generation unit 17A detects the instantaneous phase of the disturbing wave video signal and generates a video beat component that follows the instantaneous phase of the disturbing wave video signal, so that an extremely large beat suppression amount can be obtained. It is configured. The beat suppression amount is the power ratio of the residual beat component to the input beat component.
[0033]
The multiplier 21a multiplies the low-frequency component of the Q signal after analog-digital conversion extracted by the low-frequency orthogonal component extraction unit 15 by a sine wave having the same frequency as the offset frequency oscillated by the oscillator 21c. An alternating component detection output signal is obtained.
[0034]
The multiplier 21b advances the phase of a sine wave having the same frequency as the offset frequency oscillated by the oscillator 21c to the low frequency component of the Q signal after analog-digital conversion extracted by the low frequency orthogonal component extraction unit 15 by π / 2. The second quadrature component detection output signal is obtained by multiplying the cosine wave. Here, the calculation processing of the Q signal in the multiplier 21a and the multiplier 21b is equivalent to quadrature detection of the Q signal.
[0035]
The oscillator 21c generates an oscillating signal having the same frequency as the offset frequency. The offset frequency is used even when interference occurs when the same carrier frequency is used for radio waves transmitted from two broadcasting stations. In order to make the disturbance difficult to see visually, it is a frequency obtained by shifting the carrier frequency of one broadcasting station by about 10 kHz. If the sin wave output from the oscillator 21 is expressed as sin (δt + θ), the cosine wave obtained by advancing the phase of the sin wave by π / 2 is expressed as cos (δt + θ).
The multiplier 21a and the oscillator 21c correspond to the first quadrature component detection output means described in the claims. The multiplier 21b, the oscillator 21c, and a π / 2 phase shifter (π / 2 in the figure) Corresponds to the second orthogonal component detection output means described in the claims.
[0036]
An addition coefficient multiplier 23 constituting two systems of transversal filters adds an addition coefficient (C to the first quadrature component detection output signal and the second quadrature component detection output signal.₂) And an addition coefficient (C) for each output signal of the unit delay element 25 in one horizontal scanning period₀, C₁, C₂). This addition coefficient is limited by a constraint condition (to be described later) for detecting the instantaneous phase of the transversal filter output signal obtained when each tap output of the transversal filter is displayed as a vector.
[0037]
A unit delay element 25 (1 HDL in the figure) that constitutes two systems of transversal filters delays the first quadrature component detection output signal and the second quadrature component detection output signal in units of one horizontal scan period. The signal before the horizontal scanning period (1H) (1H delayed signal (corresponding to the upper part of the screen in the displayed screen)), the signal before 1H, the signal before 1H, and the output signal of the center tap of the transversal filter Signal (1H phase advance signal (corresponding to the lower part of the screen in the displayed screen)), 2T ahead signal tap output. In this embodiment, the unit delay element is used for one horizontal scanning period, but the frequency characteristics of the transversal filter can be changed by changing the delay unit. However, when changing the number of taps and the delay unit, it is necessary to change the delay time of the synchronous component delay unit 7 accordingly.
[0038]
As shown in FIG. 1, the addition coefficient multiplier 23a₁Is added to the first quadrature component detection output signal (2H phase advance signal) which is the output from the multiplier 21a.₂And the addition coefficient multiplier 23a₂Is a unit delay element 25a in one horizontal scanning period.₁Addition coefficient C to the tap output signal (1H phase advance signal) that is output from₁And the addition coefficient multiplier 23a₃Is a unit delay element 25a in one horizontal scanning period.₂Is added to the center tap output signal which is output from₀Is multiplied. Further, the addition coefficient multiplier 23a₄Is a unit delay element 25a in one horizontal scanning period.₃Is added to the tap output signal (1H delayed signal) which is the output from₁And the addition coefficient multiplier 23a₅Is a unit delay element 25a in one horizontal scanning period.₄Is added to the tap output signal (2H delay signal) which is the output from₂Is multiplied. These addition coefficient multipliers 23a₁~ 23a₅And a unit delay element 25a for one horizontal scanning period₁~ 25a₄And the adder / aggregator 27a correspond to the first instantaneous phase detecting means described in the claims.
[0039]
Similarly, addition coefficient multiplier 23b₁Is added to the second quadrature component detection output signal (2H phase advance signal) which is the output from the multiplier 21b.₂And the addition coefficient multiplier 23b₂Is a unit delay element 25b in one horizontal scanning period.₁Addition coefficient C to the tap output signal (1H phase advance signal) that is output from₁And the addition coefficient multiplier 23b₃Is a unit delay element 25b in one horizontal scanning period.₂Is added to the center tap output signal which is output from₀Is multiplied. Also, the addition coefficient multiplier 23b₄Is a unit delay element 25b in one horizontal scanning period.₃Is added to the tap output signal (1H delayed signal) which is the output from₁And the addition coefficient multiplier 23b₅Is a unit delay element 25b in one horizontal scanning period.₄Is added to the tap output signal (2H delay signal) which is the output from₂Is multiplied. These addition coefficient multipliers 23b₁~ 23b₅And unit delay element 25b for one horizontal scanning period₁~ 25b₄And the adder / aggregator 27b correspond to the second instantaneous phase detecting means described in the claims.
[0040]
The addition totalizer 27 (27a and 27b [Σ: sigma] in the figure) adds and totals the tap output signals of the first and second transversal filters multiplied by the addition coefficient. The addition totalizer 27a Adds and totals each tap output signal of the first transversal filter having the first quadrature component detection output signal as an input signal, and the summation counter 27b uses the second quadrature component detection output signal as an input signal. Each tap output signal of the second transversal filter is summed up. Addition coefficient C as described later₀, C₁, C₂Is set so as to satisfy a certain relationship, as shown in FIG. 1, the output of the adder 27a is R (t) cos (β−θ), and the output of the adder 27b is R (t). sin (β−θ), and thereby the instantaneous phase of the Q signal can be detected.
[0041]
The multiplier 29a multiplies the output R (t) cos (β−θ) from the adder / aggregator 27a and the cos wave (cos (δt + θ)).
The multiplier 29b multiplies the output R (t) sin (β−θ) from the addition totalizer 27b and the sin wave (sin (δt + θ)).
The subtracter 31 subtracts the output of the multiplier 29b from the output of the multiplier 29a, and the output of the subtractor 31 becomes R (t) cos (δt + β), that is, a video beat component. In the case of FIG. 1, the multipliers 29a and 29b and the subtractor 31 correspond to the beat component output means described in the claims.
[0042]
(Instantaneous phase detection in beat component generation unit [beat component generation circuit])
From this, the instantaneous phase detection of the beat component generation unit 17A shown in FIG. 1 will be described. Here, first, the detection of the instantaneous phase of the beat component generation unit 17B (see FIG. 2) including two 3-tap transversal filters having a simpler circuit configuration than the beat component generation unit 17A as its constituent elements will be described. Then, detection of the instantaneous phase of the beat component generation unit 17A will be described. As shown in FIG. 2, the beat component generation circuit 17B includes a multiplier 21a, a multiplier 21b, an oscillator 21c, an addition coefficient multiplier 23, a unit delay element 25 for one horizontal scanning period, and an adder / aggregator. 27, a multiplier 29a, a multiplier 29b, and a subtractor 31. In addition, when compared with the beat component generation unit 17A illustrated in FIG. 1, components that are not different in each configuration are denoted by the same reference numerals and description thereof is omitted.
[0043]
Incidentally, the difference between the beat component generator 17B shown in FIG. 2 and the beat component generator 17A shown in FIG. 1 is the number of addition coefficient multipliers 23 and the number of unit delay elements 25 in one horizontal scanning period. . That is, three addition coefficient multipliers 23a are provided on the output line of the first orthogonal component detection output signal that is an output from the multiplier 21a.₁, 23a₂, 23a₃And two unit delay elements 25a in one horizontal scanning period₁25a₂And three addition coefficient multipliers 23b on the output line of the second quadrature component detection output signal that is an output from the multiplier 21b.₁, 23b₂, 23b₃And two unit delay elements 25b in one horizontal scanning period₁25b₂And are provided. That is, in the beat component generation unit 17B, the output lines of the first quadrature component detection output signal and the second quadrature component detection output signal are multiplied by two unit delay elements 25 in one horizontal scanning period and three addition coefficients. Each of the sample values (a value obtained by multiplying the addition coefficient) is obtained as each tap output. The beat component generation unit 17A can obtain five sample values.
[0044]
(Instantaneous phase detection in beat component generation unit 17B)
The beat component generation unit 17B detects the instantaneous phase of the disturbing wave video signal when generating the video beat component. Here, X is obtained by multiplying the Q signal by a sine wave having the same frequency as the offset frequency, and Y is obtained by multiplying the cos wave having the same frequency as the offset frequency by Y. The ghost wave is included in the disturbing wave video signal. Assuming that Q is the above-described equation (3), R (t) sin (δt + β),

These can be expressed as (4).
[0045]
A signal obtained by passing a signal that can be expressed by the equation (4) through a 1H delay line (1HDL in the figure) is X_k, Y_k(In FIG. 2, 1HDL25a₁25b₁And the delay time of 1H is τ, and (t_k= T−kτ)

And Y as well_kAlso

Can be written.
[0046]
FIG. 3 illustrates signals represented by equations (5) and (6) as vectors. Each delay line output (X_k, Y_k) Is a vector rotating around (1 / 2Rcos (β-θ), 1 / 2Rsin (β-θ)) with a radius of 1 / 2R and an angular velocity of 2δ. That is, in FIG. 3, v1 is P1 (X₀, Y₀) And v2 is P2 (X₁, Y₁) And v3 is P3 (X_-1, Y_-1). Here, R, β, etc. are the sample times (t_k), The vector of the + 1H signal, the vector of the -1H signal, and 0.28 times (= -2 cos (2δτ) =-2 cos (98.06 degrees)) of the 0H signal. The vector sum remains at the fixed point P4 even if each vector rotates with time. Therefore, the phase (β) and the amplitude (R) can be known from the coordinate position of the point P4.
[0047]
The description given with reference to FIG.

[0048]
Where R (t₁) = R (t_-1) = R (t₀) (= R), β (t₁) = Β (t_-1) = Β (t₀) (= Β), and when the signal U and the signal V are obtained by adding the outputs of the delay lines, the following equation is obtained.

[0049]
In this equation (8), the coefficient C₁, C₂If the value satisfies the following equation, the phase (β−θ) and the amplitude (R) can be obtained.

[0050]
That is, from the equation (9), C₀= -2C₁cos (2δτ) = − 2C₁cos (98.06 degrees) = 0.28C₁Therefore, the sum of the vector of the + 1H signal, the vector of the -1H signal, and the vector 0.28 times the 0H signal remains at the fixed point P4 in FIG. β-θ) can be detected.
[0051]
Solve this equation (9)
C₀= -2 cos (2δτ) / (1-cos (2δτ)) = 0.24586
C₁= 1 / (1-cos (2δτ)) = 0.877707 (10) is obtained, and β and R in this case are obtained from the following equations.
U = Rcos (β−θ), V = Rsin (β−θ),
[Expression 1]

... (11) Formula
[0052]
In this embodiment, it is assumed that the carrier frequency of the unnecessary broadcast wave is shifted to the plus side by 10.01 kHz with respect to the carrier frequency of the desired broadcast wave. When the frequency is shifted to the negative side by 10.01 kHz, the carrier wave of B (t) becomes cos {(ω−δ) t + φ}, and therefore, the expression (1) representing the RF input signal is expressed as RF input signal = A (t) cos (ωt) + R (t) cos {(ω−δ) t + β}. In this case, I signal = A (t) + R (t) cos (−δt + β), Q signal = R (t) sin (−δt + β), and Equation (11) is
U = −Rcos (β + θ), V = Rsin (β + θ),
[Expression 1]

(11 ').
[0053]
In the above formulas (4) to (11 ′), β and R are set to each sample time (t_k)), The normal video signal has a large vertical correlation. Therefore, it can be assumed that this assumption is established with high accuracy. In the beat component generation unit 17B, It can be said that an accurate video beat component can be generated.
[0054]
(Instantaneous phase detection in the beat component generation unit 17A)
Next, instantaneous phase detection in the beat component generation unit 17A will be described.
The phase of the PLL oscillation wave of the offset beat canceller using the PLL described in the prior art using the instantaneous phase β (t) obtained by the equation (11) described in the description of the instantaneous phase detection of the beat component generation unit 17B. Although the problem can be solved by instantaneously controlling the image, the video beat component that follows the instantaneous phase of the jamming video signal is directly generated by the circuit shown in FIG. 1 or 3 without using a PLL. be able to. Next, this will be described.
[0055]
The output signals of the adder / aggregator 27a and the adder / aggregator 27b described in the beat component generation unit 17A correspond to the signal obtained by the expression (11), and the U signal has a cosine wave (cos (δt + θ )) Is multiplied by a multiplier 29a, a sin wave (sin (δt + θ)) having the same frequency as the offset frequency is multiplied by a multiplier 29b, and the V signal is subtracted by a subtractor 31.
U × cos (δt + θ) −V × sin (δt + θ) = R (t) cos {(β−θ) + (δt + θ)} = R (t) cos (δt + β) (12) That is, a video beat component can be generated very easily.
[0056]
Here, when the carrier frequency of the unnecessary broadcast wave is shifted to the minus side with respect to the carrier frequency of the desired broadcast wave, V × sin (δt + θ) −U × cos (δt + θ) = R (t) cos { (Β + θ) − (δt + θ)} = R (t) cos (−δt + β) (12 ′) The video beat component can be generated very easily. That is, the subtracter 31 outputs the output of the multiplier 29b from the output of the multiplier 29a depending on whether the carrier frequency of the unnecessary broadcast wave is shifted to the plus side or the minus side with respect to the carrier frequency of the desired broadcast wave. If it is determined whether to subtract or subtract the output of the multiplier 29a from the output of the multiplier 29b, the beat component generating unit 17A can output the video beat component.
[0057]
According to the equations (12) and (12 ′), the oscillation phase (θ) of the oscillator 21c does not affect the finally generated video beat component, and can be any value. Accordingly, the 10.01 kHz generated by the oscillator 21c does not need to be phase-locked to the input wave as in a conventional PLL (Phase Locked Loop) means of a beat component generation circuit (not shown). It will be good. The fact that a feedback system such as a PLL is not necessary has a great merit from the viewpoint of hardware stability and response speed.
[0058]
The beat component generation unit 17B performs 3-sample addition (-1H, 0H, + 1H), whereas the beat component generation unit 17A performs 5-sample addition (-2H, -1H, 0H, 1H, 2H). Each sample addition process is a filter process in the vertical frequency domain, and this characteristic is determined by the above-described addition coefficient.
[0059]
In the case of the beat component generation unit 17B, 3 samples are added, and the addition coefficient C₀, C₁Is uniquely determined by the constraint condition of the equation (9), and therefore, characteristics other than the frequency response shown in FIG. However, in the case of the beat component generation unit 17A, 5 samples are added, and the number of addition coefficients is one more than the constraint condition. For this reason, each coefficient can have a degree of freedom, and the frequency characteristics can be set freely to some extent as shown in FIG. Note that the constraint condition of the coefficient when using 2M delay lines (unit delay element 25 in one horizontal scanning period) and making the load sum of 2M + 1 tap outputs a transversal filter output is as follows:
[0060]
[Expression 2]

[0061]
The increase in noise due to the addition processing is given by the following equation (14).
[0062]
[Equation 3]

(14) Formula
In addition, each addition coefficient of the transversal filter is theoretically limited by the above constraint conditions, but in reality, it is adjusted while looking at the overall beat cancellation operation. It is of a nature that it only needs to be set.
[0063]
(When transversal filter is configured with an even number of taps)
In the above, the operation of the beat component generation unit 17A has been described by taking the case where the transversal filter is configured with odd taps as an example, but even if the transversal filter is configured with even taps, Next, it will be explained supplementarily that the effect can be obtained.
[0064]
FIG. 5 shows a block diagram of the broadcast wave reception tuner 1A in the case where the beat component generation unit 17C is configured by an even-tap transversal filter. As shown in FIG. 5, the beat component generator 17C includes a multiplier 21a, a multiplier 21b, an oscillator 21c, an addition coefficient multiplier 23, and a unit delay element 25 (see FIG. 5). 1HDL), an adder / aggregator 27, a multiplier 29a, a multiplier 29b, and a subtractor 31. The addition coefficient multiplier 23 adds an addition coefficient (C to the first quadrature component detection output signal and the second quadrature component detection output signal._1.5) And an addition coefficient (C) for each output signal of the unit delay element 25 in one horizontal scanning period_0.5, C_1.5). When the beat component generation unit 17C of the broadcast wave reception tuner 1A shown in FIG. 5 is compared with the beat component generation unit 17A of the broadcast wave reception tuner 1 shown in FIG. 1, there is no difference in each configuration. With respect to, the same reference numerals are given and description thereof is omitted.
[0065]
The difference between the beat component generator 17C shown in FIG. 5 and the beat component generator 17A shown in FIG. 1 is the number of addition coefficient multipliers 23 and the number of unit delay elements 25 in one horizontal scanning period. That is, four addition coefficient multipliers 23a are provided on the output line of the first orthogonal component detection output signal, which is an output from the multiplier 21a.₁, 23a₂, 23a₃, 23a₄And three unit delay elements 25a in one horizontal scanning period₁25a₂25a₃And four addition coefficient multipliers 23b on the output line of the second quadrature component detection output signal that is an output from the multiplier 21b.₁, 23b₂, 23b₃, 23b₄And three unit delay elements 25b in one horizontal scanning period₁25b₂25b₃And are provided. That is, in the beat component generation unit 17C, three unit delay elements 25 in one horizontal scanning period and four addition coefficient multipliers are provided on the output lines of the first quadrature component detection output signal and the second quadrature component detection output signal. Accordingly, four sample values (numerical values multiplied by an addition coefficient) are obtained as each tap output.
[0066]
Specifically, as shown in FIG. 5, the addition coefficient multiplier 23a₁Is an addition coefficient C to the first quadrature component detection output signal (1.5H phase advance signal) which is the output signal from the multiplier 21a._1.5And the addition coefficient multiplier 23a₂Is a unit delay element 25a in one horizontal scanning period.₁Is added to the tap output signal (0.5H phase advance signal) which is output from_0.5And the addition coefficient multiplier 23a₃Is a unit delay element 25a in one horizontal scanning period.₂Is added to the tap output signal (0.5H delayed signal) which is output from_0.5And the addition coefficient multiplier 23a₄Is a unit delay element 25a in one horizontal scanning period.₃Is added to the tap output signal (1.5H delayed signal) which is output from_1.5Is multiplied.
[0067]
Similarly, addition coefficient multiplier 23b₁Is an addition coefficient C to the second quadrature component detection output signal (1.5H phase advance signal) which is the output signal from the multiplier 21b._1.5And the addition coefficient multiplier 23b₂Is a unit delay element 25b in one horizontal scanning period.₁Is added to the tap output signal (0.5H phase advance signal) which is output from_0.5And the addition coefficient multiplier 23b₃Is a unit delay element 25b in one horizontal scanning period.₂Is added to the tap output signal (0.5H delayed signal) which is output from_0.5And the addition coefficient multiplier 23b₄Is a unit delay element 25b in one horizontal scanning period.₃Is added to the tap output signal (1.5H delayed signal) which is output from_1.5Is multiplied.
[0068]
Here, the input signal of the transversal filter of FIG. 5 is X obtained by multiplying the Q signal by a sin wave having the same frequency as the offset frequency, and Y obtained by multiplying the cos wave having the same frequency as the offset frequency. It is expressed by the formula (4). A signal X obtained by passing the signal expressed by the equation (4) through a 1H delay line (1HDL in FIG. 5)_k, Y_kIs also as shown in equations (5) and (6).
[0069]
In FIG. 5, 1HDL25a₁25b₁Each delay line output is X_-0.5, Y_-0.51HDL25a₂25b₂Each delay line output is X_0.5, Y_0.5In other words, the signals U and V obtained by adding the delay line outputs are expressed as follows.

[0070]
In this equation (15), the coefficient C_0.5, C_1.5If the numerical value satisfying the following equation is taken, the phase (β−θ) and the amplitude (R) can be obtained.

In this way, when the transversal filter is configured with even taps, there is a difference in the constraint condition of each addition coefficient for detecting the instantaneous phase as compared with the case where the transversal filter is configured with an odd number. There is no difference in operation.
[0071]
(Broadcast wave reception tuner operation)
Next, the operation of the broadcast wave receiving tuner 1 will be described with reference to the flowchart shown in FIG. 6 (refer to FIG. 1 as appropriate).
First, in the broadcast wave receiving tuner 1, it is determined whether or not the RF input signal A (t) cos (ωt) + R (t) cos {(ω + δ) t + β} is input from the receiving antenna (not shown) to the quadrature detection unit 3. (S1). The system waits until this RF input signal is input (No), and when it is determined that the RF input signal has been input (Yes), the quadrature detection unit 3 performs quadrature detection on the RF input signal, and in-phase components with the desired wave carrier wave. It is separated into a certain I signal and a Q signal which is a quadrature component. In the case of an I signal that is an in-phase component separated by the quadrature detection unit 3 (S2, Yes), the separated I signal is analog-digital converted by the in-phase component A / D conversion unit 5 (S3).
[0072]
Then, the I signal analog-digital converted by the in-phase component A / D conversion unit 5 is converted into the in-phase component delay unit 7 (2HDL in FIG. 1) in order to match the relative time relationship with the output signal of the beat component generation unit 17A. ) Is delayed by 2H (S4). The signal delayed by the in-phase component delay unit 7 is branched and, if it is a high frequency component (S5, Yes), the high frequency component extraction delay unit 9 extracts the high frequency component and delay adjusts (S6). If it is a low-frequency component (S5, No), the low-frequency component extraction delay unit 11 extracts the low-frequency component and adjusts the delay (S7).
[0073]
In the case of the Q signal that is the quadrature component separated by the quadrature detection unit 3 (S2, No), the separated Q signal is analog-digital converted by the quadrature component A / D conversion unit 13 (S8). Then, the low frequency orthogonal component extraction unit 15 (LPF in FIG. 1) extracts the low frequency component of the orthogonal component (S9). Thereafter, the low-frequency component of the orthogonal component is input to the beat component generation unit 17A, and the beat component generation unit 17A generates a video beat component R (t) cos (δt + β) (S10).
[0074]
Then, the video signal output unit 19 subtracts the video beat component generated by the beat component generation unit 17A from the low frequency component that has been delay-adjusted by the low frequency component extraction delay unit 11 (S11). Further, the video signal output unit 19 adds the signal obtained by subtracting the video beat component from the low frequency component and the high frequency component that has been delay-adjusted by the high frequency component extraction delay unit 9, and outputs the digital video output A (t ) Is output (S12).
[0075]
(Operation of beat component generation unit [beat component generation circuit])
Next, the operation of generating a video beat component (offset beat component) in the beat component generation unit 17A will be described with reference to the flowchart shown in FIG. 7 (see FIG. 1 as appropriate).
First, the beat component generation unit 17A determines whether or not there is an input of a Q signal R (t) sin (δt + β) (S21). The system waits until the Q signal is input (No), and if it is determined that the Q signal is input (Yes), the multiplier 21a offsets a sin wave having the same frequency as the offset frequency by the multiplier 21b. Cos waves having the same frequency as the frequency are respectively multiplied by the Q signal to obtain a first quadrature component detection output signal and a second quadrature component detection output signal (S22).
[0076]
The first quadrature component detection output signal and the second quadrature component detection output signal are respectively added to the addition coefficient multiplier 23a.₁, 23b₁And a unit delay element 25a in one horizontal scanning period₁25b₁(S23). Addition coefficient multiplier 23a₁, 23b₁Then, the addition coefficient C₂Is multiplied by unit delay element 25a in one horizontal scanning period₁25b₁Then, the first quadrature component detection output signal and the second quadrature component detection output signal are delayed by one horizontal scanning period.
[0077]
The first quadrature component detection output signal delayed by 1H and the second quadrature component detection output signal delayed by 1H are respectively added to the addition coefficient multiplier 23a.₂, 23b₂And a unit delay element 25a in one horizontal scanning period₂25b₂(S24). Further, the unit delay element 25a in one horizontal scanning period₂Output signal from the addition coefficient multiplier 23a₃Unit delay element 25a in one horizontal scanning period₃Subsequently, the unit delay element 25a in one horizontal scanning period₃Output signal from the addition coefficient multiplier 23a₄Unit delay element 25a in one horizontal scanning period₄Unit delay element 25a for one horizontal scanning period₄The output signal from is the addition coefficient multiplier 23a.₅Is input. Further, the unit delay element 25b in one horizontal scanning period₂Is output from the addition coefficient multiplier 23b.₃Unit delay element 25b in one horizontal scanning period₃Subsequently, the unit delay element 25b in one horizontal scanning period₃Is output from the addition coefficient multiplier 23b.₄Unit delay element 25b in one horizontal scanning period₄Unit delay element 25b for one horizontal scanning period₄The output signal from is the addition coefficient multiplier 23b.₅Is input.
[0078]
Thereafter, in the addition totalizer 27a, the addition coefficient multiplier 23a₁, 23a₂, 23a₃, 23a₄, 23a₅The tap output signals multiplied by the addition coefficient are added up and added by the addition coefficient multiplier 23b.₁, 23b₂, 23b₃, 23b₄, 23b₅The tap output signals multiplied by the addition coefficient are tabulated (S25). Furthermore, the multiplier 29a multiplies the first transversal filter output signal, which is the output signal of the adder / aggregator 27a, and the cosine wave having the same frequency as the offset frequency, and the multiplier 29b multiplies the output signal of the adder / aggregator 27b. The two-transversal filter output signal is multiplied by the sine wave having the same frequency as the offset frequency (S26). The output signals of the multipliers 29a and 29b, which are the results, are subtracted and output to the video signal output unit 19 as video beat components (S27).
[0079]
Here, when the carrier frequency of the unnecessary broadcast wave is shifted to the plus side with respect to the carrier frequency of the desired broadcast wave, the output of the multiplier 29b is subtracted from the output of the multiplier 29a, and the carrier wave of the desired broadcast wave When the carrier wave frequency of the unnecessary broadcast wave is shifted to the minus side with respect to the frequency, the output of the multiplier 29a is subtracted from the output of the multiplier 29b. Whether the carrier frequency of the unnecessary broadcast wave is shifted to the plus side or the minus side with respect to the carrier frequency of the desired broadcast wave is determined by the setting of the broadcasting station that transmits these broadcast waves. Therefore, if the receiving point where the broadcast wave receiving tuners 1 and 1A according to the embodiment of the present invention are to be installed is determined, it is naturally determined, and these relationships can be handled as known ones.
[0080]
(About the addition coefficient (sample addition) in the beat component generation unit)
Here, a supplementary description will be given of the addition coefficient (sample addition) multiplied by the addition coefficient multiplier 23 in the beat component generation units 17A and 17B. The equation (4) (the Q signal multiplied by a sine wave having the same frequency as the offset frequency and a cosine wave having the same frequency as the offset frequency) gives a general form of quadrature detection. That is, the Q signal as an input signal can be regarded as a carrier wave of frequency δ being amplitude-modulated with a signal R (t) and further phase-modulated with a signal β (t). The modulated signal (Q signal) multiplied by the carrier having the same frequency as the carrier is X and Y shown in the equation (4).
[0081]
In normal signal demodulation, the second term of each equation (4), which is a frequency component twice the carrier wave, is removed by a filter (not shown) to extract the demodulated component (first term). This operation will be described with reference to the vector diagram shown in FIG. That is, a signal obtained by multiplying a carrier (a sin wave having the same frequency as the offset frequency and a cosine wave having the same frequency as the offset frequency) is a rotation vector represented by the 0H signal in FIG. ) To remove the high-frequency component of the second term is equivalent to integrating the rotation vector over one rotation period (one cycle of 2δ). That is, the rotation component of the vector is averaged by integration and becomes zero, and the first term that is the center of rotation can be extracted as a demodulated output.
[0082]
On the other hand, in the sample addition, instead of integrating the rotation vector over one period (continuously), the vector is periodically sampled to obtain a weighted sum of each sample value. A unit delay element 25 (see FIG. 1) for one horizontal scanning period constitutes a transversal filter, and a horizontal scanning frequency (f_h= 15.73 kHz), the frequency characteristic of the weighted sum is f_hThe comb filter characteristics are repeated.
[0083]
As an example, FIG. 8 shows characteristics when a ± 2H delay line is used. In this comb filter, if the constraint condition (13) for sample addition is satisfied, the response at the frequency twice the offset frequency (high frequency carrier = 20.02 kHz) is zero (a black dot at the center in FIG. 8). Frequency around 20 kHz). Also, at this time, based on the characteristics of the comb filter, f_hThe response to frequencies separated by an integral multiple of zero is also zero (in FIG. 8, the frequency 2δ + f_hAnd 2δ−f_h)
[0084]
By the way, a video signal having a vertical correlation of 100% has a frequency component of f._hHowever, when the interfering video signal is a signal having such a vertical correlation of 100%, the video signal component (the first term of equation (4)) is f_hIt has a component only at a frequency that is an integral multiple of the frequency, and the high-frequency component (the second term in equation (4)) is_hThe signal has a component only at a frequency separated by an integer multiple of. When such a signal is passed through the filter having the characteristics shown in FIG. 8, the video signal component passes 100% (no distortion), the high frequency component is completely blocked, and only the desired video signal component can be extracted. it can.
[0085]
Conversely, a video signal whose vertical correlation is not 100% is f_hIn such a video signal, the video signal component is distorted by the influence of the frequency characteristics of the filter (linear distortion), and the high frequency component is in the filter passband. Have ingredients. For this reason, a mixture of the video signal component and the high frequency component becomes the filter output, and the separation between the video signal component and the high frequency component becomes insufficient.
[0086]
That is, the fact that the video signal component is distorted or that the video signal component and the high frequency component are not sufficiently separated means that the detection of the amplitude and phase (R and β) of the disturbing wave video signal is incomplete. Accordingly, an error is included in the video beat component generated from these, and the broadcast wave receiving tuner 1 cannot perform an accurate beat cancel operation.
[0087]
Therefore, as one method for alleviating the problem of video signal component distortion and high frequency component separation, there is a change in offset frequency. An example in which the offset frequency is changed is shown in FIG. FIG. 9B shows a case where the current offset frequency is 10.1 kHz, and FIG. 9A shows the offset frequency as f._hIn other words, the high frequency component is f / 4._hThis is a case where it is set to be in the middle of an integer multiple of (the signal energy is concentrated at this frequency). As is apparent from the comparison with FIG. 9B, the signal pass band is wide (the signal distortion is small), and the high-frequency carrier is f._hSince it is farthest from an integer multiple of the frequency, the separation of the video signal component and the high frequency component is relatively excellent. Needless to say, to change the offset frequency, it is necessary to change the setting of the existing equipment.
[0088]
(About beat component generator using frame correlation)
So far, the beat component generation units 17A and 17B that utilize the property that the vertical correlation of the video signal is high have been described. A beat component generation unit (not shown) using the frame correlation of the video signal will be described below. A normal video signal has a high frame correlation. For example, a still image has a frame correlation of 100%. In the case of this still image, the video signal component exists only at an integral multiple of the frame frequency (29.97 Hz: hereinafter abbreviated as 30 Hz).
[0089]
The case of using this frame correlation will be described with reference to FIG. When frame correlation is used, a unit delay element of one field or one frame period is used instead of the unit delay element of one horizontal scanning period used when vertical correlation is used. Specifically, “1HDL” of the beat component generation unit 17A shown in FIG. 1 is replaced with “1 field delay line (1VDL) or 1 frame delay line (1FDL)” (not shown). In order for the broadcast wave receiving tuner 1 of FIG. 1 to perform the beat cancel operation, it is necessary to change “2HDL” of the synchronous component delay unit 7 to “1FDL” or “2FDL” in accordance with the change of the unit delay element. It is. FIG. 10 is an operation explanatory diagram when “1HDL” of the beat component generation unit 17A is replaced with “1 field delay line (1VDL)”.
[0090]
Since the current offset frequency of 10.1 kHz is 167 times the field frequency (or 334 times the frame frequency), the rotation vector becomes the same as the current signal vector even if the field is delayed. As a result, no matter how these signals are added, the rotation vector component cannot be canceled. That is, the frame correlation cannot be used with the current offset frequency. Therefore, the offset frequency is 7.5 Hz (vertical frequency (f_v)). In the case of this offset frequency, as shown in FIG. 10A, the rotation vectors vr2 and vr3 of the one-frame delayed signal and the one-frame phase advance signal are in reverse phase relation with the rotation vector vr1 of the current signal, and they are added by a certain amount. It is possible to cancel the rotation vector by weighted addition with a coefficient.
[0091]
That is, when the offset frequency is shifted by 7.5 Hz, a 5-tap transversal filter having a unit delay element of one field period as a component or a 3-tap having a unit delay element of one frame period as a component, If a 5-tap transversal filter is used, a video beat component can be generated as in the case of using a transversal filter having a unit delay element in one horizontal scanning period as a component. Similarly, when the offset frequency is shifted by 15 Hz, the rotation vector of the one-field delay signal and the one-field phase advance signal has a reverse phase relationship with the rotation vector of the current signal. By using a 3-tap or 5-tap transversal filter as a component, a video beat component can be generated. However, in either case, it is necessary to change the settings of existing equipment.
[0092]
This embodiment has the following effects.
In the broadcast wave receiving tuner 1, the received broadcast wave video signal is quadrature detected by the quadrature detection unit 3 and separated into an in-phase component and a quadrature component. One in-phase component is analog-digital converted by the in-phase component A / D conversion unit 5, delayed by the in-phase component delay unit 7, and a high-frequency component is extracted and delayed by the high-frequency component extraction delay unit 9. Further, the low frequency component extraction delay unit 11 extracts and delays the low frequency component. The other orthogonal component is analog-digital converted by the orthogonal component A / D conversion unit 13, and the low frequency orthogonal component extraction unit 15 extracts the low frequency component of the orthogonal component. Then, the beat component generation unit 17A generates a video beat component that follows the instantaneous phase change of the interfering wave video signal from the low frequency component of the quadrature component, and the video signal output unit 19 generates the video beat component from the low frequency component of the in-phase component. The component is subtracted and further added to the high-frequency component of the in-phase component and output as a digital video signal. For this reason, even when the instantaneous phase of the interfering wave video signal changes, a desired wave video signal from which the video beat component has been removed is obtained, and the user of the broadcast wave receiving tuner 1 can improve the desired broadcast program broadcast from the broadcast station. You can watch in a state.
[0093]
In the beat component generation units 17A and 17B (beat component generation circuit), the first quadrature component detection output signal and the second quadrature component detection output signal obtained by the multiplier 21a, the multiplier 21b, and the oscillator 21c are used. While being multiplied by the addition coefficient by the addition coefficient multiplier 23, the delay is sequentially delayed by the unit delay element 25 in one horizontal scanning period, and the other addition coefficients are multiplied and added up. Based on these detection results, the adder / counters 27a and 27b, the multipliers 29a and 29b, and the subtractor 31 can output a video beat component corresponding to the instantaneous phase change of the disturbing wave video signal.
[0094]
Furthermore, when it is assumed that the offset frequency is slightly shifted from the current frequency, the unit delay element constituting the transversal filter is replaced with a delay element of one field period or one frame period from a delay element of one horizontal scanning period, Video beat components can be generated using frame correlation. For example, the offset frequency is 7.5 Hz (vertical frequency (f_v)), The instantaneous phase detection using the 1-frame delay signal and the 1-frame advance signal can be performed, and a complete beat cancel operation can be performed particularly in a still image.
[0095]
When three addition coefficient multipliers 23 are provided (in the case of the beat component generation unit 17B), the value (sample value) added as the multiplication result is three, and the circuit can be configured more simply. it can. When five addition coefficient multipliers 23 are provided (in the case of the beat component generation unit 17A), the value (sample value) added as a multiplication result is five, and the addition coefficient can have a degree of freedom. it can. As the sample value increases, the degree of freedom of the addition coefficient also increases.
[0096]
Furthermore, when there are a plurality of unnecessary broadcast waves, the beat component generator 17A according to an embodiment of the present invention can be used by shifting the offset frequency of each broadcast station so that the offset frequencies are not the same. , 17B, 17C (beat component generation circuit) corresponding to unnecessary broadcast waves, it becomes possible to generate video beat components separately, and beat cancel operation by linearly calculating them Can be executed.
[0097]
As mentioned above, although this invention was demonstrated based on one Embodiment, this invention is not limited to this.
The processing of each component of the beat component generation units 17A and 17B and the broadcast wave reception tuner 1 can also be regarded as a beat component generation method and a broadcast wave reception method that are regarded as each process (step). In this case, the same effects as those of the beat component generation units 17A and 17B and the broadcast wave reception tuner 1 can be obtained. In addition, as described above, even if the transversal filter is configured with an even number of taps, the frequency characteristics equivalent to those of FIGS. 8, 9, and 10 (b) can be obtained. The present invention is not limited to the odd-tap transversal filter described in detail in the embodiment.
[0098]
【The invention's effect】
According to the invention of claim 1, the first quadrature component detection output signal and the second quadrature component detection output signal obtained by the first quadrature component detection output means and the second quadrature component detection output means, The first instantaneous phase detection means and the second instantaneous phase detection means are each subjected to signal processing by a transversal filter to perform instantaneous phase detection using the vertical correlation of the interfering wave image signal. Based on this, the beat component output means outputs the video beat component. Therefore, if this video beat component is subtracted from the video signal of the in-phase component, the video beat component is removed even when the instantaneous phase of the interfering wave video signal is changed, that is, a good desired wave video signal without interfering video is obtained. Can be obtained.
[0099]
According to the second aspect of the invention, the offset frequency in the offset carrier system is changed from the current 10.01 kHz to, for example, 7.5 Hz (vertical frequency (f_v)), The instantaneous phase detection using the frame correlation of the disturbing wave video signal can be performed, and a complete beat cancel operation can be performed particularly in a still image.
[0100]
According to the third aspect of the present invention, the circuit scale can be reduced in the case of a 3-tap transversal filter, and the constraint condition for detecting the instantaneous phase in the case of a 5-tap transversal filter. It is possible to give a degree of freedom to setting each addition coefficient to be multiplied to each tap output limited by the above.
[0101]
According to the invention described in claim 4, the in-phase component separated by the quadrature detection means is extracted by the high frequency component extraction delay means and the low frequency component is extracted by the low frequency component extraction delay means and delayed. . The other orthogonal component is extracted by the orthogonal component low-frequency extraction means. Then, the beat component generation circuit generates a video beat component from the low-frequency component of the quadrature component, and the video signal output means subtracts the video beat component from the low-frequency component of the in-phase component and further adds it to the high-frequency component of the in-phase component. Is output. For this reason, interference interference inside and outside the broadcasting area of the broadcasting station can be prevented, especially when the interference wave image signal is accompanied by a ghost wave component, the interference wave image signal which is a composite wave of the direct interference wave and the ghost interference wave. The offset beat (video beat component) that changes with the amplitude fluctuation of the video signal whose instantaneous phase causes interference interference can be removed, and the user of the broadcast wave receiving tuner can improve the desired broadcast program broadcast from the broadcast station You can watch in a state.
[Brief description of the drawings]
FIG. 1 is a block diagram of a broadcast wave receiving tuner according to an embodiment of the present invention.
FIG. 2 is a block diagram showing another configuration of the beat component generation unit.
FIG. 3 is a vector diagram for explaining sample addition processing by a beat component generation unit shown in FIG. 2;
4 is a diagram illustrating frequency characteristics of a beat component generation unit illustrated in FIGS. 1 and 2. FIG.
FIG. 5 is a block diagram of a broadcast wave reception tuner when the beat component generation unit shown in FIG. 1 is configured by an even-tap transversal filter.
6 is a flowchart for explaining the operation of the broadcast wave receiving tuner shown in FIG. 1;
7 is a flowchart illustrating the operation of the beat component generation unit shown in FIG.
FIG. 8 is a diagram illustrating frequency characteristics of sample addition processing by the beat component generation unit illustrated in FIG.
FIG. 9 is a diagram for explaining the frequency characteristics of the sample addition processing by the beat component generation unit shown in FIG. 2 and the frequency characteristics when the offset frequency is shifted.
FIG. 10A is a vector diagram when the offset frequency is shifted by 7.5 Hz from the current frequency and the 5-tap transversal filter of the beat component generation unit is configured by unit delay elements of one field period. .
(B) The frequency characteristics when the offset frequency is shifted by 7.5 Hz from the current frequency and the 5-tap transversal filter of the beat component generation unit is configured by unit delay elements of one field period. .
[Explanation of symbols]
1, 1A Broadcast wave reception tuner
3 Quadrature detection unit (orthogonal detection means)
5 In-phase component A / D converter (In-phase component A / D conversion means)
7 In-phase component delay
9 High-frequency component extraction delay unit (High-frequency component extraction delay means)
11 Low frequency component extraction delay unit (low frequency component extraction delay means)
13 Orthogonal component A / D converter (orthogonal component A / D converter)
15 Low frequency orthogonal component extraction unit (low frequency orthogonal component extraction means)
17A, 17B, 17C Beat component generation unit (beat component generation circuit)
19 Video signal output section
21a, 21b multiplier (first and second orthogonal component detection output means)
21c Oscillator (first and second orthogonal component detection output means)
23. Addition coefficient multiplier (first and second instantaneous phase detection means)
25 Unit delay element for one horizontal scanning period
27a, 27b Adder / Aggregator (first and second instantaneous phase detecting means)
29a, 29b Multiplier (beat component output means)
31 Subtractor (beat component output means)

Claims (4)

A beat component generation circuit that generates a video beat component that is included in a received broadcast wave video signal and is generated by interference of a disturbing broadcast wave video signal with the broadcast wave video signal,
First quadrature component detection to obtain a first quadrature component detection output signal by multiplying the quadrature component obtained by quadrature detection of the broadcast wave video signal by a sine wave having the same frequency as the offset frequency generated by the oscillator Output means;
A second quadrature component detection output means for multiplying the quadrature component by a cos wave having the same frequency as an offset frequency obtained by shifting the phase of the sin wave by π / 2, and obtaining a second quadrature component detection output signal;
A first transversal filter to which a first quadrature component detection output signal obtained by the first quadrature component detection output means is input, and includes a plurality of unit delay elements for one horizontal scanning period, and a load of each tap output. First instantaneous phase detection means comprising a transversal filter configured to obtain a sum;
A second transversal filter to which the second quadrature component detection output signal obtained by the second quadrature component detection output means is input, and includes a plurality of unit delay elements for one horizontal scanning period, and the load sum of each tap output is calculated. Second instantaneous phase detection means comprising a transversal filter configured to obtain;
Subtracting the calculation output obtained by multiplying the second detection result obtained by the second instantaneous phase detection means by the sin wave from the calculation output obtained by multiplying the first detection result obtained by the first instantaneous phase detection means by the cos wave, or Beat component output means for subtracting a calculation output obtained by multiplying the first detection result by the cosine wave from a calculation output obtained by multiplying the second detection result by the sin wave;
A beat component generation circuit comprising: The unit delay elements of the first and second transversal filters constituting the first instantaneous phase detection means and the second instantaneous phase detection means are respectively changed from a delay element of one horizontal scanning period to one field period or one frame period. The beat component generation circuit according to claim 1, wherein the beat component generation circuit is replaced with a delay element. The number of taps of the first and second transversal filters constituting the first instantaneous phase detection unit and the second instantaneous phase detection unit, respectively, is 3 taps or 5 taps. Item 3. The beat component generation circuit according to Item 2. This is a broadcast wave receiving tuner that receives a broadcast wave video signal and removes and outputs a video beat component that is included in the broadcast wave video signal and is generated by interference of the disturbing broadcast wave video signal with the broadcast wave video signal. And
Quadrature detection means for receiving the broadcast wave video signal, performing quadrature detection on the received signal, and separating the in-phase component and the quadrature component;
In-phase component A / D conversion means for analog-to-digital conversion of the in-phase component quadrature detected by the quadrature detection means;
High-frequency component extraction delay means for taking out a high-frequency component of the video signal analog-digital converted by the in-phase component A / D conversion means and adjusting the delay;
Low frequency component extraction delay means for taking out a low frequency component of the video signal analog-digital converted by the in-phase component A / D conversion means and adjusting the delay;
Orthogonal component A / D conversion means for analog-digital conversion of the orthogonal component orthogonally detected by the orthogonal detection means;
Low-frequency orthogonal component extraction means for extracting a low-frequency component of the video signal analog-digital converted by the orthogonal component A / D conversion means;
4. A video beat component generated by interference of a disturbing broadcast wave video signal with a broadcast wave video signal included in the orthogonal component extracted by the low-frequency orthogonal component extraction unit is generated. Beat component generation circuit described in the section,
The beat component generated by the beat component generation circuit is subtracted from the low frequency component of the video signal extracted by the low frequency component extraction delay means, and added to the high frequency component of the video signal extracted by the high frequency component extraction delay means. Video signal output means for outputting as a digital video output signal,
A broadcast wave receiving tuner characterized by comprising:

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