JP3728633B2 - Vector control method and apparatus for induction motor - Google Patents

Description

ここに、Ｒ２は２次抵抗を、Ｌ２は２次インダクタンスを、Ｍは相互インダクンスを、ｓは微分演算子ｄ／ｄｔを各々示している。また、滑り角周波数生成部８ｂでは、基本的には次式に示すように、２次磁束のｄ軸成分推定値と１次電流ｑ軸成分ｉ１ｑとを用いて滑り角周波数ωｓを生成している。
【数２】

滑り周波数形１次角周波数生成器８は、次式に示すように、滑り角周波数を速度検出器から得た電動機の電気角速度ω２ｎに加算して、１次角周波数ω１を決定している。
【数３】

【０００７】
【発明が解決しようとする課題】
上記従来の滑り周波数形ベクトル制御方法及び装置においては、同図及び式（３）が明確に示すように、１次角周波数の生成に、電動機の速度情報が不可欠である。このため、エンコーダに代表される速度検出器を中心とした速度検出系を構成しなければならなかった。しかし、速度検出器を含む速度検出系は誘導電動機本体に比較すると極めて脆弱であり、誘導電動機含む全駆動システムの信頼性を著しく低下させていた。例えば、速度検出器は電動機が受ける衝撃に対し著しく弱く、また、速度検出器の信号は一般に数ボルトであり送信時にはノイズに対する特別の配慮が必要であり、更には、速度検出器の電源線や信号線の安全な配線にも配慮が求められた。また、脆弱な速度検出系が異常を起こした場合の対策も要求された。このような速度検出系の実装はコスト面でも不利であった。加えて、検出器の実装に際しては、電動機の軸方向の寸法が大きくなるなど、電動機の設置においても不利であった。速度検出の方法及び手段としては、エンコーダ等の速度検出器を用いないで、速度推定の形で速度を検出するような速度検出系を構成するものも一部報告されているが、検出速度の検出精度、検出範囲、検出迅速性、検出演算負荷等の問題を新たに誘発し、制御性能は速度検出器付きの場合に比較しかなり劣化するものとなった。しかも、このような速度検出系を構成したベクトル制御方法及び装置は、速度検出器付きの場合に比較し数段複雑化しており、この点においても不利なものとなった。
【０００８】
上に説明したように、これまで実用化が進られてきた滑り周波数形ベクトル制御方法に立脚する限り、その実現に際しては電動機の速度情報の確保は不可欠であり、速度情報の確保に起因する諸問題が従来より不可避的に残置されてきた。
【０００９】
本発明は、以上の背景のもとになされたものであり、その目的は、速度情報の確保に起因する諸問題を解決しうる、速度情報を一切必要としないベクトル制御方法および同装置を提供することにある。
【００１０】
【課題を解決するための手段】
上記目的を達成するために、請求項１の発明は、２次元ベクトル量としての２次磁束と同一方向をｄ軸に選定しこれと直交する軸をｑ軸に選定する直交ｄｑ座標系上で、１次電流をｄ軸成分とｑ軸成分に分割し制御する工程と、直交ｄｑ座標系の位相を決定する工程とを有する誘導電動機のベクトル制御方法において、１次角周波数と１次磁束のｄ軸成分の磁束推定近似値との積と、１次電圧のｑ軸成分あるいはその電圧推定近似値から１次電流のｑ軸成分あるいはその電流推定近似値を直列接続された１次抵抗と総合１次漏れインダクタンスに流した際に発生する降下電圧を減じて得られた残電圧値とを、直接的に等しくすべく、該残電圧値と該１次磁束ｄ軸成分推定近似値との演算処理により該１次角周波数を決定し、決定した該１次角周波数を、該位相決定工程で用いる１次角周波数の全部または一部として使用することを特徴とするものである。
【００１１】
請求項２の発明は、請求項１の誘導電動機のベクトル制御方法において、該１次磁束ｄ軸成分推定近似値の定常値が、該１次電流ｄ軸成分あるいはその電流推定近似値の定常値に１次インダクタンスを乗じた値に等しくなるように、該１次磁束ｄ軸成分推定近似値を定めることを特徴とするものである。
【００１２】
請求項３の発明は、請求項１及び２の誘導電動機のベクトル制御方法において、該１次磁束ｄ軸成分推定近似値に対し正の下限値を設定し、該１次磁束ｄ軸成分推定近似値が該下限値より常時大きな値を取るようにしたことを特徴とするするものである。
【００１３】
請求項４の発明は、請求項１の誘導電動機のベクトル制御方法において、該残電圧値の生成工程において、低域通過フィルタによるフィルタリング処理を行うことを特徴とするものである。
【００１４】
また、請求項５の発明は、２次元ベクトル量としての２次磁束と同一方向をｄ軸に選定しこれと直交する軸をｑ軸に選定する直交ｄｑ座標系上で、１次電流をｄ軸成分とｑ軸成分に分割し制御する手段と、直交ｄｑ座標系の位相を決定する手段とを有する誘導電動機のベクトル制御装置において、該位相決定手段の構成に、１次磁束ｄ軸成分の磁束推定近似値を生成する近似磁束生成部と、１次電圧のｑ軸成分あるいはその電圧推定近似値から１次電流のｑ軸成分あるいはその電流推定近似値を直列接続された１次抵抗と総合１次漏れインダクタンスに流した際に発生する降下電圧を減じた残電圧値を生成する残電圧生成部と、近似磁束生成部の出力信号と１次角周波数との積が該残電圧生成部の出力信号と直接的に等しくなるように該１次角周波数を決定する１次角周波数演算部とを有する１次角周波数生成手段を少なくとも具備することを特徴とするものである。
【００１５】
請求項１又は５の本発明によれば、２次磁束と同一方向をｄ軸に選定しこれと直交する軸をｑ軸に選定する直交ｄｑ座標系に必要な位相確立のための１次角周波数が、電動機の速度情報を一切用いることなく、生成されると言う作用が得れることを、先ずノイズ等を含まない正確な信号を用いて説明する。
【００１６】
１次角周波数をω１、１次磁束のｄ軸成分をφ１ｄ、１次電圧のｑ軸成分をｖ１ｑ、１次電流のｄ軸成分をｉ１ｄ、またｑ軸成分をｉ１ｑと表現するならば、本発明は、電動機の速度信号を一切含まない次の関係式（４）が成立するように、式（４）の右辺で示される信号を信号φ１ｄによる除算等の演算により、直接的に１次角周波数ω１を生成しようとするものである。
【数４】

ここに、Ｌｐ１は、１次インダクタンスＬ１、２次インダクタンスＬ２、相互インダクタンスＭを用いて、Ｌｐ１＝（Ｌ１ーＭ・Ｍ／Ｌ２）と定義される総合１次漏れインダクタンスである。上式右辺第２項の信号は、１次電流ｑ軸成分を、直列接続された１次抵抗と総合１次漏れインダクタンスに流した際に発生する降下電圧を示しており、また右辺全体の信号は、１次電圧ｑ軸成分からこの降下電圧を減じて得られた残電圧値を示している。また、このときの１次磁束ｄ軸成分φ１ｄは、直接検出することなく、１次電流ｄ軸成分ｉ１ｄを用いたつぎの式（５）、（６）に従い推定近似しようとするものである。
【数５】

【数６】

上式におけるＷ２、Ｒ２ｎは、Ｗ２＝Ｒ２／Ｌ２、Ｒ２ｎ＝（Ｍ・Ｍ・Ｒ２）／（Ｌ２・Ｌ２）と定義される電動機パラメータである。
【００１７】
ところで、誘導電動機においては、内部状態において、一般に次式（７）、（８）に示す関係が成立している。
【数７】

【数８】

ここに、φ２ｑは２次磁束のｑ軸成分である。
【００１８】
さてここで、本発明の基本を示す式（４）を式（７）に用いると、ｓφ２ｑ＝０なる結果が得られる。また、本発明で１次磁束の近似推定に利用される式（６）を式（８）に用いると、φ２ｑ＝０なる結果が得られる。ｓφ２ｑ＝０、φ２ｑ＝０なる結果は、２次磁束と同一方向をｄ軸に選定しこれと直交する軸をｑ軸に選定する直交ｄｑ座標系ための位相が確立されることを意味している。以上の説明によれば、１次磁束ｄ軸成分の推定近似値を得るには、式（５）、（６）の利用が不可欠のように考えがちであるが、この利用は緩和が可能である。これについては、請求項２の発明の作用説明の際に改めて説明する。以上の説明より明らかなように、請求項１又は５の本発明によれば、本質的に、２次磁束と同一方向をｄ軸に選定しこれと直交する軸をｑ軸に選定する直交ｄｑ座標系の位相確立に必要な１次角周波数が、電動機の速度情報を一切用いることなく、生成されると言う作用が得れる。
【００１９】
請求項１又は５の本発明では、１次電圧のｑ軸成分、１次電流のｑ軸成分がノイズ等の影響により正確な値が入手できない場合には、これらの近似推定値により１次角周波数を生成するようにしている。本発明は、直交ｄｑ座標系の位相確立に必要な本質的条件を直接満足させることを１次角周波数の生成原理としているのでいるので、ノイズ等の影響が強い場合にも、上記信号の近似推定値を利用することにより、現実的な１次角周波数を生成できると言う作用が得られる。
【００２０】
請求項２の本発明によれば、請求項１の発明におけるベクトル制御方法において、１次磁束ｄ軸成分推定近似値の定常値φ１ｄ（＝一定）を、１次電流ｄ軸成分あるいはその電流推定近似値の定常値ｉ１ｄ（＝一定）に１次インダクタンスＬ１を乗じた値に等しくなるように定めるもの、すなわち、定常値φ１ｄをつぎの関係が維持されるように定めるものである。
【数９】

上式は、定常値に関しては、式（５）、（６）と完全に等価である。しかし、式（５）、（６）に比較し、はるかに簡単である。ベクトル制御においては、内部磁束に直接関係する１次電流ｄ軸成分は例外的な場合を除き一定の定常値に保たれるので、請求項２の本発明によれば、１次磁束ｄ軸成分の推定近似値を極めて簡単に生成できると言う作用が得られる。
【００２１】
請求項３の本発明によれば、請求項１および２の発明におけるベクトル制御方法において、１次磁束ｄ軸成分推定近似値に対し正の下限値を設定し、１次磁束ｄ軸成分推定近似値が該下限値より常時大きな値を取るようにするので、１次角周波数の生成を、残電圧値の１次磁束ｄ軸成分推定近似値による除算という簡単な演算で実施する場合にも、ゼロあるいは極小値で除算を行うというゼロ割り現象を確実に回避することができるという作用が得られる。
【００２２】
請求項４の本発明によれば、請求項１の発明におけるベクトル制御方法において、残電圧値の生成工程に、低域通過フィルタによるフィルタリング処理を行うので、残電圧値を与える式（４）の右辺が示すように総合１次漏れインダクタンスＬｐ１が関与する電圧生成には純粋微分が要求されるが、この純粋微分を回避することができると言う作用が得られる。ひいては、残電圧値が現実的な信号に対して不必要な過大値を示したり、ノイズ等の外乱に対し不都合な過激反応することを回避することができると言う作用が得られる。
【００２３】
【発明の実施の形態】
以下、図面を用いて、本発明の実施形態を詳細に説明する。本発明を適用した誘導モータのベクトル制御装置の全系の１例を図１に示す。図１は、本発明に直接関係する新１次角周波数生成器１１、及びこれに関連して誘導電動機に速度検出器が装置されていない２点を除けば、他の機器は従来の滑り周波数形ベクトル制御装置の場合の図１３と同様である。すなわち、直交ｄｑ座標系の位相を決定する手段において、新１次角周波数生成器で生成された１次角周波数を積分器９で積分処理し、さらには、積分処理した信号を正弦信号発生器１０で積分値に応じた正弦信号（ｃｏｓ、ｓｉｎ信号）を生成し、これを直交ｄｑ座標系の最終的位相情報として、ベクトル回転器６ａ、６ｂへ出力する点は、従来の滑り周波数形ベクトル制御方法及び装置と同様である。このため、本実施形態では、１次角周波数成手段を具現化した新１次角周波数生成器１１についてその１例を詳しく説明する。
【００２４】
図１の例では、新１次角周波数生成器１１には、１次電流のｄ軸成分ｉ１ｄ、ｑ軸成分ｉ１ｑ、及び１次電圧ｑ軸成分の近似推定値としてその指令値ｖ１ｑ＊が、入力されている。
【００２５】
図２は、新１次角周波数生成器１１の内部の構成を示したものである。同図の１１ａは、１次電圧ｑ軸成分指令値から１次電流のｑ軸成分を直列接続された１次抵抗と総合１次漏れインダクタンスに流した際に発生する降下電圧を減じた残電圧値を生成する残電圧生成部である。１１ｂは、１次電流ｄ軸成分ｉ１ｄから１次磁束ｄ軸成分の磁束推定近似値を生成する近似磁束生成部であり、１１ｃは、近似磁束生成部の出力信号と１次角周波数との積が上記残電圧生成部の出力信号と直接的に等しくなるように１次角周波数を決定する１次角周波数演算部である。
【００２６】
図３は、残電圧値生成部１１ａの構成の１例を示したものである。図３の例では、低域通過フィルタｆ／（ｓ＋ｆ）によるフィルタリング処理を実施している点に特徴がある。式（４）が示すように総合１次漏れインダクタンスＬｐ１が関与する電圧生成には理想的状況では純粋微分が要求されるが、図３の例は、低域フィルタによりこの純粋微分を回避されていることを、明確に示している。
【００２７】
図４は、近似磁束生成部の構成の１例の詳細を示したものである。この例では、理解を容易にするため、機能別に構成を分けた３段構成で、近似磁束磁束生成部を構成している。近似磁束生成部第１段は基本部１１ｂａであり、式（５）、（６）に示した関係を忠実に具現化したものである。この基本部１１ｂａの入力である１次電流ｄ軸成分の定常値に対し、基本部の出力信号は１次インダクタンスＬ１を乗じた値に等しくなっている点を特に指摘しておく。近似磁束生成部第２段１１ｂｂはバランス部であり、前述の残電圧生成部の出力信号に対しゲインと位相のバランスを保つことを目的に用意されている。したがって、この１例では、簡単には、残電圧生成部に導入した低域通過フィルタｆ／（ｓ＋ｆ）と同一のものを設置すればよい。近似磁束生成部第３段はリミッタ部１１ｂｃであり、これにより、近似磁束生成部の最終的な出力信号である１次磁束ｄ軸成分推定近似値が予め設定した正の下限値より、常時大きな値を取ることが出来る。
【００２８】
図５は、１次角周波数演算部１１ｃの構成の１例を示したものである。１次角周波数演算部の働きは、近似磁束生成部の出力信号と１次角周波数との積が該残電圧生成部の出力信号と直接的に等しくなるように１次角周波数を決定することにあるが、本例では、残電圧生成部の出力信号を近似磁束生成部の出力信号により除算すると言う最も簡単な決定法を利用している。除算による場合にも、前述の近似磁束生成部において１次磁束ｄ軸成分推定近似値が予め設定した下限値より常時大きな値を取るようにしているので、ゼロあるいは極小値で除算を行うというゼロ割り現象を確実に回避することができる。
【００２９】
図６は、本発明を用いた上記実施形態のベクトル制御装置により、誘導電動機を駆動制御した場合の実験結果の１例を示したものである。左上が１次電流ｄ軸成分の指令値を、右上が生成された１次角周波数を、左下が発生トルクを、右下が電動機の速度を１秒間表示している。本図は、力行・回生、加速・減速というサーボの重要性能を示しており、正確なトルク発生の状況より容易に理解されるように、期待通りの優れたサーボ性能が得られている。
【００３０】
以上、１実施形態を説明したが、上記の例に代わって、以下のような実施も可能であることを指摘しておく。第１に、新１次角周波数生成器１１に使用する電流信号として上の例では１次電流のｄ軸成分ｉ１ｄ及びｑ軸成分ｉ１ｑを用いたが、この近似推定値の１つとしてこれらの電流指令を用いてもよく、また、１次電圧ｑ軸成分の信号として上の例ではその指令値ｖ１ｑ＊を用いたが、当然のことながら、電圧の実測値あるいはこの他の推定近似値を用いてよいことを指摘しておく。
【００３１】
また、残電圧生成部に使用したフィルタの次数は１次である必要はなく、２次、３次の高次であってよい。また、このフィルタの第１義的目的は、純粋微分の回避であるので、これが回避できる他の方法・手段によっても差し支えないことを指摘しておく。
【００３２】
近似磁束生成部１１ｂに関しては、第１、２、３段に分けて構成する１例を示したが、第１段と第２段は順序を入れ替えることが可能である。また、近似磁束生成部を第１、２、３段のように分けることなく、一体的に構成するこも可能であることを指摘しておく。また、近似磁束生成部内のバランス部は、残電圧生成部で使用したフィルタに対し、両生成部出力信号間のゲインと位相を保つために導入されたものであり、この帯域が十分広い場合、省略可能であることも指摘しておく。当然のことながら、電圧生成部でフィルタを使用しない場合には、バランス部は必要ない。更に、近似磁束生成部１１ｂは、図７に示したように１次インダクタンスの定数ゲインとリミッタによる構成を採用しても、あるいは、図８に示したように１次電流ｄ軸成分の指令値ｉ１ｄ＊が常時一定の場合には一定値（Ｌ１・ｉ１ｄ＊）を出力するようにしてもよいことを指摘しておく。図７、図８の実施形態による場合には、極めて簡単に近似磁束生成部を構成することができる。
【００３３】
残電圧生成部の出力信号に対し、スケーリング処理を実施する場合には、近似磁束生成部の出力信号に対しても、同様なスケーリング処理が当然必要であることを指摘しておく。
【００３４】
１次角周波数演算部１１ｃにおける演算処理に関して、除算による方法を上に例示したが、除算を一切使用しない繰り返し的方法で１次角周波数を決定できることを指摘しておく。
【００３５】
以上説明した新１次角周波数生成器１１の実現は、連続時間的な実現と同様に、離散時間的実現も可能であることを指摘しておく。離散時間的実現に際しては、ｚ演算子による離散時間表現が必要であるが、これには、例えば簡単には、これまでの説明で使用したｓ演算子をオイラー近似、台形近似等を用いてｚ演算子の表現に置換すればよい。離散時間的実現に際しては、ハードウェアー的実現のみならず、ソフトウェアー的実現も、当然ながら、可能であることを指摘しておく。特に、本発明の実行に要する演算量は、従来の滑り周波数形ベクトル制御方法と同程度であるので、最近のマイクロプロセッサ等の演算素子によれば、何等の問題なくソフトウェアー的実現を達成することができる。
【００３６】
以上の例は、直交ｄｑ座標系の位相決定の全部に本発明を用いた例である。次に、直交ｄｑ座標系の位相決定の一部に本発明の用いた場合の１実施形態について説明する。図９は、本１実施形態のベクトル制御装置の全系である。本図９と前実施形態の説明で使用した図１との違いは、本図では、位相合成器１２が付加されている点にある。このため、違いを形成する位相合成器１２を中心に説明する。
【００３７】
位相合成器１２には、本発明によって決定された位相がベクトル［ｃｏｓθ１，ｓｉｎθ１］として入力されている。この位相ベクトルの生成は、図１の実施形態の場合と同様とする。また、また直交ａｂ座標系（すなわち固定座標系）の１次電圧情報ｖ１＊と１次電流情報ｉ１が入力されている。図９では、簡明のため、これらをベクトル信号として１本の太い信号線で表示している。さらに、スケーリング係数用信号として、φ２ｎｄが入力されている。この信号φ２ｎｄは、図４で示しているように、１次磁束ｄ軸成分の推定近似値を生成する過程で生成されているので、これを単に利用すればよい。以上の入力信号対し、直交ｄｑ座標系の最終的な位相ベクトルとして［ｃｏｓθｆ，ｓｉｎθｆ］が、ベクトル回転器６ａ，６ｂに出力されている。
【００３８】
図１０は、上記位相合成器１２の内部を示したものである。図１０の１２ａは低域通過フィルタを示しており、本発明で生成された位相ベクトルをフィルタ処理したつぎの信号が出力される。
【数１０】

同図の１２ｂは、直交ｄｑ座標系の位相情報の視点からは、３種の入力信号を用いて、高域通過フィルタ処理を受けた第２の位相ベクトルの生成を示すものであり、次式のような信号が出力される。
【数１１】

直交ｄｑ座標系の最終的な位相ベクトルである［ｃｏｓθｆ，ｓｉｎθｆ］は、図１０が明快に示しているように、これらの合成として次式のように合成される。
【数１２】

【００３９】
上記に説明した位相合成器１２の使用は、本発明で生成した位相ベクトル信号を直交ｄｑ座標系の位相情報の低域領域に属する位相ベクトル信号として利用し、これにより、他の方法で生成した高域領域のみに有効な位相ベクトル信号を補足し、総合的に良好な位相ベクトル信号を生成している１例である。
【００４０】
上に示した周波数領域的に位相ベクトル信号を分割合成する方法は、本発明で生成した位相ベクトル信号を直交ｄｑ座標系の位相決定の一部に利用する方法の１例であって、本発明で生成した位相ベクトル信号を直交ｄｑ座標系の位相決定の一部に利用する方法は、当業者にとって明白なように、これに限定されるものではない。一般には、誘導電動機の動作状態に応じて、本発明で生成した位相ベクトル信号を種々の形で利用することが可能である。例えば、力行状態で本発明による位相情報を利用するように、力行・回生の状態で利用の切り替えを行う方法を用いてもよい。
【００４１】
図９の実施形態では、本発明による直交ｄｑ座標系の位相情報を、位相ベクトル［ｃｏｓθ１，ｓｉｎθ１］の段階で、他の方法によるものとの併用を行った。本発明による直交ｄｑ座標系の位相情報の利用は、これに限定されるものではなく、１次角の段階で他の方法によるものとの併用することも可能である。図１１は、この例を示したものである。同図のθ１は、本発明による１次角であり、θ２は他の方法による１次角である。同図の１３は１次角合成器であり、２つの１次角を入力信号として得て、これら２信号を処理して最終的な１次角としてθｆを出力している。θｆは正弦信号発生器１０で正弦信号に変換され、位相ベクトルが出力される。
【００４２】
本発明による直交ｄｑ座標系の位相情報の利用は、１次角周波数の段階で他の方法によるものと併用することも可能である。図１２は、この例を示したものである。同図のω１は、本発明による１次角周波数であり、ω２は他の方法による１次角周波数である。同図の１４は１次角周波数合成器であり、２つの１次角周波数を入力信号として得て、これら２信号を処理して最終的な１次角周波数ωｆを出力している。ωｆは積分器９へ送られ積分処理され、θｆが出力される。θｆは正弦信号発生器１０へ送られて正弦信号に変換され、位相ベクトルが出力される。この位相ベクトルは、これまでの説明より明らかなようにベクトル回転器へ送られる。
【００４３】
以上説明した本発明の利用方法の例は、説明の明白化のため、連続時間の表現を用い説明したが、これは、当業者にとって明白なように、連続時間的な実現のみならず、離散時間的実現も可能であることを指摘しておく。離散時間的実現に際しては、ハードウェアー的実現のみならず、ソフトウェアー的実現も、当然ながら、可能であることを指摘しておく。同様に、当業者にとって明白なように、最近の演算素子によれば、何等の問題なく、ソフトウェアー的実現を達成することができることも指摘しておく。
【００４４】
【発明の効果】
誘導電動機の滑り周波数形ベクトル制御方法及びこれを実現した同装置では、電動機の速度情報の確保は不可欠であり、速度情報の確保に起因する種々の問題が不可避的に残置されていた。しかし、以上の説明より明白なように、本発明によれば、速度情報を一切必要としないベクトル制御方法を実現でき、同装置を構成できるので、従来の方法・装置が不可避的に内臓していた速度情報の確保に起因する種々の問題をことごとく解決でき、本発明の目的を十分に達成し得る優れた効果が得られる。
【００４５】
特に、請求項１又は５の本発明によれば、２次磁束と同一方向をｄ軸に選定しこれと直交する軸をｑ軸に選定する直交ｄｑ座標系の位相確立に必要な１次角周波数が、電動機の速度情報を一切用いることなく生成されると言う作用が得れるので、この作用の結果、電動機の速度情報を一切用いないベクトル制御方法を実現及び同装置を構成できるという効果が得られる。
特に、請求項２の本発明によれば、１次磁束ｄ軸成分の推定近似値を一層簡単に生成できるようになると言う作用が得られ、この結果、電動機の速度情報を一切用いないベクトル制御方法を一層簡単に実現できるという効果が得られる。
【００４６】
特に、請求項３の本発明によれば、１次角周波数の生成を、残電圧値に１次磁束ｄ軸成分推定近似値による除算という簡単な演算で実施する場合にも、ゼロあるいは極小値で除算を行うというゼロ割り現象を確実に回避することができるという作用が得られるので、この結果、電動機の速度情報を一切用いないベクトル制御方法を更に簡単に実現できるという効果が得られる。
【００４７】
特に請求項４の本発明によれば、残電圧値の生成工程において、純粋微分を回避することができると言う作用が、ひいては、残電圧値が現実的な信号に対して不必要な過大値を示したり、ノイズ等の外乱に対し不都合な過激反応することを回避することができると言う作用が得られるので、この結果、電動機の速度情報を一切用いないベクトル制御方法の実現に、ノイズ等の外乱に対しロバスト性を付与できると言う効果が得られる。
【図面の簡単な説明】
【図１】１実施形態に係わるベクトル制御装置の概略構成を示すブロック図
【図２】同ベクトル制御装置の新１次角周波数生成器内部の概略構成を示すブロック図
【図３】新１次角周波数生成器を構成する残電圧生成部の概略構成を示すブロック図
【図４】同生成器を構成する近似磁束生成部の概略構成を示すブロック図
【図５】同生成器を構成する１次角周波数演算部の構成を示すブロック図
【図６】実施形態に係わる実験結果の１例
【図７】新１次角周波数生成器を構成する近似磁束生成部の他の実施形態での概略構成を示すブロック図
【図８】新１次角周波数生成器を構成する近似磁束生成部の他の実施形態での概略構成を示すブロック図
【図９】他の実施形態に係わるベクトル制御装置の概略構成を示すブロック図
【図１０】同ベクトル制御装置の位相合成器の概略構成を示すブロック図
【図１１】位相合成法の他の実施形態として、１次角合成器の利用を示すブロック図
【図１２】位相合成法の他の実施形態として、１次角周波数合成器の利用を示すブロック図
【図１３】従来の滑り周波数形ベクトル制御装置の概略構成を示すブロック図
【図１４】同ベクトル制御装置の滑り周波数形１次角周波数生成器内部の概略構成を示すブロック図
【符号の説明】
１ 誘導電動機
２ 速度検出器
３ 電力変換器
４ 電流検出器
５ａ ３相２相変換器
５ｂ ２相３相変換器
６ａ ベクトル回転器
６ｂ ベクトル回転器
７ 電流制御器
８ 滑り周波数形１次角周波数生成器
８ａ ２次磁束ｄ軸成分推定部
８ｂ 滑り角周波数生成部
９ 積分器
１０ 正弦信号発生器
１１ 新１次角周波数生成器
１１ａ 残電圧生成部
１１ｂ 近似磁束生成部
１１ｂａ 基本部
１１ｂｂ バランス部
１１ｂｃ リミッタ部
１１ｃ １次角周波数演算部
１２ 位相合成器
１２ａ 低域フィルタ部
１２ｂ 第２位相ベクトル生成部
１３ １次角合成器
１４ １次角周波数合成器[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a vector control method and apparatus for an induction motor. In particular, the present invention relates to a vector control method and apparatus for establishing vector control, in which the same direction as the secondary magnetic flux is selected as the base axis of the rotation orthogonal coordinate system, and the motor speed information and slip angle frequency information are not required. is there.
[0002]
[Prior art]
Conventionally, as a method for controlling an induction motor with high accuracy and high efficiency, a vector control method is known in which the state of the induction motor is captured and controlled on a rotating coordinate system synchronized with the internal magnetic flux of the motor with zero phase difference. . As the internal magnetic flux of the electric motor, a primary magnetic flux and a secondary magnetic flux are considered, but a vector control method focusing on the secondary magnetic flux is common. This vector control method is roughly divided into a direct method for directly detecting the secondary magnetic flux and an indirect method for estimating the secondary magnetic flux. At present, however, the latter slip frequency vector that does not require highly accurate magnetic flux detection. Practical use of control methods is underway.
[0003]
FIG. 13 is a block diagram schematically showing a typical example when the slip frequency vector control method is implemented as an apparatus and mounted on an induction motor. 1 is an induction motor, 2 is a speed detector of the motor, 3 is a power converter, 4 is a current detector, 5a and 5b are a 3-phase 2-phase converter, a 2-phase 3-phase converter, 6a, respectively. , 6b are vector rotators, 7 is a current controller, 8 is a slip frequency type primary angular frequency generator, 9 is an integrator, and 10 is a sine signal generator. In FIG. 13, various devices from 5a, 5b to 10 constitute a vector control device.
[0004]
In particular, the three types of devices 5a, 5b, 6a, 6b, and 7 mainly constitute a means for controlling the primary current by dividing it into a d-axis component and a q-axis component on the orthogonal dq coordinate system. It is. The three-phase current detected by the current detector 4 is converted into a two-phase current on a fixed coordinate system (hereinafter referred to as ab coordinate system) by the three-phase two-phase converter 5a and then dq by the vector rotator 6a. It is converted into a two-phase current i1d, i1q in the coordinate system. It is sent to the current controller 7. The current controller generates command voltages v1d * and v1q * on the dq coordinate system so that the conversion currents i1d and i1q follow the command values i1d * and i1q * commanded from the outside, and sends them to the vector rotator 6b. send. In 6b, this two-phase signal is converted into a two-phase command voltage in the ab coordinate system and sent to the two-phase / three-phase converter 5b. In 5 b, the two-phase signal is converted into a three-phase command voltage and output as a command value to the power converter 3. The power converter 3 generates electric power according to the command, applies it to the induction motor 1, and drives it.
[0005]
The three types of devices 8, 9, and 10 mainly constitute means for determining the phase of the orthogonal dq coordinate system. A primary current d and q-axis components i1d and i1q are sent from the vector rotator 6a to the slip frequency type primary angular frequency generator 8, and speed information is sent from the speed detector 2. The slip frequency type primary angular frequency generator 8 generates a primary angular frequency using these signals and sends it to the integrator 9 for integration. The integrated signal is sent to the sine signal generator 10, and a plurality of sine signals (cos, sin signals) corresponding to the integration values are used as final phase information of the orthogonal dq coordinate system as vector rotators 6a, Output to 6b. In the figure, for the sake of simplicity, a plurality of sine signals are captured as one phase vector and represented by a single thick line. In the following, also in other drawings, the bold lines indicate vector signal lines.
[0006]
FIG. 14 shows details of the slip frequency type primary angular frequency generator 8. In FIG. 14, reference numeral 8a denotes a secondary magnetic flux d-axis component estimator, and 8b denotes a slip angular frequency generator. The secondary magnetic flux d-axis component estimation unit 8a basically calculates and synthesizes the estimated value φ2dh of the secondary magnetic flux d-axis component according to the relationship of the following equation.
[Expression 1]

Here, R2 is a secondary resistance, L2 is a secondary inductance, M is a mutual inductance, and s is a differential operator d / dt. Further, the slip angular frequency generation unit 8b basically generates a slip angular frequency ωs using the estimated d-axis component value of the secondary magnetic flux and the primary current q-axis component i1q as shown in the following equation. Yes.
[Expression 2]

The slip frequency type primary angular frequency generator 8 determines the primary angular frequency ω1 by adding the slip angular frequency to the electrical angular velocity ω2n of the motor obtained from the speed detector, as shown in the following equation.
[Equation 3]

[0007]
[Problems to be solved by the invention]
In the conventional slip frequency vector control method and apparatus, the speed information of the motor is indispensable for generating the primary angular frequency, as clearly shown in FIG. For this reason, a speed detection system centered on a speed detector represented by an encoder has to be constructed. However, the speed detection system including the speed detector is extremely weak as compared with the induction motor main body, and the reliability of the entire drive system including the induction motor has been significantly reduced. For example, the speed detector is extremely weak against the impact received by the motor, and the signal of the speed detector is generally several volts, and special consideration must be given to noise during transmission. Consideration was also required for safe wiring of signal lines. In addition, countermeasures should also be taken in the event that a vulnerable speed detection system malfunctions. Implementation of such a speed detection system is also disadvantageous in terms of cost. In addition, when the detector is mounted, the axial dimension of the motor is increased, which is disadvantageous in the installation of the motor. As a method and means for speed detection, there are some reports that constitute a speed detection system that detects speed in the form of speed estimation without using a speed detector such as an encoder. Problems such as detection accuracy, detection range, detection speed, and detection calculation load were newly induced, and the control performance deteriorated considerably compared to the case with a speed detector. In addition, the vector control method and apparatus constituting such a speed detection system are more complicated than those with a speed detector, which is disadvantageous in this respect.
[0008]
As explained above, as long as it is based on the slip frequency vector control method that has been put into practical use, it is indispensable to secure the speed information of the motor in order to realize it. The problem has been inevitably left behind.
[0009]
The present invention has been made based on the above background, and an object thereof is to provide a vector control method and apparatus that do not require any speed information and can solve various problems caused by securing speed information. There is to do.
[0010]
[Means for Solving the Problems]
In order to achieve the above object, the invention of claim 1 is an orthogonal dq coordinate system in which the same direction as the secondary magnetic flux as a two-dimensional vector quantity is selected as the d-axis and the axis orthogonal to this is selected as the q-axis. In a vector control method for an induction motor having a step of dividing and controlling a primary current into a d-axis component and a q-axis component and a step of determining a phase of an orthogonal dq coordinate system, the primary angular frequency and the primary magnetic flux The product of the estimated magnetic flux value of the d-axis component and the primary resistance in which the q-axis component of the primary voltage or the estimated current value of the primary current is connected in series from the q-axis component of the primary voltage or the estimated voltage value thereof. Calculation of the residual voltage value and the primary magnetic flux d-axis component estimated approximate value so that the residual voltage value obtained by reducing the voltage drop generated when flowing through the primary leakage inductance is directly equalized. The primary angular frequency is determined by processing, and the determined primary angle The wave number, is characterized in that for use as all or part of the primary angular frequency used in the phase determination step.
[0011]
According to a second aspect of the present invention, in the vector control method for an induction motor according to the first aspect, the steady-state value of the primary magnetic flux d-axis component estimation approximate value is the steady-state value of the primary current d-axis component or the current estimation approximate value thereof. The primary magnetic flux d-axis component estimated approximate value is determined so as to be equal to a value obtained by multiplying the value by the primary inductance.
[0012]
According to a third aspect of the present invention, in the vector control method for an induction motor according to the first and second aspects, a positive lower limit value is set for the primary magnetic flux d-axis component estimation approximate value, and the primary magnetic flux d-axis component estimation approximation is set. The value is always larger than the lower limit value.
[0013]
According to a fourth aspect of the present invention, in the vector control method for an induction motor according to the first aspect, a filtering process using a low-pass filter is performed in the step of generating the residual voltage value.
[0014]
In the invention of claim 5, the primary current is d on the orthogonal dq coordinate system in which the same direction as the secondary magnetic flux as the two-dimensional vector quantity is selected as the d-axis and the axis orthogonal thereto is selected as the q-axis. In a vector control device for an induction motor having means for dividing and controlling an axial component and a q-axis component, and means for determining a phase of an orthogonal dq coordinate system, the phase determining means includes a primary magnetic flux d-axis component. An approximate magnetic flux generator for generating an estimated approximate value of magnetic flux, a primary resistance q-axis component of the primary voltage or a primary resistance q-axis component of the primary current or an estimated current approximate value thereof connected in series A product of a residual voltage generation unit that generates a residual voltage value obtained by reducing a voltage drop generated when flowing to the primary leakage inductance, and an output signal of the approximate magnetic flux generation unit and the primary angular frequency is the residual voltage generation unit. The first order to be directly equal to the output signal It is characterized in that at least provided with a primary angular frequency generator means having a primary angular frequency calculator for determining the frequency.
[0015]
According to the first or fifth aspect of the present invention, a primary angle for establishing a phase necessary for an orthogonal dq coordinate system in which the same direction as the secondary magnetic flux is selected as the d-axis and an axis orthogonal thereto is selected as the q-axis. The fact that the effect that the frequency is generated without using any speed information of the motor can be obtained will be described first using an accurate signal that does not include noise or the like.
[0016]
If the primary angular frequency is expressed as ω1, the d-axis component of the primary magnetic flux as φ1d, the q-axis component of the primary voltage as v1q, the d-axis component of the primary current as i1d, and the q-axis component as i1q. The invention is such that the signal represented by the right side of the equation (4) is directly subjected to an operation such as division by the signal φ1d so that the following relational equation (4) that does not include any speed signal of the motor is satisfied. The frequency ω1 is to be generated.
[Expression 4]

Here, Lp1 is a total primary leakage inductance defined as Lp1 = (L1−M · M / L2) using the primary inductance L1, the secondary inductance L2, and the mutual inductance M. The signal in the second term on the right side of the above equation indicates the voltage drop that occurs when the primary current q-axis component is passed through the series-connected primary resistance and total primary leakage inductance. Indicates a residual voltage value obtained by subtracting this drop voltage from the primary voltage q-axis component. Further, the primary magnetic flux d-axis component φ1d at this time is to be estimated and approximated according to the following equations (5) and (6) using the primary current d-axis component i1d without being directly detected.
[Equation 5]

[Formula 6]

W2 and R2n in the above formula are electric motor parameters defined as W2 = R2 / L2 and R2n = (M · M · R2) / (L2 · L2).
[0017]
By the way, in the induction motor, in the internal state, the relationship represented by the following expressions (7) and (8) is generally established.
[Expression 7]

[Equation 8]

Here, φ2q is the q-axis component of the secondary magnetic flux.
[0018]
Here, when Expression (4) indicating the basis of the present invention is used for Expression (7), a result of sφ2q = 0 is obtained. Further, when Expression (6) used for approximate estimation of the primary magnetic flux in the present invention is used for Expression (8), a result of φ2q = 0 is obtained. The result of sφ2q = 0 and φ2q = 0 means that a phase for an orthogonal dq coordinate system is established in which the same direction as the secondary magnetic flux is selected as the d-axis and the axis orthogonal to this is selected as the q-axis. Yes. According to the above explanation, in order to obtain an estimated approximate value of the primary magnetic flux d-axis component, the use of the equations (5) and (6) tends to be considered indispensable, but this use can be relaxed. is there. This will be described again when the operation of the invention of claim 2 is described. As is apparent from the above description, according to the present invention of claim 1 or 5, essentially the orthogonal dq is selected in which the same direction as the secondary magnetic flux is selected as the d-axis and the axis orthogonal thereto is selected as the q-axis. An effect is obtained that the primary angular frequency necessary for establishing the phase of the coordinate system is generated without using any speed information of the motor.
[0019]
In the present invention according to claim 1 or 5, when the q-axis component of the primary voltage and the q-axis component of the primary current cannot be obtained accurately due to the influence of noise or the like, the primary angle is calculated based on these approximate estimates. The frequency is generated. Since the present invention uses the principle of generating the primary angular frequency to directly satisfy the essential condition necessary for establishing the phase of the orthogonal dq coordinate system, the approximation of the above signal can be performed even when the influence of noise or the like is strong. By using the estimated value, an effect that a realistic primary angular frequency can be generated is obtained.
[0020]
According to the second aspect of the present invention, in the vector control method according to the first aspect of the present invention, the steady-state value φ1d (= constant) of the primary magnetic flux d-axis component estimation approximate value is changed to the primary current d-axis component or its current estimation. The approximate steady value i1d (= constant) is determined to be equal to a value obtained by multiplying the primary inductance L1, that is, the steady value φ1d is determined so that the following relationship is maintained.
[Equation 9]

The above equation is completely equivalent to the equations (5) and (6) with respect to the steady value. However, it is much simpler than equations (5) and (6). In vector control, the primary current d-axis component directly related to the internal magnetic flux is maintained at a constant steady value except in exceptional cases. It is possible to obtain an effect that it is very easy to generate the estimated approximate value.
[0021]
According to the third aspect of the present invention, in the vector control method according to the first and second aspects of the present invention, a positive lower limit value is set for the primary magnetic flux d-axis component estimation approximate value, and the primary magnetic flux d-axis component estimation approximation is performed. Since the value is always larger than the lower limit value, the primary angular frequency is generated by a simple calculation called division of the residual voltage value by the primary magnetic flux d-axis component estimation approximate value. There is an effect that the zero division phenomenon of dividing by zero or the minimum value can be surely avoided.
[0022]
According to the fourth aspect of the present invention, in the vector control method according to the first aspect of the present invention, since the filtering process using the low-pass filter is performed in the residual voltage value generation step, As shown on the right side, a pure differentiation is required for voltage generation involving the total primary leakage inductance Lp1, and an effect that this pure differentiation can be avoided is obtained. As a result, it is possible to avoid an unnecessary excessive reaction with respect to a disturbance such as noise or a residual voltage value indicating an unnecessary excessive value with respect to a realistic signal.
[0023]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 1 shows an example of the entire system of an induction motor vector control apparatus to which the present invention is applied. FIG. 1 shows a new primary angular frequency generator 11 that is directly related to the present invention, and two other components that do not have a speed detector in the induction motor. This is the same as FIG. 13 in the case of the shape vector control device. That is, in the means for determining the phase of the orthogonal dq coordinate system, the integrator 9 integrates the primary angular frequency generated by the new primary angular frequency generator, and further, the integrated signal is converted into a sine signal generator. 10, a sine signal (cos, sin signal) corresponding to the integral value is generated and output to the vector rotators 6 a and 6 b as final phase information of the orthogonal dq coordinate system. This is the same as the control method and apparatus. Therefore, in the present embodiment, an example of the new primary angular frequency generator 11 that embodies the primary angular frequency generating means will be described in detail.
[0024]
In the example of FIG. 1, the new primary angular frequency generator 11 has its command value v1q * as an approximate estimated value of the d-axis component i1d, the q-axis component i1q, and the primary voltage q-axis component of the primary current. Have been entered.
[0025]
FIG. 2 shows an internal configuration of the new primary angular frequency generator 11. 11a in the figure is a residual voltage obtained by subtracting the voltage drop generated when the q-axis component of the primary current flows from the primary voltage q-axis component command value to the primary resistance connected in series and the total primary leakage inductance. It is a residual voltage generation part which produces | generates a value. 11b is an approximate magnetic flux generator that generates a magnetic flux estimation approximate value of the primary magnetic flux d-axis component from the primary current d-axis component i1d, and 11c is the product of the output signal of the approximate magnetic flux generator and the primary angular frequency. Is a primary angular frequency calculation unit that determines the primary angular frequency so as to be directly equal to the output signal of the residual voltage generation unit.
[0026]
FIG. 3 shows an example of the configuration of the remaining voltage value generation unit 11a. The example of FIG. 3 is characterized in that a filtering process using a low-pass filter f / (s + f) is performed. As shown in equation (4), the voltage generation involving the total primary leakage inductance Lp1 requires a pure differentiation in an ideal situation. In the example of FIG. 3, this pure differentiation is avoided by a low-pass filter. It clearly shows that
[0027]
FIG. 4 shows details of an example of the configuration of the approximate magnetic flux generation unit. In this example, in order to facilitate understanding, the approximate magnetic flux generating unit is configured in a three-stage configuration in which the configuration is divided according to function. The first stage of the approximate magnetic flux generation unit is the basic unit 11ba, which faithfully embodies the relationships shown in the equations (5) and (6). It should be particularly pointed out that the output signal of the basic part is equal to the value obtained by multiplying the primary inductance L1 with respect to the steady value of the primary current d-axis component which is the input of the basic part 11ba. The approximate magnetic flux generation unit second stage 11bb is a balance unit and is prepared for the purpose of maintaining a balance between gain and phase with respect to the output signal of the residual voltage generation unit. Therefore, in this example, the same filter as the low-pass filter f / (s + f) introduced into the residual voltage generation unit may be simply installed. The third stage of the approximate magnetic flux generation unit is a limiter unit 11bc, so that the primary magnetic flux d-axis component estimated approximate value, which is the final output signal of the approximate magnetic flux generation unit, is always larger than a preset positive lower limit value. Can take a value.
[0028]
FIG. 5 shows an example of the configuration of the primary angular frequency calculation unit 11c. The function of the primary angular frequency calculation unit is to determine the primary angular frequency so that the product of the output signal of the approximate magnetic flux generation unit and the primary angular frequency is directly equal to the output signal of the residual voltage generation unit. However, in this example, the simplest determination method of dividing the output signal of the residual voltage generation unit by the output signal of the approximate magnetic flux generation unit is used. Even in the case of division, since the approximate value of primary magnetic flux d-axis component estimation is always larger than the preset lower limit value in the above-described approximate magnetic flux generation unit, zero in which division is performed with zero or a minimum value. The split phenomenon can be avoided reliably.
[0029]
FIG. 6 shows an example of experimental results when the induction motor is driven and controlled by the vector control apparatus of the above embodiment using the present invention. The upper left indicates the command value of the primary current d-axis component, the upper right indicates the generated primary angular frequency, the lower left indicates the generated torque, and the lower right indicates the motor speed for one second. This figure shows the important performances of servo such as power running / regeneration, acceleration / deceleration, and excellent servo performance as expected is obtained so that it can be easily understood from the situation of accurate torque generation.
[0030]
Although one embodiment has been described above, it should be pointed out that the following implementation is possible instead of the above example. First, in the above example, the d-axis component i1d and the q-axis component i1q of the primary current are used as the current signal used for the new primary angular frequency generator 11. However, as one of the approximate estimation values, The current command may be used, and the command value v1q * is used as the signal of the primary voltage q-axis component in the above example. However, as a matter of course, the measured value of the voltage or other estimated approximation value is used. Point out that it can be used.
[0031]
Further, the order of the filter used in the residual voltage generation unit need not be the first order, and may be the second order or the third order higher order. Also, it should be pointed out that the primary purpose of this filter is to avoid pure differentiation, so that other methods and means that can avoid this may be used.
[0032]
As for the approximate magnetic flux generation unit 11b, an example in which the first magnetic flux generation unit 11b is divided into the first, second, and third stages has been shown. Also, it should be pointed out that the approximate magnetic flux generation unit can be configured integrally without dividing it into the first, second and third stages. In addition, the balance unit in the approximate magnetic flux generation unit is introduced to maintain the gain and phase between the output signals of the two generation units with respect to the filter used in the residual voltage generation unit. It should also be pointed out that it can be omitted. As a matter of course, when the filter is not used in the voltage generation unit, the balance unit is not necessary. Further, the approximate magnetic flux generator 11b may employ a configuration using a constant gain and a limiter of the primary inductance as shown in FIG. 7, or a command value of the primary current d-axis component as shown in FIG. It should be pointed out that a constant value (L1 · i1d *) may be output when i1d * is always constant. In the case of the embodiment of FIGS. 7 and 8, the approximate magnetic flux generator can be configured very simply.
[0033]
It should be pointed out that when the scaling process is performed on the output signal of the residual voltage generation unit, a similar scaling process is naturally necessary for the output signal of the approximate magnetic flux generation unit.
[0034]
Regarding the calculation processing in the primary angular frequency calculation unit 11c, the method by division is exemplified above, but it should be pointed out that the primary angular frequency can be determined by an iterative method without using any division.
[0035]
It should be pointed out that the realization of the new primary angular frequency generator 11 described above can be realized in discrete time as well as in continuous time. In the discrete time realization, a discrete time expression by the z operator is necessary. For this purpose, for example, the s operator used in the above description is expressed by using Euler approximation, trapezoidal approximation, or the like. Replace with an operator expression. It should be pointed out that, in discrete time implementation, not only hardware implementation but also software implementation is possible. In particular, the amount of computation required to execute the present invention is comparable to that of the conventional sliding frequency vector control method, and therefore, software implementation can be achieved without any problems with recent computing elements such as microprocessors. be able to.
[0036]
The above example is an example in which the present invention is used for all the phase determination of the orthogonal dq coordinate system. Next, an embodiment in which the present invention is used as part of the phase determination of the orthogonal dq coordinate system will be described. FIG. 9 shows the entire system of the vector control apparatus according to the first embodiment. The difference between FIG. 9 and FIG. 1 used in the description of the previous embodiment is that a phase synthesizer 12 is added. For this reason, the phase synthesizer 12 that forms the difference will be mainly described.
[0037]
The phase determined by the present invention is input to the phase synthesizer 12 as a vector [cos θ1, sin θ1]. The generation of this phase vector is the same as in the embodiment of FIG. Further, primary voltage information v1 * and primary current information i1 in an orthogonal ab coordinate system (that is, a fixed coordinate system) are input. In FIG. 9, for the sake of simplicity, these are represented as vector signals by a single thick signal line. Further, φ2nd is input as a scaling coefficient signal. Since this signal φ2nd is generated in the process of generating the estimated approximate value of the primary magnetic flux d-axis component as shown in FIG. 4, it can be simply used. [Cos θf, sin θf] is output to the vector rotators 6a and 6b as the final phase vector of the orthogonal dq coordinate system for the above input signals.
[0038]
FIG. 10 shows the inside of the phase synthesizer 12. Reference numeral 12a in FIG. 10 denotes a low-pass filter, which outputs the next signal obtained by filtering the phase vector generated in the present invention.
[Expression 10]

12b in the figure shows the generation of the second phase vector subjected to the high-pass filter processing using three kinds of input signals from the viewpoint of the phase information of the orthogonal dq coordinate system. The following signal is output.
[Expression 11]

[Cos θf, sin θf], which is the final phase vector of the orthogonal dq coordinate system, is synthesized as shown in the following equation as clearly shown in FIG.
[Expression 12]

[0039]
The use of the phase synthesizer 12 described above uses the phase vector signal generated in the present invention as a phase vector signal belonging to the low-frequency region of the phase information of the orthogonal dq coordinate system, and thereby generated by another method. This is an example in which a phase vector signal effective only in the high frequency region is supplemented to generate a comprehensively good phase vector signal.
[0040]
The above-described method of dividing and synthesizing a phase vector signal in the frequency domain is an example of a method of using the phase vector signal generated in the present invention as part of the phase determination of the orthogonal dq coordinate system. As will be apparent to those skilled in the art, the method of using the phase vector signal generated in (1) as part of the phase determination of the orthogonal dq coordinate system is not limited thereto. In general, the phase vector signal generated in the present invention can be used in various forms according to the operating state of the induction motor. For example, a method of switching use in the power running / regeneration state so that the phase information according to the present invention is used in the power running state may be used.
[0041]
In the embodiment of FIG. 9, the phase information of the orthogonal dq coordinate system according to the present invention is used in combination with another phase method at the stage of the phase vector [cos θ1, sin θ1]. The use of the phase information of the orthogonal dq coordinate system according to the present invention is not limited to this, and it can be used in combination with another method at the primary angle stage. FIG. 11 shows this example. In the figure, θ1 is a primary angle according to the present invention, and θ2 is a primary angle according to another method. 13 in the figure is a primary angle synthesizer, which obtains two primary angles as input signals, processes these two signals, and outputs θf as a final primary angle. θf is converted into a sine signal by the sine signal generator 10 and a phase vector is output.
[0042]
The use of the phase information of the orthogonal dq coordinate system according to the present invention can be used in combination with another method at the primary angular frequency stage. FIG. 12 shows this example. In the figure, ω1 is a primary angular frequency according to the present invention, and ω2 is a primary angular frequency according to another method. 14 in the figure is a primary angular frequency synthesizer, which obtains two primary angular frequencies as input signals, processes these two signals, and outputs a final primary angular frequency ωf. ωf is sent to the integrator 9 and integrated, and θf is output. θf is sent to the sine signal generator 10 to be converted into a sine signal, and a phase vector is output. This phase vector is sent to the vector rotator as will be apparent from the above description.
[0043]
The above-described examples of using the present invention have been described using continuous-time expressions for clarity of explanation, but this is not limited to continuous-time implementations, as will be apparent to those skilled in the art. It should be pointed out that time realization is possible. It should be pointed out that, in discrete time implementation, not only hardware implementation but also software implementation is possible. Similarly, it should be pointed out that, as will be apparent to those skilled in the art, software implementations can be achieved without any problems with recent computing elements.
[0044]
【The invention's effect】
In the slip frequency type vector control method for an induction motor and the apparatus that realizes the slip frequency type vector control method, it is indispensable to secure the speed information of the motor, and various problems resulting from securing the speed information are inevitably left behind. However, as is clear from the above description, according to the present invention, a vector control method that does not require any speed information can be realized and the apparatus can be configured, so that conventional methods and apparatuses are inevitably incorporated. Accordingly, various problems caused by securing the speed information can be solved, and an excellent effect can be obtained in which the object of the present invention can be sufficiently achieved.
[0045]
In particular, according to the first or fifth aspect of the present invention, the primary angle necessary for establishing the phase of the orthogonal dq coordinate system in which the same direction as the secondary magnetic flux is selected as the d-axis and the axis orthogonal thereto is selected as the q-axis. Since the effect that the frequency is generated without using any speed information of the motor can be obtained, the result of this action is that the vector control method that does not use the speed information of the motor can be realized and the apparatus can be configured. can get.
In particular, according to the second aspect of the present invention, it is possible to obtain an effect that the estimated approximate value of the primary magnetic flux d-axis component can be generated more easily. As a result, vector control without using any speed information of the motor. The effect that the method can be realized more easily is obtained.
[0046]
In particular, according to the third aspect of the present invention, even when the primary angular frequency is generated by a simple operation of dividing the residual voltage value by the primary magnetic flux d-axis component estimation approximate value, zero or a minimum value is obtained. As a result, it is possible to reliably avoid the zero-dividing phenomenon of dividing by 1. As a result, it is possible to more easily realize a vector control method that does not use any motor speed information.
[0047]
In particular, according to the fourth aspect of the present invention, the action that the pure differentiation can be avoided in the step of generating the residual voltage value, and consequently the excessive value that the residual voltage value is unnecessary for a real signal. Or an adverse reaction to noise and other external disturbances can be avoided. As a result, it is possible to implement a vector control method that uses no motor speed information. The effect that the robustness can be imparted to the external disturbance is obtained.
[Brief description of the drawings]
FIG. 1 is a block diagram showing a schematic configuration of a vector control apparatus according to an embodiment.
FIG. 2 is a block diagram showing a schematic configuration inside a new primary angular frequency generator of the vector control device.
FIG. 3 is a block diagram showing a schematic configuration of a residual voltage generating unit constituting the new primary angular frequency generator.
FIG. 4 is a block diagram showing a schematic configuration of an approximate magnetic flux generation unit constituting the generator.
FIG. 5 is a block diagram showing a configuration of a primary angular frequency calculation unit constituting the generator.
FIG. 6 shows an example of experimental results according to the embodiment.
FIG. 7 is a block diagram showing a schematic configuration in another embodiment of the approximate magnetic flux generation unit constituting the new primary angular frequency generator.
FIG. 8 is a block diagram showing a schematic configuration in another embodiment of the approximate magnetic flux generation unit constituting the new primary angular frequency generator.
FIG. 9 is a block diagram showing a schematic configuration of a vector control apparatus according to another embodiment.
FIG. 10 is a block diagram showing a schematic configuration of a phase synthesizer of the vector control device.
FIG. 11 is a block diagram showing the use of a primary angle synthesizer as another embodiment of the phase synthesis method;
FIG. 12 is a block diagram showing the use of a primary angular frequency synthesizer as another embodiment of the phase synthesis method;
FIG. 13 is a block diagram showing a schematic configuration of a conventional slip frequency vector control device.
FIG. 14 is a block diagram showing a schematic configuration inside a slip frequency type primary angular frequency generator of the vector control device;
[Explanation of symbols]
1 Induction motor
2 Speed detector
3 Power converter
4 Current detector
5a 3 phase 2 phase converter
5b 2 phase 3 phase converter
6a vector rotator
6b vector rotator
7 Current controller
8 Slip frequency type primary angular frequency generator
8a Secondary magnetic flux d-axis component estimator
8b Sliding angular frequency generator
9 Integrator
10 Sine signal generator
11 New primary angular frequency generator
11a Residual voltage generator
11b Approximate magnetic flux generator
11ba Basic part
11bb balance section
11bc limiter section
11c Primary angular frequency calculator
12 Phase synthesizer
12a Low-pass filter section
12b Second phase vector generator
13 Primary angle synthesizer
14 Primary angular frequency synthesizer

Claims (5)

The primary current is divided into the d-axis component and the q-axis component on the orthogonal dq coordinate system in which the same direction as the secondary magnetic flux as the two-dimensional vector quantity is selected as the d-axis and the orthogonal axis is selected as the q-axis. In the vector control method for an induction motor, comprising the steps of controlling and determining the phase of the orthogonal dq coordinate system,
From the product of the primary angular frequency and the magnetic flux estimation approximate value of the d-axis component of the primary magnetic flux, the q-axis component of the primary voltage or the voltage estimation approximate value thereof, the q-axis component of the primary current or the current estimation approximate value thereof In order to directly equalize the residual voltage value obtained by reducing the voltage drop generated when the primary resistance connected in series and the total primary leakage inductance are passed, the residual voltage value and the primary voltage The primary angular frequency is determined by arithmetic processing with the magnetic flux d-axis component estimation approximate value, and the determined primary angular frequency is used as all or part of the primary angular frequency used in the phase determination step. A vector control method for an induction motor. The primary magnetic flux d so that the steady-state value of the primary magnetic flux d-axis component estimation approximate value is equal to the primary current d-axis component or the steady-state value of the current estimation approximate value multiplied by the primary inductance. 2. The vector control method for an induction motor according to claim 1, wherein an approximate value for estimating the axial component is determined. The positive lower limit value is set for the primary magnetic flux d-axis component estimated approximate value, and the primary magnetic flux d-axis component estimated approximate value is always larger than the lower limit value. A vector control method for induction motors 1 and 2. 2. The induction motor vector control method according to claim 1, wherein a filtering process using a low-pass filter is performed in the residual voltage value generation step. The primary current is divided into the d-axis component and the q-axis component on the orthogonal dq coordinate system in which the same direction as the secondary magnetic flux as the two-dimensional vector quantity is selected as the d-axis and the orthogonal axis is selected as the q-axis. In the induction motor vector control device having means for controlling and means for determining the phase of the orthogonal dq coordinate system,
In the configuration of the phase determining means, an approximate magnetic flux generation unit for generating a magnetic flux estimation approximate value of the primary magnetic flux d-axis component, a q-axis component of the primary voltage, or a q-axis component of the primary current from the voltage estimation approximate value or A residual voltage generator for generating a residual voltage value obtained by reducing a voltage drop generated when the estimated current approximate value is passed through a primary resistor and a total primary leakage inductance connected in series; and an output signal of the approximate magnetic flux generator Primary angular frequency generation means having a primary angular frequency calculation unit for determining the primary angular frequency so that the product of the primary angular frequency and the output signal of the residual voltage generation unit are directly equal to each other. A vector control device for an induction motor, comprising:

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