JP3759448B2 - OFDM signal combining receiver - Google Patents

Description

ここで、（ｉｍｏｄ４）はｉを４で除算したときの余りを表し、ｐは非負整数である。
【００２４】
また、任意のシンボル番号ｉ、キャリア番号ｋで特定されるＯＦＤＭ信号ベクトルｕ _k(ｉ)を以下の式（２）によって定義し、任意のシンボル番号ｉ、ブランチ番号ｌで特定されるＯＦＤＭ信号ベクトルｖ _l(ｉ)を以下の式（３）によって定義する。
【数２】

ここで、ｕ_l,k(ｉ)は、ＯＦＤＭ信号の周波数スペクトル（以下、ＯＦＤＭスペクトルという。）中のブランチ番号ｌ、キャリア番号ｋ、およびシンボル番号ｉで特定される信号であり、Ｔは転置ベクトルを表す。
【００２５】
また、ＯＦＤＭスペクトルの合成信号ｙ_k(ｉ)を以下の式（４）によって定義する。
【数３】

ここで、ｗ b _k(ｉ)はキャリア番号ｋ、ブランチ番号ｌによって特定される合成重み付け係数ベクトルであり、以下の式（５）によって定義され、Ｈは複素共役転置ベクトルを表す。
【数４】

ここで、ｗ_l,k(ｉ)はブランチ番号ｌ、スペクトル番号ｋ、およびシンボル番号ｉによって特定される重み係数である。
【００２６】
次に、上記の式（５）で表される合成重み付け係数ベクトルｗ b _k(ｉ)をＬＭＳアルゴリズムを適用して更新する方法について説明する。
時刻ｉ、すなわちシンボル番号ｉにおけるキャリア番号k_pで指定されるＳＰ信号のブランチ方向についての合計値ｄ_kp(ｉ)と、シンボル番号ｉ、キャリア番号k_pで特定されるＳＰ信号の合成信号ｙ_kp(ｉ)との差（以下、事後推定誤差という。）ｅ_kp(ｉ)は、以下の式（６）によって表される。
【数５】

【００２７】
同一のＳＰキャリア番号については、４シンボル周期でＳＰ信号が繰り返されるため、シンボル番号ｉを基準に更新される次のＳＰ信号のシンボル番号をｉ＋４とする。そして、キャリア番号ｋ_pによって特定される合成重み付け係数ベクトルの更新値ｗ b _kp(ｉ＋４)を以下の式（７）で定義する。
【数６】

ここで、μはステップ・サイズ・パラメータと呼ばれる更新の前後の変化量を規定するパラメータであり、* は複素共役を表す。
【００２８】
ＳＰ信号以外のシンボル−キャリア空間における信号であるデータ信号の合成に用いる合成重み付け係数ベクトルｗ b _k(ｉ)は、このようにして算出された、各シンボル番号ｉと各ＳＰキャリア番号ｋ_pとで特定される合成重み付け係数ベクトルｗ b _kp(ｉ)およびｗ b _kp(ｉ＋４)に基づいて補間によって生成されるものとする。補間の方法については後述する。
【００２９】
以下、ＯＦＤＭ信号合成用受信装置１００の各構成手段について説明する。
ＦＦＴ手段１０１（０）は、ブランチ（図１には、アンテナとして示す。）０で受信したＯＦＤＭ信号に、公知の高速フーリエ変換処理（以下、ＦＦＴ処理という。）を施し、ＦＦＴ処理によって得られたＯＦＤＭ信号の周波数スペクトル（以下、ＯＦＤＭスペクトルという。）から抽出したＯＦＤＭ信号ベクトルｖ _l(ｉ)（上記の式（３）を参照）、ＯＦＤＭ信号が受信された時刻であるシンボル番号ｉの情報、ブランチ番号ｌその他の必要な情報をパイロットキャリア抽出手段１１０およびキャリア合成手段１２０に出力するための手段である。
【００３０】
以下、特に断らない限り、ブランチ番号ｌ等の添字、シンボル番号ｉ等の引数が付された、ＯＦＤＭ信号ベクトル等の信号または情報等を入出力すると記載されている場合は、それを特定するために必要な添字、引数等はその信号等に伴って入出力されるものとし、その記載を省略する。
ＦＦＴ手段１０１（１）〜１０１（Ｌ−１）は、各々、ブランチ１〜（Ｌ−１）で受信したＯＦＤＭ信号に上記ＦＦＴ手段１０１（０）が行う処理と同様の処理を施す。
【００３１】
パイロットキャリア抽出手段１１０は、ＦＦＴ手段１０１（０）〜１０１（Ｌ−１）から出力された各ＯＦＤＭ信号ベクトルｖ _l(ｉ)を入力とし、シンボル番号がｉのときのＳＰ信号を、上記の式（１）に示すＳＰキャリア番号ｋ_pで特定されるキャリアの信号として、入力された各ＯＦＤＭ信号ベクトルｖ _l(ｉ)から抜き出し、抜き出した各ブランチのＳＰ信号ｕ_l,k(ｉ)を上記の式（２）に示すように配列して、ＯＦＤＭ信号ベクトルｕ _k(ｉ)を生成し、各ＯＦＤＭ信号ベクトルｕ _k(ｉ)をパイロットキャリア合成手段１３０および表示部１４０に出力するための手段である。
【００３２】
キャリア合成手段１２０は、ＦＦＴ手段１０１（０）〜１０１（Ｌ−１）から出力された各ＯＦＤＭ信号ベクトルｖ _l(ｉ)と後述する内挿手段１７０から出力される合成重み付け係数ベクトルｗ b _k(ｉ)とを入力とし、上記の式（２）、式（４）、および式（５）に基づいて合成信号ｙ_k(ｉ)を形成し、合成信号ｙ_k(ｉ)を外部に出力するための手段である。
【００３３】
以下、キャリア合成手段１２０において用いる合成重み付け係数ベクトルｗ b _k(ｉ)を生成するためにパイロットキャリア合成手段１３０、係数更新手段１４０、差分演算手段１５０、パイロット信号発生手段１６０、および内挿手段１７０が行う処理について説明する。
【００３４】
パイロットキャリア合成手段１３０は、パイロットキャリア抽出手段１１０から出力される各ＯＦＤＭ信号ベクトルｕ _k(ｉ)と後述する係数更新手段１４０から出力される合成重み付け係数ベクトルｗ b _kp(ｉ)とを入力とし、上記の式（４）に基づいて合成信号ｙ_kp(ｉ)を生成して差分演算手段１５０に出力すると共に、入力された合成重み付け係数ベクトルｗ b _kp(ｉ)を内挿手段１７０に出力し、シンボル番号ｉの情報およびキャリア番号ｋ_pの情報をパイロット信号発生手段１６０に出力するための手段である。
【００３５】
パイロット信号発生手段１６０は、予めＯＦＤＭ信号に挿入されるＳＰ信号の値に対応する所定の値（以下、基準合成値という。）ｄ_k(ｉ)をシンボル番号ｉ、ＳＰキャリア番号ｋ_pに対応させて記憶しておき、パイロットキャリア合成手段１３０から出力されたシンボル番号ｉの情報およびＳＰキャリア番号ｋ_pの情報を入力とし、入力されたこれらの情報に基づいてシンボル番号ｉおよびＳＰキャリア番号ｋ_pによって特定される基準合成値ｄ_kp(ｉ)を後述する差分演算手段１５０に出力するための手段である。
【００３６】
差分演算手段１５０は、パイロットキャリア合成手段１３０から出力される合成信号ｙ_kp(ｉ)とパイロット信号発生手段１６０から出力される基準合成値ｄ_kp(ｉ)とを入力とし、上記の式（６）に示す事後推定誤差ｅ_kp(ｉ)を算出し、後述する係数更新手段１４０に出力するための手段である。
【００３７】
係数更新手段１４０は、パイロットキャリア抽出手段１１０から出力されたＯＦＤＭ信号ベクトルｕ _kp(ｉ)および差分演算手段１５０から出力される事後推定誤差ｅ_kp(ｉ)を入力とし、上記の式（７）に基づいて合成重み付け係数ベクトルｗ b _kp(ｉ)の更新を行い、更新によって得られた合成重み付け係数ベクトルｗ b _kp(ｉ＋４)をパイロットキャリア合成手段１３０に出力するための手段である。ここで、係数更新手段１４０は、シンボル番号ｉでの更新処理が完了した場合、更新処理によって得られた合成重み付け係数ベクトルｗ b _kp(ｉ＋４)を次の処理のために記憶しておくのでも良い。なお、最初に行う更新処理に用いる合成重み付け係数ベクトルｗ b _kp(０)の値としては、所定の値を記録しておくのでも、その他の方法で生成するのでも良い。
【００３８】
内挿手段１７０は、パイロットキャリア合成手段１３０から出力された合成重み付け係数ベクトルｗ b _kp(ｉ)およびｗ b _kp(ｉ＋４)を入力とし、キャリア合成手段１２０が合成信号ｙ_k(ｉ)を形成するために用いる合成重み付け係数ベクトルｗ b _k(ｉ)を補間によって生成し、キャリア合成手段１２０に出力するための手段である。
【００３９】
ＯＦＤＭ信号合成用受信装置１００を図２に示す構成のＯＦＤＭ信号合成用受信装置２００とすることにより、内挿手段１７０をシンボル間内挿手段２７７およびキャリア間内挿手段２７８で構成し、内挿手段１７０が行う補間処理をシンボル軸方向とキャリア軸方向に分けて行うことも可能である。以下に説明する。
【００４０】
初めに、シンボル軸方向についての補間方法について説明する。シンボル軸方向についての補間方法の具体例として、最新値保持法と直線内挿法とについて説明する。
ここでの最新値保持法は、以下の式（８）に示すように、上記の式（７）に示す合成重み付け係数ベクトルｗ b _kp(ｉ＋４)の値を、合成重み付け係数ベクトルｗ b _kp(ｉ＋１)、ｗ b _kp(ｉ＋２)、ｗ b _kp(ｉ＋３)に代入するものである。言うまでもなく上記の式（７）に示された合成重み付け係数ベクトルｗ b _kp(ｉ＋４)は、シンボル軸方向の更新処理によって得られたものである。また、シンボル番号（ｉ＋１）、 (ｉ＋２)、および (ｉ＋３)は、更新前のシンボル番号ｉと更新後のシンボル番号ｉ＋４の間の点である。
【数７】

【００４１】
次に、直線内挿法は、はじめに以下の式（９）に従ってΔｗ b _kp(ｉ)を生成し、シンボル番号ｉとｉ＋４との間のシンボル番号のＳＰ信号についての合成重み付け係数ベクトルｗ b _kp(ｉ＋１)、ｗ b _kp(ｉ＋２)、ｗ b _kp(ｉ＋３)の値を式（１０）〜（１２）に従って、算出するものである。
【数８】

【００４２】
このシンボル軸方向の内挿により、受信した時にキャリア軸方向に１２キャリア周期でＳＰ信号が挿入されていたのが３キャリア周期でＳＰ信号が挿入されたのと同等になる。すなわち、補完後のＳＰキャリア番号は、上記の式（１）の替わりに以下の式（１３）によって表されることになる。
【数９】

ここで、ｐは、上記の式（１）と同様に非負整数である。
【００４３】
シンボル間内挿手段２７７は、パイロットキャリア合成手段１３０から出力された合成重み付け係数ベクトルｗ b _kp(ｉ)およびｗ b _kp(ｉ＋４)を入力とし、シンボル番号ｉとｉ＋４との間のシンボル番号の合成重み付け係数ベクトルｗ b _kp(ｉ＋１)、ｗ b _kp(ｉ＋２)、ｗ b _kp(ｉ＋３)の値を上記の式（８）、または上記の式（９）〜（１３）に示す値として生成するのでもよい。補間によって得られた合成重み付け係数ベクトルｗ b _kp(ｉ＋１)、ｗ b _kp(ｉ＋２)、ｗ b _kp(ｉ＋３)の情報は、キャリア間内挿手段２７８に出力される。なお、シンボル軸方向の補間方法については、上記の方法以外の方法でも良い。
【００４４】
次に、キャリア軸方向についての補間方法の具体例として、直線内挿法、ＦＦＴ補間法、および畳み込み補間法の３つの方法について説明する。
まず、直線内挿法について説明する。
シンボル軸方向の内挿によりＳＰ信号はキャリア軸方向に３キャリア周期に挿入される。直線内挿法は、はじめに以下の式（１４）に従ってδｗ b _kp(ｉ)を生成し、キャリア番号ｋ_pとｋ_p＋３との間のキャリア番号のＳＰ信号についての合成重み付け係数ベクトルｗ b _kp+1(ｉ)、ｗ b _kp+2(ｉ)の値を式（１４）〜（１６）に従って、算出するものである。
【数１０】

【００４５】
このキャリア軸方向の内挿により全てのＳＰ信号についての合成重み付け係数ベクトルｗ b _k(ｉ)が求められる。しかしながら、直線内挿法は１次の近似法であり、近似の精度が良好でない場合があるため、直線内挿法に替わる次のＦＦＴ補間法について以下に説明する。
【００４６】
合成重み付け係数ベクトルｗ b _k(ｉ)は、上記の式（５）に示すように、ブランチ軸方向のＬ個の要素で構成されていた。ここでは、キャリア軸方向の合成重み付け係数ベクトルｗ c _l(ｉ)を以下の式（１７）によって定義する。
【数１１】

【００４７】
ここで、上記キャリア軸方向の合成重み付け係数ベクトルｗ c _l(ｉ)の各構成要素のうち、上記の式（１３）に示すＳＰキャリア番号以外のキャリア番号によって特定される要素に０を詰める。すなわち、以下の式（１８）に示すＷ_l,k(ｉ)を用いて以下の式（１９）に示すような構成の係数ベクトルＷ c _l(ｉ)を定義する。なお、このようにすることを零内挿という。
【数１２】

【００４８】
次に、係数ベクトルＷ c _l(ｉ)に対して逆高速フーリエ変換（以下、ＩＦＦＴ（Inverse Fast Fourier Transform）という。）処理を施す。その際、ＩＦＦＴ処理対象のポイント数を N = 2ⁿ とする。ここで、νは自然数であり、Ｎ＞Ｋの条件が成立するものとする。ＩＦＦＴ処理後に得られる任意のインパルスに付した番号であるインパルス番号（整数）をｎとする。ここで、ｎの範囲を以下の式（２０）に示す範囲とする。
【数１３】

【００４９】
ここで、係数ベクトルＷ c _l(ｉ)をＮポイント数の係数ベクトルとするために、係数ベクトルＷ c _l(ｉ)の左端に（Ｎ−Ｋ−１）／２個の零からなる零ベクトルを挿入し、係数ベクトルＷ c _l(ｉ)の右端に｛（Ｎ−Ｋ−１）／２｝＋１個の零からなる零ベクトルを挿入し、以下の式（２１）に示す係数ベクトルを生成する。
【数１４】

ここで、０は以下の式（２２）で定義される０ベクトルである。
【数１５】

【００５０】
上記の式（２１）に示すように、係数ベクトルＷ c _l(ｉ)の端、すなわちＯＦＤＭ信号帯域両端に０ベクトルを詰めるのではなく、０ベクトルの替わりに帯域の端のパイロット信号にかけられる重みを詰めて外挿することで、合成重み付け係数ベクトルのＯＦＤＭ信号帯域端の誤差を低減することができる。外挿の様子を図９に示す。すなわち、図９には、以下の式（２３）〜（２５）に示すように、ＯＦＤＭ信号帯域の左端に（Ｎ−Ｋ−１）／２個の要素からなるベクトルＡを挿入し、かつ、ＯＦＤＭ信号帯域の右端に｛（Ｎ−Ｋ−１）／２｝＋１個の要素からなるベクトルＢを挿入した様子を示す。なお、以下の式（２３）に示すベクトルを改めて係数ベクトルＷ c _l(ｉ)と呼ぶことにする。
【００５１】
【数１６】

【００５２】
このようして形成した上記の式（２３）に示す係数ベクトルＷ c _l(ｉ)に対しブランチ毎にＩＦＦＴ処理を施す。このＩＦＦＴ処理によって得られたベクトルをインパルス応答ベクトルｈ _lという。換言すれば、係数ベクトルＷ c _l(ｉ)を以下の式（２６）で定義する変換をしてインパルス応答ベクトルｈ _lを生成することである。
【数１７】

【００５３】
ここで、インパルス応答ベクトルｈ _lを以下の式（２７）に示すように定義する。
【数１８】

インパルス応答ベクトルｈ _lの一例を図１０に引用記号１００１が指す曲線として模式的に示す。
【００５４】
いわゆるエイリアシングを除去するために、以下の処理を行う。その処理の説明に入る前に、フィルタ係数ベクトルｆを以下の式（２８）で定義しておく。
【数１９】

フィルタ係数ベクトルｆの一例を図１０に引用記号１００２が指す曲線として模式的に示す。
【００５５】
エイリアシングを除去するため、以下の式（２９）に示すようにインパルス応答ベクトルｈ _lとフィルタ係数ベクトルｆとを要素毎に積をとり、ＦＦＴ処理を施す（以下の式における記号「●」は、要素毎の積を表すものとする。）。
【数２０】

【００５６】
これによって、補間された合成重み付け係数ベクトルＷ c _l(ｉ)が生成される。そして、このようにして合成重み付け係数ベクトルＷ c _l(ｉ)を全てのブランチについて生成することで、生成された全てのブランチの合成重み付け係数ベクトルＷ c _l(ｉ)のデータからキャリア軸方向に補間された合成重み付け係数ベクトルＷ b _k(ｉ)を生成することができる。
【００５７】
次に、畳み込み補間法について説明する。
はじめに、フィルタ係数ベクトルｆのインパルス応答ベクトルｇ（以下、ｆのインパルス応答ベクトルｇという。）を以下の式（３０）で定義する。
【数２１】

そして、ｆのインパルス応答ベクトルｇを以下の式（３１）に示すベクトルとして表すこととする。
【数２２】

【００５８】
フィルタ係数ベクトルｆの特性を矩形ではなく両端で緩やかに０に向かって減衰するような特性とすると、ｆのインパルス応答ベクトルｇの特性も両端で概ね零が連続するような特性となり、ｆのインパルス応答ベクトルｇの要素数を以下の式（３２）に示すようにＮ個からＭ個に低減できる。
【数２３】

【００５９】
このように定義されたｆのインパルス応答ベクトルｇと上記の式（１９）で定義された係数ベクトルＷ c _l(ｉ)を用いて以下の式（３３）に示すように畳み込み演算をすることによって、任意のキャリア番号ｋおよび任意のブランチ番号ｌで特定される合成重み付け係数Ｗ_l,k(ｉ)を生成することができる。
【数２４】

【００６０】
このようにして生成した合成重み付け係数Ｗ_l,k(ｉ)を上記の式（５）のｗ_l,k(ｉ)に代入することによって、キャリア軸方向に補間された合成重み付け係数ベクトルｗ b _k(ｉ)を生成することができる。
なお、キャリア軸方向の補間方法については、上記の方法以外の方法でも良い。
【００６１】
キャリア間内挿手段２７８は、シンボル間内挿手段２７７から出力されたシンボル軸方向について補間された合成重み付け係数ベクトルを入力とし、上記の直線内挿法、ＦＦＴ補間法、畳み込み補間法その他の補間方法によってキャリア軸方向に補間された合成重み付け係数ベクトルｗ b _k(ｉ)を生成し、キャリア合成手段１２０に出力するための手段である。
【００６２】
以上説明したように、本発明の第１の実施の形態に係るＯＦＤＭ信号合成用受信装置１００は、ＳＰ信号に基づいてＯＦＤＭ信号のサブキャリア毎にアレーアンテナからの信号を最適に合成することを可能にすることによって、マルチパス成分の混入による周波数選択性ひずみが発生する状況下でも希望波の信号を良好に再生できる。
【００６３】
図３は、本発明の第２の実施の形態に係るＯＦＤＭ信号合成用受信装置３００の概略の構成を示すブロック図である。第２の実施の形態のＯＦＤＭ信号合成用受信装置３００は、ＲＬＳ（Recursive Least Squares）アルゴリズムを適用して合成重み付け係数を生成するものである。以下に、図面を参照して説明する。
【００６４】
ＯＦＤＭ信号合成用受信装置１００は、アンテナブランチ０〜（Ｌ−１）、高速フーリエ変換（以下、ＦＦＴ（Fast Fourier Transform）という。）手段１０１（０）〜１０１（Ｌ−１）、パイロットキャリア抽出手段１１０、キャリア合成手段１２０、パイロットキャリア合成手段３３０、係数更新手段３４０、差分演算手段１５０、パイロット信号発生手段１６０、シンボル間内挿手段２７７およびキャリア間内挿手段２７８によって構成される。
【００６５】
なお、ＯＦＤＭ信号合成用受信装置３００を構成する手段のうち、本発明の第１の実施の形態に係るＯＦＤＭ信号合成用受信装置１００における構成手段と同様の処理を行うものには同一の引用番号を付し、その説明を省略する。
また、ＯＦＤＭ信号ベクトルｕ _k(ｉ)、合成重み付け係数ベクトルｗ b _k(ｉ)、ＳＰ信号のキャリア番号ｋｐ等の、本発明の第１の実施の形態において定義した記号や用語等と同一の記号等は、本発明の第２の実施の形態においても同一の定義とする。
【００６６】
本発明の第２の実施の形態において、ＯＦＤＭスペクトルの合成信号ｙ_k(ｉ)を以下の式（３４）に示すように定義する。
【数２５】

合成信号ｙ_k(ｉ)の生成にシンボル番号が（ｉ−４）の合成重み付け係数ベクトルが係るのは、シンボル番号ｉの直前のＳＰ信号が存在するシンボル番号は（ｉ−４）だからである。
【００６７】
以下、この合成重み付け係数ベクトルｗ b _k(ｉ)を生成する方法について説明する。
はじめに、ＳＰ信号の合成重み付け係数ベクトルｗ b _kp(ｉ)の生成について説明する。ＳＰ信号の合成重み付け係数ベクトルは、ＲＬＳアルゴリズムを適用して以下の式（３５）に示すように生成される。
【数２６】

【００６８】
ここで、ｋ _kp(ｉ)は以下の式（３６）に示すように定義されるゲインベクトルであり、ξ_kp（ｉ）は以下の式（３７）に示すように定義される事前推定誤差である。
【数２７】

【００６９】
ただし、λ^-1は忘却係数であり、Ｐ _kp(ｉ)は以下の式（３８）に示すように定義されるＬ×Ｌの相関逆行列であり、ｄ_kp(ｉ)は本発明の第１の実施の形態において定義した基準合成値である。
【数２８】

【００７０】
以下、本発明の第１の実施の形態と同様に、特に断らない限り、ブランチ番号ｌ等の添字、シンボル番号ｉ等の引数が付された、ＯＦＤＭ信号ベクトル等の信号または情報等を入出力すると記載されている場合は、それを特定するために必要な添字、引数等はその信号等に伴って入出力されるものとし、その記載を省略する。
【００７１】
パイロットキャリア合成手段３３０は、パイロットキャリア抽出手段１１０から出力される各ＯＦＤＭ信号ベクトルｕ _k(ｉ)と後述する係数更新手段３４０から出力される合成重み付け係数ベクトルｗ b _kp(ｉ−４)とを入力とし、上記の式（３４）に基づいて合成信号ｙ_kp(ｉ)を生成して差分演算手段１５０に出力すると共に、入力された合成重み付け係数ベクトルｗ b _kp(ｉ−４)を内挿手段１７０に出力し、シンボル番号ｉの情報およびキャリア番号ｋ_pの情報をパイロット信号発生手段１６０に出力するための手段である。
【００７２】
係数更新手段３４０は、パイロットキャリア抽出手段１１０から出力されるＯＦＤＭ信号ベクトルｕ _kp(ｉ)および差分演算手段１５０から出力される事前推定誤差ξ_kp（ｉ）を入力とし、上記の式（３５）〜（３８）に基づいて合成重み付け係数ベクトルｗ b _kp(ｉ−４)の更新を行い、更新によって得られた合成重み付け係数ベクトルｗ b _kp(ｉ)をパイロットキャリア合成手段３３０に出力するための手段である。ここで、係数更新手段３４０は、シンボル番号ｉ−４での更新処理が完了した場合、更新処理によって得られた合成重み付け係数ベクトルｗ b _kp(ｉ)を次の処理のために記憶しておくのでも良い。なお、最初に行う更新処理に用いる合成重み付け係数ベクトルの値としては、所定の値を記録しておくのでも、その他の方法で生成するのでも良い。
【００７３】
シンボル間内挿手段２７７は、パイロットキャリア合成手段３３０から出力された合成重み付け係数ベクトルｗ b _kp(ｉ−４)およびｗ b _kp(ｉ)を入力とし、シンボル番号ｉ−４とｉとの間のシンボル番号のＳＰ信号についての合成重み付け係数ベクトルｗ b _kp(ｉ−３)、ｗ b _kp(ｉ−２)、ｗ b _kp(ｉ−１)の値を以下の式（７）、または以下の式（８）〜（１２）に示す値として生成するのでもよい。補間によって得られた合成重み付け係数ベクトルｗ b _kp(ｉ−１)、ｗ b _kp(ｉ−２)、ｗ b _kp(ｉ−３)の情報は、キャリア間内挿手段２７８に出力される。なお、シンボル軸方向の補間方法については、上記の方法以外の方法でも良い。
【００７４】
本発明の第２の実施の形態でも本発明の第１の実施の形態と同様に、シンボル軸方向についての補間方法として最新値保持法と直線内挿法を用いることができる。また、他の方法であっても良いことは当然のことである。
本発明の第２の実施の形態では、最新値保持法は、以下の式（３９）に示すように、上記の式（３８）に示す合成重み付け係数ベクトルｗ b _kp(ｉ)の値を、合成重み付け係数ベクトルｗ b _kp(ｉ−１)、ｗ b _kp(ｉ−２)、ｗ b _kp(ｉ−３)に代入するものである。言うまでもなく上記の式（３８）に示された合成重み付け係数ベクトルｗ b _kp(ｉ)は、シンボル軸方向の更新処理によって得られたものである。また、シンボル番号（ｉ−１）、 (ｉ−２)、および (ｉ−３)は、更新前のシンボル番号ｉ−４と更新後のシンボル番号ｉの間の点である。
【数２９】

【００７５】
次に、直線内挿法は、はじめに以下の式（４０）に従ってΔｗ b _kp(ｉ)を生成し、シンボル番号ｉとｉ−４との間のシンボル番号のＳＰ信号についての合成重み付け係数ベクトルｗ b _kp(ｉ−１)、ｗ b _kp(ｉ−２)、ｗ b _kp(ｉ−３)の値を式（４１）〜（４３）に従って、算出するものである。
【数３０】

【００７６】
このシンボル軸方向の内挿により、受信した時にキャリア軸方向に１２キャリア周期でＳＰ信号が挿入されていたのが３キャリア周期でＳＰ信号が挿入されたのと同等になる。すなわち、補完後のＳＰキャリア番号は、上記の式（１）の替わりに以下の式（４４）によって表されることになる。
【数３１】

ここで、ｐは、上記の式（１）と同様に非負整数である。
【００７７】
シンボル間内挿手段２７７は、パイロットキャリア合成手段３３０から出力された合成重み付け係数ベクトルｗ b _kp(ｉ)およびｗ b _kp(ｉ−４)を入力とし、シンボル番号ｉとｉ−４との間のシンボル番号の合成重み付け係数ベクトルｗ b _kp(ｉ−１)、ｗ b _kp(ｉ−２)、ｗ b _kp(ｉ−３)の値を上記の式（３９）、または上記の式（４０）〜（４４）に示す値として生成するのでもよい。補間によって得られた合成重み付け係数ベクトルｗ b _kp(ｉ−１)、ｗ b _kp(ｉ−２)、ｗ b _kp(ｉ−３)の情報は、キャリア間内挿手段２７８に出力される。なお、シンボル軸方向の補間方法については、上記の方法以外の方法でも良い。
【００７８】
次に、キャリア軸方向についての補間方法の具体例として、直線内挿法、ＦＦＴ補間法、および畳み込み補間法の３つの方法について説明する。
まず、直線内挿法について説明する。
シンボル軸方向の内挿によりＳＰ信号はキャリア軸方向に３キャリア周期に挿入される。直線内挿法は、はじめに以下の式（４５）に従ってδｗ b _kp(ｉ−４)を生成し、キャリア番号ｋ_pとｋ_p＋３との間のキャリア番号のＳＰ信号についての合成重み付け係数ベクトルｗ b _kp+1(ｉ−４)、ｗ b _kp+2(ｉ−４)の値を式（４５）〜（４７）に従って、算出するものである。
【数３２】

【００７９】
このキャリア軸方向の内挿により全てのＳＰ信号についての合成重み付け係数ベクトルｗ b _k(ｉ)が求められる。
次に、ＦＦＴ補間法について説明する。合成重み付け係数ベクトルｗ b _k(ｉ−４)は、上記の式（５）に示すように、ブランチ軸方向のＬ個の要素で構成されていた。ここでは、キャリア軸方向の合成重み付け係数ベクトルｗ c _l(ｉ−４)を以下の式（４８）によって定義する。
【数３３】

【００８０】
ここで、上記キャリア軸方向の合成重み付け係数ベクトルｗ c _l(ｉ−４)の各構成要素のうち、上記の式（４４）に示すＳＰキャリア番号以外のキャリア番号によって特定される要素に０を詰める。すなわち、以下の式（４９）に示すＷ_l,k(ｉ−４)を用いて以下の式（５０）に示すような構成の係数ベクトルＷ c _l(ｉ−４)を定義する。なお、このようにすることを零内挿という。
【数３４】

【００８１】
次に、係数ベクトルＷ c _l(ｉ−４)に対して逆高速フーリエ変換（以下、ＩＦＦＴ（Inverse Fast Fourier Transform）という。）処理を施す。その際、ＩＦＦＴ処理対象のポイント数を N = 2ⁿ とする。ここで、νは自然数であり、Ｎ＞Ｋの条件が成立するものとする。ＩＦＦＴ処理後に得られる任意のインパルスに付した番号であるインパルス番号（整数）をｎとする。ここで、ｎの範囲を以下の式（５１）に示す範囲とする。
【数３５】

【００８２】
ここで、係数ベクトルＷ c _l(ｉ−４)をＮポイント数の係数ベクトルとするために、係数ベクトルＷ c _l(ｉ−４)の左端に（Ｎ−Ｋ−１）／２個の零からなる零ベクトルを挿入し、係数ベクトルＷ c _l(ｉ−４)の右端に｛（Ｎ−Ｋ−１）／２｝＋１個の零からなる零ベクトルを挿入し、以下の式（５２）に示す係数ベクトルを生成する。
【数３６】

ここで、０は以下の式（５３）で定義される０ベクトルである。
【数３７】

【００８３】
上記の式（５２）に示すように、係数ベクトルＷ c _l(ｉ−４)の端、すなわちＯＦＤＭ信号帯域両端に０ベクトルを詰めるのではなく、０ベクトルの替わりに帯域の端のパイロット信号にかけられる重みを詰めて外挿することで、合成重み付け係数ベクトルのＯＦＤＭ信号帯域端の誤差を低減することができる。外挿の様子を図９に示す。すなわち、図９には、以下の式（５４）〜（５６）に示すように、ＯＦＤＭ信号帯域の左端に（Ｎ−Ｋ−１）／２個の要素からなるベクトルＡを挿入し、かつ、ＯＦＤＭ信号帯域の右端に｛（Ｎ−Ｋ−１）／２｝＋１個の要素からなるベクトルＢを挿入した様子を示す。なお、以下の式（５４）に示すベクトルを改めて係数ベクトルＷ c _l(ｉ−４)と呼ぶことにする。
【００８４】
【数３８】

【００８５】
このようして形成した上記の式（５４）に示す係数ベクトルＷ c _l(ｉ−４)に対しブランチ毎にＩＦＦＴ処理を施す。このＩＦＦＴ処理によって得られたベクトルをインパルス応答ベクトルｈ _l(ｉ−４)という。換言すれば、係数ベクトルＷ c _l(ｉ−４)を以下の式（５７）で定義する変換をしてインパルス応答ベクトルｈ _l(ｉ−４)を生成することである。
【数３９】

【００８６】
ここで、インパルス応答ベクトルｈ _l(ｉ−４)を以下の式（５８）に示すように定義する。
【数４０】

インパルス応答ベクトルｈ _l(ｉ−４)の一例を図１０に引用記号１００１が指す曲線として模式的に示す。
【００８７】
いわゆるエイリアシングを除去するために、以下の処理を行う。その処理の説明に入る前に、フィルタ係数ベクトルｆを以下の式（５９）で定義しておく。
【数４１】

フィルタ係数ベクトルｆの一例を図１０に引用記号１００２が指す曲線として模式的に示す。
【００８８】
エイリアシングを除去するため、以下の式（６０）に示すようにインパルス応答ベクトルｈ _l(ｉ−４)とフィルタ係数ベクトルｆとを要素毎に積をとり、ＦＦＴ処理を施す（以下の式における記号「●」は、要素毎の積を表すものとする。）。
【数４２】

【００８９】
これによって、補間された合成重み付け係数ベクトルＷ c _l(ｉ)が生成される。そして、このようにして合成重み付け係数ベクトルＷ c _l(ｉ)を全てのブランチについて生成することで、生成された全てのブランチの合成重み付け係数ベクトルＷ c _l(ｉ)のデータからキャリア軸方向に補間された合成重み付け係数ベクトルＷ b _k(ｉ)を生成することができる。
【００９０】
次に、畳み込み補間法について説明する。
はじめに、フィルタ係数ベクトルｆのインパルス応答ベクトルｇ（以下、ｆのインパルス応答ベクトルｇという。）を以下の式（６１）で定義する。
【数４３】

そして、ｆのインパルス応答ベクトルｇを以下の式（６２）に示すベクトルとして表すこととする。
【数４４】

【００９１】
フィルタ係数ベクトルｆの特性を矩形ではなく両端で緩やかに０に向かって減衰するような特性とすると、ｆのインパルス応答ベクトルｇの特性も両端で概ね零が連続するような特性となり、ｆのインパルス応答ベクトルｇの要素数を以下の式（６３）に示すようにＮ個からＭ個に低減できる。
【数４５】

【００９２】
このように定義されたｆのインパルス応答ベクトルｇと上記の式（５０）で定義された係数ベクトルＷ c _l(ｉ−４)を用いて以下の式（６４）に示すように畳み込み演算をすることによって、任意のキャリア番号ｋおよび任意のブランチ番号ｌで特定される合成重み付け係数Ｗ_l,k(ｉ−４)を生成することができる。
【数４６】

【００９３】
このようにして生成した合成重み付け係数Ｗ_l,k(ｉ−４)を上記の式（５）のｗ_l,k(ｉ−４)に代入することによって、キャリア軸方向に補間された合成重み付け係数ベクトルｗ b _k(ｉ−４)を生成することができる。
なお、キャリア軸方向の補間方法については、上記の方法以外の方法でも良い。
【００９４】
キャリア間内挿手段２７８は、シンボル間内挿手段２７７から出力されたシンボル軸方向について補間された合成重み付け係数ベクトルを入力とし、上記の直線内挿法、ＦＦＴ補間法、畳み込み補間法その他の補間方法によってキャリア軸方向に補間された合成重み付け係数ベクトルｗ b _k(ｉ−４)を生成し、キャリア合成手段１２０に出力するための手段である。
【００９５】
以上説明したように、本発明の第２の実施の形態に係るＯＦＤＭ信号合成用受信装置３００は、ＳＰ信号に基づいてＯＦＤＭ信号のサブキャリア毎にアレーアンテナからの信号を最適に合成することを可能にすることによって、マルチパス成分の混入による周波数選択性ひずみが発生する状況下でも希望波の信号を良好に再生できる。
【００９６】
図４は、本発明の第３の実施の形態に係るＯＦＤＭ信号合成用受信装置４００の概略の構成を示すブロック図である。本発明の第３の実施の形態のＯＦＤＭ信号合成用受信装置４００は、時間領域において各ブランチからのＯＦＤＭ信号を合成して出力するものである。以下に、図面を参照して説明する。
【００９７】
ＯＦＤＭ信号合成用受信装置４００は、アンテナブランチ（以下、単にブランチという。）０〜（Ｌ−１）、高速フーリエ変換（以下、ＦＦＴ（Fast Fourier Transform）という。）手段１０１（０）〜１０１（Ｌ−１）、パイロットキャリア抽出手段１１０、時間領域合成手段４２０、パイロットキャリア合成手段１３０、係数更新手段４４０、差分演算手段１５０、パイロット信号発生手段１６０、内挿手段４７０、および逆高速フーリエ変換（以下、ＩＦＦＴ（Inverse Fast Fourier Transform）という。）手段４８１（０）〜４８１（Ｌ−１）によって構成される。
【００９８】
なお、ＯＦＤＭ信号合成用受信装置４００を構成する手段のうち、本発明の第１の実施の形態に係るＯＦＤＭ信号合成用受信装置１００における構成手段と同様の処理を行うものには同一の引用番号を付し、その説明を省略する。
また、ＯＦＤＭ信号ベクトルｕ _k(ｉ)、合成重み付け係数ベクトルｗ b _k(ｉ)、ＳＰ信号のキャリア番号ｋ_p等の、本発明の第１の実施の形態において定義した記号や用語等と同一の記号等は、本発明の第２の実施の形態においても同一の定義とする。
【００９９】
ＩＦＦＴ手段４８１（０）は、後述する内挿手段４７０から出力される合成重み付け係数ベクトルｗ b _k(ｉ)等を入力として、入力された合成重み付け係数ベクトルｗ b _k(ｉ)等に対してＩＦＦＴ処理を施し、ＩＦＦＴ処理によって得られたベクトルから時間領域でのフィルタ処理に必要なフィルタ係数（以下、トランスバーサルフィルタ係数という。）を抽出し、時間領域合成手段４２０に出力するための手段である。
【０１００】
なお、ＩＦＦＴ手段４８１（０）に入力される合成重み付け係数ベクトルｗ b _k(ｉ)等、およびＩＦＦＴ手段４８１（０）から出力されるトランスバーサルフィルタ係数は、ブランチ０から出力されたＯＦＤＭ信号のフィルタ処理のためのものである。ＩＦＦＴ手段４８１（１）〜４８１（Ｌ−１）は、ＩＦＦＴ手段４８１（０）と同様の処理を行う手段であり、それぞれ、ブランチ１〜（Ｌ−１）から出力されたＯＦＤＭ信号のフィルタ処理のためのものである。
【０１０１】
時間領域合成手段４２０は、ブランチ０〜（Ｌ−１）から出力されたＯＦＤＭ信号とＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）から出力されたトランスバーサルフィルタ係数とを入力とし、入力されたトランスバーサルフィルタ係数に応じて各ブランチからのＯＦＤＭ信号に時間領域でフィルタ処理を施し、フィルタ処理後の各ブランチの信号を合成して出力するための手段である。
【０１０２】
具体的には、入力されたＯＦＤＭ信号Ｕ_l(nt)に対して以下の式（６５）に示す変換を施すことにより、ブランチ毎、キャリア毎（時間領域での処理であるため、式（６５）には、明確に表示されない。）に時間領域でフィルタ処理され、全ブランチについて合成された合成信号Ｙ(nt)を生成する。
【数４７】

ただし、ntは任意の時刻の番号（ここで、ｎは整数とする。）、ｌは任意のブランチの番号、ｈ_l,m(ｉ)（０≦ｍ≦Ｍ−１であり、Ｍはトランスバーサルフィルタ５２１（０）〜５２１（Ｌ−１）のタップの数である。）はトランスバーサルフィルタ係数、そしてｉはシンボル番号である。
【０１０３】
なお、ＯＦＤＭ信号合成用受信装置４００を、図５に示すＯＦＤＭ信号合成用受信装置５００のように構成し、時間領域合成手段４２０をトランスバーサルフィルタ５２１（０）〜５２１（Ｌ−１）と加算手段５２２とによって構成するのでも良い。
【０１０４】
その場合、トランスバーサルフィルタ５２１（０）〜５２１（Ｌ−１）は、図５に示すように、ブランチ０〜（Ｌ−１）から出力されたＯＦＤＭ信号および各ＩＦＦＴ手段４８１（０）〜（Ｌ−１）から出力されたトランスバーサルフィルタ係数を入力とし、入力されたＯＦＤＭ信号に対してブランチ毎に時間領域内でフィルタ処理を施し、加算手段５２２に出力する。
加算手段５２２は、トランスバーサルフィルタ５２１（０）〜５２１（Ｌ−１）から出力されたフィルタ処理後の各結果を加算して外部に出力する。
【０１０５】
以下に、ＯＦＤＭ信号合成用受信装置５００を図６に示すＯＦＤＭ信号合成用受信装置６００のように構成し、内挿手段４７０をシンボル間内挿手段６７１とキャリア間内挿手段６７２とによって構成するのでも良い。この場合、シンボル間内挿手段６７１、キャリア間内挿手段６７２、およびＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）によって、以下に示すような処理がなされる。
【０１０６】
シンボル間内挿手段６７１は、パイロットキャリア合成手段１３０から出力された合成重み付け係数ベクトルｗ b _kp(ｉ)を入力とし、上記、本発明の第１の実施の形態において説明したシンボル間補間処理（例えば、式（８）または、式（９）−（1２））を施し、キャリア間内挿手段６７２に出力する。
【０１０７】
キャリア間内挿手段６７２は、上記のシンボル間補間処理によって得られた合成重み付け係数ベクトルｗ b _kp(ｉ)を入力とし、キャリア軸方向に所定の補間処理を施して合成重み付け係数ベクトルｗ b _k (ｉ)を生成し、ＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）に出力する。ここで、「所定の補間処理」として、例えば直線補間処理（式（１４）−（１６））、本発明の第１の実施の形態において説明した畳み込み補間法（式（３０）−（３３））、その他の方法があるが、その方法の種別を問わない。
【０１０８】
以下に、ＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）によって行われる処理について具体的に説明する。
ここで、キャリア間内挿手段６７２から出力された補間後の合成重み付け係数ベクトルｗ b _k (ｉ)を以下の式（６６）に示す。
【数４８】

【０１０９】
また、ＩＦＦＴ処理の対象となるデータ数（ポイント数ともいう。）をＮ＝２^νとする。ただし、νは自然数であり、かつN＞K の条件が成立するものとする。ＩＦＦＴ処理によって得られる任意のインパルスの番号をｎとする。ｎの範囲を以下の式（６７）に示す範囲とする。
【数４９】

【０１１０】
ここで、係数ベクトルＷ c _l(ｉ)をＮポイント数の係数ベクトルとするために、係数ベクトルＷ c _l(ｉ)の左端に（Ｎ−Ｋ−１）／２個の零からなる零ベクトルを挿入し、係数ベクトルＷ c _l(ｉ)の右端に｛（Ｎ−Ｋ−１）／２｝＋１個の零からなる零ベクトルを挿入し、以下の式（６８）に示す係数ベクトル（これを新たに係数ベクトルＷ c _l(ｉ)とする。）を生成する。
【数５０】

ここで、０は以下の式（６９）で定義される０ベクトルである。
【数５１】

【０１１１】
次に、係数ベクトルＷ c _l(ｉ)に対して以下の式（７０）に示すようにＩＦＦＴ処理を施し、トランスバーサルフィルタの係数ベクトルを生成する。
【数５２】

上記で得られたトランスバーサルフィルタの係数ベクトルを以下の式（７１）のようにおく。
【数５３】

【０１１２】
上記の式（７１）のＭ個の連続した要素に注目して式（７１）を以下の式（７２）のように書き直すこともできる。
【数５４】

Ｎ個のトランスバーサルフィルタ係数からトランスバーサルフィルタのタップ長分（このタップ長をＭとする。）だけベクトルｈ _lの中の連続した要素を抜き出して、以下に示す式（７３）Ｍ次元ベクトルとし、トランスバーサルフィルタ係数ベクトルにする。
【数５５】

【０１１３】
以上のように処理することでタップ数Ｍのトランスバーサルフィルタ係数ベクトルｈ _lが生成される。このようにしてＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）は、それぞれ、ブランチランチ毎のトランスバーサルフィルタ係数ベクトルｈ _lを生成し、対応するトランスバーサルフィルタ５２１（０）〜５２１（Ｌ−１）に出力する。
【０１１４】
また、ＯＦＤＭ信号合成用受信装置５００を図７に示すＯＦＤＭ信号合成用受信装置７００のように構成し、内挿手段４７０をシンボル間内挿手段６７１と零挿入手段７７２とによって構成するのでも良い。この場合に、零挿入手段７７２およびＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）によって行われる処理について、以下に、具体的に説明する。
【０１１５】
零挿入手段７７２は、シンボル間補間処理（例えば、式（８）または、式（９）−（1２））が施された合成重み付け係数ベクトルｗ b _kp(ｉ)を入力とし、入力された合成重み付け係数ベクトルｗ b _kp(ｉ)に対して、本発明の第１の実施の形態において説明した零挿入処理（式（１７）−（２５））を施し、係数ベクトルＷ c _l(ｉ)（式（２３）−（２５））を生成し、ＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）に出力する。
【０１１６】
ＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）は、上記零挿入処理によって得られた係数ベクトルＷ c _l(ｉ)に対してＩＦＦＴ処理（式（２６））を施し、インパルス応答ベクトル（式（２７））を生成する。以下に、式（２７）を再度記載する。
【数５６】

【０１１７】
上記の式（２７）のＭ個の連続した要素に注目して式（２７）を以下の式（７４）のように書き直すこともできる。
【数５７】

【０１１８】
Ｎ個のインパルス応答ベクトルからトランスバーサルフィルタのタップ長分（このタップ長をＭとする。）だけベクトルｈ _lの中の連続した要素を抜き出して、以下に示す式（７５）Ｍ次元ベクトルとし、トランスバーサルフィルタ係数ベクトルにする。
【数５８】

これにより、Ｍタップのトランスバーサルフィルタの係数ｈ _lが求まった。
【０１１９】
以上のように処理することでタップ数Ｍのトランスバーサルフィルタ係数ベクトルｈ _lが生成される。このようにしてＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）は、それぞれ、ブランチランチ毎のトランスバーサルフィルタ係数ベクトルｈ _l（式（７５））を生成し、対応するトランスバーサルフィルタ５２１（０）〜５２１（Ｌ−１）に出力する。なお、キャリア間内挿手段６７２およびＩＦＦＴ手段４８１（０）〜４８１（Ｌ−１）においてなされる処理は、本発明の第１の実施の形態において説明したＦＦＴ補間法に対応するものである。
【０１２０】
本発明の第３の実施の形態における説明は、ＬＭＳアルゴリズムを適用する場合について行ったが、ＲＬＳアルゴリズムを適用する場合でも、ＬＭＳアルゴリズムを適用して得られた上記の結果（例えば、式（７３）、（７５）に示すトランスバーサルフィルタ係数）の引数が(ｉ)から(ｉ−４)に変わるだけであり、その他の部分は実質的に変わらない。
【０１２１】
以上説明したように、本発明の第３の実施の形態に係るＯＦＤＭ信号合成用受信装置４００、５００、６００、７００は、ＳＰ信号に基づいてＯＦＤＭ信号のサブキャリア毎にアレーアンテナからの信号を最適に合成することを可能にすることによって、マルチパス成分の混入による周波数選択性ひずみが発生する状況下でも希望波の信号を良好に再生できる。
【０１２２】
【発明の効果】
以上説明したように、本発明は、マルチパス成分の混入による周波数選択性ひずみが発生する状況下でも希望波の信号を良好に再生できるＯＦＤＭ信号合成用受信装置を実現することができる。
【図面の簡単な説明】
【図１】本発明の第１の実施の形態に係るＯＦＤＭ信号合成用受信装置の構成の一例を示すブロック図である。
【図２】本発明の第１の実施の形態に係るＯＦＤＭ信号合成用受信装置の構成の一例を示すブロック図である。
【図３】本発明の第２の実施の形態に係るＯＦＤＭ信号合成用受信装置の構成の一例を示すブロック図である。
【図４】本発明の第３の実施の形態に係るＯＦＤＭ信号合成用受信装置の構成の一例を示すブロック図である。
【図５】本発明の第３の実施の形態に係るＯＦＤＭ信号合成用受信装置の構成の一例を示すブロック図である。
【図６】本発明の第３の実施の形態に係るＯＦＤＭ信号合成用受信装置の構成の一例を示すブロック図である。
【図７】本発明の第３の実施の形態に係るＯＦＤＭ信号合成用受信装置の構成の一例を示すブロック図である。
【図８】キャリア−シンボル空間における、ＯＦＤＭ信号を構成する信号の配列を示す図である。
【図９】本発明の合成重み付け係数の外挿方法について説明するための図である。
【図１０】本発明のＦＦＴを用いた補間方法について説明するための図である。
【符号の説明】
０〜（Ｌ−１） アンテナブランチ
１００、２００、３００、４００、５００、６００、７００ ＯＦＤＭ信号合成用受信装置
１０１（０）〜（Ｌ−１） 高速フーリエ変換（ＦＦＴ）手段
１１０ パイロットキャリア抽出手段
１２０ キャリア合成手段
１３０、３３０ パイロットキャリア合成手段
１４０、３４０、４４０ 係数更新手段
１５０ 差分演算手段
１６０ パイロット信号発生手段
１７０ 内挿手段
２７７ シンボル間内挿手段
２７８ キャリア間内挿手段
４２０ 時間領域合成手段
４７０ 内挿手段
４８１（０）〜（Ｌ−１） 逆高速フーリエ変換（ＦＦＴ）手段
５２１（０）〜（Ｌ−１） トランスバーサルフィルタ
５２２ 加算手段
６７１ シンボル間内挿手段
６７２ キャリア間内挿手段
７７２ 零挿入手段
１００１ インパルス応答ベクトル
１００２ フィルタ係数ベクトル[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an OFDM signal combining receiver for digital broadcasting and digital transmission that employs an OFDM (Orthogonal Frequency Division Multiplexing) system, and particularly occurs when receiving radio waves such as digital broadcasting and wireless LAN. The present invention relates to an OFDM signal combining receiver to which adaptive array antenna technology and diversity combining technology for removing fading and interference waves are applied.
[0002]
[Prior art]
Traditionally, adaptive array antenna technology has been described by John Litva, Tystadok-Yongro, “Digital Beamforming in Wireless Communication”, Arthouse Publishers, 1996 (“Digital Beam-forming in Wireless Communication”, John Litva, Titus Kwok-Yeung Lo, Artech House Publishers, 1996), Simon Hykin, Hiroshi Suzuki, “Adaptive Filter Theory”, Science and Technology Publishing, 2001, and other books. In the adaptive array antenna technique, a weight is assigned to each antenna in accordance with a reception situation, and a signal received by each antenna is synthesized (hereinafter referred to as beam forming).
[0003]
In these books, the LMS (Least Mean Square) algorithm and the RLS (Recursive least squares) algorithm were applied to digital transmission systems such as single carrier PSK (Phase Shift Keying) and multilevel QAM (Quadrature Amplitude Modulation). An adaptive array antenna and a transmission line equalizer are described in detail.
[0004]
On the other hand, as an adaptive array antenna technology for receiving an OFDM signal, Satoshi Hori, Nobuyoshi Kikuma, Naoki Inagaki, “MMSE Adaptive Array Using Guard Section in OFDM”, IEEJ Technical Report, AP2001-50 (2001). -07) (hereinafter referred to as Document 1), Toru Nishikawa, Yoshitaka Hara, Keisuke Hara, "Examination of Adaptive Array for OFDM in Mobile Communications", IEICE Technical Report, RCS-2000-232 (2001-03) ( Hereinafter referred to as Reference 2.), Takeo Fujii, Masao Nakagawa, “OFDM indoor multi-station simultaneous frequency simultaneous transmission system using smart antenna”, IEICE Technical Report, RCS2000-83 (2000-07) (hereinafter referred to as Reference) 3)).
[0005]
[Problems to be solved by the invention]
However, in the technique disclosed in the above-mentioned document 1, when frequency selective distortion occurs due to the multipath component of the desired wave mixed in the received wave, the reception situation of each subcarrier constituting the OFDM signal is different, Since the optimum weighting coefficients for beam forming for subcarriers are different, there is a problem in that suitable beamforming cannot be performed for all subcarriers, and a desired wave signal cannot be reproduced satisfactorily.
Further, the technique disclosed in the above-mentioned document 2 monitors virtual subcarriers that are not actually used for signal transmission, and determines a combined weighting coefficient (or weighting coefficient for beam forming) of each array antenna. And has the same problem as the technique disclosed in the above-mentioned document 1.
[0006]
On the other hand, the technique disclosed in Document 3 has a configuration in which an OFDM signal is fast Fourier transformed for each array antenna, and is synthesized by applying an RLS algorithm to each subcarrier after the fast Fourier transform. Yes. By applying such an algorithm, an optimum beam pattern is formed for each subcarrier, so that it is strong against the influence of frequency selective distortion due to the multipath of the propagation path. A training (pilot) signal (hereinafter referred to as a pilot signal) in the technique disclosed in Document 3 is based on pilot signals that are uniformly arranged at a constant interval in both the carrier direction and the symbol direction. Thus, the composite weighting coefficient of each array antenna is calculated for each subcarrier.
[0007]
However, ISDB-T (Integrated Services Digital Broadcasting-Terrestrial) and DVB-T (Digital Video Broadcasting-Terrestrial), which are the current broadcasting systems for terrestrial digital television broadcasting, adopt OFDM, A signal called a pilot (Scattered Pilot, hereinafter referred to as SP) signal is inserted. As shown in FIG. 8, the SP signal has an arrangement in which pilot carriers are dispersed in both the carrier direction and the symbol direction, and there is a problem that the technique disclosed in Document 3 cannot be applied as it is.
[0008]
The present invention has been made to solve such a problem, and an object thereof is to make it possible to optimally synthesize a signal from an array antenna for each subcarrier of an OFDM signal based on an SP signal. Another object of the present invention is to provide an OFDM signal synthesizing receiver capable of satisfactorily reproducing a signal of a desired wave even in a situation where frequency selective distortion due to mixing of multipath components occurs.
[0009]
[Means for Solving the Problems]
In view of the above points, the invention according to claim 1 is configured such that each of a plurality of antenna branches constituting a receiving antenna receives an OFDM signal, and among the signals constituting the OFDM signal, mutual communication between the antenna branches is performed. In the receiving apparatus for synthesizing a signal in which both the symbol number and the carrier number, which is the subcarrier number, for identifying the signal to be compared with the OFDM signal received for each antenna branch, On the other hand, SP that is a scattered pilot signal in which both the symbol number and the carrier number are the same from the means for performing the fast Fourier transform processing to generate the OFDM spectrum and the OFDM spectrum for each antenna branch Same symbol as pilot carrier extraction means for extracting signal The combination weighting coefficient is applied to each SP signal specified by the same carrier number and the carrier number for specifying each SP signal and the branch number which is the antenna branch number, and added for each subcarrier. Predetermined information relating to a value indicated by the SP signal specified by pilot carrier combining means for generating a combined signal, a symbol number for specifying the combined signal, and a carrier number for specifying the combined signal Pilot signal generating means for generating a reference composite value, and the composite signal having the same symbol number and carrier number for identifying the reference composite value and the composite signal from the reference composite value Means for generating an estimation error which is a value obtained by subtracting the value indicated by And a coefficient updating means for generating a composite weighting coefficient for each antenna branch for each SP carrier in accordance with an SP carrier number that is a subcarrier to which the SP signal is assigned, and the coefficient updating means Based on the respective synthesis weighting coefficients, a predetermined interpolation process relating to the symbol number and the carrier number is performed, and a synthesis weighting coefficient vector identified by the same symbol number and the same branch number, and the synthesis weighting coefficient corresponding to the carrier number is an element Interpolating means for generating a composite weighting coefficient vector, and a symbol number and a carrier number equal to a symbol number and a carrier number for identifying a signal constituting the OFDM signal for each signal constituting the OFDM signal Multiply the composite weighting factor specified by Carrier synthesizing means for generating a synthesized signal by adding the values obtained in the carrier number direction, and a predetermined synthesis weighting coefficient applied by the pilot carrier synthesizing means is generated by the coefficient updating means. And a composition weighting coefficient.
[0010]
With this configuration, it is possible to mix multipath components by enabling optimal synthesis of signals from the array antenna for each subcarrier of the OFDM signal based on a scattered pilot signal in which pilot signals are not evenly inserted. It is possible to realize an OFDM signal synthesizing receiver that can satisfactorily reproduce a signal of a desired wave even in a situation where frequency selective distortion occurs due to.
[0011]
In the invention according to claim 2, each of the plurality of antenna branches constituting the receiving antenna receives an OFDM signal, and among the signals constituting the OFDM signal, signals mutually compared between the antenna branches are obtained. Fast Fourier transform processing is performed on the OFDM signal received for each antenna branch in a receiving apparatus for synthesizing a signal in which both a symbol number and a carrier number which is a subcarrier number for identification are specified. To extract an SP signal that is a scattered pilot signal having the same symbol number and the same carrier number from the OFDM spectrum for each antenna branch Same symbol number and same key as pilot carrier extraction means A composite weighting coefficient is applied to each SP signal specified by the rear number in accordance with a carrier number for specifying each SP signal and a branch number which is the number of an antenna branch, and the resultant signal is added for each subcarrier. A reference that is predetermined information relating to a value indicated by the SP signal specified by a pilot carrier combining means for generating a signal, a symbol number for specifying the combined signal, and a carrier number for specifying the combined signal Pilot signal generating means for generating a combined value, and a value indicated by the combined signal having the same symbol number and carrier number for identifying the reference combined value and the combined signal from the reference combined value , A means for generating an estimation error that is a value obtained by subtracting, the estimation error, and the S Coefficient updating means for generating a composite weighting coefficient for each antenna branch for each SP carrier according to an SP carrier number that is a subcarrier to which a signal is assigned, and the SP generated by the coefficient updating means By performing a predetermined interpolation process on the combination weighting coefficient for each antenna branch for each carrier, a combination weighting coefficient for a data carrier that is a subcarrier other than the SP carrier is generated, and for each antenna branch , Each of which constitutes the OFDM signal of the same symbol number for each antenna branch, and interpolating means for converting into a combined weighting coefficient for each carrier and performing inverse Fourier transform processing to generate a filter coefficient vector The filter unit corresponding to the value indicated by the carrier signal Time domain combining means for generating a combined signal by performing a predetermined convolution integral processing with the number vector and adding the predetermined combining weighting coefficient applied by the pilot carrier combining means is calculated by the coefficient updating means The generated composition weighting coefficient is used.
[0012]
With this configuration, it is possible to mix multipath components by enabling optimal synthesis of signals from the array antenna for each subcarrier of the OFDM signal based on a scattered pilot signal in which pilot signals are not evenly inserted. It is possible to realize an OFDM signal synthesizing receiver that can satisfactorily reproduce a signal of a desired wave even in a situation where frequency selective distortion occurs due to.
[0013]
Further, the invention according to claim 3 has a configuration in which the coefficient updating means generates the composite weighting coefficient based on an LMS (Least Mean Square) algorithm or an RLS algorithm (Recursive Least Squares).
With this configuration, it is possible to mix multipath components by enabling optimal synthesis of signals from the array antenna for each subcarrier of the OFDM signal based on a scattered pilot signal in which pilot signals are not evenly inserted. It is possible to realize an OFDM signal synthesizing receiver that can satisfactorily reproduce a signal of a desired wave even in a situation where frequency selective distortion occurs due to.
[0014]
According to a fourth aspect of the present invention, the filter coefficient vector is a transversal filter coefficient vector, the time domain synthesizing unit includes a transversal filter and an adding unit, and the transversal filter is provided for each antenna branch. The transversal filter coefficient vector and a value indicated by the OFDM signal are subjected to a predetermined convolution integration process, and the adding means adds a value obtained by the convolution integration process to form a composite signal. is doing.
[0015]
With this configuration, it is possible to mix multipath components by enabling optimal synthesis of signals from the array antenna for each subcarrier of the OFDM signal based on a scattered pilot signal in which pilot signals are not evenly inserted. It is possible to realize an OFDM signal synthesizing receiver that can satisfactorily reproduce a signal of a desired wave even in a situation where frequency selective distortion occurs due to.
[0016]
According to a fifth aspect of the present invention, the interpolation means includes an inter-symbol interpolation means and an inverse Fourier transform means, and the inter-symbol interpolation means has a predetermined symbol number with respect to a synthesis weighting coefficient. Means for performing an interpolation process, wherein the inverse Fourier transform means performs an inverse Fourier transform process on the combined weighting coefficient vector subjected to the interpolation process with respect to a symbol number to generate the transversal filter coefficient vector. It has the structure which is a means for this.
[0017]
With this configuration, it is possible to mix multipath components by enabling optimal synthesis of signals from the array antenna for each subcarrier of the OFDM signal based on a scattered pilot signal in which pilot signals are not evenly inserted. It is possible to realize an OFDM signal synthesizing receiver that can satisfactorily reproduce a signal of a desired wave even in a situation where frequency selective distortion occurs due to.
[0018]
According to a sixth aspect of the present invention, the interpolating means includes inter-symbol interpolating means, inter-carrier interpolating means, and inverse Fourier transform means, and the inter-symbol interpolating means , Means for performing a predetermined interpolation process on the symbol number, and the inter-carrier interpolation means is a means for performing a predetermined interpolation process on the carrier number with respect to the combined weighting coefficient vector obtained by the interpolation process. And the inverse Fourier transform means performs an inverse Fourier transform process on the combined weighting coefficient vector that has been subjected to the interpolation process with respect to a symbol number and has been subjected to the interpolation process with respect to a carrier number, and the transversal filter It has a configuration which is a means for generating a coefficient vector.
[0019]
With this configuration, it is possible to mix multipath components by enabling optimal synthesis of signals from the array antenna for each subcarrier of the OFDM signal based on a scattered pilot signal in which pilot signals are not evenly inserted. It is possible to realize an OFDM signal synthesizing receiver that can satisfactorily reproduce a signal of a desired wave even in a situation where frequency selective distortion occurs due to.
[0020]
DETAILED DESCRIPTION OF THE INVENTION
The OFDM signal combining receiver of the present invention will be described below with reference to the accompanying drawings.
FIG. 1 is a block diagram showing a schematic configuration of a receiving apparatus 100 for OFDM signal synthesis according to the first embodiment of the present invention. The OFDM signal combining receiver 100 includes antenna branches 0 to (L-1), fast Fourier transform (hereinafter referred to as FFT (Fast Fourier Transform)) means 101 (0) to 101 (L-1), and pilot carrier extraction. Means 110, carrier synthesizing means 120, pilot carrier synthesizing means 130, coefficient updating means 140, difference calculating means 150, pilot signal generating means 160, and interpolation means 170.
[0021]
Prior to the description of each component means, terms, symbols, definitions and others will be described. Signals transmitted in the ISDB-T (Integrated Services Digital Broadcasting-Terrestrial) system and the DVB-T (Digital Video Broadcasting-Terrestrial) system, which are broadcasting systems for digital terrestrial television broadcasting, include Scattered Pilot (hereinafter referred to as Scattered Pilot). A reference signal called SP) is inserted. FIG. 8 shows the arrangement of SP signals in the carrier-symbol space. In FIG. 8, the SP signal is represented by a black circle, and the other data signal is represented by a white circle.
[0022]
In the following description, the number L of antenna branches (hereinafter simply referred to as “branches”) constituting the receiving antenna is referred to as the number of branches, and the branch number, which is a number assigned to any branch, is set to l (0 ≦ l ≦ L). -1). The total number of subcarriers constituting the OFDM signal is K (K is an odd number), and a number assigned to any subcarrier (hereinafter simply referred to as a carrier number) is represented by k (0 ≦ k ≦ K−1). . Further, a number (generally corresponding to time, hereinafter referred to as a symbol number) assigned to an arbitrary symbol constituting the OFDM signal is i.
[0023]
In the signal arrangement shown in FIG. 8, when the symbol number is i, the carrier number of the carrier serving as the SP signal (hereinafter referred to as SP carrier) is k._p(Hereinafter, the carrier number of the SP carrier is also referred to as the SP carrier number), the SP carrier number k._pIs defined by the following equation (1).
[Expression 1]

Here, (imod4) represents the remainder when i is divided by 4, and p is a non-negative integer.
[0024]
An OFDM signal vector specified by an arbitrary symbol number i and carrier number ku _k(i) is defined by the following equation (2), and an OFDM signal vector specified by an arbitrary symbol number i and branch number lv _l(i) is defined by the following formula (3).
[Expression 2]

Where u_{l, k}(i) is a signal specified by a branch number l, a carrier number k, and a symbol number i in the frequency spectrum of the OFDM signal (hereinafter referred to as OFDM spectrum), and T represents a transposed vector.
[0025]
Also, the combined signal y of the OFDM spectrum_k(i) is defined by the following formula (4).
[Equation 3]

here,w b _k(i) is a combined weighting coefficient vector specified by the carrier number k and the branch number l, which is defined by the following equation (5), and H represents a complex conjugate transposed vector.
[Expression 4]

Where w_{l, k}(i) is a weighting factor specified by the branch number l, the spectrum number k, and the symbol number i.
[0026]
Next, the composite weighting coefficient vector represented by the above equation (5)w b _kA method of updating (i) by applying the LMS algorithm will be described.
Carrier number k at time i, that is, symbol number i_pThe total value d for the branch direction of the SP signal specified by_kp(i), symbol number i, carrier number k_pSP signal composite signal y specified by_kpDifference from (i) (hereinafter referred to as posterior estimation error) e_kp(i) is represented by the following formula (6).
[Equation 5]

[0027]
For the same SP carrier number, since the SP signal is repeated in a 4-symbol period, the symbol number of the next SP signal that is updated based on the symbol number i is i + 4. And carrier number k_pUpdate value of the composite weighting coefficient vector specified byw b _kp(i + 4) is defined by the following formula (7).
[Formula 6]

Here, μ is a parameter that defines a change amount before and after the update, called a step size parameter, and * represents a complex conjugate.
[0028]
Combining weighting coefficient vector used for combining data signals that are signals in symbol-carrier space other than SP signalsw b _k(i) represents each symbol number i and each SP carrier number k calculated in this way._pA composite weighting coefficient vector specified byw b _kp(i) andw b _kpIt is assumed that it is generated by interpolation based on (i + 4). An interpolation method will be described later.
[0029]
Hereinafter, each component of the OFDM signal combining receiver 100 will be described.
The FFT means 101 (0) is obtained by performing a known fast Fourier transform process (hereinafter referred to as FFT process) on the OFDM signal received by the branch (shown as an antenna in FIG. 1) 0 and performing the FFT process. OFDM signal vector extracted from the frequency spectrum of the OFDM signal (hereinafter referred to as OFDM spectrum)v _l(i) (refer to the above equation (3)), information of symbol number i that is the time when the OFDM signal is received, branch number l and other necessary information are output to pilot carrier extracting means 110 and carrier synthesizing means 120 It is a means to do.
[0030]
Unless otherwise specified, if it is described that a signal or information such as an OFDM signal vector with a subscript such as a branch number l and an argument such as a symbol number i is input / output, this is specified. Subscripts, arguments, etc. necessary for the above are assumed to be input / output along with the signal, etc., and their description is omitted.
The FFT means 101 (1) to 101 (L-1) perform processing similar to the processing performed by the FFT means 101 (0) on the OFDM signals received in the branches 1 to (L-1), respectively.
[0031]
The pilot carrier extraction means 110 outputs each OFDM signal vector output from the FFT means 101 (0) to 101 (L-1).v _lThe SP signal when (i) is input and the symbol number is i is the SP carrier number k shown in the above equation (1)._pEach OFDM signal vector input as a carrier signal specified inv _lSP signal u of each branch extracted from (i)_{l, k}(i) is arranged as shown in Equation (2) above, and an OFDM signal vectoru _kgenerate (i) and each OFDM signal vectoru _kThis is a means for outputting (i) to the pilot carrier combining means 130 and the display unit 140.
[0032]
The carrier synthesizing unit 120 outputs each OFDM signal vector output from the FFT units 101 (0) to 101 (L-1).v _l(i) and a synthetic weighting coefficient vector output from interpolation means 170 described later.w b _k(i) as an input, and the synthesized signal y based on the above equations (2), (4), and (5)_k(i) to form a composite signal y_kThis is means for outputting (i) to the outside.
[0033]
Hereinafter, a synthesis weighting coefficient vector used in the carrier synthesis means 120w b _kProcessing performed by the pilot carrier synthesizing unit 130, the coefficient updating unit 140, the difference calculating unit 150, the pilot signal generating unit 160, and the interpolation unit 170 to generate (i) will be described.
[0034]
The pilot carrier combining unit 130 outputs each OFDM signal vector output from the pilot carrier extracting unit 110.u _k(i) and a synthetic weighting coefficient vector output from coefficient updating means 140 described later.w b _kp(i) as an input, and based on the above equation (4), the synthesized signal y_kp(i) is generated and output to the difference calculation means 150, and the input composite weighting coefficient vectorw b _kp(i) is output to the interpolation means 170, and the information of the symbol number i and the carrier number k_pIs output to the pilot signal generating means 160.
[0035]
Pilot signal generating means 160 has a predetermined value (hereinafter referred to as a reference composite value) d corresponding to the value of the SP signal inserted in advance into the OFDM signal._k(i) is symbol number i, SP carrier number k_pInformation corresponding to the symbol number i output from the pilot carrier combining means 130 and the SP carrier number k._pAnd the symbol number i and the SP carrier number k based on the input information._pReference composite value d specified by_kpThis is means for outputting (i) to the difference calculation means 150 described later.
[0036]
The difference calculation means 150 is a combined signal y output from the pilot carrier combining means 130._kp(i) and the reference composite value d output from the pilot signal generating means 160_kp(i) as an input, and the posterior estimation error e shown in the above equation (6)_kpThis is a means for calculating (i) and outputting it to the coefficient updating means 140 described later.
[0037]
The coefficient updating means 140 is an OFDM signal vector output from the pilot carrier extracting means 110.u _kp(i) and a posteriori estimation error e output from the difference calculation means 150_kp(i) as an input and based on equation (7) above, a combined weighting coefficient vectorw b _kp(i) is updated, and the combined weighting coefficient vector obtained by the updatew b _kpThis is a means for outputting (i + 4) to the pilot carrier combining means 130. Here, when the update process with the symbol number i is completed, the coefficient update unit 140 performs the synthesis weighting coefficient vector obtained by the update process.w b _kp(i + 4) may be stored for the next processing. Note that the composite weighting coefficient vector used for the first update processw b _kpAs the value of (0), a predetermined value may be recorded or generated by other methods.
[0038]
Interpolating means 170 is a combined weighting coefficient vector output from pilot carrier combining means 130.w b _kp(i) andw b _kp(i + 4) is input, and the carrier synthesizing means 120 receives the synthesized signal y_kcomposite weighting coefficient vector used to form (i)w b _kThis is a means for generating (i) by interpolation and outputting it to the carrier synthesizing means 120.
[0039]
By using the OFDM signal combining receiver 100 having the configuration shown in FIG. 2 as the OFDM signal combining receiver 100, the interpolating means 170 is composed of the intersymbol interpolating means 277 and the intercarrier interpolating means 278. The interpolation processing performed by the means 170 can be performed separately for the symbol axis direction and the carrier axis direction. This will be described below.
[0040]
First, an interpolation method in the symbol axis direction will be described. As a specific example of the interpolation method in the symbol axis direction, the latest value holding method and the linear interpolation method will be described.
The latest value holding method here is a composite weighting coefficient vector shown in the above equation (7) as shown in the following equation (8).w b _kpThe value of (i + 4) is a composite weighting coefficient vectorw b _kp(i + 1),w b _kp(i + 2),w b _kpThis is substituted for (i + 3). Needless to say, the composite weighting coefficient vector shown in equation (7) above.w b _kp(i + 4) is obtained by the update process in the symbol axis direction. Symbol numbers (i + 1), (i + 2),and (i + 3) is a point between the symbol number i before update and the symbol number i + 4 after update.
[Expression 7]

[0041]
Next, the straight line interpolation method starts with ΔΔ according to the following equation (9):w b _kp(i) is generated, and a composite weighting coefficient vector for an SP signal having a symbol number between symbol numbers i and i + 4w b _kp(i + 1),w b _kp(i + 2),w b _kpThe value of (i + 3) is calculated according to equations (10) to (12).
[Equation 8]

[0042]
By this interpolation in the symbol axis direction, the SP signal inserted in the 12-carrier cycle in the carrier axis direction when received is equivalent to the SP signal inserted in the 3-carrier cycle. That is, the complemented SP carrier number is represented by the following formula (13) instead of the above formula (1).
[Equation 9]

Here, p is a non-negative integer as in the above formula (1).
[0043]
Intersymbol interpolation means 277 is a combined weighting coefficient vector output from pilot carrier combining means 130.w b _kp(i) andw b _kp(i + 4) as an input and a composite weighting coefficient vector of symbol numbers between symbol numbers i and i + 4w b _kp(i + 1),w b _kp(i + 2),w b _kpYou may produce | generate the value of (i + 3) as a value shown to said Formula (8) or said Formula (9)-(13). Composite weighting coefficient vector obtained by interpolationw b _kp(i + 1),w b _kp(i + 2),w b _kpThe information of (i + 3) is output to the intercarrier interpolating means 278. The interpolation method in the symbol axis direction may be a method other than the above method.
[0044]
Next, as specific examples of the interpolation method with respect to the carrier axis direction, three methods of a linear interpolation method, an FFT interpolation method, and a convolution interpolation method will be described.
First, the straight line interpolation method will be described.
By interpolation in the symbol axis direction, the SP signal is inserted in a three-carrier cycle in the carrier axis direction. The linear interpolation method starts with δ according to the following equation (14):w b _kp(i) is generated and the carrier number k_pAnd k_pComposite weighting coefficient vector for SP signals with carrier numbers between +3w b _{kp + 1}(i),w b _{kp + 2}The value of (i) is calculated according to the formulas (14) to (16).
[Expression 10]

[0045]
This weighting coefficient vector for all SP signals by interpolation in the carrier axis directionw b _k(i) is required. However, since the linear interpolation method is a first-order approximation method and the accuracy of the approximation may not be good, the following FFT interpolation method that replaces the linear interpolation method will be described below.
[0046]
Composite weighting coefficient vectorw b _k(i) was composed of L elements in the branch axis direction as shown in the above equation (5). Here, the composite weighting coefficient vector in the carrier axis directionw c _l(i) is defined by the following formula (17).
## EQU11 ##

[0047]
Here, the composite weighting coefficient vector in the carrier axis directionw c _lAmong the constituent elements of (i), the elements specified by the carrier number other than the SP carrier number shown in the above formula (13) are filled with 0. That is, W shown in the following formula (18)_{l, k}The coefficient vector having the structure as shown in the following equation (19) using (i)W c _lDefine (i). This is called zero interpolation.
[Expression 12]

[0048]
Next, the coefficient vectorW c _l(i) is subjected to inverse fast Fourier transform (hereinafter referred to as IFFT (Inverse Fast Fourier Transform)) processing. At that time, the number of points subject to IFFT processing is N = 2ⁿ And Here, ν is a natural number, and the condition of N> K is satisfied. Let n be an impulse number (integer) that is a number assigned to an arbitrary impulse obtained after the IFFT processing. Here, the range of n is a range shown in the following formula (20).
[Formula 13]

[0049]
Where the coefficient vectorW c _lTo make (i) an N-point coefficient vector, a coefficient vectorW c _lInsert a (N−K−1) / 2 zero vector at the left end of (i) to obtain a coefficient vectorW c _lA zero vector consisting of {(N−K−1) / 2} +1 zeros is inserted at the right end of (i), and a coefficient vector shown in the following equation (21) is generated.
[Expression 14]

here,0Is a zero vector defined by the following equation (22).
[Expression 15]

[0050]
As shown in equation (21) above, the coefficient vectorW c _lAt the end of (i), that is, at both ends of the OFDM signal band0The error at the OFDM signal band edge of the combined weighting coefficient vector can be reduced by packing and extrapolating the weight applied to the pilot signal at the band edge instead of zeroing the vector. The state of extrapolation is shown in FIG. That is, in FIG. 9, as shown in the following formulas (23) to (25), a vector composed of (N−K−1) / 2 elements at the left end of the OFDM signal band.AAnd a vector of {(N−K−1) / 2} +1 elements at the right end of the OFDM signal bandBThe state of inserting is shown. Note that the vector shown in the following equation (23) is changed to a coefficient vector.W c _lWe will call it (i).
[0051]
[Expression 16]

[0052]
The coefficient vector shown in the above equation (23) thus formedW c _lIFFT processing is performed for each branch for (i). The vector obtained by this IFFT processing is expressed as an impulse response vector.h _lThat's it. In other words, the coefficient vectorW c _lAn impulse response vector obtained by converting (i) by the following equation (26)h _lIs to generate
[Expression 17]

[0053]
Where the impulse response vectorh _lIs defined as shown in the following equation (27).
[Formula 18]

Impulse response vectorh _lAn example of this is schematically shown in FIG.
[0054]
In order to remove so-called aliasing, the following processing is performed. Before entering the description of the process, the filter coefficient vectorfIs defined by the following equation (28).
[Equation 19]

Filter coefficient vectorfAn example of this is schematically shown in FIG. 10 as a curve indicated by a reference symbol 1002.
[0055]
In order to remove aliasing, an impulse response vector as shown in equation (29) below:h _lAnd filter coefficient vectorfFor each element, and FFT processing is performed (the symbol “●” in the following expression represents the product for each element).
[Expression 20]

[0056]
This results in an interpolated composite weighting coefficient vectorW c _l(i) is generated. In this way, the composite weighting coefficient vectorW c _lBy generating (i) for all branches, the combined weighting coefficient vector of all the generated branchesW c _lComposite weighting coefficient vector interpolated in the carrier axis direction from the data of (i)W b _k(i) can be generated.
[0057]
Next, the convolution interpolation method will be described.
First, the filter coefficient vectorfImpulse response vectorg(Less than,fImpulse response vectorgThat's it. ) Is defined by the following equation (30).
[Expression 21]

AndfImpulse response vectorgIs represented as a vector shown in the following equation (31).
[Expression 22]

[0058]
Filter coefficient vectorfIf the characteristic is not a rectangle but a characteristic that gently attenuates toward zero at both ends,fImpulse response vectorgThe characteristic is also such that approximately zero continues at both ends,fImpulse response vectorgThe number of elements can be reduced from N to M as shown in the following equation (32).
[Expression 23]

[0059]
Defined like thisfImpulse response vectorgAnd the coefficient vector defined by equation (19) aboveW c _lBy performing a convolution operation using (i) as shown in the following equation (33), a composite weighting coefficient W specified by an arbitrary carrier number k and an arbitrary branch number l_{l, k}(i) can be generated.
[Expression 24]

[0060]
The composite weighting coefficient W generated in this way_{l, k}(i) is w of the above formula (5)_{l, k}A composite weighting coefficient vector interpolated in the carrier axis direction by substituting in (i)w b _k(i) can be generated.
The interpolation method in the carrier axis direction may be a method other than the above method.
[0061]
The carrier interpolating means 278 receives the combined weighting coefficient vector interpolated in the symbol axis direction output from the symbol interpolating means 277, and performs the above-described linear interpolation method, FFT interpolation method, convolution interpolation method, and other interpolations. Composite weighting coefficient vector interpolated in the carrier axis direction by the methodw b _kThis is a means for generating (i) and outputting it to the carrier synthesizing means 120.
[0062]
As described above, the OFDM signal combining receiver 100 according to the first embodiment of the present invention optimally combines the signals from the array antenna for each subcarrier of the OFDM signal based on the SP signal. By making it possible, the signal of the desired wave can be satisfactorily reproduced even in a situation where frequency selective distortion occurs due to mixing of multipath components.
[0063]
FIG. 3 is a block diagram showing a schematic configuration of a receiving apparatus 300 for OFDM signal synthesis according to the second embodiment of the present invention. The OFDM signal synthesizing receiver 300 according to the second embodiment generates a synthesis weighting coefficient by applying an RLS (Recursive Least Squares) algorithm. Hereinafter, description will be given with reference to the drawings.
[0064]
The OFDM signal combining receiver 100 includes antenna branches 0 to (L-1), fast Fourier transform (hereinafter referred to as FFT (Fast Fourier Transform)) means 101 (0) to 101 (L-1), and pilot carrier extraction. Means 110, carrier synthesizing means 120, pilot carrier synthesizing means 330, coefficient updating means 340, difference calculating means 150, pilot signal generating means 160, inter-symbol interpolating means 277, and inter-carrier interpolating means 278.
[0065]
Of the means constituting the OFDM signal combining receiver 300, the same reference numerals are used for the same processing as the means in the OFDM signal combining receiver 100 according to the first embodiment of the present invention. The description is omitted.
OFDM signal vectoru _k(i) Composite weighting coefficient vectorw b _k(i) The same symbols and terms as defined in the first embodiment of the present invention, such as the SP signal carrier number kp, etc., are also defined in the second embodiment of the present invention. And
[0066]
In the second embodiment of the present invention, the combined signal y of the OFDM spectrum_k(i) is defined as shown in the following formula (34).
[Expression 25]

Composite signal y_kThe generation of (i) is related to the synthesis weighting coefficient vector having the symbol number (i-4) because the symbol number in which the SP signal immediately before the symbol number i exists is (i-4).
[0067]
Hereinafter, this composite weighting coefficient vectorw b _kA method for generating (i) will be described.
First, SP signal synthesis weighting coefficient vectorw b _kpThe generation of (i) will be described. The composite weighting coefficient vector of the SP signal is generated as shown in the following formula (35) by applying the RLS algorithm.
[Equation 26]

[0068]
here,k _kp(i) is a gain vector defined as shown in the following formula (36), and ξ_kp(I) is a prior estimation error defined as shown in the following equation (37).
[Expression 27]

[0069]
Where λ^-1Is the forgetting factor,P _kp(i) is an L × L correlation inverse matrix defined as shown in the following equation (38), and d_kp(i) is a reference composite value defined in the first embodiment of the present invention.
[Expression 28]

[0070]
Hereinafter, as in the first embodiment of the present invention, unless otherwise specified, signals such as OFDM signal vectors, information, etc., with subscripts such as branch numbers l and arguments such as symbol numbers i are input / output. If so, the subscripts, arguments, etc. necessary to identify them are input / output along with the signals, etc., and the description is omitted.
[0071]
The pilot carrier combining unit 330 outputs each OFDM signal vector output from the pilot carrier extracting unit 110.u _k(i) and a synthetic weighting coefficient vector output from coefficient updating means 340 described later.w b _kp(i-4) as an input, and based on the above equation (34), the synthesized signal y_kp(i) is generated and output to the difference calculation means 150, and the input composite weighting coefficient vectorw b _kp(i-4) is output to the interpolation means 170, and the information of the symbol number i and the carrier number k_pIs output to the pilot signal generating means 160.
[0072]
The coefficient updating means 340 is an OFDM signal vector output from the pilot carrier extracting means 110.u _kp(i) and the prior estimation error ξ output from the difference calculation means 150_kp(I) as an input, and a combined weighting coefficient vector based on the above equations (35) to (38)w b _kp(i-4) is updated, and the combined weighting coefficient vector obtained by the updatew b _kpThis is means for outputting (i) to the pilot carrier combining means 330. Here, the coefficient updating unit 340, when the update process with the symbol number i-4 is completed, the composite weighting coefficient vector obtained by the update process.w b _kp(i) may be stored for the next processing. Note that a predetermined value may be recorded as a value of the synthesis weighting coefficient vector used for the update process performed first, or may be generated by other methods.
[0073]
Intersymbol interpolation means 277 is a combined weighting coefficient vector output from pilot carrier combining means 330.w b _kp(i-4) andw b _kp(i) as an input and a composite weighting coefficient vector for SP signals having symbol numbers between symbol numbers i-4 and iw b _kp(i-3),w b _kp(i-2),w b _kpYou may produce | generate the value of (i-1) as a value shown in the following formula | equation (7) or the following formula | equation (8)-(12). Composite weighting coefficient vector obtained by interpolationw b _kp(i-1),w b _kp(i-2),w b _kpThe information of (i-3) is output to the intercarrier interpolating means 278. The interpolation method in the symbol axis direction may be a method other than the above method.
[0074]
In the second embodiment of the present invention, as in the first embodiment of the present invention, the latest value holding method and the linear interpolation method can be used as the interpolation method in the symbol axis direction. Of course, other methods may be used.
In the second embodiment of the present invention, as shown in the following equation (39), the latest value holding method is a combined weighting coefficient vector represented by the above equation (38).w b _kpThe value of (i) is the composite weighting coefficient vectorw b _kp(i-1),w b _kp(i-2),w b _kpThis is substituted for (i-3). Needless to say, the composite weighting coefficient vector shown in equation (38) above.w b _kp(i) is obtained by the update process in the symbol axis direction. Symbol numbers (i-1), (i-2),and (i-3) is a point between the symbol number i-4 before update and the symbol number i after update.
[Expression 29]

[0075]
Next, the linear interpolation method starts with ΔΔ according to the following equation (40):w b _kp(i) is generated, and a composite weighting coefficient vector for an SP signal having a symbol number between symbol numbers i and i-4w b _kp(i-1),w b _kp(i-2),w b _kpThe value of (i-3) is calculated according to equations (41) to (43).
[30]

[0076]
By this interpolation in the symbol axis direction, the SP signal inserted in the 12-carrier cycle in the carrier axis direction when received is equivalent to the SP signal inserted in the 3-carrier cycle. That is, the complemented SP carrier number is represented by the following formula (44) instead of the above formula (1).
[31]

Here, p is a non-negative integer as in the above formula (1).
[0077]
Intersymbol interpolation means 277 is a combined weighting coefficient vector output from pilot carrier combining means 330.w b _kp(i) andw b _kp(i-4) as an input and a composite weighting coefficient vector of symbol numbers between symbol numbers i and i-4w b _kp(i-1),w b _kp(i-2),w b _kpYou may produce | generate the value of (i-3) as a value shown to said Formula (39) or said Formula (40)-(44). Composite weighting coefficient vector obtained by interpolationw b _kp(i-1),w b _kp(i-2),w b _kpThe information of (i-3) is output to the intercarrier interpolating means 278. The interpolation method in the symbol axis direction may be a method other than the above method.
[0078]
Next, as specific examples of the interpolation method with respect to the carrier axis direction, three methods of a linear interpolation method, an FFT interpolation method, and a convolution interpolation method will be described.
First, the straight line interpolation method will be described.
By interpolation in the symbol axis direction, the SP signal is inserted in a three-carrier cycle in the carrier axis direction. The linear interpolation method starts with δ according to the following equation (45):w b _kp(i-4) is generated and the carrier number k_pAnd k_pComposite weighting coefficient vector for SP signals with carrier numbers between +3w b _{kp + 1}(i-4),w b _{kp + 2}The value of (i-4) is calculated according to equations (45) to (47).
[Expression 32]

[0079]
This weighting coefficient vector for all SP signals by interpolation in the carrier axis directionw b _k(i) is required.
Next, the FFT interpolation method will be described. Composite weighting coefficient vectorw b _k(i-4) is composed of L elements in the branch axis direction as shown in the above equation (5). Here, the composite weighting coefficient vector in the carrier axis directionw c _l(i-4) is defined by the following formula (48).
[Expression 33]

[0080]
Here, the composite weighting coefficient vector in the carrier axis directionw c _lAmong the components of (i-4), 0 is filled in the elements specified by the carrier number other than the SP carrier number shown in the above equation (44). That is, W shown in the following formula (49)_{l, k}A coefficient vector having the structure as shown in the following equation (50) using (i-4)W c _lDefine (i-4). This is called zero interpolation.
[Expression 34]

[0081]
Next, the coefficient vectorW c _l(i-4) is subjected to inverse fast Fourier transform (hereinafter referred to as IFFT (Inverse Fast Fourier Transform)) processing. At that time, the number of points subject to IFFT processing is N = 2ⁿ And Here, ν is a natural number, and the condition of N> K is satisfied. Let n be an impulse number (integer) that is a number assigned to an arbitrary impulse obtained after the IFFT processing. Here, the range of n is a range shown in the following formula (51).
[Expression 35]

[0082]
Where the coefficient vectorW c _lIn order to make (i-4) a coefficient vector of N points, a coefficient vectorW c _lInsert a (N−K−1) / 2 zero vector at the left end of (i-4) to create a coefficient vectorW c _lA zero vector consisting of {(N−K−1) / 2} +1 zeros is inserted at the right end of (i-4) to generate a coefficient vector represented by the following equation (52).
[Expression 36]

here,0Is a zero vector defined by the following equation (53).
[Expression 37]

[0083]
As shown in equation (52) above, the coefficient vectorW c _lAt the end of (i-4), ie at both ends of the OFDM signal band0The error at the OFDM signal band edge of the combined weighting coefficient vector can be reduced by packing and extrapolating the weight applied to the pilot signal at the band edge instead of zeroing the vector. The state of extrapolation is shown in FIG. That is, in FIG. 9, as shown in the following formulas (54) to (56), a vector composed of (N−K−1) / 2 elements at the left end of the OFDM signal band.AAnd a vector of {(N−K−1) / 2} +1 elements at the right end of the OFDM signal bandBThe state of inserting is shown. Note that the vector shown in the following equation (54) is changed to a coefficient vectorW c _lIt will be called (i-4).
[0084]
[Formula 38]

[0085]
The coefficient vector shown in the above equation (54) thus formedW c _lThe IFFT process is performed for each branch on (i-4). The vector obtained by this IFFT processing is expressed as an impulse response vector.h _l(i-4). In other words, the coefficient vectorW c _lAn impulse response vector obtained by converting (i-4) defined by the following equation (57)h _l(i-4) is generated.
[39]

[0086]
Where the impulse response vectorh _l(i-4) is defined as shown in the following formula (58).
[Formula 40]

Impulse response vectorh _lAn example of (i-4) is schematically shown in FIG. 10 as a curve indicated by the reference symbol 1001.
[0087]
In order to remove so-called aliasing, the following processing is performed. Before entering the description of the process, the filter coefficient vectorfIs defined by the following equation (59).
[Expression 41]

Filter coefficient vectorfAn example of this is schematically shown in FIG. 10 as a curve indicated by a reference symbol 1002.
[0088]
In order to remove aliasing, an impulse response vector as shown in equation (60) below:h _l(i-4) and filter coefficient vectorfFor each element, and FFT processing is performed (the symbol “●” in the following expression represents the product for each element).
[Expression 42]

[0089]
This results in an interpolated composite weighting coefficient vectorW c _l(i) is generated. In this way, the composite weighting coefficient vectorW c _lBy generating (i) for all branches, the combined weighting coefficient vector of all the generated branchesW c _lComposite weighting coefficient vector interpolated in the carrier axis direction from the data of (i)W b _k(i) can be generated.
[0090]
Next, the convolution interpolation method will be described.
First, the filter coefficient vectorfImpulse response vectorg(Less than,fImpulse response vectorgThat's it. ) Is defined by the following formula (61).
[Equation 43]

AndfImpulse response vectorgIs represented as a vector shown in the following equation (62).
(44)

[0091]
Filter coefficient vectorfIf the characteristic is not a rectangle but a characteristic that gently attenuates toward zero at both ends,fImpulse response vectorgThe characteristic is also such that approximately zero continues at both ends,fImpulse response vectorgThe number of elements can be reduced from N to M as shown in the following equation (63).
[Equation 45]

[0092]
Defined like thisfImpulse response vectorgAnd the coefficient vector defined by equation (50) aboveW c _lBy performing a convolution operation using (i-4) as shown in the following equation (64), a composite weighting coefficient W specified by an arbitrary carrier number k and an arbitrary branch number l_{l, k}(i-4) can be generated.
[Equation 46]

[0093]
The composite weighting coefficient W generated in this way_{l, k}(i-4) is replaced with w in the above formula (5)._{l, k}By substituting into (i-4), the combined weighting coefficient vector interpolated in the carrier axis directionw b _k(i-4) can be generated.
The interpolation method in the carrier axis direction may be a method other than the above method.
[0094]
The carrier interpolating means 278 receives the combined weighting coefficient vector interpolated in the symbol axis direction output from the symbol interpolating means 277, and performs the above-described linear interpolation method, FFT interpolation method, convolution interpolation method, and other interpolations. Composite weighting coefficient vector interpolated in the carrier axis direction by the methodw b _kThis is a means for generating (i-4) and outputting it to the carrier synthesizing means 120.
[0095]
As described above, the OFDM signal combining receiver 300 according to the second embodiment of the present invention optimally combines the signals from the array antenna for each subcarrier of the OFDM signal based on the SP signal. By making it possible, the signal of the desired wave can be satisfactorily reproduced even in a situation where frequency selective distortion due to mixing of multipath components occurs.
[0096]
FIG. 4 is a block diagram showing a schematic configuration of an OFDM signal combining receiver 400 according to the third embodiment of the present invention. The OFDM signal synthesizing receiver 400 according to the third embodiment of the present invention synthesizes and outputs the OFDM signals from the respective branches in the time domain. Hereinafter, description will be given with reference to the drawings.
[0097]
The OFDM signal combining receiver 400 includes antenna branches (hereinafter simply referred to as branches) 0 to (L-1), fast Fourier transform (hereinafter referred to as FFT (Fast Fourier Transform)) means 101 (0) to 101 ( L-1), pilot carrier extraction means 110, time domain synthesis means 420, pilot carrier synthesis means 130, coefficient update means 440, difference calculation means 150, pilot signal generation means 160, interpolation means 470, and inverse fast Fourier transform ( Hereinafter, it is comprised of IFFT (Inverse Fast Fourier Transform) means 481 (0) to 481 (L-1).
[0098]
Of the means constituting the OFDM signal combining receiver 400, the same reference numerals are used for the same processing as the constituent means in the OFDM signal combining receiver 100 according to the first embodiment of the present invention. The description is omitted.
OFDM signal vectoru _k(i) Composite weighting coefficient vectorw b _k(i) SP carrier number k_pThe same symbols and the like as the symbols and terms defined in the first embodiment of the present invention are also defined in the second embodiment of the present invention.
[0099]
The IFFT unit 481 (0) is a composite weighting coefficient vector output from the interpolation unit 470 described later.w b _k(i) etc. as input, and input composite weighting coefficient vectorw b _k(i) etc. are subjected to IFFT processing, filter coefficients (hereinafter referred to as transversal filter coefficients) necessary for filtering in the time domain are extracted from vectors obtained by the IFFT processing, and time domain synthesizing means 420 is extracted. It is a means for outputting.
[0100]
Note that the composite weighting coefficient vector input to the IFFT means 481 (0).w b _kThe transversal filter coefficients output from (i) and the like and IFFT means 481 (0) are for filtering the OFDM signal output from branch 0. IFFT means 481 (1) to 481 (L-1) are means for performing the same processing as IFFT means 481 (0), and filter processing of OFDM signals output from branches 1 to (L-1), respectively. Is for.
[0101]
The time domain synthesizing unit 420 receives the OFDM signal output from the branches 0 to (L-1) and the transversal filter coefficients output from the IFFT units 481 (0) to 481 (L-1) as inputs. This is a means for performing a filtering process in the time domain on the OFDM signal from each branch in accordance with the transversal filter coefficient, and synthesizing and outputting the signals of each branch after the filtering process.
[0102]
Specifically, the input OFDM signal U_lBy performing the conversion shown in the following expression (65) on (nt), time is taken for each branch and for each carrier (because it is processing in the time domain, it is not clearly displayed in expression (65)). A combined signal Y (nt) that is filtered in the region and combined for all branches is generated.
[Equation 47]

Here, nt is an arbitrary time number (where n is an integer), l is an arbitrary branch number, and h_{l, m}(i) (0 ≦ m ≦ M−1, where M is the number of taps of the transversal filters 521 (0) to 521 (L−1)) is a transversal filter coefficient, and i is a symbol number is there.
[0103]
The OFDM signal combining receiver 400 is configured like the OFDM signal combining receiver 500 shown in FIG. 5, and the time domain combining means 420 is added to the transversal filters 521 (0) to 521 (L-1). It may be configured by means 522.
[0104]
In that case, as shown in FIG. 5, the transversal filters 521 (0) to 521 (L-1) are connected to the OFDM signals output from the branches 0 to (L-1) and the IFFT units 481 (0) to ( The transversal filter coefficient output from (L-1) is input, the input OFDM signal is subjected to filter processing in the time domain for each branch, and output to the adding means 522.
The adding unit 522 adds the results after the filtering process output from the transversal filters 521 (0) to 521 (L-1) and outputs the result to the outside.
[0105]
In the following, the OFDM signal combining receiver 500 is configured like the OFDM signal combining receiver 600 shown in FIG. 6, and the interpolating means 470 is configured by the inter-symbol interpolating means 671 and the inter-carrier interpolating means 672. It's okay. In this case, the following processing is performed by the symbol interpolating unit 671, the carrier interpolating unit 672, and the IFFT units 481 (0) to 481 (L-1).
[0106]
Symbol interpolating means 671 is a combination weighting coefficient vector output from pilot carrier combining means 130.w b _kp(i) is used as an input, inter-symbol interpolation processing (for example, Expression (8) or Expression (9)-(12)) described in the first embodiment of the present invention is performed, and inter-carrier interpolation is performed. Output to means 672.
[0107]
The intercarrier interpolation means 672 is a combination weighting coefficient vector obtained by the intersymbol interpolation processing described above.w b _kp(i) as an input, and a predetermined weighting process is performed in the carrier axis direction to obtain a combined weighting coefficient vectorw b _k (i) is generated and output to IFFT means 481 (0) to 481 (L-1). Here, as the “predetermined interpolation processing”, for example, linear interpolation processing (expressions (14) to (16)), the convolution interpolation method (expressions (30) to (33)) described in the first embodiment of the present invention. ), There are other methods, but the type of method does not matter.
[0108]
Hereinafter, processing performed by the IFFT units 481 (0) to 481 (L-1) will be described in detail.
Here, the interpolated composite weighting coefficient vector output from the intercarrier interpolating means 672w b _k (i) is shown in the following formula (66).
[Formula 48]

[0109]
In addition, the number of data (also referred to as the number of points) to be subjected to IFFT processing is N = 2.^νAnd However, ν is a natural number and the condition of N> K is satisfied. Let n be the number of an arbitrary impulse obtained by IFFT processing. The range of n is a range shown in the following formula (67).
[Formula 49]

[0110]
Where the coefficient vectorW c _lTo make (i) an N-point coefficient vector, a coefficient vectorW c _lInsert a (N−K−1) / 2 zero vector at the left end of (i) to obtain a coefficient vectorW c _lA zero vector consisting of {(N−K−1) / 2} +1 zeros is inserted at the right end of (i), and a coefficient vector shown in the following equation (68) (this is newly set as a coefficient vector).W c _l(i). ) Is generated.
[Equation 50]

here,0Is a zero vector defined by the following equation (69).
[Formula 51]

[0111]
Next, the coefficient vectorW c _l(i) is subjected to IFFT processing as shown in the following equation (70) to generate a coefficient vector of a transversal filter.
[Formula 52]

The coefficient vector of the transversal filter obtained above is set as shown in the following equation (71).
[53]

[0112]
The expression (71) can be rewritten as the following expression (72) by paying attention to M consecutive elements of the above expression (71).
[Formula 54]

From N transversal filter coefficients, a vector corresponding to the tap length of the transversal filter (this tap length is M).h _lThe continuous elements in are extracted and used as the equation (73) M-dimensional vector shown below, which is used as a transversal filter coefficient vector.
[Expression 55]

[0113]
The transversal filter coefficient vector with M taps is processed as described above.h _lIs generated. In this way, the IFFT means 481 (0) to 481 (L-1) respectively perform transversal filter coefficient vectors for each branch lunch.h _lAre output to the corresponding transversal filters 521 (0) to 521 (L-1).
[0114]
Further, the OFDM signal combining receiver 500 may be configured as the OFDM signal combining receiver 700 shown in FIG. 7, and the interpolation unit 470 may be configured by the inter-symbol interpolation unit 671 and the zero insertion unit 772. . In this case, processing performed by the zero insertion unit 772 and the IFFT units 481 (0) to 481 (L-1) will be specifically described below.
[0115]
The zero insertion means 772 is a composite weighting coefficient vector that has been subjected to inter-symbol interpolation processing (for example, Expression (8) or Expressions (9) to (12)).w b _kp(i) as input, and input composite weighting coefficient vectorw b _kp(i) is subjected to the zero insertion processing (formulas (17) to (25)) described in the first embodiment of the present invention to obtain a coefficient vector.W c _l(i) (Formulas (23) to (25)) are generated and output to the IFFT means 481 (0) to 481 (L-1).
[0116]
IFFT means 481 (0) to 481 (L-1) are coefficient vectors obtained by the zero insertion process.W c _lAn IFFT process (Equation (26)) is performed on (i) to generate an impulse response vector (Equation (27)). Below, Formula (27) is described again.
[56]

[0117]
The expression (27) can be rewritten as the following expression (74) by paying attention to M consecutive elements of the above expression (27).
[Equation 57]

[0118]
A vector corresponding to the tap length of the transversal filter (this tap length is assumed to be M) from N impulse response vectors.h _lThe continuous elements in are extracted and used as an expression (75) M-dimensional vector shown below to form a transversal filter coefficient vector.
[Formula 58]

Thus, the coefficient of the M-tap transversal filterh _lWanted.
[0119]
The transversal filter coefficient vector with M taps is processed as described above.h _lIs generated. In this way, the IFFT means 481 (0) to 481 (L-1) respectively perform transversal filter coefficient vectors for each branch lunch.h _l(Expression (75)) is generated and output to the corresponding transversal filters 521 (0) to 521 (L-1). The processing performed in the intercarrier interpolating means 672 and the IFFT means 481 (0) to 481 (L-1) corresponds to the FFT interpolation method described in the first embodiment of the present invention.
[0120]
The description in the third embodiment of the present invention has been given for the case where the LMS algorithm is applied. ), The argument of the transversal filter coefficient shown in (75) is only changed from (i) to (i-4), and other parts are not substantially changed.
[0121]
As described above, OFDM signal combining receivers 400, 500, 600, and 700 according to the third embodiment of the present invention receive signals from the array antenna for each subcarrier of the OFDM signal based on the SP signal. By making it possible to synthesize optimally, it is possible to satisfactorily reproduce the signal of the desired wave even in a situation where frequency selective distortion due to mixing of multipath components occurs.
[0122]
【The invention's effect】
As described above, the present invention can realize an OFDM signal synthesizing receiver that can satisfactorily reproduce a signal of a desired wave even in a situation where frequency selective distortion due to mixing of multipath components occurs.
[Brief description of the drawings]
FIG. 1 is a block diagram showing an example of a configuration of a receiving apparatus for OFDM signal synthesis according to a first embodiment of the present invention.
FIG. 2 is a block diagram showing an example of a configuration of a receiving apparatus for OFDM signal synthesis according to the first embodiment of the present invention.
FIG. 3 is a block diagram showing an example of a configuration of a receiving apparatus for OFDM signal synthesis according to a second embodiment of the present invention.
FIG. 4 is a block diagram showing an example of a configuration of an OFDM signal combining receiver according to a third embodiment of the present invention.
FIG. 5 is a block diagram showing an example of a configuration of a receiving apparatus for OFDM signal synthesis according to a third embodiment of the present invention.
FIG. 6 is a block diagram showing an example of a configuration of an OFDM signal combining receiver according to a third embodiment of the present invention.
FIG. 7 is a block diagram showing an example of a configuration of an OFDM signal combining receiver according to a third embodiment of the present invention.
FIG. 8 is a diagram showing an arrangement of signals constituting an OFDM signal in a carrier-symbol space.
FIG. 9 is a diagram for explaining an extrapolation method of a synthesis weighting coefficient according to the present invention.
FIG. 10 is a diagram for explaining an interpolation method using FFT of the present invention.
[Explanation of symbols]
0 to (L-1) Antenna branch
100, 200, 300, 400, 500, 600, 700 OFDM signal combining receiver
101 (0) to (L-1) Fast Fourier transform (FFT) means
110 Pilot carrier extraction means
120 Carrier synthesis means
130, 330 Pilot carrier combining means
140, 340, 440 coefficient update means
150 Difference calculation means
160 Pilot signal generating means
170 Interpolation means
277 Interpolating means between symbols
278 Intercarrier interpolating means
420 Time domain synthesis means
470 Interpolation means
481 (0) to (L-1) Inverse Fast Fourier Transform (FFT) means
521 (0) to (L-1) Transversal filter
522 Adding means
671 Symbol interpolating means
672 Intercarrier interpolating means
772 Zero insertion means
1001 Impulse response vector
1002 Filter coefficient vector

Claims (6)

Each of a plurality of antenna branches constituting a receiving antenna receives an OFDM signal, and among the signals constituting the OFDM signal, a symbol number and a subcarrier for identifying signals to be compared with each other between the antenna branches In the receiving apparatus for synthesizing signals having the same carrier number, the OFDM signal received for each antenna branch is subjected to fast Fourier transform processing to generate an OFDM spectrum. And a pilot carrier extracting means for extracting an SP signal which is a scattered pilot signal having the same symbol number and carrier number from the OFDM spectrum for each antenna branch, and the same symbol number and Identified by the same carrier number A pilot carrier for generating a composite signal by multiplying the SP signal by a composite weighting coefficient in accordance with a carrier number for specifying each SP signal and a branch number which is an antenna branch number and adding each of the subcarriers Generating a reference composite value that is predetermined information relating to a value indicated by the SP signal specified by a combining means, a symbol number for specifying the combined signal, and a carrier number for specifying the combined signal A value obtained by subtracting a value indicated by the combined signal having the same symbol number and carrier number for identifying the reference combined value and the combined signal from the pilot signal generating means and the reference combined value Means for generating an estimation error, the estimation error, and a subkey to which the SP signal is assigned. Coefficient updating means for generating a composite weighting coefficient for each antenna branch for each SP carrier according to the rear SP carrier number, and a symbol based on each composite weighting coefficient generated by the coefficient update means A predetermined interpolation process relating to the number and the carrier number is performed to generate a composite weighting coefficient vector identified by the same symbol number and the same branch number, and including a composite weighting coefficient corresponding to the carrier number as an element And interpolating means for multiplying each signal constituting the OFDM signal by a symbol number for identifying the signal constituting the OFDM signal, a symbol number equal to the carrier number, and a composite weighting factor specified by the carrier number Value obtained in the carrier number direction. Carrier combining means for generating a combined signal by adding together, and the predetermined combining weighting coefficient applied by the pilot carrier combining means is a combining weighting coefficient generated by the coefficient updating means. An OFDM signal synthesizing receiver. Each of a plurality of antenna branches constituting a receiving antenna receives an OFDM signal, and among the signals constituting the OFDM signal, a symbol number and a subcarrier for identifying signals to be compared with each other between the antenna branches In the receiving apparatus for synthesizing signals having the same carrier number, the OFDM signal received for each antenna branch is subjected to fast Fourier transform processing to generate an OFDM spectrum. And a pilot carrier extracting means for extracting an SP signal which is a scattered pilot signal having the same symbol number and carrier number from the OFDM spectrum for each antenna branch, and the same symbol number and Identified by the same carrier number A pilot carrier for generating a composite signal by multiplying the SP signal by a composite weighting coefficient in accordance with a carrier number for specifying each SP signal and a branch number which is an antenna branch number and adding each of the subcarriers Generating a reference composite value that is predetermined information relating to a value indicated by the SP signal specified by a combining means, a symbol number for specifying the combined signal, and a carrier number for specifying the combined signal A value obtained by subtracting a value indicated by the combined signal having the same symbol number and carrier number for identifying the reference combined value and the combined signal from the pilot signal generating means and the reference combined value Means for generating an estimation error, the estimation error, and a subkey to which the SP signal is assigned. Coefficient updating means for generating a composite weighting coefficient for each antenna branch for each SP carrier according to the rear SP carrier number, and each of the SP carriers generated by the coefficient updating means. By performing a predetermined interpolation process on the combination weighting coefficient for the antenna branch, a combination weighting coefficient for a data carrier that is a subcarrier other than the SP carrier is generated, and for each carrier for each antenna branch Interpolation means for converting to a combination weighting coefficient and performing an inverse Fourier transform process to generate a filter coefficient vector, and a value indicated by a signal of each carrier constituting the OFDM signal having the same symbol number for each antenna branch And a predetermined convolution with the corresponding filter coefficient vector Time domain combining means for performing integration processing and generating a combined signal by adding, and the predetermined combining weighting coefficient applied by the pilot carrier combining means is a combining weighting coefficient generated by the coefficient updating means An OFDM signal combining receiver characterized by comprising: 3. The OFDM signal combining receiver according to claim 1, wherein the coefficient updating unit generates the combining weighting coefficient based on an LMS (Least Mean Square) algorithm or an RLS algorithm (Recursive Least Squares). 4. . The filter coefficient vector is a transversal filter coefficient vector, and the time domain synthesis unit includes a transversal filter and an addition unit, and the transversal filter includes the transversal filter coefficient vector and the antenna for each antenna branch. 3. The OFDM signal synthesis according to claim 2, wherein a predetermined convolution integration process with a value indicated by the OFDM signal is performed, and the addition means integrates the values obtained by the convolution integration process into a synthesized signal. Receiving device. The interpolating means is composed of inter-symbol interpolating means and inverse Fourier transform means, and the inter-symbol interpolating means is means for performing a predetermined interpolation process with respect to a symbol number for a synthesis weighting coefficient, The inverse Fourier transform means is means for performing the inverse Fourier transform process on the combined weighting coefficient vector subjected to the interpolation process with respect to the symbol number to generate the transversal filter coefficient vector. The receiving apparatus for combining OFDM signals according to claim 4. The interpolating means includes inter-symbol interpolating means, inter-carrier interpolating means, and inverse Fourier transform means, and the inter-symbol interpolating means performs a predetermined interpolation process on the symbol number with respect to the composite weighting coefficient. The carrier interpolating means is means for performing a predetermined interpolation process on a carrier weight number with respect to the combined weighting coefficient vector obtained by the interpolation process, and the inverse Fourier transform means is Means for generating a transversal filter coefficient vector by performing an inverse Fourier transform process on the combined weighting coefficient vector subjected to the interpolation process with respect to a symbol number and subjected to the interpolation process with respect to a carrier number; 5. The OFDM signal synthesizing receiver according to claim 4.

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