JP4482541B2 - Circularly polarized array antenna - Google Patents

Description

  The present invention relates to a circularly polarized array antenna configured by arranging a plurality of radiating elements and capable of radiating circularly polarized waves from each radiating element.

In wireless communication typified by mobile phones, microwaves are used as carrier waves. In recent years, not only mobile phones, but also wireless LAN (Local Area Network) and wireless USB (
Universal Serial Bus) is also being researched and developed so that microwaves can be used. At present, wireless communication using microwaves has a low data transmission speed, and thus is not suitable for transmitting large amounts of data, for example, data representing video such as still images and moving images. In the microwave band, the band allocated for communication is narrow, and therefore, wireless communication using microwaves has a limited possibility of being used for transmission of a large amount of data.

  Therefore, wireless communication using electromagnetic waves having a frequency higher than that of microwaves, for example, quasi-millimeter waves and millimeter waves of 20 GHz or more, has been attracting attention as a means for transmitting a large amount of data, and research and development have been promoted. ing. Particularly in the 60 GHz band, a wide band is allocated for communication in the world, and wireless communication using such 60 GHz band electromagnetic waves has been put into practical use, and communication between offices is used instead of optical fiber communication. It is used for such as and is spreading.

  There are two types of electromagnetic waves: linearly polarized waves and circularly polarized waves. When using quasi-millimeter waves and millimeter waves, circular polarization is superior to linear polarization in wireless communication for information communication applications. When circularly polarized light is reflected by surrounding obstacles, the turning direction is reversed. The circularly polarized antenna prevents the primary reflected wave from being captured from an obstacle, thereby suppressing the influence of the reflected wave and the multipath. In the case of a circularly polarized antenna, it is not necessary to match the polarization plane of the transmission antenna with the polarization plane of the reception antenna. Such a circularly polarized antenna exhibits power in satellite mobile communications. The circularly polarized antenna can be easily aligned even in communication between the personal computer and the wireless LAN base station, thereby increasing the degree of freedom in installing the personal computer.

  Patent Documents 1 and 2 disclose circularly polarized array antennas. The circularly polarized array antenna disclosed in Patent Document 1 has a plurality of waveguides that are feed line forming bodies, and each waveguide is formed with a plurality of slots to form a radiating element. Set to generate waves. Further, the circularly polarized array antenna disclosed in Patent Document 2 is provided with a plurality of dielectric resonators that are radiating elements on the upper surface of a dielectric waveguide that is a feed line forming body, and circularly polarized waves are generated. Is set as follows.

JP 2002-353727 A JP 2000-36711 A

  In the circularly polarized array antenna disclosed in Patent Document 1, no consideration is given to the fact that the transmission wave is reflected at the end of the waveguide. In the waveguide, the transmission wave travels from the feeding point of the waveguide toward the end of the waveguide. The electric power of this transmission wave is supplied to each radiating element, and thereby circularly polarized waves are radiated from each radiating element. Part of the transmitted wave reaches the end of the waveguide and is reflected at the end of the waveguide. The reflected transmission wave (hereinafter referred to as “reflected wave”) travels toward the feeding point of the waveguide. The power of this reflected wave is supplied to each radiating element, and thereby circularly polarized waves are radiated from each radiating element. The turning direction of the circularly polarized wave by the reflected wave is opposite to the turning direction of the circularly polarized wave by the transmission wave traveling from the feeding point of the waveguide toward the end of the waveguide.

  The circularly polarized wave array antenna disclosed in Patent Document 1 has a problem that the axial ratio increases due to the circularly polarized wave caused by such a reflected wave. Further, the circularly polarized waves caused by the reflected waves strengthen each other in a specific direction different from the front direction, and as a result, there arises a problem that the side lobe rises as a whole.

  The circularly polarized array antenna disclosed in Patent Document 2 has the same problem as the circularly polarized array antenna disclosed in Patent Document 1.

  An object of the present invention is to provide a circularly polarized array antenna that can suppress an increase in the axial ratio of circularly polarized waves and suppress an increase in side lobes.

The present invention is arranged in a matrix along a virtual plane, and a plurality of pairs of radiating elements that radiate circularly polarized waves that rotate in the same direction when supplied with power,
A feed line forming body that is connected to each radiating element and feeds each radiating element when fed;
The feeder line forming body is:
A pair of radiating elements are provided apart from each other in the extending direction so that circularly polarized waves can be radiated from the pair of radiating elements in the same phase. There are a plurality of line parts that are fed to various positions, and each line part is fed in the same phase,
When the wavelength of the transmission wave in the line portion is represented by λ, a constant that is greater than or equal to 0 and less than half the wavelength of the transmission wave in the line portion is represented by L, and an integer that is greater than or equal to 0 is represented by n.
The distance A from the one end in the extending direction of the line portion to the arrangement position of the radiation element arranged on the one end side is:
A = L + n · (λ / 2) (a)
Or
A = L + {n + (1/2)}. (Λ / 2) (b)
Satisfied,
The distance B from the other end in the extending direction of the line portion to the arrangement position of the radiating element arranged on the other end side is:
B = L + n · (λ / 2) (c)
Or
B = L + {n + (1/2)}. (Λ / 2) (d)
Satisfied,
The plurality of pairs of radiating elements are arranged so as to satisfy the expression (a) or the expression (c) and the first radiating element arranged to satisfy the expression (b) or the expression (d). Two radiating elements,
At least one of the first radiating elements is adjacent to another first radiating element in the row direction and the column direction, or at least one of the second radiating elements is other in the row direction and the column direction. The second radiating element is adjacent to the circularly polarized array antenna.

  According to the present invention, each line portion is provided with a first radiating element and a second radiating element as a pair of radiating elements.

  In the invention, it is preferable that the first radiating element and the second radiating element are arranged in plane symmetry with respect to two symmetry planes orthogonal to the virtual plane and orthogonal to each other.

  According to the present invention, a plurality of pairs of radiating elements are arranged in a matrix along a virtual plane. A feed line forming body is connected to each radiating element. The feed line forming body feeds each radiating element when fed. Each radiating element radiates circularly polarized waves that rotate in the same direction when fed from a feeder line formation.

  The feed line forming body has a plurality of line portions formed to extend. The line portion is provided with a pair of radiating elements spaced apart from each other in the extending direction. The line portion is fed to a position between the pair of radiating elements so that circularly polarized waves can be radiated in the same phase from the pair of radiating elements. Moreover, power is supplied to each line portion with the same phase. Since each line portion is fed in this way, all of the plurality of pairs of radiating elements can radiate circularly polarized waves in the same phase.

  In the line portion, the transmission wave travels from the feeding point of the line portion toward both ends in the extending direction of the line portion (hereinafter referred to as “both ends of the line portion”). The electric power of this transmission wave is supplied to a pair of radiating elements, and thereby circularly polarized waves are radiated from the pair of radiating elements. Part of the transmission wave reaches both ends of the line portion and is reflected at both ends of the line portion. The reflected transmission wave (hereinafter referred to as “reflected wave”) travels toward the feeding point of the line portion. The power of the reflected wave is supplied to the pair of radiating elements, and thereby circularly polarized waves are radiated from the pair of radiating elements.

  The turning direction of the circularly polarized wave by the reflected wave is opposite to the turning direction of the circularly polarized wave by the transmission wave traveling from the feeding point of the line part toward both ends of the line part. The circularly polarized wave (hereinafter referred to as “inverse circularly polarized wave”) due to such a reflected wave increases the axial ratio of the circularly polarized wave radiated from the pair of radiating elements. As a result, the axial ratio of circularly polarized waves increases as a whole.

  Considering such points, in the present invention, the distance A from the one end in the extending direction of the line portion to the arrangement position of the radiation element arranged on the one end side is A = L + n · (λ / 2) Or, it is selected so that A = L + {n + (1/2)} · (λ / 2) is satisfied, and the arrangement position of the radiating element arranged on the other end side from the other end in the extending direction of the line portion Is selected so as to satisfy B = L + n · (λ / 2) or B = L + {n + (1/2)} · (λ / 2). The plurality of pairs of radiating elements include a first radiating element arranged to satisfy A = L + n · (λ / 2) or B = L + n · (λ / 2), and A = L + {n + (1 / 2)} · (λ / 2) or B = L + {n + (1/2)} · (λ / 2). In other words, both the first radiating element and the second radiating element exist as the plurality of pairs of radiating elements.

  Since the first radiating element and the second radiating element are arranged so as to satisfy the above-described formulas, the first radiating element reaches the first radiating element from the feeding point of the line portion via one end or the other end of the line portion. The absolute value of the difference between the distance to the second line and the distance from the feeding point of the line portion to the second radiating element via one end or the other end of the line portion is an odd multiple of λ / 2. Thus, since the absolute value of the difference between the distances is an odd multiple of λ / 2, the phase of the reverse circular polarization radiated from the first radiating element and the reverse circular polarization radiated from the second radiating element. Therefore, the reverse circular polarization radiated from the first radiating element and the reverse circular polarization radiated from the second radiating element cancel each other. Accordingly, as a whole, an increase in the axial ratio of circularly polarized waves can be suppressed.

  When the first radiating element and the second radiating element are alternately arranged at equal intervals in a certain direction, the reverse circular polarization radiated from the first radiating element and the reverse circle radiated from the second radiating element are provided. Polarizations reinforce each other in a specific direction different from the front direction. As a result, the side lobe rises as a whole.

  In view of such points, in the present invention, at least one of the first radiating elements is adjacent to another first radiating element in the row direction and the column direction, or of the second radiating element. At least one of the first radiating element and the second radiating element is arranged so that another second radiating element is adjacent in the row direction and the column direction. By arranging the first radiating element and the second radiating element in this manner, the reverse circular polarization radiated from all the first radiating elements and the reverse radiated from all the second radiating elements are arranged. It is prevented that circularly polarized waves strengthen each other in a specific direction different from the front direction. Therefore, as a whole, the side lobe can be prevented from rising.

  According to the present invention, each line portion is provided with a first radiating element and a second radiating element as a pair of radiating elements. The waves cancel each other. As described above, since the reverse circularly polarized waves cancel each other out for each pair of radiating elements provided in each line portion, the increase in the axial ratio of the circularly polarized waves can be suitably suppressed as a whole.

  According to the invention, the first radiating element and the second radiating element are arranged in plane symmetry with respect to two symmetry planes orthogonal to the virtual plane and orthogonal to each other. In this case, since the arrangement of each radiating element is determined for the other three regions only by designing the arrangement of each radiating element for one of the four regions partitioned by the two symmetry planes, the design is easy. It becomes.

FIG. 1 is a perspective view showing a configuration of a circularly polarized array antenna 11 according to an embodiment of the present invention. In FIG. 1, the circularly polarized array antenna 11 is partially cut away for easy understanding. The circularly polarized array antenna 11 of this embodiment is a wireless LAN (Local Area
In a wireless system such as a network system and a video transmission system, it is used as a transmission antenna and a reception antenna. The circularly polarized array antenna 11 can emit electromagnetic waves and can capture electromagnetic waves. The electromagnetic wave radiated and captured by the circularly polarized array antenna 11 is an electromagnetic wave such as a quasi-millimeter wave band and a millimeter wave band.

  The circularly polarized array antenna 11 is arranged along a virtual plane with the orientations aligned, and is connected to each of the radiating elements 12 and a plurality of pairs of radiating elements 12 that radiate circularly polarized waves that rotate in the same direction when fed. And a feed line forming body 13 (see FIG. 2 to be described later) that feeds the radiation elements 12 in parallel when fed.

  Each radiating element 12 is arranged in a matrix. Here, assuming two directions orthogonal to each other in the virtual plane, one of these two directions is an X direction and the other is a Y direction. In the X direction, which is the row direction, two or more even numbers, in the present embodiment, 16 radiating elements 12 are aligned at intervals, and in the Y direction, which is the column direction, two or more even numbers or odd numbers. In the present embodiment, 16 radiating elements 12 are aligned at intervals. In the present embodiment, when i is an odd number of 1 or more, among the radiating elements 12 aligned in a line in the X direction, the i-th radiating element 12 and the (i + 1) th radiating element from the upstream in the X direction. 12 makes a pair.

  FIG. 2 is a perspective view schematically showing the vicinity of the pair of radiating elements 12. FIG. 3 is a perspective view schematically showing a part of the feeder line forming body 13. In FIG. 3, for easy understanding, the feeder line forming body 13 is partially cut away.

  The radiating element 12 is realized by a dielectric resonator. The radiating element 12 includes a columnar portion 16 formed in a columnar shape and a peripheral wall portion 17 formed in a cylindrical shape along the peripheral surface of the columnar portion 16. The cylindrical portion 16 is made of a dielectric material. The peripheral wall part 17 consists of a conductor, and such a peripheral wall part 17 becomes an electromagnetic wave shield. An opening surface 18 that radiates electromagnetic waves into the space is formed on one end surface in the axial direction of the cylindrical portion 16.

  The feed line forming body 13 is configured by a rectangular waveguide that is a waveguide formed along the feed line. The feed line forming body 13 may be constituted by a hollow waveguide or may be constituted by a dielectric waveguide. In the present embodiment, the feed line forming body 13 is constituted by a dielectric waveguide.

  The feeder line forming body 13 has a plurality of line portions (hereinafter referred to as “first line portions”) 21 that are formed to extend and in which the pair of radiating elements 12 are provided apart in the extending direction C. And a second line portion 22 connected to each first line portion 21. The pair of radiating elements 12 are provided on the first line portion 21 such that the other end surface in the axial direction of the cylindrical portion 16 faces the H surface 23 on one side of the first line portion 21. The second line portion 22 continues to the H surface 24 on the other side of the first line portion 21. The H plane is a plane parallel to the magnetic field of the transmission wave in the first line portion 21.

  The first line portion 21 is fed to a position between the pair of radiating elements 12 so that circularly polarized waves can be radiated from the pair of radiating elements 12 in the same phase. Specifically, a coupling slot 25 is formed between the first line portion 21 and the second line portion 22. The first line portion 21 and the second line portion 22 are coupled by the coupling slot 25. The coupling slot 25 is disposed at a position that is equidistant from each of the pair of radiating elements 12 in the extending direction C of the first line portion 21. Accordingly, the first line portion 21 is fed to a position at equal intervals in each of the pair of radiating elements 12 in the extending direction C. Moreover, the first line portions 21 are fed with the same phase. Since each first line portion 21 is fed in this way, all of the plurality of pairs of radiating elements 12 can radiate circularly polarized waves in the same phase.

  The second line portion 22 is connected to the port portion 26 for supplying power from the outside, and the port portion 26, branched as many times as the number of each first line portion 21, and connected to each first line portion 21. And a branch line portion 27. In the present embodiment, the branch line portion 27 branches seven times, and the power supplied to each radiating element is adjusted by the branch line portion 27. The branch line portion 27 is terminated at a position extending a distance S from the position of the coupling slot 25. This distance S is selected to be about one half of the wavelength of the transmission wave in the branch line portion 27. The branch line portion 27 may be formed of a single layer or a plurality of layers. When the branch line portion 27 is formed of a plurality of layers, the layers are connected by the small holes 28.

  Since the power supplied to each radiating element 12 is adjusted by the branch line portion 27 that is the front stage of the feed line forming body 13 and then supplied to each radiating element 12 by parallel feeding, each radiating element 12 is fed like series feeding. The radiation pattern can be controlled without adjusting the coupling amount for each element. Thus, since adjustment of each coupling amount is unnecessary, the design for controlling the radiation pattern can be facilitated.

  FIG. 4 is a front view showing the vicinity of the radiating element 12. A coupling hole 31 coaxial with the radiating element 12 is formed between the radiating element 12 and the first line portion 21. The radiation element 12 and the first line portion 21 are coupled by the coupling hole 31. The coupling hole 31 is formed in a substantially circular or polygonal shape. In the present embodiment, the coupling hole 31 is formed in a rectangular shape, and is provided so that its longitudinal direction is parallel to the extending direction C of the first line portion 21. The coupling hole 31 is arranged at a position shifted from the central axis L1 of the first line portion 21 when viewed from the front. By arranging the coupling hole 31 in this way, circularly polarized waves can be radiated from the radiating element 12.

  The coupling holes 31 for the pair of radiating elements 12 are shifted to opposite sides with respect to the central axis L1 when viewed from the front. Since the coupling holes 31 are displaced in this way, circularly polarized waves rotating in the same direction can be radiated from the pair of radiating elements 12. In the present embodiment, as shown in FIG. 4, the coupling hole 31 is shifted from the central axis L <b> 1 of the first line portion 21 to the left side with respect to the traveling direction G of the transmission wave in the first line portion 21. Formed.

  FIG. 5 is a diagram showing a magnetic field distribution of a transmission wave in the first line portion 21. FIG. 5 (1) shows a reference state, and FIG. 5 (2) shows a state where the phase is advanced by 90 ° from the reference state. 5 (3) shows a state in which the phase is advanced by 180 ° from the reference state, and FIG. 5 (4) shows a state in which the phase is advanced by 270 ° from the reference state. In FIG. 5, an arrow 32 indicates the magnetic field distribution of the transmission wave in the first line portion 21, and arrows 33 a to 33 d indicate magnetic field components that are coupled to the coupling hole 31.

  In the reference state, as shown in FIG. 5A, the coupling hole 31 is coupled to the magnetic field component 33 a in the direction opposite to the traveling direction G of the transmission wave in the first line portion 21. When the phase advances by 90 ° from the reference state, the coupling hole 31 is coupled to the right magnetic field component 33b with respect to the traveling direction G of the transmission wave in the first line portion 21, as shown in FIG. When the phase advances by 180 ° from the reference state, the coupling hole 31 is coupled to the magnetic field component 33c in the same direction as the traveling direction G of the transmission wave in the first line portion 21, as shown in FIG. When the 270 ° phase advances from the reference state, the coupling hole 31 is coupled to the left magnetic field component 33d with respect to the traveling direction G of the transmission wave in the first line portion 21, as shown in FIG. In this way, the left-handed circularly polarized wave is radiated from the radiating element 12.

  When radiating right-handed circularly polarized light from the radiating element 12, the coupling hole 31 is shifted from the central axis L <b> 1 of the first line portion 21 to the right side with respect to the traveling direction G of the transmission wave in the first line portion 21. That's fine.

  FIG. 6 is a cross-sectional view schematically showing the first line portion 21 and the pair of radiating elements 12 provided in the first line portion 21. In the first line portion 21, the transmission wave is transmitted from the feeding point 36 of the first line portion 21 to both ends in the extending direction C of the first line portion 21 (hereinafter referred to as “both ends of the first line portion 21”) 21 a and 21 b. Proceed toward. The electric power of this transmission wave is supplied to the pair of radiating elements 12, whereby circularly polarized waves are radiated from the pair of radiating elements 12.

  At the same frequency as the center frequency of the circularly polarized array antenna 11 (for example, the center frequency is 61.5 GHz for a circularly polarized array antenna for 61.5 GHz), all the power of the transmission wave is supplied to the pair of radiating elements 12. Therefore, the transmission wave does not reach both ends 21a and 21b of the first line portion 21. However, at a frequency (for example, 63 GHz) shifted from the center frequency of the circularly polarized array antenna 11, a part of the power of the transmission wave is not supplied to the pair of radiating elements. To both ends 21a and 21b.

  A part of the transmission wave reaches both ends 21 a and 21 b of the first line portion 21 and is reflected at both ends 21 a and 21 b of the first line portion 21. The reflected transmission wave (hereinafter referred to as “reflected wave”) travels toward the feeding point 36 of the first line portion 21. The power of the reflected wave is supplied to the pair of radiating elements 12, whereby circularly polarized waves are radiated from the pair of radiating elements 12.

  The turning direction of the circularly polarized wave by the reflected wave is a circularly polarized wave (hereinafter referred to as “circularly polarized wave”) transmitted from the feeding point 36 of the first line portion 21 toward both ends 21a and 21b of the first line portion 21. Is the direction opposite to the turning direction. Due to the circularly polarized wave (hereinafter referred to as “inverse circularly polarized wave”) due to such a reflected wave, the axial ratio of the circularly polarized wave radiated from the pair of radiating elements 12 is increased. As a result, the axial ratio of circularly polarized waves increases as a whole.

In consideration of such points, in the present embodiment, the first distance A from the one end 21a in the extending direction C of the first line portion 21 to the arrangement position of the radiation element 12 arranged on the one end 21a side is The arrangement position of the radiation element 12 that is selected so as to satisfy the following expression (1) or (2) and is arranged on the other end 21b side from the other end 21b in the extending direction C of the first line portion 21. The second distance B is selected so as to satisfy the following formula (3) or formula (4). The plurality of pairs of radiating elements 12 are disposed so as to satisfy the first radiating element disposed so as to satisfy Expression (1) or Expression (3) and satisfying Expression (2) or Expression (4). A second radiating element. In other words, both the first radiating element and the second radiating element exist as the plurality of pairs of radiating elements 12.
A = L + n · (λ / 2) (1)
A = L + {n + (1/2)}. (Λ / 2) (2)
B = L + n · (λ / 2) (3)
B = L + {n + (1/2)}. (Λ / 2) (4)

  In the formulas (1) to (4), the wavelength of the transmission wave in the first line portion 21 is represented by λ, and is a constant that is 0 or more and less than half the wavelength of the transmission wave in the first line portion 21. Is represented by L, and an integer of 0 or more is represented by n.

  Since the first radiating element and the second radiating element are arranged so as to satisfy the above-described formulas, one end 21a or the other end 21b of the first line portion 21 is connected from the feeding point 36 of the first line portion 21. And the distance from the feeding point 36 of the first line portion 21 to the second radiating element via the one end 21a or the other end 21b of the first line portion 21. The absolute value of the difference is an odd multiple of λ / 2. Thus, since the absolute value of the difference between the distances is an odd multiple of λ / 2, the phase of the reverse circular polarization radiated from the first radiating element and the reverse circular polarization radiated from the second radiating element. Therefore, the reverse circular polarization radiated from the first radiating element and the reverse circular polarization radiated from the second radiating element cancel each other. Accordingly, as a whole, an increase in the axial ratio of circularly polarized waves can be suppressed.

  In the present embodiment, the plurality of pairs of radiating elements 12 include the same number of first radiating elements and second radiating elements. Thereby, as a whole, an increase in the axial ratio of circularly polarized waves can be suitably suppressed.

Further, in the present embodiment, each first line portion 21 is provided with a first radiating element and a second radiating element as a pair of radiating elements 12. In this embodiment, the first distance A and the second distance B satisfy the following formula (5). In Expression (5), the wavelength of the transmission wave in the first line portion 21 is represented by λ, and an integer greater than or equal to 0 is represented by n.
| A−B | = λ / 4 + λ · n / 2 (5)

  Since the distances A and B satisfy the expression (5) in this way, the phases of the reverse circularly polarized waves radiated from the pair of radiating elements 12 are shifted by 180 °, thereby the inverse radiated from the pair of radiating elements 12 Circularly polarized waves cancel each other. Therefore, an increase in the axial ratio of circularly polarized waves radiated from the pair of radiating elements 12 can be suitably suppressed. As a result, an increase in the axial ratio of circularly polarized waves can be suitably suppressed as a whole.

  When the first radiating element and the second radiating element are alternately arranged at equal intervals in a certain direction, the reverse circular polarization radiated from the first radiating element and the reverse circle radiated from the second radiating element are provided. Polarizations reinforce each other in a specific direction different from the front direction. As a result, the side lobe rises as a whole.

  In consideration of such a point, in the present embodiment, at least one of the first radiating elements is adjacent to another first radiating element in the X direction and the Y direction, or the second radiating element is used. At least one of the elements is arranged with the first radiating element and the second radiating element such that another second radiating element is adjacent in the X direction and the Y direction. By arranging the first radiating element and the second radiating element in this manner, the reverse circular polarization radiated from all the first radiating elements and the reverse radiated from all the second radiating elements are arranged. It is prevented that circularly polarized waves strengthen each other in a specific direction different from the front direction. Therefore, as a whole, the side lobe can be prevented from rising.

  Table 1 shows an example of the phase shift of the reverse circular polarization radiated from each radiating element 12. In Table 1, the horizontal direction corresponds to the X direction, and the vertical direction corresponds to the Y direction. In Table 1, the phase of the reverse circular polarization radiated from the first radiating element is 0 °, and the phase of the reverse circular polarization radiated from the second radiating element is 180 °. The arrangement relationship of each numerical value representing the phase of the reverse circular polarization corresponds to the arrangement relationship of each radiation element 12. The numerical values 1 to 16 described at the top of Table 1 indicate the order from the upstream in the X direction, and the numerical values 1 to 16 described at the leftmost portion of Table 1 are the order from the downstream in the Y direction. Indicates. The unit of the phase of the reverse circular polarization is “°”.

  FIG. 7 is a front view showing a part of the circularly polarized array antenna 11 shown in FIGS. 1 to 6 in detail. 8 is a cross-sectional view taken along section line S8-S8 in FIG. 7 and 8, a part of them is omitted for convenience. In FIG. 7, in order to facilitate understanding, a part is shown through.

  The circularly polarized wave array antenna 11 is integrally formed and realized as an antenna substrate. Such a circularly polarized array antenna 11 can be easily carried and handled.

  The circularly polarized array antenna 11 is provided with a resonator unit dielectric substrate 42 formed by laminating resonator unit dielectric layers 41, a resonator unit dielectric substrate 42, and a power supply unit dielectric layer 51. And a power feeding part dielectric substrate 52 formed under the power feeding part dielectric substrate 52, and another power feeding part dielectric substrate having a configuration similar to that of the power feeding part dielectric substrate 52. Description of the other power feeding unit dielectric substrate is omitted.

  A resonator unit upper main conductor layer 43 is formed on the upper surface of the resonator unit dielectric substrate 42, and a resonator unit lower main conductor layer 44 is formed on the lower surface of the resonator unit dielectric substrate 42. An opening 45 is formed in the resonator unit upper main conductor layer 43. Around the opening 45, a plurality of resonator portion through conductors 46 that electrically connect the resonator portion upper main conductor layer 43 and the resonator portion lower main conductor layer 44 with a predetermined interval therebetween. Is formed. The resonator part through conductors 46 are arranged at intervals of less than one half of the signal wavelength of the high frequency signal. The interval between the resonator part through conductors 46 is not necessarily a constant value, and may be set by combining various values at less than one half of the signal wavelength. Further, each resonator part through conductor 46 may be provided in double or triple.

  Each resonator section through conductor 46 is provided between the resonator section upper main conductor layer 43 and the resonator section lower main conductor layer 44 in parallel with the main conductor layers 43, 44. Is electrically connected. An opening 48 having a shape similar to the opening 45 is formed in the resonator unit sub-conductor layer 47. The resonator unit sub-conductor layer 47 is formed as a single layer or a plurality of layers as necessary. Each resonator part through conductor 46 and the resonator part sub-conductor layer 47 form an electromagnetic wave shield, which forms the peripheral wall part 17 of the radiating element 12 shown in FIG.

  The resonator part dielectric substrate 42 is surrounded by a resonator part upper main conductor layer 43, a resonator part lower main conductor layer 44, each resonator part through conductor 46, and a resonator part sub-conductor layer 47, and A space filled with the body is formed. Thus, the space filled with the dielectric forms the cylindrical portion 16 of the radiating element 12 shown in FIG.

  A power feeding unit upper main conductor layer 53 is formed on the upper surface of the power feeding unit dielectric substrate 52, and a power feeding unit lower main conductor layer 54 is formed on the lower surface of the power feeding unit dielectric substrate 52. The power feeding unit upper main conductor layer 53 is formed in common with the resonator unit lower main conductor layer 44. An opening 45 is formed in the power supply unit upper main conductor layer 53 (resonator unit lower main conductor layer 44), thereby forming the coupling hole 31.

  Between the power feeding unit upper main conductor layer 53 and the power feeding unit lower main conductor layer 54, a plurality of power feeding unit through conductors 56 that electrically connect the main conductor layers 53 and 54 are arranged in two rows. ing. The respective feed portion through conductors 56 are arranged at an interval of less than half the signal wavelength of the high frequency signal. The interval between the power feed portion through conductors 56 is not necessarily a constant value, and may be set by combining various values with less than half of the signal wavelength. In addition, each of the power feeding portion through conductors 56 may be provided in a double or triple manner.

  Each feed portion through conductor 56 is electrically connected by a feed portion sub-conductor layer 57 provided in parallel with the main conductor layers 53 and 54 between the feed portion upper main conductor layer 53 and the feed portion lower main conductor layer 54. Connected. The power feeding portion sub-conductor layer 57 is formed as a single layer or a plurality of layers as necessary. An electromagnetic wave shielding body is formed by each of the power feeding portion through conductors 56 and the power feeding portion sub-conductor layer 57, and this forms an E-plane conductor of the power feeding line forming body 13 shown in FIGS.

  The power feeding unit dielectric substrate 52 is surrounded by a power feeding unit upper main conductor layer 53, a power feeding unit lower main conductor layer 54, each power feeding unit through conductor 56, and a power feeding unit sub-conductor layer 57, and is filled with a dielectric. A space is formed. Thus, the space filled with the dielectric forms the feed line of the feed line forming body 13 shown in FIGS.

  The resonator unit dielectric substrate 42 and the power feeding unit dielectric substrate 52 are not particularly limited as long as they function as a dielectric and have characteristics that do not hinder the transmission of high-frequency signals. From the viewpoint of accuracy in manufacturing the body 13 and ease of manufacture, it is desirable to be made of ceramics.

  The resonator unit dielectric substrate 42 and the power supply unit dielectric substrate 52 preferably have a relative dielectric constant εr of about 4 to 100, for example, low-temperature fired ceramics such as alumina ceramics, aluminum nitride ceramics, and glass ceramics. In particular, low-temperature fired ceramics such as glass ceramics are preferably used. The resonator part dielectric substrate 42 and the power feeding part dielectric substrate 52 are made into a slurry by adding an appropriate organic solvent and solvent to the ceramic raw material powder, for example, and adopting a doctor blade method, a calendar roll method, or the like. Then, a plurality of ceramic green sheets are obtained by forming into a sheet shape, and these are laminated and fired.

  Further, each resonator section through conductor 46 and each power feeding section through conductor 56 may be formed of a via hole conductor and a through hole conductor. For example, in a through hole made by punching a ceramic green sheet, A conductive paste is embedded and then fired simultaneously with the ceramic green sheet. Each resonator part through conductor 46 and each power supply part through conductor 56 have an appropriate diameter of 50 to 300 μm.

  FIG. 9 is a front view showing a configuration of a circularly polarized array antenna 61 according to another embodiment of the present invention. FIG. 10 is a front view showing the circularly polarized array antenna 61 with a part cut away. FIG. 11 is a perspective view schematically showing the vicinity of the pair of radiating elements 12. Since the circularly polarized wave array antenna 61 of the present embodiment is similar to the circularly polarized wave array antenna 11 shown in FIGS. 1 to 8, the corresponding parts are denoted by the same reference numerals, the description thereof is omitted, and different points are provided. I will explain only.

  In the present embodiment, the radiating elements 12 provided in the first line portions 21 adjacent in the extending direction C of the first line portions 21 are aligned on a straight line. As in the circularly polarized wave array antenna 11 shown in FIGS. 1 to 8, the radiating elements 12 provided in the first line portions 21 adjacent in the extending direction C of the first line portions 21 are arranged in a so-called zigzag pattern. With such a configuration, unnecessary electromagnetic waves are radiated in the direction in which the distance between the radiating elements 12 is large. In consideration of this point, in the present embodiment, by arranging the radiating elements 12 in a straight line, the difference in the interval between the radiating elements 12 is reduced, thereby suppressing the emission of unnecessary electromagnetic waves. be able to.

  In the X direction, the radiating elements 12 are aligned at equal intervals, and in the Y direction, the radiating elements 12 are aligned at equal intervals. The interval D1 between the radiating elements 12 aligned in the X direction and the interval D2 between the radiating elements 12 aligned in the Y direction may be the same or different, but suppress the emission of unnecessary electromagnetic waves. From the point of view of doing, it is preferable that they are the same.

  In the present embodiment, the first radiating element and the second radiating element are arranged in plane symmetry with respect to two symmetry planes orthogonal to the virtual plane and orthogonal to each other. In this case, since the arrangement of each radiating element 12 is determined for the other three regions only by designing the arrangement of each radiating element 12 for one of the four regions partitioned by the two symmetry planes. Becomes easy.

  In FIG. 9, there are places where the adjacent first line portions 21 appear to communicate with each other, but in actuality, they are blocked by the walls of the via-hole conductors as shown in FIG.

  FIG. 12 is a front view schematically showing the vicinity of a pair of radiating elements 12 provided in a circularly polarized array antenna which is still another embodiment of the present invention. Since the circularly polarized wave array antenna of the present embodiment is similar to the circularly polarized wave array antenna 61 shown in FIGS. 9 to 11, the corresponding parts are denoted by the same reference numerals, description thereof is omitted, and different points are described. Just explain.

  In the present embodiment, the coupling slot 25 is formed such that the longitudinal direction E thereof is orthogonal to the extending direction C of the first line portion 21 in the virtual one plane. By forming the coupling slot 25 in this way, the power reflected by the coupling slot 25 when power is supplied to the first line portion 21 via the coupling slot 25 can be reduced. Therefore, power loss can be suppressed, and thereby the power supply efficiency to the first line portion 21 can be improved.

  FIG. 13 is a perspective view schematically showing the vicinity of a pair of radiating elements provided in a circularly polarized array antenna which is still another embodiment of the present invention. FIG. 14 is a front view schematically showing the vicinity of the pair of radiating elements shown in FIG. Since this embodiment is similar to each of the above-described embodiments, only different points will be described. The circularly polarized array antenna of the present embodiment is realized by a patch antenna.

  The circularly polarized array antenna includes a first dielectric layer 72, a second dielectric layer 73, a ground conductor layer 74 interposed between the first dielectric layer 72 and the second dielectric layer 73, A first strip conductor layer 75 formed on the opposite side of the first dielectric layer 72 from the ground conductor layer 74 and a first strip conductor formed on the opposite side of the first dielectric layer 72 from the ground conductor layer 74. A pair of patch conductor layers 76a and 76b electrically connected to the layer 75; a second strip conductor layer 77 formed on the opposite side of the second dielectric layer 73 from the ground conductor layer 74; It includes a connecting portion 78 that penetrates through the two dielectric layers 72 and 73 and the ground conductor layer 74 and electrically connects the first strip conductor layer 75 and the second strip conductor layer 77.

  In the present embodiment, a pair of radiating elements is configured by the pair of patch conductor layers 76a and 76b. Further, the first strip conductor layer 75, the first dielectric layer 72, and the ground conductor layer 74 constitute a first line portion. A second line portion is configured including the second strip conductor layer 77, the second dielectric layer 73, and the ground conductor layer 74.

  The first strip conductor layer 75 is formed to extend. One of the pair of patch conductor layers 76a and 76b is connected to the first strip conductor layer 75 near the one end 75a in the extending direction C1 of the first strip conductor layer 75, and the width direction C2 of the first strip conductor layer 75 is connected. Arranged on one side, the other 76b is connected to the first strip conductor layer 75 near the other end 75b in the extending direction C1 of the first strip conductor layer 75, and arranged on the other side in the width direction C2 of the first strip conductor layer 75. Is done. Each patch conductor layer 76a, 76b is formed in a square shape. One patch conductor layer 76a is fed with a phase difference of 90 ° with respect to each center of its two adjacent edges, and the other patch conductor layer 76b is 90 ° with respect to each center of its two adjacent edges. Power is supplied with a phase difference. In addition, in order to match the phase of the circularly polarized waves radiated from the pair of patch conductor layers 76a and 76b, the position of the connecting portion 78 is set to one patch conductor layer by λ / 4 from the center between the patch conductor layers 76a and 76b. The distance from the connecting portion 78 to the other patch conductor layer 76b is longer by λ / 2 than the distance from the connecting portion 78 to one patch conductor layer 76a.

  FIG. 15 is a perspective view schematically showing the vicinity of a pair of radiating elements provided in a circularly polarized array antenna which is still another embodiment of the present invention. FIG. 16 is a front view schematically showing the vicinity of the pair of radiating elements shown in FIG. Since this embodiment is similar to each of the above-described embodiments, only different points will be described. The circularly polarized array antenna of the present embodiment is realized by a slot antenna.

  The circularly polarized wave array antenna includes a first dielectric layer 81, a second dielectric layer 82, and a first strip conductor layer 83 interposed between the first dielectric layer 81 and the second dielectric layer 82. And a first ground conductor layer 84 formed on the opposite side of the first dielectric layer 81 from the first strip conductor layer 83 and a second dielectric layer 82 formed on the opposite side of the first strip conductor layer 83. The second ground conductor layer 85, the third dielectric layer 86 formed on the opposite side of the second ground conductor layer 85 from the second dielectric layer 82, and the second ground conductor of the third dielectric layer 86. A second strip conductor layer 87 formed on the opposite side of layer 85.

  In the present embodiment, the first strip conductor layer 83, the first and second dielectric layers 81 and 82, and the first and second ground conductor layers 84 and 85 constitute a first line portion. A second line portion is configured including the second strip conductor layer 87, the third dielectric layer 86, and the second ground conductor layer 85.

  The first strip conductor layer 83 is formed to extend. In the first ground conductor layer 84, slots 88a and 88b are formed near the one end 83a and the other end 83b in the extending direction C3 of the first strip conductor layer 83. In the present embodiment, a pair of radiating elements is formed by the portions 89a and 89b where the slots 88a and 88b of the first ground conductor layer 84 are formed. A coupling slot 90 is formed in the second ground conductor layer 84. The coupling line 90 couples the first line portion and the second line portion.

  In order to evaluate the axial ratio characteristics, simulation was performed by a high frequency structure simulator (HFSS) method. In this simulation, a pair of radiating elements 12 is assumed with the same configuration as the circularly polarized array antenna 11 shown in FIGS.

Specifically, the dielectric constant εr of the dielectric of the feed line forming body 13 is 4.9. The dimensions of the feeder line forming body 13 were as follows.
・ Width dimension of first line portion 21 = 1.89 mm
・ Width dimension of the second line portion 22 = 1.64 mm
・ Thickness dimension of the first and second line portions 21 and 22 = 0.6 mm
-Via diameter dimension of feed section through conductor 56 = 0.2 mm
・ Via spacing of feed through conductor 56 = 0.60 mm

  Note that both end faces and both side faces of the first and second line portions 21 and 22 are indicated by via centers. The wavelength λ of the transmission wave at this time was 2.64 mm, and the center frequency was about 61.5 GHz.

The dimensions of the radiating element 12 were as follows.
・ Dimension of coupling hole 31 = 1.12 × 1.06 mm
・ Diameter size of the opening surface 18 = 1.47 mm
-Thickness dimension of the radiating element 12 = 0.6 mm
-Diameter dimension of the radiating element 12 = 1.61 mm

  The first distance A and the second distance B were variously changed. As a result, the first and second distances A when the phase of the reverse circularly polarized wave radiated from the radiation element 12 in the front direction by the transmission waves reflected at both ends 21a and 21b of the first line portion 21 becomes 0 °. , B was 1.32 mm. For convenience of explanation, of the pair of radiating elements 12, the radiating element 12 disposed on the one end 21 a side of the first line portion 21 is referred to as one radiating element 12 a and is disposed on the other end 21 b side of the first line portion 21. The radiating element 12 may be referred to as the other radiating element 12b.

  Table 2 shows the relationship between the phase of the reverse circular polarization from one radiating element 12a and the first distance A, and the relationship between the phase of the reverse circular polarization from the other radiating element 12b and the second distance B.

  Table 3 shows the phase difference between the phase of the reverse circular polarization from one radiating element 12a and the phase of the reverse circular polarization from the other radiating element 12b, and the first distance A and the second distance B. The relationship with the difference between them is shown.

  In Tables 2 and 3, the phase difference between the phase of the reverse circular polarization from one radiating element 12a and the phase of the reverse circular polarization from the other radiating element 12b was set to 0 ° to 180 °. . In Table 3, the difference | A−B | between the first distance A and the second distance B is expressed using the wavelength λ of the transmission wave. The phase difference is a phase difference in the front direction.

Here, when the center frequency is 61.5 GHz, when the axial ratio is 4.00 dB or less, the circularly polarized array antenna characteristics of the present invention are satisfied, and when the center frequency is 62.0 GHz, ± 2.0 GHz, It is assumed that the circularly polarized array antenna characteristics of the present invention are satisfied when the axial ratio is 10.00 dB or less.
Table 4 shows the simulation results.

  From the results of Table 4, the first distance A and the second distance B are 45 ° in phase difference between the reverse circular polarization from one radiating element 12a and the reverse circular polarization from the other radiating element 12b. It can be seen that the axial ratio satisfies 4.00 dB or less when the difference between is and λ / 16 (0.17 mm). Further, the phase difference between the reverse circular polarization from one radiating element 12a and the reverse circular polarization from the other radiating element 12b is 180 °, and the difference between the first distance A and the second distance B is It can be seen that the axial ratio can be reduced most when λ / 4 (6.6 mm).

  Based on this result, the relationship between the first distance A and the second distance B is preferably λ / 4 ≧ | A−B | ≧ λ / 16, where the axial ratio is 4.00 dB or less. The relationship between the first distance A and the second distance B is most preferably | A−B | = λ / 4, which can most reduce the axial ratio.

  Table 5 shows the simulation result of the frequency characteristics with respect to the axial ratio at ± 2.0 GHz centering on the frequency of 61.5 GHz.

  From the result of Table 5, the phase difference between the reverse circular polarization from one radiating element 12a and the reverse circular polarization from the other radiating element 12b is 135 °, and the first distance A and the second distance B Is λ / 5.33 (0.49 mm), the axial frequency is less than 4.00 dB at a center frequency of 61.5 GHz, and the axis is within a range of ± 2.0 GHz centered on 61.5 GHz. It can be seen that the ratio satisfies 10.00 dB or less.

  The phase difference between the reverse circular polarization from one radiating element 12a and the reverse circular polarization from the other radiating element 12b is 170 °, and the difference between the first distance A and the second distance B is Is λ / 4.24 (0.62 mm), it can be seen that the axial ratio satisfies 4.00 dB or less in the range of ± 2.0 GHz centering on the frequency of 61.5 GHz.

  Further, the phase difference between the reverse circular polarization from one radiating element 12a and the reverse circular polarization from the other radiating element 12b is 180 °, and the difference between the first distance A and the second distance B is Is λ / 4 (0.66 mm), it can be seen that the axial ratio satisfies 4.00 dB or less in the range of ± 2.0 GHz centering on the frequency of 61.5 GHz, and the axial ratio is most reduced.

  Based on this result, the relationship between the first distance A and the second distance B is λ / 4 ≧ | A−B | ≧ λ / 5., Where the axial ratio is 10.00 dB or less in the 4.0 GHz band. 33 is desirable. The relationship between the first distance A and the second distance B is more preferably λ / 4 ≧ | A−B | ≧ λ / 4.24 in which the axial ratio is 4.00 dB or less in the 4.0 GHz band. The relationship between the first distance A and the second distance B is most desirably | A−B | = λ / 4, in which the axial ratio is 4.00 dB or less in the 4.0 GHz band and can be reduced most.

  Although the first distance A is 1.32 mm, the difference between the first distance A and the second distance B is important, and the same tendency was observed even if the value of the first distance A was changed.

  Hereinafter, the side lobe rise other than the front direction due to the radiation of the reverse circular polarization will be described based on the simulation result of the radiation pattern.

  In this simulation, the same configuration as the circularly polarized array antenna 61 shown in FIGS. 9 to 11 is assumed. Specifically, a circular polarization array antenna 61 for 61.5 GHz is assumed. The distance D1 between the radiating elements 12 aligned in the X direction was 3.72 mm, and the distance D2 between the radiating elements 12 aligned in the Y direction was 3.97 mm.

  Further, the electric field intensity of the circularly polarized wave radiated from each radiating element 12 is as shown in Table 6. In Table 6, the horizontal direction corresponds to the X direction, and the vertical direction corresponds to the Y direction. The arrangement relationship of the numerical values representing the electric field intensity corresponds to the arrangement relationship of the radiating elements 12. Numerical values 1 to 16 described at the top of Table 6 indicate the order from the upstream in the X direction, and numerical values 1 to 16 described at the leftmost part of Table 6 are the order from the downstream in the Y direction. Indicates. In Table 6, the electric field intensity of the circularly polarized wave radiated from each radiating element 12 is expressed as a ratio.

  FIG. 17 is a diagram for explaining FIGS. 18, 20, and 22 to 25. Referring also to FIG. 9, the center of the circularly polarized array antenna 61 in the virtual plane is the origin O, the axis extending from the origin O in the X direction is the X axis, and the axis extending in the Y direction is the Y axis. The axis extending in the front direction of the circularly polarized array antenna 61 was taken as the Z axis. Here, a unit vector V expressed by polar coordinates (θ, φ) starting from the origin O is assumed. This unit vector V represents the radiation direction. 18, 20, and 22 to 25, the X coordinate and the Y coordinate of the end point of the unit vector V are associated with the electric field strength in the radial direction represented by the unit vector V. The electric field strength in each radiation direction is displayed in decibels based on the electric field strength in the front direction.

  In this simulation, simulation was performed for Comparative Examples 1 to 4 and Examples 1 to 8 described later. Since Tables 7 to 9 showing Comparative Examples 2 to 4 and Tables 10 to 16 showing Examples 1, 2, 4 to 8 are similar to Table 1, description of Tables 7 to 16 is omitted. To do.

  First, in the case of the frequency of 61.5 GHz, verification was performed as Comparative Example 1 in the case where only the circularly polarized wave was radiated without radiating the reversely polarized wave. FIG. 18 is a diagram illustrating a simulation result of Comparative Example 1. FIG. 19 is a diagram showing only the radiation direction in a virtual plane including the X axis and the Z axis in the simulation results of Comparative Example 1. In FIG. 19, the horizontal axis indicates the radiation direction in a virtual plane, and the vertical axis indicates the electric field strength. As for the radiation direction, the front direction is 0 °, and the case where the front direction is inclined to the downstream side in the X direction is positive, and the case where the front direction is inclined to the upstream side in the X direction is negative. The electric field strength is displayed in decibels based on the electric field strength in the front direction. In Comparative Example 1, the side lobe level was −25.8 dB.

  Next, in the case of a frequency of 63 GHz, that is, when reverse circular polarization is radiated, specifically, the electric field intensity of reverse circular polarization radiated from each radiating element 12 is a circularly polarized wave radiated from each radiating element 12. About the case where it is 20% of the electric field strength of a wave, it verified as Comparative Examples 2-4 and Examples 1-8.

First, Comparative Examples 2 to 4 will be described in order.
FIG. 20 is a diagram illustrating a simulation result of Comparative Example 2. FIG. 21 is a diagram illustrating only the radiation direction in a virtual plane including the X axis and the Z axis among the simulation results of Comparative Example 2. Since FIG. 21 is similar to FIG. 19, the description of FIG. 21 is omitted. In Comparative Example 2, it was assumed that reverse circularly polarized waves were radiated from each radiating element 12 with a phase difference as shown in Table 7. In other words, in the second comparative example, the first radiating element rows arranged in the Y direction and the second radiating element rows arranged in the Y direction are alternately arranged in the X direction. In Comparative Example 2 like this, as shown in FIGS. 20 and 21, side lobes that are prominent in a direction other than the front direction, specifically, in a specific radial direction within a virtual plane including the X-axis and the Z-axis. Is seen. In Comparative Example 2, the side lobe level was -14.0 dB. In Comparative Example 2, it can be seen that the side lobe rises due to the reverse circular polarization radiated from each radiating element 12.

  FIG. 22 is a diagram illustrating a simulation result of Comparative Example 3. In Comparative Example 3, it was assumed that reverse circularly polarized waves were radiated from each radiating element 12 with a phase difference as shown in Table 8. In other words, in Comparative Example 3, the first radiating element rows arranged in the X direction and the second radiating element rows arranged in the X direction are alternately arranged in the Y direction. In Comparative Example 3 like this, as shown in FIG. 22, significant side lobes are seen in directions other than the front direction, specifically in a specific radial direction within a virtual plane including the Y axis and the Z axis. . In Comparative Example 3, the side lobe level was -14.0 dB. Also in Comparative Example 3, it can be seen that the side lobe rises due to the reverse circular polarization radiated from each radiating element 12.

  FIG. 23 is a diagram illustrating a simulation result of Comparative Example 4. In Comparative Example 4, it is assumed that the reverse circularly polarized wave is radiated from each radiating element 12 with a phase difference as shown in Table 9. In other words, in Comparative Example 4, the first radiating element and the second radiating element are alternately arranged in the X direction and alternately in the Y direction. In such a comparative example 4, as shown in FIG. 23, a remarkable side lobe is seen in a specific radiation direction other than the front direction. In Comparative Example 4, the side lobe level was -14.0 dB. Also in Comparative Example 4, it can be seen that the side lobe rises due to the reverse circular polarization radiated from each radiating element 12.

  Next, with respect to these Comparative Examples 2 to 4, Examples 1 to 8 of the present invention in which the rise of the side lobe is suppressed will be described in order.

  FIG. 24 is a diagram illustrating a simulation result of the first embodiment. In the first embodiment, it is assumed that reverse circular polarized waves are radiated from the radiating elements 12 with a phase difference as shown in Table 10. In other words, in the first embodiment, the eleventh radiating element 12 from the upstream in the X direction is replaced with the second radiating element from the first radiating element to the twelfth from the upstream in the X direction. The radiating element 12 was changed from the second radiating element to the first radiating element. In the first embodiment, at least one of the first radiating elements is adjacent to another first radiating element in the X direction and the Y direction, or at least one of the second radiating elements is in the X direction. And the condition that another second radiating element is adjacent in the Y direction is satisfied. In Example 1, the side lobe level was −17.2 dB, which was reduced by about 3 dB compared to Comparative Example 2. In such Example 1, it turns out that the raise of a side lobe is suppressed as shown in FIG.

  FIG. 25 is a diagram illustrating a simulation result of the second embodiment. In Example 2, it was assumed that reverse circularly polarized waves were radiated from each radiating element 12 with a phase difference as shown in Table 11. In other words, in Example 2, each of the radiating elements 12 that is 1st to 8th from the upstream in the X direction and 9th to 16th from the downstream in the Y direction, compared to Comparative Example 4, In the case of the first radiating element, the second radiating element is used instead of the radiating elements which are ninth to sixteenth from the upstream in the X direction and first to eighth from the downstream in the Y direction. In the case of the element, it was replaced with the first radiating element. In the second embodiment, at least one of the first radiating elements is adjacent to another first radiating element in the X direction and the Y direction, or at least one of the second radiating elements is in the X direction. And the condition that another second radiating element is adjacent in the Y direction is satisfied. In Example 2, the side lobe level was −21.3 dB, which was reduced by about 7 dB compared to Comparative Example 4. In Example 2 as described above, it can be seen that the rise of the side lobe is suppressed as shown in FIG.

  In Example 3, it was assumed that reverse circularly polarized waves were radiated from each radiating element 12 with a phase difference as shown in Table 1 above. In the third embodiment, at least one of the first radiating elements is adjacent to another first radiating element in the X direction and the Y direction, or at least one of the second radiating elements is in the X direction. And the condition that another second radiating element is adjacent in the Y direction is satisfied. In Example 3 as described above, the side lobe level was −25.8 dB, which was reduced by about 12 dB compared to Comparative Example 4. In such Example 3, it turns out that the raise of a side lobe is suppressed suitably.

  In Examples 4-8, it was assumed that reverse circularly polarized waves were radiated from each radiating element 12 with the phase differences as shown in Tables 12-16. In Examples 4 to 8, at least one of the first radiating elements is adjacent to another first radiating element in the X direction and the Y direction, or at least one of the second radiating elements is The condition that another second radiating element is adjacent in the X direction and the Y direction is satisfied.

  The side lobe level is -16.3 dB in Example 4, -16.5 dB in Example 5, -16.7 dB in Example 6, -16.9 dB in Example 7, and In Example 8, it was -16.4 dB. From these results, it can be seen that also in Examples 4 to 8, the rise in side lobe is suppressed compared to Comparative Examples 2 to 4.

  Further, a radiation pattern was simulated assuming an 8 × 8 circularly polarized array antenna as a configuration similar to the circularly polarized array antenna 61 shown in FIGS. In this simulation, it is assumed that eight radiating elements are aligned at intervals in the X direction and eight radiating elements are aligned at intervals in the Y direction. The electric field intensity of the electromagnetic wave radiated from each radiating element is as shown in Table 17, and the electric field intensity of the reverse circular polarization radiated from each radiating element is 20% of the electric field intensity of the circularly polarized wave radiated from each radiating element. It was said that. Since Table 17 is similar to Table 6, the description of Table 17 is omitted. Others were the same as in the above simulation.

  In this simulation, simulation was performed for Comparative Examples 5 to 7 and Examples 9 to 17 described later. 26 to 28 showing the simulation results of Comparative Examples 5 to 7 are similar to FIG. 18 and the like described above, and thus description of these FIGS. 26 to 28 is omitted. Since Tables 18 to 20 showing Comparative Examples 5 to 7 and Tables 21 to 29 showing Examples 9 to 17 are similar to Table 1, description of Tables 18 to 29 is omitted.

  FIG. 26 is a diagram illustrating a simulation result of Comparative Example 5. In Comparative Example 5, it is assumed that the reverse circularly polarized wave is radiated from each radiating element with a phase difference as shown in Table 18. In Comparative Example 5 like this, as shown in FIG. 26, significant side lobes are seen in directions other than the front direction. In Comparative Example 5, the side lobe level was -14.0 dB.

  FIG. 27 is a diagram illustrating a simulation result of Comparative Example 6. In Comparative Example 6, it is assumed that the reverse circular polarized wave is radiated from each radiating element with a phase difference as shown in Table 19. In Comparative Example 6 as described above, as shown in FIG. 27, significant side lobes are seen in directions other than the front direction. In Comparative Example 6, the side lobe level was -14.0 dB.

  FIG. 28 is a diagram illustrating a simulation result of Comparative Example 7. In Comparative Example 7, it is assumed that the reverse circular polarized wave is radiated from each radiating element with a phase difference as shown in Table 20. In such a comparative example 7, as shown in FIG. 28, a remarkable side lobe is seen in directions other than the front direction. In Comparative Example 7, the side lobe level was -14.0 dB.

  In Examples 9 to 17, it was assumed that reverse circularly polarized waves were radiated from each radiating element with a phase difference as shown in Tables 21 to 29. In Examples 9 to 17, at least one of the first radiating elements is adjacent to another first radiating element in the X direction and the Y direction, or at least one of the second radiating elements is The condition that another second radiating element is adjacent in the X direction and the Y direction is satisfied.

  The side lobe level is -14.3 dB in Example 9, -14.7 dB in Example 10, -14.9 dB in Example 11, and -14.9 dB in Example 12. Example 13 is -15.3 dB, Example 14 is -15.1 dB, Example 15 is -16.9 dB, Example 16 is -17.7 dB, and Example 17 is -15. It was 6 dB. From these results, it can be seen that in Examples 9 to 17, an increase in side lobe is suppressed as compared with Comparative Examples 5 to 7.

It is a perspective view which shows the structure of the circularly polarized wave array antenna 11 which is one Embodiment of this invention. 3 is a perspective view schematically showing the vicinity of a pair of radiating elements 12. FIG. 3 is a perspective view schematically showing a part of a feeder line forming body 13. FIG. 3 is a front view showing the vicinity of a radiating element 12. FIG. 3 is a diagram showing a magnetic field distribution of a transmission wave in the first line portion 21. FIG. 2 is a cross-sectional view schematically showing a first line portion 21 and a pair of radiating elements 12 provided in the first line portion 21. FIG. It is a front view which shows a part of circularly polarized array antenna 11 shown in FIGS. 1-6 in detail. It is sectional drawing seen from cut surface line S8-S8 of FIG. It is a front view which shows the structure of the circularly polarized array antenna 61 which is the other form of implementation of this invention. FIG. 3 is a front view showing a partially polarized circular array antenna 61. 3 is a perspective view schematically showing the vicinity of a pair of radiating elements 12. FIG. It is a front view which shows typically the vicinity of a pair of radiation element 12 with which the circularly polarized array antenna which is further another form of implementation of this invention is provided. It is a perspective view which shows typically the vicinity of a pair of radiation element with which the circularly polarized array antenna which is further another form of implementation of this invention is provided. It is a front view which shows typically the vicinity of a pair of radiation | emission element shown in FIG. It is a perspective view which shows typically the vicinity of a pair of radiation element with which the circularly polarized array antenna which is further another form of implementation of this invention is provided. FIG. 16 is a front view schematically showing the vicinity of a pair of radiating elements shown in FIG. 15. FIG. 26 is a diagram for explaining FIGS. 18, 20, and 22 to 25. It is a figure which shows the simulation result of the comparative example 1. It is a figure shown only about the radiation | emission direction in the virtual one plane containing the X-axis and Z-axis among the simulation results of the comparative example 1. FIG. It is a figure which shows the simulation result of the comparative example 2.

It is a figure shown only about the radiation direction in the virtual one plane containing the X-axis and Z-axis among the simulation results of the comparative example 2. It is a figure which shows the simulation result of the comparative example 3. It is a figure which shows the simulation result of the comparative example 4. It is a figure which shows the simulation result of Example 1. FIG. It is a figure which shows the simulation result of Example 2. FIG. It is a figure which shows the simulation result of the comparative example 5. It is a figure which shows the simulation result of the comparative example 6. It is a figure which shows the simulation result of the comparative example 7.

Explanation of symbols

DESCRIPTION OF SYMBOLS 11,61 Circularly polarized array antenna 12 Radiation element 13 Feed line formation body 21 1st line part 22 2nd line part

Claims (3)

A plurality of pairs of radiating elements that radiate circularly polarized waves that are arranged in a matrix along a virtual plane and that rotate in the same direction when fed.
A feed line forming body that is connected to each radiating element and feeds each radiating element when fed;
The feeder line forming body is:
A pair of radiating elements are provided apart from each other in the extending direction so that circularly polarized waves can be radiated from the pair of radiating elements in the same phase. There are a plurality of line parts that are fed to various positions, and each line part is fed in the same phase,
When the wavelength of the transmission wave in the line portion is represented by λ, a constant that is greater than or equal to 0 and less than half the wavelength of the transmission wave in the line portion is represented by L, and an integer that is greater than or equal to 0 is represented by n.
The distance A from the one end in the extending direction of the line portion to the arrangement position of the radiation element arranged on the one end side is:
A = L + n · (λ / 2) (a)
Or
A = L + {n + (1/2)}. (Λ / 2) (b)
Satisfied,
The distance B from the other end in the extending direction of the line portion to the arrangement position of the radiating element arranged on the other end side is:
B = L + n · (λ / 2) (c)
Or
B = L + {n + (1/2)}. (Λ / 2) (d)
Satisfied,
The plurality of pairs of radiating elements are arranged so as to satisfy the expression (a) or the expression (c) and the first radiating element arranged to satisfy the expression (b) or the expression (d). Two radiating elements,
At least one of the first radiating elements is adjacent to another first radiating element in the row direction and the column direction, or at least one of the second radiating elements is other in the row direction and the column direction. A circularly polarized array antenna, wherein the second radiating elements are adjacent to each other.   The circularly polarized array antenna according to claim 1, wherein each line portion is provided with a first radiating element and a second radiating element as a pair of radiating elements.   3. The circularly polarized wave according to claim 1, wherein the first radiating element and the second radiating element are arranged in plane symmetry with respect to two symmetry planes orthogonal to the virtual plane and orthogonal to each other. Wave array antenna.

Images