JP4677366B2 - State detection device in motor control device - Google Patents

Description

  The present invention relates to a state detection device that detects a state of a motor in a motor control device that performs PWM control of the motor.

  In many cases, a motor control device is equipped with a state detection device that detects a motor state (for example, position, speed, magnetic pole, temperature, etc.). This state detection device is disposed on the motor side and is connected to the control circuit via a cable.

By the way, when the motor control device is configured to PWM control the winding current of the motor, PWM switching noise is generated. This switching noise is often superimposed on the output signal of the state detection device via the motor body or cable. In this case, it is difficult to accurately detect the motor state from this output signal. In particular, when the signal cable and the power cable are arranged close to each other and the lengths of these cables are large, the switching noise induced in the signal cable becomes large. The detection accuracy will be further reduced.
Therefore, means for reducing the influence of the switching noise has been proposed (for example, Patent Documents 1 to 3).

JP 2001-8482 A JP 2004-364460 A JP 2003-153696 A

  However, with the technique described in Patent Document 1, the frequency (several KHz to several tens KHz) of the PWM carrier signal near the top of the PWM triangular wave where noise is difficult to enter, with respect to the detection signal (AM modulated wave signal) that changes at high speed. A / D conversion is performed in synchronization with (1), so that a high-speed A / D converter is required. In the technique disclosed in Patent Document 1, in principle, the excitation carrier frequency of the resolver is limited to the frequency of the PWM carrier signal or a multiple of the frequency.

  On the other hand, the technique described in Patent Document 2 is to make the sampling period of A / D conversion for the detection signal of the resolver coincide with the PWM switching period so that the influence of switching noise in the A / D conversion result is always constant. is there. According to this technique, the influence of the switching noise on the rotation angle of the motor calculated from the A / D conversion result is theoretically canceled. However, when the motor rotates, the timing of PWM switching with respect to the PWM carrier changes, so that it becomes practically difficult to cancel the influence of the switching noise.

  In the technique according to Patent Document 3, it is necessary to additionally use a common mode coil. Since components such as coils are difficult to downsize, a large mounting area must be secured in order to use the common mode coil. This hinders the miniaturization of the motor control device. In addition, since the technique according to Patent Document 3 is based on the premise that the signal is differentially transmitted, it also has a drawback such as an increase in the number of signal cables.

  The present invention has been made in view of such conventional problems, and its purpose is to provide PWM switching noise for motor state detection without using a high-speed signal processing function or an element requiring a large mounting area. An object of the present invention is to provide a state detection device in a motor control device capable of reliably removing the influence of the above.

The present invention provides a state detection device for detecting a state of the motor in a motor control device that performs PWM control of the motor, and generates an analog state signal corresponding to the state of the motor, and the analog state signal Signal generating means for generating an amplitude modulated signal obtained by amplitude modulating a carrier signal; demodulating means for synchronously demodulating the analog state signal from the amplitude modulated signal using the first carrier signal; and the synchronously demodulated analog state Smoothing means for smoothing the PWM switching noise component included in the signal, and the frequency f _am of the first carrier signal and the frequency f _pwm of the second carrier signal used for the PWM control are The above object is achieved by setting so as to have the following relationship.
When f _am > f _pwm ,
f _am = f _pwm × (n + 0.5)
When f _am <f _pwm ,
f _am = f _pwm / (2n)
Where n is a positive integer

  The signal generating means may include an angle sensor that generates a signal corresponding to the rotation angle of the rotor of the motor as the analog state signal. In that case, for example, a resolver or an MR sensor is used as the angle sensor. The signal generating means may include a temperature sensor that generates a signal corresponding to the temperature of the motor as the analog state signal.

According to the present invention, the relationship between the frequency f _am of the first carrier signal that is amplitude-modulated with the analog state signal and the frequency f _pwm of the second carrier signal used for PWM control of the motor is _expressed as PWM switching noise. Since it is set so as to be canceled, the influence of PWM switching noise on motor state detection can be reliably removed without using a high-speed signal processing function or an element requiring a large mounting area.

FIG. 1 is a block diagram of a motor control device including a state detection device according to an embodiment of the present invention.
In FIG. 1, the resolver 10 detects the rotation angle of the rotor of the motor 20, and the control circuit 30 controls the current flowing through the windings of the motor 20 based on the rotation angle of the rotor.
The resolver 10 includes an excitation winding 11 and detection windings 12 and 13, and is fastened to the rotor shaft of the motor 20.
The control circuit 30 includes an oscillation circuit 31, a frequency dividing circuit 32 that divides the frequency of the clock signal output from the oscillation circuit 31, and the excitation winding 11 of the resolver 10 at the frequency output from the frequency dividing circuit 32. And an excitation circuit 33 for exciting.

In this embodiment, since the oscillation frequency of the oscillation circuit 31 is set to 100 KHz, the excitation frequency of the excitation winding 11 of the resolver 10 is 50 KHz obtained by dividing 100 KHz by two. Assuming that the output voltage of the excitation circuit 33, that is, the applied voltage of the excitation winding 11, is F (t), the output voltages V _a and V _b of the detection windings 12 and 13 of the resolver 10 are respectively expressed as follows: Is done.
V _a = K · sin θ · F (t)
V _b = K · cos θ · F (t)
Where K: proportional constant, θ: rotor rotation angle

The applied voltage F (t) of the excitation winding 11 is a voltage defined as a function of time. A sine wave voltage is generally used as the applied voltage F (t). However, a triangular wave voltage or a square wave voltage can be used as the voltage F (t), and in this embodiment, a square wave voltage as shown in FIG. 2A is used.
When the excitation winding 11 of the resolver 10 is excited with the square wave voltage F (t), for example, the voltage V _a as shown in FIG. 2B from the detection winding 12, that is, the voltage F that is an excitation carrier signal. A voltage V _{a obtained} by amplitude-modulating (t) with the sin phase modulation signal shown in FIG. A voltage V _{b obtained} by amplitude-modulating the voltage F (t) with a cos phase modulation signal is output from the detection winding 13.
Since the resolver 10 is grounded at one end of the excitation winding 11 and the detection windings 12 and 13, it outputs a signal at a single end.

The band-pass filters 34A and 34B connected to the detection windings 12 and 13 of the resolver 10 obtain the excitation carrier frequency (the frequency of the excitation voltage of the excitation winding 11 ₎ from the output voltages V _a and V _{b of the} detection windings 12 and 13. ) To remove frequency components other than the components.
The synchronous demodulation circuit 35A outputs the output signal of the bandpass filter 34A as it is during the “H” period of the clock signal output from the frequency divider circuit 32 (a signal corresponding to the signal in FIG. 2A). In the “L” period, the output signal of the filter 34A is inverted and output.
Similarly, the synchronous demodulation circuit 35B outputs the output signal of the bandpass filter 34B as it is during the “H” period of the output signal of the frequency divider circuit 32, and inverts the output signal of the filter 34B during the “L” period. Output.

Therefore, the synchronous demodulation circuit 35A outputs a sin phase signal shown in FIG. 2C, and the synchronous demodulation circuit 35B outputs a corresponding cos phase signal. The sin phase signal and the cos phase signal are input to the arithmetic circuit 37 after the excitation carrier frequency component is removed by the low-pass filters 36A and 36B, respectively.
The voltages V _a ′ and V _b ′ output from the low-pass filters 36A and 36B are expressed as follows, respectively.
V _a '= K · sin θ
V _b '= K · cos θ

The arithmetic circuit 37 performs A / D conversion on the voltages V _a ′ and V _b ′, and further executes R / D conversion processing to calculate the rotor rotation angle θ. The arithmetic circuit 37 forms a motor winding current command based on the deviation between the rotor rotation angle command θ _com and the rotor rotation angle θ. The frequency dividing circuit 38 generates a 20 KHz clock signal obtained by dividing the 100 KHz clock signal output from the oscillating circuit 31 by 5, and outputs this as a PWM (pulse width modulation) carrier signal.
The PWM inverter 39 controls the motor current so that a current corresponding to the motor winding current command flows through the winding of the motor 20 by a switching operation based on the PWM carrier signal.

Next, the relationship between the switching noise generated by the PWM control and the output signal of the resolver 10 will be described with reference to FIG. Here, it is assumed that the motor 50 is stopped and the rotation angle θ does not change. For the sake of easy explanation, the operation of the bandpass filters 34A and 34B is ignored.
FIG. 3A shows an excitation signal (frequency f _am = 50 KHz) of the resolver 10, and FIG. 3B shows a PWM carrier signal (frequency f _pwm = 20 KHz). In the PWM inverter 39, switching noise synchronized with the PWM carrier signal is generated. Since this switching noise wraps around the resolver 10 via the motor 50 main body, cable, etc., the PWM switching noise indicated by hatching in FIG. 3C is included in the output signal of the resolver 10 when the PWM carrier signal rises. Superimpose with.

  In the synchronous demodulation circuit 35A (35B), the polarity of the output signal of the resolver 10 is inverted during the “L” period of the output signal of the frequency dividing circuit 32, that is, the “L” period of the excitation signal of the resolver 10. From the synchronous demodulation circuit 35A (35B), as shown in FIG. 3D, the non-inversion component and the inversion component of the switching noise appear alternately. The non-inverted component and the inverted component of the switching noise are smoothed and canceled by the low-pass filter 36A (36B). As a result, as shown in FIG. 3E, no noise component appears in the output of the low-pass filter 36A (36B).

FIG. 4 shows the relationship between the switching noise and the output signal of the resolver 10 when the frequency f _am of the excitation signal of the resolver 10 is 40 KHz and the frequency f _pwm of the PWM carrier signal is 20 KHz.
In this case, since only the positive polarity switching noise shown in FIG. 4C is superimposed on the output signal of the resolver 10, the outputs of the synchronous demodulation circuits 35A and 35B are not as shown in FIG. Inverted switching noise appears. As a result, the smoothed switching noise components appear in the outputs of the low-pass filters 36A and 36B, as shown by hatching in FIG. This means that when the motor 20 is rotated, the original output signal levels of the low-pass filters 36A and 36B change under the influence of the switching noise.

After all, when f _am > f _pwm is satisfied, the following conclusion is drawn.
_{f am = f pwm × (n} + 0.5) ··· (1)
Here, when n is a positive integer, the influence of PWM switching noise does not appear on the outputs of the low-pass filters 36A and 36B. Also,
f _am = f _pwm × n (2)
In this case, the switching noise component included in the outputs of the low-pass filters 36A and 36B is maximized.

Next, a case where f _am <f _pwm is considered. For example, as shown in FIG. 5 (a), (b) , when the f _am = f _pwm / 2, the switching noise is superimposed in the form shown in FIG. 5 (c) to the output signal of the resolver 10. Accordingly, as shown in FIG. 5D, the non-inverted component and the inverted component of the switching noise are alternately output from the synchronous demodulator circuit 35A (35B). This non-inversion component and inversion component are canceled by (36B). Therefore, as shown in FIG. 5D, the output of the low-pass filter 36A (36B) is not affected by switching noise.

On the other hand, when the relationship between the frequencies f _am and f _pwm is, for example, f _am = f _pwm / 3, switching noise is superimposed on the output signal of the resolver 10 in a form similar to the case of FIG. Therefore, as in the case of FIG. 4, the switching noise component included in the outputs of the low-pass filters 36A and 36B is maximized.

After all, in the relationship of f _am <f _pwm ,
f _am = f _pwm / (2n) (3)
Switching noise is canceled in the case of
f _am = f _pwm / (2n + 1) (4)
In this case, the effect of switching noise is maximized.

In the present embodiment, based on the above considerations, the frequency f _pwm frequency f _am and PWM carrier signal of the excitation signal of the resolver 10, so as to satisfy the relationship of formula (1) in f _am> f _pwm, also, It is set so as to satisfy the relationship of the expression (3) when f _am <f _pwm .
Therefore, according to the present embodiment, the rotor rotation angle of the motor 20 is detected without being affected by the PWM switching noise included in the output of the resolver 10, and the control accuracy of the rotation angle of the motor 20 is improved.

Note that, when the excitation signal of the resolver 10 and the PWM carrier signal are not synchronized, the phases of both signals are shifted with time, so the direction and magnitude of the switching noise component periodically change. However, even in this case, the effect of canceling the PWM switching noise can be obtained by setting the ratio of the frequencies f _am and f _{pwm in} accordance with the relationship of the above formulas (1) and (3). Therefore, instead of obtaining the PWM carrier signal from the frequency divider 38, it is possible to obtain the PWM carrier signal from an independent oscillator that generates a carrier signal having a predetermined frequency.

The state detection device according to the above embodiment uses the resolver 10 as means for detecting the rotation angle of the rotor of the motor 20, but an MR (Magneto resistance) sensor can be used instead of the resolver 10.
As shown in FIG. 6, the MR sensor 40 includes bridge-connected magnetoresistive elements 41 to 44 and a magnet (not shown) attached to the rotor of the motor 20, and is common to the magnetoresistive elements 41 and 43. The connection point is connected to the output of the excitation circuit 33, and the common connection point of the magnetoresistive elements 42 and 44 is grounded.
In this MR sensor 40, the resistance values R _{1 to} R _{4 of the} magnetoresistive elements 41 to 44 change as follows in accordance with changes in the strength and direction of the magnetic field due to the rotational displacement of the magnet.
R ₁ = R _C −R _V · sin θ
R ₂ = R _C + R _V · sin θ
R ₃ = R _C −R _V · cos θ
R ₄ = R _C + R _V · cos θ
Here, R _C : resistance constant component, R _V : resistance change

Therefore, from the midpoint of the sin phase (the common connection point of the magnetoresistive elements 41 and 42),
V _ma = R ₂ / (R ₁ + R ₂ ) × F (t) = [(1/2) + R _V · sin θ / (2R
_C )] × F (t)
From the midpoint of the cos phase,
V _mb = R ₄ / (R ₃ + R ₄ ) × F (t) = [(1/2) + R _V · cos θ / (2R
_C )] × F (t)
Is output.

The output voltages V _ma and V _mb of the MR sensor 40 are amplitude-modulated excitation carrier signals, like the output voltages V _a and V _{b of the} resolver 10. Therefore, the frequency f _pwm frequency f _am and PWM carrier signal of the excitation signal of the MR sensor 40, so as to satisfy the relationship of formula (1) in f _am> f _pwm, also, f _am <formula in f _pwm (3 ), The rotor rotation angle of the motor 20 can be detected without being affected by PWM switching noise, as in the case where the resolver 10 is used.
It is possible to use a variable magnetoresistive resolver instead of the MR sensor 40. In this case, each winding resistance of the variable magnetoresistive resolver is the same as the magnetoresistive elements 41 to 44 of the MR sensor 40. It is bridge-connected in the form of

In the state detection device according to the above embodiment, the detected state of the motor 20 is the rotation angle of the rotor of the motor. However, the present invention detects the state of the motor 20 different from the rotation angle of the rotor. Can also be applied effectively.
That is, for example, when detecting the temperature of the motor 20, as shown in FIG. 7, the output voltage of the excitation circuit is applied to the series circuit of the thermistor 51 and the resistor 52 constituting the temperature sensor 50. In this case, a signal obtained by amplitude-modulating the excitation carrier with a signal corresponding to the temperature is output from the common connection point of the thermistor 51 and the resistor 52. Therefore, the frequency f _{am of the} excitation signal and the frequency f _pwm of the PWM carrier signal are set so as to satisfy the relationship of the equation (1) when f _am > f _pwm , and the relationship of the equation (3) is satisfied when f _am <f _pwm . By setting so as to satisfy, the temperature of the motor 20 can be accurately detected without being affected by PWM switching noise.
The detected motor temperature is used for detecting an overload of the motor.

It is a block diagram of a motor control apparatus provided with the state detection apparatus which concerns on one embodiment of this invention. (A) illustrates the waveform diagram of the excitation voltage, (b) illustrates the waveform diagram of the amplitude modulation output signal of the resolver, and (c) illustrates the waveform diagram of the demodulated resolver output signal. It is a wave form diagram which shows the relationship between the switching noise when the frequency of an excitation carrier signal and the frequency of a PWM carrier signal have a certain relationship, and the output signal of a resolver. It is a wave form diagram which shows the relationship between the switching noise and the output signal of a resolver when the frequency of an excitation carrier signal and the frequency of a PWM carrier signal have another relationship. It is a wave form diagram which shows the relationship between the switching noise and the output signal of a resolver when the frequency of an excitation carrier signal and the frequency of a PWM carrier signal have another relationship. It is a connection diagram of the magnetoresistive element of MR sensor. It is a connection diagram of the thermistor used as a temperature sensor.

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 Resolver 11 Excitation winding 12, 13 Detection winding 20 Motor 30 Control circuit 31 Oscillation circuit 32, 38 Dividing circuit 33 Excitation circuit 35A, 35B Synchronous demodulation circuit 36A, 36B Low-pass filter 37 Operation circuit 39 PWM inverter 40 MR sensor 50 Temperature sensor

Claims (5)

A state detection device that detects a state of the motor in a motor control device that performs PWM control of the motor,
A signal generating means for generating an analog state signal corresponding to the state of the motor and generating an amplitude modulated signal obtained by amplitude modulating the first carrier signal with the analog state signal;
Demodulating means for synchronously demodulating the analog state signal from the amplitude modulation signal using the first carrier signal;
Smoothing means for smoothing a PWM switching noise component included in the synchronously demodulated analog state signal,
Motor control, wherein the frequency f _am of the first carrier signal and the frequency f _pwm of the second carrier signal used for the PWM control are set so as to have the following relationship: A state detection device in the device.
When f _am > f _pwm ,
f _am = f _pwm × (n + 0.5)
When f _am <f _pwm ,
f _am = f _pwm / (2n)
Where n is a positive integer   2. The state detection device in a motor control device according to claim 1, wherein the signal generation means includes an angle sensor that generates a signal corresponding to a rotation angle of the rotor of the motor as the analog state signal.   The state detection device in the motor control device according to claim 2, wherein the angle sensor is a resolver.   The state detection device in the motor control device according to claim 2, wherein the angle sensor is an MR sensor.   2. The state detection device in a motor control device according to claim 1, wherein the signal generating means includes a temperature sensor that generates a signal corresponding to the temperature of the motor as the analog state signal.

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