JP4882816B2 - Rotating machine control device - Google Patents

Description

  The present invention relates to a control device for a rotating machine that acquires information related to a rotation angle of the rotating machine based on an electrical state quantity of the rotating machine when the rotating machine is driven by operating a switching element of an inverter.

  As this type of control device, for example, as seen in Patent Document 1 below, a high-frequency voltage signal is superimposed in the estimated d-axis direction of the output signal of the inverter for a three-phase motor having saliency, and is actually propagated at this time. There has also been proposed a method for calculating a rotation angle based on a high-frequency current signal. That is, since the inductance of the three-phase motor is the smallest in the d-axis direction, current tends to flow in the d-axis direction compared to the q-axis direction. Therefore, regardless of the phase angle of the superimposed high frequency voltage signal, the actually propagated high frequency current signal is a signal having a large value in the d-axis direction. Therefore, if the high-frequency current signal that actually propagates when the high-frequency voltage signal is superimposed in the estimated d-axis direction is deviated from the estimated d-axis, it is understood that the estimated d-axis is different from the actual d-axis. In the control device, paying attention to this point, it is possible to acquire information about the rotation angle by correcting the estimated d-axis so as to reduce the difference between the actually propagated high-frequency current signal and the estimated d-axis.

Specifically, the control device first calculates a rotation speed by performing a proportional-integral operation on the rotation angle error equivalent amount quantified based on the detected value of the actually propagated high-frequency current signal. Then, the rotation angle is calculated by the integral calculation of the rotation speed.
JP 2003-348896 A

  By the way, the inventors have found that in the above control device, there is a possibility that the tolerance to noise is lowered when the rotation angle is calculated. For this reason, even when the rotation speed of the three-phase motor is constant, there is a possibility that a phenomenon in which an error equivalent amount of the calculated value of the rotation angle vibrates may occur. On the other hand, it may be possible to avoid such a problem by performing filter processing when calculating the rotation angle, but in this case, the responsiveness when calculating the rotation angle and the rotation speed may be reduced. As described above, regarding the calculation of the rotation angle, there is a problem of improving both resistance to noise and improving responsiveness.

  Furthermore, the inventors have also found that a steady deviation occurs in the calculated rotation angle when the rotation speed changes rapidly in the control device.

  In addition to the control device described above, when driving a rotating machine by operating a switching element of an inverter, there is a method for acquiring information on the rotation angle of the rotating machine based on an electrical state quantity of the rotating machine. As a result, the above-mentioned problem may arise.

  The present invention has been made in order to solve the above-mentioned problems, and its purpose is to improve resistance to noise when acquiring information on the rotation angle of the rotating machine based on the electrical state quantity of the rotating machine. It is an object of the present invention to provide a control device for a rotating machine that can achieve a suitable balance with improved responsiveness. In addition, the present invention has been made to solve the above-described problems, and the object of the present invention is to suitably suppress or avoid occurrence of a steady deviation in the calculated rotation angle even when the rotation speed changes rapidly. It is an object of the present invention to provide a control device for a rotating machine that can be used.

  Hereinafter, means for solving the above-described problems and the operation and effects thereof will be described.

  According to a first aspect of the present invention, there is provided an error equivalent amount calculating means for calculating an error equivalent amount for the currently acquired rotation angle based on the electrical state quantity, and rotation of the rotating machine based on the error equivalent amount. A high response angle calculation means for calculating an angle; and a noise resistant angle calculation means for calculating a rotation angle of the rotating machine based on the error equivalent amount by a process having a higher degree of noise removal than the high response angle calculation means; And a switching unit that switches between the output of the noise-resistant angle calculation unit and the output of the high response angle calculation unit as a rotation angle used for driving the rotating machine.

  When the rotation angle is calculated by processing with a strong degree of noise removal, the responsiveness when calculating the rotation angle is lowered. On the other hand, when the rotation angle is calculated by processing with a low degree of noise removal, the responsiveness is improved, but the resistance to noise is reduced. In this regard, in the above-described invention, the noise resistance angle calculation unit and the high response angle calculation unit are provided, and which is used depending on the situation is suitable for both the improvement of resistance to noise and the improvement of response. Can be achieved.

  According to a second aspect of the present invention, there is provided an error equivalent amount calculating means for calculating an error equivalent amount for the currently acquired rotation angle based on the electrical state quantity, a filter means, and the error equivalent amount. A noise-resistant angle calculating means for calculating a rotation angle of the rotating machine based on the above, a high response angle calculating means for calculating the rotation angle of the rotating machine based on the error equivalent amount without providing a filter means, and the rotating machine. The rotation angle used for driving includes switching means for switching between the output of the noise-resistant angle calculation means and the output of the high response angle calculation means.

  When the rotation angle is calculated with the filter means, the resistance to noise is improved, but the responsiveness in calculating the rotation angle is reduced. On the other hand, when the rotation angle is calculated without providing the filter means, the responsiveness is improved, but the resistance to noise is reduced. In this regard, in the above-described invention, the noise resistance angle calculation unit and the high response angle calculation unit are provided, and which is used depending on the situation is suitable for both the improvement of resistance to noise and the improvement of response. Can be achieved.

  According to a third aspect of the present invention, in the first or second aspect of the invention, the high response angle calculating means calculates a rotational speed and one of its equivalent values based on a multiple integral operation with the error equivalent amount as an input. And a means for calculating the rotation angle based on an integral operation of the output of the speed calculation means.

  In the above invention, the transfer function having the error equivalent amount and the rotation angle as input and output is a transfer function of the third order or higher. For this reason, even when the rotation speed changes in a ramp shape, it is possible to avoid a steady deviation in the rotation angle.

  According to a fourth aspect of the present invention, in the third aspect of the invention, the high response angle calculating unit further includes a correcting unit that corrects the rotation angle calculated based on the integral calculation based on the error equivalent amount. Features.

  In the above invention, by providing the correction means, it is possible to improve the degree of freedom of control design, such as making it possible to individually design stability and responsiveness.

  According to a fifth aspect of the present invention, in the invention according to any one of the first to fourth aspects, the noise-resistant angle calculation unit is configured to perform the rotation by a differential operation of a rotation angle output from the noise-resistant angle calculation unit and a filtering process. A speed calculating means for calculating either the rotational speed of the machine or an equivalent value thereof, a correcting means for correcting the output of the speed calculating means based on the error equivalent amount, and the anti-resistance based on an integral operation of the corrected output. Means for calculating a rotation angle as an output of the noise angle calculation means.

  In the above invention, even if noise is mixed in the rotation angle, the noise component is removed by performing the filtering process when calculating the rotation speed by the differential operation. And since a rotation angle is calculated based on this, the noise component of a rotation angle comes to be removed suitably.

  According to a sixth aspect of the present invention, in the fifth aspect of the invention, the noise proof angle calculation unit corrects the rotation angle calculated based on the integral calculation based on the error equivalent amount, thereby calculating the noise proof angle calculation. The rotation angle as the output of the means is calculated.

  In the above-described invention, since the rotation angle calculated based on the integral calculation has a term for correcting based on the error equivalent amount, the degree of freedom in design of the control is improved as compared with the case where the rotation angle is directly calculated from the integral calculation. Can be made.

  The invention according to claim 7 is the invention according to any one of claims 1 to 6, wherein each of the high response angle calculation means and the noise resistant angle calculation means is calculated based on the electrical state quantity. The rotation angle is calculated based on an integral calculation of any one of a rotation speed and an equivalent value thereof, and the switching means is an integral calculation that is adopted after switching as a means for calculating the rotation angle at the time of the switching. As an initial value, the value of the integral operation that was adopted before switching is set.

  When the rotation angle is calculated based on the integral calculation of the rotation speed or its equivalent value, feedback correction is applied to the rotation angle by actually using the rotation angle. For this reason, when the rotation angle is actually used, the result of the integral calculation converges to an appropriate value. However, when the rotation angle is not actually used, the result of the integral calculation is not an appropriate value. For this reason, immediately after switching, the integration calculation result of the newly selected angle calculation means does not become an appropriate value, and the calculation accuracy of the rotation angle may be reduced. In this regard, in the above-described invention, by setting the integral calculation value of the angle calculation means adopted before the switching as the initial value of the integral calculation of the angle calculation means newly adopted by the switching, such a problem can be suitably achieved. Can be suppressed.

  When the high response angle calculation unit in the invention according to claim 7 includes the speed calculation unit according to claim 3 and the noise-resistant angle calculation unit includes the speed calculation unit exemplified in claims 5 and 6, When switching from the noise resistant angle calculating means to the high response angle calculating means, it is desirable to set the rotation speed calculated by the noise resistant angle calculating section as the initial value of the speed calculating section.

  The invention according to claim 8 is the invention according to any one of claims 1 to 6, wherein each of the high response angle calculation means and the noise resistant angle calculation means is calculated based on the electrical state quantity. The rotation angle is calculated based on any integral operation of the rotation speed and its equivalent value, and means for performing the processing related to the integral operation is shared.

  When the rotation angle is calculated based on the integral calculation of the rotation speed or its equivalent value, feedback correction is applied to the rotation angle by actually using the rotation angle. For this reason, when the rotation angle is actually used, the result of the integral calculation converges to an appropriate value. However, when the rotation angle is not actually used, the result of the integral calculation is not an appropriate value. For this reason, when a means for performing integral calculation processing is provided between the high response angle calculation means and the noise-resistant angle calculation means, the integration calculation result of the newly selected angle calculation means is appropriate immediately after switching. There is a risk that the calculation accuracy of the rotation angle may be lowered. In this regard, in the above invention, by sharing the means for performing the integral calculation, the continuity of the integral calculation result can be maintained regardless of the switching.

  When the high response angle calculation unit according to the invention of claim 8 includes the speed calculation unit according to claim 3 and the noise-resistant angle calculation unit includes the speed calculation unit exemplified in claims 5 and 6, When switching from the noise resistant angle calculating means to the high response angle calculating means, it is desirable to set the rotation speed calculated by the noise resistant angle calculating section as the initial value of the speed calculating section.

  According to a ninth aspect of the present invention, in the invention according to any one of the first to eighth aspects, the switching means is a rotation used to drive the rotating machine when a rotation angle of the rotating machine is equal to or larger than a predetermined value. The output of the noise-resistant angle calculation means is selected as an angle.

  When starting the rotating machine, the rotational speed increases. Further, the rotational speed decreases during the deceleration operation of the rotating machine. For this reason, when the rotational speed is a low rotational speed, the rotational speed tends to fluctuate easily. On the other hand, when the rotation speed is high, the rotation speed tends to be in a steady state. In the above invention, in view of this point, switching can be appropriately performed by using the output of the noise resistant angle calculation means in a high rotation speed region where the rotation speed tends to be stable.

  In the case where the high response angle calculation means and the noise resistant angle calculation means include speed calculation means, the calculated values of these speed calculation means may be used as the rotation speed.

  According to a tenth aspect of the present invention, in the invention according to any one of the first to ninth aspects, the switching means is a rotation angle used for driving the rotating machine in a situation where the rotating speed of the rotating machine is stable. The output of the noise-resistant angle calculation means is selected as follows.

  In a situation where the rotational speed of the rotating machine is stable, the angle error by the high response angle calculation means may have a vibration component due to noise or the like that is several times the rotational speed. On the other hand, such a vibration component can be sufficiently suppressed by the noise resistant angle calculation means. In this regard, in the above configuration, the fluctuation of the angle error when the rotational speed is stable can be suitably suppressed by using the noise-proof angle calculation means in a situation where the rotational speed is stable.

  The invention according to claim 11 is the invention according to any one of claims 1 to 10, wherein the switching means drives the rotating machine when the amount of change in the rotating speed of the rotating machine is not less than a predetermined value. The output of the high response angle calculating means is selected as the rotation angle used for the purpose.

  When the rotational speed changes, high responsiveness is required particularly for calculating the rotational angle and rotational speed. In this regard, in the above invention, switching can be appropriately performed by using the output of the high-responsiveness angle calculation means when the amount of change in the rotation speed is greater than or equal to a predetermined value.

  In the case where the high response angle calculation means and the noise resistant angle calculation means include speed calculation means, the calculated values of these speed calculation means may be used as the rotation speed.

  The invention according to claim 12 is the invention according to any one of claims 1 to 11, wherein the switching means switches from the noise-resistant angle calculation means to the high response angle calculation means, and the high response angle. The conditions for switching from the calculating means to the noise-proof angle calculating means are set so as to be different from each other.

  In the above invention, the condition for switching from the noise resistant angle calculating means to the high response angle calculating means and the condition for switching from the high response angle calculating means to the noise resistant angle calculating means are made different from each other, thereby switching between these two means. Can be avoided frequently.

  A thirteenth aspect of the present invention is the invention according to any one of the first to twelfth aspects of the present invention, wherein the rotating machine has a saliency in its structure, and based on the information about the rotation angle, the rotating machine The apparatus further includes superimposing means for superimposing a frequency signal having a period different from the rotation period of the electrical angle and oscillating in an arbitrary phase angle direction on the output signal of the inverter, and the error equivalent amount calculating means is The error equivalent amount is calculated based on the vibration direction of the actually propagated frequency signal.

  In the above invention, the frequency signal is superimposed on the output signal of the inverter. Thereby, a predetermined frequency signal propagates through the rotating machine. However, when the acquired rotation angle and the actual rotation angle are shifted, it is considered that a shift occurs between the assumed frequency signal and the actually propagated frequency signal. In the above invention, paying attention to this point, the error equivalent amount is calculated based on the deviation between the assumed frequency signal and the actually propagated frequency signal.

  Note that the frequency signal actually propagated by superimposition is a state quantity that is not the same as the state quantity that is directly operated as the output of the inverter in order to superimpose the frequency signal among the detected values of the electrical state quantity of the rotating machine. The signal is calculated based on the detected value. For example, when the output voltage of the inverter (phase voltage of the rotating machine) is manipulated to the command voltage to superimpose the frequency signal, the actually propagated frequency signal is a state quantity (for example, current) other than the voltage. The signal is calculated based on the value.

  According to a fourteenth aspect of the present invention, in the thirteenth aspect of the present invention, the moving average processing means for applying a moving average processing to at least one of the detected value of the actually propagated frequency signal and the output of the error equivalent amount calculating means is further provided. The moving average processing means performs the moving average processing based on a time interval that is an integral multiple of the period of the frequency signal superimposed by the superimposing means.

  By performing the moving average process, the influence of noise can be suppressed. However, when detecting the response to the superposition of the frequency signal, it is difficult to appropriately perform the moving average process if the moving average process of the detection result is performed regardless of the cycle of the frequency signal. In the above invention, in view of this point, it is possible to appropriately perform this by performing the moving average process at a time interval that is an integral multiple of the period of the frequency signal.

(First embodiment)
Hereinafter, a first embodiment in which a control device for a rotating machine according to the present invention is applied to a control device for a three-phase motor mounted on a hybrid vehicle will be described with reference to the drawings.

  FIG. 1 shows an overall configuration of a control system according to the present embodiment.

  The illustrated electric motor 10 is an embedded magnet synchronous motor (IPMSM). That is, as shown in FIG. 2, the rotor 10a of the electric motor 10 is configured by embedding a permanent magnet in an iron body.

  The αβ converter 20 shown in FIG. 1 converts the current flowing through the motor 10 into the current in the stationary coordinate system based on the U-phase actual current iu and the V-phase actual current iv out of the current actually flowing through the motor 10. That is, it is a part for converting into current vector components of the α axis and the β axis. Here, for example, the U phase is set to the same phase angle as the α axis, and the β axis is determined so as to be orthogonal thereto.

  The dq converter 22 is a part that converts the actual current iα on the α axis and the actual current iβ on the β axis into the current of the rotating coordinate system, that is, the actual current id on the d axis and the actual current iq on the q axis. is there. In this conversion, the rotation angle θ of the output shaft of the electric motor 10 is used. More precisely, the rotation angle θ is an electrical angle and is a rotation angle in the positive direction of the d axis with respect to the α axis. At this time, the low-pass filter also performs a process of removing a high-frequency component described later from the output of the αβ converter 20. For this reason, the dq conversion unit 22 extracts the current of the d-axis component and the q-axis component used when driving the electric motor 10 from the current that actually flows through the electric motor 10.

  The command current setting unit 24 is a part for setting the command current idc on the d axis and the command current iqc on the q axis based on the required torque Td for the electric motor 10.

  The command voltage setting unit 26 is a part that calculates the command voltage vdc on the d axis and the command voltage vqc on the q axis based on the command current idc, the command current iqc, the actual current id, and the actual current iq. This conversion is basically performed by feedback control of the actual current id on the d axis to the command current idc and feedback control of the actual current iq on the q axis to the command current iqc. This feedback control may be proportional integral control, for example.

  The αβ converter 27 converts the command voltage vdc on the d axis and the command voltage vqc on the q axis into a command voltage vαc on the α axis and a command voltage vβc on the β axis. In this conversion, the rotation angle θ is used.

  The three-phase conversion unit 28 converts the output of the adder 30a according to the command voltage vαc on the α axis and the output of the adder 30b according to the command voltage vβc on the β axis into the u-phase command voltages vuc, v This is a part for converting into a phase command voltage vvc and a w-phase command voltage vwc.

  The PWM signal generation unit 32 is a part that generates an operation signal of the inverter 34 for applying the command voltages vuc, vvc, vwc to the electric motor 10. As a result, the switching element SW of the inverter 34 is operated, and the voltage of the high voltage battery 36 is applied to the electric motor 10.

  Next, processing related to acquisition of the rotation angle θ of the electric motor 10 according to the present embodiment will be described.

  In the present embodiment, when driving the electric motor 10, a high-frequency signal having a cycle shorter than the rotation cycle of the electric angle of the electric motor 10 is superimposed on the output of the inverter 34. In other words, a high-frequency signal having a cycle shorter than the cycle of the current that actually flows through the electric motor 10 is superimposed according to the command currents idc and iqc. Then, the rotation angle θ of the electric motor 10 is calculated based on the high-frequency signal that actually propagates through the electric motor 10. This is done in view of the electric motor 10 having saliency.

  That is, since the electric motor 10 has saliency due to its structure, the inductance in the d-axis direction is minimum and the inductance in the q-axis direction is maximum. Accordingly, since the current flows more easily in the d-axis direction than in the q-axis direction, the high-frequency signal actually propagated through the electric motor 10 is deflected in the d-axis direction when the high-frequency signal is superimposed. Specifically, as shown in FIG. 3A, when the estimated d-axis (estimated d-axis) is advanced with respect to the actual d-axis (real d-axis), the estimated d-axis When a high-frequency signal (one-dot chain line in the figure) is superimposed on the direction, the direction of the actually propagated high-frequency signal (solid line in the figure) is delayed with respect to the estimated d-axis in order to deflect toward the real d-axis side. It shifts to the corner side. As shown in FIG. 3B, when the estimated d-axis and the actual d-axis coincide with each other, when the high-frequency signal (one-dot chain line in the figure) is superimposed in the estimated d-axis direction, it is actually propagated. The direction of the high-frequency signal to be performed (solid line in the figure) coincides with the estimated d-axis. Furthermore, as shown in FIG. 3C, when the estimated d-axis is retarded with respect to the actual d-axis, a high-frequency signal (one-dot chain line in the figure) is superimposed in the estimated d-axis direction. The direction of the actually propagating high-frequency signal (solid line in the figure) is shifted to the advance side with respect to the estimated d-axis to be deflected to the actual d-axis.

  If the above property is used, the d-axis can be estimated and calculated, and thus the rotation angle θ can be calculated. That is, by repeating the superposition of the high frequency signal while making the direction in which the high frequency signal actually propagates be the estimated d-axis direction, the phase angle of the superposed high frequency signal can be matched with the phase angle of the actually propagated high frequency signal, As a result, the estimated d-axis can be matched with the actual d-axis.

  Specifically, as shown in FIG. 1, the high frequency voltage setting unit 40 outputs a voltage signal vhdc as a high frequency signal in the d-axis direction to the αβ conversion unit 41. The αβ conversion unit 41 converts the voltage signal vhdc into a voltage signal vhαc on the α axis and a voltage signal vhβc on the β axis, and outputs them to the adders 30a and 30b. For this reason, a signal in which the voltage signal vhdc is superimposed on the command voltages vdc and vqc is input to the three-phase conversion unit 28. On the other hand, the high frequency current detector 42 extracts only high frequency components of the actual current iα on the α axis and the actual current iβ on the β axis. That is, the current signal ihα on the α-axis and the current signal ihβ on the β-axis are generated and output as high-frequency signals that are actually propagated to the electric motor 10. Based on the vector signals (voltage signals vhαc, vhβc) output from the αβ converter 41 and the vector signals (current signals ihα, ihβ) output from the high-frequency current detector 42, the error equivalent amount detector 44 generates an outer product of them. Is calculated. This outer product value is a parameter having a correlation with the phase angle difference between the voltage signal vhdc and the current signals ihα and ihβ. For this reason, if this outer product is set to zero, the voltage signal vhdc output from the high-frequency voltage setting unit 40 can be superimposed in the direction with the smallest inductance. In the present embodiment, the position / velocity calculation unit 46 calculates and outputs the temporary rotation angle θp so that the error equivalent amount Δθ output from the error equivalent amount detection unit 44 is zero. Then, the αβ conversion unit 41 performs αβ conversion based on the temporary rotation angle θp. Thereby, the voltage signal vhdc is superimposed in the direction in which the inductance is assumed to be minimum.

  By the way, when the output torque of the electric motor 10 increases, magnetic saturation may partially occur depending on the current flow mode in the electric motor 10. This will be described below with reference to FIG. FIG. 4B shows a high-frequency signal that is actually propagated when a high-frequency signal is superimposed in all directions while keeping the amplitude constant as shown in FIG. That is, FIG. 4B shows that when the current vector for driving the electric motor 10 (command current idc, iqc) becomes the current vector on the q-axis, the direction in which the inductance becomes the minimum from the d-axis direction. An example in which a phenomenon that shifts toward the current vector direction occurs is shown. In this case, the vector signal that actually propagates through the electric motor 10 is deflected to the driving current vector side by superposition of the high-frequency signal. When the phenomenon shown in FIG. 4B occurs, the temporary rotation angle θp cannot be set in the d-axis direction. Here, a method for dealing with the phenomenon shown in FIG. 4B in the present embodiment will be described.

  FIG. 5 shows the relationship between the phase angle of the actual current vector (actual current id, iq) for driving, the output torque of the electric motor 10, and the amplitude (target amplitude) of the frequency signal that actually propagates through the electric motor 10. . Specifically, this relationship indicates a relationship when a high-frequency signal in a direction in which the inductance of the electric motor 10 is minimized is superimposed on the output signal of the inverter 34. As shown in the figure, the target amplitude value changes depending on the phase angle of the driving current vector and the output torque. For this reason, when the actual amplitude value deviates from the target amplitude value, it is considered that the value deviated from the actual phase angle is recognized as the phase angle of the driving current. In other words, it is considered that the rotation angle shifted from the actual rotation angle is recognized as the rotation angle (more precisely, the electrical angle) of the electric motor 10. Therefore, in order to correct the temporary rotation angle θp output from the position / velocity calculation unit 46 so as to reduce the difference between the actual amplitude value and the target amplitude value, as shown in FIG. 48. FIG. 6 shows the processing of the position correction unit 48.

  The torque estimation unit 48a is a part that calculates an estimated value Te of the output torque of the electric motor 10 based on the actual currents id and iq. The phase angle calculation unit 48b is a part that calculates the phase angle φ of the current vector (current vector generated by the actual currents id and iq) flowing through the electric motor 10 based on the actual currents id and iq. The target amplitude value setting unit 48c is a part that calculates a target amplitude value (the amplitude value shown in FIG. 5 above) based on the estimated value Te of the output torque and the phase angle φ of the actual current vector. The deviation amount calculation unit 48d is a part that calculates the difference between the detected value of the amplitude value of the high-frequency signal output from the high-frequency current detection unit 42 and the target amplitude value. The correction amount calculation unit 46e is a part that calculates the correction amount cor of the temporary rotation angle θp for causing the actual amplitude value to follow the target amplitude value. Here, for example, the correction amount cor may be calculated by the sum of the proportional term and the integral term of the output signal of the deviation amount calculation unit 46d.

  With the above processing, even when magnetic saturation occurs in the electric motor 10, the rotation angle θ can be set with high accuracy. That is, as shown in FIG. 7, the phase angle at which the inductance is minimized due to magnetic saturation is shifted from the d-axis, so the direction in which the temporary rotation angle θp is zero is not the d-axis direction. However, by correcting the temporary rotation angle θp with the correction amount cor, the direction in which the rotation angle θ is zero can be matched with the d-axis direction.

  At this time, in the present embodiment, the high frequency signal (voltage signal vhdc) is set based on the temporary rotation angle θp. As a result, the rotation angle θ can be converged as a true value. On the other hand, when the high frequency signal (voltage signal vhdc) is set based on the final rotation angle θ, the superposition direction of the high frequency signal is true d even when the motor 10 is magnetically saturated. It is in the axial direction and is not in the direction with the smallest inductance. Therefore, there is a phase difference between the superimposed high frequency signal (voltage signal vhdc) and the actually propagated high frequency signal (current signals ihα, ihβ). For this reason, in the position / velocity calculation unit 46, the temporary rotation angle θp is changed, and as a result, the rotation angle θ is changed despite the correct value.

  Next, the position / velocity calculation unit 46 will be described in detail. FIG. 8 shows the processing of the position / velocity calculation unit 46. As illustrated, the position / velocity calculation unit 46 includes a noise-resistant angle calculation unit 50 and a high response angle calculation unit 60. These will be described in detail below.

<Noise resistant angle calculation unit 50>
Here, in the noise-proof angle calculation unit 50, the rotation angle θ1 that is the output is differentially calculated by the differential calculation unit 51, and then this is filtered by the low-pass filter 52 that is a first-order lag transfer function. Temporary rotational speed ω1p is calculated. On the other hand, the speed correction amount calculation unit 53 calculates the speed correction amount Δω1 by proportional integral calculation of the error equivalent amount Δθ. The speed correction unit 54 calculates the rotational speed ω1 by adding the speed correction amount Δω1 to the temporary rotational speed ω1p. The integral calculation unit 55 calculates a temporary rotation angle θ1p by integrating the rotation speed ω1. On the other hand, the angle correction amount calculation unit 56 calculates the angle correction amount Δθ1 by proportional integral calculation of the error equivalent amount Δθ. Then, the angle correction unit 57 calculates the rotation angle θ1 by adding the correction amount Δθ1 of the angle correction amount calculation unit 56 to the temporary rotation angle θ1p.

  The noise-proof angle calculation unit 50 is a third-order transfer function having the error equivalent amount Δθ and the rotation angle θ1 as input and output. For this reason, even if it is a case where rotation speed (omega) 1 changes rapidly, the steady state deviation of a rotation angle can be suppressed or eliminated suitably. This will be described in detail below.

When the proportional gain Kp1 and integral gain Ki1 of the speed correction amount calculation unit 53 and the proportional gain Kp2 and integral gain Ki2 of the angle correction amount calculation unit 56 are used, the error equivalent amount Δθ and the rotation angle θ1 are expressed by the following transfer function: Is related.

Δθ = θ1 × Ts × s × s / (1 + Ts) (Kp1 × s + Ki1 + Kp × s × s + Ki1 × s)

Here, assuming a ramp-like change as the change in the rotation speed ω, the change in the rotation speed can be expressed as “ωinc / s × s” using a constant ωinc, and the rotation angle θ is expressed as “ωinc / s × s”. s × s ”. For this reason, the steady-state deviation Δθ (∞) of the error equivalent amount Δθ is zero as follows according to the Laplace final value theorem.

Δθ (∞)
= Lim s × (ωinc / s × s × s) × Ts × s × s / (1 + Ts) (Kp1 × s + Ki1 + Kp × s × s + Ki1 × s)
= 0

For this reason, even if the rotational speed changes in a ramp shape, the steady deviation can be made zero. It is desirable that the proportional gain Kp2 and the integral gain Ki2 of the angle correction amount calculation unit 56 are set to values smaller than the proportional gain Kp1 and the integral gain Ki1 of the speed correction amount calculation unit 53.

The low-pass filter 52 is designed to remove higher-order vibration components that may occur at the rotation angle θ1 when the rotation speed is steady. That is, as shown in FIG. 9, the gain of the low-pass filter 52 gradually decreases to less than “1” when the gain is equal to or higher than a specific frequency fc (cutoff frequency). For this reason, the vibration component can be suitably removed by designing the specific frequency fc to be smaller than the frequency of the higher-order vibration component. Here, as the higher-order vibration component, the sixth-order vibration component of the rotation period of the electrical angle is mainly used, and thus changes according to the rotation speed. For this reason, in the present embodiment, the cut-off frequency fc is variably set according to the rotational speed ω1. This sixth-order vibration component tends to be particularly noticeable when the winding specification of the electric motor 10 is concentrated winding.
<High response angle calculation unit 60>
In the high response angle calculation unit 60, the speed calculation unit 61 calculates the rotation speed ω2 as the sum of the double integration calculation, the integration calculation, and the proportional calculation for the error equivalent amount Δθ. Then, the integral calculation unit 62 calculates the rotation angle θ2 by integrating the rotation speed ω2. The high response angle calculation unit 60 is also a third-order transfer function having the error equivalent amount Δθ and the rotation angle θ2 as input and output. For this reason, even if it is a case where rotational speed (omega) 2 changes rapidly, a steady-state deviation can be suppressed or eliminated suitably. This will be described in detail below.

When the proportional gain Kp3, integral gain Ki3, and double integral gain Kii of the speed calculation unit 61 are used, the error equivalent amount Δθ and the rotation angle θ2 are related by the following transfer function.

Δθ = θ2 × s × s × s / (Kp3 × s × s × s + Ki3 × s + Kii)

Here, assuming a ramp-like change as the change in the rotation speed ω, the change in the rotation speed can be expressed as “ωinc / s × s” using a constant ωinc, and the rotation angle θ is expressed as “ωinc / s × s”. s × s ”. For this reason, the steady-state deviation Δθ (∞) of the error equivalent amount Δθ is zero as follows according to the Laplace final value theorem.

Δθ (∞)
= Lim s × (ωinc / s × s × s) × s × s × s / (Kp3 × s × s × s + Ki3 × s + Kii)
= 0

For this reason, even if the rotational speed changes in a ramp shape, the steady deviation can be made zero.

  Incidentally, since the error equivalent amount Δθ is calculated based on the detected value of the current output from the high-frequency current detector 42, noise is likely to be mixed therein. Here, since the noise-proof angle calculation unit 50 includes the low-pass filter 52, the degree of noise removal is strong. That is, even if the error equivalent amount Δθ includes noise or the noise becomes large due to the differential operation of the rotation angle θ1, this noise is suitably removed by the low-pass filter 52. Then, the rotational speed ω1 is calculated by correcting the temporary rotational speed ω1p from which the noise is removed by the correction amount Δω1, and the rotational angle θ1 is calculated from the calculated rotational speed ω1, thereby suitably removing the noise from the rotational angle θ1. be able to. However, since the noise-resistant angle calculation unit 50 includes the low-pass filter 52, the responsiveness is low. For this reason, when the rotational speed ω changes, a delay occurs in the estimation of the rotational speed ω1 and the rotational angle θ1.

  On the other hand, since the high response angle calculation part 60 is not provided with a filter means, it is highly responsive. For this reason, even when the rotational speed changes, there is little estimation delay of the rotational speed ω2 and the rotational angle θ2. However, since the high response angle calculation unit 60 does not include a filter unit, it is significantly affected by noise.

  FIG. 10 shows a rotation angle estimation error (position error) and a speed estimation error when the rotation speed changes in a ramp shape. Specifically, FIG. 10A shows a case where the noise resistant angle calculation unit 50 is used, and FIG. 10B shows a case where the high response angle calculation unit 60 is used. As shown in the figure, when the noise-resistant angle calculation unit 50 is used, an estimated delay of the rotational speed ω1 occurs with respect to the true rotational speed indicated by a one-dot chain line in the figure. In particular, this delay becomes prominent at the start of the change in rotational speed. On the other hand, when the high response angle calculation unit 60 is used, the response delay is preferably eliminated.

  FIG. 11 shows an estimated error equivalent amount of the rotation angle when the rotation speed is in a steady state. Specifically, FIG. 11A shows a case where the noise resistant angle calculation unit 50 is used, and FIG. 11B shows a case where the high response angle calculation unit 60 is used. As shown in the figure, when the high response angle calculation unit 60 is used, since the filter effect is weak, a vibration component is included in the position error.

  Incidentally, in FIG. 12A, in place of the speed calculation unit 61 of FIG. 8, the rotation speed when the rotation speed is calculated based on the sum of the integral calculation and the proportional calculation of the error equivalent amount Δ is in a ramp shape. The estimation error (position error) of the rotation angle and the estimation error of the speed when changing are shown. In this case, a steady error occurs in the estimation of the rotation angle as compared with the case of using the high response angle calculation unit 60 shown in FIG. This is because the transfer function having the error equivalent amount Δθ and the rotation angle as input and output is quadratic.

  As described above, both the noise-resistant angle calculation unit 50 and the high response angle calculation unit 60 have no steady deviation in the estimation of the rotation angle when the speed changes in a ramp shape. However, the anti-noise angle calculation unit 50 has a strong filter effect but low response, and the high response angle calculation unit 60 has high responsiveness but weak filter effect. ing. Therefore, in the present embodiment, as shown in FIG. 8, the switching unit 70 that switches between the noise-resistant angle calculation unit 50 and the high response angle calculation unit 60 according to the rotation speed, and the selector 72 are provided. Prepare. That is, the switching unit 70 operates the selector 72 based on the rotational speeds ω1 and ω2, so that the temporary rotation angle θp output from the position / speed calculation unit 46 is changed to the rotation angle θ1 output from the noise-resistant angle calculation unit 50. Or the rotation angle θ2 output from the high response angle calculation unit 60 is switched.

  FIG. 13 shows a processing procedure for switching between the noise-resistant angle calculation unit 50 and the high response angle calculation unit 60. This process is repeatedly executed by a microcomputer or the like, for example, at a predetermined cycle.

  In this series of processing, first, in step S10, it is determined whether or not the noise-resistant angle calculation unit 50 is selected. If it is determined that the noise resistant angle calculation unit 50 is selected, it is determined in step S12 whether or not the rotational speed ω1 is equal to or lower than a predetermined speed α. This process is to determine whether or not it is time to switch to the high response angle calculation unit 60. That is, when the electric motor 10 is started up or when the electric motor 10 is decelerated, the operating state of the electric motor 10 enters the low rotational speed region, so that it is considered that the rotational speed is not stable in the low rotational speed region. For this reason, in such a situation, the high response angle calculation unit 60 is used.

  That is, in step S <b> 14, first, the value of the integral calculation unit 55 of the anti-noise angle calculation unit 50 is substituted into the integral calculation unit 62 of the high response angle calculation unit 60. This process is a process for suppressing a decrease in calculation accuracy of the rotation angle θ2 at the time of switching. That is, in order for the integral calculation unit 62 to calculate a correct value, the integral calculation needs to be performed over a certain time in a state where the rotation speed θ2 is adopted. The value is not correct. For this reason, the value of the integral calculation part 62 is substituted. At this time, as the initial value of the double integral term of the speed calculation unit 61, the rotational speed ω1 of the noise resistant angle calculation unit 50 is set, and the integral term of the speed calculation unit 61 is initialized. In step S <b> 16, the output of the selector 72 is switched to the output of the high response angle calculation unit 60.

  On the other hand, when it is determined in step S10 that the output of the high response angle calculation unit 60 is selected, it is determined in step S20 whether or not the rotational speed ω2 is equal to or higher than the predetermined speed β. This process determines whether or not it is time to switch to the noise-resistant angle calculation unit 50. The predetermined speed β is higher than the predetermined speed α. As described above, the condition for switching from the high response angle calculation unit 60 to the noise resistance angle calculation unit 50 is different from the condition for switching from the noise resistance angle calculation unit 50 to the high response angle calculation unit 60, thereby enabling the high response angle calculation unit. 60 and frequent switching between the noise-proof angle calculation unit 50 are avoided.

  When it is determined that the rotational speed ω2 is greater than the predetermined speed β, in step S22, as the initial value of the integration calculation unit 55 of the noise-resistant angle calculation unit 50, the integration of the integration calculation unit 62 of the high response angle calculation unit 60 is performed. Assign a value. The purpose of this process is the same as the purpose of the process in step S14. At this time, the speed correction amount calculation unit 53 and the angle correction amount calculation unit 56 are initialized. In step S <b> 24, the output of the selector 72 is switched to the output of the noise resistant angle calculation unit 50.

  According to the embodiment described in detail above, the following effects can be obtained.

  (1) A noise-resistant angle calculation unit 50 and a high response angle calculation unit 60 are provided, and by selectively using these, it is possible to achieve both the improvement of resistance to noise and the improvement of responsiveness. .

  (2) In the high response angle calculation unit 60, the rotational speed ω2 is calculated based on the multiple integration calculation with the error equivalent amount Δθ as an input, and the rotation angle θ2 is calculated based on the integration calculation. Thereby, even when the rotation speed changes in a ramp shape, it is possible to avoid a steady deviation in the rotation angle.

  (3) In the noise-proof angle calculation unit 50, the temporary rotation speed ω1p is calculated by differential calculation of the rotation angle θ1, and the value subjected to the filter processing by the low-pass filter 52 is corrected according to the error equivalent amount Δ. Thereafter, the rotation angle θ1 was calculated by performing an integral calculation. Thereby, higher-order noise components can be suitably removed from the rotation angle θ1.

  (4) The noise-proof angle calculation unit 50 calculates the rotation angle θ1 by correcting the temporary rotation angle θ1p calculated based on the integral calculation according to the error equivalent amount Δθ. Thereby, the degree of freedom in control design can be improved as compared with the case where the rotation angle θ1p calculated from the integral calculation is directly used.

  (5) At the time of switching, it was adopted before switching as the initial value of the integral calculation (the initial value of the integral calculation unit 55 or the initial value of the integral calculation unit 62) which is adopted after switching as a means for calculating the rotation angle. The value of the integral calculation (the value of the integral calculation unit 62 or the value of the integral calculation unit 55) was set. Thereby, the value of the integral calculation part 55 or the value of the integral calculation part 62 can be made into an appropriate value immediately after switching, and the calculation accuracy of the rotation angle can be maintained high.

  (6) When the rotation speed of the electric motor 10 is equal to or higher than a predetermined value, the output of the noise resistant angle calculation unit 50 is selected. Thereby, switching can be performed appropriately.

  (7) The conditions for switching from the noise resistant angle calculation unit 50 to the high response angle calculation unit 60 and the conditions for switching from the high response angle calculation unit 60 to the noise resistance angle calculation unit 50 are different from each other. As a result, frequent switching can be avoided.

(8) The motor 10 having a saliency in its structure is used to superimpose a high-frequency signal that vibrates in a direction in which the inductance is minimized, and the error equivalent amount Δθ is calculated based on the vibration direction of the high-frequency signal actually propagated by the superposition. Calculated. Thereby, the error equivalent amount Δθ can be appropriately calculated.
(Second Embodiment)
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 14 shows processing of the position / velocity calculation unit 46 according to the present embodiment. As shown in the figure, in the present embodiment, the high response angle calculation unit 60 includes an angle correction amount calculation unit 63. The angle correction amount calculation unit 63 calculates the angle correction amount by performing a proportional integral operation on the error equivalent amount Δθ. Then, the angle correction unit 64 outputs the rotation angle θ2 obtained by correcting the output of the integration calculation unit 62 with the angle correction amount.

  Thus, by providing the angle correction amount calculation unit 63, the proportional gain and integral gain of the angle correction amount calculation unit 63 can be newly used as parameters for control design. For this reason, it is possible to improve the degree of freedom of control design, for example, it is possible to individually design stability and responsiveness together with other gains.

  According to this embodiment described above, in addition to the effects (1) to (8) of the first embodiment, the following effects can be obtained.

(9) In the high response angle calculator 60, the output of the integral calculator 62 is corrected based on the error equivalent amount Δθ. Thereby, the freedom degree of design of control can be improved.
(Third embodiment)
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 15 shows a processing procedure for switching between the noise-resistant angle calculation unit 50 and the high response angle calculation unit 60 according to the present embodiment. This process is repeatedly executed by a microcomputer or the like, for example, at a predetermined cycle. In FIG. 15, the same processes as those shown in FIG. 13 are denoted by the same reference numerals for the sake of convenience.

  As illustrated, in the present embodiment, switching is performed according to the amount of change in rotational speed. That is, when the amount of change | Δω1 | of the rotational speed ω1 is equal to or greater than the predetermined amount γ, switching is made to the high response angle calculation unit 60, and when the amount of change | Δω2 | Switch to the calculation unit 50. Thereby, the high response angle calculation unit 60 can be used when the rotation speed changes and responsiveness is required.

  According to this embodiment described above, in addition to the effects (1) to (5), (7), and (8) of the first embodiment, the following effects can be obtained. .

(10) The high response angle calculation unit 60 was used when the amount of change in rotational speed was greater than or equal to a predetermined value. Accordingly, the high response angle calculation unit 60 can be used when high responsiveness is required.
(Fourth embodiment)
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the second embodiment.

  In the present embodiment, part of the processing is shared between the noise-resistant angle calculation unit 50 and the high response angle calculation unit 60. FIG. 16 shows processing of the position / velocity calculation unit 46 according to the present embodiment. In FIG. 16, processes corresponding to the processes shown in FIG. 14 are given the same reference numerals for convenience.

As shown in the figure, in the present embodiment, either the output of the speed correction unit 54 of the noise resistant angle calculation unit 50 or the output of the speed calculation unit 61 of the high response angle calculation unit 60 is input to the integration calculation unit 62. Is switched by the selector 72. Thereby, it is not necessary to initialize the integral calculation unit 62 when switching. For this reason, when switching from the noise resistant angle calculation unit 50 to the high response angle calculation unit 60, the initial value of the double integration of the speed calculation unit 61 is used as the output of the selector 72 immediately before switching, and the integral term is initialized. do it. Further, when switching from the high response angle calculation unit 60 to the noise resistant angle calculation unit 50, the integral term of the speed correction amount calculation unit 53 may be initialized. As described above, according to the present embodiment, the integration calculation unit 62 and the angle correction amount calculation unit 63 are shared between the noise-resistant angle calculation unit 50 and the high response angle calculation unit 60, thereby performing the processing for switching. It can be simplified.
(Fifth embodiment)
Hereinafter, a fifth embodiment will be described with reference to the drawings, focusing on differences from the first embodiment.

  FIG. 17 shows the overall configuration of the control system according to the present embodiment. Note that, in FIG. 17, portions corresponding to those in FIG.

  As shown in the figure, in this embodiment, the output of the error equivalent amount detection unit 44 is subjected to moving average processing by the moving average processing unit 80, and the output of the moving average processing unit 80 is used as the error equivalent amount Δθ to calculate the position / velocity. Input to the unit 46. As a result, for example, the error equivalent amount Δθ can be calculated while suitably removing noise mixed in the current signals ihα and ihβ that are the output of the high-frequency current detection unit 42 due to noise included in the current sensor or the like. . Here, the moving average processing unit 80 performs a moving average process based on sampling values at time intervals “n (n: integer)” times the period of the voltage signal vhdc for the output of the error equivalent amount detection unit 44. Thereby, noise can be sufficiently removed from the error equivalent amount Δθ. Here, the noise removal effect increases as the integer “n” that defines the time interval for performing the moving average process is increased, but this increases the delay amount. For this reason, the integer “n” is reduced as the rotational speed of the electric motor 10 increases.

  According to this embodiment described above, in addition to the effects (1) to (8) of the first embodiment, the following effects can be obtained.

(11) The error equivalent amount Δθ is calculated by subjecting the output of the error equivalent amount detection unit 44 to a moving average process. Thereby, the error amount Δθ can be calculated while suitably removing the noise component included in the current sensor or the like.
(Sixth embodiment)
Hereinafter, the sixth embodiment will be described with reference to the drawings with a focus on differences from the fifth embodiment.

  FIG. 18 shows the overall configuration of the control system according to the present embodiment. Note that, in FIG. 17, portions corresponding to those in FIG.

  As shown in the figure, in the present embodiment, the current signals ihα and ihβ, which are the outputs of the high-frequency current detection unit 42, are subjected to a moving average processing by the moving average processing unit 82, and the output of the moving average processing unit 82 corresponds to the above error equivalent amount. Input to the detection unit 44. As a result, for example, the error equivalent amount Δθ can be calculated while suitably removing noise mixed in the current signals ihα and ihβ that are the output of the high-frequency current detection unit 42 due to noise included in the current sensor or the like. . Here, the moving average processing unit 82 also performs moving average processing based on sampling values at time intervals “n” times the period of the voltage signal vhdc for the current signals ihα and ihβ. Thereby, noise can be sufficiently removed from the error equivalent amount Δθ. Here, the noise removal effect increases as the integer “n” that defines the time interval for performing the moving average process is increased, but this increases the delay amount. For this reason, the integer “n” is reduced as the rotational speed of the electric motor 10 increases.

Also by this embodiment described above, the same effects as those of the fifth embodiment can be obtained.
(Seventh embodiment)
Hereinafter, the seventh embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  FIG. 19 shows the overall configuration of the control system according to the present embodiment. In FIG. 19, portions corresponding to those in FIG. 1 are given the same reference numerals for convenience.

  As illustrated, in the present embodiment, the amplitude calculation unit 84 calculates the amplitude N of the three-phase command current based on the command currents idc and iqc output from the command current setting unit 24. Then, the division unit 86 outputs a value obtained by dividing the output Δ of the error equivalent amount detection unit 44 by the amplitude N to the position / speed calculation unit 46 as the error equivalent amount Δθ. Thereby, the calculation gain of the temporary rotation angle θ can be made substantially constant regardless of the values of the command currents idc and iqc.

  That is, as the amplitude of the three-phase command current increases, magnetic saturation is likely to occur, and thus the amplitude of the current that actually propagates through the electric motor 10 tends to increase. For this reason, even if the error of the rotation angle θ is constant, the output Δ of the error equivalent amount detection unit 44 tends to increase as the amplitude of the command current increases. This means that the calculation gain of the rotation angle θ varies according to the amplitude of the command current. On the other hand, by dividing the output Δ of the error equivalent amount detection unit 44 by the amplitude N, the output Δ is pseudo-normalized, and the fluctuation of the error equivalent amount Δθ caused by the fluctuation of the amplitude N can be suppressed. it can. For this reason, regardless of the amplitude N of the command current, the calculation gain of the rotation angle θ can be made substantially constant.

  According to this embodiment described above, in addition to the effects (1) to (8) of the first embodiment, the following effects can be obtained.

(12) The error equivalent amount Δθ is calculated by dividing the output Δ of the error equivalent amount detection unit 44 by the amplitude N of the command current. Thereby, the change of the error equivalent amount Δθ due to the change of the amplitude N can be suitably suppressed or eliminated.
(Eighth embodiment)
Hereinafter, the eighth embodiment will be described with reference to the drawings with a focus on differences from the previous seventh embodiment.

  FIG. 20 shows the overall configuration of the control system according to the present embodiment. In FIG. 20, portions corresponding to those in FIG. 19 are given the same reference numerals for convenience.

  As shown in the figure, in the present embodiment, the division unit 88 divides the output Δ of the error equivalent amount detection unit 44 by the amplitude ihn of the high frequency signal determined from the high frequency currents ihα and ihβ, thereby obtaining the error equivalent amount Δθ. calculate. Thereby, the calculation gain of the temporary rotation angle θ can be made substantially constant regardless of the values of the command currents idc and iqc.

  That is, as the amplitude of the three-phase command current increases, magnetic saturation is likely to occur, and thus the amplitude of the current that actually propagates through the electric motor 10 tends to increase. For this reason, even if the error of the rotation angle θ is constant, the amplitude ihn tends to increase as the amplitude of the command current increases, and as a result, the output Δ of the error equivalent amount detection unit 44 tends to increase. This means that the calculation gain of the rotation angle θ varies according to the amplitude of the command current. On the other hand, by dividing the output Δ of the error equivalent amount detection unit 44 by the amplitude ihn, the output Δ is pseudo-normalized, and the fluctuation of the error equivalent amount Δθ caused by the fluctuation of the command currents idc and iqc is suppressed. can do.

Also by this embodiment described above, the same effects as those of the previous seventh embodiment can be obtained.
(Ninth embodiment)
Hereinafter, the ninth embodiment will be described with reference to the drawings, centering on differences from the previous seventh embodiment.

  FIG. 21 shows the overall configuration of the control system according to the present embodiment. In FIG. 21, the same reference numerals are given for the sake of convenience for the portions corresponding to FIG. 19.

  As shown in the figure, in this embodiment, the multipliers 90 and 92 input the high frequency currents ihα and ihβ divided by the amplitude N of the command current output from the amplitude calculator 84 to the error equivalent amount detector 44. To do. As a result, the error equivalent amount detection unit 44 calculates the error equivalent amount Δθ based on the outer product value of the outputs of the multipliers 90 and 92 and the voltage signals vhαc and vhβc.

Also by this embodiment described above, the same effects as those of the previous seventh embodiment can be obtained.
(Tenth embodiment)
Hereinafter, the tenth embodiment will be described with reference to the drawings with a focus on differences from the previous eighth embodiment.

  FIG. 22 shows the overall configuration of the control system according to the present embodiment. Note that, in FIG. 22, the same reference numerals are given for the sake of convenience for the portions corresponding to FIG.

  As shown in the figure, in the present embodiment, the multipliers 94 and 96 divide the high-frequency currents ihα and ihβ by the amplitude ihn calculated from the high-frequency currents ihα and ihβ and input the error equivalent amount detection unit 44. As a result, the error equivalent amount detection unit 44 calculates the error equivalent amount Δθ based on the outer product value of the outputs of the multipliers 94 and 96 and the voltage signals vhαc and vhβc.

According to the present embodiment described above, the same effect as that of the eighth embodiment can be obtained.
(Eleventh embodiment)
Hereinafter, the eleventh embodiment will be described with reference to the drawings, centering on differences from the first embodiment.

  FIG. 23 shows the overall configuration of the control system according to the present embodiment. In FIG. 23, portions corresponding to those in FIG. 1 are given the same reference numerals for convenience.

  As illustrated, in the present embodiment, a high-frequency current setting unit 100 is provided to superimpose a high-frequency signal on the output signal of the inverter 34. In the high-frequency current setting unit 100, the αβ conversion unit 41a is superimposed on the current signal ihdc1 as a high-frequency signal in the direction estimated as the d-axis direction by the temporary rotation angle θp, that is, the direction in which the inductance is assumed to be minimum. The current signal ihdc1 is output. The αβ converter 41a converts the current signal ihdc1 into current signals ihαc and ihβc on the αβ axis based on the temporary rotation angle θp. These current signals ihαc and ihβc are output to the dq converter 41b. The dq converter 41b converts the current signals ihαc and ihβc into current signals ihdc2 and ihqc2 on the dq axis based on the rotation angle θ. These current signals ihdc2 and ihqc2 are superimposed on the command currents idc and iqc through the adders 25a and 25b.

  As described above, the command voltage setting unit 26 determines the command voltage vdc on the d axis based on the difference between the command current idc (specifically, the command current idc on which the current signal ihdc2 is superimposed) and the actual current id on the d axis. The command voltage vqc on the q axis is set based on the difference between the command current iqc (specifically, the command current iqc on which the current signal ihqc2 is superimposed) and the actual current iq on the q axis. Thereby, the command voltages vdc and vqc are used to superimpose the current signal ihdc1 in the direction assumed to be the d-axis direction by the temporary rotation angle θp.

  On the other hand, the dq converter 22a converts these into actual currents id and iq without removing high frequency components from the actual currents iu and iv. Therefore, when the current signal ihdc is superimposed on the command voltages vdc and vqc (more precisely, when the command voltage is set to superimpose the current signal ihdc2), the frequency signal that actually flows through the motor 10 is It is mixed.

  The command voltages vdc and vqc are converted by the αβ converter 27 into a command voltage vαc on the α axis and a command voltage vβc on the β axis based on the rotation angle θ. Then, the high frequency voltage detection unit 102 extracts the voltage signal vhα on the α axis and the voltage signal vβc on the β axis according to the high frequency signal actually propagated through the electric motor 10 from the command voltages vαc and vβc.

  On the other hand, the error equivalent amount detection unit 44 calculates an outer product value of the vector signals of the voltage signals vhαc and vhβc and the vector signals of the current signals ihαc and ihβc. Then, the position / velocity calculation unit 46 calculates the temporary rotation angle θp so that the outer product value is zero. As a result, the temporary rotation angle θp is set to a value where the position where the inductance of the electric motor 10 is minimized is zero. The calculation method of the temporary rotation angle θp is in accordance with the calculation method by the position / velocity calculation unit 46 shown in FIG.

  The position correction unit 104 calculates the estimated value Te of the output torque and the phase φ of the current vector of the actual currents iq and iq based on the actual currents id and iq, and sets the target amplitude values of the voltage signals vhα and vhβ based on these. . Then, a correction amount cor for setting the difference between the actual amplitude vhn detected by the high-frequency voltage detection unit 102 and the target amplitude value to zero is set.

  Also according to the present embodiment described above, it is possible to obtain effects according to the above-described effects of the first embodiment.

(Other embodiments)
Each of the above embodiments may be modified as follows.

  In each of the above embodiments, when switching from the noise resistant angle calculating unit 50 to the high response angle calculating unit 60, the initial value of the double integral term of the speed calculating unit 61 is the rotational speed calculated by the noise resistant angle calculating unit 50. However, it is not limited to this. For example, the initial value of the integral term may be used as the rotational speed calculated by the noise resistant angle calculation unit 50 and the double integral term may be initialized.

  The switching method between the high response angle calculation unit 60 and the noise resistant angle calculation unit 50 is not limited to the above. For example, in the case of controlling the rotational speed of the electric motor 10 to the command speed, the switching may be performed based on the command speed.

  -You may change previous 11th Embodiment by the change of the 2nd-10th Embodiment with respect to previous 1st Embodiment.

  The high response angle calculation unit 60 in the previous third, fifth to eleventh embodiments may be configured as in the previous second embodiment.

  In the fourth embodiment, the angle correction amount calculation unit 63 may be omitted.

  The switching method in the previous fifth to eleventh embodiments may be changed to the method in the previous third embodiment.

  -You may change previous 2nd-4th embodiment by the change of either of the 5th-10th embodiment with respect to previous 1st Embodiment.

  In the sixth and eighth embodiments, the amplitude of the actual current may be used instead of the amplitude of the command current.

  The low pass filter 52 may be provided upstream of the differential calculation unit 51.

  The low-pass filter 52 is not limited to a first-order lag filter. For example, a secondary delay filter may be used. At this time, it is desirable to have a design that can remove higher-order vibration components such as sixth-order harmonic noise of the rotation period of the electrical angle. Again, as in the above embodiment, it is desirable to variably set the cutoff frequency according to the rotational speed. However, the filter means is not limited to one that variably sets the cutoff frequency according to the rotation speed.

  The high response angle calculation unit 60 is not limited to one that does not include filter means. For example, in the previous fifth embodiment, since the moving average processing unit 80 is a filter means, if the moving average processing unit 80 is included in the position / velocity calculating unit 46, the noise-resistant angle calculating unit 50 and Both high response angle calculation units 60 are configured to include filter means. However, even in this case, the anti-noise angle calculation unit 50 performs processing with a higher degree of noise removal than the high response angle calculation unit 60, but the responsiveness decreases. Switching is effective.

  The speed correction amount calculation unit 53 is not limited to outputting the sum of the proportional calculation and the integral calculation of the error equivalent amount Δθ, and may be configured to output the sum of the proportional calculation, the integral calculation, and the differential calculation, for example. . Alternatively, an integration calculation result of the error equivalent amount Δθ may be output.

  The angle correction amount calculation unit 56 is not limited to one that outputs the sum of the proportional calculation and integral calculation of the error equivalent amount Δθ. For example, the structure which outputs the sum of a proportional calculation, an integral calculation, and a differential calculation may be sufficient. Alternatively, an integration calculation result of the error equivalent amount Δθ may be output. Further, a proportional calculation result of the error equivalent amount Δθ may be output.

  The configuration of the speed calculation unit 61 is not limited to the one that calculates the sum of the double integral calculation, the proportional calculation, and the integral calculation of the error equivalent amount Δθ. For example, a sum of double integral calculation, proportional calculation, integral calculation, and differential calculation of the error equivalent amount Δθ may be output. Alternatively, the sum of the double integral calculation and the integral calculation of the error equivalent amount Δθ may be output. Furthermore, the sum of the double integral calculation and the proportional calculation of the error equivalent amount Δθ may be output. Further, a triple integral calculation result of the error equivalent amount Δθ may be output.

  The angle correction amount calculation unit 63 is not limited to the one that outputs the sum of the proportional calculation and integral calculation of the error equivalent amount Δθ. For example, it may be configured to output the sum of proportional calculation, integral calculation, and differential calculation of the error equivalent amount Δθ. Alternatively, an integration calculation result of the error equivalent amount Δθ may be output. Further, a proportional calculation result of the error equivalent amount Δθ may be output.

  -Due to the structure, the electric motor having saliency is not limited to the electric motor 10 described above. For example, an embedded magnet synchronous motor (IPMSM) or a synchronous reluctance motor (SynRM) may be used.

  -The rotating machine is not limited to an electric motor but may be a generator.

  ・ As a method for acquiring information related to the rotation angle of the rotating machine based on the electrical state quantity of the rotating machine, it is based on the frequency signal that is actually propagated through the motor when the frequency signal is superimposed on the output signal of the inverter. Not exclusively. For example, it may be detected based on an induced voltage (see, for example, “Japanese Patent Laid-Open No. 2006-230120” or “Electronic Engineering Handbook 6th Edition, Electrical Society of Japan, Volume 21, 3.6”).

  In each of the above embodiments, the control device according to the present invention is applied to a hybrid vehicle. Furthermore, the control device of the present invention may be applied to an electric motor as power transmission means such as power steering in a vehicle using an internal combustion engine as a power source.

The figure which shows the whole structure of the control system of the electric motor concerning 1st Embodiment. The figure which shows the rotor of the electric motor concerning the embodiment. The figure which shows the problem regarding the detection of a rotation angle. The figure which shows the problem which the calculation method of the rotation angle using the deflection | deviation of the electric current in an electric motor crawls. The figure which shows the relationship between the phase (phi) of the electric current for a drive, output torque, and a target amplitude value. The figure which shows the calculation process of the correction amount of the rotation angle based on the said target amplitude value. The figure which shows the aspect of the setting process of the said rotation angle. The block diagram which shows the process of the position / speed calculation part concerning the said embodiment. The figure which shows the gain characteristic of the low pass filter concerning the embodiment. The time chart which shows the calculation result of the rotational speed and angle by the noise-proof angle calculation part and high response angle calculation part concerning the embodiment. The figure which shows the calculation aspect of the rotation angle by the noise-proof angle calculation part and high response angle calculation part concerning the embodiment. The time chart which compares the calculation aspect of the rotation angle and rotation speed by a high response angle calculation part and a prior art. The flowchart which shows the procedure of the switching process of the noise-proof angle calculation part in the said embodiment, and a high response angle calculation part. The block diagram which shows the process of the position / velocity calculation part concerning 2nd Embodiment. The flowchart which shows the procedure of the switching process of the noise-proof angle calculation part and high response angle calculation part in 3rd Embodiment. The block diagram which shows the process of the position / velocity calculation part concerning 4th Embodiment. The figure which shows the whole structure of the control system of the electric motor concerning 5th Embodiment. The figure which shows the whole structure of the control system of the electric motor concerning 6th Embodiment. The figure which shows the whole structure of the control system of the electric motor concerning 7th Embodiment. The figure which shows the whole structure of the control system of the electric motor concerning 8th Embodiment. The figure which shows the whole structure of the control system of the electric motor concerning 9th Embodiment. The figure which shows the whole structure of the control system of the electric motor concerning 10th Embodiment. The figure which shows the whole structure of the control system of the electric motor concerning 11th Embodiment.

Explanation of symbols

  DESCRIPTION OF SYMBOLS 10 ... Electric motor, 44 ... Error equivalent amount detection part (one embodiment of error equivalent amount calculation means), 50 ... Noise resistance angle calculation part (one embodiment of noise resistance angle calculation means), 60 ... High response angle calculation part ( One embodiment of high response angle calculation means).

Claims (14)

In driving a rotating machine by operating a switching element of an inverter, in a control device for a rotating machine that acquires information on a rotation angle of the rotating machine based on an electrical state quantity of the rotating machine,
An error equivalent amount calculating means for calculating an error equivalent amount for the currently acquired rotation angle based on the electrical state amount;
High response angle calculation means for calculating a rotation angle of the rotating machine based on the error equivalent amount;
Noise-resistant angle calculation means for calculating a rotation angle of the rotating machine based on the error equivalent amount by processing having a higher degree of noise removal than the high response angle calculation means;
The rotating machine includes: a switching unit that switches between the output of the noise-resistant angle calculating unit and the output of the high response angle calculating unit as a rotation angle used for driving the rotating machine. Control device. In driving a rotating machine by operating a switching element of an inverter, in a control device for a rotating machine that acquires information on a rotation angle of the rotating machine based on an electrical state quantity of the rotating machine,
An error equivalent amount calculating means for calculating an error equivalent amount for the currently acquired rotation angle based on the electrical state amount;
Noise-resistant angle calculation means comprising a filter means and calculating the rotation angle of the rotating machine based on the error equivalent amount;
High response angle calculation means for calculating the rotation angle of the rotating machine based on the error equivalent amount without providing a filter means;
The rotating machine includes: a switching unit that switches between the output of the noise-resistant angle calculating unit and the output of the high response angle calculating unit as a rotation angle used for driving the rotating machine. Control device.   The high response angle calculation means includes a speed calculation means for calculating one of a rotational speed and an equivalent value based on a multiple integration calculation with the error equivalent amount as an input, and an integral calculation of an output of the speed calculation means. The rotating machine control device according to claim 1, further comprising a means for calculating a rotation angle.   4. The control device for a rotating machine according to claim 3, wherein the high response angle calculation means further comprises correction means for correcting the rotation angle calculated based on the integral calculation based on the error equivalent amount.   The noise resistant angle calculating means includes speed calculating means for calculating either the rotational speed of the rotating machine or an equivalent value thereof by differential calculation and filtering of the rotational angle output from the noise resistant angle calculating means, and the error equivalent Correction means for correcting the output of the speed calculation means based on the amount; and means for calculating a rotation angle as an output of the noise-proof angle calculation means based on an integral operation of the corrected output. The control apparatus of the rotary machine in any one of Claims 1-4.   The noise-resistant angle calculation unit calculates a rotation angle as an output of the noise-proof angle calculation unit by correcting a rotation angle calculated based on the integral calculation based on the error equivalent amount. Item 6. A rotating machine control device according to Item 5. The high response angle calculation means and the noise-resistant angle calculation means both calculate the rotation angle based on an integral operation of a rotation speed calculated based on the electrical state quantity and an equivalent value thereof. Yes,
The switching means is characterized in that, at the time of the switching, as the initial value of the integration calculation that is adopted after switching as the means for calculating the rotation angle, the value of the integration calculation that was adopted before the switching is set. The control device for a rotating machine according to any one of claims 1 to 6.   The high response angle calculation means and the noise-resistant angle calculation means both calculate the rotation angle based on an integral operation of a rotation speed calculated based on the electrical state quantity and an equivalent value thereof. The rotating machine control device according to claim 1, wherein a unit for performing the processing related to the integration calculation is shared.   The switching unit selects an output of the noise-resistant angle calculation unit as a rotation angle used for driving the rotating machine when a rotation speed of the rotating machine is equal to or higher than a predetermined value. The rotating machine control device according to claim 8.   10. The switching means selects an output of the noise-proof angle calculation means as a rotation angle used for driving the rotating machine under a situation where the rotation speed of the rotating machine is stable. The control apparatus of the rotary machine in any one of.   The switching unit selects an output of the high response angle calculation unit as a rotation angle used for driving the rotating machine when a change amount of the rotating speed of the rotating machine is equal to or larger than a predetermined value. The control device for a rotating machine according to any one of claims 1 to 10.   The switching unit sets a condition for switching from the noise resistant angle calculating unit to the high response angle calculating unit and a condition for switching from the high response angle calculating unit to the noise resistant angle calculating unit. The control device for a rotating machine according to claim 1, wherein the control device is a rotary machine. The rotating machine has saliency in its structure,
And a superimposing unit that superimposes on the output signal of the inverter a frequency signal having a period different from the rotation period of the electrical angle of the rotating machine and oscillating in an arbitrary phase angle direction based on the information on the rotation angle. ,
The control apparatus for a rotating machine according to any one of claims 1 to 12, wherein the error equivalent amount calculating means calculates the error equivalent amount based on a vibration direction of a frequency signal actually propagated by the superposition. . A moving average processing means for applying a moving average process to at least one of the detected value of the actually propagated frequency signal and the output of the error equivalent amount calculating means;
14. The rotating machine control device according to claim 13, wherein the moving average processing means performs the moving average processing based on a time interval that is an integral multiple of a cycle of the frequency signal superimposed by the superimposing means.

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