JP4943383B2 - Brushless motor control device and electric steering device - Google Patents

Description

  The present invention relates to a brushless motor control device and an electric steering device.

2. Description of the Related Art Conventionally, for example, when estimating a rotation angle in a state where a brushless motor is rotating, a method of estimating a rotation angle from fluctuations in induced voltage is known (for example, see Non-Patent Document 1).
Further, conventionally, for example, when estimating the rotation angle of the magnetic pole of the rotor from the predetermined reference rotation position when the brushless motor is stopped, a pulse voltage that does not contribute to the rotation of the rotor is applied. Applying to the motor, the self-inductance is obtained by Fourier transform from the current and voltage applied to the brushless motor at this time, and using the fact that the self-inductance changes according to the rotation angle of the rotor due to the saliency of the rotor, A method for estimating a rotor angle, that is, a rotation angle of a magnetic pole of a rotor from a predetermined reference rotation position is known (see, for example, Non-Patent Document 2).
Conventionally, for example, two different estimation calculation processes for estimating the rotation angle of a brushless motor, that is, the rotation angle is determined according to the response current (or response voltage) when a high frequency voltage (or high frequency current) is applied to the brushless motor. One of the two estimated values obtained by the estimation process and the process of estimating the rotation angle according to the induced voltage and energization current of the brushless motor is used as the vehicle steering torque, speed, engine speed, steering angle. There is known an electric power steering device that is switched and selected in accordance with, for example (see Patent Document 1).
Shinnaka, “Proposal of Minimum D-factor State Observer for Permanent Magnet Synchronous Motor and Sensorless Vector Control Method Using It”, IEEE Trans. IA, Vol. 123, no. 12, 2003, p1446-1460 Yamamoto, et al., “Initial magnetic pole position estimation method for surface magnet synchronous motor using pulse voltage”, IEEE Trans. IA, Vol. 125, no. 3,2005, p253-258 JP 2007-307940 A

By the way, like the electric power steering device according to Patent Document 1 of the above-described prior art, a state in which the brushless motor is stopped in a state where the brushless motor is rotating according to the driving state of the vehicle. Therefore, when switching between suitable estimation calculation processes, there is a possibility that a problem that the switching angle cannot be properly switched and the estimation accuracy of the rotation angle is lowered may occur.
For example, when a small lane change is performed while the vehicle is running, the brushless motor (that is, the steering wheel) is rotating even if the steering torque is relatively small. In the case of steady circular turning, the brushless motor (that is, the steering handle) is stopped even if the steering torque is relatively large.
When it is determined that the brushless motor is stopped when the steering torque is small, and the rotation angle estimated value by a suitable estimation calculation process is employed in the stopped state despite the rotation, There is a problem in that it is difficult to obtain an accurate rotation angle without obtaining a precise rotation angle, and steering performance is deteriorated.
This is based on the response current (or response voltage) when a high-frequency voltage (or high-frequency current) is applied to the brushless motor in a state where it is stopped. The motor rotates in response to this response. This is because the influence of the induced voltage caused by the operation is added, so that a precise rotation angle cannot be obtained.

  The present invention has been made in view of the above circumstances, and while the rotation angle of the brushless motor is estimated quickly and accurately, the brushless motor steps out and excessive torque fluctuation occurs or the brushless motor stops. By providing a control device for a brushless motor capable of preventing the occurrence of the problem and by using a rotation sensor, the vehicle mountability is improved, the reliability is high, and the driver can steer while ensuring the desired quietness. An object of the present invention is to provide an electric steering device capable of preventing a feeling of discomfort from feeling.

In order to solve the above-described problems and achieve the object, a brushless motor control device according to a first aspect of the present invention includes a rotor (for example, the rotor 63 in FIG. 3 of the embodiment) and a multi-phase stator winding. Brushless motor (eg, in FIG. 3 of the embodiment) having a stator (eg, stator 64 in FIG. 3 of the embodiment) having a wire (eg, stator winding 64a in FIG. 3 of the embodiment) Control of brushless motor provided with motor 31) and energization switching means (for example, PWM signal generation unit 94 and FET bridge 72 in FIG. 4 of the embodiment) that switches and controls energization to the stator windings of each phase. an apparatus sets the γδ coordinate system having a phase difference with respect to the dq coordinate system, and the induced voltage estimation unit that estimates an induced voltage component generated on the induced voltage component and δ-axis generated γ-axis Based on whether the sum of the squares of the induced voltage component generated in the square and the δ-axis induced voltage component generated in the γ axis are estimated by the induced voltage estimation unit is larger than a predetermined value, the brushless Determination means for determining rotation and stop of the motor (for example, stop determination unit 163 in FIG. 14 of the embodiment).

  Further, the brushless motor control device according to the second aspect of the present invention provides a rotation angle estimating means for estimating a rotation angle when the brushless motor is rotating and outputting an estimated rotation angle signal (for example, an embodiment). 4 when the brushless motor is determined to be stopped by the determination means, the rotation angle is estimated by a process different from the rotation angle estimation means. Drive control means (for example, the stop time estimator 96a and the control unit 73 in FIG. 4 of the embodiment) for driving and controlling the brushless motor.

The brushless motor control device according to the third aspect of the present invention includes a rotor (for example, the rotor 63 in FIG. 3 of the embodiment) and a multi-phase stator winding (for example, in FIG. 3 of the embodiment). A brushless motor (for example, the motor 31 in FIG. 3 of the embodiment) having a stator (for example, the stator 64 in FIG. 3 of the embodiment) having the stator winding 64a) and the stator windings of each phase 4 is a brushless motor control device comprising energization switching means (for example, the PWM signal generation unit 94 and the FET bridge 72 in FIG. 4 of the embodiment) for switching and controlling energization to the dq coordinate system. set the γδ coordinate system with phase difference, the induced voltage estimation unit that estimates an induced voltage component generated on the induced voltage component and δ-axis generated in the γ axis, convergence calculation for converging the phase difference to zero Based on the rotation angle estimation means (for example, the rotation time estimator 96b in FIG. 4 of the embodiment) that estimates the rotation angle of the brushless motor and outputs an estimated rotation angle signal, and is output from the rotation angle estimation means. Based on the estimated rotation angle signal, current control means for controlling energization by vector control using the γδ coordinate system (for example, the control unit 73 in FIG. 4 of the embodiment) and estimation by the induced voltage estimation means It has been based on whether the sum of the squares of the induced voltage component generated in the square and the δ-axis induced voltage component generated in the γ axis is larger than a predetermined value, determining the rotation and stop of the brushless motor Determination means (for example, stop determination unit 163 in FIG. 14 of the embodiment).

  Further, the brushless motor control device according to the fourth aspect of the present invention provides the rotation angle by a process different from the rotation angle estimation means when the determination means determines that the brushless motor is stopped. A stop-time rotation angle estimation unit (for example, a stop-time estimator 96a in FIG. 4 of the embodiment) that estimates and outputs a stop-time estimated rotation angle signal is provided.

In addition, in the electric steering apparatus according to the fifth aspect of the present invention, the brushless motor control apparatus according to the fourth aspect and steering torque detection means for detecting the steering torque and outputting a steering torque signal (for example, the embodiment) 1), the steering torque signal output from the steering torque detection means, and the estimated rotation angle signal or the stoppage estimated rotation angle signal output from the control device. Steering control means (for example, the control unit 73 in FIG. 4 of the embodiment) that drives and controls the brushless motor and generates auxiliary torque that assists the steering torque from the brushless motor.

  According to the brushless motor control device of the first aspect of the present invention, the induced voltage generated by the brushless motor can be accurately detected, and when the induced voltage is smaller than a predetermined value, the brushless motor is stopped. It can be determined appropriately.

More specifically, in the γδ coordinate system having the phase difference θe with respect to the dq coordinate system, the induced voltage generated in the γ axis due to the absolute value Eex of the induced voltage generated in the q axis (torque current axis) of the actual brushless motor. Is (Eex × sin θe), and the induced voltage generated on the δ axis is (Eex × sin θe).
By observing the sum of the square of the induced voltage (Eex × sin θe) generated on the γ axis and the square of the induced voltage (Eex × cos θe) generated on the δ axis, the induced voltage can be accurately determined regardless of the phase difference θe. Therefore, it is possible to detect the rotation and stop of the brushless motor appropriately. Then, an accurate estimated value of the rotation angle can be obtained according to the determination result.

  Furthermore, according to the control apparatus for a brushless motor according to the second aspect of the present invention, a precise rotation angle can be obtained regardless of whether the brushless motor is stopped or rotating. It can be estimated, and smooth drive control can be performed without stepping out the brushless motor or causing torque fluctuation.

  According to the brushless motor control device of the third aspect of the present invention, when the phase difference does not converge to zero during the execution of the rotation angle estimation calculation (particularly, the induced voltage generated by the brushless motor is small and the phase difference is small). Even if it is difficult to converge to zero), it is possible to accurately detect the induced voltage generated by the brushless motor. If this induced voltage is smaller than the predetermined value, it is appropriate that the brushless motor is stopped. Can be determined.

More specifically, in the γδ coordinate system having the phase difference θe with respect to the dq coordinate system, the induced voltage generated in the γ axis due to the absolute value Eex of the induced voltage generated in the q axis (torque current axis) of the actual brushless motor. Is (Eex × sin θe), and the induced voltage generated on the δ-axis is (Eex × cos θe).
In the process of estimating the rotation angle when the brushless motor is rotating, the induced voltage (Eex × sin θe) generated on the γ-axis and the induced voltage (Eex × cos θe) generated on the δ-axis are estimated. By calculating the ratio of the components of the induced voltage ((Eex × sin θe) / (Eex × cos θe)), (tan θe) is calculated, and the phase angle θe is estimated by the inverse calculation. For example, as shown in FIG. 23, the phase angle θe is estimated so as to converge to zero from an appropriate initial value between (−180 °) and (+ 180 °). Therefore, even if the induced voltage component generated on the γ-axis or δ-axis is observed, if the phase angle θe does not converge to zero, there is a problem that an induced voltage with high accuracy cannot be detected.

  Further, when the induced voltage generated by the brushless motor is relatively high, for example, as shown in FIG. 24, when a sufficient time has elapsed by the convergence calculation, the phase difference θe converges to zero, but the rotation speed is low. In some cases, the phase difference θe may not converge to zero even after a sufficient time has elapsed, and the induced voltage component generated on the γ axis or the induced voltage component generated on the δ axis has not converged. There arises a problem that it cannot be determined whether or not the vehicle is stopped.

  On the other hand, according to the brushless motor control device of the third aspect of the present invention, the square of the induced voltage (Eex × sin θe) generated on the γ axis and the induced voltage (Eex × cos θe) generated on the δ axis. By observing the sum with the square, the induced voltage can be accurately detected regardless of the phase difference θe, and the rotation and stop of the brushless motor can be appropriately determined. Then, an accurate estimated value of the rotation angle can be obtained according to the determination result.

  Furthermore, according to the brushless motor control device of the fourth aspect of the present invention, it is possible to estimate the rotation angle with high accuracy when the brushless motor is stopped and rotating, to step out the brushless motor, Smooth drive control can be performed without causing fluctuations.

  In addition, according to the electric steering apparatus of the fifth aspect of the present invention, the apparatus configuration can be simplified and the apparatus can be reduced in size without the need for providing a rotation angle sensor such as a resolver. In addition to being able to improve, for example, the brushless motor will step out and torque fluctuation will occur, the steering feeling will be lowered, and the brushless motor will step out and stop, for example. It is possible to prevent the driver from feeling uncomfortable with the steering feeling and to suppress the running behavior of the vehicle from becoming unstable.

DESCRIPTION OF EXEMPLARY EMBODIMENTS Hereinafter, an embodiment of a brushless motor control device and an electric steering device of the invention will be described with reference to the accompanying drawings.
The brushless motor control device 70 according to the present embodiment is mounted on an ECU (Electronic Control Unit) 50 of the electric steering device 1 as the vehicle steering device shown in FIG.

  As shown in FIG. 1, for example, the electric steering device 1 includes a steering shaft 3 connected to a steering wheel 2 of a vehicle and a universal shaft joint 4 connected to the steering shaft 3 to steering wheels (wheels) 5 and 5. The steering system includes a steering mechanism 7 housed in a housing 6 that constitutes a steering gear box, and a steering assist mechanism 8 that generates a steering assist force in the steering mechanism 7.

The steering mechanism 7 includes a rack and pinion mechanism 10, and a pinion shaft 11 of the rack and pinion mechanism 10 is connected to the universal joint 4.
The pinion 12 provided on the pinion shaft 11 and the rack 14 provided on the rack shaft 13 that can reciprocate in the vehicle width direction are meshed with each other.

The pinion shaft 11 is rotatably supported at its lower, middle, and upper portions by bearings 15 a, 15 b, and 15 c, for example, and the pinion 12 is provided at the lower end of the pinion shaft 11.
The rack shaft 13 is supported in a substantially cylindrical rack housing 6 a extending in the vehicle width direction of the housing 6 through a bearing 16 so as to be capable of reciprocating in the longitudinal direction of the shaft.

Both ends of the rack housing 6a have openings that open, and the end 13a of the rack shaft 13 protrudes from the opening.
A rack end plate 17 having an outer diameter larger than that of the rack shaft 13 is fixed to each end portion 13 a of the rack shaft 13, and a rack end head 18 is fixed to the rack end plate 17.
The rack end head 18 includes a ball joint 19, a tie rod 20 is connected to the ball joint 19, and a steering wheel (front wheel) 5 is linked to the tie rod 20.

On the outer peripheral surface near the opening at both ends of the rack housing 6a, an annular groove 6b protruding inward in the radial direction is formed.
An end of a bellows-shaped rack end cover 21 that can be expanded and contracted in the longitudinal direction of the rack shaft 13 is attached to the annular groove 6b of the rack housing 6a, and the end 13a of the rack shaft 13 and the rack end plate are mounted. 17, the rack end head 18, and the ball joint 19 are accommodated in a rack end cover 21, and the tie rod 20 penetrates the rack end cover 21 and protrudes outward.

The steering assist mechanism 8 includes a motor 31 including a brushless motor that generates a steering assist force for reducing the steering force by the steering wheel 2, a worm gear 32, and a worm wheel gear 33. The wheel gear 33 is accommodated in the housing 6 constituting the steering gear box.
The motor 31 is connected to a worm gear 32 that is pivotally supported by the housing 6, and the worm gear 32 meshes with a worm wheel gear 33 that is provided integrally with the pinion shaft 11. The worm gear 32 and the worm wheel gear 33 constitute a reduction mechanism, and torque generated by the motor 31 is boosted by the worm gear 32 and the worm wheel gear 33 and transmitted to the pinion shaft 11.

A magnetostrictive steering torque sensor 40 that detects a steering torque (steering input) based on a change in magnetic characteristics caused by magnetostriction is provided between the intermediate bearing 15b and the upper bearing 15c in the pinion shaft 11. Has been placed.
The steering torque sensor 40 includes two magnetostrictive films (for example, magnetic anisotropic films such as Ni-Fe plating) provided on the outer peripheral surface of the pinion shaft 11 so as to have anisotropy in opposite directions at a predetermined interval in the axial direction. Magnetostrictive films) 41, 42, two detection coils 43, 44 disposed opposite to the magnetostrictive films 41, 42, and detection circuits 45, 46 connected to the respective detection coils 43, 44. . Each detection circuit 45, 46 converts the change in inductance of each detection coil 43, 44 caused by the inverse magnetostriction characteristic generated when the steering torque is applied to each magnetostrictive film 41, 42 into a voltage change, and then changes the ECU. (Electric Control Unit) 50. The ECU 50 calculates the steering torque that acts on the steering shaft 3 based on the outputs of the detection circuits 45 and 46.

  Then, the ECU 50 determines a target current to be supplied to the motor 31 in accordance with the magnitude of the steering torque detected by the steering torque sensor 40 (that is, the steering torque input from the steering wheel 2 by the driver). For example, by performing control such as PID control so that the current flowing through the target 31 matches the target current, an auxiliary torque corresponding to the steering torque is generated from the motor 31 and this auxiliary torque is transmitted to the pinion shaft via the speed reduction mechanism. 11 is transmitted. Thereby, the steering assist force by the motor 31 acts in the same direction as the steering input by the driver, and the steered wheels 5 are steered by the combined torque obtained by adding the assist torque of the motor 31 to the driver's steering torque.

For example, as shown in FIG. 2, the motor 31 is attached to the side portion of the housing 6 so as to protrude from the housing 6 with a bolt, and closes the side opening of the housing 6. The motor 31 is attached to the lid 61 with a bolt. A bottom cylindrical motor case 62, a rotor 63 provided around the rotation axis O, and having a permanent magnet 63 a, are arranged opposite to each other in the radial direction so as to cover the outer periphery of the rotor 63, and rotate the rotor 63. And a stator 64 having a plurality of stator windings 64a for generating a rotating magnetic field.
The stator 64 is accommodated in, for example, a motor case 62 by press fitting or the like, and an output shaft 65 disposed coaxially with the rotation shaft O is fixed to the inner peripheral portion of the rotor 63.
The lid 61 and the motor case 62 of the motor 31 support the output shaft 65 through two bearings 66 so as to be rotatable.

For example, as shown in FIG. 3, the stator 64 of the motor 31 includes a plurality of divided cores 64b arranged in an annular shape, an insulating bobbin 64c, and a stator winding 64a wound around the bobbin 64c in a multiple manner. And is accommodated by press-fitting or the like in a motor case 62 molded by press molding or the like.
The split core 64b is configured, for example, by laminating a plurality of substantially T-shaped silicon steel plates in the direction of the rotation axis O, and includes an outer yoke portion 64b1 and an inner teeth portion 64b2. On both end surfaces in the circumferential direction of the yoke portion 64b1, a convex portion protruding in the circumferential direction is provided on one end surface, and a concave portion in which the convex portion can be fitted is provided on the other end surface, and adjacent in the circumferential direction. An annular yoke is formed by fitting the convex portion of one yoke portion 64b1 of the divided cores 64b and 64b into the concave portion of the other yoke portion 64b1. The teeth portion 64b2 has a smaller circumferential width than the yoke portion 64b1, and protrudes from the yoke portion 64b1 toward the radially inward rotor 63. And the bobbin 64c which consists of insulating resin materials etc. is mounted | worn with the teeth part 64b2.

The rotor 63 of the motor 31 includes, for example, a permanent magnet 63a, a magnet cover 63b, a back yoke 63c, and an output shaft 65.
The substantially cylindrical back yoke 63c is configured, for example, by laminating a plurality of substantially annular silicon steel plates in the direction of the rotation axis O, the output shaft 65 is mounted on the inner peripheral portion, and a predetermined interval in the circumferential direction on the outer peripheral surface. A plurality of permanent magnets 63a are arranged. And the magnet cover 63b is arrange | positioned so that the outer peripheral surface of the some permanent magnet 63a may be covered.

The output shaft 65 of the motor 31 is connected to the worm shaft 32a of the worm gear 32 through a coupling 67, for example, as shown in FIG.
The worm shaft 32 a is arranged coaxially with the output shaft 65 of the motor 31 and is rotatably supported by the housing 6 via two bearings 68. Of the two bearings 68 mounted in the housing 6, one of the bearings 68 on the motor 31 side is restricted by the retaining ring 69 from moving toward the motor 31 in the longitudinal direction of the shaft.

  In the electric steering device 1 according to the present embodiment, the brushless motor control device 70 includes, for example, an FET bridge 72 using a battery 71 as a DC power source and a control unit 73 as shown in FIG. It is equipped with.

In the brushless motor control device 70, the motor 31 is driven by the FET bridge 72 in response to a control command output from the control unit 73.
For example, as shown in FIG. 5, the FET bridge 72 includes a bridge circuit formed by bridge connection using a plurality of FETs (for example, MOSFETs: Metal Oxide Semiconductor Field Effect Transistors), and the bridge circuit performs pulse width modulation. It is driven by the (PWM) signal.
The bridge circuit includes, for example, a high-side and low-side U-phase transistor UH, UL, a high-side and low-side V-phase transistor VH, VL, and a high-side and low-side W-phase transistor WH, WL that are paired for each phase. Each transistor UH, VH, WH has a drain connected to the battery 71 (+ B) to form a high side arm, and each transistor UL, VL, WL has a source grounded and a low side arm For each phase, the sources of the high-side arm transistors UH, VH, WH and the low-side arm transistors UL, VL, WL are connected to each other.

  The FET bridge 72 is a gate signal (that is, a PWM signal) that is a switching command that is output from the control unit 73 and is input to the gates of the transistors UH, VH, WH, UL, VL, and WL when the motor 31 is driven, for example. ), The DC power supplied from the battery 71 is converted into the three-phase AC power by switching the on (off) / off (off) state of each transistor paired for each phase. By sequentially commutating the energization to the winding 64a, AC U-phase current Iu, V-phase current Iv and W-phase current Iw are energized to the stator winding 64a of each phase.

Note that the booster circuit 74 includes, for example, a capacitor and a charge pump circuit composed of a transistor, and instructs a boost signal of the booster circuit 74 to switch the on (conductive) / off (shutoff) state of each transistor. Signal) is input from the control unit 73.
The booster circuit 74 boosts the gate voltages of the transistors UH, VH, and WH constituting the high side arm of the FET bridge 72.
Further, stator windings 64a and 64a between the battery 71 and the FET bridge 72 and the booster circuit 74, and any two phases (for example, U phase and V phase) of the FET bridge 72 and the motor 31 are used. Is provided with a relay 75b that is opened and closed by a relay drive circuit 75a. A relay drive signal for controlling the opening / closing operation of the relay 75b is input from the control unit 73 to the relay drive circuit 75a.

  The control unit 73 performs current feedback control (vector control) on the γ-δ coordinates forming the rotation orthogonal coordinates. For example, the steering torque sensor 40 according to the steering torque input from the steering wheel 2 by the driver. The target γ-axis current Iγc and the target δ-axis current Iδc are calculated from the signal (torque detection signal Tq) output from the vehicle, the vehicle speed V output from the vehicle speed sensor 78, and the like, and based on the target γ-axis current Iγc and the target δ-axis current Iδc. The three-phase output voltages Vu, Vv, Vw are calculated, and a PWM signal as a gate signal is input to the FET bridge 72 according to the respective phase output voltages Vu, Vv, Vw. Γ-axis current Iγ and δ-axis current Iδ obtained by converting the detected values of the phase currents Iu, Iv, Iw supplied to 31 onto the γ-δ coordinates, and the target γ-axis current Control is performed so that each deviation between Iγc and target δ-axis current Iδc becomes zero. Details of feedback control of current on the γ-δ coordinates will be described later.

For example, when the motor 31 is started, the control unit 73 compares each phase output voltage Vu, Vv, Vw with a carrier signal such as a triangular wave in order to pass a sinusoidal current, and each transistor of the FET bridge 72. A gate signal (that is, a PWM signal) for driving on / off of UH, VH, WH, UL, VL, WL is generated. Then, the DC power supplied from the battery 71 is converted into the three-phase AC power by switching the on (off) / off (off) state of each pair of transistors in the FET bridge 72 for each of the three phases. By sequentially commutating energization to each stator winding 64a of the three-phase motor 31, alternating U phase current Iu, V phase current Iv and W phase current Iw are energized to each stator winding 64a.
Note that the duty ratio of the PWM signal for turning on / off the transistors UH, UL and VH, VL and WH, WL by pulse width modulation (PWM), that is, a map (data) of the on / off ratio is controlled in advance. Stored in the unit 73.

  The control unit 73 includes at least two of the phase currents Iu, Iv, and Iw supplied from the FET bridge 72 to each phase stator winding 64a of the motor 31 (for example, the U-phase current Iu and the W-phase current). Detection signal (for example, U-phase detection current Ius, W-phase detection current Iws, etc.) output from the current sensor 76 for detecting Iw, etc., and when the rotor 63 of the motor 31 used in, for example, coordinate conversion processing is stopped. Of each phase voltage Vu, Vv required to estimate the rotation angle θm (that is, the rotation angle of the magnetic pole of the rotor 63 from the predetermined reference rotation position and the rotation position of the output shaft 65 of the motor 31). , Vw (for example, U-phase voltage Vu, V-phase voltage Vv, etc.) and a midpoint voltage (midpoint voltage) Vn to which a plurality of stator windings 64a of motor 31 are connected are detected. Detection signals (for example, U-phase voltage Vu, V-phase voltage Vv, midpoint voltage Vn, etc.) output from voltage sensor 77, and detection signals output from vehicle speed sensor 78 that detects vehicle speed (vehicle speed) V Is entered.

  The control unit 73 includes, for example, a phase correction unit 81, a target current setting unit 82, a first correction calculation unit 83, an inertia correction unit 84, a differential calculation unit 85, a second correction calculation unit 86, and a damper. Correction unit 87, field control unit 88, current deviation calculation unit 89, current control unit 90, non-interference controller 91, voltage correction unit 92, γδ-3 phase conversion unit 93, and PWM signal generation unit 94 The first and second inter-phase voltage calculation units 95a and 95b, a rotation angle estimator 96, a switching unit 97, and a three-phase-γδ conversion unit 98.

  The phase correction unit 81 performs phase correction processing on the torque detection signal Tq output from the steering torque sensor 40 for each vehicle speed V output from the vehicle speed sensor 78.

  The target current setting unit 82 is supplied from the FET bridge 72 to the motor 31 based on the torque detection signal Tq subjected to the phase correction processing by the phase correction unit 81 and the vehicle speed V output from the vehicle speed sensor 78. A current command for designating each phase current Iu, Iv, Iw is calculated, and this current command is, among γ-axis target current Iγc and δ-axis target current Iδc on rotating orthogonal coordinates, in particular, δ This is the shaft target current Iδc.

  The γ-δ coordinates forming the rotation orthogonal coordinates are, for example, the field magnetic flux direction of the permanent magnet of the rotor 63 as the γ axis (field axis), and the direction orthogonal to the γ axis as the δ axis (torque axis). And rotating in synchronization with the rotational phase of the rotor 63. As a result, the γ-axis target current Iγc and the δ-axis target current Iδc, which are DC signals, are given as current commands for the AC signals supplied from the FET bridge 72 to each phase of the motor 31.

The first correction calculation unit 83 newly adds a value obtained by adding the inertia correction term output from the inertia correction unit 84 to the δ-axis target current Iδc calculated by the target current setting unit 82, and newly sets the δ-axis target current. Output as Iδc.
The inertia correction unit 84 is, for example, a time differential value of the torque detection signal Tq output from the steering torque sensor 40, the vehicle speed V output from the vehicle speed sensor 78, and the rotational speed ωm (= dθm / dt) output from the differential calculation unit 85. Based on (= dωm / dt), an inertia correction term related to the moment of inertia is calculated.
As the rotational speed ωm, an estimated rotational speed ωr output from a rotation time estimator 96b described later is employed.

The second correction calculation unit 86 newly calculates a value obtained by subtracting the damper correction term output from the damper correction unit 87 from the δ-axis target current Iδc corrected by the first correction calculation unit 83. Output as current Iδc.
The damper correction unit 87 performs damping based on, for example, the torque detection signal Tq output from the steering torque sensor 40, the vehicle speed V output from the vehicle speed sensor 78, and the rotational speed ωm (= dθm / dt) output from the differential calculation unit 85. A damper correction term related to the coefficient is calculated.

  The field control unit 88 is a field weakening of field weakening control that controls the current phase so as to weaken the field amount of the rotor 63 equivalently in order to suppress an increase in the counter electromotive voltage accompanying an increase in the rotational speed ωm of the motor 31, for example. The target value for the magnetic current is a γ-axis correction current, and the γ-axis target current Iγc calculated based on the δ-axis target current Iδc corrected by the second correction calculation unit 86 is further corrected based on the rotational speed ωm. The correction current is newly output as the γ-axis target current Iγc.

The current deviation calculation unit 89 calculates a deviation ΔIγ between the γ-axis target current Iγc and the γ-axis current Iγ, and calculates a deviation ΔIδ between the δ-axis target current Iδc and the δ-axis current Iδ. and a δ-axis current deviation calculation unit 89b.
The γ-axis current Iγ and the δ-axis current Iδ are three-phase-γδ that calculates the γ-axis current Iγ and the δ-axis current Iδ by converting the detected values of the phase currents Iu, Iv, and Iw onto γ-δ coordinates. Output from the converter 98.

  The current control unit 90 controls and amplifies the deviation ΔIγ to calculate the γ-axis voltage command value ΔVγ by, for example, PID (proportional integral differentiation) operation, and controls and amplifies the deviation ΔIδ to control the δ-axis. And a δ-axis current PI controller 90b for calculating the voltage command value ΔVδ.

  Further, the non-interference controller 91, for example, a γ-axis current Iγ and a δ-axis current Iδ, a γ-axis inductance Lγ stored in advance (a d-axis inductance Ld described later may be substituted), and a δ-axis inductance Lδ ( Based on the rotation speed ωm (equivalent to the estimated rotational speed ωr) output from the rotation time estimator 96b of the rotation angle estimator 96 to be described later). In order to cancel the velocity electromotive force component that interferes with the δ axis and control the γ axis and the δ axis independently, the γ axis compensation term Vγc (= ωr · Lq · Iδ) and δ-axis compensation term Vδc (= ωr · Lq · Iγ) are calculated.

  The voltage correction unit 92 includes a γ-axis voltage calculation unit 92a that sets a value obtained by adding the γ-axis compensation term Vγc to the γ-axis voltage command value ΔVγ, and a δ-axis voltage command value ΔVδ to δ. And a δ-axis voltage calculation unit 92b that uses a value obtained by adding the axis compensation term Vδc as a δ-axis voltage command value Vδ.

  The γδ-3 phase conversion unit 93 generates the γ-axis voltage command value Vγ and the δ-axis voltage command value Vδ on the γ-δ coordinates based on the rotation angle θm corresponding to the rotation position of the motor 31 output from the switching unit 97. Then, it is converted into a U-phase output voltage Vu, a V-phase output voltage Vv, and a W-phase output voltage Vw, which are voltage command values on a three-phase AC coordinate that is a stationary coordinate.

  For example, when the motor 31 is driven, the PWM signal generation unit 94 compares each phase output voltage Vu, Vv, Vw with a carrier signal such as a triangular wave in order to pass a sine wave current, and the FET bridge 72 A gate signal (that is, a PWM signal) for driving each transistor UH, VH, WH, UL, VL, WL on / off is generated.

  Further, the PWM signal generation unit 94, for example, at the time of estimating the rotation angle when the motor 31 is stopped, the FET bridge 72 according to a command signal Vsa output from a stop time estimator 96a of the rotation angle estimator 96 described later. Each of the transistors UH, VH, WH, UL, VL, WL is output a predetermined gate signal composed of pulses for driving on / off. This predetermined gate signal is generated between the phase terminals of the motor 31 (for example, between the U-phase and V-phase terminals, etc.) with a predetermined rectangular wave, for example, an audible frequency, with an energization pattern as shown in FIGS. As an external frequency, a rectangular wave AC voltage or a pulse having a predetermined voltage value (for example, 12 V) having a frequency (for example, 40 kHz) twice as high as a PWM frequency (for example, 20 kHz) at the time of driving the motor 31 is used. The FET bridge 72 is instructed to apply a rectangular wave AC voltage (for example, 12 V or the like) having a shape (for example, about 10 μsec).

The PWM signal generation unit 94 will also be described later when selecting a single estimated value from a plurality of candidates for the rotation angle θm, as will be described later, for example, when the rotation angle is estimated when the motor 31 is stopped. When the command signal Vsb output from the stop time estimator 96a of the rotation angle estimator 96 is input, an FET bridge is generated according to each phase output voltage Vu, Vv, Vw generated according to the steering torque. 72 outputs a predetermined gate signal composed of pulses for turning on / off the transistors UH, VH, WH, UL, VL, WL.
More specifically, the predetermined gate signal uses a single estimated value θm as a temporary estimated value from a plurality of candidates for a rotation angle θm, which will be described later, and uses this temporary estimated value to generate a predetermined minute current within the assist dead zone. The FET bridge 72 is instructed to energize and drive the motor 31, and whether or not the temporary estimated value is correct is determined by whether or not the steering assist force by the motor 31 acts in the same direction as the steering input by the driver. To do. If the provisional estimated value is correct, the motor is driven and controlled using this estimated value from the time when the steering torque exceeds the range within the assist dead zone. If it is not correct, the motor is driven and assisted using the other estimated value. Details will be described later.

  Further, the PWM signal generation unit 94 is a signal for instructing the boosting operation of the booster circuit 74 (for example, a gate signal for switching the on (conductive) / off (shutoff) state of each transistor of the charge pump circuit included in the booster circuit 74. Etc.) is output.

  The first and second inter-phase voltage calculation units 95a and 95b include operation amplifiers, and the first inter-phase voltage calculation unit 95a uses the U-phase voltage based on the phase voltages Vu and Vv and the midpoint voltage Vn detected by the voltage sensors 77. Vun (= Vu−Vn) is calculated, and the second interphase voltage calculation unit 95b calculates the V phase voltage Vvn (= Vv−Vn).

The rotation angle estimator 96 includes a stop time estimator 96a and a rotation time estimator 96b.
Then, the switching unit 97 selects the stop time estimator 96a or the rotation time estimator 96b according to the state of the motor 31, and the stop time rotation angle θs or the rotation time estimator 96b output from the stop time estimator 96a. Is output as the rotation angle θm.
For example, the switching unit 97 selects the stop time estimator 96a when the motor 31 is stopped, and selects the rotation time estimator 96b when the motor 31 rotates.

The switching unit 97 outputs the output from the stop time estimator 96a and the output from the rotation time estimator 96b based on a switching signal output from a stop determination unit 163 in FIG. 15 described later in the rotation time estimator 96b. Switch. Specifically, the magnitude of the electromotive voltage shown in FIG. 15 (the sum of the square of the δ-axis induced voltage Eex · cos θe and the square of the γ-axis induced voltage Eex · sin θe (Eex ² · cos ² θe + Eex ² · sin ² θe)) Is smaller than a predetermined value, it is determined that the motor 31 is stopped and a signal for selecting the stop time estimator 96a is generated, and the magnitude of the electromotive voltage (the square of the δ-axis induced voltage Eex · cos θe) is generated. And the sum of the squares of the γ-axis induced voltage Eex · sin θe (Eex ² · cos ² θe + Eex ² · sin ² θe)) are larger than a predetermined value, it is determined that the motor 31 is rotating and is estimated at the time of rotation. A signal for selecting the device 96b is generated.

The stop-time estimator 96a of the rotation angle estimator 96 estimates the stop-time rotation angle θs by using the fact that the inductance of the motor 31 varies with the rotation angle θm when the motor 31 is stopped.
Further, the rotation time estimator 96b of the rotation angle estimator 96 estimates the rotation-time estimated rotation angle θr by utilizing the fact that the induced voltage generated by the motor 31 varies with the rotation speed ωm when the motor 31 rotates.

  Specifically, the stop-time estimator 96a of the rotation angle estimator 96 includes a plurality of candidates for the stop-time rotation angle θs based on the interphase voltages Vun and Vvn output from the first and second interphase voltage calculators 95a and 95b. Is selected. A single estimated value from the plurality of candidates is used as a temporary estimated value, and the temporary bridge is used to instruct the FET bridge 72 to drive the motor 31 by energizing a predetermined minute current within the assist dead zone. From the torque detection signal Tq output from the steering torque sensor 40, whether or not the temporary estimated value is correct is determined depending on whether or not the steering assist force by the motor 31 acts in the same direction as the steering input by the driver. When the temporary estimated value is correct, this value is output as an estimated value of the rotation angle θs at the time of stop, and the motor 31 is driven and controlled using this estimated value from the time when the steering torque exceeds the range in the assist dead zone. Assist. If it is not correct, the other estimated value is output, and the motor 31 is driven and controlled using this estimated value from the time when the steering torque exceeds the range within the assist dead zone.

Specifically, the square of the δ-axis induced voltage Eex · cos θe output from a δ-axis induced voltage estimation unit 151 in FIG. 15 described later in the rotation time estimator 96b and the γ-axis output from the γ-axis induced voltage estimation unit 150 Based on the magnitude of the electromotive voltage by the sum of the square of the induced voltage Eex · sin θe (Eex ² · cos ² θe + Eex ² · sin ² θe), the value of the induced voltage caused by the rotation of the motor 31 is greater than a predetermined value. When the time is smaller, the stop determination unit 163 in FIG. 15 estimates that the motor 31 is stopped and outputs a switching signal.
Then, the switching unit 97 selects the rotation angle estimation by the stop time estimator 96a.

For example, when estimating the rotation angle when the motor 31 is stopped, the transistors UH, VH, WH, UL, VL, WL of the FET bridge 72 are connected to the high-side U-phase transistor UH as shown in FIG. And a state in which the low-side V-phase transistor VL is turned on and the other transistors VH, WH, UL, WL are turned off; for example, as shown in FIG. 6B, the high-side U-phase transistor UH is turned on; By repeating and driving two states of turning off the other transistors VH, WH, UL, VL, WL, a predetermined rectangular wave (for example, 40 kHz and between the U phase and V phase terminals of the motor 31 is driven. A command signal Vsa that instructs application of an AC voltage of 12 V) is output.
Alternatively, after the energization pattern shown in FIG. 6 (a) is performed once, the energization pattern shown in FIG. 6 (b) is performed once so that a pulse-like (for example, about 10 μsec) rectangular wave is generated. A command signal Vsa instructing application of an alternating voltage (for example, 12 V) is output.

And based on the ratio (interphase voltage ratio) Vun / Vvn between the U-phase voltage Vun and the V-phase voltage Vvn when a predetermined rectangular wave is applied between the U-phase and V-phase terminals of the motor 31, for example, is set in advance. The rotation angle θs at the time of stop is acquired by map search with respect to the predetermined first map.
The first map is a map showing a predetermined correspondence between the phase voltage ratio Vun / Vvn and the rotation angle θs at the time of stop, for example, as shown in FIG. 7, for example, in electrical angle (edeg). Within the range of 0 ° to 360 °, the four values θ1,..., Θ4 of the stop rotation angle θs correspond to an appropriate single value of the interphase voltage ratio Vun / Vvn. That is, two cycles of the interphase voltage ratio Vun / Vvn are 360 ° in electrical angle (edeg). For example, when the phase voltage ratio Vun / Vvn = 1.5, the rotation angle θs = θ1 (= 100 °), θ2 (= 150 °), θ3 (= 280 °), θ4 (= 330 °) corresponds to ing.

Then, the stop time estimator 96a searches for a map with respect to a predetermined second map set in advance, for example, based on the V-phase voltage Vvn when a predetermined rectangular wave is applied between the U-phase and V-phase terminals of the motor 31. Thus, any two of the four values θ1,..., Θ4 of the stop rotation angle θs searched by the first map are selected.
This second map is a map showing a predetermined correspondence between, for example, the V-phase voltage Vvn and the stop rotation angle θs. For example, as shown in FIG. 8, the second map is 0 ° to 360 ° in electrical angle (edeg). Within the range of °, the four values φ1,..., Φ4 of the stop rotation angle θs correspond to an appropriate single value of the V-phase voltage Vvn. That is, two cycles of the interphase voltage Vvn are 360 ° in electrical angle (edeg).

In order to obtain an accurate stop rotation angle θs even if the battery voltage (that is, the power supply voltage of the FET bridge 72) fluctuates, the battery voltage Vb is detected and the V-phase voltage Vvn detected using this value is detected. , And the second map is searched using the corrected V-phase voltage Vvn to obtain four values φ1,..., Φ4 of the rotation angle θs at the time of stop. Therefore, a voltage follower circuit 71a having an operational amplifier is provided between the stop time estimator 96a and the battery 71 (+ B), for example, as shown in FIG. 4, and the output of the voltage follower circuit 71a is stopped. It is input to the estimator 96a.
For example, if the V-phase voltage Vvn when the inter-phase voltage ratio Vun / Vvn = 1.5 is Vvn = 2.3 (V), the rotation angle that satisfies this is the rotation angle θs = φ1 (= 100 °), φ2 (= 175 °), φ3 (= 280 °), and φ4 (= 355 °).
For this reason, when the four values θ1 (= 100 °), θ2 (= 150 °), θ3 (= 280 °), and θ4 (= 330 °) of the rotation angle θs at the time of stop are retrieved by the first map, Two values θ1 (= 100 °) and θ3 (= 280 °) that are equal to the search result of the second map are selected as estimated value candidates for the stop rotation angle θs.

In the energization switching by the FET bridge 72, for example, when the U-phase and V-phase stator windings 64a are energized as shown in FIGS. Since the magnitudes of the currents flowing through the stator windings 64a are equal, the interphase voltage ratio Vun / Vvn is equal to the impedance ratio Zun / Zvn as shown in the following formula (1). In each impedance Zun (= Run + j · ω · Lun) and Zvn (= Rvn + j · ω · Lvn), the angular frequency ω (ω = 2πf, f = 40 kHz) is high, and the angular frequency ω is sufficiently high. Since the winding resistances Run and Rvn are sufficiently small compared to the reactances (ω · Lun) and (ω · Lvn), the interphase voltage ratio Vun / Vvn is substantially equal to the interphase inductance ratio Lun / Lvn.
Alternatively, since the pulse width Δt is short (10 μsec), the rate of change of current (ΔI / Δt) is large, and the voltage drop at each winding resistance Run, Rvn is sufficiently smaller than the voltage drop at each inductance Lun, Lvn. The interphase voltage ratio Vun / Vvn is substantially equal to the interphase inductance ratio Lun / Lvn as shown in the following formula (2).

The interphase inductances Lun, Lvn, and Lwn have a phase difference of 120 ° in electrical angle (edeg) due to the saliency of the motor 31, for example, as shown in FIG. Accordingly, two cycles of this change are 360 ° in electrical angle (edeg).
In FIG. 9 showing the inductance change of the motor 31, for example, the average value of the interphase inductances Lun, Lvn, and Lwn is about 72 μH, and the interphase inductances Lun, Lvn, and Lwn are the minimum value (for example, 58 μH) and the maximum value. (For example, 86 μH).
Therefore, the rotation angle θs at the time of stopping can be detected from the phase voltage ratio Vun / Vvn approximated to the phase inductance ratio Lun / Lvn.

For example, with respect to the winding resistance Run (for example, 10 mΩ) of the motor 31 and the angular frequency ω (for example, 2π × 40 × 10 ³ rad / sec), the winding resistance Run (= 10 × 10 ⁻³ Ω). << impedance ω · Lun (= 18100 × 10 ⁻³ Ω), and the winding resistance Run can be ignored as shown in the above formula (1).
Further, since the impedances Zun and Zvn are relatively high with respect to the voltage of the battery 71, the magnitude of the current flowing through the U-phase and V-phase stator windings 64a (for example, about 0.1 A) is relative. It is possible to prevent unnecessary torque from being generated in the motor 31 due to energization of a rectangular wave applied between the phase terminals of the motor 31 when the rotation angle is estimated.

Further, the stop-time estimator 96a uses the steering torque Tq output from the steering torque sensor 40 to calculate the stop-time rotation angle θs selected based on, for example, the first map shown in FIG. 7 and the second map shown in FIG. One of the two values (for example, θ1, θ3) is selected as the temporary estimated value.
The two estimated value candidates (for example, θ1, θ3) of the selected rotation angle θs at the time of stop correspond to each value (for example, θ1, θ3) because the electrical angle (edeg) has a phase difference of 180 °. The field direction of the rotor 63, that is, the direction of the magnetic poles, is opposite to each other.
Therefore, when the same energization is performed on the motor 31 in each of the two estimated value candidates (for example, θ1 and θ3), on the other hand, the steering torque of the driver is assisted to be the same direction as the driver's steering direction. This is because an auxiliary torque is generated by the motor 31 and, on the other hand, the steering torque of the driver is increased, and the auxiliary torque is generated by the motor 31 in the direction opposite to the steering direction of the driver. Thereby, it is possible to determine whether or not the estimated value candidate is appropriate by observing the steering torque.

  For example, as shown in the time chart of FIG. 10 or FIG. 11, in a period in which the steering torque (torque detection signal) Tq detected by the steering torque sensor 40 is zero (that is, a period before time t1), the motor 31 In the stop state, the stop-time estimator 96a acquires two estimated value candidates (for example, θ1, θ3) as the stop-time rotation angle θs based on the first map and the second map. Here, the stop-time estimator 96a selects one of the two estimated value candidates (for example, θ1 and θ3) (for example, θ1) of the stop-time rotation angle θs, that is, the temporary stop-time rotation angle θs. Select as estimated value (provisional estimated value). In the example shown in FIG. 10 or FIG. 11, the motor 31 is in a stopped state during a period in which the detected steering torque (torque detection signal) Tq is zero. The same applies when 31 is stopped.

Then, for example, after time t1 shown in FIG. 10 or FIG. 11, the steering torque (torque detection signal) Tq detected by the steering torque sensor 40 in response to the driver's steering input starts to change from zero. Then, the stop-time estimator 96a causes the steering assist force by the motor 31 to act in the same direction as the steering input by the driver according to the temporary estimated value of the stop-time rotation angle θs, and the motor via the FET bridge 72. A command signal Vsb instructing to energize 31 a predetermined minute current temporarily (period of time t2 to time t3) is output to the PWM signal generation unit 94.
Note that the energization of the predetermined minute current is a state where the steering torque (torque detection signal) Tq detected by the steering torque sensor 40 is equal to or less than a predetermined auxiliary lower limit torque (assist dead band) (for example, as shown in FIG. 10 or FIG. 11). (Period from time t1 to time t4).

  The stop time estimator 96a is caused by, for example, a predetermined minute current (motor current) being applied to the motor 31 during a period from time t2 to time t3 shown in FIG. 10 or FIG. As shown in FIG. 10, when the increase speed of the steering torque (torque detection signal) Tq detected by the steering torque sensor 40 decreases or the steering torque (torque detection signal) Tq decreases, the steering direction of the driver is changed. Auxiliary torque is generated by the motor 31 in the same direction, and it is determined that the temporary estimated value (for example, θ1) of the rotation angle θs at the time of stop is appropriate, and this temporary estimated value (for example, θ1) is It is set as an estimated value of the rotation angle θs at the time of stopping when the motor 31 is stopped before energization with a predetermined minute current.

  On the other hand, for example, as shown in FIG. 11, the increasing speed of the steering torque (torque detection signal) Tq detected by the steering torque sensor 40 due to the application of a predetermined minute current (motor current) to the motor 31. If it increases, an auxiliary torque is generated by the motor 31 in a direction different from the steering direction of the driver (that is, opposite), and the provisional estimated value (for example, θ1) of the rotation angle θs at the time of stop is not appropriate. In other words, the other of the two values (for example, θ3) of the stop rotation angle θs other than the temporary estimated value (for example, θ1), that is, the two values (for example, θ1, θ3) of the stop rotation angle θs. It is determined that it is appropriate to set the estimated value, and, for example, after the time t3 shown in FIG. 11, “1” is set to the flag value of the driving direction inversion flag that instructs to reverse the driving direction of the motor 31. Set Further, one of the two values (for example, θ3, θ3) of the rotation angle θs at the time of stop is used to estimate the rotation angle θs at the time of stop when the motor 31 is stopped before energizing a predetermined minute current. Set as a value.

  Further, when the steering torque (torque detection signal) Tq exceeds the auxiliary lower limit torque, for example, after time t4 shown in FIG. 10 or FIG. 11, the motor 31 is started using the set stop rotation angle θs. To do.

  In a driving state in which the rotational speed ωm of the motor 31 is equal to or higher than a predetermined speed, the U-phase voltage Vun or the V-phase voltage Vvn increases from a predetermined value during the stop due to an induced voltage caused by the rotation of the motor 31. An error occurs in the method of detecting the inductance that changes depending on the rotation angle θm using the phase-to-phase voltage ratio when an AC voltage is applied between the lines of the motor 31, making estimation difficult. Therefore, in this case, the rotation time estimator 96b of the rotation angle estimator 96 estimates that the motor 31 is rotating, and the switching unit 97 selects rotation angle estimation by the stop time estimator 96a. Then, the rotation time estimator 96b estimates the rotation time estimated rotation angle θr based on the induced voltage that varies according to the magnetic pole position of the rotor 63.

For example, in the vector control block 200 using a conventional dq axis using the rotation sensor (resolver) 201 shown in FIG. Further, a γ-δ axis having a phase difference θe (= actual rotation angle θ−rotation estimated rotation angle θr) and rotation speed ωe as shown in the following mathematical formula (3) is set.
Then, the induced voltage generated on the γ-axis and the δ-axis having a phase angle of the phase difference θe with respect to the d-axis and the q-axis is estimated as shown in FIG. 15, and the induced voltage generated on the estimated γ-axis and δ-axis is estimated. Further, the phase difference θe is obtained as shown in FIG. 15, and the actual rotation angle (actual rotation angle) θ and the estimated rotation angle θr during rotation are equal as shown in FIG. 14 so that the phase difference θe converges to zero. Control is performed as follows.

  Further, the current (d-axis current Id and q-axis current Iq) and voltage (d-axis voltage command value Vd and q-axis voltage command value Vq) on the dq axis, and the current on the γ-δ axis (γ-axis current) Iγ and δ-axis current Iδ) and voltage (γ-axis voltage Vγ and δ-axis voltage Vδ) are described as shown in the following equations (4) and (5).

FIG. 14 shows a control block for estimating a rotation angle (estimated rotation angle) θr during rotation of the motor 31 and vector-controlling the motor 31 using the estimated rotation angle θr during rotation. This is a detail of the vector control unit in the control unit 73 shown in FIG.
The rotation time estimator 96b includes a γ-axis induced voltage estimator 150, a δ-axis induced voltage estimator 151, a tan θe calculator 152, a θe calculator 153, a deviation calculator 154, a PI controller 155, a differential It comprises a calculator 156, an Ld multiplier 157, a rotation speed calculator 158, an integrator 159, multipliers 160 and 161, an adder 162, a stop determiner 163, and a square sum calculator 164.

The γ-axis induced voltage estimator 150 and the δ-axis induced voltage estimation unit 151 include a γ-axis current Iγ and a δ-axis current Iδ, a γ-axis voltage Vγ and a δ-axis voltage Vδ, a value (−ωe · Ld · Iγ), and a value. From (ωe · Ld · Iδ), the γ-axis induced voltage Eex · sinθe that is the induced voltage appearing on the γ-axis and the δ-axis induced voltage Eex · cosθe that is the induced voltage appearing on the δ-axis are obtained as shown in the block diagram of FIG. Calculate and output.
The tan θe calculator 152 calculates and outputs a ratio tan θe that is a ratio of the obtained γ-axis induced voltage Eex · sin θe and δ-axis induced voltage Eex · cos θe.
The θe calculator 153 calculates and outputs the phase difference θe by obtaining an arctangent value tan ⁻¹ of the value of the ratio tan θe.

The deviation calculation unit 154 calculates the deviation between the phase difference θe calculated in this way and the phase difference θe = 0, which is a convergence target value of the phase difference θe, and the PI control unit 155 controls the deviation to be zero. PI control is performed and a control amount Δθ is output.
Further, the differentiator 156 differentiates the phase difference θe calculated in this way to calculate and output the rotational speed ωe.
Further, the Ld multiplier 157 multiplies the rotational speed ωe and the d-axis inductance Ld and outputs the result.

On the other hand, the square sum calculator 164 calculates the square of the δ-axis induced voltage Eex · cos θe output from the δ-axis induced voltage estimation unit 151 and the square of the γ-axis induced voltage Eex · sin θe output from the γ-axis induced voltage estimation unit 150. The sum (Eex ² · cos ² θe + Eex ² · sin ² θe) is calculated and output.
The number-of-rotations calculator 158 is a square root (√ (Eex ² · cos ² θe + Eex ² · sin ² θe) = Eex) of the value (Eex ² · cos ² θe + Eex ² · sin ² θe) output from the square sum calculator 164. That is, by dividing the value of the induced voltage by the previously stored induced voltage constant ke, the estimated rotational speed ωr is calculated and output.
Further, the integrator 159 integrates the estimated rotational speed ωr based on the initial rotational angle θ0 output from the stop time estimator 96a, and calculates and outputs the rotational angle (actual rotational angle) θ.
The multiplier 160 outputs a value (ωe · Ld · Iδ) obtained by multiplying the multiplication value (ωe · Ld) of the rotational speed ωe and the d-axis inductance Ld and the δ-axis current Iδ.

The multiplier 161 outputs a value (−ωe · Ld · Iγ) obtained by multiplying the multiplication value (ωe · Ld) of the rotational speed ωe and the d-axis inductance Ld and the γ-axis current Iγ.
The adder 162 adds the control amount Δθ and the rotation angle θ, and outputs this as the rotation estimated rotation angle θr.
Thereafter, the vector control of the block diagram shown in FIG. 14 is performed using the estimated rotation angle θr and the estimated rotation speed ωr so that the phase difference θe converges to zero.

  14 and 15 show an example in which the control amount Δθ is added to the rotation angle θ, and this is output as the estimated rotation angle θr at the time of rotation to perform vector control to converge the phase difference θe to zero. However, the present invention is not limited to this. For example, as shown in FIG. 25, the phase difference θe is reduced to zero by changing the characteristics of the γ-axis induced voltage estimator 150 and the δ-axis induced voltage estimator 151 using the control amount Δθ. It may be converged.

  For example, as shown in FIG. 26, one or both of the resistance value R and the d-axis inductance Ld in the γ-axis induced voltage estimator 150 and the δ-axis induced voltage estimator 151 is changed by the control amount Δθ. Specifically, a map of the resistance value R and the d-axis inductance Ld with respect to the control amount Δθ is prepared in advance, and the resistance value R and the d-axis inductance Ld are changed by searching this map.

14 and 15, and the values (Eex ² · cos ² θe + Eex ² · sin ² θe) in FIGS. 25 and 26, that is, the square of the δ-axis induced voltage Eex · cos θe and the γ-axis induced voltage Eex · sin θe. Since the sum of the square of (sin ² θe + cos ² θe = 1), for example, as shown in FIG. 16, the square value of the induced voltage Eex of the motor 31 is shown regardless of the phase angle θe.
For example, as shown in FIG. 17A, the stop determination unit 163 compares the value (Eex ² · cos ² θe + Eex ² · sin ² θe) with a predetermined value C1 stored in advance, and compares the value (Eex ² · When cos ² θe + Eex ² · sin ² θe) is less than the predetermined value C1, it is determined that the motor 31 is stopped.
Further, as shown in FIG. 17B, for example, the stop determination unit 163 compares the square root of the value (Eex ² · cos ² θe + Eex ² · sin ² θe) with a predetermined value C2 that is stored in advance. When the square root of (Eex ² · cos ² θe + Eex ² · sin ² θe) is less than the predetermined value C2, it is determined that the motor 31 is stopped.
When the stop determination unit 163 determines that the motor 31 is stopped, the stop determination unit 163 outputs a switching signal Vc to the switching unit 97.

The electric steering device 1 according to this embodiment has the above-described configuration. Next, operations of the electric steering device 1, particularly operations of the rotation angle estimator 96 and the switching unit 97 will be described.
First, for example, in step S01 shown in FIG. 18, it is determined whether or not the ignition switch of the vehicle is turned on (IG ON).
If this determination is “YES”, the flow proceeds to step S 02.
On the other hand, when the determination result is “NO”, the process does not proceed.

In step S02, each detected current (that is, γ-axis current Iγ and δ-axis current Iδ) and each voltage command value (that is, γ-axis voltage Vγ and δ-axis voltage Vδ) are acquired.
In step S03, the γ-axis induced voltage Eex · sin θe and the δ-axis induced voltage Eex · cos θe are calculated from the acquired γ-axis current Iγ, δ-axis current Iδ, γ-axis voltage Vγ, and δ-axis voltage Vδ.
In step S04, the sum of the square of the δ-axis induced voltage Eex · cos θe and the square of the γ-axis induced voltage Eex · sin θe (Eex ² · cos ² θe + Eex ² · sin ² θe) is calculated, and this value (Eex ² · cos ² θe + Eex ² · sin ² θe), it is determined whether or not the motor 31 is in a stopped state.
For example, as shown in FIGS. 17A and 17B, when the value (Eex ² · cos ² θe + Eex ² · sin ² θe) is smaller than a predetermined value C1 (= (C2) ² ), the motor 31 It is determined that no induced voltage is generated due to rotation and the motor 31 is stopped.
If this determination is “NO”, the flow proceeds to step S 24 described later.
On the other hand, if this determination is “YES”, the flow proceeds to step S 05.
In step S05, application of a predetermined rectangular wave AC voltage is started between the phase terminals of the motor 31 (for example, between the U-phase and V-phase terminals, etc.), and the process of estimating the rotation angle θs at the stop is started.
In step S06, the U-phase voltage Vun (= Vu−Vn) and the V-phase voltage Vvn (= Vv−) calculated based on the phase voltages Vu, Vv and the midpoint voltage Vn detected by the voltage sensors 77. Vn) is obtained.

In step S07, an interphase voltage ratio Vun / Vvn is calculated.
In step S08, four values θ1,..., Θ4 of the stop rotation angle θs are acquired by map search for the first map based on the interphase voltage ratio Vun / Vvn.
In step S09, four values φ1,..., Φ4 of the rotation angle θs at the time of stop are acquired by map search for the second map based on the V-phase voltage Vvn.
In step S10, two values equivalent to any one of the four values φ1,..., Φ4 are selected from the four values θ1,.

In step S11, application of the predetermined rectangular wave AC voltage is terminated.
Then, in step S12 shown in FIG. 19, one of the two estimated value candidates of the stop rotation angle θs is selected as a temporary estimated value.
In step S13, it is determined whether the steering torque Tq detected by the steering torque sensor 40 is greater than zero.
When the determination result is “NO”, the determination process of step S13 is repeatedly executed.
On the other hand, if the determination result is “YES”, that is, if the driver's steering input is started, the process proceeds to step S14.

In step S14, it is determined whether or not the steering torque Tq is a predetermined set torque (<auxiliary lower limit torque) smaller than the auxiliary lower limit torque.
If this determination is “YES”, the flow proceeds to step S15.
On the other hand, when the determination result is “NO”, the determination process of step S14 is repeatedly executed.
In step S15, the steering assist force by the motor 31 acts in the same direction as the steering input by the driver in accordance with the steering torque Tq detected by the steering torque sensor 40 and the temporary estimated value of the rotation angle θs at the time of stop. In this manner, the motor 31 is driven and controlled with a predetermined minute current via the FET bridge 72.

In step S <b> 16, it is determined whether or not the increasing speed of the steering torque Tq detected by the steering torque sensor 40 has decreased due to the application of a predetermined minute current to the motor 31.
If this determination is “NO”, the flow proceeds to step S 18 described later.
On the other hand, if the determination result is “YES”, it is determined that the provisional estimated value of the stop rotation angle θs is appropriate, and the process proceeds to step S17. In step S17, the stop rotation angle θs is set. The temporary estimated value is set as an estimated value of the rotation angle θs at the time of stopping when the motor 31 is stopped before the predetermined minute current is applied, and the process proceeds to step S20 described later.

In step S18, it is determined whether or not the increasing speed of the steering torque Tq detected by the steering torque sensor 40 has increased due to the application of a predetermined minute current to the motor 31.
If this determination is “NO”, the flow returns to step S 16 described above.
On the other hand, if the determination result is “YES”, it is determined that the temporary estimated value of the rotation angle θs at the time of stop is not appropriate, and the process proceeds to step S19. In this step S19, two estimated value candidate candidates are determined. Any one of them is set as an estimated value of the rotation angle θs at the time of stop in the stop state of the motor 31 before energization with a predetermined minute current. This corresponds to the aforementioned driving direction inversion flag.
In step S20, the execution of the drive control of the motor 13 with a predetermined minute current is finished, the estimated value of the stop rotation angle θs is set as the rotation angle θm, and the stop rotation angle θs estimation process is finished.

In step S21 shown in FIG. 20, each detected current (that is, γ-axis current Iγ and δ-axis current Iδ) and each voltage command value (that is, γ-axis voltage Vγ and δ-axis voltage Vδ) are acquired again. .
In step S22, the acquired γ-axis current Iγ and δ-axis current Iδ and γ-axis voltage Vγ and δ-axis voltage Vδ, motor resistance R, d-axis inductance Ld, which will be described in advance, From the phase difference speed (rotational speed) ωe, the γ-axis induced voltage Eex · sin θe and the δ-axis induced voltage Eex · cos θe are calculated.
In step S23, the sum of the square of the δ-axis induced voltage Eex · cos θe and the square of the γ-axis induced voltage Eex · sin θe (Eex ² · cos ² θe + Eex ² · sin ² θe) is calculated, and this value (Eex ² · cos ² θe + Eex ² · sin ² θe), it is determined whether or not the motor 31 is in a rotating state.
If this determination is “NO”, the flow proceeds to the end, and the series of processing is ended.
On the other hand, if this determination is “YES”, the flow proceeds to step S24.

In step S24, a ratio (tan θe) between the obtained γ-axis induced voltage Eex · sin θe and δ-axis induced voltage Eex · cos θe is obtained, and an arctangent value tan ^{−1 of the} resulting ratio (tan θe) is obtained. Is calculated to obtain the phase difference θe, and the phase difference θe is differentiated to calculate the phase difference speed (rotational speed) ωe.
In step S25, the deviation of zero, which is the convergence target of the phase difference θe and the phase difference θe, is calculated, and the control amount Δθ is calculated via the PI controller 155.
In step S26, the estimated rotational speed ωr is calculated by dividing the δ-axis induced voltage Eex · cos θe in advance by a known induced voltage constant ke, and the estimated rotational speed ωr is integrated to obtain a rotation angle (actual rotation). Angle) θ is obtained.
In step S27, the rotation angle θ and the control amount Δθ are added to calculate the rotation estimated rotation angle θr. Then, brushless motor drive control is performed using the estimated rotation angle θr during rotation so that the phase difference θe becomes zero.

In step S28, it is determined whether or not the motor 31 has just been started.
If this determination is “NO”, the flow proceeds to step S 30 described later.
On the other hand, if this determination is “YES”, the flow proceeds to step S29.
In step S29, it is determined whether or not there is a difference exceeding a predetermined value between the rotation angle θs at the stop and the estimated rotation angle θr at the rotation, thereby determining the rotation angle θs at the stop and the estimated rotation at the rotation. It is determined whether or not the angle θr is substantially equal.
When the determination result is “YES”, the process proceeds to step S30, and in this step S30, drive control of the motor 31 based on the estimated rotation angle θr is executed as normal time control, and a series of processes is ended. To do.
On the other hand, when the determination result is “NO”, the process proceeds to step S31. In step S31, for example, a command for stopping the motor 31 that is rotationally driven is given as the predetermined abnormality time control, for example, during rotation. It is prohibited to output the estimated value of the estimated rotation angle θr as the rotation angle θm, and regardless of detection signals (for example, U-phase detection current Ius, W-phase detection current Iws, etc.) output from the current sensor 76. The execution of another known estimation process for estimating the rotation angle θm based on the detection signal output from the voltage sensor 77 is instructed, and the series of processes ends.
In the control loop in which the motor 31 is vector-controlled, the series of processing in steps S01 to S31 is repeated to estimate the rotation angle θm of the motor 31, and the motor 31 is controlled.

  As described above, according to the brushless motor control device 70 according to the present embodiment, when the phase difference θe does not converge to zero during the execution of the estimation calculation of the rotation angle θm (particularly, induction induced by the motor 31). Even when the voltage is small and the phase difference θe is difficult to converge to zero, the induced voltage generated by the motor 31 can be detected accurately regardless of the phase difference θe. If it is smaller, it can be properly determined that the motor 31 is stopped.

  Specifically, by observing the sum of the square of the induced voltage (Eex × sin θe) generated on the γ-axis and the square of the induced voltage (Eex × cos θe) generated on the δ-axis, regardless of the phase difference θe, The induced voltage can be detected with high accuracy, and the rotation and stop of the motor 31 can be appropriately determined. According to the determination result, an accurate estimated value of the rotation angle θm can be obtained, and smooth drive control can be performed without causing the motor 31 to step out or cause torque fluctuation. it can.

  In addition, according to the electric steering device 1 according to the present embodiment, it is possible to simplify the device configuration and reduce the size of the device without the need of providing a rotation angle sensor such as a resolver, for example, and improve the vehicle mountability. It is possible to prevent the motor 31 from stepping out, causing torque fluctuations and lowering the steering feeling, and preventing the motor 31 from stepping out and stopping, for example. In addition to preventing the driver from feeling uncomfortable with the steering feeling, the driving behavior of the vehicle can be prevented from becoming unstable.

  In the above-described embodiment, the rotation time estimator 96b is configured such that the γ-axis current Iγ, the δ-axis current Iδ, the γ-axis voltage command value Vγ, and the δ-axis voltage command value Vδ are known in advance. The phase difference θe is calculated based on the d-axis inductance Ld and the q-axis inductance Lq, and the convergence control is executed so that the phase difference θe becomes zero, so that the estimated rotation angle θr during rotation is estimated. However, the present invention is not limited to this. For example, U is calculated by a first interphase voltage calculation unit 95a including an operation amplifier, such as the brushless motor control device 70 of the electric steering device 1 according to the first modification shown in FIG. Based on -N-phase line voltage Vun (= Vu-Vn), U-phase detection current Ius output from current sensor 76, U-N-phase inductance Lun, and resistance Run that are known in advance. Then, the induced voltage Ve is calculated by the following formula (6), and the rotational speed ωr is estimated from the induced voltage Ve by utilizing the fact that the induced voltage Ve is proportional to the rotational speed of the motor 31, and this is integrated over time. The value obtained in this way may be the estimated value of the estimated rotation angle θr during rotation. Then, control on the dq axis is performed as shown in FIG. 21 using the estimated rotation angle θr during rotation.

  In the first modification, as a main point different from the above-described embodiment, the target current setting unit 82 outputs a q-axis target current Iqc. The current control unit 90 controls and amplifies the deviation ΔId to calculate the d-axis voltage command value ΔVd, and the current control unit 90 controls and amplifies the deviation ΔIq to calculate the q-axis voltage command value ΔVq. And a shaft current PI controller 90e. Then, the dq-3 phase conversion unit 93A, which replaces the γδ-3 phase conversion unit 93, converts the d-axis voltage command value Vd and the q-axis voltage command value Vq on the dq coordinate into three-phase AC coordinates that are stationary coordinates. The U-phase output voltage Vu, the V-phase output voltage Vv, and the W-phase output voltage Vw, which are the above voltage command values, are converted and output. Further, a three-phase-γδ conversion unit 98A replacing the three-phase-γδ conversion unit 98 converts the detected values of the respective phase currents Iu, Iv, Iw onto dq coordinates, and d-axis current Id and q-axis current Iq. Is calculated and output. The rotation time estimator 96c instead of the rotation time estimator 96b includes, for example, a U-phase voltage Vun (= Vu−Vn) output from the first interphase voltage calculation unit 95a and a U-phase detection output from the current sensor 76. Based on the current Ius, the estimated rotational speed ωr and the estimated rotational angle θr during rotation are calculated and output.

In the above-described embodiment, the stop-time estimator 96a estimates two estimated value candidates (for example, θ1, θ3) of the stop-time rotation angle θs based on the interphase voltage ratio Vun / Vvn. Without limitation, for example, two estimated value candidates (for example, θ1, θ3) of the rotation angle θs at the time of stop may be estimated based on the line voltage ratio Vuv / Vwu.
In the brushless motor control apparatus 70 according to the second modified example, as shown in FIG. 22, for example, the stop time estimator 96a has a detection output from a voltage sensor 77 for detecting each phase voltage Vu, Vv, Vw. A signal is being input.

The stop time estimator 96a calculates the line voltages Vuv (= Vu−Vv) and Vwu (= Vw−Vu), and the line voltage ratio Vuv / Vwu is substantially equal to the line inductance ratio Luv / Lwu. By utilizing this, the rotation angle θs at the time of stop is obtained by map search with respect to the third map showing a predetermined correspondence relationship between the line voltage ratio Vuv / Vwu and the rotation angle θs at the time of stop.
In addition, this 3rd map is a map which shows the predetermined | prescribed correspondence of line voltage ratio Vuv / Vwu, for example, and rotation angle (theta) s at a time of stop, Comprising: The range of 0 degree to 360 degrees in an electrical angle (edeg) The four values α1,..., Α4 of the rotation angle θs at the time of stop correspond to an appropriate single value of the line voltage ratio Vuv / Vwu. That is, two cycles of the line voltage ratio Vuv / Vwu are 360 ° in electrical angle (edeg).
Then, the stop-time estimator 96a selects the line voltage Vwu and the stop voltage in order to select any two of the four values α1,..., Α4 of the stop-time rotation angle θs searched by the third map. The stop rotation angle θs is obtained by map search for the fourth map showing a predetermined correspondence relationship with the hour rotation angle θs.
This fourth map is a map showing a predetermined correspondence between, for example, the line voltage Vwu and the rotation angle θs at the time of stop, and the line-to-line is within the range of 0 ° to 360 ° in electrical angle (edeg). Four values β1,..., Β4 of the rotation angle θs at the stop correspond to an appropriate single value of the voltage Vwu. That is, two cycles of the line voltages Vuv and Vwu are 360 ° in electrical angle (edeg).
Among the four values α1,..., Α4 of the stop rotation angle θs searched by the third map, any one of the four values β1,..., Β4 of the stop rotation angle θs corresponding to the line voltage Vwu. Two values equivalent to two are selected as estimated value candidates of the rotation angle θs at the time of stop.
In order to obtain an accurate stop rotation angle θs even if the battery voltage (that is, the power supply voltage of the FET bridge 72) fluctuates, the battery voltage Vb is detected, and the line voltage Vwu detected using this value is detected. The fourth map is searched using the corrected line voltage Vwu, and four values β1,..., Β4 of the rotation angle θs at the time of stop are obtained. For this reason, a voltage follower circuit 71a having an operational amplifier is provided between the stop time estimator 96a and the battery 71 (+ B), for example, as shown in FIG. 22, and the output of the voltage follower circuit 71a is stopped. It is input to the estimator 96a.

  Note that the line voltages Vuv (= Vu−Vv), Vvw (= Vv−Vw), and Vwu (= Vw−Vu) according to the second modification are as described above, for example, as shown in the following formula (7). The magnitude is multiplied by √3 with respect to each interphase voltage Vun, Vvn, Vwn in the embodiment, and the phase is delayed by (π / 6).

  In the above-described embodiment, the stop-time estimator 96a determines that the provisional estimated value (for example, θ1) of the stop-time rotation angle θs is not appropriate when a predetermined minute current is applied to the motor 31. When the flag value of the driving direction reversal flag is set to “1”, one of the two estimated value candidates (for example, θ1, θ3) of the rotation angle θs at the time of stop selected based on the second map (for example, θ1, θ3) For example, θ3) is set as an estimated value of the rotation angle θs at the time of stopping the motor 31 in a stopped state before energization with a predetermined minute current. However, the present invention is not limited to this. Even if it is determined that the setting of (for example, θ1) is not appropriate, this temporary estimated value is set as an estimated value of the rotation angle θs at the time of stopping in a stopped state of the motor 31 before energization with a predetermined minute current. As shown in FIGS. 10 and 11 When the flag value of the driving direction reversal flag is set to “1” and the steering torque Tq detected by the steering torque sensor 40 becomes equal to or higher than a predetermined auxiliary lower limit torque, the auxiliary torque is supplied from the motor 31 according to the steering torque Tq. When generating the auxiliary torque, only the direction of generation of the auxiliary torque may be reversed without changing the magnitude of the auxiliary torque.

It is a lineblock diagram of the electric steering device concerning one embodiment of the present invention. It is a lineblock diagram of the steering auxiliary mechanism of the electric steering device concerning one embodiment of the present invention. It is the sectional view on the AA line shown in FIG. It is a block diagram of the control apparatus of the brushless motor which concerns on one Embodiment of this invention. It is a block diagram of FET bridge | bridging shown in FIG. FIG. 5 is a diagram illustrating an on (conduction) / off (cutoff) state of each transistor of the FET bridge illustrated in FIG. 4. It is a graph which shows the correspondence of the voltage ratio Vun / Vvn and rotation angle (theta) m which concern on one Embodiment of this invention. It is a graph which shows the correspondence of the interphase voltage Vvn and rotation angle (theta) m which concern on one Embodiment of this invention. It is a graph which shows the correspondence of each phase inductance Lun, Lvn, Lwn and rotation angle (theta) m which concerns on one Embodiment of this invention. It is a graph which shows an example of the change of the steering torque Tq which concerns on one Embodiment of this invention, a motor current, and the flag value of a drive direction inversion flag. It is a graph which shows an example of the change of the steering torque Tq which concerns on one Embodiment of this invention, a motor current, and the flag value of a drive direction inversion flag. It is a figure which shows the structure of the motor control block which concerns on an example of a prior art as a comparative example with respect to embodiment of this invention. It is a figure which shows an example of the correspondence of the dq axis | shaft and (gamma) -delta axis which concern on one Embodiment of this invention. It is a figure which shows the structure of the motor control block which concerns on one Embodiment of this invention. It is a block block diagram of the rotation time estimator which concerns on one Embodiment of this invention. It is a figure which shows an example of (delta) axis induced voltage Eex * cos (theta) e and (gamma) axis induced voltage Eex * sin (theta) e which concern on one Embodiment of this invention. The motor rotation speed (rotation speed), the value (Eex ² · cos ² θe + Eex ² · sin ² θe) and the square root (√ (Eex ² · cos ² θe + Eex) output from the sum of squares calculator according to one embodiment of the present invention is a graph showing one example of a correspondence relationship between ^{2 · sin 2 θe) = Eex} ). It is a flowchart which shows operation | movement of the electric steering apparatus which concerns on one Embodiment of this invention, especially operation | movement of a rotation angle estimator and a switching part. It is a flowchart which shows operation | movement of the electric steering apparatus which concerns on one Embodiment of this invention, especially operation | movement of a rotation angle estimator and a switching part. It is a flowchart which shows operation | movement of the electric steering apparatus which concerns on one Embodiment of this invention, especially operation | movement of a rotation angle estimator and a switching part. It is a block diagram of the control apparatus of the brushless motor which concerns on the 1st modification of one Embodiment of this invention. It is a block diagram of the control apparatus of the brushless motor which concerns on the 2nd modification of one Embodiment of this invention. It is a graph which shows an example of the convergence state of phase difference (theta) e and (delta) axis induced voltage Eex * cos (theta) e which concerns on embodiment of this invention. It is a graph which shows an example of the correspondence of motor rotation speed (rotation speed) and phase difference (theta) e which concerns on embodiment of this invention. It is a figure which shows the structure of the motor control block which concerns on the modification of one Embodiment of this invention. It is a block block diagram of the rotation time estimator which concerns on the modification of one Embodiment of this invention.

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 Electric steering apparatus 31 Motor 40 Steering torque sensor (steering torque detection means)
63 Rotor 64 Stator 64a Stator winding 70 Brushless motor control device 72 FET bridge (energization switching means)
73 Control unit (current control means, steering control means, drive control means)
76 Current sensor 77 Voltage sensor 94 PWM signal generator (energization switching means)
96a Stop time estimator (stop rotation angle estimation means, drive control means)
96b Rotation estimator (rotation angle estimation means)
163 Stop determiner (determination means)

Claims (5)

A brushless motor control device comprising: a brushless motor including a rotor and a stator having a plurality of phases of stator windings; and energization switching means for switching and controlling energization of the stator windings of each phase,
an induced voltage estimating means for setting a γδ coordinate system having a phase difference with respect to the dq coordinate system and estimating an induced voltage component generated on the γ axis and an induced voltage component generated on the δ axis;
Based on whether the sum of the squares of the induced voltage component generated in the square and the δ-axis induced voltage component generated in the γ axis are estimated by the induced voltage estimation unit is larger than a predetermined value, the brushless control device for a brushless motor, characterized in that it comprises a a <br/> determining means for determining the rotation and stopping of the motor. A rotation angle estimating means for estimating a rotation angle when the brushless motor is rotating and outputting an estimated rotation angle signal;
A drive control unit configured to estimate the rotation angle by a process different from the rotation angle estimation unit and to drive and control the brushless motor when the determination unit determines that the brushless motor is stopped; The brushless motor control device according to claim 1. A brushless motor control device comprising: a brushless motor including a rotor and a stator having a plurality of phases of stator windings; and energization switching means for switching and controlling energization of the stator windings of each phase,
an induced voltage estimating means for setting a γδ coordinate system having a phase difference with respect to the dq coordinate system and estimating an induced voltage component generated on the γ axis and an induced voltage component generated on the δ axis;
A rotation angle estimating means for estimating a rotation angle of the brushless motor and outputting an estimated rotation angle signal based on a convergence calculation for converging the phase difference to zero;
Current control means for controlling energization by vector control using the γδ coordinate system based on the estimated rotation angle signal output from the rotation angle estimation means;
Based on whether the sum of the squares of the induced voltage component generated in the square and the δ-axis induced voltage component generated in the γ axis are estimated by the induced voltage estimation unit is larger than a predetermined value, the brushless A control device for a brushless motor, comprising: determination means for determining rotation and stop of the motor. When the determination means determines that the brushless motor is stopped, the rotation angle estimation at stop is performed by estimating the rotation angle by a process different from the rotation angle estimation means and outputting an estimated rotation angle signal at stop. 4. The brushless motor control apparatus according to claim 3, further comprising means. The brushless motor control device according to claim 4 ;
Steering torque detection means for detecting steering torque and outputting a steering torque signal;
The brushless motor is driven and controlled according to the steering torque signal output from the steering torque detection means and the estimated rotation angle signal or the estimated rotation angle signal at the time of stop output from the control device, and the steering An electric steering apparatus comprising: a steering control means for generating auxiliary torque for assisting torque from the brushless motor.

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