JP5167697B2 - Motor control method and motor control apparatus - Google Patents

Description

  The present invention relates to a motor control method for generating a PWM pulse based on a current command value of a motor and a switching frequency of the inverter, and controlling the PWM pulse generated for generating an inverter for driving the motor at a variable speed, and a motor control device for executing the control method It is.

  In the present invention, “switching frequency” means a frequency for controlling ON / OFF of a switch element constituting an inverter for driving a motor, and conventionally generates a PWM pulse. This is the same as the carrier frequency used in the above description, and is also referred to as the carrier frequency in the following description.

Conventionally, a motor control method for generating a PWM pulse based on a current command value of a motor and a switching frequency of the inverter and controlling the PWM pulse generated by an inverter that drives the motor at a variable speed, and a motor control device for executing the control method Various types are known. In the motor control method and the motor control device described above, for example, in Patent Document 1, the carrier frequency of the inverter is set low in order to reduce noise, and the carrier frequency is increased according to the current deviation. There is shown a method that makes it possible to maintain the response characteristics even when the value is lowered.
JP 2001-37248 A

  However, as shown in FIG. 11, when a large harmonic current is included in the motor current, it remains as a current control deviation even in a steady state. Therefore, in the above-described method for deriving the carrier frequency according to the current control deviation, Although the carrier frequency can be increased along with the increase of the control deviation, there is a problem that it is difficult to decrease the carrier frequency. In addition to the influence of the harmonic current generated by the induced voltage of the motor and the spatial harmonic of the inductance, this harmonic current has a lower carrier frequency, that is, lower harmonics such as the fifth and seventh orders. There was also a problem that would increase the current.

  An object of the present invention is to solve the above-described problems and reduce the switching loss of the inverter by lowering the carrier frequency even when the harmonic current is included in the motor current as described above. It is an object of the present invention to provide a motor control method and a motor control device that enable this.

  The motor control method of the present invention is a motor control method in which a PWM pulse is generated based on a motor current command value and an inverter switching frequency, and an inverter that drives the motor at a variable speed is controlled. The switching frequency is increased as the amount of change in at least one of the value, torque command value, voltage command value, speed command value, and position command value increases, and the at least one command value becomes constant and becomes a steady state. When it reaches, the switching frequency is lowered.

  The motor control device of the present invention generates a motor, an inverter for supplying electric power to the motor to drive the motor at a variable speed, and a PWM pulse for controlling the inverter based on the current command value of the motor and the switching frequency of the inverter. And a control means for controlling the switching frequency according to a change amount of at least one of the current command value, torque command value, voltage command value, speed command value, and position command value of the motor. And a calculating means for increasing the switching frequency as the amount of change in the at least one command value increases, and lowering the switching frequency when the at least one command value becomes constant and reaches a steady state. It is characterized by this.

  According to the motor control method and motor control apparatus of the present invention described above, the amount of change in the command value is detected, and when high responsiveness is required according to the change, the switching frequency / control cycle is changed to be high. High responsiveness can be obtained. Further, when the command value becomes constant and reaches a steady state, the switching frequency can be lowered and the switching loss of the inverter can be reduced. Therefore, even if the detected current value contains large harmonic currents or noise, the switching frequency can be changed according to the change in the command value. Can be reduced.

  Further, as a preferred example of the motor control method and motor control apparatus of the present invention, the amount of change in at least one command value among the current command value, torque command value, voltage command value, speed command value, and position command value of the motor In some cases, the switching frequency is obtained from the control deviation, the control frequency of the current control or the switching frequency is obtained from the estimated disturbance value, and one of the two switching frequencies is selected as the switching frequency. By configuring in this way, not only when the amount of change in the command value is detected and a high responsiveness is required according to the change, but also when the disturbance of current control such as induced voltage changes due to a change in load Even if disturbance suppression is required, it is possible to change the switching frequency and control period to be high by obtaining the switching frequency from either the estimated value of the disturbance or the current control deviation, thereby obtaining high responsiveness. it can. Further, when the command value and the disturbance become constant and reach a steady state, the switching frequency can be lowered and the switching loss of the inverter can be reduced.

  Further, as a preferred example of the motor control method and the motor control apparatus of the present invention, when the switching frequency is obtained from the change amount of at least one command value, the switching frequency is obtained by multiplying the change amount of the command value by a gain. Sometimes. With this configuration, by changing the switching frequency by multiplying the change amount of the command value by the gain, the switching frequency can be further increased and the response is high when the change amount of the command value is large. Can be secured.

  Furthermore, as a preferred example of the motor control method and motor control apparatus of the present invention, when the switching frequency is determined from the control deviation of the current control, a value obtained by removing the harmonic component from the deviation of the current control is calculated, and the calculated value There is a case where the switching frequency is calculated by multiplying the magnitude of the signal by a gain. With this configuration, by removing the harmonic component from the current control deviation, the switching frequency can be changed to be lower even in the case where a large amount of harmonic current is included even during steady state. The normal inverter switching loss can be reduced.

  Further, as a preferred example of the motor control method and motor control apparatus of the present invention, when obtaining the switching frequency from the control deviation of the current control, a low-pass having a cutoff frequency calculated from the deviation of the fundamental current control from the electrical angular frequency. The switching frequency may be calculated by inputting to the filter and multiplying the output value by the gain. With this configuration, even when the harmonic component is removed from the current control deviation by the low-pass filter, the cut-off frequency of the low-pass filter is changed according to the change in the motor speed. Harmonic components can be sufficiently removed, and switching loss of the inverter can be reduced.

  Furthermore, as a preferred example of the motor control method and motor control apparatus of the present invention, an observer for estimating a disturbance voltage value from a voltage command value, a current detection value, and a motor circuit constant when obtaining a switching frequency from the disturbance estimated value. Thus, the change amount of the disturbance voltage value may be calculated, and the switching frequency may be calculated by multiplying the absolute value by the gain. With this configuration, the amount of change in the command value is detected, and not only when responsiveness is required to be high in response to the change, but also by estimating the disturbance, current such as an induced voltage due to a change in load Even when disturbance disturbance is required when the disturbance of control changes, the carrier frequency / control period can be changed to be high, and high responsiveness can be obtained. Further, when the command value and the disturbance are constant and reach a steady state, the carrier frequency can be lowered and the switching loss of the inverter can be reduced.

  Furthermore, as a preferred example of the motor control method and motor control apparatus of the present invention, a d-axis disturbance voltage value and a q-axis disturbance voltage value are calculated from a dq-axis voltage command value and a current detection value, respectively, and a dq-axis disturbance is obtained. In some cases, a scalar of a disturbance voltage value is calculated from the voltage value, and a switching frequency is calculated by multiplying the absolute value of the change amount of the scalar by a gain. By configuring in this way, by estimating the disturbance on the dq axis, when the disturbance of the current control such as the dq axis induced voltage is changed by the change of the load, by estimating the change of the disturbance on both axes, High responsiveness can be obtained by changing the carrier frequency / control cycle to a high level. Further, when the command value and the disturbance are constant and reach a steady state, the carrier frequency can be lowered and the switching loss of the inverter can be reduced.

  Moreover, as a suitable example of the motor control method and motor control apparatus of this invention, it may comprise so that the calculated switching frequency may be restrict | limited between the preset upper limit and lower limit. With this configuration, the switching frequency obtained in the above example has an upper limit, and not only ensures stable operation within the drive frequency of the inverter device, but also stable at the switching frequency that is the limit of PWM drive. Can be realized. Moreover, at the time of low rotation, control responsiveness can be ensured by setting the lower limit value.

  Furthermore, as a suitable example of the motor control method and the motor control device of the present invention, any one of a switching frequency obtained from a change amount of a command value, a control deviation of current control, or a switching frequency obtained from an estimated disturbance value is used. May be configured to select a larger value. By configuring in this way, by selecting a switching frequency obtained from the command value and a large value of the switching frequency obtained from the estimated disturbance value or control deviation, both responsiveness when changing the command value and disturbance suppression are ensured. Can be obtained.

  Embodiments of a motor control method and a motor control apparatus according to the present invention will be described below with reference to the drawings.

<First embodiment>
In the first embodiment, a description will be given of a basic motor control method and a motor control device, and an example in which a change in carrier frequency, which is a feature of the present invention, is performed based on a motor current command value.

  FIG. 1 is a block diagram showing an example of the configuration of the control device in the first embodiment of the present invention. In the example shown in FIG. 1, a motor 101 is a three-phase permanent magnet synchronous motor, and drives a wheel 103 via a speed reducer 102. Although not shown, the inverter 104 that supplies power to the motor 101 converts DC power from the battery into UVW three-phase AC. The current of each phase is detected by the current sensor 105, and the detected values iu, iv, iw are input to the control device. Further, an electrical angle θe obtained by a position sensor attached to the motor 101, for example, a resolver is also input to the control device.

  Arithmetic processing in the control device is performed by digital control at the same frequency as the carrier frequency (switching frequency) fc of the inverter 104. First, using the detected three-phase alternating currents iu, iv, iw and the electrical angle θe, the coordinate converter 200 converts them into dq-axis current values id, iq. At this time, the input value is sampled and held at the carrier frequency. Although not shown in the figure, a motor controller that outputs a dq axis current command value from the torque command value and motor speed of the motor 101 obtains motor current command values id_cmd and iq_cmd. The difference between id and iq is calculated, and the control voltage is calculated by the PI controller 201. Further, non-interference control 205 for feedforward compensation of the induced voltage of the permanent magnet synchronous motor 101 is performed, and the control voltage is calculated from the current command values id_cmd and iq_cmd and the electrical angular velocity ωe. The electrical angular velocity ωe is obtained by differentiating the electrical angle θe by the electrical angular velocity calculator 208.

  The control voltage of the PI controller 201 and the control voltage of the non-interference control 205 are added to obtain the d-axis control voltage vd and the q-axis control voltage vq. The obtained vd and vq are subjected to coordinate transformation 202 to a three-phase AC voltage to obtain AC voltage command values vu, vv and vw. Next, in modulation factor calculation 203, each AC voltage command value is divided by Vdc / 2 using the DC voltage value Vdc of the inverter to calculate the modulation factor command value. The calculation result of the dead time compensation calculation 207 is added to this modulation factor command value in each phase. In the dead time compensation calculation 207, first, the coordinate converter 206 calculates a three-phase current command value from the dq axis current command values id_cmd and iq_cmd, and the dead time is calculated from the sign of the current command, the dead time of the switching element, and the carrier frequency fc. The voltage error equivalent is calculated and obtained as the value of the modulation factor.

  In the triangular wave PWM generation 204, as shown in the description of PWM pulse generation in FIG. 2 and an example of the configuration of the inverter 104 and the motor 101, the modulation factor command value of each phase has a value of −1 to +1, and the carrier cycle Tc. Compared with a triangular wave with a width of (1 / fc), a PWM pulse is generated. The modulation rate command value is held at the carrier period. Each switch of the inverter is driven using this PWM pulse as a gate signal.

  The above description is the same as the configuration of a general motor control method and motor control device.

Next, a description will be given of a calculation unit that determines the control period Tc of the motor current control and the carrier frequency fc for driving the inverter, which are characteristic parts of the motor control method and motor control apparatus of the present invention. In the example shown in FIG. 1, the current command values id_cmd and iq_cmd are set to 301, and the command value of the current vector is converted to a scalar Ia.
Ia = (id_cmd ² + iq_cmd ² ) ^1/2
This Ia is input to the high-pass filter (HPF) 302, and the change amount at a high frequency of the command value equal to or higher than the cut-off frequency among the change of the current command value is obtained as an output. This change amount is multiplied by a preset gain Kf303 to obtain a command value for the carrier frequency fc.

Simply multiplying the amount of change by a gain may result in an excessively high carrier frequency when the command value changes stepwise. In order to prevent this, an upper limit value is determined by the carrier frequency limiter 305. Further, in order to generate a three-phase alternating current with PWM, six on / off states are required within one voltage cycle, so the lower limit is set so that at least six carriers are obtained.
fc_min = 6 × fe = (3 / π) × ωe
Note that when the motor speed is low and ωe is small, this lower limit value is reduced and the control response is also lowered. Therefore, a lower limit value of the carrier frequency from the viewpoint of controllability is prepared separately, and fc_min is less than that value. It is restricted not to become.

  The carrier frequency fc calculated in this way is reflected in the triangular wave carrier frequency of the next output. In addition, since the control cycle is changed, the control cycle Tc necessary for the control gain, integration, HPF, and differentiation 208 of the PI controller is recalculated by the control gain / cycle calculation 306. The control gain and control cycle of this PI controller reflect values from the next output. Note that the dead time compensation calculation is performed using the carrier frequency at the next triangular wave output, and the dead time compensation calculation is performed using the updated fc within the period in which the fc is updated.

  An example of calculating the carrier frequency in this way is shown in FIG. In the example shown in FIG. 3, when the torque command value increases stepwise and the q-axis current command value iq_cmd changes stepwise, Ia similarly changes stepwise. The HPF output represents the amount of change in the current command value. When the current command value reaches a steady state, the HPF output also becomes zero. When this value is input to the carrier frequency limiter, the value is limited between a lower limit value fc_min obtained from the electrical angular velocity and an upper limit value fc_max obtained from a limit such as the switching speed of the switching element of the inverter, thereby obtaining fc.

  By calculating in this way, the amount of change in the current command value is detected, and when high responsiveness is required according to the change, the carrier frequency / control cycle is changed to be high and high responsiveness can be obtained. it can. Further, when the current command value becomes constant and reaches a steady state, the carrier frequency can be lowered and the switching loss of the inverter can be reduced. Even if the detected current value includes a large harmonic current or noise, the carrier frequency can be changed according to the change in the command value, thus reducing inverter switching loss during steady state. be able to.

<Second embodiment>
As a second embodiment, an example will be described in which a configuration for performing disturbance estimation with a disturbance observer is added to the first embodiment described above. In the description of the second embodiment, only differences from the first embodiment shown in FIG. 1 are shown.

  FIG. 4 shows a block diagram of the control device of the second embodiment. The observer 307 is composed of a disturbance observer and a block for calculating the amount of disturbance change, and receives the control voltages vd0 and vq0 of the PI controller 201 and the detected currents id and iq, and outputs the estimated amount of disturbance change. Details of the observer 307 are shown in FIG. In FIG. 5, a continuous system using the Laplace operator s is used, but when it is mounted on the control device, for example, bilinear transformation is performed to discretize it. In FIG. 5, R represents the resistance value of the motor, Ld and Lq represent inductance values in the respective dq axes, ωa represents the cutoff frequency of the low-pass filter, and ωh represents the cutoff frequency of the high-pass filter.

  A disturbance observer is constituted by an operation composed of vd, id and a motor circuit equation and a low-pass filter. The estimated disturbance voltage value {circumflex over (τ)} d is an estimation result of the disturbance voltage caused by the induced voltage generated on the d-axis, the on-voltage of the dead time device, or the like. The cut-off frequency of this low-pass filter is set so that a 6ωe disturbance voltage component that causes a harmonic current on the dq axis can be removed. Similarly, the disturbance voltage value ^ τq on the q axis is estimated. From the obtained ^ τd and ^ τq, the scalar voltage ^ τa of the disturbance voltage value is passed through an operator and passed through a high-pass filter to obtain a disturbance estimated value change amount.

  Returning to FIG. 4 again, a method for calculating the carrier frequency will be described. The carrier frequency is obtained by multiplying the output of the observer 307 by the gain Kfd308. This carrier frequency increases in accordance with the change in disturbance, and a carrier frequency limiter 309 is provided to set the upper limit and lower limit. This has the same upper and lower limit values as the limiter 305 set in the first embodiment. Thereafter, at 310, the maximum value is selected from the outputs of the carrier frequency limiters 305 and 309, and the next carrier frequency is calculated.

  An example of calculation of such a carrier frequency is shown in FIG. The operation using the current command value Ia is the same as in FIG. 3 of the first embodiment. Here, when the disturbance estimated value ^ τa changes, for example, when the wheel accelerates and the induced voltage on the dq axis increases, the amount of change of ^ τa is obtained from the output of the HPF. The fc obtained from the output of the HPF is larger than the carrier frequency fc obtained from the current command value Ia, and is reflected in the final carrier frequency fc.

  By calculating in this way, the amount of change in the current command value is detected, and not only when high responsiveness is required in response to the change, but also when current control disturbances such as induced voltage change due to load changes In addition, even when disturbance suppression is required, the carrier frequency / control period can be changed to be high and high responsiveness can be obtained. Further, when the current command value and the disturbance are constant and reach a steady state, the carrier frequency can be lowered and the switching loss of the inverter can be reduced.

<Third embodiment>
As a third embodiment, an example will be described in which a configuration in which the control deviation is passed through the LPF instead of the observer 307 in the second embodiment described above will be described. In the description of the third embodiment, only differences from the second embodiment shown in FIG. 4 are shown.

The configuration of the third embodiment is shown in FIG. In the third embodiment, current control deviations Δid and Δiq on the dq axis are respectively input to the low-pass filter LPF 401, and harmonic components included in the control deviation are removed. The cut-off frequency of the low-pass filter 401 is set to 6ωe / 10 so that the frequency component of 6ωe can be removed on the dq axis, and is set to 1/10 of the target frequency.
If the output of the low-pass filter is Δid_LPF and Δiq_LPF, the LPF output value of these control deviations is converted from a vector to a scalar ΔIa_LPF at 402.
ΔIa_LPF = (Δid_LPF ² + Δiq_LPF ² ) ^1/2
The carrier frequency is calculated by multiplying this ΔIa_LPF by the gain Kfe 403.

  By calculating in this way, the amount of change in the current command value is detected, and not only when the response is required to be high in response to the change, but also when the current control deviation changes due to a change in load, that is, disturbance suppression Can be obtained, a high response can be obtained by changing the carrier frequency and the control period to be high. Even in the case where it is difficult to configure the observer because the parameter information of the motor is not obtained, such as exchanging the motor to be controlled, the present embodiment can be configured.

<Fourth embodiment>
As a fourth embodiment, an example in which a configuration for performing harmonic control instead of the low-pass filter 401 in the third embodiment described above will be described. In the description of the fourth embodiment, only differences from the third embodiment shown in FIG. 7 are shown.

  The configuration of the fourth embodiment is shown in FIG. In the fourth embodiment, the current control deviations Δid and Δiq are converted by the coordinate converter 501 into a coordinate system dhqh coordinate that rotates in synchronization with the harmonics. Here, conversion to coordinates that rotate in synchronization with the fifth harmonic current in three-phase alternating current is performed, and conversion is performed using −6θe. The difference between the harmonic current command values idh_cmd, iqh_cmd and the harmonic current component in the dhqh coordinates is calculated, and the control voltage is obtained by the PI controller 502. Here, the same control gain as that of the dq coordinate PI controller is used, and idh_cmd and iqh_cmd are both 0. The obtained control voltage is coordinate-converted to the dq coordinate by the coordinate converter 503 and added to the dq-axis control voltage. In this way, the fifth harmonic current can be controlled to zero.

Using the current control deviations Δidh and Δiqh of the harmonic current control system, a scalar amount ΔIah of the control deviation of the harmonic current control is calculated at 504.
ΔIah = (Δidh ² + Δiqh ² ) ^1/2
The carrier frequency is calculated by multiplying ΔIah by the gain Kfe 403.

  In this way, when performing fundamental wave current and harmonic current control, the principal of the controlled object can be obtained in a reduced amount of current frequency by using the current control deviation in the rotating coordinates synchronized with the harmonic. For this reason, it can be obtained as an amount obtained by removing the harmonic component compared with the case where the control deviation on the dq axis is used. For this reason, it is not necessary to newly add an LPF, and an increase in the amount of calculation can be suppressed.

<Fifth embodiment>
As a fifth embodiment, an example in which a low-pass filter 401 is added to the above-described fourth embodiment will be described. In the description of the fifth embodiment, only differences from the fourth embodiment shown in FIG. 7 are shown.

  The configuration of the fifth embodiment is shown in FIG. FIG. 9 shows a configuration in which a low-pass filter LPF 401 is added to the fourth embodiment shown in FIG. When the motor includes not only the fifth order but also the seventh order harmonic current, the LPF 401 is provided in order to remove the influence of the remaining seventh order harmonic current. In the coordinate system synchronized with the fifth order, the seventh order has each frequency of 12ωe, and the cut-off frequency of the LPF 401 can be set to a higher value than when LPF is set with the dq coordinates. For this reason, only the fluctuation of the current due to the disturbance can be detected with higher responsiveness, and high responsiveness can be obtained by changing the carrier frequency.

<Regarding the upper control system of the current command value in the embodiment described above>
FIG. 10 shows an example of a control system higher than the current command values id_cmd and iq_cmd in the above-described embodiments (the first to fifth embodiments). In the example shown in FIG. 10, the speed controller 601 and the torque controller 602 constitute speed control, and the torque controller 602 refers to a current command value map created in advance as an output based on the speed command value ωe_cmd. Then, id_cmd and iq_cmd are derived. In this case, instead of the scalar amount obtained from the command values id_cmd and iq_cmd, the torque command value Te_cmd is obtained. Moreover, the structure which uses speed command value (omega) e_cmd instead of Te_cmd can also be used. Here, although not shown, when a position control feedback loop is configured in addition to the speed control loop, the position command value can be used as an input of the HPF.

  Therefore, in the first to fifth embodiments described above, an example in which the switching frequency (carrier frequency) fc is changed based on the current command value is shown, but in the present invention, instead of the current command value, A torque command value, a speed command value, and a position command value that constitute a higher-level control system of the current command value can be used. Further, since the current command value and the voltage command value are correlated, in the first to fifth embodiments described above, the voltage command value can be used instead of the current command value.

  According to the motor control method and the motor control device of the present invention, the switching frequency can be changed according to the change in the command value even when the detected current value includes a large harmonic current or noise. Therefore, it can be suitably applied to a motor for the purpose of reducing inverter switching loss during normal operation.

It is the block diagram which showed an example of the structure of the control apparatus in 1st Example of this invention. FIG. 6 is a diagram for explaining a method for obtaining a PWM pulse from a modulation factor command value in the first embodiment of the present invention. It is a figure for demonstrating the example which calculated the carrier frequency in 1st Example of this invention. It is the block diagram which showed an example of the structure of the control apparatus in 2nd Example of this invention. It is a block diagram for demonstrating the detail of the observer in the 2nd Example of this invention. It is a figure for demonstrating the example of a calculation of the carrier frequency in 2nd Example of this invention. It is the block diagram which showed an example of the structure of the control apparatus in the 3rd Example of this invention. It is the block diagram which showed an example of the structure of the control apparatus in the 4th Example of this invention. It is the block diagram which showed an example of the structure of the control apparatus in the 5th Example of this invention. It is a block diagram for demonstrating an example of the high-order control system of the electric current command value in the Example mentioned above. It is a figure for demonstrating the state in which a big harmonic current is contained in a motor current.

Explanation of symbols

DESCRIPTION OF SYMBOLS 101 Motor 102 Reduction gear 103 Wheel 104 Inverter 105 Current sensor 200, 202, 206, 501, 503 Coordinate converter 201, 502 PI controller 203 Modulation rate calculation 204 Triangular wave PWM generation 205 Non-interference control 207 Dead time compensation calculation 208 Electric angular velocity Arithmetic units 301, 402, 504 Converter 302 High-pass filter 303 Gain Kf
304 Gain Kfw
305, 309 Carrier frequency limiter 306 Control gain / cycle calculation 307 Observer 308 Gain Kfd
310 selection unit

Claims (10)

In a motor control method of generating a PWM pulse based on a motor current command value and an inverter switching frequency and controlling an inverter that drives the motor at a variable speed by the generated PWM pulse, the motor current command value, torque command value, voltage command value, velocity command value, at least one of increasing the switching frequency with increasing the amount of change in the command value, wherein if the at least one command value has reached a steady state becomes constant switching of the position command value Reduce the frequency ,
The switching frequency is determined from a change amount of at least one command value among the current command value of the motor, the torque command value, the voltage command value, the speed command value, and the position command value, and from the estimated disturbance value, Obtaining a switching frequency, selecting one of the two switching frequencies as the switching frequency;
When determining the switching frequency from the disturbance estimated value, the observer calculates the disturbance voltage value from the voltage command value, the current detection value, and the motor circuit constant, and calculates the amount of change in the disturbance voltage value to obtain the absolute value. A motor control method, wherein a switching frequency is calculated by multiplying a gain. In a motor control method of generating a PWM pulse based on a motor current command value and an inverter switching frequency and controlling an inverter that drives the motor at a variable speed by the generated PWM pulse, the motor current command value, torque command value, voltage The switching frequency is increased with an increase in the change amount of at least one of the command value, the speed command value, and the position command value, and the switching is performed when the at least one command value becomes constant and reaches a steady state. Reduce the frequency,
The switching frequency is determined from a change amount of at least one command value among the current command value of the motor, the torque command value, the voltage command value, the speed command value, and the position command value, and from the estimated disturbance value, Obtaining a switching frequency, selecting one of the two switching frequencies as the switching frequency;
A motor control method comprising: selecting a switching frequency that is larger of a switching frequency obtained from a change amount of the command value and a switching frequency obtained from the estimated disturbance value as the switching frequency. 3. The motor control method according to claim 1, wherein when the switching frequency is obtained from the change amount of the at least one command value, the switching frequency is obtained by multiplying the change amount of the command value by a gain. A d-axis disturbance voltage value and a q-axis disturbance voltage value are calculated from the dq-axis voltage command value and the current detection value, respectively, a scalar of the disturbance voltage value is calculated from the dq-axis disturbance voltage value, and the change amount of the scalar the motor control method according to claim 1 or 2, characterized in that for calculating the switching frequency multiplied by the gain to the absolute value of. The motor control method according to any one of claims 1 to 4, characterized in that to limit between the upper limit value and the lower limit value set the computed switching frequency in advance. A motor control device comprising: a motor; an inverter for supplying electric power to the motor to drive the motor at a variable speed; and a control means for generating a PWM pulse for controlling the inverter based on the current command value of the motor and the switching frequency of the inverter And a means for obtaining a switching frequency in accordance with a change amount of at least one command value among a current command value, a torque command value, a voltage command value, a speed command value, and a position command value of the motor, wherein the at least one command An arithmetic means for increasing the switching frequency with an increase in the amount of change in value, and lowering the switching frequency when the at least one command value becomes constant and reaches a steady state;
The switching frequency is determined from a change amount of at least one command value among the current command value of the motor, the torque command value, the voltage command value, the speed command value, and the position command value, and from the estimated disturbance value. Means for determining the switching frequency;
Means for selecting any one of the two switching frequencies as the switching frequency,
The means for obtaining the switching frequency from the estimated disturbance value is calculated from an observer for calculating a change amount of the disturbance voltage value estimated by estimating the disturbance voltage value from the voltage command value, the current detection value, and the motor circuit constant. Means for multiplying the change amount of the disturbance voltage value by a gain, calculating the change amount of the disturbance voltage value by the observer, and calculating the switching frequency by multiplying the absolute value by the gain. . A motor control device comprising: a motor; an inverter for supplying electric power to the motor to drive the motor at a variable speed; and a control means for generating a PWM pulse for controlling the inverter based on the current command value of the motor and the switching frequency of the inverter And a means for obtaining a switching frequency in accordance with a change amount of at least one command value among a current command value, a torque command value, a voltage command value, a speed command value, and a position command value of the motor, wherein the at least one command An arithmetic means for increasing the switching frequency with an increase in the amount of change in value, and lowering the switching frequency when the at least one command value becomes constant and reaches a steady state;
Together with means for obtaining the switching frequency from a change amount of at least one command value of the motor current command value, the torque command value, the voltage command value, the speed command value, and the position command value, from the estimated disturbance value Means for determining the switching frequency;
The calculation means includes means for selecting, as the switching frequency, a larger value of the switching frequency obtained from the change amount of the command value and the switching frequency obtained from a disturbance estimated value.
The motor control apparatus characterized by the above-mentioned. It means for determining a switching frequency from the change amount of the at least one command value includes means for multiplying a gain to the variation of the command value, characterized in that by multiplying the gain variation of the command value determining the switching frequency The motor control device according to claim 6 or 7 . The calculation means calculates a d-axis disturbance voltage value and a q-axis disturbance voltage value from the dq-axis voltage command value and the current detection value, and calculates a scalar of the disturbance voltage value from the dq-axis disturbance voltage value. and means, and means for multiplying a gain to the absolute value of the scalar amount of change, according to claim 6 or 7, characterized in that by multiplying a gain to the absolute value of the scalar change amount computing the switching frequency The motor control device described in 1. The motor control device according to any one of claims 6 to 9 , wherein the calculation means includes a limiter that limits the calculated switching frequency between a preset upper limit value and a lower limit value.

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