JP5169014B2 - Phase estimation method for AC motors - Google Patents

Description

The present invention generates a rotor phase estimation value or a rotor magnetic flux phase estimation value necessary for sensorless drive control of an AC motor including a synchronous motor such as a permanent magnet synchronous motor and a synchronous reluctance motor and an induction motor. It relates to a phase estimation method.

The rotor phase (N pole phase, negative salient pole phase) is required to achieve high control performance by using AC motors (synchronous motors such as permanent magnet synchronous motors and synchronous reluctance motors) that rotate synchronously at the same frequency as the power supply frequency. Etc.) is necessary. In addition, the phase of the rotor magnetic flux is necessary for an AC motor such as a permanent magnet synchronous motor or an induction motor having a rotor magnetic flux to exhibit high control performance. For this reason, position and velocity sensors such as encoders and resolvers are mounted on the rotor, and these phases are obtained directly or indirectly. However, the mounting of the position / speed sensor is problematic in terms of reliability, cost, and the like, and the rotor phase is estimated using the equivalent values of the stator voltage and current for driving the motor. As is well known to those skilled in the art, in a synchronous motor having a rotor magnetic flux such as a permanent magnet synchronous motor, the rotor phase and the rotor magnetic flux phase can be regarded as the same.

For rational drive control of an AC motor, a rotor phase is required for a synchronous motor, and a rotor magnetic flux phase is required for an induction motor. In driving without a position / velocity sensor, it is necessary to estimate these phases. In the following, the most traditional phase estimation method will be introduced by taking permanent magnet synchronous motor, synchronous reluctance motor, and induction motor as examples.

In the case of a non-salient permanent magnet synchronous motor, a mathematical model describing the relationship between the stator voltage and current and the rotor magnetic flux (usually called a voltage model) is expressed by the following equation on the fixed αβ coordinate system.

Here, 2 × 1 vectors ν ₁ and i ₁ mean the voltage and current of the stator, respectively. The 2 × 1 vector φ _i means the stator reaction magnetic flux, and the 2 × 1 vector φ _m means the rotor magnetic flux caused by the rotor permanent magnet. R ₁ is the resistance of the stator winding, L _d is the stator (d-axis) inductance, and s is the differential operator d / dt. In the case of a permanent magnet synchronous motor, the rotor phase θ _α is the same as the phase of the rotor magnetic flux φ _m , and this phase is included in the rotor magnetic flux as shown in equations (2) and (3). It is. Therefore, if the rotor magnetic flux can be estimated, the rotor phase estimated value can be obtained immediately from the rotor magnetic flux estimated value.

式右辺の四角括弧の中が、誘起電圧相当値を意味している。The most traditional method of estimating the rotor magnetic flux is to generate an induced voltage equivalent value by using the voltage / current of the stator, and approximate integration of the generated value as follows.

The square brackets on the right side of the equation mean the induced voltage equivalent value.

In principle, the processing for the induced voltage equivalent value needs to be a pure integration having a phase delay of −π / 2 (rad). However, in practice, various errors including voltage, current, parameters, etc. act as disturbances on the rotor flux estimate (especially this phase), and pure integration cannot be used for this relaxation. Considering the rotational speed of the motor, approximate integration with a large constant filter coefficient ω _c has been performed. In the approximate integration, in order to theoretically have a phase delay of −π / 2 (rad), the frequency of the signal to be processed must be infinite. In order to have a phase delay of approximately −π / 2 (rad), the frequency of the signal to be processed must be at least 10 times the constant filter coefficient. Therefore, in the estimation at the time of driving in the low to medium speed range, if approximate integration with a large constant filter coefficient is used, an appropriate phase delay of −π / 2 (rad) cannot be obtained, and the phase delay is insufficient. To do. A large shortage of phase delay means a large phase shift in the phase advance direction in the rotor phase estimation value. The phase error of the phase estimation value in the phase estimation method is a fatal defect. In order to reduce the phase advance in the phase estimation value, the above-described constant filter coefficient may be reduced. However, in this case, the above-described problem of mitigating the disturbance influence becomes obvious. Actually, this estimation method has been used in a limited manner by limiting the driving speed range while compromising the phase deviation of the phase estimation value and the disturbance influence mitigation.

In the case of a synchronous reluctance motor having salient pole characteristics, a voltage model describing the relationship of the stator is expressed by the following equation on the fixed αβ coordinate system.

また、ｉ_ｄ、ｉ_ｑは固定子のｄ軸、ｑ軸電流であり、ω_２ｎは回転子の電気速度である。また、Ｊは次式で定義された２ｘ２交代行列である。

Here, L _i and L _m are in-phase inductance and mirror-phase inductance having the following relationship with the stator dq inductance.

Further, i _d and i _q are the d-axis and q-axis currents of the stator, and ω _2n is the electric speed of the rotor. J is a 2 × 2 alternating matrix defined by the following equation.

In synchronous reluctance motor, (6) the magnetic flux phi _m of Formula is part of the stator reaction magnetic flux. However, as is clear from the comparison between Equations (5) and (6) and Equations (1) and (2), the magnetic flux of Equation (6) is equivalent to the rotor magnetic flux of Equation (2) on the model. Can be handled. In the present specification, in consideration of this equivalence, hereinafter, the magnetic flux of the formula (6) is referred to as a pseudo rotor magnetic flux or simply a rotor magnetic flux.

(5), which means the theta _alpha in equation (6), the rotor phase (negative salient pole or positive salient pole of the rotor phase). That is, if the designer selects the d-axis as a negative salient pole, it means the phase of the negative salient pole, and if the designer selects the d-axis as a positive salient pole, it means the phase of the positive salient pole. In either case, it means the d-axis phase selected by the designer. As is clear from the comparison between the equations (5) and (6) and the equations (1) and (2), the voltage model of the synchronous reluctance motor has a high similarity to the voltage model of the permanent magnet synchronous motor. . As understood from this similarity, the relationship of the equation (4) for the permanent magnet synchronous motor can be used for the rotor phase estimation of the synchronous reluctance motor. As long as the relationship of equation (4) is used, the problem of phase shift in the phase estimation value also occurs. In the case of a synchronous reluctance motor, the value corresponding to the right parenthesis in equation (4) (that is, the integrated value or approximated integrated value) is an extended induced voltage equivalent value.

ここに、φ_ｍは誘導電動機における回転子磁束であり、φ_ｉは固定子反作用磁束である。Ｌ_１ｔは固定子の総合漏れインダクタンスである。誘導電動機のセンサレス駆動においては、回転子磁束の位相を把握する必要があるが、当業者には周知にように、本把握は回転子磁束の推定を通じ直ちに行うことができる。（９）式と（１）、（２）式との比較より明白なように、誘導電動機の電圧モデルは、永久磁石同期電動機の電圧モデル（特に、非突極の電圧モデル）と高い類似性を有している。本類似性より理解されるように、誘導電動機の回転子磁束位相推定に関しては、永久磁石同期電動機のための（４）式の関係が利用可能である。
（４）式の関係を利用する限りは、位相推定値における位相の狂いの問題も、同様に発生する。なお、誘導電動電動機の場合には、（４）式の右辺括弧に対応する値（すなわち、被積分値あるいは被近似積分値）は、誘起電圧相当値となる。The voltage model describing the relationship of the induction motor stator is expressed by the following equation on the fixed αβ coordinate system.

Here, φ _m is the rotor magnetic flux in the induction motor, and φ _i is the stator reaction magnetic flux. L _1t is the total leakage inductance of the stator. In the sensorless drive of the induction motor, it is necessary to grasp the phase of the rotor magnetic flux, but as is well known to those skilled in the art, this grasp can be immediately performed by estimating the rotor magnetic flux. As is clear from the comparison between the expression (9) and the expressions (1) and (2), the voltage model of the induction motor is highly similar to the voltage model of the permanent magnet synchronous motor (particularly, the voltage model of a non-salient pole). have. As understood from this similarity, the relationship of the formula (4) for the permanent magnet synchronous motor can be used for the rotor magnetic flux phase estimation of the induction motor.
As long as the relationship of equation (4) is used, the problem of phase shift in the phase estimation value also occurs. In the case of an induction motor, the value corresponding to the right parenthesis in equation (4) (that is, the integrated value or approximated integrated value) is an induced voltage equivalent value.

A phase shift in the phase estimate is a fatal defect. The only way to fix a fatal defect is to correct the phase error. Phase correction generally uses a dynamic phase corrector, as is often used in controller design. This type of dynamic phase compensator is useful for rough phase correction such as “generally phase advance in a certain frequency band”, but it is not as precise as rotor phase estimation or rotor flux phase estimation. It is not suitable for accurate phase estimation. In fact, in this type of phase estimation method, it seems that the phase correction method using a dynamic phase corrector is hardly utilized.

Problems to be solved by the invention

The present invention has been made under the above-mentioned background, and its purpose is to change the rotor phase or rotor magnetic flux phase required for sensorless drive control of an AC motor including a synchronous motor and an induction motor. It is to provide a novel phase estimation method that can be estimated in a state that does not appear and in which a disturbance influence is suppressed.

Means for solving the problem

In order to achieve the above object, the invention described in claim 1 is a method for estimating a phase of a rotor of an AC motor in which a rotor rotates in synchronization with a stator power supply frequency. the corresponding value was at least using generated, and the equivalent value of the induced voltage or the extended electromotive force caused by the rotation of the rotor, speed応帯frequency response and the rotor of the non-changing the pass bandwidth in accordance with the rotation of the rotor Processing with a high-order filter having a phase lag characteristic of -π / 2 (rad) at a frequency corresponding to zero rotation speed and an attenuation characteristic of -40 (dB / dec) or higher in the high band, and non-zero rotation of the rotor A filtered signal mainly including a fundamental wave component having a frequency corresponding to the speed is generated, and a phase estimation value of the rotor is generated from the filtered signal.

The invention according to claim 2 is a method for estimating the phase of a rotor magnetic flux of an AC motor having a rotor magnetic flux, wherein the rotation of the rotor magnetic flux is generated using at least the equivalent values of the voltage and current of the stator. -Π / 2 at the frequency corresponding to the non-zero rotation speed of the rotor flux and the speed response band characteristic that changes the passband width according to the rotation of the rotor magnetic flux. The fundamental wave component of the frequency corresponding to the non-zero rotational speed of the rotor magnetic flux is processed using a high-order filter having a phase delay characteristic of (rad) and an attenuation characteristic of −40 (dB / dec) or higher in the high frequency range. And a phase estimation value of the rotor magnetic flux is generated from the filtered signal.

The invention of claim 3 is the phase estimation method for an AC motor according to any one of claims 1 to 2, wherein the high-order filter having a quick response band characteristic is The low-pass filter has an attenuation characteristic of 40 (dB / dec).

Next, the operation of the present invention will be described. According to the first aspect of the present invention, for a synchronous motor whose rotor rotates in synchronization with the stator power supply frequency, an equivalent of an induced voltage or an extended induced voltage generated using at least the equivalent values of the stator voltage and current. The value is processed by a high-order filter having a rapid response band characteristic that changes the pass band width according to the rotation of the rotor and an attenuation characteristic of −40 (dB / dec) or higher in the high band, and from the signal to be filtered The phase estimation value of the rotor is generated.

ここに、ζ，ω_ｆはフィルタ係数である。In order to simplify the description, the following second-order low-pass filter is considered as a high-order filter (hereinafter, abbreviated as a fast-response band high-order filter) having a rapid response band characteristic according to the present invention.

Here, ζ and ω _f are filter coefficients.

FIG. 1 shows the frequency characteristics (amplitude characteristics and phase characteristics) of the equation (10) under the conditions of ζ = 2 and ω _f = 100π (rad / s). As can be easily understood from the figure, in the second-order low-pass filter of equation (10), the pass bandwidth is approximately ω _f , and the amplitude attenuation is −40 (dB / dec) in the high frequency range beyond ω _f. And the phase lag at the frequency ω = ω _f is exactly −π / 2 (rad). This feature in the frequency characteristics, coefficient zeta, a value how involved not immutable omega _f. phase lag in omega = omega _f is the point is the same as the phase lag -π / 2 (rad) of the pure integration is to be particularly noted.

In the synchronous motor in which the rotor rotates in synchronization with the stator power supply frequency, it is assumed that other than the rotor magnetic flux (including the equivalent rotor magnetic flux) is known, and according to the invention of claim 1, the voltage and current of the stator are used. When the induced voltage or the equivalent value of the extended induced voltage is processed by the fast response band high-order filter, the following relationship is obtained.

すなわち、中心周波数では、位相偏移（位相進み・位相遅れ）は発生しない。純粋積分処理と同一のローパスフィルタ位相遅れ−π／２（ｒａｄ）が、回転子磁束に代わって誘起電圧あるいは拡張誘起電圧を処理することにより生じた等価な微分処理に伴う位相進み＋π／２（ｒａｄ）により、相殺された形となっている。(11) Equation F (s.ω _f ) s in the third equation means a band pass characteristic having a pass band with ω = ω _f as a center frequency, and has the following property at the center frequency.

That is, no phase shift (phase advance / phase lag) occurs at the center frequency. The same low-pass filter phase lag -π / 2 (rad) as in the pure integration process is the phase lead + π / 2 (with the equivalent differential process generated by processing the induced voltage or the extended induced voltage instead of the rotor magnetic flux. rad) is offset.

FIG. 2 shows the frequency characteristic of F (s.ω _f ) s corresponding to FIG. For rotor flux, (12) was added to the characteristics of the type, in the high frequency range of omega _f beyond, that amplitude attenuation of -20 (dB / dec) is maintained is confirmed. On the other hand, the effect of the second-order low-pass filter is effective against various error components that are included in the induced voltage equivalent value or the extended induced voltage equivalent value generated using the stator voltage / current and belong to the high frequency range. Yes, it will be attenuated by -40 (dB / dec) (see FIG. 1).

If it is considered that the fundamental wave component of the rotor magnetic flux is the same as the electric speed of the rotor, the filter coefficient ω _f is selected equal to the absolute value of the electric speed (that is, ω _f = | ω _2n FIG. 2 shows that the rotor magnetic flux fundamental wave component can be detected without phase error. Further, when estimating the rotor magnetic flux including the harmonic component caused by non-sinusoidal magnetization, the amplitude characteristic of FIG. 2 is attenuated by the harmonic component included in the estimated value of the rotor magnetic flux. The value means more sinusoidal. Specifically, the nth harmonic component of the rotor magnetic flux is attenuated by −20 logn (dB). For example, the third and fifth harmonic components that are the main components are attenuated by −9.5 (dB) and −14.0 (dB), respectively. As is apparent from the above description using the expression (11), the signal to be filtered by the high-order filter is the rotor magnetic flux estimated value (including the equivalent rotor magnetic flux estimated value) itself.

The rotor phase estimation value is immediately determined through the arc tangent process of the rotor magnetic flux estimation value, which is the filtered signal by the high-order filter. If this is taken into consideration, it is important to estimate the rotor magnetic flux for this purpose without estimating the phase shift of only the fundamental wave component. When ω _f = | ω _2n | is selected, the frequency characteristic of FIG. 2 showing the characteristic without phase shift at the frequency ω = ω _f and the attenuation characteristic in the high frequency range beyond ω _f is the rotor magnetic flux estimation. From above, it is a particularly preferable characteristic. According to the present invention of claim 1, this characteristic is obtained by processing an induced voltage equivalent value or an extended induced voltage equivalent value with a high-order filter (a filter having an amplitude attenuation of −40 (dB / dec) or higher in a high band). Special attention should be paid to the fact that it was obtained.

The selection of ω _f = | ω _2n | in the above example, that is, the change of the low-pass filter coefficient ω _f according to the rotor electrical speed ω _2n is that the filter coefficient ω _f and the filter pass bandwidth are substantially the same. Is taken into account, it means that the pass bandwidth of the low-pass filter changes according to the speed, which means that the quick response bandwidth is established. Thus, according to the first aspect of the present invention, the high-order filter has a quick response band characteristic that changes the pass band width according to the rotation of the rotor. As a result, the effect | action which reduces and suppresses various errors contained in the voltage, electric current, parameter, etc. which act as a disturbance to a phase estimated value at the speed according to a rotational speed is acquired.

As apparent from the above description, according to the invention of claim 1, in the phase estimation of the rotor of the synchronous motor in which the rotor rotates in synchronization with the stator power supply frequency, in the AC motor, the rotor Estimates the fundamental wave component of magnetic flux (including equivalent rotor magnetic flux) with no phase error, and with various errors in voltage, current, parameters, etc. reduced / suppressed according to the rotational speed The effect that it can be obtained.

Next, the operation of claim 2 will be described. The invention of claim 2 is directed to a synchronous motor having a rotor magnetic flux, such as a permanent magnet synchronous motor, or an AC motor including an induction motor, and an induced voltage generated using an equivalent value of a voltage and a current of a stator. The equivalent value of the expansion induced voltage is processed by a high-order filter having a quick response band characteristic that changes the pass band width according to the rotation of the rotor magnetic flux and an attenuation characteristic of −40 (dB / dec) or higher in the high band. The phase estimation value of the rotor magnetic flux is generated from the filtered signal.

The only difference between the invention of claim 2 and the invention of claim 1 resides in that the high-speed filter has a quick response band characteristic according to the rotor magnetic flux speed instead of the rotor speed. For the sake of simplicity, if a description is given using the secondary filter of the equation (11), the variable filter coefficient ω _f that defines the filter bandwidth is changed to “rotor magnetic flux speed” instead of “rotor speed”. The point is to change it accordingly. With respect to other aspects, there is no difference between the invention of claim 2 and the invention of claim 1. Therefore, the explanation of the operation of the invention of claim 2 can be made by simply replacing “rotor speed” with “rotor magnetic flux speed” formally in the explanation after the expression (10) relating to the invention of claim 1. To establish. As a matter of course, in this replacement, ω _2n indicating the rotor speed needs to be replaced with the rotor magnetic flux speed. As is clear from the above replacement, the operation of the invention of claim 2 is basically the same as the operation of the invention of claim 1 except for the difference in applicable electric motors. Also in the case of the invention of claim 2, the signal to be filtered by the high-order filter is the rotor magnetic flux estimation value itself.

The validity of the action by the above replacement is clearer when a synchronous motor having a rotor magnetic flux such as a permanent magnet synchronous motor is considered as an AC motor. As is well known to those skilled in the art, in a synchronous motor having a rotor magnetic flux, the rotor phase and the rotor magnetic flux phase are the same, and the rotor speed and the rotor magnetic flux speed are the same. Therefore, when the synchronous motor having the rotor magnetic flux is targeted, there is no difference between the invention of claim 2 and the invention of claim 1. Naturally, both inventions exhibit the same effect.

As apparent from the above, according to the invention of claim 2, in the phase estimation of the rotor magnetic flux of the AC motor having the rotor magnetic flux, the fundamental wave component of the rotor magnetic flux is further reduced in a state without phase error. Thus, an effect is obtained that estimation can be performed in a state where various errors included in the voltage, current, parameter, and the like are reduced / suppressed according to the rotation speed.

Next, the operation of the invention of claim 3 will be described. As is apparent from the description of the operation of claim 1, particularly the description using the equation (10), the harmonic component contained in the rotor magnetic flux (including the equivalent rotor magnetic flux) is removed, and the fundamental wave component is extracted. In order to estimate, it is necessary to provide a high-order filter having a quick response band characteristic with an attenuation characteristic of −40 (dB / dec) in the high band at least. In other words, if the high-order filter has an attenuation characteristic of −40 (dB / dec) in the high band, the harmonic component contained in the rotor magnetic flux (including the equivalent rotor magnetic flux) is removed, and the fundamental wave It becomes possible to extract and estimate the components. According to the invention of claim 3, since the high-order filter having the quick response band characteristic is a low-pass filter having an attenuation characteristic of −40 (dB / dec) in the high band, the effect of claim 1 or claim 2 is achieved. Can be obtained with a minimum processing load.

測定値を、固定子電圧相当値としては電圧指令値を利用する例となっている。Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. FIG. 3 shows an embodiment of a sensorless vector control system (torque control mode) using the phase estimation method of the present invention for a permanent magnet synchronous motor. In FIG. 3, the devices other than the phase determiner 2 have the same configuration as that of a normal sensor utilization vector control system. Since the functions of these devices are easily understood by those skilled in the art, description thereof is omitted. For reference, the meaning of the device in the drawing is given in “Explanation of Symbols” at the end of this specification, so please refer to it. The core of the present invention is the phase determiner 2. The purpose of the phase determiner is to use the fixed voltage and current equivalent values of the stator to determine the fixed αβ coordinates.

In this example, the voltage command value is used as the measured value corresponding to the stator voltage equivalent value.

FIG. 4 shows an example of the internal configuration of the phase determiner in FIG. The phase determiner 2 includes a fast response band high-order filter 2a, a speed estimator 2b, and a coefficient generator 2c according to the present invention.

子電圧・電流の相当値としての固定子電流実測値と固定子電圧指令値に加えて、最終回転

請求項１乃至請求項３のすべての発明に合致した、更には（１０）〜（１２）式の説明に利用した２次フィルタを採用した例を示している。本例では、固定子の電圧・電流相当値に対する処理を、次の（１３）式のように行なっている。

すなわち、固定子反作用磁束微分値の入力に代わって、固定子反作用磁束そのものを速応帯域高次フィルタに入力するようにしている。当然のことながら、この場合の固定子反作用磁束の入力すべき端子は、同図に明示したように、電圧相当値を入力する端子と異なることになる。The fast response band high-order filter 2a outputs a rotor magnetic flux estimated value on a fixed αβ coordinate system as a filtered signal, and through this arctangent calculation, a rotor phase estimated value (initial rotor phase estimated value).

In addition to the actual stator current value and the stator voltage command value as the equivalent values of the child voltage and current, the final rotation

An example is shown in which a second-order filter that matches all the inventions of claims 1 to 3 and that is used in the explanation of the expressions (10) to (12) is employed. In this example, processing for the voltage / current equivalent value of the stator is performed as shown in the following equation (13).

That is, instead of inputting the stator reaction magnetic flux differential value, the stator reaction magnetic flux itself is input to the fast response band high-order filter. As a matter of course, the terminal to which the stator reaction magnetic flux is to be input in this case is different from the terminal to which the voltage equivalent value is input as clearly shown in FIG.

ここに、Ｉは２ｘ２単位行列であり、Ｑ（・）は次式で定義された鏡行列である。

図５では、最終位相推定値を利用した固定子反作用磁束生成の様子（ブロック２ａ−１）を破線の貫徹矢印で表現している。非突極の場合には、固定子反作用磁束φ_ｉは、最終位相推定値の要なく、次の（１６）式のように生成できる。

When the target permanent magnet synchronous motor has salient pole characteristics, the stator reaction magnetic flux φ _i is

Here, I is a 2 × 2 unit matrix, and Q (•) is a mirror matrix defined by the following equation.

In FIG. 5, the state of the stator reaction magnetic flux generation using the final phase estimated value (block 2a-1) is expressed by a penetrating broken arrow. In the case of a non-salient pole, the stator reaction magnetic flux φ _i can be generated as in the following equation (16) without the need for the final phase estimation value.

The speed estimator 2b is configured according to an integral feedback type speed estimation method. The details of the integral feedback type speed estimation method are described in detail in the following document, and the description thereof will be omitted.
(1) Shinnaka Shinji: “Proposal of minimum dimension D-factor state observer of permanent magnet synchronous motor and sensorless vector control method using it”, IEEJ Transactions D Volume, Vol. 123, no. 12, pp. 1446-1460 (2003-12)
The speed estimator 2b receives the initial rotor phase estimated value as an input, and outputs the final rotor phase estimated value and the same electric speed estimated value (see FIG. 4). The estimated value of the rotor electrical speed is used as feedback to the coefficient generator 2c. In the case of the speed control mode, the estimated electric speed value is also output to the outside of the phase determiner 2. In FIG. 4, this state is indicated by a broken line.

リミッタ処理の３処理を施して得た最終処理信号をフィルタ係数ω_ｆとし、実質的に本関係を達成している。The coefficient generator 2c processes the electrical speed estimation value obtained from the speed estimator, and quickly generates the variable coefficient ω _f of the fast response band high-order filter. In order to obtain the desired effect with the rapid response band high-order filter according to the present invention, it is necessary to substantially satisfy the relationship of ω _f = | ω _2n | with respect to the filter coefficient ω _f (note that the rapid response band) In the case of the second-order filter, the coefficient ζ is constant and has no problem). coefficient

The final processed signal obtained by applying the 3 treatment of the limiter processing and the filter coefficient omega _f, have achieved substantially the relationship.

り、簡単には、次の一定係数をもつローパスフィルタによる処理でよい。

このときのフィルタは次の１次のものでよい。

脈動除去処理は、ローパスフィルタ処理に代わって、次の機能をもつマルチレイトサンプルホールダ処理でも達成できる。

絶対値処理は、フィルタ係数ω_ｆが正であることを保証するためのものである。下限リミッタ処理は、フィルタ係数ω_ｆの下限値をある正の微小値に設定し、計算誤差が存在する場合にも、速応帯域高次フィルタの不安定化を防止するためのものである。

In short, processing by a low-pass filter having the following constant coefficient may be used.

The filter at this time may be the following primary filter.

The pulsation removing process can be achieved by a multi-rate sample holder process having the following functions instead of the low-pass filter process.

The absolute value processing is for assuring that the filter coefficient ω _f is positive. Lower limiter processing is set to a positive small value with the lower limit value of the filter coefficients omega _f, when the calculation error is present is also intended to prevent the destabilization of the speed応帯range order filter.

からのフィルタ係数ω_ｆの生成例として、（１７）式のローパスフィルタを用いた例を示している。また、同図では、絶対値処理と下限リミッタ処理とを、１つのブロックで表現している。

As an example of generating the filter coefficient ω _f from, an example using the low-pass filter of the equation (17) is shown. In the figure, the absolute value process and the lower limiter process are expressed by one block.

In the above embodiment, the fast response band high-order filter is a second-order low-pass filter in order to ensure the simplicity and concreteness of the description. In general, it should be pointed out that an n-order low-pass filter can be used as the fast response band high-order filter. For example, when the fast response band high-order filter is third order, the following is useful.

In the above embodiment, the fast response band high-order filter is configured on the fixed αβ coordinate system (see FIG. 3). The fast response band high-order filter on this coordinate system is one in which two one-input one-output filters are arranged in parallel (see equation (11)). Alternatively, the fast response band high-order filter can also be configured on a quasi-synchronous γδ coordinate system aimed at synchronization without a phase difference in the rotor phase. FIG. 7 shows an example of this embodiment. The phase determiner 2 is different from the embodiment in FIG. 3 in that the phase determiner 2 is arranged on the left side of the vector rotators 6a and 6b.

FIG. 8 shows an example of the internal configuration of the phase determiner in FIG. The phase determiner 2 includes a fast response band high-order filter 2a according to the present invention, a phase synchronizer 2d, and a coefficient generator 2c.

The fast response band high-order filter for the quasi-synchronous γδ coordinate system can be obtained by formally replacing the Laplace operator of the fast response band high-order filter for the fixed αβ coordinate system with the D factor. As a result, the fast response band high-order filter on the quasi-synchronous γδ coordinate system becomes a single 2-input 2-output D-factor filter. FIG. 9 shows a fast response band high-order filter for the quasi-synchronous γδ coordinate system corresponding to the fast response band high-order filter for the fixed αβ coordinate system of FIG. The D factor in the figure is a 2 × 2 matrix defined as follows.

当然のことながら、非突極の場合には、Ｌ_ｄ＝Ｌ_ｑ＝Ｌ_ｉであり、固定子反作用磁束は、固定子電流相当値にスカラーのインダクタンスを乗ずることにより、直ちに生成される（（１６）式参照）。準同期γδ座標系用の速応帯域高次フィルタは、準同期γδ座標系上での回転子磁束を推定しており、最終的には、被フィルタ処理信号である回転子磁束推定値を逆正接処理し、同位相を生成・出力している。本位相は、準同期γδ座標系上からみた回転子位相推定値となっている。In FIG. 9, the factor D is used as an inverse matrix, but since this realization method has already been introduced in detail in the above document (1), this description is omitted. In FIG. 9, the stator reaction magnetic flux input to the fast response band high-order filter is generated as follows in consideration of the quasi-synchronization of the coordinate system.

Of course, in the case of a non-salient pole, L _d = L _q = L _i and the stator reaction magnetic flux is immediately generated by multiplying the value corresponding to the stator current by the scalar inductance (( (See 16)). The fast response band high-order filter for the quasi-synchronous γδ coordinate system estimates the rotor magnetic flux on the quasi-synchronous γδ coordinate system, and finally reverses the rotor flux estimation value that is the filtered signal. Tangent processing is performed to generate and output the same phase. This phase is a rotor phase estimation value viewed from the quasi-synchronous γδ coordinate system.

The phase synchronizer 2d is configured according to a generalized integral PLL method. The phase synchronizer 2d receives the rotor phase estimation value viewed from the quasi-synchronous γδ coordinate system, and outputs the rotor phase estimation value and the electric speed estimation value on the fixed αβ coordinate system. The details of the generalized integral PLL method are described in detail in the above-mentioned document (1), and thus the description thereof is omitted.

また、準同期γδ座標系で速応帯域高次フィルタを構成する場合には、これへの入力である拡張誘起電圧の相当値を、固定子の電圧・電流相当値を用いて以下の関係に基づき生成すればよい。

The embodiment described with reference to FIGS. 3 to 9 is an embodiment of the phase estimation method for a permanent magnet synchronous motor. However, even in the case of a synchronous reluctance motor, the stator reaction magnetic flux This can be used as is, with only minor changes to the generation. Specifically, when a fast response band high-order filter is configured in a fixed αβ coordinate system, an equivalent value of an extended induced voltage that is an input to this is expressed as follows using a voltage / current equivalent value of a stator. What is necessary is just to produce | generate based on a relationship.

Also, when configuring a fast response band high-order filter in a quasi-synchronous γδ coordinate system, the equivalent value of the extended induced voltage that is input to this is expressed as follows using the voltage / current equivalent value of the stator. It may be generated based on this.

As described above, the embodiment described with reference to FIGS. 3 to 9 is the case of the synchronous reluctance motor only by changing the input to the fast response band high-order filter to the equation (23) or (24). This is an example embodiment. In the case of synchronous reluctance synchronization, it is more convenient to apply the input signal to the fast response band high-order filter without dividing it into two according to the left side of equation (13).

Subsequently, an embodiment example in the case of targeting an induction motor will be described. The rotor magnetic flux estimation method for the induction motor is also related to the generation of the stator reaction magnetic flux in comparison with the embodiment example of the phase estimation method for the permanent magnet synchronous motor described with reference to FIGS. You can use it as it is with just a few changes. Specifically, in FIGS. 5 and 9 showing the configuration of the fast response band high-order filter, the following relation is applied to the stator reaction magnetic flux generation block 2a-1 to generate the stator reaction magnetic flux. (See equation (9)).

回転子磁束速度推定値と読み替える必要がある。繰返し説明しているように、永久磁石同期電動においては、回転子速度も回転子磁束速度も同一である。誘導電動機においては、この同一性は一般には成立しない。回転子磁束の推定に必要な速度情報は、回転子磁束の速度である。As is clear from the comparison between the equations (25) and (16), the embodiment in the case of targeting an induction motor is the embodiment and the form in the case of a non-salient permanent magnet synchronous motor. It becomes completely the same. However, the phase estimation for the permanent magnet synchronous motor described with reference to FIGS.

It should be read as the rotor flux velocity estimate. As described repeatedly, in the permanent magnet synchronous motor, the rotor speed and the rotor magnetic flux speed are the same. In an induction motor, this identity generally does not hold. The speed information necessary for estimating the rotor magnetic flux is the speed of the rotor magnetic flux.

As described above, when the rotor phase estimation method and the rotor magnetic flux phase estimation method according to the present invention are configured on a fixed αβ coordinate system for permanent magnet synchronous motors, synchronous reluctance motors, and induction motors, The case where it is configured with is described in detail. It should be pointed out that the present invention estimates these phases using the electrical relationship of the stator and is applicable to other AC motors having the same stator. Other AC motors having the same stator include, for example, a wound synchronous motor, a hybrid field synchronous motor having a winding and a permanent magnet in the rotor, and a synchronous permanent magnet and an induction cage in the rotor There are hybrid electric motors.

In the embodiment described with reference to FIGS. 3 to 9, the voltage command value is used as the voltage equivalent value, but an actual voltage measurement value or other approximate value may be used. Moreover, although the current measured value is used as the current equivalent value, a current command value or another approximate value may be used.

In the embodiment using FIG. 3 and FIG. 7, the present invention has been described for the torque control mode in which the estimation of the rotor phase or the rotor magnetic flux phase is the key to the drive control. As will be readily understood by those skilled in the art, the present invention is not limited to the torque control mode, but can be applied to driving in other modes such as a speed control mode.

In the embodiment using FIG. 3 and FIG. 7, the current control is performed in a feedback manner. It should be pointed out that the present invention is applicable even when the current control is performed in a feedforward manner or when the current control is not performed.

Although the phase determiner according to the present invention can be implemented in an analog manner, it is preferably constructed digitally in view of the remarkable progress of recent digital technology. Although the digital configuration includes a hardware configuration and a software configuration, as will be apparent to those skilled in the art, any of the present invention can be configured. The present invention has been described in detail with reference to various embodiments using various drawings.

Effect of the invention

As is clear from the above description, the present invention has the following effects. According to the first aspect of the present invention, in the estimation of the phase of the rotor of the synchronous motor in which the rotor rotates in synchronization with the stator power supply frequency among the AC motors, the rotor magnetic flux (including the equivalent rotor magnetic flux) is used. The fundamental wave component can be estimated in a state where there is no phase error, and further, in a state where various errors included in the voltage, current, parameter, and the like are reduced / suppressed according to the rotational speed. As a result, as a result of this action, according to the first aspect of the present invention, the rotor phase is calculated from the rotor magnetic flux estimated value (including the equivalent rotor magnetic flux estimated value) of the synchronous motor in a state where no phase deviation occurs. The effect that it can estimate in the state which suppressed disturbance influence is acquired.

Next, the effect of the present invention of claim 2 will be described. According to the invention of claim 2, in the phase estimation of the rotor magnetic flux of the AC motor having the rotor magnetic flux, the fundamental wave component of the rotor magnetic flux is in a state without phase error, and further, the voltage, current, parameter, etc. The effect that it is possible to estimate in a state where various errors included in is reduced / suppressed according to the rotation speed was obtained. As a result, as a result of this action, according to the invention of claim 2, from the estimated value of the rotor magnetic flux of the AC motor having the rotor magnetic flux, the phase of the rotor magnetic flux is affected in a state in which there is no phase deviation and the influence of disturbance. The effect that it can estimate in the suppressed state is acquired.

Next, the effect of the present invention of claim 3 will be described. According to the invention of claim 3, the effect that the action of claim 1 or claim 2 can be obtained with a minimum processing load is obtained. As a result, according to the present invention of claim 3, as a result of this action, the effect of claim 1 or claim 2 can be obtained with a minimum processing load.

Example of one-frequency characteristics of high-speed fast response band filter Example of 1-frequency characteristics when differential processing is applied to a high-speed response band high-order filter 1 is a block diagram of a drive control system including a drive control device according to an embodiment. The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment Configuration example of fast response band high-order filter in one embodiment Configuration example of coefficient generator in one embodiment 1 is a block diagram of a drive control system including a drive control device according to an embodiment. The block diagram which shows the basic composition of the phase determiner in the example of 1 embodiment Configuration example of fast response band high-order filter in one embodiment

Explanation of symbols

DESCRIPTION OF SYMBOLS 1 AC motor 2 Phase determiner 2a Fast response band high-order filter 2a-1 Stator reaction magnetic flux generation block 2b Speed estimator 2c Coefficient generator 2d Phase synchronizer 3 Power converter 4 Current detector 5a Three-phase two-phase converter 5b 2-phase 3-phase converter 6a Vector rotator 6b Vector rotator 7 Cosine sine signal generator 8 Current controller

Claims (3)

A phase estimation method for a rotor of an AC motor in which a rotor rotates in synchronization with a stator power supply frequency,
An equivalent value of the induced voltage or extended induced voltage caused by the rotation of the rotor, which is generated using at least the equivalent value of the stator voltage and current, is a speed response that changes the passband width according to the rotation of the rotor. It is processed with a high-order filter that has a phase lag characteristic of -π / 2 (rad) at a frequency corresponding to the non-zero rotation speed of the band characteristics and a rotor, and an attenuation characteristic of -40 (dB / dec) or higher at high frequencies. Generating a filtered signal mainly including a fundamental wave component having a frequency corresponding to a non-zero rotation speed of the rotor, and generating a rotor phase estimation value from the filtered signal. For rotor phase estimation. A phase estimation method for a rotor magnetic flux of an AC motor having a rotor magnetic flux,
The pass band width is changed according to the rotation of the rotor magnetic flux, with the equivalent value of the induced voltage or the expansion induced voltage caused by the rotation of the rotor magnetic flux generated using at least the equivalent value of the stator voltage and current. High-order filter with fast response band characteristics and phase lag characteristics of -π / 2 (rad) at frequencies corresponding to non-zero rotational speed of rotor flux and attenuation characteristics of -40 (dB / dec) or higher at high frequencies To generate a filtered signal mainly including a fundamental wave component having a frequency corresponding to the non-zero rotation speed of the rotor magnetic flux, and to generate a phase estimation value of the rotor magnetic flux from the filtered signal. The rotor magnetic flux phase estimation method for AC motors. 3. The high-order filter having a quick response band characteristic is a low-pass filter having an attenuation characteristic of −40 (dB / dec) in a high band. Phase estimation method for AC motors.

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