JP5510071B2 - Actuator drive device and translation mechanism, interferometer and spectrometer using the same - Google Patents

Description

  The present invention relates to a drive device for an actuator that performs mechanical resonance drive, and a translation mechanism, an interferometer, and a spectroscope using the same, and more particularly to a drive device that uses a PLL (Phase Locked Loop).

  Conventionally, there is a general case where an actuator including a movable part and a drive part (electromagnetic force generation mechanism such as a PZT piezoelectric vibrator or VCM) coupled thereto is AC driven to obtain its mechanical output. At a specific frequency, a mechanical resonance phenomenon occurs in which a large mechanical output (mechanical vibration amplitude) is obtained in the movable part. FIG. 15 is a graph showing characteristics of the mechanical output (displacement Vm) with respect to the drive frequency fd and the phase θm of the displacement Vm with respect to the drive signal. FIG. 15B is an enlarged view around the mechanical resonance frequency fo of FIG. As apparent from FIG. 15, a large machine output and a sudden phase change appear at the machine resonance frequency fo.

  This is due to the mechanical resonance phenomenon caused by the equivalent mass M and elastic coefficient k of the movable part material viewed from the fixed part of the actuator and the loss coefficient μ caused by internal loss, friction, air friction, etc. of the material, and simple. Is understood. The characteristic of the machine output at the time of the machine resonance is that a simple vibration with a small distortion by a kind of disturbance filtering based on the resonance Q (quality factor) and a large machine vibration amplitude can be obtained. Therefore, various actuators have been devised using such mechanical features.

  Such a drive device for a resonance drive type actuator needs to be driven (drive signal Vd) with an AC signal corresponding to the mechanical resonance frequency fo. Further, since the mechanical resonance frequency fo is assumed to fluctuate due to environmental changes, changes with time, etc., the drive device always uses the frequency of the drive signal Vd (drive frequency fd) during the drive. The function of tracking to fo and obtaining a stable machine output is further required.

  One of the driving methods having such a function is the PLL control described above. This utilizes the fact that the phase of the drive signal Vd and the phase of the machine output Vm change sharply in the vicinity of the resonance frequency fo as shown in FIG. From FIG. 15B, it is understood that if the vibration displacement x is taken as the machine output Vm, the phase θm is delayed from the drive signal Vd by 90 degrees at the resonance frequency fo. At higher frequencies, it will be delayed by a maximum of 180 degrees. Therefore, if the difference of the phase θm between the drive signal Vd and the machine output Vm (displacement x) is detected and the drive frequency fd is controlled so that it is always 90 deg, the machine resonance frequency fo can be tracked. Stable machine output can be obtained. In this respect, the PLL control has a function of performing negative feedback control of the oscillation frequency of a built-in voltage controlled oscillator (VCO) so that the phase difference between the two signals is always maintained at a predetermined value. It can be adapted for use in the drive control of a resonance drive type actuator.

  By the way, in the drive device using the PLL, the VCO normally performs free-run oscillation (oscillation state at an undecided frequency) when the actuator is activated, such as when the power is turned on. In general, the free-run oscillation frequency does not coincide with the mechanical resonance frequency fo of the actuator, and the drive unit drives at a frequency other than the mechanical resonance frequency fo. Therefore, although one drive signal (an AC signal corresponding to the free-run oscillation frequency) is input to the phase comparator, the mechanical output of the other mechanical output Vm (displacement x) detection signal is very small. The value is below the detection level and is not input. For this reason, the phase comparator does not function, the phase is not drawn in the control of the PLL, and the actuator remains in a state where it is not driven by mechanical resonance.

  Therefore, in order to avoid such a state, in Patent Document 1, when the drive device is started, the negative feedback loop of the PLL is opened, and the oscillation frequency control terminal of the VCO is controlled by a sweep circuit. The oscillation frequency fvco is swept toward the machine resonance frequency fo, and at the same time, the machine output Vm (displacement x) is detected. As the oscillation frequency fvco approaches the machine resonance frequency fo, the machine output Vm (displacement x) is detected. When the level rises and reaches a predetermined value, the control circuit determines that the actuator is in a mechanical resonance drive state, and switches the oscillation frequency control terminal of the VCO from the sweep circuit to the output of the phase comparator, thereby The negative feedback loop is closed to achieve phase pull-in (phase lock-in).

JP 2002-78368 A

  However, the above-described measures have the following problems. First, when it is determined that the actuator is in the mechanical resonance drive state and the oscillation frequency control terminal of the VCO is switched from the sweep circuit to the output of the phase comparator, the difference between the output potential of the sweep circuit and the output potential of the phase comparator. As a result, a sudden voltage change occurs at the oscillation frequency control terminal of the voltage controlled oscillator VCO. This is equivalent to closing the negative feedback loop and simultaneously applying a kind of noise to the oscillation frequency control terminal, and the phase pull-in operation of the PLL becomes unstable for a moment. Therefore, depending on the negative feedback loop gain design (gain, phase margin settings, etc.) and loop filter design, overshoot and ringing occur in the change in the oscillation frequency of the VCO. There are cases where time is required and phase lock-in may not be possible.

  Thus, it is considered that the above-described problems can be improved by increasing the detection accuracy of the mechanical resonance driving state, but there are problems that an increase in circuits (peak hold circuit) and adjustment of the detection level are required.

  SUMMARY OF THE INVENTION An object of the present invention is to provide an actuator drive device that can perform a reliable phase pull-in operation in a short time and can suppress an increase in circuit scale and cost, and a parallel movement mechanism and an interferometer using the same. And providing a spectrometer.

  The actuator driving device according to the present invention is a driving device for an actuator that causes a mechanical resonance drive to be performed by an actuator including a movable portion and a driving portion that drives the movable portion. A voltage-controlled oscillator variably generated according to the detection unit, a detection unit that detects a mechanical output of the actuator, a phase comparator that detects a difference between the phase of the drive signal and the detection signal of the detection unit, and the phase comparison A loop filter unit that smoothes the phase comparison signal of the detector and applies the signal to the voltage controlled oscillator as the control signal. The loop filter unit substantially reduces the resonance frequency of the actuator with respect to the integration circuit and the integration circuit. A reference voltage source for providing a reference voltage separated by a predetermined voltage from a target voltage which is the level of the matched control signal, and the integrating circuit Resetting means for resetting at the start of the integration operation and sweeping the level of the control signal from the reference voltage toward the target voltage, thereby bringing the actuator into the mechanical resonance state. And

  According to the above configuration, when driving an actuator including a movable part that is a driven member and a drive part that drives the movable part, the frequency of the drive signal for driving the drive part is determined according to the control signal. A voltage-controlled oscillator that variably generates, a detection unit that detects a mechanical output of the actuator, a phase comparator that detects a difference between the phase of the drive signal and the phase of the detection signal of the detection unit, and the phase comparator And a loop filter unit that smoothes the phase comparison signal and supplies the control signal to the voltage-controlled oscillator as a control signal, thereby forming a PLL loop and driving the mechanical resonance of the actuator. Is a low-pass filter that smoothes the phase comparison signal of the phase comparator and applies the DC voltage as the control signal to the voltage controlled oscillator. In addition to functioning as a filter, it keeps the can sweep the level of the control signal. Specifically, a reference voltage source and a reset unit are provided in an integration circuit that realizes the function as the low-pass filter, and the reference voltage source substantially matches the resonance frequency of the actuator with respect to the integration circuit. A reference voltage separated by a predetermined voltage from the target voltage which is the level of the control signal, that is, the sweep start voltage is given, and the reset means resets the integration circuit at the start of the integration operation, and after releasing the reset, The control signal is swept from the reference voltage toward the target voltage. When the frequency of the drive signal from the voltage controlled oscillator coincides with the resonance frequency that varies for each actuator by the sweep, the phase comparison signal of the phase comparator is also in a predetermined state (predetermined voltage or open state). ) And the movable part actuator can be pulled into the mechanical resonance state.

  Therefore, when the control signal provided outside the PLL loop is swept and when the output of the loop filter is switched and applied to the voltage controlled oscillator, a kind of noise is generated at the time of discontinuous switching, and the PLL In contrast to the unstable operation of the loop, in the present invention, the control signal given to the voltage controlled oscillator is smoothly (continuously) from the reference voltage to the vicinity of the target voltage of the resonance frequency along with the integration operation of the integration circuit. Therefore, there is no occurrence of overshoot or ringing at the time of transition, the oscillation frequency of the voltage controlled oscillator is stabilized, and phase pull-in operation (phase lock-in) can be performed in a short time and surely. Become. Further, without providing a special configuration for sweeping the control signal outside the PLL loop, the control signal is swept only by providing a reference voltage source and a reset means in association with the integration circuit of the loop filter unit, Furthermore, since the sweep stop operation is performed using the phase comparison signal of the phase comparator, it is not necessary to use a dedicated circuit for determining the mechanical resonance driving state, and an increase in circuit scale and cost increase can be suppressed.

  In the actuator driving device of the present invention, the integrating circuit includes a Miller integrating circuit including an operational amplifier, a capacitor that negatively feeds back an output thereof, and an input resistor. It is characterized by comprising a switching element for short-circuiting between terminals.

  According to the above configuration, the integration circuit uses a Miller integration circuit, the output from the phase comparator is connected to the non-inverting input terminal of the operational amplifier via the input resistor, and the output of the operational amplifier is connected to the non-inverting input via the short-circuit switch or capacitor. If the short-circuit switch is switched from “ON” to “OFF” by applying to the inverting input terminal of the operational amplifier, the feedback capacitor can be charged with a constant current by the input / output potential difference and the input resistance. (Ramp signal voltage).

  Therefore, the control voltage for the voltage controlled oscillator, which is the output voltage of the integration circuit, also becomes the ramp signal voltage, and the sweep operation of the oscillation frequency of the voltage controlled oscillator can be performed smoothly and linearly over time, and the sweep speed can be set. It becomes easy. As a result, the machine output can be detected with high accuracy, and the pull-in operation can be performed in a short time.

  Furthermore, in the actuator driving apparatus according to the present invention, the integration circuit is a lag / lead filter.

  According to the above configuration, the lag filter including the RC first-order lag element has an effect of attenuating unnecessary harmonic components in the phase comparison signal from the phase comparator, but has a stabilizing effect in the negative feedback loop. On the other hand, the lag / lead filter has a function of advancing the phase, and has the effect of stabilizing by compensating for the phase delay generated in the negative feedback loop (ensure gain margin and phase margin). Therefore, it is preferable to use this lag / lead filter as a low-pass filter.

  In the parallel movement mechanism of the present invention, the actuator includes a pair of leaf spring members formed in a strip shape as the movable portion, and the opposite surfaces of the leaf spring member are coupled between both ends. A pair of rigid bodies that support the leaf spring members in parallel with each other, and by fixing one end side, the other end side can be translated in the thickness direction of the leaf spring member. One of the leaf spring members is composed of a drive member that drives the bending in the thickness direction thereof, and the movable member is mechanically driven by being given a drive signal from the actuator drive device to the drive member. And

  According to said structure, the parallel displacement mechanism used for a Michelson interferometer etc. is realizable.

  Furthermore, in the parallel movement mechanism of the present invention, the leaf spring member is made of an SOI substrate.

  According to said structure, a parallel displacement mechanism can be manufactured using what is called MEMS technology, ie, the technique which combined semiconductor manufacturing techniques, such as photolithography and an etching, and joining techniques, such as anodic bonding.

  Therefore, variation in the length of the leaf spring member can be suppressed, the inclination of the parallel movement mechanism during assembly and parallel movement can be suppressed, and individual differences can be eliminated.

  The interferometer according to the present invention includes the parallel movement mechanism, and a moving mirror is mounted on the other end side of the leaf spring member.

  According to said structure, installation of a corner cube is unnecessary, an interferometer can be reduced in size, and highly accurate interference can be implement | achieved.

  Furthermore, the spectroscope of the present invention includes the interferometer described above.

  According to said structure, a small and high-resolution spectrometer can be implement | achieved.

  As described above, the actuator driving device according to the present invention drives an actuator including a movable portion serving as a driven member and a driving portion that drives the driven member with a PLL circuit, and performs mechanical resonance driving. In the PLL circuit, the reference signal source and the reset circuit are integrated into the integration circuit that realizes the function as a low-pass filter that smoothes the phase comparison signal of the phase comparator and applies the DC voltage as a control signal to the voltage controlled oscillator. And the reference voltage source applies a predetermined reference voltage that becomes a sweep start voltage to the integration circuit, the reset unit resets the integration circuit at the start of the integration operation, and the control signal Is swept from the reference voltage toward a target voltage corresponding to the resonance frequency of the actuator.

  Therefore, the control signal given to the voltage-controlled oscillator smoothly (continuously) transitions from the reference voltage to the vicinity of the target voltage of the resonance frequency in accordance with the integration operation of the integration circuit. Thus, the oscillation frequency of the voltage controlled oscillator is stabilized, and a reliable phase pull-in operation (phase lock-in) can be performed in a short time. Further, without providing a special configuration for sweeping the control signal outside the PLL loop, the control signal is swept only by providing a reference voltage source and a reset means in association with the integration circuit of the loop filter unit, Furthermore, since the sweep stop operation is performed using the phase comparison signal of the phase comparator, it is not necessary to use a dedicated circuit for determining the mechanical resonance driving state, and an increase in circuit scale and cost increase can be suppressed.

  In the parallel movement mechanism of the present invention, as described above, the movable portion of the actuator is connected between the pair of leaf spring members formed in a strip shape and the opposite surface side of the leaf spring member. And a pair of rigid bodies that support the leaf spring members in parallel with each other, and by fixing one end side thereof, the other end side can be translated in the thickness direction of the leaf spring member. On the other hand, a drive member that drives one of the leaf spring members to bend in the thickness direction is used as the drive unit, and a drive signal from the actuator drive device is applied to the drive member, so that the movable unit is a machine. Resonate drive.

  Therefore, it is possible to realize a translation mechanism used in a Michelson interferometer or the like.

  Moreover, the interferometer of the present invention includes the parallel movement mechanism as described above, and includes a moving mirror mounted on the other end side of the leaf spring member.

  Therefore, it is not necessary to install a corner cube, the interferometer can be miniaturized, and high-precision interference can be realized.

  Furthermore, the spectroscope of the present invention includes the interferometer as described above.

  Therefore, a small and high resolution spectrometer can be realized.

It is a block diagram of the drive device concerning one embodiment of the present invention. It is a graph which shows the change of the oscillation frequency with respect to the control voltage change of PLL VCO. It is a block diagram of the conversion circuit which shows an example of the drive circuit of a mechanical resonance actuator, and converts a rectangular wave into a triangular wave. FIG. 4 is a waveform diagram for explaining the operation of the conversion circuit shown in FIG. 3. It is a wave form diagram which shows the example of the input / output signal waveform of a typical phase comparator. It is a graph for demonstrating the machine output with respect to the change of the drive frequency of a mechanical resonance actuator, and its phase change. It is a block diagram of one structural example of the loop filter part in PLL of the drive device of this Embodiment. It is a wave form diagram for demonstrating operation | movement of the said PLL. It is a figure which shows the schematic structure of the spectrometer using the said drive device. It is a perspective view which shows schematic structure of the said parallel displacement mechanism. It is sectional drawing which shows schematic structure of the said parallel displacement mechanism. It is sectional drawing which shows typically the structure and operation | movement of the said parallel displacement mechanism. It is a block diagram of the loop filter part in the drive device concerning other embodiments of the present invention. It is a graph which shows the structure of a lag filter and a lag lead filter, and the frequency characteristic of a gain and a phase. It is a graph for demonstrating the machine output with respect to the change of the drive frequency of a mechanical resonance actuator, and its phase change.

(Embodiment 1)
FIG. 1 is a block diagram of a driving apparatus 1 according to an embodiment of the present invention. The drive device 1 mechanically drives an actuator 2 including a movable part that is a driven member and a drive part that drives the driven part. In general, the drive circuit 1 is connected to the PLL circuit 3 from the PLL circuit 3. Is converted into a waveform and amplitude suitable for driving the actuator 2 and output as a drive signal Vd for obtaining an expected machine output Vm, and a machine output Vm (displacement) of the movable part. and a detection unit 5 that obtains a detection signal Vs by detecting x).

  The main components of the PLL circuit 3 are similar to the configuration of the existing PLL circuit, and include a VCO 11, a frequency divider 12, a phase comparator 13, and a loop filter unit 14, and constitute a negative feedback loop. To do. However, in the present embodiment, the frequency fd of the drive signal Vd for driving the actuator 2, that is, the mechanical resonance frequency fo is relatively low, and the oscillation signal Vvco from the VCO 11 has the frequency fvco at the frequency divider 12. The oscillation signal Vn divided by 1 / N is supplied to the drive circuit 4 and is input to the phase comparator 13. Therefore, depending on the frequency (mechanical resonance frequency fo) for driving the actuator 2, the oscillation signal Vvco from the VCO 11 may be directly supplied to the drive circuit 4 as the oscillation signal Vn without providing the frequency divider 12. Alternatively, the frequency divider 12 may be provided on the input side of the phase comparator 13.

  The VCO 11 is a signal generator that can vary the oscillation frequency fvco in accordance with the control voltage Vc applied to the control terminal Vcnt. FIG. 2 shows the relationship between the control voltage Vc and the oscillation frequency fvco. Thus, in the VCO 11, the direct-current control voltage Vc and the oscillation frequency fvco are in a monotonically increasing relationship.

  In the drive circuit 4, the input signal (oscillation signal Vn) is normally a rectangular wave, but if the adverse effect due to the response at the mechanical resonance frequency fo higher than the mechanical resonance frequency fo of the actuator 2 is taken into account, the harmonics can be obtained. Since the drive signal fd having a small wave component is desirable, the input signal (oscillation signal Vn) is converted into a triangular wave or sine wave waveform to amplify the power. FIG. 3 shows an example of a conversion circuit 4a to a triangular wave using a Miller integration circuit as the drive circuit 4. The conversion circuit 4 a is configured by including an input resistor 42 and a feedback capacitor 43 in an inverting amplifier 41. Here, a rectangular wave as shown in FIG. 4A is input as the input signal (oscillation signal Vn), and the direction of the current It flowing through the input resistor 42 changes at the rising and falling timings of the oscillation signal Vn. Then, the inverting amplifier 41 charges and discharges the current It with a constant current with respect to the feedback capacitor 43, and a triangular wave Vo can be obtained at the output (FIG. 4C). Based on the triangular wave Vo, the amplitude is appropriately changed to be a drive signal Vd. By the conversion, the phase of the fundamental wave component Vdf (FIG. 4D) of the drive signal Vd is advanced by 90 deg phase with respect to the phase of the fundamental wave component Vnf of the oscillation signal Vn (FIG. 4B). .

  The actuator 2 outputs a machine output Vm corresponding to the drive signal Vd created as described above. The detection unit 5 detects the mechanical output Vm using a piezoelectric vibration sensor or the like coupled in the vicinity of the movable unit of the actuator 2, performs waveform shaping after amplification, and outputs a rectangular wave signal (detection signal Vs, main signal) In the example, the displacement x is detected).

  The phase comparator 13 detects a difference between the phase of the oscillation signal Vn and the phase of the detection signal Vs and outputs a phase comparison signal Vpd corresponding to the phase difference. FIG. 5 shows an example of input / output signal waveforms of a typical phase comparator (in a No. 4046 PLL IC). There are three types of phase difference Δθ between the oscillation signal Vn and the detection signal Vs. That is, when the phase of the detection signal Vs is advanced from the phase of the oscillation signal Vn ((A) in the figure), the phase comparison signal Vpd outputs the power supply voltage Vcc value. When the phase of the detection signal Vs coincides with the phase of the oscillation signal Vn ((B) in the figure), the output circuit of the phase comparator 13 is in a high impedance state. Furthermore, when the phase of the detection signal Vs is delayed from the phase of the oscillation signal Vn ((C) in the figure), the phase comparison signal Vpd outputs the ground potential value (0 V). In addition, even when only the oscillation signal Vn is input and the detection signal Vs is not input (“H” or “L” signal is input and does not change), the circuit configuration outputs the ground potential value (0 V). ((D) in the figure).

  The loop filter unit 14 removes unnecessary high-frequency components included in the phase comparison signal Vpd (FIG. 5) (LPF function) and outputs a signal (control voltage Vc) of a change component of the phase difference Δθ to the VCO 11. To the control terminal Vcnt. The loop filter unit 2 also has a phase compensation function for ensuring the feedback amount setting and stability (gain margin, phase margin) of the PLL loop described below.

  Here, as shown in FIG. 6, if the mechanical resonance frequency fo of the actuator 2 in the mechanical resonance driving state (mechanical resonance frequency fo1, driving frequency fd1) is changed by the environmental temperature or the like (fo1 → fo2), the actuator Due to the mechanical resonance characteristic of 2, the phase delay of the mechanical output Vm (displacement x) with respect to the drive signal Vd (drive frequency fd1) is smaller than 90 deg (θm1 in the figure). Therefore, the phase of the detection signal Vs advances from the phase difference of 0 deg from the phase of the oscillation signal Vn (Δθm in the figure). As a result, the phase comparison signal Vpd moves from the state shown in FIG. 5B to the state shown in FIG. 5A, and the phase comparator 13 periodically outputs the voltage Vcc to the loop filter unit 14. . As a result, the output (control signal Vc) of the loop filter unit 14 increases, and the oscillation frequency fvco of the VCO 11 also increases. When the oscillation frequency fvco increases, the drive frequency fd1 of the drive signal Vn that is the frequency-divided signal changes to fd2. Then, the phase of the detection signal Vs returns to 90 deg and returns to the stable state again (negative feedback operation). It can be understood that a series of these operations follows the fluctuation of the mechanical resonance frequency fo of the actuator 2.

  It should be noted that in the present embodiment, the loop filter unit 14 has a frequency sweep function of the VCO 11 in addition to the LPF function as described above. Specifically, FIG. 7 is a block diagram of a loop filter unit 14a, which is a configuration example of the loop filter unit 14. The loop filter unit 14a includes an integration circuit 21 that functions as the LPF, and a target voltage Vct that is a level of a control voltage Vc that substantially matches the mechanical resonance frequency fo of the actuator 2 with respect to the integration circuit 21. A reference voltage Vref separated by a predetermined voltage ΔVc from the reference voltage source 22, that is, a reference voltage source 22 for providing a sweep start voltage, which will be described later, and the integration circuit 21 are reset at the start of the integration operation, and after the reset is released, the control voltage Vc A short-circuit switch 23 that pulls the actuator 2 into the mechanical resonance state by sweeping the level from the reference voltage Vref toward the target voltage Vct.

  The integration circuit 21 includes a Miller integration circuit including an amplifier 24, an input resistor R, and a feedback capacitor C. The output of the phase comparator 13 (phase comparison signal Vpd) is connected to the negative input terminal of the amplifier 24 via a resistor R. The amplifier 24 is an amplifier having a differential input configuration, and its output Va is fed back to the negative input terminal by a feedback capacitor C, and the reference voltage Vref is supplied from the reference voltage source 22 to the positive input terminal, Become. The output Va of the amplifier 24 is inverted by the subsequent amplifier 25 to become the control voltage Vc of the VCO 11. The reason for the inversion is to return to the positive phase because the phase is inverted by the Miller integrating circuit.

  The integration circuit 21 functions as an LPF in a phase lock-in state, which will be described later, and removes unnecessary high frequency components as described above, as well as setting the feedback amount of the negative feedback loop and stability (gain margin, Phase compensation is performed to ensure a phase margin. As a reset means, the short-circuit switch 23 is provided between the terminals of the feedback capacitor C. By setting this to “ON”, the charge of the feedback capacitor C is discharged (reset), It is possible to set integration operation standby.

  FIG. 8 is a waveform diagram for explaining the operation of the driving apparatus 1 configured as described above. First, in the waiting period W1, the short-circuit switch 23 is turned on to wait for the integration operation. At this time, the output Va of the integrating circuit 21 holds the same voltage as the reference voltage Vref. At this time, the offset is adjusted so that the control voltage Vc from the amplifier 25 becomes a predetermined start voltage Vc1. As a result, an oscillation signal having a frequency fvco1 corresponding to the start voltage Vc1 is output from the VCO 11, and an oscillation signal Vn whose frequency is multiplied by 1 / N by the frequency divider 12 is sent from the drive circuit 4 to the actuator 2. A drive signal Vd having a frequency fd = fvco1 / N is applied. The reference voltage Vref and the start voltage Vc1 are voltages for setting the oscillation frequency fvco of the VCO 11 to a predetermined start frequency fvco1, and a target frequency ft = fo set from the mechanical resonance frequency fo and the frequency division ratio N Any voltage may be used as long as it can oscillate at a frequency sufficiently away from × N (which may be high or low, but is shown on the high frequency side in FIG. 8).

  Therefore, in the period W1 in the standby state, the actuator 2 is not in the mechanical resonance driving state, so the input of the phase comparator 13 is not input because the detection signal Vs from the detection unit 5 is below the detection level, Only the oscillation signal Vn from the frequency divider 12 is provided, and the phase comparator 13 outputs “L (= 0 V)” to the phase comparison signal Vpd. Therefore, a current I (= Vref / R) flows through the input resistor R of the integrating circuit 21 from the output terminal of the amplifier 24 via the short-circuit switch 23 (see FIG. 7).

  Next, in the period W2 of the start-up state, the short-circuit switch 23 is switched from “ON” to “OFF”, whereby the current I passes from the short-circuit switch 23 (I1 in FIG. 7) to the feedback capacitor C. (I2 in the figure), and the feedback capacitor C flows as a charging current. Therefore, the output Va of the amplifier 24 indicates a ramp signal voltage (Va = I / C · t + Vref, t: elapsed time) due to constant current charging of the feedback capacitor C. At the same time, the control voltage Vc decreases with time from the start voltage Vc1, so the oscillation frequency fvco of the VCO 11 also decreases. Thus, the frequency of the oscillation signal Vn (drive signal Vd) approaches the mechanical resonance frequency fo while performing the sweep operation. The ramp signal voltage slope I / C corresponds to the sweep speed of the oscillation frequency fvco of the VCO 11, but this magnitude is obtained when the frequency fd = fvco / N of the drive signal Vd to be swept crosses the mechanical resonance frequency fo. It is assumed that the detection unit 5 is set to a speed at which the machine output Vm grows and can be sufficiently detected.

  Subsequently, in the transient period W3, the sweep operation of the drive frequency fd = fvco / N proceeds, approaches the mechanical resonance frequency fo of the actuator 2, and the mechanical output Vm gradually grows. Then, when the period W4 of the phase pull-in operation (phase lock-in) is reached, the growth of the mechanical output Vm exceeds the detection level of the detection unit 5, and the detection signal Vs whose waveform has been shaped is input to the phase comparator 13. The phase comparator 13 performs a phase comparison operation between the detection signal Vs and the oscillation signal Vn, and starts outputting the result. That is, the phase comparison signal Vpd of the phase comparator 13 shifts from the “L” output state to the repetitive pulse output state (“L” ⇔ “high impedance”). The following operations start with this transition.

  That is, first, a change occurs in the input (phase comparison signal Vpd) of the integrating circuit 21, and the charging current I becomes 0 at high impedance, so the average value of the charging current I decreases. As a result, the rate of decrease of the control voltage Vc decreases, and the sweep rate of the drive frequency fd = fvco / N also decreases. The drive frequency fd = fvco / N gradually approaches the mechanical resonance frequency fo, the phase difference between the detection signal Vs and the divided signal Vn is reduced, and the high impedance period of the phase comparison signal Vpd is increased. By repeating the above operation, the drive frequency fd = fvco / N coincides with the mechanical resonance frequency fo of the actuator 2 at that time, and the phase is drawn (locked in).

  Finally, in the lock state period W5, the phase difference between the detection signal Vs and the frequency-divided signal Vn becomes 0 deg (the phase difference between the drive signal Vd and the detection signal Vs is 90 deg). As a result, the phase comparison signal Vpd shifts to the “high impedance” state, the charging current I also becomes 0, and the output Va of the integrating circuit 21 is a constant value Va2 (the control voltage Vc from the amplifier 25) that is the target voltage Vct. Holds Vc2). That is, it can be said that the operation of generating the sweep signal by the integration circuit 21 is stopped by the phase comparison signal Vpd. Therefore, the negative feedback loop is stabilized, and the drive frequency fd = fvco / N substantially coincides with the mechanical resonance frequency fo of the actuator 2, and the phase lock-in operation is completed.

  Therefore, when the control signal provided outside the PLL loop is swept as in the prior art and when the output of the loop filter is switched and applied to the VCO, a kind of noise is generated at the time of discontinuous switching. While the operation of the PLL loop becomes unstable, in the present embodiment, the control voltage Vc (Va) applied to the VCO 11 is changed from the reference voltage Vref to the target of the resonance frequency fo along with the integration operation of the integration circuit 21. Since the transition to the voltage Vct (= Vct2 (Vat2)) is smooth (continuously), no overshoot or ringing occurs at the transition, the oscillation frequency fvco of the VCO 11 is stable, and the phase is short and reliable. A pull-in operation (phase lock-in) can be performed. Further, a reference voltage source 22 and a short-circuit switch 23 as a reset means are provided in association with the integrating circuit 21 of the loop filter unit 14 without providing a special configuration for sweeping the control signal outside the PLL loop. Only the control voltage Vc (Va) is swept, and further, the sweep stop operation is performed using the phase comparison signal Vpd of the phase comparator 13, so there is no need to use a dedicated circuit for determining the mechanical resonance driving state. Increase in circuit scale and cost increase can be suppressed.

  Further, by using a Miller integrating circuit for the integrating circuit 21, the feedback capacitor C can be charged with a constant current (current value I = (Vref−Vpd) / R, in this embodiment, Vpd = 0V). The output Va of the circuit 21 changes linearly (ramp signal voltage). Therefore, the control voltage Vc is also a ramp signal voltage, and as shown in FIG. 2, the sweep operation of the oscillation frequency fvco of the VCO 11 can be performed smoothly and linearly with time, so that the sweep speed can be easily set. This facilitates the setting of the sweep speed during the start-up period W2, so that the mechanical output Vm can be accurately detected during the phase pull-in operation period W4, and the pull-in operation can be performed in a short time.

  Although there are various configurations of the phase comparator 13 other than the present example, the phase comparison signal Vpd when only one signal (oscillation signal Vn) is input and two signals are input. Any phase comparator may be used as long as it has a signal difference (potential difference) with the phase comparison signal Vpd when the PLL control is locked to the target phase difference. In this example, since the detection unit 5 detects the displacement of the machine output Vm, the phase difference between the drive signal Vd (fd = fo) and the detection signal Vs during the mechanical resonance drive is set to 90 deg. On the other hand, when the detection unit 5 is configured to detect the speed of the mechanical output Vm, the phase difference is 0 deg. Therefore, the lock phase of the PLL control may be set to the 0 deg.

  FIG. 9 is a diagram showing a schematic configuration of the spectroscope 101 using the driving device 1 configured as described above. The driving device 1 is used as an actuator 2 for mechanical resonance driving of a later-described parallel movement mechanism 121 of the spectroscope 101. In addition, the parallel movement mechanism 121 of the present embodiment can be applied to a field where high-accuracy translational driving is required. In addition to an interferometer and a spectroscope, a refractive index measuring device, an objective lens actuator of an optical pickup, a small camera It can also be applied to an AF (autofocus) mechanism.

  The spectroscope 101 is a temporal Fourier transform spectroscope using a Michelson interferometer as the interferometer 102, and includes a calculation unit 103 and an output unit 104. The interferometer 102 includes a light source 111, a collimator lens 112, a beam splitter (for example, a half mirror) 113, a fixed mirror 114, a moving mirror 115, a condenser lens 116, and a detector 117. The moving mirror 115 constitutes a reflection part that reflects incident light while moving, and constitutes a part of the translation mechanism 121. Details of the translation mechanism 121 will be described later.

  In the above configuration, the light emitted from the light source 111 is converted into parallel light by the collimator lens 112 and then branched into two optical paths by the beam splitter 113. Each light beam is reflected by the fixed mirror 114 and the moving mirror 115, returned to the original optical path, overlapped by the beam splitter 113, and irradiated to the sample S as interference light. At this time, the sample S is irradiated with light while moving the moving mirror 115 continuously. When the optical paths from the beam splitter 113 to each mirror (fixed mirror 114 and moving mirror 115) are equal to each other, they are superimposed. The intensity of the emitted light is maximized. On the other hand, when there is a difference between the two optical paths due to the movement of the movable mirror 115, the intensity of the superimposed light changes. The light transmitted through the sample S enters the detector 117 through the condenser lens 116, and is detected as a temporal interferogram there.

  The signal output from the detector 117 of the interferometer 102 is A / D-converted and Fourier-transformed by the calculation unit 103, and as a result, a spectrum is generated. This spectrum is displayed and output by the output unit 4. Since the intensity of light of each wavelength (wave number (= 1 / wavelength)) can be known from the output spectrum, the characteristics (material, structure, component amount, etc.) of the sample S can be known from this.

  FIG. 10 is a perspective view showing a schematic configuration of the parallel movement mechanism 121, and FIG. 11 is a sectional view thereof. The parallel movement mechanism 121 includes a pair of leaf spring members 131 and 132 formed in a band shape, and the opposite ends of the leaf spring members 131 and 132, by connecting the two end portions to each other. A pair of rigid bodies 133 and 134 that support 132 in parallel with each other, a driving member 135 that drives one of the leaf spring members (131 in FIGS. 10 and 11) in the thickness direction as described later, and the above-described And a moving mirror 115.

  The leaf spring members 131 and 132 are formed using, for example, an SOI (Silicon on Insulator) substrate, and support layers 131a and 132a made of silicon, insulating oxide film layers (BOX layers) 131b and 132b made of silicon oxide, After the active layers 131c and 132c made of silicon are laminated, in the support layers 131a and 132a and the insulating oxide film layers (BOX layers) 131b and 132b, both end portions corresponding to the rigid bodies 133 and 134 are left, and the intermediate portion Is removed and formed using techniques such as photolithography and dry etching. The plate spring members 131 and 132 are opposed to the surfaces on which the support layers 131a and 132a and the insulating oxide film layers (BOX layers) 131b and 132b are formed, so that the support layers 131a and 132a and the rigid bodies 133 and 134 Are joined to form flat and parallel flat plate portions 131p and 132p.

The rigid bodies 133 and 134 are made of glass that is thicker than the flat plate portions 131p and 132p of the leaf spring portions 131 and 132. As the glass, alkali glass containing sodium oxide (Na ₂ O) or potassium oxide (K ₂ O) can be used. In this case, as described above, the support layers 131a and 132a made of silicon and the rigid bodies 133 and 134 can be connected by anodic bonding. The anodic bonding is a technique in which a direct voltage of several hundred volts is applied to silicon and glass at a temperature of several hundred degrees Celsius to form Si—O covalent bonds, thereby directly bonding the two.

  One of the rigid bodies 133 and 134 (134 in FIGS. 10 and 11) is fixed to the inner wall of the interferometer 102, and the other (133) is deformed by the deformation of the leaf spring member 131 by the drive member 135. It can be displaced. The drive member 135 bends and deforms the leaf spring member 131 to cause the rigid body 134 to be fixed to the rigid body 133 by the two leaf spring members 131 and 132. It is to be translated in the opposite direction. In the present embodiment, the drive member 135 is provided above the rigid body 134 of the leaf spring member 131 and on the surface opposite to the rigid body 134. The movable mirror 115 is provided above the rigid body 133 in the leaf spring member 131 and on the surface opposite to the rigid body 133. The driving member 135 and the moving mirror 115 may be provided on the leaf spring member 132 side. The sizes of the driving member 135 and the moving mirror 115 may be set as appropriate.

  The moving mirror 115 is formed, for example, by sputtering Au on the active layer 131c made of silicon. Alternatively, the movable mirror 115 may be formed by forming a metal material such as Al or Pt on the active layer 131c by vapor deposition or adhesion.

  The driving member 135 is composed of a PZT element or the like. As shown in FIG. 12, the PZT element has a structure in which electrodes 142 and 143 are attached to both surfaces of a piezoelectric material PZT (lead zirconate titanate) 141. The drive member 135 is bonded to the leaf spring member 131 using an adhesive. In the leaf spring member 131, an extraction electrode 153 and a fixed electrode 154 are formed in advance by sputtering of a metal material, and the upper electrode 142 is connected to the lower electrode 143 through the fixed electrode 154. The electrode pads from the drive circuit 4 are connected via wire bonding.

  By applying a voltage between the electrodes 142 and 143, the PZT 41 expands and contracts in the longitudinal direction, bends and deforms the leaf spring member 131, and can displace the moving mirror 115 together with the rigid body 133. . For example, when the PZT 141 is extended by applying a positive voltage between the electrodes 142 and 143, as shown in FIG. 12A, the leaf spring member 131 is deformed to be convex upward, The movable mirror 115 is displaced downward together with the rigid body 133. On the other hand, when the PZT 41 is contracted by applying a negative voltage between the electrodes 142 and 143, the leaf spring member 131 is deformed so as to protrude downward as shown in FIG. The movable mirror 115 is displaced upward together with the rigid body 133. In this way, the leaf spring member 131 is bent and deformed by the drive member 135, whereby the movable mirror 115 is translated together with the rigid body 133 with respect to the fixed rigid body 134 as shown in FIG. The mechanism 121 can be realized.

  Thus, by configuring the leaf spring members 131 and 132 using the SOI substrate, as described above, so-called MEMS (Micro Electro Mechanical Systems) technology, that is, semiconductor manufacturing technology such as photolithography and etching, and anodic bonding The translation mechanism 121 can be manufactured using a technique that combines a joining technique such as the above. Therefore, as long as the lithography mask accuracy is ensured with high accuracy, it is possible to avoid variations in the lengths of the two flat plate portions 131p and 132p in one translation mechanism 121. As a result, it is possible to suppress the inclination of the movable portion (the rigid body 133 and the movable mirror 115) when the parallel movement mechanism 121 is assembled or translated. In addition, since it is possible to avoid individual differences, that is, to prevent the length of the flat plate portions 131p and 132p from varying for each individual of the plurality of parallel movement mechanisms 121, it is possible to stably manufacture the plurality of parallel movement mechanisms 121.

Moreover, the thermal expansion coefficient of silicon is about 3 × 10 ⁻⁶ (/ K), and the thermal expansion coefficient of alkali glass is about 3.2 × 10 ⁻⁶ (/ K). Since the support layers 131a and 132a made of silicon and the rigid bodies 133 and 134 made of glass are connected, the two members connected to each other have close thermal expansion coefficients, so that deformation of the parallel movement mechanism 121 due to temperature change is suppressed. be able to. Thereby, it can suppress that a movable part (the rigid body 133 and the movable mirror 115) inclines due to a temperature change.

  Further, the plate spring members 131 and 132 and the rigid bodies 133 and 134 are connected by bonding (in the above example, anodic bonding). That is, both of the above are directly joined without using an adhesive. As a result, manufacturing errors (inclinations and positional deviations of the rigid bodies 133 and 134 due to shrinkage of the adhesive) that occur when the adhesive is used can be reliably eliminated, and a highly accurate parallel movement mechanism can be realized.

  Further, since the movable mirror 115 is provided on one of the leaf spring members 131 and 132 (the leaf spring member 131 in this embodiment), the movable mirror 115 is moved together with the rigid body 133 by the bending deformation of the leaf spring member 131 by the drive member 135. Can be translated. Therefore, the parallel movement mechanism 121 of this embodiment can be easily applied to the interferometer 2 shown in FIG. 9, that is, the Michelson interferometer.

  In addition, since the interferometer 102 of the present embodiment includes the parallel movement mechanism 121 described above, it is possible to reduce the size of the interferometer 102 without requiring the installation of a corner cube, and to realize highly accurate interference. it can. And since the spectroscope 101 of this embodiment is provided with the interferometer 102 mentioned above, a small-sized and high-resolution spectroscope can be realized.

(Embodiment 2)
FIG. 13 is a block diagram of a loop filter unit 14b in a driving apparatus according to another embodiment of the present invention. The loop filter unit 14b is similar to the loop filter unit 14a described above, and corresponding portions are denoted by the same reference numerals, and description thereof is omitted. The loop filter unit 14a described above realizes the LPF function with a lag filter including a first-order lag element of RC, but it should be noted that this loop filter unit 14b is realized with a lag / lead filter. Is.

  FIG. 14A is a graph showing the configuration of the lag filter and the frequency characteristics of gain and phase, and FIG. 14B is a graph showing the configuration of the lag-lead filter and the frequency characteristics of gain and phase. As shown in FIG. 14A, the lag filter comprising the first-order lag element of RC has an effect of attenuating unnecessary harmonic components in the phase comparison signal Vpd from the phase comparator 13, but a negative feedback loop. In contrast, the lag and reed filter has a function of advancing the phase as shown in FIG. 14 (b), and compensates for the phase lag generated in the negative feedback loop. Therefore, it is preferable to use this lag / reed filter as a low-pass filter.

  Therefore, in the loop filter unit 14b of the present embodiment, a resistor Rc (r2 in FIG. 14B) is inserted in series with the feedback capacitor C (c1 in FIG. 14B), so that a lag / lead filter is obtained. Is realized. Thereby, the gain margin and the phase margin can be ensured.

  In the loop filter units 14a and 14b, the integration circuit 21 performs the integration operation by accumulating charges in the feedback capacitor C, but may perform the integration operation by extracting the charges. In that case, the resetting means is a means for giving an initial charge. The short-circuit switch 23 as the reset means is not limited to a single transistor, and may be configured by providing a multistage switch element with its drive circuit.

DESCRIPTION OF SYMBOLS 1 Drive apparatus 2 Actuator 3 PLL circuit 4 Drive circuit 4a Conversion circuit 41 Inverting amplifier 42 Input resistance 43 Feedback capacitor 5 Detection part 11 VCO
12 frequency divider 13 phase comparator 14, 14a, 14b loop filter unit 21 integrating circuit 22 reference voltage source 23 short-circuit switch 24 amplifier 101 spectroscope 102 interferometer 103 calculation unit 104 output unit 111 light source 112 collimator lens 113 beam splitter 114 fixed Mirror 115 Moving mirror 116 Condenser lens 117 Detector 121 Parallel movement mechanism 131, 132 Leaf spring member 133, 134 Rigid body 135 Drive member 131a, 132a Support layer 131b, 132b Insulating oxide film layer 131c, 132c Active layer 131p, 132p Flat plate portion C Feedback capacitor R Input resistance Rc Resistance S Sample

Claims (7)

A voltage controlled oscillator that variably generates a frequency of a drive signal for driving the drive unit according to a control signal in an actuator drive device that causes a mechanical resonance drive to be performed by an actuator including a movable unit and a drive unit that drives the movable unit A detection unit that detects a mechanical output of the actuator, a phase comparator that detects a difference between a phase of the drive signal and a detection signal of the detection unit, and a phase comparison signal of the phase comparator is smoothed A loop filter unit that supplies the voltage-controlled oscillator as the control signal.
The loop filter unit is
An integration circuit;
A reference voltage source that provides a reference voltage that is a predetermined voltage apart from a target voltage that is a level of a control signal that substantially matches the resonance frequency of the actuator;
Resetting means for pulling the actuator into the mechanical resonance state by resetting the integrating circuit at the start of the integrating operation, and sweeping the level of the control signal from the reference voltage toward the target voltage; An actuator driving device comprising:   The integration circuit comprises a Miller integration circuit comprising an operational amplifier, a capacitor that negatively feeds back its output, and an input resistor, and the reset means comprises a switch element that short-circuits between the terminals of the capacitor. The actuator driving device according to claim 1.   2. The actuator driving apparatus according to claim 1, wherein the integrating circuit is a lag / lead filter. The actuator is
As the movable part, a pair of leaf spring members formed in a strip shape, and a pair of rigid bodies that support the leaf spring members in parallel with each other by connecting the opposite ends on the opposite surface side of the leaf spring member, The one end side is fixed, and the other end side can be translated in the thickness direction of the leaf spring member.
The drive unit includes a drive member that drives one of the leaf spring members to bend in the thickness direction thereof.
A parallel movement mechanism, wherein the movable member is mechanically driven by being given a drive signal from the actuator drive device according to any one of claims 1 to 3 to the drive member.   The parallel movement mechanism according to claim 4, wherein the leaf spring member is made of an SOI substrate.   An interferometer comprising the parallel movement mechanism according to claim 4 or 5, wherein a moving mirror is mounted on the other end side of the leaf spring member.   A spectrometer comprising the interferometer according to claim 6.

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