JP5511531B2 - Control device for synchronous motor - Google Patents

Description

  The present invention relates to a control device for driving a synchronous motor, and more particularly to a control device for a synchronous motor that stably performs drive control without using a position sensor or the like.

  2. Description of the Related Art Conventionally, as a supercharging system for an internal combustion engine, a system is known in which a compressor is connected in series or in parallel with a turbocharger, and supercharging is performed while assisting driving of the compressor with an AC motor. This supercharging assist AC motor is connected to a storage battery as a drive source via a power conversion circuit. Further, the control device used in this case drives the AC motor to assist supercharging when the vehicle accelerates by controlling the power conversion circuit, and stops assist during other constant speed running.

  In addition, since the on-vehicle AC motor uses a storage battery as a drive source, it is desired to reduce current consumption as much as possible. Even in the supercharging assist system as described above, when acceleration / deceleration work is not performed It is necessary to control so as not to consume as much current as possible.

  By the way, there are various types of AC motors such as field winding type synchronous motors and permanent magnet synchronous motors, and induction motors, but synchronous motors are used when assisting supercharging of an internal combustion engine with an electric motor. The Also, when controlling such a synchronous motor, it is possible to perform vector control using a position sensor, but the position sensor is expensive and has problems such as reliability due to failure. A sensorless control without a sensor is desired.

  Here, for example, when driving a synchronous motor to assist supercharging when the vehicle accelerates, the power of the storage battery is supplied to the AC motor, but at that time, if an overcurrent flows through the power conversion circuit or the synchronous motor, a failure occurs. Cause. Therefore, conventionally, a protection circuit such as a fuse is provided between the storage battery and the power conversion circuit, or when the overcurrent flows through the power conversion circuit, the operation of the power conversion circuit is stopped.

  However, if the control for stopping the operation of the power conversion circuit when an overcurrent flows through the power conversion circuit is simply performed, for example, there is a problem that supercharging cannot be assisted when it is desired to accelerate the vehicle. Therefore, in the conventional technology, control is performed to correct the primary frequency command value so as to prevent overcurrent of the power conversion circuit and the motor, and to perform overcurrent protection when performing acceleration / deceleration operations that require the maximum current. An apparatus has been proposed (see, for example, Patent Document 1 below).

  On the other hand, when assisting supercharging of the internal combustion engine with a synchronous motor as described above, various load disturbances affect the operation of the synchronous motor. Examples of such load disturbance include disturbance torque such as a change in storage battery voltage and deterioration of internal resistance due to aging, or mechanical vibration transmitted from a device such as a turbine connected to a synchronous motor. In particular, when a supercharging compressor is connected to the synchronous motor, the turbine connected to the compressor is driven by the exhaust gas of the internal combustion engine, so the output fluctuation of the exhaust gas becomes the disturbance torque as it is.

  In the sensorless control device, if the power conversion circuit is controlled only in accordance with the speed command, the acceleration / deceleration operation or the like of the vehicle is not performed, and the torque is affected by the influence of the load disturbance even when the synchronous motor does not need to assist the compressor drive As a result, there arises a problem that unnecessary current flows through the synchronous motor and the power of the storage battery is wasted.

  In addition, in the prior art, in order to suppress torque pulsation, there has been proposed one in which current control is performed using a deviation between the speed command and the rotation speed of the electric motor. That is, in Patent Document 2, torque control is performed based on torque compensation characteristics calculated according to the electrical angle of the motor and the like, and torque ripple generated from a load or the like is reduced at high speed using a table or the like ( For example, see Patent Document 2 below).

JP-A-5-68398 Japanese Patent Laid-Open No. 2005-224051

  However, in the conventional technique described in Patent Document 1 described above, the primary frequency command value is corrected so as to prevent overcurrent of the power conversion circuit and the electric motor. Therefore, the acceleration / deceleration operation that requires the maximum current is performed. Although overcurrent protection is possible, it does not take into account current control under conditions in which acceleration / deceleration operations are not performed, such as the idling state of an internal combustion engine of a vehicle. Therefore, when the configuration as in Patent Document 1 is used for a supercharging compressor, it is difficult to suppress unnecessary current when it is affected by a load disturbance during operation without acceleration / deceleration. That is, unlike the overcurrent protection control, the control for suppressing the current due to the load disturbance cannot be stably controlled with the configuration as in Patent Document 1.

  Further, in the conventional technique described in Patent Document 2 described above, current control is performed using a deviation between the speed command and the rotation speed of the electric motor in order to suppress torque pulsation, but the constant speed without performing the acceleration / deceleration operation. The effect of reducing the extra current consumption due to the influence of the load disturbance cannot be exhibited sufficiently during traveling.

  The present invention has been made in order to solve the above-described problems, and in the case where the synchronous motor is driven and controlled while being affected by the load disturbance, the current flowing through the synchronous motor is optimally controlled. While avoiding interruption of the power conversion circuit while protecting the conversion circuit from overcurrent, it is possible to reduce unnecessary power consumption by suppressing unnecessary current from flowing to the synchronous motor due to the influence of load disturbance, etc., and It is an object of the present invention to provide a control apparatus for a synchronous motor that can stably control driving without using a position sensor or the like.

A control apparatus for a synchronous motor according to the present invention includes a power conversion circuit that converts DC power supplied from a DC power source into AC power of variable voltage and variable frequency and outputs the power, and a primary supplied from the power conversion circuit to the synchronous motor. A current detector for detecting a phase current, a current component calculation circuit for calculating a current component based on the primary phase current and rotation angle information detected by the current detector, and a current setting used for controlling the synchronous motor A current setting circuit for setting a value , a current setting value set by the current setting circuit so as to flow a current of the current setting value during acceleration / deceleration operation, and to suppress a current when the acceleration / deceleration operation is not performed; A frequency correction calculation circuit for obtaining a frequency correction amount from the current component calculated by the current component calculation circuit, and the frequency correction calculation circuit calculated from the primary frequency command value by the frequency correction calculation circuit. A subtraction circuit that subtracts the wave number correction amount to obtain a primary frequency, an integration circuit that integrates the primary frequency obtained by the subtraction circuit and converts it into the rotation angle information, a primary frequency obtained by the subtraction circuit, and A voltage command calculation circuit that calculates a primary voltage command value based on the rotation angle information obtained by the integration circuit and outputs the primary voltage command value to the power conversion circuit.

  According to the present invention, it is possible to always set and control an optimal current value for a state in which the current of the synchronous motor is likely to fluctuate due to a load disturbance caused by fluctuations in the exhaust gas or the like of the internal combustion engine. As a result, wasteful current consumption of the DC power source such as a storage battery can be suppressed, and the power conversion circuit and the synchronous motor are protected from overcurrent, and the power conversion circuit is stopped by the overcurrent, or the synchronous motor is controlled. Can be prevented from becoming unstable. Further, since the synchronous motor can be controlled stably without using a position sensor or the like, the control circuit can be configured at low cost.

It is a block diagram which shows the structure of the control apparatus of the synchronous motor in Embodiment 1 of this invention. It is a block diagram which shows the detailed structure of the electric current component calculating circuit of the control apparatus. It is a block diagram which shows the detailed structure of the frequency correction calculating circuit of the same control apparatus. It is a wave form diagram which shows the relationship between the frequency command value in this invention, and a primary frequency. It is a block diagram which shows the structure of the control apparatus of the synchronous motor in Embodiment 2 of this invention. It is a block diagram which shows the detailed structure of the electric current component calculating circuit of the control apparatus. It is a block diagram which shows the detailed structure of the frequency correction calculating circuit of the same control apparatus. It is a block diagram which shows the structure of the control apparatus of the synchronous motor in Embodiment 3 of this invention.

Embodiment 1 FIG.
FIG. 1 is a block diagram showing a configuration of a control device for a synchronous motor according to Embodiment 1 of the present invention, FIG. 2 is a block diagram showing a detailed configuration of a current component calculation circuit of the control device, and FIG. 3 is a frequency of the control device It is a block diagram which shows the detailed structure of a correction | amendment arithmetic circuit.

  In FIG. 1, symbol SM is a synchronous motor that performs field using a permanent magnet or a field winding, and a compressor (not shown) is connected to the synchronous motor SM as a load to assist supercharging.

  The control device for driving and controlling the synchronous motor SM includes a frequency command circuit 1, a subtraction circuit 2, a voltage command calculation circuit 3, a power conversion circuit 4, a current setting circuit 5, an integration circuit 6, a current component calculation circuit 7, and a frequency correction calculation circuit. 8 and current detectors 9 and 10 are mainly configured.

  Here, for example, speed information from an accelerator petal or the like (not shown) is input to the frequency command circuit 1, and the frequency command circuit 1 uses a primary frequency command value for driving the synchronous motor SM according to the speed information. Set f * and output.

Subtraction circuit 2 subtracts the frequency correction amount Δf given from frequency correction computing circuit 8 will be described later with respect to the frequency command circuit 1 of the primary frequency instruction f *, and outputs the subtracted value as the primary frequency f _1.

The voltage command calculation circuit 3 uses the information on the primary frequency f ₁ from the subtraction circuit 2 and the rotation angle θ of the rotor from the integration circuit 6 described later to obtain the three-phase AC voltage command values Vu *, Vv *, and Vw *. Calculate and output. Various control methods have been proposed for calculating the three-phase AC voltage command values Vu *, Vv *, and Vw *.

For example, in the generally well-known V / f control method, the primary frequency f ₁ and the primary line voltage value V ₁ are controlled to have a proportional relationship. For example,

The primary voltage command value V * is determined as follows. Here, K1 is a predetermined proportional coefficient.

  In actual control, Vd * is set to zero in the dq coordinate system, and the voltage command V * in the equation (1) is given as the q-axis voltage Vq *.

Note that when V * is calculated from a frequency f ₁ of zero [Hz] with a simple proportional relationship, the voltage drop due to the primary winding resistance is not taken into account, and there is a concern about torque shortage at low rotational speeds. For this reason, control is usually performed so as to give an offset voltage or the like in a low rotation speed region.

  The coordinate conversion formula from the two-phase dq coordinate system to the uvv three-phase AC coordinate system is generally

Represented as: The rotation angle θ in the equation (2) indicates the direction of the d-axis magnetic pole position when the direction of the field magnetic flux of the rotor is the d-axis.

  The rotation angle θ in this case is

Therefore, in sensorless control, θ [rad] is generally obtained by integrating the primary frequency f ₁ [Hz]. Here, assuming that Vd * = 0 and Vq * is the expression (1), the three-phase AC voltage commands Vu *, Vv *, Vw * are calculated from the above expression (2) using the rotation angle θ. The

  The power conversion circuit 4 converts a DC voltage Vdc supplied from a DC power source such as a storage battery (not shown) into a triangular wave comparison method or an instantaneous mode according to the three-phase AC voltage values Vu *, Vv *, Vw * input from the voltage command calculation circuit 3 Pulse width modulation (PWM) is performed by a voltage vector selection method or the like, and converted into a three-phase AC voltage having a variable frequency and variable voltage for output.

The current setting circuit 5 sets a current value i ₁ used for controlling the synchronous motor SM. That is, for the current setting circuit 5, information for determining a condition for performing an acceleration / deceleration operation that requires the maximum current or a condition for not performing an acceleration / deceleration operation such as an idling state of an internal combustion engine of a vehicle (for example, torque Information) is entered. Based on this information, the current setting circuit 5 sets an allowable maximum current value in a state where the power conversion circuit 4 does not trip when performing maximum acceleration / deceleration operation, and does not require acceleration / deceleration operation. When it is necessary to suppress the current consumption, a minimum current value is set so that the rotation operation can be maintained. In the case of a non-salient electric motor, since the q-axis current and the torque are in a proportional relationship, the torque can be directly controlled by the current value i ₁ set by the current setting circuit 5.

Integrator circuit 6, as shown in the aforementioned (3), by integrating the primary frequency instruction f ₁ at a predetermined cycle to calculate the rotation angle θ of the rotor of the synchronous motor SM.

The current detectors 9 and 10 detect two current values iu and iv of the U phase and the V phase flowing through the armature of the synchronous motor SM. In the first embodiment, sensorless control is performed by referring only to the current values iu and iv of the current detectors 9 and 10 without using a rotation sensor or a position sensor. In addition,
Here, two current values iu and iv of U phase and V phase are used, but not limited to this, any two current values iu, iv and iw of U phase, V phase and W phase are used. A value can be used.

  The current component calculation circuit 7 converts the current values iu and iv input from the current detectors 9 and 10 and information on the rotation angle θ obtained by the integration circuit 6 into a two-phase coordinate space to convert the current on the q axis. The value iq is output. That is, when considering the control of the synchronous motor SM, it is usually performed in the dq coordinate which is a rotating coordinate system, not the three-phase AC coordinate system. For this reason, the current component calculation circuit 7 converts the current in the three-phase coordinate system u−v−w into the current in the two-phase coordinate system d−q. Only the current value iq is calculated.

  In general, the determinant for converting from a three-phase coordinate system to a two-phase coordinate system is

It is expressed.

  Now, assuming that iw = −iu−iv, only the current value iq on the q axis is obtained from the above equation (4),

It becomes.

  FIG. 2 shows a detailed configuration of the current component calculation circuit 7 in the case of obtaining the current value iq on the q axis of the equation (5).

  In FIG. 2, reference numeral 12 denotes an input terminal to which the U-phase current value iu detected by the current detector 9 in the three-phase alternating current is input, and reference numeral 13 denotes V detected by the current detector 10 in the three-phase alternating current. An input terminal to which the phase current value iv is input, 14 is an input terminal to which the rotation angle θ obtained by the integrating circuit 6 is input, and 15 is an output terminal of the current iq on the q axis. Reference numerals 16, 17, and 18 denote multiplication circuits that multiply predetermined coefficients, 19 denotes an addition circuit that adds two inputs, and 21 denotes a trigonometric function that calculates and outputs sin θ and cos θ from information on the input rotation angle θ. The generation circuit, 20 and 22 are multiplication circuits for multiplying sin θ and cos θ, respectively, and 23 is a subtraction circuit for subtracting two signals.

  The operation of the current component calculation circuit 7 is based on the above-described equation (5), using information on two current values iu and iv of the three-phase current and the rotation angle θ of the rotor, on the q axis. The current value iq is calculated. Therefore, the coefficient of the multiplier 16 is √ (3/2), the coefficient of the multiplier 17 is 1 / √2, and the coefficient of the multiplier 18 is √2.

The frequency correction calculation circuit 8 uses the current setting value i ₁ input from the current setting circuit 5 and the current value iq on the q axis input from the current component calculation circuit 7 to provide a frequency correction amount for the primary frequency command value f *. Δf is calculated and output. That is, the frequency correction calculation circuit 8 performs a PI control calculation based on the deviation Δi between the current setting value i ₁ and the q-axis current value iq to calculate the frequency correction amount Δf. At that time, the condition determination using the values of the primary frequency f ₁ and the primary frequency command f * is performed, and when the acceleration / deceleration operation is performed, the current set value i ₁ is not corrected, and the acceleration / deceleration operation is not performed. When the condition is satisfied, the current set value i ₁ is corrected, and PI control calculation is performed using the corrected current set value i ₁ .

A detailed configuration of the frequency correction arithmetic circuit 8 is shown in FIG.
In FIG. 3, reference numeral 24 is an input terminal to which the primary frequency f ₁ obtained by the subtraction circuit 2 is input, 25 is an input terminal to which the primary frequency command f * from the frequency command circuit 1 is input, and 26 is a current setting circuit 5. Is an input terminal to which the current set value i ₁ given from is input, 27 is an input terminal to which the current value iq on the q axis given from the current component calculation circuit 7 is inputted, and 28 is an output from which the frequency correction amount Δf is outputted. Terminal. Also, 29 and 34 are subtracting circuits for subtracting two signals, 30 is a threshold comparing circuit for performing a threshold comparison on the difference between the primary frequency f ₁ and the primary frequency command f *, and 31 is a differential value of the primary frequency command f *. Is a determination circuit that determines whether or not the current setting value i ₁ needs to be corrected, and 33 is a correction circuit that corrects the current setting value i ₁ according to the output result of the determination circuit 33. Reference numeral 38 denotes a PI controller that performs PI control including multiplication of the proportional gain Kp and integral operation of the integral gain Ki, a multiplication circuit 35 that multiplies the proportional gain, and an integration circuit 36 that performs an integration process using a predetermined integral gain. And an adder circuit 37 for adding two input signals.

  Next, the operation of the control device for the synchronous motor SM having the above configuration, particularly the operation of the frequency correction arithmetic circuit 8 will be described in detail here.

  As described above, the current setting circuit 5 sets an allowable maximum current value in a state where the power conversion circuit 4 does not trip when it is desired to perform maximum acceleration / deceleration operation, and consumes current without requiring acceleration / deceleration operation. When it is necessary to suppress this, a minimum current value is set so that the rotation operation can be maintained.

  In response to this, the frequency correction arithmetic circuit 8 operates in two types of operation modes. That is, when accelerating / decelerating at the maximum rate, the current setting circuit 5 sets a maximum current value that is allowed within a range in which the power conversion circuit 4 does not stop operating due to overcurrent. Therefore, in the first operation mode, Then, control is performed such that the acceleration / deceleration operation is performed using the maximum current value (hereinafter, this operation mode is referred to as a current control mode).

On the other hand, when it is necessary to suppress current consumption without requiring acceleration / deceleration operations, the current setting circuit 5 sets the minimum current value that can hold the rotation operation. Therefore, in the second operation mode, When it is determined that the primary frequency f ₁ converges to the primary frequency command f * and is operating stably, the frequency correction amount Δf is gradually returned to a value near zero, and at the same time, the current set value i ₁ Is controlled to a value close to zero so that unnecessary current is not consumed (hereinafter, this operation mode is referred to as a recovery processing mode).

Therefore, in the former current control mode, the q-axis current value iq begins to be significantly different from the current set value i _{1 due to} a new setting of the current set value i _1, and the current value deviation Δi of (iq−i ₁ ). Is used to calculate the frequency correction amount Δf. That is, in this current control mode, the correction circuit 33 directly outputs the current set value i ₁ input without correction. Then, the PI controller 38 uses the deviation Δi (= iq−i ₁ ) of the current value obtained by the subtraction circuit 34 to adjust the frequency correction amount Δf so that the q-axis current value iq matches the current set value i ₁ . Perform the calculation.

  That is, the PI controller 38

Perform the operation. Here, Kp is a proportional gain, and Ki is an integral gain.

In this current control mode, when the acceleration / deceleration operation based on the current setting value i ₁ is performed and the primary frequency f ₁ and the primary frequency command f * converge to a state that substantially matches, the mode shifts to the recovery processing mode. In this recovery processing mode, the frequency correction amount Δf is gradually returned to a value near zero as described above, and at the same time, the current setting value i ₁ is corrected to a value near zero so that unnecessary current is not consumed. To control.

  Here, the determination circuit 32 determines whether or not the current control mode has shifted to the recovery processing mode based on the output of the threshold comparison circuit 30 and the output of the differentiation circuit 31.

That is, the subtraction circuit 29 calculates the difference between the primary frequency f ₁ input to the input terminal 25 and the primary frequency command f * input to the input terminal 26. Next, the threshold determination circuit 30 performs a size comparison determination with the threshold f _th for the difference between the primary frequency f ₁ that is the output of the subtraction circuit 29 and the primary frequency command f *.

  That is, the threshold comparison circuit 30

It is determined whether or not holds.

As described above, the threshold comparison circuit 30 performs the size comparison determination with the threshold f _th for the difference between the primary frequency f ₁ and the primary frequency command f * for the following reason.

When the synchronous motor SM is operating stably without being out of step, the primary frequency f ₁ is substantially equal to the motor rotational frequency, and the primary frequency f ₁ is treated as equivalent to the rotational frequency of the synchronous motor SM. It is possible. However, under the condition that the rotational speed fluctuates due to the presence of a load disturbance, even when the primary frequency f ₁ substantially converges to the primary frequency command f *, as shown in FIG. 4A, the primary frequency f ₁ It fluctuates with respect to the primary frequency command f * under the influence of disturbance. In this state, simply strictly determine the match / mismatch of the primary frequency f ₁ and the primary frequency instruction f *, the operation mode in a short period and factors switches misjudgment alternately to the current control mode and recovery mode Become.

Therefore, here, as shown in FIG. 4B, a fixed threshold value ± f _th is set as a determination margin for the primary frequency command f *, and the primary frequency f ₁ converges within this determination margin. Occasionally, so that be regarded as a primary frequency f ₁ is matched to the primary frequency instruction f *.

Although not shown here, when the threshold comparator circuit 30 to determine a match / mismatch of the primary frequency instruction f * and the primary frequency f ₁ is one primary frequency f ₁ is without such divergent primary frequency instruction f In order to confirm that it has converged stably to *, the condition of whether or not the primary frequency f ₁ has reached the primary frequency command f * is also determined at the same time, and the primary frequency f ₁ and the primary frequency command f * are The determination shown in the above equation (7) is performed within a predetermined time after the coincidence.

Further, the differentiating circuit 31 performs a differentiation operation on the primary frequency command f * to determine whether or not the primary frequency command f * has changed. That is, in determining whether the transition from current control mode to the recovery process mode, the primary frequency instruction f * is under conditions not changed, match / mismatch of the primary frequency f ₁ and the primary frequency instruction f * It is necessary to judge. Therefore, determination using a differential operation result of the primary frequency instruction f _1.

Then, the determination circuit 32 determines based on both the output of the threshold comparator circuit 30 and the differentiating circuit 31, whether the transition from current control mode to the recovery mode, namely whether to correct the current setpoint i _1. That is, the determination circuit 32
(1) | f * −f ₁ | <f _th (after f ₁ reaches f *)
(2) It is determined whether the AND condition of (1) and (2) that the output of the differentiation circuit 31 is zero is satisfied.

When the determination circuit 32 determines that the AND condition of (1) and (2) is satisfied, the correction circuit 33 corrects the current set value i ₁ to a value close to zero accordingly. The minimum current value for maintaining the rotation operation is taken. Therefore, when shifting to the recovery processing mode, the frequency correction amount Δf gradually becomes a value near zero, and the synchronous motor SM is driven at a speed corresponding to the primary frequency command f * set by the frequency command circuit 1. At the same time, the minimum current value (substantially zero) for maintaining the rotation operation is maintained.

  As described above, in the first embodiment, an optimal current value is always set and controlled in a state where the current of the synchronous motor SM is likely to fluctuate due to a load disturbance caused by exhaust gas of the internal combustion engine or the like. it can. Thereby, useless current consumption of a DC power source such as a storage battery can be suppressed, and the power conversion circuit 4 and the synchronous motor SM are protected from overcurrent, and the power conversion circuit 4 is stopped or synchronized by the overcurrent. It becomes possible to prevent the control operation of the electric motor SM from becoming unstable. Further, since the synchronous motor SM can be stably controlled without using a position sensor or the like, the control circuit can be configured at low cost.

  Here, as a transition condition from the current control mode to the recovery processing mode, the determination circuit 32 makes a determination based on the outputs of both the threshold comparison circuit 30 and the differentiation circuit 31, but the present invention provides such a method. It is not limited to only, and other mode transition determinations can also be made.

Embodiment 2. FIG.
FIG. 5 is a block diagram showing the configuration of the synchronous motor control device according to Embodiment 2 of the present invention, FIG. 6 is a block diagram showing the detailed configuration of the current component calculation circuit of the control device, and FIG. 7 is the frequency of the control device. FIG. 4 is a block diagram illustrating a detailed configuration of a correction arithmetic circuit, and components corresponding to or corresponding to those of the first embodiment illustrated in FIGS. 1 to 3 are denoted by the same reference numerals.

  The second embodiment is different from the first embodiment in the configuration of the current component calculation circuit 45 and the frequency correction calculation circuit 46. That is, in the first embodiment, the current component calculation circuit 45 calculates only the current value iq on the q-axis in the two-phase coordinate space. However, in the second embodiment, the d-axis in the two-phase coordinate space. Both the current values id and iq on the q axis are calculated. Further, the frequency correction calculation circuit 46 calculates the frequency correction amount Δf from the current values id and iq on the d axis and the q axis.

  Next, details of the current component calculation circuit 45 will be described with reference to FIG. In FIG. 6, the same components as those in Embodiment 1 (FIG. 2) are denoted by the same reference numerals, and detailed description thereof is omitted here.

  Now, when the current value id on the d-axis is obtained from the relational expression of iw = −iu−iv and the above-described expression (4),

It becomes.

  Therefore, the current component calculation circuit 48 in FIG. 6 obtains both the current values id and iq on the d-axis and the q-axis at the same time by the equations (5) and (8). Therefore, the current component calculation circuit 45 further includes multiplication circuits 40 and 41 for multiplying the values of sin θ and cos θ, and an addition circuit 42 for adding two signals to the configuration of the current component calculation circuit 7 shown in FIG. , And an output terminal 39 of the adder circuit 42 is added.

  Next, details of the frequency correction circuit 49 will be described with reference to FIG. In FIG. 7, the same components as those in Embodiment 1 (FIG. 3) are denoted by the same reference numerals, and detailed description thereof is omitted here.

  The frequency correction circuit 49 uses both the current values id and iq on the d axis and the q axis, for example,

The absolute current value I is obtained as follows, and the frequency correction amount Δf is calculated using the absolute current value I and the current setting value i ₁ . Therefore, this frequency correction circuit 49 is different from the configuration of the frequency correction arithmetic circuit 8 shown in FIG. An absolute value circuit 44 for obtaining the absolute value of the two-phase current using both the current values id and iq is added.

As described above, in the second embodiment, the current is controlled using the current absolute value I based on the two current values id and iq on the dq axis subjected to the 2-3 phase conversion. Therefore, the effective value of the phase current can be controlled more optimally, and it becomes possible to reliably prevent the control operation of the synchronous motor SM from becoming unstable.
Since other configurations, operations, and effects are the same as those in the first embodiment, detailed description thereof is omitted here.

Embodiment 3 FIG.
FIG. 8 is a block diagram showing the configuration of the synchronous motor control apparatus according to Embodiment 3 of the present invention, and the same reference numerals are given to components corresponding to or corresponding to those of Embodiment 2 shown in FIG.

The third embodiment is different from the second embodiment in the configuration of the voltage command calculation circuit 47. That is, in the first and second embodiments, the voltage command calculation circuit 3 uses the information on the primary frequency f ₁ and the rotation angle θ from the integration circuit 6 to calculate the three-phase AC voltage command values Vu *, Vv *, and Vw *. Although the calculation is performed, the voltage command calculation circuit 47 according to the third embodiment includes the current component calculation circuit 45 in addition to the information on the primary frequency f ₁ from the subtraction circuit 2 and the rotation angle θ from the integration circuit 6. The three-phase voltage command values Vu *, Vv *, and Vw * are calculated and output based on the current values id and iq on the d-axis and the q-axis.

  In this case, there are several methods for obtaining the voltage command value of the two-phase coordinate system from the current value of the two-phase coordinate system. For example,

The method of calculating as can be considered. Here, R1 is a value of the primary winding resistance, and is a term for correcting a voltage drop when a current flows. K2 is the same as the coefficient of V / f constant control, and in particular, is the same coefficient as the q-axis voltage command at no load.

As described above, in the third embodiment, the two-phase voltage command values Vd * and Vq * based on the currents id and iq on the dq axis that have undergone the 2-3-phase conversion by the voltage command calculation circuit 47 are used. Thus, since the synchronous motor SM is configured to be driven, the synchronous motor SM can be controlled with higher accuracy and stability.
Since other configurations, operations, and effects are the same as those in the second embodiment, detailed description thereof is omitted here.

  The control circuit of the synchronous motor SM described above is an example of the present invention, and may be appropriately changed as long as the same operation and effect can be obtained. The synchronous motor SM may be a field winding type synchronous motor or a permanent magnet synchronous motor, and the voltage command calculation circuits 3 and 47 are not limited to V / f constant control. The same sensorless control may be used.

  In each of the first to third embodiments, the case where the present invention is applied to an assist device for a vehicle internal combustion engine has been described as an example. However, the present invention is not limited to this and can be applied to all control devices for a synchronous motor SM. . In any case, the present invention is not limited to these first to third embodiments, and various modifications and applications can be made without departing from the spirit of the present invention.

SM synchronous motor, 1 frequency command circuit, 2 addition circuit,
3, 47 Voltage command calculation circuit, 4 Power conversion circuit, 5 Current setting circuit, 6 Integration circuit,
7,45 Current component calculation circuit, 8,46 Frequency correction calculation circuit, 9,10 Current detector.

Claims (4)

A power conversion circuit for converting DC power supplied from a DC power source into AC power having a variable voltage and variable frequency and outputting the AC power;
A current detector for detecting a primary phase current supplied from the power conversion circuit to the synchronous motor;
A current component calculation circuit that calculates a current component based on the primary phase current and rotation angle information detected by the current detector;
A current setting circuit for setting a current set value used for controlling the synchronous motor;
The current setting value set by the current setting circuit and the current component calculation circuit are calculated so that the current of the current setting value flows when the acceleration / deceleration operation is performed, and the current is suppressed when the acceleration / deceleration operation is not performed. A frequency correction arithmetic circuit for obtaining a frequency correction amount from the current component;
A subtraction circuit for subtracting the frequency correction amount calculated by the frequency correction arithmetic circuit from a primary frequency command value to obtain a primary frequency;
An integration circuit that integrates the primary frequency obtained by the subtraction circuit and converts it into the rotation angle information;
A voltage command calculation circuit that calculates a primary voltage command value based on the primary frequency obtained by the subtraction circuit and the rotation angle information obtained by the integration circuit and outputs the primary voltage command value to the power conversion circuit;
The control apparatus of the synchronous motor provided with. It said frequency correction computing circuit, before the primary frequency reaches the primary frequency command value described above, so that by maintaining the current set value set by the current setting circuit is the primary frequency reaches the primary frequency command value above After the frequency correction amount is calculated and the primary frequency reaches the primary frequency command value , the current setting value set by the current setting circuit is corrected, and the primary frequency is reduced with less current. 2. The synchronous motor control device according to claim 1, wherein the control device maintains the primary frequency command value. The frequency correction arithmetic circuit has a condition that a difference between the primary frequency command value and the primary frequency is kept below a certain threshold during a predetermined period after the primary frequency reaches the primary frequency command value. The synchronous motor control device according to claim 2, wherein when the condition is satisfied, the current set value is corrected. In addition to the primary frequency obtained by the subtraction circuit and the rotation angle information obtained by the integration circuit, the voltage command computation circuit is configured to perform the primary operation based on the current component obtained by the current component computation circuit. The synchronous motor control device according to any one of claims 1 to 3, wherein the voltage command value is calculated and output.

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