JP5656750B2 - Single carrier receiver - Google Patents

Description

  The present invention relates to a single carrier transmission system, and more particularly to a single carrier receiver suitable for wireless relay using an antenna having a relatively sharp directional characteristic.

  As part of the television broadcasting business, there is a relay service. At this time, the single carrier QAM method has been widely adopted as a kind of FPU (Field Pickup Unit) modulation method for wirelessly transmitting video at a relay site to a television station. Therefore, it is standardized as ARIB STD-B11 “Portable microwave band digital radio transmission system for transmitting television broadcast program material”.

By the way, this transmission method is mainly used for fixed wireless relay, but in this case, in order to establish a transmission line, at the relay site (transmission point) and reception point, the direction of sending out radio waves and the direction of arrival of radio waves Therefore, it is necessary to adjust the direction of the antenna so that the antenna is directed accurately toward the antenna (for example, see Patent Document 1).
Therefore, a method conventionally used for adjusting the antenna direction of the FPU will be described with reference to FIG.

In FIG. 3, first, the transmission baseband unit 31, the transmission high-frequency unit 32, and the transmission antenna 33 are installed at the relay site, and then the reception antenna 1, the reception high-frequency unit 2, the reception baseband unit 37, and the reception signal level indicator. 38 is installed on a relay receiving side such as a television broadcasting station.
At the relay site, transmission signals generated in the transmission baseband unit 31 and the transmission high frequency unit 32 are transmitted as radio waves by the transmission antenna 33.

As a result, the radio wave radiated from the transmission antenna 33 reaches the reception antenna 1 on the relay reception side such as a television broadcasting station via the transmission path 34.
However, at this time, in the transmission line 34, the radio wave level is attenuated due to the blocking of the propagation path by a building or the like, the atmosphere, or rainfall.

  Therefore, the signal received by the receiving antenna 1 is input to the receiving high-frequency unit 2, where the received signal level is detected by an analog detector and subjected to automatic gain control and amplified to an optimum level. Finally, the signal is supplied to the reception baseband unit 37, demodulated here, and used as a signal for relay broadcasting.

At this time, the reception signal obtained from the reception high-frequency unit 2 is supplied to the reception signal level indicator 38 for adjusting the direction of the antenna.
The received signal level indicator 38 converts the level of the received signal into a shake of the meter pointer, the loudness or tone color of the sound emitted from the speaker, etc. so that the operator can recognize the level of the received signal.

  Therefore, the operator at the relay site and the operator on the relay reception side communicate with each other, and while looking at the level of the received signal displayed on the received signal level indicator 38, the direction of the transmitting antenna 33 and the receiving antenna 1 is moved up, down, left and right. The direction of the transmitting antenna 33 and the receiving antenna 1 is adjusted by searching slightly for the direction in which the received signal level becomes maximum.

Japanese Patent No. 4354004

The above prior art does not give consideration to the point that a received signal of a certain level or more is required for antenna direction adjustment, and there is a problem that direction adjustment is extremely difficult in the initial stage.
Here, in the antenna direction adjustment in the conventional FPU, the reception signal level is detected by the analog detector of the reception high frequency section.

  However, in this case, if the level of the received signal is lowered, the received signal is buried in the noise of the first stage amplifier of the received high-frequency section, and it is difficult to detect with high accuracy. For this reason, the conventional technique performs accurate detection. For this purpose, for example, a C / N of 7 dB or more is required, although it depends on the accuracy of the detector.

  For example, when the received signal level is about -90 dBm and the C / N is about 7 dB, it is difficult to detect a level at which the received signal is less than -90 dBm with the prior art. Even if it reaches the antenna on the relay reception side, if it is a reception signal of a slight level, the reception signal cannot be captured on the relay reception side. There wasn't.

  Moreover, in the case of the FPU system, an antenna having a narrow directivity angle is often used in order to obtain a high antenna gain. For this reason, in the initial stage of the antenna direction adjustment in which the antenna direction adjustment between transmission and reception has not yet been performed, it is natural that the reception signal level is very low and is less than −90 dBm.

  An object of the present invention is to provide a single carrier receiver capable of detecting the presence of a received signal even when the level of the received signal is considerably low.

In order to solve the above-mentioned problems, the present invention is a transmission apparatus for transmitting a single carrier signal, wherein the frame structure of the transmission signal is composed of a preamble signal whose amplitude and phase are known and a data signal for transmitting data. Transmit the signal.
In the receiving apparatus of the transmitting apparatus that receives the transmission signal, N _P (N _P is an integer of 2 or more) samples of a predetermined time interval in the received sample sequence converted from the single carrier signal is held and output. storage means, said N _P samples column by the storage means, and N _P-number of signal sequences of said predetermined constant-time interval in the known preamble signal, a complex multiplying means for complex conjugate multiplier in accordance with a sequential, the complex multiplication for each of N _P-number of the complex conjugate multiplication result by the means and the multiplying means multiplying means for multiplying the corresponding weighting factor, and integrating means for adding N _P number of operation result of the multiplication means is provided, wherein said N _P samples string storing means outputs, while shifting one sample corresponding to said predetermined constant-time intervals, the complex multiplication means, the multiplication means and Provided is a receiving device that operates the integrating means and estimates a received signal level based on an integration result by the integrating means.

To improve the S / N in the receiving apparatus, the storage means, the at predetermined time intervals, a shift register for one sample shifts N _P samples of the holding and outputting the weight coefficient, at least partially consists of N _P number of coefficients corresponding to the center portion of the impulse response of the low pass filter divides the multiplication result obtained by multiplying the weight coefficient K (K is an integer of 2 or more) sets said integration means Is a local addition in each of each set, and further includes a powerizing means for calculating the square of the absolute value of the integration result of each set, and an adding means for adding the square of the absolute value of each set. And the received signal level is estimated based on the result of the adding means.

To further improve the S / N in the receiving apparatus, the weighting factor coefficients become a central portion of the impulse response of the low-pass filter the K or K-1 pieces of from approximately N _P pieces linked a column has the coefficient sequence of a different delay time from each other B (B is an integer of 2 or more) sets said multiplier means, said N _P-number of complex conjugate multiplication results, and the corresponding weighting factor For each of the B sets, the power generating means calculates the integral result squared for each of the B sets, and the adding means adds the absolute value squared results obtained from the B set, The reception signal level is estimated based on the addition result by the adding means.

  Further, a method for adjusting the direction of the receiving antenna is generated based on the estimated received signal level, and the direction of the receiving antenna is adjusted using the generated direction adjusting signal. And an apparatus for the same.

It should be noted that the present invention can also include a modification of a mode for carrying out the invention to be described later while maintaining mathematical equivalence on the premise of a linear time invariant (LTI) system. For example, in order to include those having a value of 0 to N _P number of coefficients, or contain the same value for N _P-number of coefficients are interchangeable the order of addition and multiplication, complex conjugate multiplication result even when N may _P number less than, is represented by generalizing the N _P pieces like. The weighting factor corresponds to a finite impulse response, and the number of multiplications can be reduced by polyphase decomposition. Such expressions do not preclude the invention from including equivalent variations of the above.

According to the present invention, at the initial stage of antenna direction adjustment, the presence of a signal can be detected even if the CN ratio of the received signal is about −10 dB, that is, the received signal level is about −107 dBm. Become.
Therefore, according to the present invention, it is possible to search for a direction in which the received signal level is maximized while changing the direction of the antenna, and as a result, the antenna direction can be easily adjusted.

The block block diagram of the single carrier receiver which concerns on Example 1 of this invention. Explanatory drawing of the transmission frame format in a single carrier system. The block diagram of the conventional FPU. The block diagram of the complex multiplication part in one embodiment of this invention. Explanatory drawing which shows rotation of the complex multiplication signal by carrier frequency shift | offset | difference. The block diagram of the weighting coefficient multiplication part and integrator in one embodiment of this invention. The figure explaining an example of a weighting coefficient. The figure explaining the output signal of a weighting coefficient multiplication part. The figure explaining the frequency characteristic of the output signal of a weighting coefficient multiplication part, and the frequency characteristic of a weighting coefficient. The figure explaining the output signal of a weighting coefficient multiplication part when a sampling clock phase shifts. The figure explaining the correction characteristic of the output of a maximum value average part, and a received power converter. The block block diagram which shows 2nd embodiment of the single carrier receiver by this invention. The figure explaining the weighting coefficient in the weighting coefficient multiplication part in 2nd embodiment of this invention. The block diagram of the weighting coefficient multiplication part in one embodiment of this invention, a local integrator, and a local absolute value unit. The figure explaining the output level characteristic of C / N versus absolute value machine in the 2nd example. The block block diagram which shows 3rd embodiment of the single carrier receiver by this invention. The figure explaining the weighting factor in the weighting factor multiplication part in 3rd embodiment of this invention. The figure explaining the output level characteristic of C / N vs. absolute value device in the third embodiment.

Hereinafter, a single carrier receiver according to the present invention will be described in detail with reference to an embodiment shown in FIG.
First, before describing specific embodiments, the signal format of a transmission signal in a single carrier scheme will be described. As shown in FIG. 2, the transmission signal is used for facilitating equalization processing at a receiving unit. and duration of the preamble signal P (t) consisting of N _P samples, the frame is constituted by the period for transmitting the data signal D (t) consisting of the N _D sample. Here, t is the sample _{number, N P,} N _D is an integer of 2 or more, respectively.

The preamble signal P (t) is generated as a signal sequence having a known amplitude and phase during the preamble period.
As a known signal sequence generation method at this time, a known pseudorandom signal (PN) or the like is often used, and a signal (symbol) using a modulation scheme such as BPSK or QPSK is often used.

Next, the signal D (t) in the data period transmits data using a modulation method corresponding to the transmission rate, such as BPSK method to 64QAM method. 1 the frame length is 19200 symbols example, N _P and N _D Depending on the oversampling ratio, the corresponding symbol as many or twice, to quadruple the like.
Then, the framed signal is repeatedly transmitted.

<Example 1>
FIG. 1 is a block diagram of a single carrier receiving apparatus according to Embodiment 1 of the present invention.
As already described, the signal transmitted as a radio wave from the transmitting antenna at the relay site is received by the receiving antenna 1 and converted into a baseband signal (or an intermediate frequency signal) by the receiving high-frequency unit 2.

The baseband signal is input to the A / D 3 and sampled and quantized at a time interval having a predetermined oversample ratio with respect to the symbol rate to obtain a received sampling sequence Rin (t).
The obtained received sampling sequence Rin (t) is input to the quadrature detector 4, where a conversion process from a real signal to an IQ (In-phase / Quadrature) complex signal is performed, and the received complex sampling sequence Z ( t). Here, as indicated above, t is a sample number.

This quadrature detection processing may be performed by digital signal processing, but may be performed by using an analog quadrature mixer and sampling each IQ using an A / D converter.
The obtained reception complex sampling sequence Z (t) is input to the complex multiplier 5.

The complex multiplier 5, as shown in FIG. 4 includes a storage element 41, such as a shift register having a number of samples N _P about the same length of the preamble period, and N _P-number of the complex multipliers 42. The storage element 41 receives the received complex sampling sequence Z (t), and each time the sample number t increases by 1, the value Z (t, m) output from each stage is shifted to the right by one. Note that m is an integer from 0 to NP-1.
N _P-number of the complex multipliers 42, as shown in the following equation (1), the value of each stage of the shift register 41 Z (t, m), predefined preamble signal P at the transmitting side (m) of the complex conjugate signal P ^* and (m), complex multiplication according to their arrangement, each time a sample number t increases 1, N _P-number of complex multipliers signal M (t, m) obtained.

Formula (1)

Here, when the preamble signal is BPSK modulated, the complex multiplication process can be easily realized by the process of inverting / non-inverting the sign of Z (t, m).
At this time, in the receiving apparatus, the clock frequency and the carrier frequency on the transmission side can be accurately reproduced, and the timing input to the shift register 41 coincides with the preamble period of the received sample sequence Z (t). In this case, the complex multiplication signals M (t, m) are all in-phase signals.

However, in the present invention, since the purpose is to estimate the received signal power at a low C / N, it is difficult to perform a correct carrier regeneration process in a region where the C / N is low, and the carrier frequency is likely to shift. In such a case, the complex multiplication signal M (t) has a constant phase shift within the NP sample period, and rotates as shown in FIG.
The amount of rotation at this time is expressed by the following equation (2), where the carrier frequency deviation amount is Δf, and the complex reception sampling sequence Z (t) in the preamble period.

Formula (2)

Here, fCLK is a reception clock frequency, θ is a fixed phase, and N (t) is a noise signal.
Therefore, the complex multiplication signal M (t, m) shown in the equation (1) when the rotation occurs is expressed by the following equation (3).

Formula (3)

  However, at this time, the following equation (4) was used.

Formula (4)

Incidentally, a general correlation is, M (t, m) and although integration process in the integration range N _P, if such a rotation occurs signals, it being integration converted to offset the signal between the antiphase As a result, the result is close to 0, and it is difficult to estimate the received signal level from the correlation result.

For this reason, in this embodiment, the complex multiplication signal M (t, m) is input to the weighting coefficient multiplication unit 6 so that the received signal level can be estimated with high accuracy even when rotation occurs.
Here, in the weighting coefficient multiplier unit 6, 6, as shown in equation (5) and (6), N _P-number of the complex multipliers 61 complex multiplication signal M (t, m) and the weighting factor W Multiplication with (m) is performed in the range of m from 0 to N _P −1, and the integrator 7 adds them.

Formula (5)
Formula (6)

An example of the weight coefficient W (m) is shown in FIG. Weighting factor in this example W (m) is a real number, which the impulse response of an ideal low-pass filter, the same finite length fraction and the preamble period around its maximum amplitude position (i.e. N _P samples), were excised As long as it functions as a low-pass filter, it may be used. This filter operation will be described in detail with reference to FIG.

  FIG. 8 shows the values (real parts) of the complex multiplication signal M (t, m) at each sample time t when the vertical axis is the shift register position m. In the example of FIG. 8, the timing input to the shift register 41 at the sample time t = t ′ + 0 coincides with the preamble period of the received sample sequence Z (t), and as shown in FIG. M (t ′ + 0, m) is a signal with a substantially constant phase rotation. At sample times other than t = t ′ + 0, the contents of the shift register 41 and the preamble signal P (m) are uncorrelated, so the phase is random.

  Considering the frequency characteristics of the complex multiplication signal M (t, m) at each sampling time t in these respects, as shown in FIG. 9, the complex multiplication signal M (t, t, at the sampling time t = t ′ + 0, as shown in FIG. The frequency characteristic of m) has a peak at a frequency position in the vicinity of the direct current that matches the rotational speed (solid line in FIG. 9), and signals at times other than t = t ′ + 0 are substantially flat frequencies from low to high frequencies. It has characteristics (dotted line in FIG. 9). Further, with respect to the frequency characteristic signal of such a low-pass filter, the signal at the time when the low frequency energy is large due to the frequency characteristic (chain line in FIG. 9) formed by the weighting factor multiplier 6 and the integrator 7, Only the signal at the time when the preamble signal arrives (t = t ′ + 0) increases the output level of the integrator 7. At other times t, the output is suppressed by the low-pass filter, and the output level of the integrator 7 becomes small.

  As for the characteristics of the low-pass filter, it is desirable that the maximum preamble rotation component generated by the frequency deviation of the synthesizer in the baseband part and the high-frequency part is the passband width, and the gain between them is constant. If the passband width is narrow and the frequency deviation is large, the frequency of the rotated preamble signal becomes high, deviates from the passband of the filter, and is not detected as a preamble signal. In addition, if the passband width is wide, the noise component mixed in increases and the S / N decreases, so it is necessary to have an appropriate frequency characteristic.

As described above, the present invention is greatly different from a general correlator in that a filter is formed by further multiplying the output result of the complex multiplier 5 by a weighting factor.
In addition, the weight coefficient multiplication unit 6 applies the low-pass filter to the received signal itself that is frequency-spread with the preamble pattern even if a low-pass filter is inserted on the input side of a general correlator. Therefore, it is meaningless at all just to lose the broadband information of the preamble signal. Even if it is inserted on the output side of the correlator, the integration result of the rotation signal is already close to 0 as described above, and this also does not make sense. Thus, the configuration is different from that of a general correlator.

Next, the output value of the integrator 7 is subjected to an absolute value square calculation by the absolute value unit 8, and an absolute value (power) signal A (t) is output.
The absolute value signal A (t) is input to the filter unit 9. The filter unit 9 performs a convolution operation between the filter coefficient F (n) and the absolute value signal A (t), and outputs a filter result C (t). This is in order to reduce the influence when clock recovery (symbol synchronization) is not performed, and will be described below.

  As described above, the C / N is low at the initial stage of antenna direction adjustment, and correct carrier reproduction is difficult. However, correct clock reproduction is also difficult. When the clock recovery is correctly realized, as described in FIG. 8, the preamble component appears only at a certain sample time t, and the other time becomes the data component. However, when the clock recovery is not performed correctly and the sampling clock phase is shifted, as shown in FIG. 10, the preamble component appears in the time t when A (t) is maximum, and the samples before and after that, As the sample time increases, the preamble component decreases.

  Thus, the signal energy of the preamble is not concentrated only at a certain time t, but the signal energy of the preamble has a temporal spread. Therefore, by adding the maximum value and the signals on both sides thereof, the signal energy of the spread preamble is integrated to compensate for the sampling clock phase shift. To add these adjacent signals, the filter unit 9 may perform a convolution operation with the absolute value signal A (t) using the filter coefficient F (n) as a 3-tap rectangular signal.

By the way, the above is a case where a simple additive white noise model (AWGN) is assumed as a model of a propagation path (transmission path), but a multipath environment in which a plurality of reflected waves exist in an actual propagation path. The case is assumed.
In such an environment, a rectangular signal having a predetermined time width W may be used as the filter coefficient F (n) in order to calculate the sum of the energy of a plurality of existing multipaths. As the window width W at this time, it is desirable that the window width W be equal to or greater than L, where L is the longest multipath delay time assumed in advance.

However, if the window width W is set longer than necessary, noise components in the window width W increase and S / N deteriorates. Therefore, the window width W needs to be designed to an appropriate value.
Further, when the multipath delay time can be sequentially calculated on the receiver side, the window width W may be adaptively controlled according to the reception environment.

The output signal C (t) of the filter unit 9 is input to the maximum value detection unit 10, and the maximum value detection unit 10 searches for the maximum value of C (t) within one frame period and uses the maximum value signal MAX as the frame period. To output.
Then, the maximum value MAX is averaged by the maximum value averaging unit 11, where the disturbance component is removed.
The time constant for averaging at this time is preferably short enough to quickly follow the antenna direction adjustment control, for example, within a few hundred msec.

  Therefore, according to this embodiment, a signal level corresponding to the received signal level is output even with a low C / N. As a result, a weak received signal at the initial stage of antenna direction adjustment can be easily captured. be able to. If frame synchronization is established, the above-described operation may be performed only at t near the frame synchronization timing.

The output signal of the maximum value averaging unit 11 is input to the reception power conversion unit 12, where it is converted into a reception power level corresponding to the input value and output. Therefore, details of this conversion will be described with reference to FIG.
Normally, the receiving device performs various signal processing after controlling the level of the received signal, which varies greatly depending on the receiving conditions, to a constant level by an automatic gain control (AGC) circuit. Is used.

This is no exception in this embodiment.
Then, in this case, the power of the signal Rin (t) input to the A / D 3 is also kept constant. As a result, the output signal of the maximum value averaging unit 11 is indicated by a broken line in FIG. As shown by the solid line, the output signal level of the maximum value averaging unit 11 gradually approaches a certain constant value when the C / N increases, that is, when the received signal level increases, as indicated by the solid line. .
Further, in the region where C / N is low, the noise component is dominant over the signal component, and when C / N is low, that is, even when the received signal level is small, the output signal level of the maximum value averaging unit 11 is constant. Asymptotic to the value.

  Therefore, the reception power conversion unit 12 multiplies the output level of the maximum value averaging unit 11 by the correction characteristic which is the reverse characteristic as shown by the solid line in FIG. The converted reception level is output as a direction adjustment signal A. At this time, as shown in FIG. 11, the value of the reception level signal is converted into a unit of dB, but may be converted into a value of an arbitrary unit system that facilitates antenna direction adjustment.

Through the processing described above, a signal level corresponding to the received signal level can be output from the received power converter 12 even at a low C / N, and a weak received signal can be captured even in the initial stage of the direction adjustment of the receiving antenna. become.
The direction adjustment signal A thus output from the reception power conversion unit 12 is input to the reception signal level indicator 13.

The received signal level indicator 13 converts the level of the direction adjustment signal A into visual information such as the deflection of the meter pointer, the waveform on the monitor surface, and the color so that the operator can easily adjust the antenna direction. So that it can be recognized by the operator.
At this time, you may convert into auditory information, such as a musical scale and a sound volume.

  Therefore, according to this embodiment, the level of the direction adjustment signal A can be easily recognized by the operator, and at this time, as described above, even if the C / N of the reception signal is about −10 dB, the level of the reception signal is accurate. As a result, according to this embodiment, the initial stage of the direction adjustment of the receiving antenna which cannot detect even the presence of the received signal by the conventional technique using the received detection level at the received signal level of C / N of about −10 dB. In this case, the received signal level can be easily detected.

  Therefore, according to this embodiment, by using the calculated direction adjustment signal A, the operator can easily find the direction in which the received signal level is maximized while changing the direction of the receiving antenna.

<Example 2>
Next, a second embodiment will be described with reference to FIG. The second embodiment is the same as the first embodiment shown in FIG. 1 except that the integrator 7 is replaced with a local integrator 7a and the absolute value unit 8 is replaced with a local absolute value unit 8a. The configuration is the same as that of the embodiment. In the following description, the description of the processes denoted by the same reference numerals as in the first embodiment is omitted.

In the first embodiment, as shown in FIG. 7, the value of the weighting factor W (m) constituting the low-pass filter decreases as the distance from the center m = N _P / 2 increases. The coefficient at the end contributes to a decrease in ringing at the end of the pass band. However, in the present invention, rather than the ringing characteristic at the end of the pass band, the problem is that the S / N improvement is reduced due to the small value of the end coefficient. It becomes. Therefore, in the second embodiment, as shown in FIG. 13, the end of the weighting coefficient W (m) (the broken line portion in FIG. 13) is removed, and K sets of effective weighting coefficients near the center (K is an integer of 2 or more). ) Are combined to define a new weighting factor W ′ (m) of the same length, and W ′ (m) is used as the weighting factor of the weighting factor multiplier 6 to improve the S / N. The S / N improvement technique will be described below.

  The output from the weighting factor multiplier 6 is input to the local integrator 7a. As shown in FIG. 14, the local integrator 7a performs addition for each block of the weight coefficient W ′ (m), and outputs the addition result. Since the phases of the outputs of the local integrators 7a are different from each other, if they are coherently added in the complex region, cancellation due to different phases occurs, so that coherent addition cannot be performed. Therefore, the addition result from the local integrator 7a is input to the local absolute value unit 8a, and the local absolute value unit 8a performs an absolute value square operation on each addition result and then performs non-coherent addition. Non-coherent addition is inferior to coherent addition, but an S / N improvement effect can be obtained. With these processes, it is possible to reduce the noise floor as compared with the first embodiment even in a low C / N environment as shown in FIG.

<Example 3>
Next, a third embodiment will be described. The third embodiment provides an antenna direction adjustment method in which the S / N is further improved in the first embodiment or the second embodiment.

  FIG. 16 shows a configuration in which the third embodiment is applied to the second embodiment shown in FIG. When the third embodiment is applied to the first embodiment, the local integrator 7a is replaced with the integrator 7 and the local absolute value unit 8a is replaced with the absolute value unit 8, and the rest is the same. It is the composition. Further, in FIG. 16, the description of the processes denoted by the same reference numerals as those in the first embodiment or the second embodiment is omitted.

In the third embodiment, B (B is an integer of 2 or more) sets of weight coefficient multiplier 6, local integrator 7a and local absolute value unit 8a are prepared, and the output of complex multiplier 5 is used as a weight coefficient multiplier of each group. Connect to 6. Here, when the weighting factor of the weighting factor multiplier 6 in the b-th stage (b is an integer from 1 to B) is W′b (m), as shown in FIG. 17, the weighting factor W′b ( The peak position of m) is not overlapped with other weight coefficient peak positions. Furthermore, if the weighting factors W′b (m) at each stage are orthogonal to each other, a more effective weighting factor is obtained.
The output of the weight coefficient multiplication unit 6 at each stage is non-coherently added by the adder 14 via the local integrator 7a and the local absolute value unit 8a. By these processes, as shown in FIG. 18, it is possible to improve the S / N in a low C / N environment as compared with the first embodiment or the second embodiment.

  For example, when focusing on the weighting factor W′0 (m) in FIG. 17, the S / N is improved by effectively using the signal energy of the preamble at the position m where the value of W′0 (m) is small. I can't let you. For this reason, in the weight coefficients W′1 (m) and W′2 (m) of the other stages, a large value coefficient is arranged at a position m where the value of W′0 (m) is small, so that the signal energy of the preamble The S / N can be improved by fully utilizing the above. The weighting factor multiplication unit 6 at each stage has different group delay times when viewed as a filter by these weighting factors W′0 (m), W′1 (m), and W′2 (m). become. It should be noted that the weighting factor W′0 (m) contains just three effective weighting factor sequences (K), and W′1 (m) and W′2 (m) are two (K−1). Although both sides are filled with coefficients of 0, it is desirable to adjust the scale appropriately before adding by the adder 14 so that the detected value does not depend on the position m.

In this example, the weighting factor multipliers 6 of each stage are configured with different weighting factor peak positions, but may be configured with different passbands.
As described above, according to the second or third embodiment, it is possible to calculate the highly accurate direction adjustment signal A even under a C / N environment lower than that of the first embodiment. It becomes possible.

  The present invention can be used in a single carrier transmission system called an FPU system for wirelessly transmitting video at a relay site to a television station.

1 receiving antenna 2 receiving high frequency part 3 A / D (analog / digital converter)
4 Quadrature Detector 5 Complex Multiplication Unit 6 Weighting Factor Multiplication Unit 7 Integrator 8 Absolute Value Unit 9 Filter Unit 10 Maximum Value Detection Unit 11 Maximum Value Average Unit 12 Received Power Conversion Unit 13 Received Signal Level Indicator 31 Transmission Baseband Unit 32 Transmission high-frequency unit 33 Transmitting antenna 34 Transmission path 37 Reception baseband unit 38 Received signal level indicator 41 Shift register 42 Multiplier 7a Local integrator 8a Local absolute value unit

Claims (5)

In a receiving apparatus of a transmission apparatus that transmits a single carrier signal having a known preamble signal and a data signal in one frame,
Storage means for holding and outputting N _P (N _P is an integer of 2 or more) samples of a predetermined time interval in a sequence of received samples converted from the single carrier signal;
Said N _P samples column by the storage means, and N _P-number of signal sequences of said predetermined constant-time interval in the known preamble signal, a complex multiplying means for complex conjugate multiplier in accordance with a sequential,
For each of N _P-number of the complex conjugate multiplication result of the complex multiplication means, and multiplying means for multiplying the corresponding weighting factor,
An integration means for adding N _P number of operation result of the multiplication means is provided,
The complex multiplication means, the multiplication means and the integration means are operated while shifting the NP sample strings output from the storage means by one sample corresponding to the predetermined time interval, and the integration result by the integration means is obtained. A receiving apparatus that estimates a received signal level based on the received signal level. The receiving device according to claim 1,
Said storage means, N and _P number of the complex conjugate multiplication result K by the complex multiplier means (K is an integer of 2 or more) is divided into sets, at the predetermined time interval for each set, said holding and output N A shift register that shifts _P samples by one sample;
The weighting factor is made N _P number of coefficients at least partially corresponding to the center portion of the impulse response of the low-pass filter,
The multiplication result obtained by multiplying the weighting factor is divided into K (K is an integer of 2 or more) pairs,
The integrating means adds locally in each of the sets, and
Power generation means for calculating the square of the absolute value of each set of integration results;
Adding means for adding the absolute value squared results of each set,
A receiving apparatus that estimates a received signal level based on a result of the adding means. The receiving device according to claim 1,
The weighting factor has B (B is an integer of 2 or more) sets of coefficient sequences having different peak positions,
The multiplication means includes a said N _P-number of the complex conjugate multiplication result, a multiplication with the corresponding weighting factor, performs for each of the B group,
The integration means integrates the multiplication results of the multiplication means obtained for each of the B groups for each B group, and
A power generation means for calculating an absolute value square of each of the B sets of integration results;
Adding means for adding the B sets of absolute value square calculation results,
A receiving apparatus that estimates a received signal level based on an addition result of the B sets by the adding means. The receiving device according to claim 2,
The weighting factors are the K or K-1 pieces of, a coefficient sequence composed of substantially N _P number linked, the peak positions are different from each other B (B is an integer of 2 or more) have a said coefficient sequence of sets And
The multiplication means includes a said N _P-number of the complex conjugate multiplication result, a multiplication with the corresponding weighting factor, performs for each of the B group,
The power generating means performs an absolute value square operation on the integration result for each of the B groups,
The adding means adds the absolute value square results obtained from the set B,
A receiving apparatus that estimates a received signal level based on an addition result obtained by the adding means. The receiving device according to claim 1, wherein
A receiving apparatus that generates a direction adjustment signal for a reception antenna based on the estimated reception signal level, and performs direction adjustment of the reception antenna using the generated direction adjustment signal.