JP6432476B2 - Radio wave arrival direction estimation device, radio wave arrival direction estimation system - Google Patents

Description

  The present invention relates to a radio wave arrival direction estimation device that estimates from which direction a radio wave emitted by a radio tag comes in order to estimate a direction in which the radio tag exists, and a radio wave arrival direction including the device and the radio tag It relates to an estimation system.

  The Pseudo-doppler method is known as one of the methods for estimating the radio wave arrival direction (for example, Non-Patent Document 1). In the Pseudo-doppler method, an antenna is mounted on a rotating plate or the like, and the antenna is moved to create a Doppler shift for radio waves emitted from a radio wave source. Since the signal received by the antenna changes due to the created Doppler shift, the change of the signal is analyzed to estimate the direction of the radio wave source. In Non-Patent Document 1, Fourier transform is used for analysis of a signal received by an antenna.

Chang, H.-L., Tian, J.-B., Lai, T.-T., Chu, H.-H., and Huang, P., Spinning beacons for precise indoor localization, to appear in ACM Sensys '08.

  In order to use the Pseudo-doppler method to install a device with an antenna fixed on a rotating disk indoors, it is necessary to separate the direct wave and the reflected wave caused by multipath.

  The direct wave and the reflected wave generated by multipath have different arrival directions of radio waves. That is, the direction of the radio wave source is apparently different between the direct wave and the reflected wave. A radio wave received by a moving antenna undergoes a Doppler shift reflecting the direction of the radio wave source. Therefore, the direct wave and the reflected wave can be separated by increasing the frequency resolution.

  In the Fourier transform, the frequency resolution Δf is given by the reciprocal of the window width. That is, if the window width to be analyzed is widened, the frequency resolution is high (Δf is small), and conversely if the window width is narrowed, the frequency resolution is low (Δf is large).

  Of course, the greater the frequency difference between the direct wave and the reflected wave, the easier the separation of the direct wave and the reflected wave. Therefore, it is necessary to increase the Doppler shift amount.

  For these reasons, the apparatus described in Non-Patent Document 1 rotates a large disk with a slow rotation cycle. Even if the rotation period is slow, if the disk is large, the speed of the antenna installed in the vicinity of the outer periphery of the disk increases, so the Doppler shift increases. Further, since the rotation period is slow, the time window can be widened. Therefore, the frequency resolution Δf can be increased.

  However, since a large disk is used, it cannot be easily installed in various indoor places. In order to be easily installed in various indoor places, it is desired to be small.

  To increase the Doppler shift while reducing the size of the disk, the angular velocity should be increased. However, if the angular velocity is increased, the frequency analysis window cannot be widened. This is because the time window T has a relationship of T = N / fs (N is the number of sampling points, fs is the sampling frequency), and N is reduced when the angular velocity is increased. If the angular velocity is increased, the frequency analysis window cannot be widened, so that the frequency resolution is lowered.

  Instead of frequency analysis by Fourier transform, prepare a measurement signal model and change the parameter of the model, and use a method to determine the degree of coincidence with the measurement signal. Thus, the angular resolution can be increased.

  However, the distance between the antenna and the wireless tag changes continuously due to the rotation of the antenna, and the model of the measurement signal representing the Doppler shift caused by the change in the distance becomes a complicated model when expressed strictly. As a result, the amount of calculation increases. If the model is simplified using approximation, the amount of calculation is reduced, but if the approximation is not appropriate, the estimation accuracy of the radio wave arrival direction is lowered.

  The present invention has been made based on this circumstance, and the object of the present invention is the direction of arrival of radio waves that can be downsized, have high angular resolution, high accuracy, and can reduce the amount of calculation. To provide an estimation device and a radio wave arrival direction estimation system.

  The above object is achieved by a combination of the features described in the independent claims, and the subclaims define further advantageous embodiments of the invention. Reference numerals in parentheses described in the claims indicate a correspondence relationship with specific means described in the embodiments described later as one aspect, and do not limit the technical scope of the present invention. .

The present invention for achieving the above object is a radio wave arrival direction estimation device that estimates the arrival direction of a radio wave having a preset constant frequency transmitted by the wireless tag (300),
A turntable (112), a drive unit (113) that rotates the turntable at a predetermined period, and a radio wave that is fixed to a position other than the center of rotation on the turntable and transmitted by the wireless tag is received. An antenna (111), and a receiving unit (100) that outputs a measurement signal that is a signal determined based on radio waves received by the antenna;
A model that represents a measurement signal using an approximation in which the radio wave transmitted by the wireless tag is a plane wave, and the deviation square of the difference between the approximate model including the azimuth angle and phase at which the radio wave arrives as an unknown parameter and the measurement signal The deviation sum of squares corresponding value, which is one of the values that change depending on the sum or the deviation sum of squares, is calculated while changing the unknown parameters of the approximation model, and the approximation model that matches the measurement signal is calculated based on the deviation sum of squares. And a direction determining unit (230) that determines an azimuth angle in the determined approximate model as an azimuth angle at which radio waves arrive.

  In the present invention, the rotating disk can be downsized for the following reason. The frequency transmitted by the wireless tag is set to a constant frequency, but naturally, a certain amount of fluctuation occurs in the transmitted frequency in the actual device. If the magnitude of the Doppler shift caused by the rotation of the antenna is about the same as the fluctuation range of the frequency transmitted by the wireless tag, the frequency fluctuation caused by the Doppler shift cannot be distinguished from the fluctuation of the frequency transmitted by the wireless tag. Therefore, regardless of the frequency analysis method, the Pseudo-doppler method needs to move the antenna at a speed at which a certain amount of Doppler shift occurs.

  If the turntable is rotated at high speed, the window width becomes narrow. However, in the present invention, it is not Fourier transform, but is either the deviation sum of squares of the difference between the measurement signal output from the receiving unit and the approximate model obtained by modeling the measurement signal, or a value that changes in accordance with the deviation sum of squares. The azimuth angle from which the radio wave arrives is determined using the deviation sum of squares corresponding value. Therefore, it is possible to increase the Doppler shift by rotating the rotating disk at high speed. That is, it is not necessary to enlarge the rotating disk in order to increase the speed of the antenna fixed to the rotating disk. Therefore, it is possible to reduce the size of the turntable. Further, in order to increase the angular resolution, it is only necessary to narrow the pitch for numerical search, so it is easy to increase the angular resolution.

  In the present invention, the azimuth angle can be accurately estimated for the following reason while reducing the amount of calculation. In the present invention, since the antenna rotates and repeats approaching and separating from the wireless tag, Doppler shift occurs in the radio wave received by the antenna. Therefore, the frequency of the radio wave received by the antenna increases with the rotational movement of the antenna. fluctuate. For this reason, if the radio wave received by the antenna is modeled strictly, a complicated model is generated and the amount of calculation increases.

  Therefore, in the present invention, an approximate model representing the measurement signal is considered. The radio wave transmitted by the wireless tag is actually a spherical wave, but the approximate model according to the present invention obtains the radio wave transmitted by the wireless tag by approximating it as a plane wave.

  The approximation as a plane wave is that the wireless tag is assumed to exist in the same direction with respect to the antenna regardless of the position of the antenna. As the distance from the antenna to the wireless tag is longer than the rotation radius of the antenna, the direction change of the wireless tag relative to the antenna is smaller even if the antenna rotates. In the present invention, as described above, the turntable can be reduced in size. If the turntable is small, the radius of rotation of the antenna is also small. If the turning radius of the antenna is reduced, the distance from the antenna to the wireless tag is likely to be longer than the turning radius of the antenna. Therefore, the approximation to the plane wave can estimate the azimuth angle with an accuracy close to that when the radio wave received by the antenna is strictly modeled in the present invention in which the rotating disk can be downsized. Since the azimuth angle can be estimated with an accuracy close to that when strictly modeled, the approximation using a plane wave can accurately estimate the azimuth angle in the present invention in which the rotating disk can be downsized.

  An approximate model using approximation as a plane wave will be described in detail later, but there is no square root that exists when it is modeled strictly. Therefore, the amount of calculation can be greatly reduced as compared with the case of strict modeling.

  Furthermore, in the present invention, as a value for evaluating the coincidence between the approximate model and the measurement signal, the deviation sum of squares that is either the deviation sum of squares of the difference between the approximate model and the measurement signal or a value that changes in accordance with the sum of deviation squares. Use the corresponding value. The reason for using the deviation sum of squares value is to evaluate the coincidence between the approximate model and the measurement signal with a value that is not easily affected by the fluctuation of the DC offset component included in the measurement signal.

  Here, unlike the present invention, it is assumed that the amount of the DC offset component included in the measurement signal is measured in advance and the DC offset component is removed from the measurement signal. The magnitude of the DC offset component varies depending on environmental change factors such as temperature. For this reason, when removing the DC offset component from the measurement signal, it is necessary to dynamically change the amount of the DC offset component to be removed at certain time intervals such as time intervals at which the environmental temperature cannot be considered constant. However, it is difficult to dynamically change the amount of the DC offset component to be removed with high accuracy. In addition, it is conceivable to add this DC offset component to the unknown parameter of the approximate model, but in this case, the amount of calculation increases due to an increase in the unknown parameter.

  However, the DC offset component does not fluctuate significantly as long as it is a time sufficient to determine one set of estimated values of the arrival direction of radio waves. Therefore, as in the present invention, by using the value representing the deviation sum of squares as a value for evaluating the coincidence between the approximate model and the measurement signal, the DC offset component included in the measurement signal varies depending on the environmental change factor. The azimuth angle can be determined with high accuracy. Moreover, since the DC offset component is not used as an unknown parameter, an increase in the amount of computation can be suppressed.

In the invention according to claim 2, the receiving unit generates a low frequency signal in which the frequency of the radio wave received by the antenna is reduced so that the center frequency is lower than the maximum Doppler shift determined by the rotation speed of the antenna. A signal generator (120) is provided to output a low frequency signal as a measurement signal,
The approximate model is a model that represents a low-frequency signal generated by the low-frequency signal generation unit using an approximation in which the radio wave transmitted by the wireless tag is a plane wave.
The direction determining unit changes the unknown parameter of the approximate model as a change range of 180 degrees or less with respect to the azimuth, and the antenna rotates more than 180 degrees by the deviation sum of squares corresponding value for the approximate model and the low-frequency signal. Calculate over the interval to determine the approximate model that matches the low frequency signal.

  In the present invention, the low frequency signal generator generates the low frequency signal. Since the low frequency signal has a lower center frequency than the maximum Doppler shift determined by the rotation speed of the antenna, the frequency may be negative.

  By approximating the radio wave transmitted by the wireless tag to a plane wave, a phase error occurs when the approximate model is not approximated. Here, since the frequency fluctuates due to the Doppler shift due to the rotation of the antenna, the state where the frequency is negative can be regarded as a state where the rotation direction of the antenna is reversed with respect to the state where the frequency is positive.

  The phase error generated in the approximate model is caused by a difference in distance between the distance from the wireless tag to the center of rotation of the antenna and the distance from the wireless tag to the antenna. Therefore, if the rotation direction of the antenna is reversed, a phase error also occurs in the opposite direction.

  Since the phase errors are opposite to each other when the frequency is positive and when the frequency is negative, the interval for calculating the deviation sum of squares includes the interval where the frequency is positive and the interval where it is negative In this case, the phase error can be canceled out.

  The sign of the frequency is reversed when the antenna, the rotation center of the antenna, and the wireless tag are aligned. Therefore, every time the antenna rotates 180 degrees, the positive and negative frequencies are reversed.

  Since the direction determining unit of the present invention sets the section for calculating the deviation sum of squares corresponding to the section where the antenna rotates more than 180 degrees, the phase error when the frequency is positive and the phase when the frequency is negative The error will be offset. Therefore, an approximate model with a small phase error with respect to the actual measurement signal can be determined as an approximate model that matches the measurement signal. As a result, the accuracy of the azimuth angle determined from the approximate model is further improved.

In the invention according to claim 3, the receiving unit is an I component of a low-frequency signal obtained by lowering the frequency of the radio wave received by the antenna so that the center frequency is lower than the maximum Doppler shift determined by the rotation speed of the antenna. A low-frequency signal generation unit (120) that generates a Q component signal that is a Q component of the I component signal and the low frequency signal, and outputs the I component signal and the Q component signal as a measurement signal;
The approximation model is an I component signal generated by the low frequency signal generation unit using an approximation in which the radio wave transmitted by the wireless tag is a plane wave, an I component approximation model representing the Q component signal, and a Q component approximation model.
The direction determining unit changes the unknown parameter of the I component approximate model and the Q component approximate model as a change range wider than 180 degrees with respect to the azimuth, and the deviation square sum corresponding value for the I component approximate model and the I component signal, And the deviation sum of squares corresponding values for the Q component approximate model and the Q component signal are calculated over the section where the antenna rotates more than 180 degrees, and the I component approximate model and the Q component signal matching the I component signal are calculated. A matching Q component approximate model is determined.

  The invention according to claim 3 divides the low frequency signal in the invention according to claim 2 into an I component signal and a Q component signal. The approximate models are also an I component approximate model and a Q component approximate model. Then, the direction determining unit calculates a deviation square sum corresponding value for the I component signal and the I component approximate model, and a deviation square sum corresponding value for the Q component signal and the Q component approximate model.

  The reason for calculating the deviation sum of squares corresponding to the I component signal and the Q component signal for the I component signal and the Q component signal is that the azimuth angle, which is an unknown parameter in the I component approximate model and the Q component approximate model, is 180. It is because it changes over a wider range than the degree. That is, the azimuth angle search range is wider than 180 degrees.

  As will be described in detail later, the I component approximation model and the Q component approximation model may have the same waveform as a waveform having a different azimuth angle and a phase difference. For this reason, when the azimuth angle search range is wider than 180 degrees, the azimuth angle cannot be determined using only one of the I component approximate model and the Q component approximate model.

  However, if one of the I component approximation model and the Q component approximation model substitutes a parameter that becomes the same waveform as the waveform having an azimuth angle different by 180 degrees in the other approximation model, the azimuth angle in the other approximation model is Two waveforms that are 180 degrees different from each other are different waveforms. Therefore, even if the azimuth angle search range is wider than 180 degrees, the azimuth angle can be determined by using the two models of the I component approximate model and the Q component approximate model.

In the invention of claim 4, the low frequency signal generation unit generates a low frequency signal having a center frequency of 0 Hz,
The direction determining unit sets a section for calculating the deviation sum of squares corresponding value as a section in which the antenna rotates 360 degrees.

In the invention of claim 5, the low frequency signal generation unit generates an I component signal and a Q component signal having a center frequency of 0 Hz,
The direction determining unit sets a section for calculating the deviation sum of squares corresponding value as a section in which the antenna rotates 360 degrees.

  According to the fourth and fifth aspects of the present invention, the low-frequency signal generator generates a low-frequency signal or an I component signal and a Q component signal with a center frequency of 0 Hz. When the center frequency is 0 Hz, the absolute value of the phase error in the positive frequency section of the approximate model and the phase error in the negative frequency of the approximate model are equal to each other with the opposite sign while the antenna rotates once. . Since the section for calculating the deviation sum of squares corresponding value is the section in which the antenna rotates 360 degrees, the phase error can be canceled with high accuracy. Therefore, the accuracy of the azimuth angle determined from the approximate model is further improved.

方向決定部は、サンプリング番号をｋ、総サンプリング数をＫ、測定信号と近似モデルとの差をＶ_ｋ、差の平均値をＶバーとしたとき、式６で表す偏差平方和ｅが最小となる複数の到来波の方位角および仰角の組み合わせを探索するものであって、

式６をＢで偏微分した式と、式６をＣで偏微分した式をそれぞれ０とすることで立式される連立方程式を解くことによりＢ、Ｃを算出し、算出したＢ、Ｃを用いて、偏差平方和ｅが最小となる複数の到来波の方位角および仰角の組み合わせを探索する。 In the invention according to claim 6, the approximate model is a model in which a synthesized wave of a plurality of incoming waves is converted into a low frequency signal, the frequency of the radio wave transmitted by the wireless tag is f _RF , the radius of rotation of the antenna is R, The speed of light is v _c , the time is t, the azimuth angle of each incoming wave is φ _m , the elevation angle of each incoming wave is δ _m , the amplitude of each incoming wave is A _m , the phase of each incoming wave is Ψ _m , and the number of incoming waves _Is N and the approximate model is V _ref , the model is represented by Formula 1 or Formula 2, Formula 3, Formula 4, and Formula 5,

When the sampling number is k, the total number of samplings is K, the difference between the measurement signal and the approximate model is V _k , and the average value of the differences is V bar, the direction determination unit has a minimum deviation square sum e expressed by Equation 6 as follows. Search for a combination of azimuth and elevation angles of a plurality of incoming waves,

B and C are calculated by solving simultaneous equations formed by partial differentiation of Equation 6 by B and partial differentiation of Equation 6 by C to 0, and calculating the calculated B and C. The combination of azimuth angles and elevation angles of a plurality of incoming waves with the smallest deviation sum of squares e is searched.

  According to the sixth aspect of the present invention, since B and C having the amplitude A and the phase Ψ as parameters can be calculated by simultaneous equations, the amplitude A and the phase Ψ of the incoming wave can be excluded from the unknown parameters in Expression 6. Therefore, since there are fewer unknown parameters that need to be searched, the calculation can be performed quickly.

方向決定部は、サンプリング番号をｋ、総サンプリング数をＫ、Ｉ成分信号とＩ成分近似モデルとの差をＩ_ｋ、Ｉ成分信号とＩ成分近似モデルとの差の平均値をＩバー、Ｑ成分信号とＱ成分近似モデルとの差をＱ_ｋ、Ｑ成分信号とＱ成分近似モデルとの差の平均値をＱバーとしたとき、式１２で表す偏差平方和ｅが最小となる複数の到来波の方位角および仰角の組み合わせを探索するものであって、

式１２をＢで偏微分した式と、式１２をＣで偏微分した式をそれぞれ０とすることで立式される連立方程式を解くことによりＢ、Ｃを算出し、算出したＢ、Ｃを用いて、偏差平方和ｅが最小となる複数の到来波の方位角および仰角の組み合わせを探索する。 In the invention according to claim 7, the I component approximation model and the Q component approximation model are models obtained by converting the I component and the Q component of a composite wave of a plurality of incoming waves into low frequency signals, respectively, and transmitted by the wireless tag The frequency of the radio wave to be transmitted is f _RF , the radius of rotation of the antenna is R, the speed of light is v _c , the time is t, the azimuth angle of each incoming wave is φ _m , the elevation angle of each incoming wave is δ _m , and the amplitude of each incoming wave is A _m , the phase of each incoming wave is Ψ _m , the number of incoming waves is N, the I component approximate model is I _ref , and the Q component approximate model is Q _ref , the I component approximate models are Equations 7, 9, and 10 , And the Q component approximation model is a model represented by Equation 8, Equation 9, Equation 10, and Equation 11,

The direction determining unit sets the sampling number to k, the total number of samplings to K, the difference between the I component signal and the I component approximate model to I _k , the average value of the difference between the I component signal and the I component approximate model to I bar, Q When the difference between the component signal and the Q component approximate model is Q _k , and the average value of the difference between the Q component signal and the Q component approximate model is Q bar, a plurality of arrivals in which the deviation sum of squares e expressed by Equation 12 is minimized Searching for a combination of azimuth and elevation of a wave,

B and C are calculated by solving simultaneous equations formed by partial differentiation of Equation 12 with B and equations obtained by partial differentiation of Equation 12 with C to 0, and calculate the calculated B and C. The combination of azimuth angles and elevation angles of a plurality of incoming waves with the smallest deviation sum of squares e is searched.

  In the invention according to claim 7 as well, since B and C can be calculated by simultaneous equations, the amplitude A and the phase Ψ of the incoming wave can be excluded from the unknown parameters in Expression 12. Therefore, since there are fewer unknown parameters that need to be searched, the calculation can be performed quickly.

  In the invention according to claim 8, the direction determining unit calculates a value calculated by an expression obtained by removing the term of only the measurement signal from the expression for calculating the deviation square sum e as the deviation square sum corresponding value.

  In this way, it is possible to search for a combination of azimuth angles and elevation angles of a plurality of incoming waves that minimize the deviation sum of squares e without calculating the term of only the measurement signal, so that the amount of calculation can be reduced.

  In the ninth aspect of the invention, the direction determining unit calculates the deviation square sum corresponding value by an equation obtained by removing the term of only the I component signal and the term of only the Q component signal from the equation for calculating the deviation square sum e. Calculate the value.

  According to the ninth aspect of the present invention, it is possible to search for a combination of azimuth angles and elevation angles of a plurality of incoming waves that minimize the deviation sum of squares e without calculating a term of only the measurement signal.

In the invention of claim 10, the calculation is performed by inputting a plurality of azimuth angles and times into a z-factor term which is a part of an equation for calculating the deviation sum of squares corresponding value and can be calculated by determining z. a storage unit (220) that stores a calculated value of the z-factor term;
The direction determination unit calculates a deviation sum-of-squares corresponding value using the calculated value of the z-factor term stored in the storage unit.

  In this way, if the value corresponding to the deviation sum of squares is calculated using the calculated value of the z-factor term, the amount of computation when searching for the azimuth and elevation is reduced, so the azimuth and elevation can be estimated quickly. it can.

  An eleventh aspect of the invention is a radio wave arrival direction estimation system including the radio wave arrival direction estimation device according to any one of claims 1 to 10 and a wireless tag.

It is a block diagram of the radio wave arrival direction estimation system of the embodiment. 6 is a diagram illustrating a relative position between an antenna 111 and a wireless tag 300. FIG. It is a figure explaining plane wave approximation. It is a figure showing the plane containing the s-axis and z-axis of FIG. It is a figure which compares and shows the waveform of an approximate model, and the waveform without approximation. It is a figure explaining the reason for which the phase of an approximate model advances. It is a figure which compares the waveform of an I component approximation model, and the I component signal of the waveform without approximation. It is a figure which compares the waveform of a Q component approximation model, and the Q component signal of the waveform without approximation. FIG. 10 is a diagram illustrating a configuration of a receiving unit 100 in Modification 1.

  Hereinafter, embodiments of the present invention will be described with reference to the drawings. The radio wave arrival direction estimation system according to this embodiment includes the wireless tag reader 1 and the wireless tag 300 shown in FIG. The wireless tag reader 1 functions as a radio wave arrival direction estimating device.

The wireless tag 300 transmits an unmodulated wave having a predetermined carrier frequency f ₀ set in advance. The wireless tag 300 is of an active type and may transmit radio waves continuously, but it is preferable to transmit radio waves intermittently from the viewpoint of battery life. The wireless tag 300 is carried by a person and has a size that can be easily accommodated in a pocket of clothes.

  The wireless tag reader 1 includes a reception unit 100 and a signal processing unit 200, and the reception unit 100 includes an antenna unit 110 and a low frequency signal generation unit 120.

[Description of Antenna Unit 110]
The antenna unit 110 includes an antenna 111, a turntable 112, and a drive unit 113. The antenna 111 is fixed to the outer peripheral edge of the turntable 112. There is no particular limitation on the shape and size of the antenna 111 as long as it can receive an unmodulated wave transmitted by the wireless tag 300 and can be fixed to a position other than the rotation center on the turntable 112.

  The turntable 112 is rotated by the drive unit 113. The shape of the rotating disk 112 is not limited to the disk shape, but is preferably not eccentric with respect to the driving unit 113. The turntable 112 is sized so that it can easily be set indoors. For example, a disk having a diameter of 10 cm. When the turntable 112 rotates, the antenna 111 fixed on it also rotates at the same time.

  The drive unit 113 includes a motor, and rotates the turntable 112 at a constant cycle. This fixed period is determined from the rotation speed of the antenna 111 determined from the Doppler shift to be secured and the rotation radius of the antenna 111.

[Description of Low Frequency Signal Generation Unit 120]
The low frequency signal generation unit 120 includes a band pass filter 121, a local oscillator 122, a mixer 123, a low pass filter 124, an A / D converter 125, a phase shifter 126, a mixer 127, a low pass filter 128, and an A / D converter 129. I have.

  The bandpass filter 121 uses a frequency range determined from a Doppler shift generated by rotation of the antenna 111 around the frequency of the radio wave transmitted by the wireless tag 300 as a pass frequency band. The band-pass filter 121 receives a reception signal received by the antenna 111 and removes noise from the reception signal. In addition, on the transmission path where the received signal is sent from the antenna 111 to the band pass filter 121, between one end of the rotation side transmission path rotating with the turntable 112 and one end of the non-rotation side transmission path facing the one end, The antenna patterns are opposed to each other and transmitted by radio. However, a slip ring may be used instead of wireless transmission.

The local oscillator 122 generates a local oscillation signal having the same frequency as the carrier frequency f ₀ transmitted by the wireless tag 300. The mixer 123 mixes the local oscillation signal and the signal output from the band-pass filter 121, and outputs a signal having a frequency that is the sum and difference of the frequency of the local oscillation signal and the frequency output from the band-pass filter 121. .

The low-pass filter 124 extracts a signal having a frequency that is the difference between the frequency of the local oscillation signal and the frequency output from the band-pass filter 121 from the signal output from the mixer 123. Since the frequency of the local oscillation signal is the same as the carrier frequency f ₀ transmitted by the wireless tag 300, the signal extracted by the low-pass filter 124 has a center frequency of 0 Hz. Hereinafter, the signal output from the low-pass filter 124 is referred to as an I component signal. The A / D converter 125 converts the I component signal that is an analog signal extracted by the low-pass filter 124 into a digital signal.

  The phase shifter 126 shifts the phase of the local oscillation signal by 90 °. The mixer 127 mixes the local oscillation signal whose phase is shifted by 90 ° by the phase shifter 126 and the signal output from the bandpass filter 121. The low pass filter 128 extracts a signal having a frequency that is the difference between the frequency of the local oscillation signal and the frequency output from the band pass filter 121 from the signal output from the mixer 127. However, the signal output from the low-pass filter 128 is 90 ° out of phase with the I component signal. A signal output from the low-pass filter 128 is hereinafter referred to as a Q component signal. This Q component signal also has a center frequency of 0 Hz. The A / D converter 129 converts the Q component signal, which is an analog signal extracted by the low-pass filter 128, into a digital signal. The I component signal and the Q component signal output from the A / D converters 125 and 129 correspond to the low frequency signal and the measurement signal in the claims.

[Description of Signal Processing Unit 200]
The signal processing unit 200 includes a signal acquisition unit 210, a storage unit 220, and a direction determination unit 230. The signal acquisition unit 210 acquires I component signals and Q component signals from the A / D converters 125 and 129, and stores the acquired signals in the storage unit 220.

  The wireless tag 300 transmits an unmodulated wave. However, the distance between the antenna 111 and the wireless tag 300 changes as the turntable 112 rotates. Therefore, the frequency of the radio wave received by the antenna 111 varies. Therefore, the frequency of the I component signal and the Q component signal acquired by the signal acquisition unit 210 also varies.

  The storage unit 220 stores a formula for calculating E shown in the formula 70 described later, which is a value corresponding to the sum of deviations of the claims. In addition, the storage unit 220 also stores calculated values calculated in advance by changing the time t, the azimuth angle φ, and the elevation angle δ for the variable S used in the equation for calculating E.

  E is a value calculated by an expression obtained by removing the term of only the measurement signal from the expression for calculating the deviation sum of squares e. The deviation sum of squares e is a deviation sum of squares calculated for a residual that is a difference between the measurement signal and an approximation model obtained by modeling the measurement signal by plane wave approximation.

  In the present embodiment, the measurement signal is specifically an I component signal and a Q component signal. Therefore, the approximate model is also an I component approximate model obtained by modeling the I component signal and a Q component signal. It is a Q component approximation model. First, the I component approximate model and the Q component approximate model will be described.

[Explanation of I component approximate model and Q component approximate model]
In order to describe the I component approximate model and the Q component approximate model, it is first necessary to describe a model (hereinafter, a strict model) in which the received signal received by the antenna 111 is not approximated.

FIG. 2 is a diagram illustrating the relative position between the antenna 111 and the wireless tag 300. As shown in FIG. 2, in the following description, it is assumed that the radio wave transmitted by the wireless tag 300 has a frequency f _RF , an amplitude A, and a phase Ψ _T. The rotation angular velocity of the antenna 111 is ω, and the rotation angle is θ. If time t is used, θ = ωt. Further, the azimuth angle of the wireless tag 300 with respect to the reference azimuth is φ, and the angle formed by the line connecting the center position of the turntable 112 and the wireless tag 300 with respect to the plane including the turntable 112 is the elevation angle δ. Let radius be R.

The distance between the wireless tag 300 and the center of the turntable 112 is L ₀ , the distance between the wireless tag 300 and the antenna 111 is _LR, and the position of the wireless tag 300 is P _t (x _t , y _t , z _t ), the center position of the turntable 112 is P ₀ (x ₀ , y ₀ , z ₀ ), and the position of the antenna 111 is P _R (x _R , y _R , z _R ).

If the initial angle of the antenna 111 and a x-axis direction, the position P _R of the antenna 111 can be expressed by the following equation 13.

Further, the position P _t of the wireless tag 300 can be expressed by Expression 14 using the center position P ₀ (x ₀ , y ₀ , z ₀ ), the elevation angle δ, and the azimuth angle φ of the turntable 112.

Distance _{L R} between the wireless tag 300 and the antenna 111 can be expressed by Equation 15. By substituting Equations 13 and 14 into Equation 15 and rearranging, Equation 16 is obtained.

Since Equation 16 is obtained when the velocity of light and _{v C,} the received signal _{V R} by the antenna 111 receives can be expressed by Equation 17.

Usually, the frequency of radio waves is high because it radiates into the air. Therefore, since the high frequency of the received signal V _R, lowering the frequency and local signals and mixing. Frequency _{f LO,} the mixing with the signal of the phase [psi _LO, the received signal _{V R} after mixing the formula 18. Equation 18 is an exact model.

_LR increases or decreases as the antenna 111 moves circularly. Thus, from Equation 18, the frequency of the received signal V _R after mixing, it can be seen that vary with time. Therefore, in the Fourier transform methods require a certain amount of window width to do good analysis accuracy can not be analyzed accurately received signal V _R after mixing.

Therefore, in this embodiment, the azimuth angle φ and the elevation angle δ are estimated by model matching. However, exact model shown in Equation 18 includes a distance L _R between the wireless tag 300 and the antenna 111, the distance L _R, as shown in Equation 15, the entire expression is in the square root. Therefore, using the exact model of Equation 18 requires complex calculations. In the present embodiment, in order to simplify the calculation, an approximate model is used in which the radio wave transmitted by the wireless tag 300 is approximated as a plane wave and the exact model shown in Expression 18 is approximated.

  Considering that the radio wave transmitted by the wireless tag 300 is a plane wave, the radio wave received by the antenna 111 is parallel to the radio wave from the wireless tag 300 toward the rotation center of the antenna 111 regardless of the position of the antenna 111. Can think.

In this case, as shown in FIG. 3, the radio wave reaches the antenna 111, to the radio wave from the wireless tag 300 to the rotational center of the antenna 111 and transmitted from the approximate wave source 300a on the plane perpendicular P _L Can be considered.

In FIG. 3, L _R ′ is a distance from the approximate radio wave source 300a to the antenna 111. S is an axis representing the direction of the azimuth angle φ. FIG. 4 is a diagram showing a plane including the s-axis and the z-axis.

4 that the approximate distance L _R ′ from the approximate radio wave source 300a to the antenna 111 can be expressed by Equation 19.

_'Is used in place of L _R, i.e., the distance L _R approximate distance L _R for formula _18' approximate distance L _R shown in equation 19 by substituting, further summarized phase as [psi ', has the formula 20 can get. Equation 20 is hereinafter referred to as an approximate model.

  Unlike Equation 17, Equation 20 has a simple shape with no square root. In the present embodiment, an I component approximate model and a Q component approximate model derived based on the approximate model shown in Expression 20 are used.

  The reason why the I component approximate model and the Q component approximate model are used is that the approximate model of Expression 20 causes a phase error with respect to the actual received waveform. FIG. 5 shows a comparison between a waveform of an approximate model in which appropriate parameters are set and a waveform without approximation, that is, a waveform of an exact model.

  Compared to the waveform without approximation, the waveform of the approximate model is advanced in phase. The reason why the waveform phase of the approximate model advances is that, as shown in FIG. 6, the distance until the radio wave reaches is short in the approximate model.

6, the difference between the distance L _R and the approximate distance L _{R 'between} the wireless tag 300 and the antenna 111, the larger the perpendicular to the DOA angle of the antenna 111. In addition, the maximum value of the difference is larger as the wireless tag 300 is closer to the antenna 111, and is larger as the rotation radius R of the antenna 111 is larger. Therefore, the error between the azimuth angle φ and the elevation angle δ of the wireless tag 300 obtained using the approximate model is larger as the angle of the antenna 111 is perpendicular to the direction of arrival of radio waves, and the wireless tag 300 is closer to the antenna 111. The larger the rotation radius R of the antenna 111, the larger the antenna 111 becomes.

  The I component approximate model and the Q component approximate model used in the present embodiment are models of signals obtained by reducing the center frequency of the approximate model of Expression 20 to 0 Hz and dividing it into an I component and a Q component.

In order to reduce the center frequency of the received signal V _R to 0Hz it may be mixed signals having the same frequency as the frequency f _RF of the radio tag 300 transmits the received signal V _R. An expression representing the I component of the signal after mixing is obtained by setting f _IF = 0 in Expression 18. Further, the Q component of the signal after mixing is orthogonal to the I component of the signal after mixing. Therefore, the equations representing the I component and Q component of the signal after mixing can be represented by Equation 21.

Since the phase ψ _T and the phase ψ _LO are fixed values, the equations representing the I component and the Q component of the signal after mixing can be expressed by Equation 22.

By substituting _LR ′ shown in Equation 19 for _LR in Equation 22, Equation 23 is obtained. Expression 23 is an I component approximate model and a Q component approximate model.

  FIG. 7 shows a comparison between the waveform of the I component approximate model corresponding to the waveform of the approximate model of FIG. 5 and the I component signal of the waveform without approximation. FIG. 8 shows a comparison between the waveform of the approximate Q model corresponding to the waveform of the approximate model of FIG. 5 and the Q component signal of the waveform without approximation. 7 and 8 are cases where the wireless tag 300 is present in the 0 degree direction.

  As shown in FIGS. 7 and 8, both the I component approximate model and the Q component approximate model have a phase error with respect to the waveform without approximation. However, unlike FIG. 5, in both the I component approximate model and the Q component approximate model, the phase is advanced or delayed with respect to the waveform without approximation.

  More specifically, from 0 degrees to 180 degrees, the waveforms of the I component approximate model and the Q component approximate model are delayed from the waveforms without approximation, and the phases are advanced from 180 degrees to 360 degrees.

  The reason is as follows. Since the center frequency is set to 0 Hz, when the antenna 111 is moving away from the wireless tag 300, the I component signal and the Q component signal have negative frequencies due to Doppler shift. In the section where the frequency is negative, the rotation direction is reversed from the section where the frequency is positive. Since the rotation direction is reversed, the phase shift direction is also opposite to the section where the positive frequency is obtained.

  In this embodiment, the center frequency is set to 0 Hz in order to cause both of the phase lag and the phase advance and cancel each other out of the phase lag and the phase advance.

[Reason for using two approximate models]
However, if the center frequency is 0 Hz, the azimuth angle φ may not be uniquely determined unless the measurement signal is divided into an I component signal and a Q component signal and the I component signal and the Q component signal are not matched together.

  This is because both the I component approximation model and the Q component approximation model may have the same waveform even if the azimuth angle φ is 180 degrees. This will be explained by changing the formula.

In Expression 23, when φ + π is substituted for φ and −Ψ is substituted for Ψ, Expression 24 is obtained.

Since cos (a−π) = − cos (a), Expression 24 can be transformed into Expression 25.

Since cos (−a) = cos (a) and sin (−a) = − sin (a), Expression 25 can be transformed into Expression 26.

  The I component approximation model in Equation 26 is the same as the I component approximation model in Equation 23. In other words, even if the azimuth angle φ is different by 180 degrees, the I component approximation model cannot be distinguished from each other if the phase ψ is reversed. However, since the Q component signal model in Equation 23 and the Q component signal model in Equation 26 are different, it can be seen from the Q component signal model that (φ, ψ) and (φ + π, −ψ) can be distinguished.

  However, the Q component approximation model cannot distinguish between (φ, ψ) and (φ + π, −ψ + π). This will be described next.

In Expression 23, when φ + π is substituted for φ and −Ψ + π is substituted for Ψ, Expression 27 is obtained.

Since cos (a−π) = − cos (a), Expression 27 can be transformed into Expression 28.

Since cos (a + π) = − cos (a) and sin (a + π) = − sin (a), Expression 28 can be transformed into Expression 29.

Since cos (−a) = cos (a) and sin (−a) = − sin (a), Expression 29 can be transformed into Expression 30.

  The Q component approximation model in Equation 30 is the same as the Q component approximation model in Equation 23. Therefore, the Q component approximate model cannot distinguish between (φ, ψ) and (φ + π, −ψ + π). However, since the I component signal model in Equation 23 and the I component signal model in Equation 30 are different, it can be seen from the I component signal model that (φ, ψ) and (φ + π, −ψ + π) can be distinguished.

  As described above, the I component approximation model and the Q component approximation model of Equation 23 cannot be distinguished from the case where the azimuth angle φ is shifted by 180 degrees by itself, and therefore, when searching over an angle range wider than 180 degrees. Requires both an I component approximation model and a Q component approximation model.

[Explanation of approximate model of synthesized wave]
The approximate model shown in Equation 30 is a single wave model. In an environment where the wireless tag reader 1 estimates the arrival direction of radio waves, it is assumed that multipath occurs. Therefore, it is necessary to consider an approximate model of a combined wave in which a plurality of incoming waves generated by multipath are combined.

Since the I component approximate model and the Q component approximate model of the composite wave are models obtained by adding a plurality of waves of the model shown in Expression 30 to each other, they can be expressed by Expression 31.

When Expression 31 is generalized with N as the number of incoming waves to be combined, Expression 32 is obtained.

[Explanation of residual sum of squared deviations]
In the present embodiment, the residual deviation sum of squares of the I component approximation model and the I component signal and the residual deviation sum of squares of the Q component approximation model and the Q component signal are considered. Specifically, a deviation sum of squares e obtained by adding the two deviation sums of squares is considered. The reason for considering this deviation sum of squares e is to remove the influence of the DC offset component included in the I component signal and the Q component signal.

  The DC offset component included in the I component signal and the Q component signal is generated due to temperature characteristics of various elements constituting the receiving unit 100. On the other hand, the DC offset component is not considered in the I component approximate model and the Q component approximate model. Therefore, it is necessary to remove the influence of the DC offset component and evaluate the coincidence between the I component signal and the I component approximate model and the coincidence between the Q component signal and the Q component approximate model. Even if there is a difference in absolute value between the I component approximate model and the I component signal and between the Q component approximate model and the Q component signal due to the influence of the DC offset, if the waveform shapes are similar to each other, The residuals are similar at any time. Accordingly, if the waveform shapes are similar to each other, the residual deviation sum of squares becomes small. Therefore, in the present embodiment, the degree of coincidence between these signals and the model is evaluated based on the sum of squared deviations of the residuals of the models.

  In order to make the residual deviation sum of squares e a simple calculation, first, the I component approximate model and the Q component approximate model are simplified.

When a part of Expression 32 is replaced with a variable z shown in Expression 33, Expression 34 is obtained.

Further, when the amplitude A and the phase Ψ are converted into the variable B _m and the variable C _m shown in Expression 35, a part of Expression 34 can be converted into Expression 36.

Using Expression 36, Expression 34 can be expressed as Expression 37.

In a continuous time system, the residual I _t between the I component signal I _get (t) and the I component approximate model I _ref (t) is expressed by Equation 38, and the Q component signal Q _get (t) and the Q component approximate model Q _ref The residual Q _t of (t) is expressed by Equation 39.

Therefore, in the continuous time system, the average I bar of the residual I _t is expressed by Equation 40, and the average Q bar of the residual Q _t is expressed by Equation 41. Here, t is a sampling time, T is a sampling period, in other words, a period for calculating the deviation sum of squares e.

From these equations 40 and 41, in the discrete time system, the deviation square sum e which is the sum of the deviation square sum of the residual I _{t and} the deviation square sum of the residual Q _t is expressed by the equation 42. Note that k and K correspond to t and T in the continuous time system, respectively, k is a sampling number, and K is the total number of samplings.

The approximate model in which the deviation sum of squares e shown in Equation 42 is the minimum represents the measured I component signal and Q component signal. As can be seen from the equations 38 to 41, the equation 42 is obtained by determining the I component approximate model I _ref (t) and the Q component approximate model Q _ref (t).

Then, by replacing the characters shown in Expression 35, unknown parameters in the I component approximate model I _ref and Q component approximate model Q _ref , that is, unknown parameters in Expression 42 are azimuth angle φ _m , elevation angle δ _m , B _m, the _{C m.}

Here, consider obtaining B _m and C _m . If the deviation sum of squares e is partially differentiated with respect to B _n and C _n of the n-th incoming wave, Expression 43 and Expression 44 are obtained.

In Equations 43 and 44, the partial differential term is a variable of only z, that is, a variable of only the azimuth angle φ and the elevation angle δ, which are arrival directions, as shown in Equations 45 and 46.

When Expression 43 is replaced by Expression 45, Expression 46, and Expression 38 to Expression 41, Expression 47 is obtained.

When Formula 47 is transformed using Formula 37, Formula 48 is obtained.

Rearranging Formula 48 for B _m and C _m gives Formula 49.

The same equation is modified for C. That is, when Formula 44 is replaced by Formula 45, Formula 46, and Formula 38 to Formula 41, Formula 50 is obtained.

When Formula 50 is transformed using Formula 37, Formula 51 is obtained.

When formula 51 is arranged for B _m and C _m , formula 52 is obtained.

In order to simplify Expression 49 and Expression 52 by replacing characters, the following variables shown in Expression 53 to Expression 58 are considered. The variable D is a variable that can be calculated if a measurement signal is obtained, and the variable S is a variable of z.

When Expression 49 and Expression 52 are replaced by the variables shown in Expression 53 to Expression 58, Expression 59 and Expression 60 are obtained.

Since we want to find B _m and C _m that minimize E, formulas with the left side of formula 59 and formula 60 set to 0 are considered. That is, Equation 61 is considered.

In Expression 61, 2N simultaneous equations are obtained for B _m and C _m for each of the n-th (n = 1 to N) incoming waves. By applying azimuth angles φ _m , φ _n , elevation angles δ _m , δ _n , z _m , z _n can be obtained. Therefore, when azimuth angles φ _m , φ _n , elevation angles δ _m , δ _n are applied, simultaneous equations of Formula 61 For each n, 2N equations are obtained for 2N unknown parameters B _m , C _m . Therefore, if the azimuth angles φ _m , φ _n , elevation angles δ _m , δ _n that are search values are applied, the simultaneous equations can be solved to calculate the unknown parameters B _m , C _m . Further, since Expression 62 is obtained from Expression 35, when B _m and C _m are obtained, the amplitude A _m and the phase Ψ _m can also be determined.

Since B _m and C _m can be calculated by solving the simultaneous equations of Equation 61 in this way, an approximate model in which E is minimized by simply searching for a combination of azimuth angle φ _m and elevation angle δ _m , that is, deviation square An approximate model that minimizes the sum e can be determined.

Furthermore, in the present embodiment, the following equation modification is performed in order to simplify the calculation for determining the approximate model that minimizes the deviation sum of squares e. When Formula 42 is expanded, Formula 63 is obtained.

When Expression 38, Expression 39, Expression 40, and Expression 41 are substituted into Expression 63, Expression 64 is obtained.

When Expression 64 is expanded and arranged, Expression 65 is obtained.

Among the terms shown in Expression 65, the term of only the I component signal I _get (t) and the term of only the Q component signal Q _get (t) are the values of the I component approximate model I _ref and the Q component approximate model Q _ref . Not affected. Therefore, consider E shown in Expression 66 from which the term of only the I component signal I _get (t) and the term of only the Q component signal Q _get (t) are excluded from the right side of Expression 65.

When E is minimized, the deviation sum of squares e is also minimized. Therefore, in this embodiment, an I component approximate model I _ref and a Q component approximate model Q _ref that minimize E are searched.

In the present embodiment, Expression 66 is used as it is, instead of searching for the I component approximate model I _ref and the Q component approximate model Q _ref that minimize E, the expression is further modified.

From Expression 37, Expression 67, Expression 68, and Expression 69 are obtained.

Substituting these equations 67, 68, and 69 into equation 66 yields equation 70.

The expression shown in Expression 70 is stored in the storage unit 220. Further, the equation 70 includes the variable S shown in the equation 71.

  The storage unit 220 also stores a table of calculated values calculated by variously changing the value of z for each variable S shown in Expression 71. This table is hereinafter referred to as a pre-calculation table. Each variable S shown in Formula 71 is a variable whose value can be calculated if z is determined, and corresponds to a z-factor term in the claims.

  As shown in Equation 33, z has time t, azimuth angle φ, and elevation angle δ as variables. In the pre-calculation table, the time t is changed by 2π / ω for each sampling period, and the azimuth angle ω and the elevation angle δ are changed at an angular pitch determined based on the required angular resolution, and the time t, For each combination of the azimuth angle ω and the elevation angle δ, a calculated value obtained by calculating the variable S shown in Expression 71 is included.

The direction determination unit 230 determines a combination of arrival directions (that is, an azimuth angle φ _m and an elevation angle δ _m ) that minimizes E, and this combination of arrival directions is an arrival direction in which radio waves actually arrive. To do. Specifically, the direction determination part 230 performs the process 1-the process 8 shown below, and determines the combination of the arrival direction from which E becomes the minimum.

  In step 1, the I component signal and the Q component signal for the interval in which E is calculated from the latest I component signal and Q component signal are acquired from the storage unit 220. In this embodiment, this section is a time section in which the antenna 111 rotates 360 degrees.

In step 2, the extracted I component signal and Q component signal are corrected as shown in Expression 72. In Expression 72, I _get ′ and Q _get ′ are the I component signal and the Q component signal extracted in step 1. _Ac is the non-contact portion amplitude ratio. The non-contact portion is between one end of the rotation-side transmission path on the transmission path of the antenna 111 and one end of the non-rotation-side transmission path facing the one end, and the non-contact portion amplitude ratio is the rotation-side transmission path It is the ratio of the amplitude of the signal at one end to the amplitude at one end of the non-rotation side transmission path. This non-contact portion amplitude ratio _Ac changes with time. The reason why the non-contact portion amplitude ratio _Ac changes with time is that there is a circuit or the like around the turntable 112, and therefore the radio wave environment of the noncontact portion varies depending on the rotational position of the turntable 112. The non-contact portion amplitude ratio _Ac is measured and set in advance.

  The reason why the calculation in step 2 is performed is that the I component approximate model and the Q component approximate model are models that do not consider the change in amplitude in the non-contact portion, and therefore the I component signal and Q component signal to be compared with these models. This is because it is preferable to remove the influence of the amplitude change in the non-contact portion.

  In step 3, for the incoming waves for N waves, a combination of arrival directions, that is, a combination of the azimuth angle φ and the elevation angle δ to be searched is determined. This combination may be determined in advance.

  In step 4, for each combination of arrival directions of incoming waves determined in step 3, calculated values are obtained from the pre-calculation table.

In Step 5, for each combination of arrival directions determined in Step 3, the simultaneous equations shown in Equation 61 are solved using the calculated values obtained in Step 2 and Step 4, and B _m , _Cm is determined.

  In step 6, using the calculated value obtained in step 4 and the calculated value calculated in steps 2 and 5, E shown in Equation 70 is calculated for each arrival direction of the incoming wave determined in step 3. In step 7, the minimum value of E calculated in step 6 is determined. In step 8, the combination of arrival directions corresponding to the minimum value of E determined in step 7 is set as the direction in which radio waves actually arrive. Of the directions in which radio waves actually arrive, for example, the direction having the maximum amplitude is the direction in which the wireless tag 300 is present.

In the present embodiment described above, the turntable 112 can be downsized. The reason is as follows. As already described, in the Pseudo-doppler method, it is necessary to rotate the antenna 111 at a speed at which a Doppler shift having a magnitude that can be distinguished from the fluctuation of the frequency f _RF transmitted by the wireless tag 300 occurs.

  If the turntable 112 is rotated at a high speed, the window width is narrowed. However, in this embodiment, instead of the Fourier transform, the deviation sum of squares of the residuals of the I component signal and the Q component signal output from the receiving unit 100 and the I component approximate model and the Q component approximate model respectively corresponding thereto are calculated. E representing the total value is calculated. That is, the azimuth angle φ and the elevation angle δ are obtained by a method that is not restricted by the window width. Therefore, since the rotating disk 112 can be rotated at a high speed to increase the Doppler shift, it is not necessary to increase the rotating disk 112 in order to increase the speed of the antenna 111 fixed to the rotating disk 112. Therefore, the rotating disk 112 can be downsized. Further, in order to increase the angular resolution, it is only necessary to narrow the pitch for numerical search, so it is easy to increase the angular resolution.

  Further, as already described, if the radio wave received by the antenna 111 is modeled strictly, the models 18 and 16 become complicated models, and the amount of calculation increases. Therefore, in this embodiment, an I component approximate model and a Q component approximate model are used. The radio wave transmitted by the wireless tag 300 is actually a spherical wave, but the I component approximate model and the Q component approximate model are obtained by approximating the radio wave transmitted by the wireless tag 300 as a plane wave.

The approximation as a plane wave is that the wireless tag 300 is considered to exist in the same direction with respect to the antenna 111 regardless of the position of the antenna 111. As the distance from the antenna 111 to the wireless tag 300 is longer than the rotation radius R of the antenna 111, the direction change of the wireless tag 300 relative to the antenna 111 is smaller even if the antenna 111 rotates. In the present embodiment, as described above, the turntable 112 can be reduced in size. If the turntable 112 is small, the rotation radius R of the antenna 111 is also small. The smaller the turning radius R of the antenna 111, compared to the rotation radius R of the antenna 111, the distance _{L R} from the antenna 111 to the wireless tag 300 is likely to be longer. Therefore, in the present embodiment in which the turntable 112 can be downsized, the approximation with a plane wave is less likely to be less accurate than when the radio wave received by the antenna 111 is modeled strictly. Since there is little decrease in accuracy with respect to the case of strict modeling, the approximation to the plane wave can estimate the radio wave arrival direction with high accuracy in the present embodiment in which the turntable 112 can be reduced in size.

  Further, the I component approximate model and the Q component approximate model of the present embodiment do not have a square root existing in the exact model as shown in Expression 23. Therefore, the amount of calculation can be greatly reduced as compared with the case where the strict model is used.

Further, in the present embodiment, E is used as a value for evaluating the coincidence between the approximate model and the I component signal I _get and the Q component signal Q _get , and the azimuth angle φ _m and the elevation angle _m in the approximate model that minimizes E are determined. To do.

This E is a value obtained by removing the term of only the I component signal I _{get and} the term of only the Q component signal Q _get from the deviation sum of squares e of the residual. Therefore, E also indicates the variation in the residual between the approximate model and the I component signal I _get or the Q component signal Q _get .

Since the DC offset component contained in the I component signal and the Q component signal hardly fluctuates in a short time, the influence of the DC offset component on the residual can be regarded as constant in the short time. Therefore, the DC offset component included in the I component signal and the Q component signal has little influence on E, which is a value indicating the variation of the residual. Therefore, by using E as a value for evaluating the degree of coincidence between the approximate model and the I component signal I _get or the Q component signal Q _get , the DC offset component included in the I component signal and the Q component signal is changed to the substrate temperature or the like. Even if it fluctuates due to the influence, the azimuth angle φ _m and the elevation angle δ _m can be accurately determined, and since the DC offset component is not used as an unknown parameter, an increase in the amount of calculation can be suppressed.

Further, in the present embodiment, B _m and C _m can be calculated by the simultaneous equations shown in Expression 61, so that the amplitude A _m and the phase Ψ _m of the incoming wave are excluded from the unknown parameters that need to be searched. it can. Therefore, since there are fewer unknown parameters that need to be searched, the calculation can be performed quickly.

Further, in the present embodiment, instead of calculating the deviation square sum e of the residual shown in Expression 65, a term of only the I component signal I _{get and} a term of only the Q component signal Q _get are calculated from the deviation square sum e. E, which is the removed value, is calculated. As a result, the azimuth angle φ _m and the elevation angle δ of a plurality of incoming waves that minimize the residual deviation square sum e without calculating the term of only the I component signal I _{get and} the term of only the Q component signal Q _{get. Since} the combination of _m can be searched, the amount of calculation can be reduced.

Furthermore, in the present embodiment, the calculated value of the variable S shown in Expression 71 calculated in advance by changing the time t, the azimuth angle φ _m , and the elevation angle δ _m is stored in the storage unit 220, and this calculated value is used. E is calculated. This also azimuth angle phi _m, since the amount of computation when estimating the elevation angle [delta] _m is reduced, the azimuth angle phi _m, the elevation angle [delta] _m can be quickly estimated.

  As mentioned above, although embodiment of this invention was described, this invention is not limited to the above-mentioned embodiment, The following modification is also contained in the technical scope of this invention, Furthermore, the summary other than the following is also included. Various modifications can be made without departing from the scope. In the following description, elements having the same reference numerals as those used so far are the same as the elements having the same reference numerals in the previous embodiments unless otherwise specified. Further, when only a part of the configuration is described, the above-described embodiment can be applied to the other parts of the configuration.

<Modification 1>
Instead of the local oscillator 122 of the above-described embodiment, the first modification includes a reference antenna 130 and a band-pass filter 131 as shown in FIG. The reference antenna 130 is fixed near the turntable 112. Further, it may be fixed to the rotation center of the turntable 112. This is because the position of the rotation center of the turntable 112 does not change even if the turntable 112 rotates.

  When the local oscillator 122 is used, the frequency oscillated by the local oscillator 122 and the frequency transmitted by the wireless tag 300 do not completely match. Therefore, while it is difficult to accurately set the center frequency of the I component signal and the Q component signal to 0 Hz, when the reference antenna 130 is used, the center frequency of the I component signal and the Q component signal is accurately set to 0 Hz. can do.

<Modification 2>
In the above-described embodiment, the section for calculating E is a section in which the antenna 111 rotates 360 degrees. However, this section may be a section in which the antenna 111 rotates more than 180 degrees.

  The positive and negative frequencies of the I component signal and the Q component signal are inverted when the antenna 111, the rotation center of the antenna 111, and the wireless tag 300 are aligned. Therefore, every time the antenna 111 rotates 180 degrees, the positive and negative frequencies of the I component signal and the Q component signal are inverted. In the examples of FIGS. 7 and 8, the positive and negative frequencies are inverted at 0 degrees and 180 degrees.

  If the section for calculating E is a section in which the antenna 111 rotates more than 180 degrees, both a section in which the frequency is positive and a section in which the frequency is negative are included. Therefore, at least a section in which E is calculated is a section in which the antenna 111 rotates more than 180 degrees, it is possible to cancel at least a part of the phase error.

<Modification 3>
In the above-described embodiment, two measurement signals of the I component signal and the Q component signal are used. The reason for using two measurement signals is that the I component approximate model and the Q component approximate model may have the same waveform even if the azimuth angle φ is 180 degrees different.

  Therefore, if the search range of the azimuth angle φ is 180 degrees or less, the azimuth angle φ and the elevation angle δ may be determined using only one of the I component signal and the Q component signal. In this case, of course, only one of the I component approximate model and the Q component approximate model is used. The I component approximate model or the Q component approximate model used at this time corresponds to the low frequency approximate model in the claims. Further, the I component signal or the Q component signal used at this time corresponds to the low frequency signal and the measurement signal in the claims.

  Further, since only one of the I component signal and the Q component signal is sufficient, it is not necessary to divide the signal into an I component signal and a Q component signal. When the signal is not divided into the I component signal and the Q component signal, the I component signal in the above-described embodiment is handled as the measurement signal V as it is.

Therefore, the approximate model V _ref of the measurement signal can be represented by the following formula 73 in which I in the I component approximate model shown in formula 37 is replaced with V.

Expression 37 is an expression obtained by replacing characters in the I component approximate model shown in Expression 34. As is well known, since sin and cos are only 90 degrees out of phase with each other and have the same shape, the approximate model V _ref of the measurement signal is considered as an expression of the Q component approximate model shown in Expression 34. You can also. In this case, the approximate model V _ref of the measurement signal can be expressed by the following formula 74 in which Q in the Q component approximate model shown in formula 37 is replaced with V.

When the approximate model V _ref shown in the equation 73 or 74 is used, the deviation sum of squares e is expressed by the equation 75 because the right side shown in the equation 42 is only the first term or the second term. The

<Modification 4>
In the above-described embodiment, the I component signal and the Q component signal are signals having a center frequency of 0 Hz, but the center frequency of the I component signal and the Q component signal may not be 0 Hz. However, it is preferably lower than the maximum Doppler shift. The maximum Doppler shift is a maximum value of the Doppler shift that is generated only by the rotation of the antenna 111 on the assumption that the wireless tag 300 is stationary. The maximum Doppler shift is a Doppler shift when the velocity vector of the antenna 111 is maximum in the direction toward the wireless tag 300 and maximum in the direction away from the wireless tag 300.

  If the center frequency of the I component signal and the Q component signal is lower than the maximum Doppler shift, a negative frequency is generated in the I component signal and the Q component signal. If a negative frequency occurs, the interval in which E is calculated can include both the interval in which the frequency is positive and the interval in which the frequency is negative. Therefore, the center frequencies of the I component signal and the Q component signal need not be 0 Hz as long as they are lower than the maximum Doppler shift.

When the center frequency is not 0 Hz, if the center frequency is f _IF , the I component approximate model and the Q component approximate model are models in which the term of 2πf _IF t is placed in parentheses outside Expression 23.

<Modification 5>
The shorter the distance L _R between the antenna 111 and the wireless tag 300 with respect to the rotation radius R of the antenna 111, the phase error by using an approximation model increases. In other words, conditions for determining the azimuth angle φ of the radio tag 300 is primarily, if the distance L _R is long state between the antenna 111 and the wireless tag 300 with respect to the rotation radius R of the antenna 111, the phase error azimuth The error given to the estimation accuracy of the angle φ is small. When the influence of the phase error is small, it is not necessary to lower the center frequency so that a negative frequency is generated.

  Therefore, the receiving unit 100 may output a measurement signal whose center frequency is higher than the maximum Doppler shift, and the direction determining unit 230 may use the approximate model of Expression 20 as an approximate model.

<Modification 6>
In the above-described embodiment, the elevation angle δ is also an unknown parameter, but the elevation angle δ is often not important when determining the direction of arrival of radio waves from the wireless tag 300 carried by a person. Therefore, the elevation angle δ may be constant, for example, 0 degrees. When the elevation angle δ is set to 0 degree, cos δ shown so far is set to 1.

<Modification 7>
Direction decision unit 230 of the above embodiment, I component signal extracted in the step 1 by a non-contact area amplitude ratio A _c in step 2, had been corrected Q component signal, must be omitted this step 2 If accurate direction estimation accuracy is obtained, step 2 may be omitted.

<Modification 8>
In the above embodiment, E is calculated as the deviation sum of squares corresponding value, but of course, the deviation sum of squares e may be calculated as the deviation sum of squares corresponding value. Further, a value obtained by dividing the deviation sum of squares e by the total number of samplings, that is, variance may be calculated as a value corresponding to the deviation sum of squares.

1: wireless tag reader, 100: receiving unit, 110: antenna unit, 111: antenna, 112: turntable, 113: driving unit, 120: low frequency signal generating unit, 121: bandpass filter, 122: local oscillator, 123: Mixer, 124: low-pass filter, 125: A / D converter, 126: phase shifter, 127: mixer, 128: low-pass filter, 129: A / D converter, 130: reference antenna, 131: band-pass filter, 200 : Signal processing unit, 210: signal acquisition unit, 220: storage unit, 230: azimuth angle determination unit, 300: wireless tag, 300a: approximate radio wave source

Claims (11)

A radio wave arrival direction estimation device for estimating the arrival direction of a predetermined frequency radio wave transmitted by the wireless tag (300),
A turntable (112), a drive unit (113) for rotating the turntable at a predetermined fixed period, and a radio wave transmitted by the wireless tag that is fixed at a position other than the center of rotation on the turntable. And a receiving unit (100) that outputs a measurement signal that is a signal determined based on radio waves received by the antenna;
A model representing the measurement signal using an approximation in which a radio wave transmitted by the wireless tag is a plane wave, and an approximate model including an azimuth angle and a phase at which the radio wave arrives as an unknown parameter, and the measurement signal A deviation sum of squares corresponding value, which is either a deviation sum of squares of differences or a value that changes according to the sum of deviations of squares, is calculated while changing an unknown parameter of the approximate model, and based on the deviation sum of squares, A direction determining unit (230) that determines the approximate model that coincides with a measurement signal, and determines the azimuth angle in the determined approximate model as the azimuth angle from which the radio wave has arrived. Radio wave arrival direction estimation device. In claim 1,
The reception unit generates a low-frequency signal generation unit (120) that generates a low-frequency signal in which a frequency of a radio wave received by the antenna is lowered so that a center frequency is lower than a maximum Doppler shift determined by a rotation speed of the antenna. ), And output the low frequency signal as the measurement signal,
The approximate model is a model that represents a low-frequency signal generated by the low-frequency signal generation unit using an approximation in which a radio wave transmitted by the wireless tag is a plane wave,
The direction determination unit changes the unknown parameter of the approximate model as a change range of 180 degrees or less with respect to the azimuth, and calculates the deviation square sum corresponding value for the approximate model and the low-frequency signal as the antenna. A radio wave arrival direction estimation apparatus, wherein the approximate model that is calculated over a section rotating more than 180 degrees and coincides with the low frequency signal is determined. In claim 1,
The receiving unit includes an I component signal that is an I component of a low frequency signal obtained by reducing the frequency of the radio wave received by the antenna so that the center frequency is lower than the maximum Doppler shift determined by the rotation speed of the antenna, and A low frequency signal generator (120) for generating a Q component signal that is a Q component of the low frequency signal, and outputting the I component signal and the Q component signal as the measurement signal;
The approximation model uses an approximation in which the radio wave transmitted by the wireless tag is a plane wave, and the I component signal generated by the low frequency signal generator and the I component approximation model and Q component approximation respectively representing the Q component signal. Model
The direction determining unit changes the unknown parameter of the I component approximate model and the Q component approximate model as a change range wider than 180 degrees with respect to the azimuth angle, and changes the I component approximate model and the I component signal. The deviation sum of squares corresponding value, and the deviation sum of square correspondence value of the Q component approximate model and the Q component signal are calculated over a section where the antenna rotates more than 180 degrees, and the I component signal A radio wave arrival direction estimation apparatus, wherein the I component approximate model that matches the Q component and the Q component approximate model that matches the Q component signal are determined. In claim 2,
The low frequency signal generation unit generates the low frequency signal having a center frequency of 0 Hz,
The radio wave arrival direction estimation device, wherein the direction determining unit sets a section for calculating the deviation sum of squares corresponding value as a section in which the antenna rotates 360 degrees. In claim 3,
The low frequency signal generation unit generates the I component signal and the Q component signal having a center frequency of 0 Hz,
The radio wave arrival direction estimation device, wherein the direction determining unit sets a section for calculating the deviation sum of squares corresponding value as a section in which the antenna rotates 360 degrees. In claim 4,
The approximate model is a model obtained by converting the composite wave of a plurality of incoming waves in the low-frequency signal, the radio tag is the frequency of the radio wave f _RF to be _transmitted, the rotation radius of the antenna R, light velocity v _c , time, t, the azimuth angle phi _m of each incoming _wave, elevation of [delta] _m of each incoming _wave, amplitude a _m of the incoming _wave, the phase [psi _m of the incoming _waves, the number of arrival waves N, wherein When the approximate model is V _ref , the model is represented by Formula 1 or Formula 2, Formula 3, Formula 4, and Formula 5,

When the sampling number is k, the total number of samplings is K, the difference between the measurement signal and the approximate model is V _k , and the average value of the differences is V bar, the direction determining unit calculates the deviation sum of squares expressed by Equation 6. searching for a combination of the azimuth angle and the elevation angle of the plurality of incoming waves where e is minimized,

The B and C are calculated by solving the simultaneous equations formed by subtracting the expression 6 obtained by partial differentiation of the expression 6 from the B and the expression obtained by partially differentiating the expression 6 from the C by 0. A radio wave arrival direction estimation device that searches for a combination of the azimuth angle and the elevation angle of the plurality of incoming waves that minimizes the deviation square sum e using the B and C. In claim 5,
The I component approximate model and the Q component approximate model are models obtained by converting the I component and the Q component of a composite wave of a plurality of incoming waves into the low frequency signal, respectively, and the frequency of the radio wave transmitted by the wireless tag F _RF , the radius of rotation of the antenna is R, the speed of light is v _c , the time is t, the azimuth angle of each incoming wave is φ _m , the elevation angle of each incoming wave is δ _m , the amplitude of each incoming wave is A _m , When the phase of the incoming wave is Ψ _m , the number of incoming waves is N, the I component approximate model is I _ref , and the Q component approximate model is Q _ref , the I component approximate model is expressed by Equations 7, 9, and 10 The Q component approximate model is a model represented by Equation 8, Equation 9, Equation 10, and Equation 11,

The direction determining unit includes a sampling number k, a total sampling number K, a difference between the I component signal and the I component approximate model, I _k , and an average value of the difference between the I component signal and the I component approximate model Is the deviation square represented by Equation 12, where Q is the difference between the Q component signal and the Q component approximate model, and Q _k is the difference between the Q component signal and the Q component approximate model. Searching for a combination of the azimuth angle and the elevation angle of the plurality of incoming waves for which the sum e is minimized,

B and C are calculated by solving simultaneous equations formed by subtracting the equation 12 from the partial differentiation of the equation 12 by B and the equation 12 by partially differentiating the equation 12 by the C, respectively. A radio wave arrival direction estimation device that searches for a combination of the azimuth angle and the elevation angle of the plurality of incoming waves that minimizes the deviation square sum e using the B and C. In claim 6,
The direction determining unit calculates, as the deviation sum of squares corresponding value, a value calculated by an expression obtained by removing the term of only the measurement signal from an expression for calculating the deviation sum of squares e. Estimating device. In claim 7,
The direction determining unit calculates, as the deviation square sum corresponding value, a value calculated by an expression obtained by subtracting the term of only the I component signal and the term of only the Q component signal from the equation for calculating the deviation sum of squares e. A radio wave arrival direction estimating device. In any one of Claims 6-9,
The z-factor term that is a part of an equation for calculating the deviation-sum-corresponding value and that is calculated by inputting a plurality of the azimuth angles and the time to a z-factor term that can be calculated by determining z. A storage unit (220) that stores the calculated value of
The direction deciding unit calculates the deviation sum-of-squares corresponding value using the calculated value of the z-factor term stored in the storage unit.   A radio wave arrival direction estimation system comprising the radio wave arrival direction estimation device according to any one of claims 1 to 10 and the wireless tag.

Images