JP6502158B2 - Switching power supply - Google Patents

Description

  The present invention relates to a switching power supply.

  The resonant switching power supply device dramatically controls switching loss and noise by controlling the current or voltage in a sinusoidal manner and switching the switching element (so-called soft switching) with either current or voltage being approximately zero. Can be reduced to

  Among the resonance type isolated switching power supply devices, there is one called an LLC method, which performs adjustment of an output voltage by frequency modulation control (PFM (pulse frequency modulation) control) of a switching element. However, when using a Si-based high-voltage IGBT [insulated gate bipolar transistor] that operates in bipolar as a switching element, switching loss due to the tail current can not be made zero even by soft switching, so switching due to frequency fluctuation controlling output power There is a problem that element loss fluctuates, and in some cases, high-frequency operation generates a large amount of heat. From this, a method in which a Si-based medium-voltage MOSFET [metal-oxide-semiconductor field effect transistor] is used becomes common, and it has been difficult to apply to the field of handling high voltage and high power.

  On the other hand, in the phase control type switching power supply device, the output voltage is adjusted by phase control. Unlike the switching power supply apparatus of the LLC system, the switching power supply apparatus of the same control system has a constant switching frequency. Therefore, a Si-based IGBT having a relatively high element withstand voltage can be used as a switching element. Therefore, switching power supply devices of the same type are introduced in many fields such as power supplies for industrial equipment as circuits capable of realizing high efficiency while handling high voltage and large power.

Japanese Patent Laid-Open No. 1-295675 (FIG. 3) Japanese Patent Application Laid-Open No. 10-308510 JP, 2012-213260, A Patent No. 3981181 Specification

“DC / DC converter using low loss snubber”, Origin Technical Journal, No. 66 (2003), p19-28.

  There are two main points to be noted in a phase control type switching power supply. The first point is that a smoothing reactor (choke coil) is required on the secondary side of the isolation transformer. The second point is that the secondary winding of the isolation transformer is momentarily shorted when a reverse recovery current occurs in the rectification diode provided on the secondary side of the isolation transformer, and a large current flows from the primary side, The excess energy stored in the isolation transformer appears as a large surge voltage across the rectifying diode.

  Patent Document 1 discloses a method in which a surge voltage is absorbed by a capacitor and consumed by a resistor by connecting a snubber circuit in parallel to a rectification diode on the secondary side.

  Here, as the switching element on the primary side and the rectifying diode on the secondary side, by using a SiC device (see Patent Document 2) or the like which has a larger dielectric breakdown field in a wide band gap than a Si device, a further high voltage / large Consider the case of handling power. In this case, the surge voltage on the secondary side also increases as the input voltage and the output voltage rise. Therefore, it is necessary to provide a snubber circuit for surge voltage suppression as well as to design an element that can withstand a surge voltage more than twice the transformer rated output as a secondary side rectification diode for safety. is there.

  However, if the breakdown voltage of the rectifying diode is increased, the on-resistance value is contradictoryly increased, so that the conduction loss is increased. In addition, if the semiconductor chip area is increased to reduce the on resistance value, the parasitic capacitance is increased and the energy consumption in the snubber circuit is increased, so that the power conversion efficiency is lowered.

  Also, the fact that the snubber circuit absorbs and consumes the surge voltage is nothing but that it loses part of the input power. In particular, as the switching frequency is increased by using a SiC device suitable for high frequency operation, the operation frequency of the snubber circuit becomes higher, so that the power loss becomes apparent.

  Patent Document 3 discloses a switching power supply device provided with a snubber circuit with little power loss. However, in the conventional method, since the voltage applied to the snubber capacitor vibrates at a frequency twice the switching frequency, the capacitance value of the snubber capacitor can not be increased from the viewpoint of impedance. Therefore, as the energy handled by the snubber circuit is increased, there is a concern that vibration may be generated at a large frequency and at a high frequency, and a countermeasure for eliminating this may be required separately.

  Further, Patent Document 4 and Non-Patent Document 1 use a step recovery diode to suppress a surge voltage generated in a secondary side rectification diode, and return a reverse recovery current of the step recovery diode to reduce power loss. A switching power supply with reduced power is disclosed. However, this conventional method is applied to a forward type or flyback type switching power supply device (in particular, a snubber circuit provided on the primary side thereof) using a single switching element, and a phase control type switching power supply It could not be applied to the device as it was.

  In view of the above problems found by the present inventors of the present invention, the invention disclosed in the present specification provides a phase control switching power supply device capable of reducing power loss while suppressing surge voltage. Intended to be provided.

  The switching power supply disclosed in the present specification includes a full bridge circuit including a plurality of switching elements, a control circuit for driving the full bridge circuit in a phase control method, and a primary winding connected to the full bridge circuit. , An isolation transformer including a wire and a secondary winding magnetically coupled thereto, a rectifier circuit including a plurality of rectifying elements connected to the secondary winding, and a first output end of the positive side of the rectifier circuit Connected in parallel to the first reactor, the first capacitor connected between the second end of the first reactor and the negative output end of the rectifier circuit, and the first reactor And a snubber circuit, the snubber circuit being connected between a first diode of which the anode is connected to the first end of the first reactor, a cathode of the first diode and a second end of the first reactor. To And a connection has been CR circuit portion, said first diode has a structure is a step recovery diode (first configuration).

  In the switching power supply device having the first configuration, the snubber circuit includes a second diode having an anode connected to a cathode of the first diode and a cathode connected to the CR circuit portion, and the second diode. It is preferable that the second reactor further includes a second reactor connected in parallel with the second reactor.

  Further, in the switching power supply device having the first configuration, the control circuit performs phase control on the secondary side output time of the isolation transformer while keeping the switching frequency fixed, in accordance with the load within a range that does not fall below a predetermined lower limit. A second mode in which the switching frequency is adjusted according to the load while the secondary side output time is fixed to the lower limit value, and the lower limit value is the operation of the first diode The configuration (third configuration) may be set to a value longer than time.

  Further, in the switching power supply device having any one of the first to third configurations, at least one of the plurality of switching elements and the plurality of rectifying elements is a configuration (fourth configuration) that is a SiC-based semiconductor device. It is good to do.

  Further, in the switching power supply having any one of the first to fourth configurations, the input voltage input to the full bridge circuit may be configured to have DC 600 V or more (fifth configuration).

  Further, in the switching power supply having the first to fourth configurations, the input voltage input to the full bridge circuit is a voltage obtained by smoothing an AC voltage of AC 400 V through a PFC [Power-Factor-Correction] circuit. It is preferable that the configuration is the sixth configuration.

  Further, in the switching power supply device having any one of the first to sixth configurations, the CR circuit portion may have a configuration (seventh configuration) including a second capacitor and a resistor connected in parallel to each other.

  Further, in the switching power supply having the second configuration, the first diode and the second diode may both be sealed in a single package (eighth configuration).

  Further, in the switching power supply having the eighth configuration, it is preferable that the package is a full mold type in which the lead frame is entirely resin-sealed except for the terminal portion (a ninth configuration).

  According to the invention disclosed in the present specification, it is possible to provide a phase control type switching power supply capable of reducing power loss while suppressing surge voltage.

Circuit diagram showing a first embodiment of the switching power supply device 1 Reverse recovery characteristics of step recovery diode Timing chart showing an example of switching operation Operation state diagram at time t11 Operation state diagram at time t12 Operation state diagram at time t13 Operation state diagram at time t14 Operation state diagram at time t15 Snubber operation waveform diagram (without SRD applied) Snubber operation waveform diagram (with SRD applied) A circuit diagram showing a second embodiment of the switching power supply device 1 Operation state diagram at time t14 (during operation) Operation state diagram at time t15 (during L operation) Snubber operation waveform diagram (D + L added) A schematic diagram showing one configuration example of the package 100 External view of package 100 (surface) External view of package 100 (back side) A diagram showing an example of switching control implemented in the third embodiment

First Embodiment
FIG. 1 is a circuit diagram showing a first embodiment of the switching power supply device 1. The switching power supply device 1 according to the first embodiment receives the supply of the input voltage Vin (for example, 800 V) from the DC power supply 2 of the primary circuit system 1 p, and outputs a desired output voltage Vout (for example, 800 V) An isolation type DC / DC converter that outputs the full bridge circuit 10, the control circuit 20, the isolation transformer 30, the rectification circuit 40, the first reactor 50, the first capacitor 60, and the snubber circuit 70; Have.

  The full bridge circuit 10 includes switching elements 11 to 14 and resonant capacitors 15 to 18. Switching elements 11 and 12 are, as the first arm of full bridge circuit 10, in series between the positive terminal (= input terminal of input voltage Vin) and the negative terminal (= ground terminal of primary circuit system 1p) of DC power supply 2 It is connected. On the other hand, switching elements 13 and 14 are connected in series between the positive electrode end and the negative electrode end of DC power supply 2 as a second arm of full bridge circuit 10. A connection node between the switching elements 11 and 12 corresponds to a first output end of the full bridge circuit 10, and a connection node between the switching elements 13 and 14 corresponds to a second output end of the full bridge circuit 10.

  When the input voltage Vin is a high voltage (for example, a DC voltage of DC 600 V or higher, or a DC voltage obtained by smoothing an AC voltage of 400 V AC through the PFC circuit), the switching elements 11 to 14 have high withstand voltage (for example, It is desirable to use SiC devices (e.g., SiC-based MOSFETs) with a low on-resistance of 1200 V). With such a configuration, even when the switching power supply 1 handles a high voltage, it is possible to prevent the destruction of the switching elements 11 to 14 and reduce the power loss in the full bridge circuit 10. Become. However, if the input voltage Vin is not so high, Si-based MOSFETs or IGBTs may be used as the switching elements 11-14.

  The resonant capacitors 15 to 18 (e.g., 1 nF, respectively) are connected in parallel to the switching elements 11 to 14, respectively. By providing such resonant capacitors 15 to 18, voltage fluctuation due to resonance can be made slower to facilitate soft turn-off (zero voltage switching), so that switching loss and noise generated in the full bridge circuit 10 can be reduced. It is possible to reduce dramatically. As the resonant capacitors 15 to 18, parasitic capacitors attached to the switching elements 11 to 14 may be used.

  The control circuit 20 drives the full bridge circuit 10 by a phase control method (details will be described later) such that the output voltage Vout matches the target value. As a method of monitoring the output voltage Vout in the control circuit 20, for example, a feedback signal corresponding to the output voltage Vout is returned from the secondary circuit system 1s to the primary circuit system 1p via a photocoupler or the like. The method to monitor with can be mentioned.

  Insulating transformer 30 includes a primary winding 31 and a secondary winding 32 magnetically coupled to each other, and electrically isolates primary circuit system 1p from secondary circuit system 1s while maintaining isolation from primary circuit system 1p. The power is transmitted to the next circuit system 1s. The primary winding 31 is connected to the full bridge circuit 10 of the primary circuit system 1 p. The secondary winding 32 is connected to the rectifier circuit 40 of the secondary circuit system 1s.

  The rectifying circuit 40 includes rectifying elements (for example, diodes) 41 to 44. The anode of the rectifying element 41 and the cathode of the rectifying element 42 are connected to the first end of the secondary winding 32. The anode of the rectifying element 43 and the cathode of the rectifying element 44 are connected to the second end of the secondary winding 32. The cathodes of the rectifying elements 41 and 43 correspond to the positive output end of the rectifying circuit 40. The anodes of the rectifying elements 42 and 44 correspond to the negative output terminal of the rectifying circuit 40.

  When the input voltage Vin is a high voltage (for example, a DC voltage of DC 600 V or higher, or a DC voltage obtained by smoothing an AC voltage of 400 V AC through the PFC circuit), the high breakdown voltage is used as the rectifying elements 41 to 44. It is desirable to use a SiC device (eg, a SiC-based Schottky barrier diode) (eg, 1700 V withstand voltage).

  The first reactor 50 is a smoothing reactor (e.g., 850 μH) whose first end is connected to the positive output end of the rectifier circuit 40 and whose second end is connected to the output end of the output voltage Vout.

  The first capacitor 60 is a smoothing capacitor (for example, 840 μF) whose first end is connected to the output end of the output voltage Vout and whose second end is connected to the negative output end of the rectifier circuit 40.

  The rectifying circuit 40, the first reactor 50, and the second capacitor 60 connected in this manner function as a full-wave rectifying type rectifying and smoothing unit that generates an output voltage Vout from an induced voltage that appears in the secondary winding 32. . However, the configuration of the rectifying and smoothing unit is not necessarily limited to this, and for example, a half wave rectification type may be adopted.

  The snubber circuit 70 is a surge voltage absorbing circuit connected in parallel to the first reactor 50, and includes a first diode 71, a second capacitor 72, and a resistor 73.

  The first diode 71 is a snubber diode (for example, 1600 V withstand voltage) for operating the snubber circuit 70 in a forward bias state when a surge voltage exceeding the output voltage Vout appears at the positive output terminal of the rectifier circuit 40. . The second capacitor 72 is a snubber capacitor (for example, 0.1 μF) for absorbing energy due to the surge voltage. The resistor 73 is a snubber resistor (for example, 2.5 kΩ) for consuming energy due to a surge voltage.

  The anode of the first diode 71 is connected to the first end of the first reactor 50 (= the positive output end of the rectification circuit 40). The cathode of the first diode 71 is connected to the first end of the second capacitor 72 and the first end of the resistor 73, respectively. The second end of the second capacitor 72 and the second end of the resistor 73 are both connected to the second end of the first reactor 50 (= the output end of the output voltage Vout). Thus, the second capacitor 72 and the resistor 73 connected in parallel to each other function as a CR circuit unit connected between the cathode of the first diode 71 and the second end of the first reactor 50.

  The snubber circuit 70 configured as described above is characterized in that a step recovery diode is used as the first diode 71. The reverse recovery characteristics of the step recovery diode will be described below with reference to FIG.

  FIG. 2 is a reverse recovery characteristic diagram of the step recovery diode used as the first diode 71. As shown in FIG. The horizontal axis of this figure shows the time t, and the vertical axis shows the diode current I71 flowing through the first diode 71. The diode current I71 is defined as a forward current If flowing from the anode to the cathode as positive, and a reverse recovery current Ir flowing from the cathode to the anode as negative.

  The step recovery diode has a life of minority carriers (so-called carrier lifetime τ) longer than that of a normal diode, and can store a large charge in the forward conduction period Tf (= period in which the forward current If flows). Therefore, when the step recovery diode switches from forward bias to reverse bias at time t1, the reverse recovery current Ir continues to flow for a relatively long recovery period Tr (= time t1 to t2).

  In a normal diode, the longer the recovery period Tr, the longer the transition period Tt until the reverse recovery current Ir returns to 0A. On the other hand, in the step recovery diode, the element is designed to exhibit a transition period Tt (= time t2 to t3) sufficiently short as compared with the recovery period Tr.

  That is, the step recovery diode is designed to extend the recovery period Tr and shorten the transition period Tt, and at the time of switching from the forward bias to the reverse bias, the relatively large reverse recovery current Ir is relatively long It can be said that the diode can flow through the The technical significance of adopting such a step recovery diode as the first diode 71 will be described in detail below.

  FIG. 3 is a timing chart showing an example of the switching operation in the switching power supply device 1. The gate-source voltages Vgs11 to Vgs14 applied to the switching elements 11 to 14 and the switching elements 11 to 14 sequentially from the top Drain currents Id11 to Id14 flowing, reverse bias voltages V41 and V42 applied to the rectifying elements 41 and 42, diode currents I41 and I42 flowing to the rectifying elements 41 and 42, diode current I71 flowing to the first diode 71, and The secondary side output voltage V32 appearing in the secondary winding 32 of the isolation transformer 30 is depicted.

  As for the drain currents Id11 to Id14, the forward current flowing from the drain to the source is positive, and the reverse current flowing from the source to the drain is negative. With regard to the diode currents I41 and I42 and the diode current I71, the forward current flowing from the anode to the cathode is positive, and the reverse recovery current flowing from the cathode to the anode is negative. Further, with regard to the secondary side output voltage V30, when the first end of the secondary winding 32 has a potential higher than that of the second end is positive, and the first end of the secondary winding 32 is more than the second end. The low potential is defined as negative.

  In the phase control method adopted by the control circuit 20, the gate-source voltages Vgs11 to Vgs14 of the switching elements 11 to 14 are driven at a constant duty ratio (for example, 50%). The gate-source voltage Vgs11 on the first arm side, the gate-source voltage Vgs14 on the second arm side, the gate-source voltage Vgs12 on the first arm side, and the gate-source voltage Vgs13 on the second arm side A shift time Tshift (= phase difference) is provided between them. By variably controlling the shift time Tshift, the secondary side output time Tout of the isolation transformer 30 changes. Therefore, according to this phase control method, feedback control of the output voltage Vout is basically performed while the switching frequency fsw (and hence the switching cycle Tsw) is fixed at a constant value.

  Further, in the switching operation of the first arm, the switching elements 11 and 12 are complementarily turned on / off. Similarly, in the switching operation of the second arm, switching elements 13 and 14 are complementarily turned on / off. Note that the term "complementary" in the present specification does not mean only when the on / off states of the upper and lower switches forming each arm are completely reversed, and the upper switch and Also included is a case where a predetermined delay is given to the ON / OFF transition timing of the lower switch (when the simultaneous OFF time (dead time) Td of the upper switch and the lower switch is provided). By providing this simultaneous off time Td, it becomes possible to prevent an excessive through current flowing through both switches.

  Hereinafter, the operation state of the switching power supply 1 at times t11 to t15 will be specifically described with reference to FIGS. 4 to 8.

  4 to 8 are operation state diagrams at time t11 to t15 in FIG. 3 described above, respectively. At time t11, as shown in FIG. 4, the switching elements 11 and 14 are on, and the switching elements 12 and 13 are off. At this time, in the primary circuit system 1p, the primary current is supplied via the current path (a positive terminal of the DC power supply 2 → switching element 11 → primary winding 31 → switching element 14 → negative terminal of DC current 2) indicated by a thin broken line. Flow. Also, at this time, in the secondary circuit system 1s, a current path similarly indicated by a thin broken line (first end of secondary winding 32 → rectifying element 41 → first reactor 50 → load 3 → rectifying element 44 → secondary winding A secondary current flows through the second end of the line 32).

  At time t12, as shown in FIG. 5, the on / off states of the switching element 11 and the switching element 12 are switched. That is, at time t12, the switching elements 12 and 14 are on, and the switching elements 11 and 13 are off. When the switching element 11 is first turned off between time t11 and time 12, in the primary circuit system 1p, the primary current that has been flowing is maintained by the primary winding 31. As a result, a primary current flows through the current path (second end of primary winding 31 → switching element 14 → switching element 12 → first end of primary winding 31) indicated by the thin broken line in FIG. At this time, by turning on the switching element 12, soft turn-on can be realized. At this time, in the secondary circuit system 1s, the secondary current which has been flowing is maintained by the first reactor 50. As a result, the first current path (first end of secondary winding 32 → rectifying element 41 → first reactor 50 → load 3 → rectifying element 44 → second end of secondary winding 32) indicated by a thin broken line, And a second current path (second end of secondary winding 32 → rectifying element 43 → first reactor 50 → load 3 → rectifying element 42 → first end of secondary winding 32) indicated by a thin solid line Secondary current flows. Thus, at time t12, the rectifying elements 41 and 44 are forward biased by the electromotive force of the first reactor 50.

  At time t13, as shown in FIG. 6, the on / off states of the switching element 13 and the switching element 14 are switched. That is, at time t13, the switching elements 12 and 13 are on, and the switching elements 11 and 14 are off. At this time, in the primary circuit system 1p, the primary current is obtained via the current path (a positive terminal of the DC power supply 2 → switching element 13 → primary winding 31 → switching element 12 → negative terminal of DC current 2) indicated by a thin broken line. Flow. That is, as compared with times t11 to t12, the directions of the voltage applied to the primary winding 31 and the primary current are reversed. As a result, even in the secondary circuit system 1s, the direction of the voltage (secondary output voltage V30 of the insulating transformer 30) generated in the secondary winding 32 and the direction of the secondary current are reversed, and the rectification was previously in a forward biased state Elements 41 and 44 switch to the reverse bias state. At this time, the rectifying elements 41 and 44 are in a recovery state (= reverse recovery state) in which a reverse recovery current flows from the cathode to the anode for a very short time.

  The rectifying elements 41 and 44 in the recovery state have extremely low impedance compared to the first reactor 50. Therefore, in the secondary circuit system 1s, the first short circuit path indicated by the thin broken line (not the first current path through the first reactor 50 or the load 3) (the second end of the secondary winding 32 → rectifying element 43 → rectifying Element 41 → first end of secondary winding 32), and a second short circuit path indicated by a thin solid line (second end of secondary winding 32 → rectifying element 44 → rectifying element 42 → secondary winding 32 A short circuit current flows through the first end).

  That is, at time t13, both ends of the secondary winding 32 are instantaneously shorted. Therefore, at this time, a current is supplied from the primary circuit system 1p to the secondary circuit system 1s, whereby the energy accumulated in excess in the primary winding 31 is transmitted to the secondary circuit system 1s. Because of the high impedance to the pulse signal, energy can not be released to the load 3, and an excessive surge voltage appears across the ends of the rectifying elements 41 and 44, which may lead to destruction. This is a problem that inevitably occurs when no special measures are taken in the secondary circuit system 1 s of the switching power supply 1 adopting the phase control method.

  The switching power supply 1 according to the first embodiment has a snubber circuit 70 as means for solving the above-mentioned problems. Hereinafter, the operation of the snubber circuit 70 after time t14 will be described in detail with reference to FIGS. 7 and 8.

  When the reverse recovery current flows through the rectifying element 41 and the reverse bias voltage V41 rises and exceeds the output voltage Vout, the first diode 71 is forward biased, and the snubber circuit 70 is activated. As a result, at time t14, as shown in FIG. 7, the current path indicated by the thin broken line (second end of secondary winding 32 → rectifying element 43 → first diode 71 → second capacitor 72 → load 3 → A current is drawn into the snubber circuit 70 via the rectifying element 42 → the first end of the secondary winding 32. As a result, since the excess energy is absorbed by the second capacitor 72, it is possible to suppress the surge voltage that appears between both ends of the rectifying element 41. At time t14, the current is also supplied to the load 3 by the current path via the first reactor 50.

  However, when all the charge absorbed by the second capacitor 72 is consumed by the resistor 73, the consumed amount is nothing but the power loss in the switching power supply device 1. Therefore, at time t15, as indicated by the thin solid line in FIG. 8, as the reverse recovery current at the time of recovery of the first diode 71, part of the charge absorbed by the second capacitor 72 is 71 → 43 → 32 → 42 → It is regenerated via 3 and returned to the primary circuit system 1p via 13 → 2 → 12 → 31. In particular, if a step recovery diode is used as the first diode 71, a larger reverse recovery current can flow than when a normal diode is used, so that more power can be regenerated. Therefore, it is possible to suppress the above-mentioned power loss, and as a result, it is possible to improve the power conversion efficiency of the switching power supply device 1.

  FIGS. 9 and 10 are both snubber operation waveform diagrams, and the diode current I 71 flowing through the first diode 71 and the positive side output voltage of the rectifying circuit 40 (= reverse bias voltage of the rectifying element 41) in order from the top A voltage obtained by adding V41 and the reverse bias voltage V42 of the rectifying element 42 is depicted. Note that FIG. 10 shows the behavior when a step recovery diode is used as the first diode 71, and FIG. 9 shows a case where a step recovery diode is not used as the first diode 71 (= a normal diode is used). Behavior is shown for comparative reference.

  As is apparent from the two figures, when a step recovery diode is used as the first diode 71, the surge voltage appearing at the positive output end of the rectifier circuit 40 is more effective than when a normal diode is used. Can be suppressed. Further, as described above, when the step recovery diode is used as the first diode 71, more power can be regenerated from the snubber circuit 70 compared to the case where a normal diode is used. .

  That is, according to the switching power supply device 1 of the first embodiment, it is possible to reduce the power loss in the snubber circuit 70 while effectively suppressing the surge voltage of the secondary circuit system 1s.

  However, if the reverse recovery current at the time of recovery of the first diode 71 is too large, the switching elements 11 to 14 and the rectifying elements 41 to 44 may be destroyed. In particular, when the secondary side output voltage V30 of the isolation transformer 30 becomes 0 V during the recovery period of the first diode 71 due to the phase control condition of the full bridge circuit 10, the reverse recovery current of the first diode 71 is flowing Since both ends of the secondary winding 32 are short-circuited and an excessive current is returned to the primary circuit system 1p, the switching elements 11 to 14 may be destroyed. In the following, the inventive idea for eliminating such a concern will be described in detail by taking the second to third embodiments as an example.

Second Embodiment
FIG. 11 is a circuit diagram showing a second embodiment of the switching power supply device 1. The second embodiment is characterized in that a second diode 74 and a second reactor 75 are added as components of the snubber circuit 70 based on the first embodiment described above. Therefore, the same components as those in the first embodiment will be assigned the same reference numerals as those in FIG. 1 to omit redundant descriptions, and in the following, the main features of the second embodiment will be mainly described.

  The second diode 74 is a rectifying diode having an anode connected to the cathode of the first diode 71 and a cathode connected to the first ends of the second capacitor 72 and the resistor 73. As the second diode 74, it is desirable to use a SiC device (eg, a SiC-based Schottky barrier diode) having a high withstand voltage (eg, 1200 V withstand voltage) and a short reverse recovery time.

  The second reactor 75 is a snubber reactor (for example, 100 μH) connected in parallel to the second diode 74.

  Even when the secondary side output voltage V30 of the isolation transformer 30 becomes 0 V during the recovery period of the first diode 71, the second diode 74 and the second reactor 75 generate a short circuit current at that time. It works to suppress excessive growth.

  Also in the second embodiment, the switching operation itself of the switching power supply device 1 is basically the same as that of the first embodiment. Therefore, the suppression operation of the short circuit current by the snubber circuit 70 will be described in detail below with reference to FIGS. 12 and 13 together with FIG. 3 described above.

  12 and 13 are operation state diagrams at time t14 and t15 in FIG. 3 described above, respectively. The non-operation time (time t11 to t13) of the snubber circuit 70 is the same as that of the first embodiment (FIG. 4 to FIG. 6), and thus the redundant description will be omitted.

  When the reverse recovery current flows through the rectifying element 41 and the reverse bias voltage V41 rises, the first diode 71 is in a forward bias state, and the snubber circuit 70 is in an operating state. As a result, at time t14, as shown in FIG. 12, the current path indicated by the thin broken line (second end of secondary winding 32 → rectifying element 43 → first diode 71 → second diode 74 → second capacitor A current is drawn into the snubber circuit 70 through 72 → load 3 → rectifying element 42 → first end of the secondary winding 32). That is, in the second embodiment, when the current flows into the snubber circuit 70, the second capacitor 72 is charged via the second diode 74. Therefore, when the conduction loss of the second diode 74 is small, the excess current is absorbed by the second capacitor 72 in the same manner as in the first embodiment (FIG. 7) described above. It is possible to suppress the surge voltage appearing on the

  On the other hand, when the power is regenerated from the snubber circuit 70, the second diode 74 is in the reverse bias state. Therefore, at time t15, the second current path shown by the thin solid line in FIG. 13 (the second reactor 75 → the first diode 71 → the rectifying element 43 → the secondary winding 32 → the rectifying element 42 → the load 3) The capacitor 72 is discharged. That is, the magnitude of the reverse recovery current flowing through the first diode 71 is limited by the impedance component of the second reactor 75. Therefore, even if the secondary side output voltage V30 of the isolation transformer 30 becomes 0 V during the recovery period of the first diode 71, it is possible to suppress an excessive increase in the short circuit current flowing at that time. .

  FIG. 14 is a snubber operation waveform chart according to the second embodiment, showing, in order from the top, the diode current I71 flowing through the first diode 71, the positive side output voltage of the rectifying circuit 40 (= the reverse bias voltage V41 of the rectifying element 41 A voltage obtained by adding the reverse bias voltage V42 of the rectifying element 42) is depicted. As can be seen from this figure, also in the second embodiment, the surge voltage appearing at the positive output terminal of the rectifier circuit 40 is effectively suppressed as in the first embodiment (FIG. 10) described above. The figure shows the secondary winding 32 during normal operation, but the negative peak value of the reverse recovery current of the diode current I71 flowing through the first diode 71 is slightly reduced compared to FIG. It can be seen that the effects of the second reactor 75 appear.

  The current flowing through the second reactor 75 increases with the passage of time. Therefore, the second reactor 75 exerts the current suppressing effect in the range where the recovery period of the first diode 71 is within several μs (for example, 10 μs or less) (= the range in which the current of the second reactor 75 does not increase much).

  That is, the second reactor 75 effectively suppresses an excessive short circuit current caused by the instantaneous short circuit of the secondary winding 32 during the recovery period of the first diode 71, while the switching power supply At the time of steady operation of the device 1, the reverse recovery current of the first diode 71 can be passed through with almost no suppression. Therefore, at the time of steady operation of the switching power supply device 1, it is possible to suppress the power loss in the snubber circuit 70 and improve the power conversion efficiency of the switching power supply device 1 as in the first embodiment.

  Although the snubber circuit 70 of the second embodiment requires the first diode 71 and the second diode 72, the heat dissipating fins can be combined into one for these two diodes, so that both are single. It is desirable to seal the package 100.

  FIG. 15 is a schematic view showing a configuration example of the package 100 in which both the first diode 71 and the second diode 72 are sealed. 16 and 17 are external views of the package 100 (FIG. 16 is the front side, and FIG. 17 is the back side).

  The package 100 has three terminal portions 101 to 103 and one screw hole 104. The anode of the first diode 71 is connected to the terminal portion 101. The cathode of the first diode 71 and the anode of the second diode 74 are both connected to the terminal portion 102. The cathode of the second diode 74 is connected to the terminal portion 103.

  As described above, by sealing both the first diode 71 and the second diode 72 in a single package 100, cooling of both elements can be performed collectively, thus contributing to space saving. It becomes possible.

  The package 100 is a 3-pin full mold type in which the lead frame is entirely resin-sealed except for the three terminal portions 101 to 103. As described above, if the full mold type in which not only the front surface of the package 100 but also the back surface of the package 100 is reliably insulated, the heat dissipation is somewhat sacrificed, but the package 100 is safe against the chassis or heat sink It is possible to screw it on. In particular, in the case of application to the switching power supply 1 that handles high voltage, it is important to ensure the insulation of the package 100, so it can be said that it is desirable to make the package 100 a full mold type. It is desirable to select a package having a necessary creepage distance in consideration of the value of the input / output voltage.

Third Embodiment
In the above-described second embodiment (FIG. 11), by adding the second diode 74 and the second reactor 75 to the snubber circuit 70, a short circuit current that may occur during the recovery period of the first diode 71 is effectively achieved. Proposed a configuration that can be suppressed.

  On the other hand, in the following, as the third embodiment, the above-mentioned short circuit current is generated by devising the switching control of the control circuit 20 while adopting the same configuration as the first embodiment (FIG. 1) described above. The situation (= the situation in which the secondary side output voltage V30 of the isolation transformer 30 becomes 0 V during the recovery period of the first diode 71 and the secondary winding 32 instantaneously shorts) itself in advance Suggest a configuration that can be

  FIG. 18 is a diagram showing an example of switching control implemented in the third embodiment. In this figure, a variable control example of the secondary side output time Tout and the switching frequency fsw according to the weight condition of the load 3 and switching drive waveforms (gate-source voltages Vgs11 to Vgs14) in each load state are depicted. It is done.

  In the switching power supply 1 of the third embodiment, the control circuit 20 has two operation modes, that is, a first mode (MODE1) and a second mode (MODE2).

  The control circuit 20 in the first mode sets the secondary side output time Tout of the isolation transformer 30 according to the weight condition of the load 3 by the above-described phase control while fixing the switching frequency fsw to the steady value fsw0 (for example 100 kHz). The output voltage Vout is adjusted to be constant. Specifically, the control circuit 20 in the first mode extends the secondary side output time Tout as the load 3 increases, and conversely reduces the secondary side output time Tout as the load 3 decreases. In addition, the phase difference between the gate-source voltages Vgs11 and Vgs12 and the gate-source voltages Vgs13 and Vgs14 is variably controlled. That is, in the control circuit 20 of the first mode, the output feedback control which is the same as the first embodiment described above is performed.

  However, as the secondary side output time Tout becomes shorter, the secondary side output voltage V30 of the isolation transformer 30 becomes 0 V during the recovery period of the first diode 71, and the secondary winding 32 is likely to be shorted. Therefore, the adjustment of the secondary side output time Tout in the first mode is performed in a range in which the secondary side output time Tout does not fall below the predetermined lower limit value ToutL.

  The above lower limit value ToutL is a value (for example, 2 μs) longer than the operation time of the first diode 71 (= the sum of the time in which the forward current of the first diode 71 flows and the time in which the reverse recovery current flows) It is good to set to.

  On the other hand, even if the secondary side output time Tout is shortened to the lower limit value ToutL, when the light load state where the increase of the output voltage Vout continues is reached, the control circuit 20 further shortens the secondary side output time Tout. Switching from the first mode to the second mode.

  The control circuit 20 in the second mode adjusts the switching frequency fsw according to the weight condition of the load 3 while fixing the secondary side output time Tout to the lower limit value ToutL. Specifically, the control circuit 20 of the second mode drives the gate-source voltages Vgs11 to Vgs14 so that the switching frequency fsw is lowered from the steady value fsw0 as the load 3 becomes lighter.

  FIG. 18 exemplifies three load states RL1 to RL3 (the load 3 has a weight of RL1 <RL2 <RL3).

  In the load state RL1, the control circuit 20 is in the first mode. At this time, the secondary side output time Tout1 is adjusted according to the weight of the load 3 while the switching frequency fsw1 (= 1 / Tsw1) is fixed to the steady value fsw0. Therefore, the output duty D1 (= Tout1 / Tsw1) of the isolation transformer 30 becomes a variable value according to the secondary side output time Tout1.

  The load state RL2 corresponds to a switching point between the first mode and the second mode. At this time, the switching frequency fsw2 (= 1 / Tsw2) becomes the steady value fsw0, and the secondary side output time Tout2 becomes the lower limit value ToutL. Therefore, the output duty D2 (= Tout2 / Tsw2) of the isolation transformer 30 becomes a fixed value smaller than the output duty D1.

  In the load state RL3, the control circuit 20 is in the second mode. At this time, the switching frequency fsw3 (= 1 / Tsw3) is adjusted according to the weight of the load 3 while the secondary side output voltage Vout3 is fixed at the lower limit value ToutL. Therefore, the output duty D3 (= Tout3 / Tsw3) of the isolation transformer 30 is smaller than the output D2 and becomes a variable value according to the switching frequency fsw3.

  Thus, in the case of the switching power supply device 1 of the third embodiment, unlike the second embodiment (FIG. 11), the second diode 74 and the second reactor 75 need not be added to the snubber circuit 70. The circuit scale can be reduced.

<Other Modifications>
In addition to the embodiments described above, various technical features disclosed in the present specification can be modified in various ways without departing from the scope of the technical creation. That is, the above embodiment should be considered as illustrative in all points and not restrictive, and the technical scope of the present invention is shown by the claims rather than the description of the above embodiment. It is to be understood that the present invention includes all modifications that fall within the meaning and scope equivalent to the claims.

  The invention disclosed in the present specification can be used, for example, in isolated DC / DC converters for general industrial equipment and infrastructures.

DESCRIPTION OF SYMBOLS 1 switching power supply device 1p primary circuit system 1s secondary circuit system 2 DC power supply 3 load 10 full bridge circuit 11-14 switching element 15-18 resonance capacitor 20 control circuit 30 insulation transformer 31 primary winding 32 secondary winding 40 rectification circuit 41 to 44 Rectifying element 50 1st reactor 60 1st capacitor 70 Snubber circuit 71 1st diode (step recovery diode)
72 second capacitor 73 resistance 74 second diode 75 second reactor 100 package 101 to 103 terminal portion 104 screw hole

Claims (5)

A full bridge circuit including multiple switching elements,
A control circuit for driving the full bridge circuit by a phase control method;
An isolation transformer including a primary winding connected to the full bridge circuit and a secondary winding magnetically coupled thereto;
A rectifying circuit including a plurality of rectifying elements connected to the secondary winding;
A first reactor whose first end is connected to the positive output end of the rectifier circuit;
A first capacitor connected between the second end of the first reactor and the negative output end of the rectifier circuit;
A snubber circuit connected in parallel to the first reactor;
Have
The snubber circuit is
A first diode whose anode is connected to the first end of the first reactor;
A CR circuit unit connected between the cathode of the first diode and the second end of the first reactor;
Including
Said first diode, Ri Oh in step recovery diode,
The snubber circuit is
A second diode having an anode connected to the cathode of the first diode and a cathode connected to the CR circuit portion;
A second reactor connected in parallel to the second diode;
Further include ,
Switching power supply characterized in that. A full bridge circuit including multiple switching elements,
A control circuit for driving the full bridge circuit by a phase control method;
An isolation transformer including a primary winding connected to the full bridge circuit and a secondary winding magnetically coupled thereto;
A rectifying circuit including a plurality of rectifying elements connected to the secondary winding;
A first reactor whose first end is connected to the positive output end of the rectifier circuit;
A first capacitor connected between the second end of the first reactor and the negative output end of the rectifier circuit;
A snubber circuit connected in parallel to the first reactor;
Have
The snubber circuit is
A first diode whose anode is connected to the first end of the first reactor;
A CR circuit unit connected between the cathode of the first diode and the second end of the first reactor;
Including
Said first diode, Ri Oh in step recovery diode,
The control circuit
A first mode in which the secondary side output time of the isolation transformer is adjusted according to the load within a range not falling below a predetermined lower limit value by phase control while fixing the switching frequency;
A second mode in which the switching frequency is adjusted according to a load while the secondary side output time is fixed to the lower limit value;
Equipped with
The lower limit, that is set to a value longer than the operating time of the first diode,
Switching power supply characterized in that. At least one of the switching power supply device according to claim 1 or claim 2 characterized in that it is a semiconductor device of the SiC base of said plurality of switching elements and said plurality of rectifying elements. The switching power supply device according to any one of claims 1 to 3 , wherein an input voltage input to the full bridge circuit is DC 600 V or more. The switching power supply device according to any one of claims 1 to 3 , wherein the input voltage input to the full bridge circuit is a voltage obtained by smoothing an AC voltage of AC 400 V through a PFC circuit.

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