JP6784166B2 - Electric motor control device - Google Patents

Description

The present invention relates to a control device for an electric motor.

Examples of the above invention include the inventions described in Patent Documents 1 to 5. The control device for a rotary electric machine described in Patent Document 1 includes a carrier wave control unit that sets a reference frequency of a carrier wave based on a predetermined synchronization speed and the rotation speed of the rotary electric machine. The carrier wave control unit sets the frequency of the carrier wave to an integral multiple of the reference frequency (higher carrier) based on the current command. Further, the carrier wave control unit does not continuously switch to a high frequency for the same switching element in the high current phase region where the current command becomes high. As a result, the invention described in Patent Document 1 attempts to suppress electromagnetic noise caused by a low-frequency carrier wave and suppress heat generation of the switching element.

The inverter device of a rotary electric machine described in Patent Document 2 includes a frequency setting means capable of switching between variable carrier frequency control and fixed carrier frequency control. The frequency setting means sets the carrier frequency of each phase by carrier frequency variable control when the rotation speed of the rotary electric machine is equal to or higher than a predetermined rotation speed, and when the rotation speed of the rotary electric machine is less than the predetermined rotation speed. The carrier frequency of each phase is set by the fixed carrier frequency control. As a result, the invention described in Patent Document 2 attempts to suppress noise at low rotation of the rotary electric machine and reduce switching loss at high rotation of the rotary electric machine in which the ambient noise becomes relatively loud.

The air conditioner described in Patent Document 3 includes a carrier frequency changing means for changing a carrier frequency during PWM control according to a usage environment. The carrier frequency changing means lowers the carrier frequency when the usage environment is bright, and raises the carrier frequency when the usage environment is dark. As a result, the invention described in Patent Document 3 attempts to reduce noise due to the carrier frequency of the inverter and reduce the loss of the switching element.

The inverter device for driving an electric motor described in Patent Document 4 includes an inverter control unit that PWM-controls the inverter. Further, the inverter control unit is provided with a loss estimator that calculates the motor loss and the inverter loss in consideration of fluctuations in both the output voltage command and the carrier frequency. The inverter control unit determines the output voltage command and the carrier frequency using the calculation result of the loss estimator. As a result, the invention described in Patent Document 4 attempts to reduce inverter loss and motor loss by performing PWM control in consideration of not only carrier frequency but also loss fluctuation due to a change in output voltage command.

The vehicle power conversion device described in Patent Document 5 sets the carrier frequency at high speeds such as when the detected noise becomes large and the noise caused by the carrier frequency of PWM control is drowned out by the noise. Set low. On the other hand, the power conversion device of the vehicle sets the carrier frequency high at low speeds such as when the detected noise in the vehicle becomes small and the noise caused by the carrier frequency becomes relatively loud. As a result, the invention described in Patent Document 5 attempts to achieve both reduction of noise noise and improvement of power conversion efficiency of the switching unit.

Japanese Unexamined Patent Publication No. 2012-235619 Japanese Unexamined Patent Publication No. 2014-23350 Japanese Unexamined Patent Publication No. 2011-158170 JP-A-2015-177696 Japanese Unexamined Patent Publication No. 2006-333572

However, there is a trade-off relationship between loss (loss of power converter and loss of motor) and noise (noise associated with driving the power converter and motor) when one is improved and the other is worsened. In the inventions described in Patent Documents 1 to 5, it is possible that both reduction of loss and reduction of noise are not sufficiently considered. Therefore, in the inventions described in Patent Documents 1 to 5, there is a possibility that the optimum carrier frequency that achieves both reduction of loss and reduction of noise is not set.

The present invention has been made in view of such circumstances, and sets an optimum carrier frequency for reducing the loss of the power converter, the loss of the electric motor, and the noise associated with the driving of the power converter and the electric motor. An object of the present invention is to provide an electric motor control device capable of opening and closing a power converter.

The electric motor control device according to the present invention includes a power converter that converts DC power into AC power by controlling the opening and closing of a plurality of switching elements and supplies the converted AC power to the electric motor, and the power converter. A motor control device including a control device for controlling the opening and closing of the plurality of switching elements by pulse width modulation control, wherein the control device at least factors a carrier frequency which is a carrier frequency in the pulse width modulation control. According to the response curved surface method included in, the power converter loss estimated value which is the loss estimated value of the power converter, the electric motor loss estimated value which is the loss estimated value of the electric motor, and the driving of the power converter and the electric motor. Based on the loss noise estimation unit that calculates the drive noise estimation value, which is the noise estimation value, the power converter loss estimation value, the motor loss estimation value, and the drive noise estimation value calculated by the loss noise estimation unit. A carrier frequency control unit that calculates a determination value for determining an appropriate value of the carrier frequency and outputs the carrier frequency when the determination value is minimized, and the carrier frequency output from the carrier frequency control unit are used. It is provided with a pulse width modulation signal generation unit that generates open / close signals of the plurality of switching elements of the power converter.

According to the electric motor control device according to the present invention, the control device includes a loss noise estimation unit, a carrier frequency control unit, and a pulse width modulation signal generation unit. As a result, the electric motor control device according to the present invention sets an optimum carrier frequency for reducing the loss of the power converter, the loss of the electric motor, and the noise associated with the driving of the power converter and the electric motor, and sets the power converter. Opening and closing can be controlled.

It is a block diagram which shows an example of the electric motor control device 10. It is a block diagram which shows an example of a rotor 52. It is a block diagram which shows an example of a control device 60. It is a block diagram which shows an example of the control block of the control device 60. It is a figure which shows an example of the relationship between a torque and an armature current. It is a figure which shows an example of the relationship between the motor loss estimate Lm and the carrier frequency Fc. It is a figure which shows an example of the relationship between the power converter loss estimate Li and the carrier frequency Fc. It is a figure which shows an example of the relationship between the drive noise estimated value Ls and the carrier frequency Fc. It is a figure which shows an example of the relationship between a characteristic Def and a carrier frequency Fc. It is a figure which shows an example of the relationship between the motor loss estimated value Lm and torque Trq. It is a figure which shows an example of the relationship between the power converter loss estimate Li and torque Trq. It is a figure which shows an example of the relationship between the driving noise estimated value Ls and torque Trq. It is a figure which shows an example of the relationship between a characteristic Def and a torque Trq. It is a figure which shows an example of the relationship between the motor loss estimated value Lm and the rotation speed Nm. It is a figure which shows an example of the relationship between the power converter loss estimate Li and the rotation speed Nm. It is a figure which shows an example of the relationship between the drive noise estimated value Ls and the rotation speed Nm. It is a figure which shows an example of the relationship between a characteristic Def and a rotation speed Nm. It is a figure which shows an example of the relationship between the motor loss estimated value Lm, and DC voltage Vdc. It is a figure which shows an example of the relationship between the power converter loss estimate value Li and DC voltage Vdc. It is a figure which shows an example of the relationship between the drive noise estimated value Ls and DC voltage Vdc. It is a figure which shows an example of the relationship between a characteristic Def and a DC voltage Vdc. It is a figure which shows an example of the relationship between a carrier frequency Fc, a torque Trq, and a characteristic Def with respect to the motor loss estimated value Lm. It is a figure which shows an example of the relationship between a carrier frequency Fc, a torque Trq, and a characteristic Def concerning the power converter loss estimate value Li. It is a figure which shows an example of the relationship between a carrier frequency Fc, a torque Trq, and a characteristic Def with respect to the driving noise estimated value Ls. It is a flowchart which shows an example of the control procedure in loss noise estimation unit 76 and carrier frequency control unit 77.

Hereinafter, embodiments of the present invention will be described with reference to the drawings. The drawings are conceptual drawings and do not specify the dimensions of the detailed structure.

<Overview of power conversion system 1>
As shown in FIG. 1, the power conversion system 1 of the present embodiment includes an electric motor control device 10, a DC power supply 20, and a smoothing capacitor 30. Further, the electric motor control device 10 includes a power converter 40 and a control device 60. The electric motor 50 is electrically connected to the power converter 40.

(DC power supply 20)
The DC power supply 20 outputs DC power. The DC power supply 20 is not limited as long as it can output DC power. As the DC power source 20, for example, a lead storage battery (battery), a lithium ion battery, a power generation device (for example, a fuel cell), or the like can be used. Further, the DC power supply 20 can also generate DC power by using a known AC generator or the like. In this case, the DC power supply 20 can generate DC power by rectifying and smoothing the AC power output from the AC generator by using a known rectifying circuit, smoothing circuit, or the like.

Further, the DC power supply 20 can boost low-voltage DC power by using, for example, a known boost converter. In this case, as the DC power supply 20, for example, a non-isolated boost converter such as a known boost chopper converter can be used. Further, as the DC power supply 20, for example, an isolated boost converter such as a known flyback type converter or forward type converter can be used.

(Smoothing capacitor 30)
The smoothing capacitor 30 smoothes the DC power output from the DC power supply 20. The positive electrode side 20p of the DC power supply 20 is connected to the positive electrode side 30p of the smoothing capacitor 30. The negative electrode side 20n of the DC power supply 20 is connected to the negative electrode side 30n of the smoothing capacitor 30, and is connected to the power ground (reference potential of the circuit on the high voltage side including the DC power supply 20). As the smoothing capacitor 30, for example, an electrolytic capacitor can be used. The DC power supplied from the DC power supply 20 is smoothed by the smoothing capacitor 30 to reduce ripple.

(Power converter 40)
The power converter 40 has DC power (in this embodiment, DC power smoothed by the smoothing capacitor 30) by controlling the opening and closing of a plurality of switching elements (three sets of pair of switching elements 41 in this embodiment). ) Is converted into AC power, and the converted AC power is supplied to the electric motor 50. As shown in FIG. 1, three sets of a pair of switching elements 41 are fully bridge-connected. In each of the three sets of the pair of switching elements 41, the positive electrode side switching element 4xp connected to the positive electrode side 30p of the smoothing capacitor 30 and the negative electrode side switching element 4xn connected to the negative electrode side 30n of the smoothing capacitor 30 are connected in series. Has been done. The power converter 40 of the present embodiment is a three-phase power converter, and x is any one of u, v, and w. For example, the positive electrode side switching element 4up indicates a U-phase positive electrode side switching element, and the negative electrode side switching element 4un indicates a U phase negative electrode side switching element.

As the positive electrode side switching element 4xp and the negative electrode side switching element 4xn, known power switching elements can be used. As the positive side switching element 4xp and the negative side switching element 4xn, for example, a known insulated gate bipolar transistor (IGBT: Integrated Gate Bipolar Transistor), field effect transistor (FET: Field Effect Transistor), or the like can be used.

As shown in FIG. 1, each of the plurality (three) positive electrode side switching elements 4xp includes a control terminal 4g, an input terminal 4c, an output terminal 4e, and a freewheeling diode 4d. For example, in an insulated gate bipolar transistor (IGBT), the control terminal 4g corresponds to the gate terminal, the input terminal 4c corresponds to the collector terminal, and the output terminal 4e corresponds to the emitter terminal. The control terminal 4g is connected to the control device 60 via the drive circuit 61b. Each of the plurality (three) positive electrode side switching elements 4xp is open / closed controlled based on the drive signal output from the control device 60.

The voltage between the control terminal 4g and the output terminal 4e is defined as the control voltage Vge. For example, when the control voltage Vge is at a low level (a state of a predetermined voltage value or less), the input terminal 4c and the output terminal 4e are controlled to be electrically cut off in an open state. On the other hand, when the control voltage Vge is at a high level (a state exceeding a predetermined voltage value), the input terminal 4c and the output terminal 4e are controlled to be electrically conducted in a closed state.

As the freewheeling diode 4d, for example, a body diode (parasitic diode) of a switching element can be used. Further, instead of the body diode, a freewheeling diode may be separately provided and connected in parallel between the input terminal 4c and the output terminal 4e. The freewheeling diode 4d forms a current path from the output terminal 4e side to the input terminal 4c side when the switching element is in the open state. As a result, the switching element can be protected from the reverse current generated by opening and closing the switching element.

The same applies to the plurality (three) positive electrode side switching elements 4xp for the plurality (three) negative electrode side switching elements 4xn. The control device 60 controls the power converter 40 by opening and closing a plurality of switching elements (three sets of a pair of switching elements 41) of the power converter 40.

For example, the power converter 40 includes a positive electrode side switching element 4mx of one of a plurality (three) positive electrode side switching elements 4xp and a plurality (three) negative electrode side switching elements based on a command of the control device 60. The negative electrode side switching element 4xn of one of the 4xn is closed, and the other switching element is opened. The phases (U phase, V phase, W phase) of one positive electrode side switching element 4xp and one negative electrode side switching element 4xn to be closed are different. By sequentially changing the combination of switching elements that the control device 60 closes, the power converter 40 can convert the DC power smoothed by the smoothing capacitor 30 into AC power.

As shown in FIG. 1, an output terminal 42x is provided between the positive electrode side switching element 4xp and the negative electrode side switching element 4xn. The output terminal 42x and the phase terminal 43x of the electric motor 50 are electrically connected by a power cable 44x. The power cable 44x supplies the AC power converted by the power converter 40 to the electric motor 50. Note that x is any one of u, v, and w.

(Electric motor 50)
The electric motor 50 includes a stator 51 and a rotor 52. The electric motor 50 is, for example, a radial gap type cylindrical electric motor in which the stator 51 and the rotor 52 are coaxially arranged. The electric motor 50 may be an axial void type cylindrical motor. Further, the electric motor 50 may be an inner type cylindrical electric motor or an outer type cylindrical electric motor. In the inner type cylindrical motor, the rotor 52 is provided inside the stator 51 (on the axial side of the motor 50). In the outer type cylindrical electric motor, the rotor 52 is provided on the outside of the stator 51.

The stator 51 includes a stator core (not shown) in which a plurality of slots are formed, and armature windings (U-phase coil 51u, V-phase coil 51v, W-phase coil 51w). The stator core is formed by laminating a plurality of thin plate-shaped electromagnetic steel sheets (for example, silicon steel sheets) in the axial direction. Armature windings are wound around the plurality of slots. The armature winding is formed by winding a conductor (coil) such as copper, and the surface of the conductor is covered with an insulating layer such as enamel. The cross-sectional shape of the armature winding is not limited, and any cross-sectional shape can be used. For example, as the armature winding, conductors (coils) having various cross-sectional shapes such as a round wire having a circular cross-section and a square wire having a polygonal cross-section can be used. Further, as the armature winding, a parallel thin wire obtained by combining a plurality of thinner coil strands can also be used. When the parallel thin wire is used, the eddy current loss generated in the armature winding is reduced as compared with the case of the single wire, and the efficiency of the motor 50 is improved. Further, since the force required for coil forming can be reduced, the formability of the coil is improved and the coil production becomes easy.

The armature winding can be wound by a known method such as distributed winding (for example, concentric winding, corrugated winding, lap winding, etc.) or centralized winding. Further, as shown in FIG. 1, the armature windings (U-phase coil 51u, V-phase coil 51v, W-phase coil 51w) can be connected by Y connection. In the figure, the neutral point is indicated by the neutral point 51n. The armature windings (U-phase coil 51u, V-phase coil 51v, W-phase coil 51w) can also be connected by Δ connection.

As shown in FIG. 2, the rotor 52 includes a rotor core 52a, a plurality of (eight in this embodiment) permanent magnets 52b, and a shaft 52c. The figure is a schematic view of the rotor 52 viewed in the axial direction (the direction perpendicular to the paper surface of the figure), and schematically shows the arrangement thereof. The rotor core 52a is formed in a columnar shape by laminating a plurality of thin plate-shaped electromagnetic steel plates (for example, silicon steel plates) in the axial direction (vertical to the paper surface in the figure). A shaft 52c is provided on the rotor core 52a, and the shaft 52c penetrates the axial center of the rotor core 52a along the axial direction. Both ends of the shaft 52c in the axial direction are rotatably supported by bearing members (not shown) such as bearings.

A plurality of (eight) permanent magnets 52b are embedded in the rotor core 52a. Specifically, the rotor core 52a is provided with a plurality of magnet accommodating portions (not shown) at equal intervals in the circumferential direction. Permanent magnets 52b of a predetermined number of magnetic pole pairs (four magnetic pole pairs in this embodiment, eight) are embedded in the plurality of magnet accommodating portions. As the permanent magnet 52b, for example, a known ferrite magnet or rare earth magnet can be used. The manufacturing method of the permanent magnet 52b is not limited. As the permanent magnet 52b, for example, a resin bond magnet or a sintered magnet can be used. The resin-bonded magnet can be formed by, for example, mixing a ferrite-based raw material magnet powder and a resin, and casting the resin-bonded magnet into the rotor core 52a by injection molding or the like. The sintered magnet can be formed, for example, by forming a rare earth-based raw material magnet powder under pressure in a magnetic field and baking it at a high temperature. The number of slots of the stator 51 and the number of magnetic poles of the rotor 52 are not limited.

(Control device 60)
The control device 60 controls the power conversion system 1 including the power converter 40. More specifically, the control device 60 controls the opening and closing of a plurality of switching elements (three sets of a pair of switching elements 41) of the power converter 40 by pulse width modulation (PWM) control. As shown in FIG. 3, the control device 60 includes a known central processing unit 60a, a storage device 60b, and an input / output interface 60c, which are connected via a bus 60d.

The central arithmetic unit 60a is a CPU: Central Processing Unit, and can perform various arithmetic processes. The storage device 60b includes a first storage device 60b1 and a second storage device 60b2. The first storage device 60b1 is a volatile storage device (RAM: Random Access Memory) capable of reading and writing, and the second storage device 60b2 is a read-only non-volatile storage device (ROM: Read Only Memory). is there.

Further, as shown in FIG. 1, the control device 60 includes a DC voltage detector 61a, a drive circuit 61b, a current detector 61c, and a position detector 61d. The DC voltage detector 61a detects the DC voltage of the DC power smoothed by the smoothing capacitor 30. Specifically, the DC voltage detector 61a divides the DC voltage by, for example, a plurality of resistors having known resistance values, and outputs the divided DC voltage to the control device 60. The control device 60 knows the DC voltage value divided by a known A / D converter (not shown) or the like, and inputs the DC voltage of the DC power smoothed by the smoothing capacitor 30 (input to the power converter 40). (DC voltage) can be known.

The drive circuit 61b is a drive circuit that amplifies the drive signal output from the control device 60, and for example, a known driver circuit can be used. Note that in FIG. 1, the connection between the control terminal 4g of each switching element of the power converter 40 and the drive circuit 61b is omitted.

The current detector 61c detects the output current output from the power converter 40. In the present embodiment, the current detector 61c is provided in the power cable 44u and the power cable 44v, and detects the U-phase current Iu and the V-phase current Iv. In the present specification, the U-phase current Iu detected by the current detector 61c is referred to as a U-phase current detection value Iu_fb, and the V-phase current Iv detected by the current detector 61c is referred to as a V-phase current detection value Iv_fb. The W-phase current Iw can be calculated by subtracting the U-phase current detection value Iu_fb and the V-phase current detection value Iv_fb from 0, respectively. As the current detector 61c, a known current detector (for example, a current detector using a current transformer, a current detector using a shunt resistor, or the like) can be used.

The position detector 61d detects the position of the rotor 52 with respect to the stator 51. As the position detector 61d, a known position detector (for example, a resolver, an encoder, a hall sensor, etc.) can be used. The control device 60 may be provided with various detectors other than the above-mentioned detectors.

The central processing unit 60a shown in FIG. 3 reads the control program of the power converter 40 stored in the second storage device 60b2 into the first storage device 60b1 and executes the control program. Further, the above-mentioned detected values and the like are input to the control device 60 via the insulating portion (not shown) and the input / output interface 60c. The central processing unit 60a outputs an open / close signal to each switching element of the power converter 40 via the input / output interface 60c, the insulating portion (not shown), and the drive circuit 61b shown in FIG. 1, and causes the power converter 40 to operate. Open / close control. The insulating portion electrically insulates the circuit on the low voltage side including the control device 60 and the circuit on the high voltage side including the DC power supply 20. For the insulating portion, for example, a known photocoupler or the like can be used.

<Electric motor control device 10>
As shown in FIG. 4, when the control device 60 of the present embodiment is regarded as a control block, the three-phase / two-phase conversion unit 71, the rotation speed calculation unit 72, the current command value setting unit 73, and the current control unit It includes a 74, a two-phase / three-phase conversion unit 75, a loss noise estimation unit 76, a carrier frequency control unit 77, and a pulse width modulation signal generation unit 78.

As shown in FIG. 2, the main magnetic flux direction of the permanent magnet 52b is the d-axis direction, and the direction electrically orthogonal to the d-axis direction is the q-axis direction. The voltage equation in the dq coordinate system of the electric motor 50 can be expressed by the following equation 1.

However, the voltage in the d-axis direction is represented by the d-axis voltage Vd, and the voltage in the q-axis direction is represented by the q-axis voltage Vq. Further, the winding resistance of each phase coil of the armature winding (U-phase coil 51u, V-phase coil 51v, W-phase coil 51w) is represented by the winding resistance R. Further, the d-axis inductance, which is the inductance in the d-axis direction, is represented by the d-axis inductance Ld, and the q-axis inductance, which is the inductance in the q-axis direction, is represented by the q-axis inductance Lq. Further, the current in the d-axis direction is represented by the d-axis current Id, and the current in the q-axis direction is represented by the q-axis current Iq. Further, the angular velocity of the rotor 52 is represented by the angular velocity ω, and the induced voltage constant is represented by the induced voltage constant Φ. The differential operator is represented by the differential operator p (= d / dt).

The control device 60 controls the power converter 40 by setting the d-axis current command value Id_ref and the q-axis current command value Iq_ref, respectively. More specifically, the control device 60 controls the power converter 40 by setting the d-axis voltage command value Vd_ref and the q-axis voltage command value Vq_ref based on the d-axis current command value Id_ref and the q-axis current command value Iq_ref. To do. However, the d-axis current command value Id_ref refers to the current command value in the d-axis direction, and the q-axis current command value Iq_ref refers to the current command value in the q-axis direction. The d-axis voltage command value Vd_ref refers to the voltage command value in the d-axis direction, and the q-axis voltage command value Vq_ref refers to the voltage command value in the q-axis direction.

(Three-phase / two-phase converter 71)
The three-phase / two-phase conversion unit 71 calculates the d-axis current calculated value Id_fb and the q-axis current calculated value Iq_fb. The d-axis current calculated value Id_fb refers to the current calculated value in the d-axis direction, and the q-axis current calculated value Iq_fb refers to the current calculated value in the q-axis direction. As shown in FIG. 4, the U-phase current detection value Iu_fb and the V-phase current detection value Iv_fb detected by the current detector 61c are input to the three-phase / two-phase conversion unit 71. As described above, the W-phase current Iw can be calculated by subtracting the U-phase current detection value Iu_fb and the V-phase current detection value Iv_fb from 0, respectively. Let the detected value (calculated value) of the W-phase current Iw be the W-phase current detected value Iw_fb. Further, the rotation angle (rotation position θ) of the rotor 52 detected by the position detector 61d is input to the three-phase / two-phase conversion unit 71.

The three-phase / two-phase converter 71 uses the U-phase current detection value Iu_fb, the V-phase current detection value Iv_fb, the W-phase current detection value Iw_fb, and the rotation position θ to calculate the d-axis current based on the following equation 2. The value Id_fb and the q-axis current calculated value Iq_fb are calculated. The rotation position θ can also be estimated from, for example, changes over time in the U-phase current detection value Iu_fb, the V-phase current detection value Iv_fb, and the W-phase current detection value Iw_fb. Further, the rotation position θ can be estimated from, for example, a change over time in each phase voltage of the armature winding (U-phase coil 51u, V-phase coil 51v, W-phase coil 51w).

(Rotation speed calculation unit 72)
The rotation speed calculation unit 72 calculates the rotation speed of the rotor 52. As shown in FIG. 4, the rotation speed calculation unit 72 is input with the rotation angle (rotation position θ) of the rotor 52 detected by the position detector 61d. The rotation speed calculation unit 72 can calculate the rotation speed of the rotor 52 by time-differentiating the rotation position θ, for example. The control device 60 can also measure the rotation speed of the rotor 52 by using a known rotation speed detector. The rotation speed of the rotor 52 calculated by the rotation speed calculation unit 72 is represented by the rotation speed calculation value Nm_fb. The rotation speed calculation value Nm_fb calculated by the rotation speed calculation unit 72 is output to the loss noise estimation unit 76 described later.

(Current command value setting unit 73)
The current command value setting unit 73 calculates the d-axis current command value Id_ref and the q-axis current command value Iq_ref. The calculation method of the d-axis current command value Id_ref and the q-axis current command value Iq_ref is not limited. It is preferable that the current command value setting unit 73 sets the d-axis current command value Id_ref and the q-axis current command value Iq_ref so that the armature current is minimized with respect to the required torque.

FIG. 5 shows an example of the relationship between torque and armature current. The curve L11 shows an example of the relationship between the d-axis current command value Id_ref and the q-axis current command value Iq_ref (relationship of the armature current current vector) when the required torque (torque command value Trq_ref) is constant. The required torque is obtained by setting the current vector of the armature current on the curve L11. The arrow L12 indicates a current vector that minimizes the armature current when the required torque indicated by the curve L11 is obtained. The straight line L13 shows a tangent line at the extremum of the curve L11. That is, when the current vector of the armature current is orthogonal to the tangent line indicated by the straight line L13, the armature current for obtaining the required torque is minimized.

When the required torque increases, the torque command value Trq_ref is increased, and the curve L11 moves to the curve L11a. The arrow L12a indicates the current vector at which the armature current is minimized at this time. On the other hand, when the required torque decreases, the torque command value Trq_ref is reduced, and the curve L11 moves to the curve L11b. The arrow L12b indicates the current vector at which the armature current is minimized at this time. The horizontal axis of the figure shows the d-axis, and the vertical axis shows the q-axis.

The control device 60 sets the torque command value Trq_ref according to the required torque. As shown in FIG. 4, the torque command value Trq_ref is input to the current command value setting unit 73. The current command value setting unit 73 sets the d-axis current command value Id_ref and the q-axis current command value Iq_ref so that the armature current is minimized with respect to the torque command value Trq_ref. The d-axis current command value Id_ref and the q-axis current command value Iq_ref corresponding to the required torque (torque command value Trq_ref) are calculated in advance and converted into, for example, a map, a table, a relational expression (polynomial), or the like, and are shown in FIG. It is stored in the second storage device 60b2 shown. The current command value setting unit 73 reads a map or the like from the second storage device 60b2 to the first storage device 60b1 at startup together with the control program of the power converter 40. As a result, the current command value setting unit 73 can easily set the d-axis current command value Id_ref and the q-axis current command value Iq_ref corresponding to the required torque (torque command value Trq_ref).

(Current control unit 74)
The current control unit 74 calculates the d-axis voltage command value Vd_ref based on the d-axis current command value Id_ref. Further, the current control unit 74 calculates the q-axis voltage command value Vq_ref based on the q-axis current command value Iq_ref. Specifically, the d-axis voltage command value Vd_ref is calculated so that the d-axis current calculated value Id_fb matches the d-axis current command value Id_ref. Further, the q-axis voltage command value Vq_ref is calculated so that the q-axis current calculated value Iq_fb matches the q-axis current command value Iq_ref. The current control unit 74 is not limited as long as it can calculate the d-axis voltage command value Vd_ref and the q-axis voltage command value Vq_ref.

As shown in FIG. 4, in the present embodiment, the current control unit 74 includes a subtractor 74a, a PI control unit 74b, a subtractor 74c, and a PI control unit 74d. The d-axis current command value Id_ref and the d-axis current calculated value Id_fb are input to the subtractor 74a. The subtractor 74a calculates the deviation ΔId by subtracting the d-axis current calculation value Id_fb from the d-axis current command value Id_ref. The deviation ΔId calculated by the subtractor 74a is output to the PI control unit 74b. In the present embodiment, the PI control unit 74b performs proportional control and integral control so that the d-axis current calculated value Id_fb matches the d-axis current command value Id_ref.

The PI control unit 74b includes a proportional calculation unit 74b1, an integration calculation unit 74b2, and an adder 74b3. The proportional calculator 74b1 outputs a calculation result obtained by multiplying the deviation ΔId by the proportional gain Kpd. The integral calculator 74b2 outputs a calculation result obtained by multiplying the integral value obtained by integrating the deviation ΔId by the integral gain Kid. The adder 74b3 adds the calculation result of the proportional calculation unit 74b1 and the calculation result of the integration calculation unit 74b2. Then, the PI control unit 74b outputs the calculation result of the adder 74b3 as the d-axis voltage command value Vd_ref. The PI control unit 74b may also include a differential calculator that outputs a calculation result obtained by multiplying the differential value obtained by differentiating the deviation ΔId by the differential gain. That is, the PI control unit 74b can be a PID control unit that performs proportional control, integral control, and differential control. In this case, the adder 74b3 adds the calculation result of the proportional calculation unit 74b1, the calculation result of the integration calculation unit 74b2, and the calculation result of the differential calculation unit.

On the other hand, the q-axis current command value Iq_ref and the q-axis current calculation value Iq_fb are input to the subtractor 74c. The subtractor 74c calculates the deviation ΔIq by subtracting the q-axis current calculation value Iq_fb from the q-axis current command value Iq_ref. The deviation ΔIq calculated by the subtractor 74c is output to the PI control unit 74d. In the present embodiment, the PI control unit 74d performs proportional control and integral control so that the q-axis current calculated value Iq_fb matches the q-axis current command value Iq_ref.

The PI control unit 74d includes a proportional calculation unit 74d1, an integration calculation unit 74d2, and an adder 74d3. The proportional calculator 74d1 outputs a calculation result obtained by multiplying the deviation ΔIq by the proportional gain Kpq. The integral calculator 74d2 outputs a calculation result obtained by multiplying the integral value obtained by integrating the deviation ΔIq by the integral gain Kiq. The adder 74d3 adds the calculation result of the proportional calculation unit 74d1 and the calculation result of the integration calculation unit 74d2. Then, the PI control unit 74d outputs the calculation result of the adder 74d3 as the q-axis voltage command value Vq_ref. Similar to the PI control unit 74b, the PI control unit 74d may also include a differential calculator that outputs a calculation result obtained by multiplying the differential value obtained by differentiating the deviation ΔIq by the differential gain.

As described above, the current control unit 74 is subjected to, for example, at least proportional control (P control) and integral control (I control) of proportional control (P control), integral control (I control) and differential control (D control). , The d-axis voltage command value Vd_ref and the q-axis voltage command value Vq_ref can be calculated. The proportional gain Kpd and the integrated gain Kid, and the proportional gain Kpq and the integrated gain Kiq are stored in the second storage device 60b2 shown in FIG. These control gains are read from the second storage device 60b2 to the first storage device 60b1 at startup together with the control program of the power converter 40.

The larger the proportional gain Kpd, the shorter the deviation ΔId can be. Further, as the integrated gain Kid is increased, the offset (steady state deviation) due to the deviation ΔId can be eliminated in a short time. Further, as the differential gain is increased, the vibration of the deviation ΔId can be converged in a short time. These control gains may be acquired in advance by, for example, simulation or adjustment by an actual machine. The same can be said for the control gains (proportional gain Kpq, integral gain Kiq and differential gain) for calculating the q-axis voltage command value Vq_ref.

The d-axis voltage command value Vd_ref can be expressed by the following equation 3 using the proportional gain Kpd and the integrated gain Kid. Further, the q-axis voltage command value Vq_ref can be expressed by the following equation 4 using the proportional gain Kpq and the integral gain Kiq. In each case, the Laplace operator is represented by the Laplace operator s.

(Two-phase / three-phase converter 75)
The two-phase / three-phase conversion unit 75 has a three-phase voltage command value (U-phase voltage command value Vu_ref, V-phase voltage command) from a two-phase voltage command value (d-axis voltage command value Vd_ref and q-axis voltage command value Vq_ref). The value Vv_ref and the W-phase voltage command value Vw_ref) are calculated. As shown in FIG. 4, the d-axis voltage command value Vd_ref and the q-axis voltage command value Vq_ref calculated by the current control unit 74 are input to the two-phase / three-phase conversion unit 75. Further, the rotation angle (rotation position θ) of the rotor 52 detected by the position detector 61d is input to the two-phase / three-phase conversion unit 75. The two-phase / three-phase conversion unit 75 performs inverse conversion of the dq coordinate axes using the d-axis voltage command value Vd_ref, the q-axis voltage command value Vq_ref, and the rotation position θ, and performs U-phase voltage command values Vu_ref and V-phase voltage commands. The value Vv_ref and the W-phase voltage command value Vw_ref are calculated.

(Loss noise estimation unit 76)
The loss noise estimation unit 76 calculates the power converter loss estimation value Li, the motor loss estimation value Lm, and the drive noise estimation value Ls by the response surface methodology including the carrier frequency Fc at least as a factor. The power converter loss estimate Li refers to the loss estimate of the power converter 40. The loss of the power converter 40 corresponds to the power difference between the input power and the output power of the power converter 40. The motor loss estimated value Lm refers to the loss estimated value of the motor 50. The loss of the electric motor 50 corresponds to the power difference between the input power and the output power of the electric motor 50. The driving noise estimated value Ls refers to the noise estimated value associated with the driving of the power converter 40 and the electric motor 50. In the present embodiment, the noise associated with driving the power converter 40 and the electric motor 50 includes noise in the frequency band of the carrier frequency Fc.

The response surface methodology (RSM: Response Surface Methodology) is a method of approximating discrete data with a continuous curved surface, and the created approximate expression is called a response surface model. By creating a response surface model, the characteristics can be visualized, and it becomes easy to study factors (design variables) that have a large influence. Examples include polynomial approximation by the least squares method, RBF (Radial Basis Function) that interpolates discrete data, and the like. In the present embodiment, a method based on polynomial approximation will be described as an example, but the present invention is not limited thereto.

Moreover, it is difficult to acquire sampling data for all conditions. Therefore, design of experiments is used in the acquisition of sampling data. Design of experiments can efficiently acquire a large amount of information with a small number of experiments (measurement data). An example is the L27 orthogonal array. The L27 orthogonal array is a three-level orthogonal array, and sampling data is acquired under 27 conditions. It should be noted that the sampling data may be acquired a plurality of times (for example, three times) for the 27 conditions shown in the L27 orthogonal array, and the variation between the sampling data may be taken into consideration. Further, the fitting degree between the sampling data and the response surface model, for example, using indicators such as known coefficient of determination R ^2, can also be verified. As described above, various known methods can be used for the acquisition of sampling data by the design of experiments and the calculation method of the optimum solution by the response surface methodology (RSM).

In the present embodiment, at least the carrier frequency Fc, which is the frequency of the carrier wave in the pulse width modulation control (PWM control), is included in the factor. Further, the loss noise estimation unit 76 includes a torque command value Trq_ref of the electric motor 50, a rotation speed detection value or a rotation speed calculation value Nm_fb of the electric motor 50, a DC voltage detection value Vdc_fb of DC power input to the power converter 40, and It is preferable to use the carrier frequency Fc as a factor.

The number of switchings of a plurality of switching elements (in this embodiment, three sets of a pair of switching elements 41) increases or decreases according to the carrier frequency Fc. Therefore, the loss (switching loss) of the power converter 40 increases or decreases according to the increase or decrease of the carrier frequency Fc. Further, the noise associated with driving the power converter 40 and the electric motor 50 (particularly, the noise associated with driving the power converter 40) fluctuates according to the carrier frequency Fc. In the present embodiment, the carrier frequency Fc is at least a factor because the optimum carrier frequency Fc for reducing the loss of the power converter 40 and the loss of the motor 50 and the noise associated with the driving of the power converter 40 and the motor 50 is set. Must be included in.

Further, examples of the loss of the electric motor 50 include copper loss, iron loss and mechanical loss. The armature current increases or decreases according to the output torque (torque command value Trq_ref) of the electric motor 50. When the armature current increases or decreases, the copper loss increases or decreases, and the loss of the motor 50 increases or decreases. Further, the counter electromotive force (induced voltage) increases or decreases according to the rotation speed (rotation speed detection value or rotation speed calculation value Nm_fb) of the electric motor 50. When the counter electromotive force increases or decreases, the iron loss increases or decreases, and the loss of the motor 50 increases or decreases. Therefore, it is preferable that the loss noise estimation unit 76 uses the torque command value Trq_ref of the electric motor 50, the rotation speed detection value of the electric motor 50, or the rotation speed calculation value Nm_fb as factors.

As will be described later, the modulation factor in the pulse width modulation control (PWM control) is calculated based on the DC voltage detection value Vdc_fb of the DC power input to the power converter 40. Then, based on the modulation factor and the carrier wave (triangle wave), open / close signals of a plurality of switching elements (three sets of a pair of switching elements 41 in this embodiment) are generated. Therefore, even if the carrier frequency Fc is the same, the open state time of the plurality of switching elements fluctuates according to the DC voltage (DC voltage detection value Vdc_fb), and the loss (switching loss) of the power converter 40 increases or decreases. To do. Therefore, it is preferable that the loss noise estimation unit 76 uses the DC voltage detection value Vdc_fb of the DC power input to the power converter 40 as a factor.

As shown in FIG. 4, the loss noise estimation unit 76 is provided with a torque command value Trq_ref set by the control device 60, a rotation speed calculation value Nm_fb calculated by the rotation speed calculation unit 72, and a DC voltage detector 61a. The detected DC voltage detection value Vdc_fb is input. Further, the loss noise estimation unit 76 includes a frequency table (not shown) in the first storage device 60b1 shown in FIG. In the frequency table, carrier frequencies Fc are stored in ascending order at a predetermined pitch (for example, a pitch of several kilohertz). The loss noise estimation unit 76 sequentially selects one carrier frequency Fc from a plurality of carrier frequency Fcs stored in the frequency table. Then, the loss noise estimation unit 76 uses the selected carrier frequency Fc and the above-mentioned input values (torque command value Trq_ref, rotation speed calculation value Nm_fb, and DC voltage detection value Vdc_fb) to obtain a response curved surface model (described later). (Including polynomials).

FIG. 6A shows an example of the relationship between the motor loss estimate Lm and the carrier frequency Fc. The vertical axis of the figure shows the motor loss estimated value Lm, and the horizontal axis shows the carrier frequency Fc. The curve L21 shows an example of fluctuation of the motor loss estimated value Lm calculated by the response surface methodology. FIG. 6B shows an example of the relationship between the power converter loss estimate Li and the carrier frequency Fc. The vertical axis of the figure shows the power converter loss estimated value Li, and the horizontal axis shows the carrier frequency Fc. The curve L22 shows an example of fluctuation of the power converter loss estimated value Li calculated by the response surface methodology. FIG. 6C shows an example of the relationship between the drive noise estimate Ls and the carrier frequency Fc. The vertical axis of the figure shows the drive noise estimated value Ls, and the horizontal axis shows the carrier frequency Fc. The curve L23 shows an example of fluctuation of the driving noise estimated value Ls calculated by the response surface methodology. FIG. 6D shows an example of the relationship between the characteristic Def and the carrier frequency Fc. The vertical axis of the figure shows the characteristic Def, and the horizontal axis shows the carrier frequency Fc. Region DZ1 shows an example of the characteristic Def calculated by the response surface methodology.

As shown in the curve L21 of FIG. 6A, the fluctuation range of the motor loss estimated value Lm with respect to the increase / decrease of the carrier frequency Fc is smaller than the fluctuation range of the power converter loss estimated value Li and the drive noise estimated value Ls. The first motor loss estimated value Lm1 indicates the motor loss estimated value Lm when the carrier frequency Fc is the first carrier frequency Fc1. As shown in the curve L22 of FIG. 6B, the power converter loss estimated value Li has a minimum value with respect to the increase / decrease of the carrier frequency Fc. The first power converter loss estimated value Li1 indicates the power converter loss estimated value Li when the carrier frequency Fc is the first carrier frequency Fc1, and corresponds to a minimum value. As shown in the curve L23 of FIG. 6C, the drive noise estimated value Ls decreases as the carrier frequency Fc increases. The first drive noise estimated value Ls1 indicates the drive noise estimated value Ls when the carrier frequency Fc is the first carrier frequency Fc1.

As shown in the region DZ1 of FIG. 6D, as the carrier frequency Fc increases, the characteristic value (characteristic Def) gradually decreases up to the first carrier frequency Fc1. Even if the carrier frequency Fc increases from the first carrier frequency Fc1, the fluctuation range of the characteristic value (characteristic Def) is small and substantially constant. As described above, the characteristics when the protrusion of the region DZ1 is the lowest are the desirable characteristics that the loss of the power converter 40 and the loss of the electric motor 50 and the noise associated with the driving of the power converter 40 and the electric motor 50 are reduced. It can be said that. In this way, the loss noise estimation unit 76 can acquire the optimum carrier frequency Fc (first carrier frequency Fc1), and the power converter loss estimation value Li (first power converter loss estimation value) at this time. Li1), the electric motor loss estimated value Lm (first electric motor loss estimated value Lm1) and the driving noise estimated value Ls (first driving noise estimated value Ls1) can be acquired.

The loss noise estimation unit 76 can also obtain characteristics for torque Trq, rotation speed Nm, and DC voltage Vdc. FIG. 7A shows an example of the relationship between the motor loss estimated value Lm and the torque Trq. The vertical axis of the figure shows the motor loss estimated value Lm, and the horizontal axis shows the torque Trq. The curve L31 shows an example of fluctuation of the motor loss estimated value Lm calculated by the response surface methodology. FIG. 7B shows an example of the relationship between the power converter loss estimate Li and the torque Trq. The vertical axis of the figure shows the power converter loss estimated value Li, and the horizontal axis shows the torque Trq. Curve L32 shows a variation example of the power converter loss estimated value Li calculated by the response surface methodology. FIG. 7C shows an example of the relationship between the drive noise estimated value Ls and the torque Trq. The vertical axis of the figure shows the drive noise estimated value Ls, and the horizontal axis shows the torque Trq. The curve L33 shows an example of fluctuation of the driving noise estimated value Ls calculated by the response surface methodology. FIG. 7D shows an example of the relationship between the characteristic Def and the torque Trq. The vertical axis of the figure shows the characteristic Def, and the horizontal axis shows the torque Trq. Region DZ2 shows an example of the characteristic Def calculated by the response surface methodology.

As shown in the curve L31 of FIG. 7A, the motor loss estimated value Lm increases as the torque Trq increases. The first motor loss estimated value Lm1 indicates the motor loss estimated value Lm when the torque Trq is the first torque Trq1. As shown in the curve L32 of FIG. 7B, the power converter loss estimated value Li increases as the torque Trq increases. The first power converter loss estimated value Li1 indicates the power converter loss estimated value Li when the torque Trq is the first torque Trq1. As shown in the curve L33 of FIG. 7C, the fluctuation range of the drive noise estimated value Ls with respect to the increase / decrease of the torque Trq is smaller than the fluctuation range of the power converter loss estimated value Li and the motor loss estimated value Lm. The first drive noise estimated value Ls1 indicates the drive noise estimated value Ls when the torque Trq is the first torque Trq1. As shown in the region DZ2 of FIG. 7D, the characteristic value (characteristic Def) increases as the torque Trq increases. This is because all of the power converter loss estimated value Li, the motor loss estimated value Lm, and the drive noise estimated value Ls increase with respect to the increase in torque Trq.

FIG. 8A shows an example of the relationship between the motor loss estimated value Lm and the rotation speed Nm. The vertical axis of the figure shows the motor loss estimated value Lm, and the horizontal axis shows the rotation speed Nm. The curve L41 shows an example of fluctuation of the motor loss estimated value Lm calculated by the response surface methodology. FIG. 8B shows an example of the relationship between the power converter loss estimate Li and the rotation speed Nm. The vertical axis of the figure shows the power converter loss estimated value Li, and the horizontal axis shows the rotation speed Nm. The curve L42 shows an example of fluctuation of the power converter loss estimated value Li calculated by the response surface methodology. FIG. 8C shows an example of the relationship between the drive noise estimated value Ls and the rotation speed Nm. The vertical axis of the figure shows the drive noise estimated value Ls, and the horizontal axis shows the rotation speed Nm. The curve L43 shows an example of fluctuation of the driving noise estimated value Ls calculated by the response surface methodology. FIG. 8D shows an example of the relationship between the characteristic Def and the rotation speed Nm. The vertical axis of the figure shows the characteristic Def, and the horizontal axis shows the rotation speed Nm. Region DZ3 shows an example of the characteristic Def calculated by the response surface methodology.

As shown in the curve L41 of FIG. 8A, the fluctuation range of the motor loss estimated value Lm with respect to the increase / decrease in the rotation speed Nm is smaller than the fluctuation range of the driving noise estimated value Ls. The first motor loss estimated value Lm1 indicates the motor loss estimated value Lm when the rotation speed Nm is the first rotation speed Nm1. As shown in the curve L42 of FIG. 8B, the fluctuation range of the power converter loss estimated value Li with respect to the increase / decrease in the rotation speed Nm is smaller than the fluctuation range of the driving noise estimated value Ls. The first power converter loss estimated value Li1 indicates the power converter loss estimated value Li when the rotation speed Nm is the first rotation speed Nm1. As shown in the curve L43 of FIG. 8C, the drive noise estimated value Ls decreases as the rotation speed Nm increases. The first drive noise estimated value Ls1 indicates the drive noise estimated value Ls when the rotation speed Nm is the first rotation speed Nm1. As shown in the region DZ3 of FIG. 8D, the characteristic value (characteristic Def) is substantially constant even if the rotation speed Nm increases or decreases. When the rotation speed Nm becomes extremely large, the characteristic value (characteristic Def) decreases.

FIG. 9A shows an example of the relationship between the motor loss estimated value Lm and the DC voltage Vdc. The vertical axis of the figure shows the motor loss estimated value Lm, and the horizontal axis shows the DC voltage Vdc. The curve L51 shows an example of fluctuation of the motor loss estimated value Lm calculated by the response surface methodology. FIG. 9B shows an example of the relationship between the power converter loss estimate Li and the DC voltage Vdc. The vertical axis of the figure shows the power converter loss estimated value Li, and the horizontal axis shows the DC voltage Vdc. The curve L52 shows an example of fluctuation of the power converter loss estimated value Li calculated by the response surface methodology. FIG. 9C shows an example of the relationship between the drive noise estimated value Ls and the DC voltage Vdc. The vertical axis of the figure shows the drive noise estimated value Ls, and the horizontal axis shows the DC voltage Vdc. The curve L53 shows an example of fluctuation of the driving noise estimated value Ls calculated by the response surface methodology. FIG. 9D shows an example of the relationship between the characteristic Def and the DC voltage Vdc. The vertical axis of the figure shows the characteristic Def, and the horizontal axis shows the DC voltage Vdc. Region DZ4 shows an example of the characteristic Def calculated by the response surface methodology.

As shown in the curve L51 of FIG. 9A, the fluctuation range of the motor loss estimated value Lm with respect to the increase / decrease of the DC voltage Vdc is smaller than the fluctuation range of the driving noise estimated value Ls. The first motor loss estimated value Lm1 indicates the motor loss estimated value Lm when the DC voltage Vdc is the first DC voltage Vdc1. As shown in the curve L52 of FIG. 9B, the power converter loss estimated value Li increases as the DC voltage Vdc increases. The first power converter loss estimated value Li1 indicates the power converter loss estimated value Li when the DC voltage Vdc is the first DC voltage Vdc1. As shown in the curve L53 of FIG. 9C, the drive noise estimated value Ls increases as the DC voltage Vdc increases. The first drive noise estimated value Ls1 indicates the drive noise estimated value Ls when the DC voltage Vdc is the first DC voltage Vdc1. As shown in the region DZ4 of FIG. 9D, the fluctuation range of the characteristic value (characteristic Def) is small and substantially constant until the DC voltage Vdc is up to the first DC voltage Vdc1. When the DC voltage Vdc increases from the first DC voltage Vdc1, the characteristic value (characteristic Def) gradually increases.

The first torque Trq1 corresponds to the torque command value Trq_ref, the first rotation speed Nm1 corresponds to the rotation speed calculation value Nm_fb, and the first DC voltage Vdc1 corresponds to the DC voltage detection value Vdc_fb. Further, in FIGS. 6A, 7A, 8A and 9A, the vertical axis is shown in the same range. This also applies to FIGS. 6B, 7B, 8B and 9B, the same applies to FIGS. 6C, 7C, 8C and 9C, and for FIGS. 6D, 7D, 8D and 9D. Is the same. Further, the characteristic (characteristic Def) can be illustrated in three dimensions.

FIG. 10A shows an example of the relationship between the carrier frequency Fc, the torque Trq, and the characteristic Def with respect to the motor loss estimated value Lm. The first axis of the figure shows the carrier frequency Fc, the second axis shows the torque Trq, and the third axis shows the characteristic Def. The curved surface Cs1 shows the interaction between the carrier frequency Fc and the torque Trq at the motor loss estimated value Lm. FIG. 10B shows an example of the relationship between the carrier frequency Fc, the torque Trq, and the characteristic Def with respect to the power converter loss estimate Li. The first axis of the figure shows the carrier frequency Fc, the second axis shows the torque Trq, and the third axis shows the characteristic Def. The curved surface Cs2 shows the interaction between the carrier frequency Fc and the torque Trq in the power converter loss estimate Li. FIG. 10C shows an example of the relationship between the carrier frequency Fc, the torque Trq, and the characteristic Def with respect to the drive noise estimated value Ls. The first axis of the figure shows the carrier frequency Fc, the second axis shows the torque Trq, and the third axis shows the characteristic Def. The curved surface Cs3 shows the interaction between the carrier frequency Fc and the torque Trq at the drive noise estimated value Ls.

In this way, the loss noise estimation unit 76 can acquire the characteristic Def related to the interaction between the carrier frequency Fc and the torque Trq (corresponding to the torque command value Trq_ref). The loss noise estimation unit 76 also acquires the characteristic Def related to the interaction between the carrier frequency Fc and the rotation speed Nm (corresponding to the rotation speed calculated value Nm_fb) of the other interaction (for example, the characteristic Def related to the interaction). be able to. Further, the loss noise estimation unit 76 calculates secondary estimated values (power converter loss estimated value Li, motor loss estimated value Lm, and drive noise estimated value Ls) for each of the above-mentioned factors, and determines the characteristic Def. You can also get it.

Further, the loss noise estimation unit 76 calculates the power converter loss estimate Li, the electric motor loss estimate Lm, and the drive noise estimate Ls, respectively, using a polynomial composed of a stationary term, a linear term, and an interaction term. It is preferable that each coefficient of the polynomial is set by using the partial regression coefficient calculated by the response surface methodology. The polynomial showing the power converter loss estimated value Li can be represented by the following equation 5 using the partial regression coefficients Ki1 to Ki10. Similarly, the polynomial showing the motor loss estimated value Lm can be represented by the following equation 6 using the partial regression coefficients Km1 to Km10. Further, the polynomial showing the drive noise estimated value Ls can be represented by the following equation 7 using the partial regression coefficients Ks1 to Ks10.

In the equations 5 to 7, the carrier frequency Fc, the torque Trq (corresponding to the torque command value Trq_ref), the rotation speed Nm (corresponding to the rotation speed calculation value Nm_fb), and the DC voltage Vdc (corresponding to the DC voltage detection value Vdc_fb) are It is a variable. Further, the steady-state term in the power converter loss estimated value Li is represented by the power converter loss steady-state term Ci, the steady-state term in the motor loss estimated value Lm is represented by the motor loss steady-state term Cm, and the steady-state term in the drive noise estimated value Ls is driven. It is represented by the noise steady-state term Cs. Further, the partial regression coefficients Ki1 to Ki4 are coefficients of the linear term of the polynomial, and the partial regression coefficients Ki5 to Ki10 are coefficients of the interaction term of the polynomial. Similarly, the partial regression coefficients Km1 to Km4 are the coefficients of the first-order terms of the polynomial, and the partial regression coefficients Km5 to Km10 are the coefficients of the interaction terms of the polynomial. The partial regression coefficients Ks1 to Ks4 are the coefficients of the first-order terms of the polynomial, and the partial regression coefficients Ks5 to Ks10 are the coefficients of the interaction term of the polynomial.

Further, the loss noise estimation unit 76 uses a polynomial composed of a stationary term, a primary term, an interaction term, and a quadratic term to obtain the power converter loss estimation value Li, the motor loss estimation value Lm, and the drive noise estimation value Ls, respectively. It can also be calculated. In this case as well, it is preferable that each coefficient of the polynomial is set using the partial regression coefficient calculated by the response surface methodology. In this case, the polynomial showing the power converter loss estimated value Li can be represented by the following equation 8 using the partial regression coefficients Ki1 to Ki14. Similarly, the polynomial showing the motor loss estimated value Lm can be represented by the following equation 9 using the partial regression coefficients Km1 to Km14. Further, the polynomial showing the drive noise estimated value Ls can be represented by the following equation 10 using the partial regression coefficients Ks1 to Ks14.

The equation 8 is different from the equation 5 in that it includes a quadratic term of the polynomial. The partial regression coefficients Ki11 to Ki14 are coefficients of the quadratic term of the polynomial. Similarly, Equation 9 differs from Equation 6 in that it includes a quadratic term of the polynomial. The partial regression coefficients Km11 to Km14 are coefficients of the quadratic term of the polynomial. Equation 10 differs from Equation 7 in that it includes a quadratic term of the polynomial. The partial regression coefficients Ks11 to Ks14 are coefficients of the quadratic term of the polynomial.

(Carrier frequency control unit 77)
The carrier frequency control unit 77 determines an appropriate value of the carrier frequency Fc based on the power converter loss estimate Li, the motor loss estimate Lm, and the drive noise estimate Ls calculated by the loss noise estimation unit 76. J is calculated, and the carrier frequency Fc when the determination value J becomes the minimum is output. Further, the carrier frequency control unit 77 normalizes the power converter loss estimated value Li, the motor loss estimated value Lm, and the drive noise estimated value Ls, respectively, and adds each of the normalized estimated values to calculate the determination value J. Then it is suitable.

In the examples shown in FIGS. 6A to 6C, for example, the assumed maximum value of the power converter loss estimated value Li is set to 1. At this time, the carrier frequency control unit 77 converts (normalizes) the motor loss estimated value Lm so that the assumed maximum value of the motor loss estimated value Lm becomes 1. Further, the carrier frequency control unit 77 converts (normalizes) the drive noise estimated value Ls so that the assumed maximum value of the drive noise estimated value Ls becomes 1. The first motor loss estimated value Lm1 shown in FIG. 6A, the first power converter loss estimated value Li1 shown in FIG. 6B, and the first drive noise estimated value Ls1 shown in FIG. 6C are all normalized estimated values. The carrier frequency control unit 77 adds the first electric motor loss estimated value Lm1, the first power converter loss estimated value Li1, and the first drive noise estimated value Ls1 to calculate the determination value J.

Further, the carrier frequency control unit 77 weights at least one estimated value of the power converter loss estimated value Li, the motor loss estimated value Lm, and the drive noise estimated value Ls, and each estimation includes the weighted estimated value. The determination value J can also be calculated by adding the values. For example, it is assumed that the power conversion system 1 of the present embodiment is used in a power conversion system including a driving electric motor for a vehicle such as a hybrid vehicle. In this case, the carrier frequency control unit 77 can perform weighting according to the surrounding environment of the vehicle, the usage state (for example, the traveling state), and the like.

For example, at night, noise is often more important than during the day. In this case, the carrier frequency control unit 77 may weight the estimated drive noise value Ls. As a result, the carrier frequency control unit 77 can output an optimum carrier frequency Fc that emphasizes noise associated with driving the power converter 40 and the electric motor 50. Further, for example, when the outside air temperature is high and the traveling state of the vehicle is low speed, it is assumed that the loss of the power converter 40 is emphasized. In this case, the carrier frequency control unit 77 may weight the estimated power converter loss value Li. As a result, the carrier frequency control unit 77 can output the optimum carrier frequency Fc with an emphasis on the loss of the power converter 40. Similarly, weighting can be performed with an emphasis on the loss of the electric motor 50.

FIG. 11 is a flowchart showing an example of the control procedure in the loss noise estimation unit 76 and the carrier frequency control unit 77. First, the loss noise estimation unit 76 acquires the torque command value Trq_ref, the rotation speed calculation value Nm_fb, and the DC voltage detection value Vdc_fb among the factors (step S11). Next, the loss noise estimation unit 76 acquires the carrier frequency Fc (i) among the factors (step S12). The carrier frequency Fc (i) is a carrier frequency Fc represented by using a counter i which is a variable of 0 or more and n (n is an integer) or less. As described above, the carrier frequency Fc is stored in the frequency table in ascending order at a predetermined pitch (for example, a pitch of several kilohertz). The loss noise estimation unit 76 selects one carrier frequency Fc indicated by the counter i from a plurality of carrier frequency Fcs stored in the frequency table. The initial value of the counter i is set to 0 (zero), and the loss noise estimation unit 76 first acquires the carrier frequency Fc (0).

Next, the loss noise estimation unit 76 estimates the characteristics (loss and noise) (step S13). Specifically, the loss noise estimation unit 76 calculates the power converter loss estimation value Li, the motor loss estimation value Lm, and the drive noise estimation value Ls by the response surface methodology including the above-mentioned factors. Then, the carrier frequency control unit 77 calculates the determination value J based on the power converter loss estimation value Li, the motor loss estimation value Lm, and the drive noise estimation value Ls calculated by the loss noise estimation unit 76. The calculated determination value J is set as the determination value J1 this time (step S14).

Next, the carrier frequency control unit 77 determines whether or not the determination value J1 is smaller than the minimum determination value J0 this time (step S15). The minimum determination value J0 is a variable (buffer) that stores the determination value J calculated by the carrier frequency control unit 77. When the determination value J1 this time is smaller than the minimum determination value J0 (in the case of Yes), the carrier frequency control unit 77 substitutes the carrier frequency Fc (i) of this time into the output carrier frequency Fc_out which is the carrier frequency Fc to be output ( Step S16). Further, the carrier frequency control unit 77 substitutes the determination value J1 this time for the minimum determination value J0 (step S17). Then, the control shifts to step S18. When the determination value J1 is larger than the minimum determination value J0 (No) this time, the control shifts to step S18 without performing the processes shown in steps S16 and S17.

Next, the carrier frequency control unit 77 determines whether or not the counter i has the counter maximum value IMAX (step S18). The counter maximum value IMAX is a constant set according to the number of carrier frequency Fcs stored in the frequency table. When the counter i matches the counter maximum value IMAX (in the case of Yes), the control is temporarily terminated. When the counter i does not match the counter maximum value IMAX (No), the carrier frequency control unit 77 counts up the counter i (step S19). Specifically, the carrier frequency control unit 77 increases the counter i by one. Then, the control returns to step S12, and the above-described processing and determination are repeated until the counter i matches the counter maximum value IMAX. In this way, the carrier frequency control unit 77 can output the carrier frequency Fc when the determination value J becomes the minimum.

(Pulse width modulation signal generation unit 78)
The pulse width modulation signal generation unit 78 uses the carrier frequency Fc output from the carrier frequency control unit 77 to form a plurality of switching elements of the power converter 40 (in this embodiment, three sets of a pair of switching elements 41). Generate an open / close signal. As described above, the carrier frequency Fc output from the carrier frequency control unit 77 is defined as the output carrier frequency Fc_out. As shown in FIG. 4, the pulse width modulation signal generation unit 78 has three-phase voltage command values (U-phase voltage command value Vu_ref, V-phase voltage command value Vv_ref, and V-phase voltage command value Vv_ref) calculated by the two-phase / three-phase conversion unit 75. The W-phase voltage command value Vw_ref), the DC voltage detection value Vdc_fb detected by the DC voltage detector 61a, and the carrier frequency Fc (output carrier frequency Fc_out) output from the carrier frequency control unit 77 are input. The pulse width modulation signal generation unit 78 sets the frequency of the carrier wave (triangle wave) in the pulse width modulation control (PWM control) to the output carrier frequency Fc_out.

Further, the pulse width modulation signal generation unit 78 calculates the modulation factor by dividing the three-phase voltage command value by the DC voltage detection value Vdc_fb. The pulse width modulation signal generation unit 78 generates a pulse signal (open / close signal) by pulse width modulation control (PWM control) based on the calculated modulation rate and the carrier (triangular wave) set at the output carrier frequency Fc_out. To do. Specifically, when the voltage command value is larger than the carrier wave, the switching element is set to the closed state, and when the voltage command value is smaller than the carrier wave, the switching element is set to the open state. The generated pulse signal (open / close signal) is applied to the control terminal 4g of each switching element of the power converter 40 via the drive circuit 61b shown in FIG.

According to the electric motor control device 10 according to the aspect 1, the control device 60 includes a loss noise estimation unit 76, a carrier frequency control unit 77, and a pulse width modulation signal generation unit 78. The loss noise estimation unit 76 uses a response curved surface method in which the carrier frequency Fc, which is the frequency of the carrier wave in pulse width modulation control, is included in at least a factor. The motor loss estimated value Lm, which is the loss estimated value of the above, and the drive noise estimated value Ls, which is the noise estimated value associated with the driving of the power converter 40 and the motor 50, are calculated. The carrier frequency control unit 77 determines an appropriate value of the carrier frequency Fc based on the power converter loss estimate Li, the motor loss estimate Lm, and the drive noise estimate Ls calculated by the loss noise estimation unit 76. J is calculated, and the carrier frequency Fc when the determination value J becomes the minimum is output. The pulse width modulation signal generation unit 78 uses the carrier frequency Fc (output carrier frequency Fc_out) output from the carrier frequency control unit 77 to use a plurality of switching elements (three sets of a pair of switching elements 41) of the power converter 40. Generates an open / close signal. As a result, the electric motor control device 10 according to the aspect 1 sets an optimum carrier frequency Fc that reduces the loss of the power converter 40 and the loss of the electric motor 50, and the noise associated with the driving of the power converter 40 and the electric motor 50. Therefore, the power converter 40 can be controlled to open and close.

According to the electric motor control device 10 according to the aspect 2, in the electric motor control device 10 according to the aspect 1, the loss noise estimation unit 76 determines the torque command value Trq_ref of the electric motor 50, the rotation speed detection value or the rotation speed calculation value Nm_fb of the electric motor 50. , The DC voltage detection value Vdc_fb of the DC power input to the power converter 40, and the carrier frequency Fc are factors. Therefore, the loss noise estimation unit 76 considers the loss of the power converter 40 and the loss of the electric motor 50, and the main factors of the noise associated with the driving of the power converter 40 and the electric motor 50 as factors, and the power converter loss estimation value Li , The estimated motor loss Lm and the estimated drive noise Ls can be calculated.

When the approximate expression (response surface model) created by the response surface methodology is represented by a polynomial containing terms of degree 2 or higher, the calculation accuracy of the estimated value is improved under certain conditions, but other than the specific conditions. Under the conditions, on the contrary, the calculation accuracy of the estimated value may decrease, and the versatility may decrease. Specifically, when the approximate expression is expressed by a polynomial containing terms of degree of degree 2 or higher, constants related to the motor 50 (for example, winding resistance R, d-axis inductance Ld, q-axis inductance Lq, induced voltage constant Φ, etc.) It becomes easy to be affected by the variation of.

According to the electric motor control device 10 according to the aspect 3, in the electric motor control device 10 according to the aspect 1 or the aspect 2, the loss noise estimation unit 76 uses a polynomial composed of a stationary term, a primary term, and an interaction term to generate electric power. The converter loss estimated value Li, the motor loss estimated value Lm, and the drive noise estimated value Ls are calculated, and each coefficient of the polynomial is set using the partial regression coefficient calculated by the response surface methodology. Therefore, the loss noise estimation unit 76 is not easily affected by variations in constants related to the motor 50 (for example, winding resistance R, d-axis inductance Ld, q-axis inductance Lq, induced voltage constant Φ, etc.), and is not easily affected by variations of the motor 50. It is possible to reduce the deterioration of the estimation accuracy due to individual differences.

According to the motor control device 10 according to the aspect 4, in the motor control device 10 related to any one of the aspects 1 to 3, the carrier frequency control unit 77 has a power converter loss estimated value Li and an electric motor loss estimated value. Lm and the drive noise estimated value Ls are normalized, and the normalized estimated values are added to calculate the determination value J. Therefore, the carrier frequency control unit 77 can calculate the determination value J from a plurality of estimated values having different unit systems, and can easily set the optimum carrier frequency Fc.

According to the motor control device 10 according to the aspect 5, in the motor control device 10 related to any one of the aspects 1 to 3, the carrier frequency control unit 77 has a power converter loss estimated value Li and an electric motor loss estimated value. At least one estimated value of Lm and the driving noise estimated value Ls is weighted, and each estimated value including the weighted estimated value is added to calculate the determination value J. Therefore, the carrier frequency control unit 77 can easily set the optimum carrier frequency Fc in consideration of, for example, the ambient environment and the usage state.

<Others>
The present invention is not limited to the embodiment described above and shown in the drawings, and can be appropriately modified and implemented without departing from the gist. For example, the control method of the electric motor 50 is not limited to vector control. The control device 60 can perform known drive control (for example, rectangular wave drive). Further, the electric motor control device 10 is suitable for use in, for example, a power conversion system 1 including an electric motor for driving a vehicle such as a hybrid vehicle.

<Additional notes>
The present invention can also be regarded as a control method for the electric motor control device 10. The loss noise estimation step corresponds to the control performed by the loss noise estimation unit 76, the carrier frequency control step corresponds to the control performed by the carrier frequency control unit 77, and the pulse width modulation signal generation step corresponds to the pulse width modulation signal generation unit. Corresponds to the control performed by 78. Also in the control method of the electric motor control device 10, it is possible to obtain the same action and effect as those described above for the electric motor control device 10.

(Appendix 1)
A power converter that converts DC power into AC power by controlling the opening and closing of a plurality of switching elements and supplies the converted AC power to the electric motor.
A control device that controls opening and closing of the plurality of switching elements of the power converter by pulse width modulation control.
It is a control method of an electric motor control device provided with
By the response curved surface method in which the carrier frequency, which is the frequency of the carrier in the pulse width modulation control, is included in at least a factor, the power converter loss estimated value, which is the loss estimated value of the power converter, and the motor loss, which is the loss estimated value of the motor, are used. The loss noise estimation process for calculating the estimated value and the driving noise estimated value which is the noise estimated value associated with the driving of the power converter and the electric motor, and
Based on the power converter loss estimated value, the motor loss estimated value, and the drive noise estimated value calculated by the loss noise estimation step, a determination value for determining an appropriate value of the carrier frequency is calculated, and the determination is made. A carrier frequency control step that outputs the carrier frequency when the value becomes the minimum, and
A pulse width modulation signal generation step of generating an open / close signal of the plurality of switching elements of the power converter using the carrier frequency obtained by the carrier frequency control step.
A control method for an electric motor control device.

(Appendix 2)
In the loss noise estimation step, the torque command value of the electric motor, the rotation speed detection value or the rotation speed calculation value of the electric motor, the DC voltage detection value of the DC power input to the power converter, and the carrier frequency are used. The method for controlling an electric motor control device according to Appendix 1, which is a factor.

(Appendix 3)
In the loss noise estimation step, the power converter loss estimation value, the motor loss estimation value, and the drive noise estimation value are calculated by using a polynomial composed of a stationary term, a primary term, and an interaction term, respectively.
The control method for an electric motor control device according to Appendix 1 or Appendix 2, wherein each coefficient of the polynomial is set using the partial regression coefficient calculated by the response surface methodology.

(Appendix 4)
In the carrier frequency control step, the power converter loss estimate, the motor loss estimate, and the drive noise estimate are normalized, and the normalized estimates are added to calculate the determination value. Item 3. The method for controlling an electric motor control device according to any one of Items 1 to 3.

(Appendix 5)
The carrier frequency control step weights at least one of the power converter loss estimate, the motor loss estimate, and the drive noise estimate, and each estimate includes the weighted estimate. The control method of the electric motor control device according to any one of Supplementary Items 1 to 3, wherein the determination value is calculated by adding.

10: Electric motor control device,
40: Power converter,
41: Multiple switching elements (three sets of pair of switching elements),
50: Electric motor,
60: Control device,
76: Loss noise estimation unit, 77: Carrier frequency control unit,
78: Pulse width modulation signal generator,
Li: Power converter loss estimate, Lm: Motor loss estimate, Ls: Drive noise estimate,
Fc: carrier frequency, J: judgment value,
Trq_ref: torque command value, Nm_fb: rotation speed calculation value,
Vdc_fb: DC voltage detection value.

Claims (5)

A power converter that converts DC power into AC power by controlling the opening and closing of a plurality of switching elements and supplies the converted AC power to the electric motor.
A control device that controls opening and closing of the plurality of switching elements of the power converter by pulse width modulation control.
It is an electric motor control device equipped with
The control device is
By the response curved surface method in which the carrier frequency, which is the frequency of the carrier in the pulse width modulation control, is included in at least a factor, the power converter loss estimated value, which is the loss estimated value of the power converter, and the motor loss, which is the loss estimated value of the motor, are used. The loss noise estimation unit that calculates the estimated value and the driving noise estimated value that is the noise estimated value associated with the driving of the power converter and the electric motor, and
Based on the power converter loss estimate, the motor loss estimate, and the drive noise estimate calculated by the loss noise estimation unit, a determination value for determining an appropriate value of the carrier frequency is calculated, and the determination is made. A carrier frequency control unit that outputs the carrier frequency when the value becomes the minimum, and
A pulse width modulation signal generation unit that uses the carrier frequency output from the carrier frequency control unit to generate an open / close signal for the plurality of switching elements of the power converter.
An electric motor control device equipped with. The loss noise estimation unit determines the torque command value of the electric motor, the rotation speed detection value or the rotation speed calculation value of the electric motor, the DC voltage detection value of the DC power input to the power converter, and the carrier frequency. The electric motor control device according to claim 1, which is a factor. The loss noise estimation unit calculates the power converter loss estimation value, the motor loss estimation value, and the drive noise estimation value, respectively, using a polynomial composed of a stationary term, a primary term, and an interaction term.
The electric motor control device according to claim 1 or 2, wherein each coefficient of the polynomial is set by using the partial regression coefficient calculated by the response surface methodology. The carrier frequency control unit normalizes the power converter loss estimated value, the motor loss estimated value, and the drive noise estimated value, and adds the normalized estimated values to calculate the determined value. The electric motor control device according to any one of items 1 to 3. The carrier frequency control unit weights at least one estimated value of the power converter loss estimated value, the motor loss estimated value, and the driving noise estimated value, and each estimated value including the weighted estimated value. The electric motor control device according to any one of claims 1 to 3, wherein the determination value is calculated by adding the above.

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