JP6787042B2 - Feedback amplifier and transmitter using it - Google Patents

Description

The present invention relates to a Cartesian loop type feedback amplification device and a transmitter using the same, and more particularly to the phase difference adjustment between an input signal and a feedback signal.

When a power amplifier used in a transmitter or the like is operated in a non-linear region in order to improve power efficiency, non-linear distortion occurs, and it is necessary to compensate the power amplifier for distortion in order to compensate for this distortion.
As one of the distortion compensation, there is a medical record cyan loop distortion compensation. This Cartesian loop distortion compensation causes a feedback baseband signal (I', Q'), which is a part of the output signal of the power amplifier orthogonally demodulated to the input baseband signal (I, Q), to be negatively fed back for each Cartesian coordinate component. .. Specifically, power is obtained by subtracting the feedback baseband signal (I', Q') from the input baseband signal (I, Q) and adding the difference value to the input baseband signal (I, Q). The distortion component generated by the amplifier is canceled and the distortion is compensated. When subtracting the feedback baseband signal (I', Q') from the input baseband signal (I, Q), the phase of the input baseband signal (I, Q) and the feedback baseband signal (I', Q') If there is a phase difference in phase, the operation of the Cartesian loop type feedback amplifier becomes unstable, and in the worst case, the power amplifier may oscillate. Causes of this phase difference include the loop length of the feedback path, the phase characteristics of the power amplifier, and the temperature characteristics. Therefore, it is necessary to adjust the phase of the input baseband signal (I, Q) and the phase of the feedback baseband signal (I', Q') to eliminate the phase difference. In the conventional phase adjustment, the phase of the input baseband signal and the phase of the feedback baseband signal are detected and adjusted so that the phase difference is made zero. However, the phase detection accuracy is poor and the phase difference is made zero. There was a problem that it was difficult to do. Therefore, a method of adjusting the phase difference without detecting the phase of the input baseband signal and the phase of the feedback baseband signal has been proposed (see, for example, Patent Document 1).

In this prior art, instead of detecting the phase of the input baseband signal and the phase of the feedback baseband signal, the spurious power generated by the phase difference between the phase of the input baseband signal and the phase of the feedback baseband signal is detected. By adjusting the phase of the local signal so that the detected spurious power is minimized, this phase difference is made almost zero.

Specifically, a part of the output signal of the power amplifier is used as a feedback signal, and this feedback signal is down-converted to a predetermined frequency and passes through a frequency band above and below a predetermined frequency from the down-converted frequency. The high-pass filter and the low-pass filter are used to extract the power of the lower spurious and the power of the upper spurious generated by the power amplifier, and the power of the extracted spurious is detected by the power detector.
Then, the phase of the local signal is adjusted by the phase device so that the power of each detected spurious becomes small.

JP-A-2002-344555

However, in the above-mentioned prior art, in order to detect the spurious power contained in the feedback signal, a low-pass filter and a high-pass filter and two power detectors for detecting the spurious power from the filter outputs of the low-pass filter and the high-pass filter are used. The problem is that it is necessary and the circuit configuration becomes complicated.
Therefore, the present invention has been made by paying attention to the above-mentioned problems of the prior art, and is a Cartesian loop type feedback amplification that can simplify the circuit for adjusting the phase difference between the input baseband signal and the feedback baseband signal. It is an object of the present invention to provide a device and a transmitter using the device.

The feedback amplification device according to one aspect of the present invention amplifies an orthogonal modulation unit that generates an orthogonal modulation signal from a baseband signal and a local signal, and an orthogonal modulation signal output from the orthogonal modulation unit, and outputs the signal as a transmission signal. A power amplifier is provided, a part of the transmission signal is used as a feedback signal, an orthogonal demodulator that generates an orthogonal demographic signal from the feedback signal and a local signal, and the orthogonal demographic signal are negatively fed back to the baseband signal. A feedback amplification device that compensates for non-linear distortion includes a current detection unit that detects the current consumption of the power amplifier, and a phase adjustment unit that adjusts the phase of a local signal input to the orthogonal modulation unit or the orthogonal demodulation unit.

Further, one aspect of the transmitter according to the present invention is a baseband signal generation unit that generates a baseband signal from transmission data, and the feedback amplification described above in which the baseband signal generated by the baseband signal generation unit is input. It is equipped with a transmission antenna that transmits a transmission signal output from the device.

According to one aspect of the present invention, the current consumption of the power amplifier is detected, and the phase of the local signal input to the quadrature modulation unit or the quadrature demodulation unit is adjusted based on the detected current consumption. Can provide a simplified Cartesian loop feedback amplifier and transmitter.

It is a block diagram which shows the structural example of the transmitter including the feedback amplification apparatus which concerns on 1st Embodiment of this invention. It is a flowchart which shows an example of a phase control procedure. It is a characteristic diagram which shows the measured value of a phase difference, a current consumption and a spurious. It is a characteristic diagram which shows the relationship between a phase difference and a current consumption.

Next, an embodiment of the present invention will be described with reference to the drawings. In the description of the drawings below, the same or similar parts are designated by the same or similar reference numerals. However, it should be noted that the drawings are schematic and differ from the actual ones. Therefore, the specific components should be judged in consideration of the following explanation.
Further, the embodiments shown below exemplify devices and methods for embodying the technical idea of the present invention, and the technical idea of the present invention describes the shape, structure, arrangement, etc. of components. It is not specific to the following. The technical idea of the present invention may be modified in various ways within the technical scope specified by the claims stated in the claims.

Hereinafter, a transmitter using the feedback amplification device according to the embodiment of the present invention will be described.
As shown in FIG. 1, the transmitter 10 is configured as a transmitter having a Cartesian loop type feedback amplification device. The transmitter 10 is a Cartesian loop system that amplifies a modulated signal by orthogonally modulating a baseband signal generation unit 21 that generates a baseband signal from transmission data and a baseband signal output from the baseband signal generation unit 21. It includes a feedback amplification device 30 and a transmission antenna 50 that transmits a transmission signal output from the feedback amplification device 30.

The baseband signal generation unit 21 generates an in-phase component signal I and an orthogonal component signal Q of the baseband signal from the input transmission data, and feedback-amplifies the generated in-phase component signal I and the orthogonal component signal Q of the baseband signal. Output to device 30.
The feedback amplification device 30 includes a phase adjustment unit 31, a distortion compensation unit 32, a low-pass filter 33a and 33b, a quadrature modulation unit 34, a local oscillator 35, an upconverter 36, a power amplifier 37, a directional coupler 38, an attenuator 39, and a downconverter. 40, a quadrature demodulation unit 41 and a current detection unit 43 are provided.

The phase adjusting unit 31 includes a phase shifting unit 31a and a phase control unit 31b. The phase control unit 31b outputs a phase command value φt to the phase shift unit 31a based on the current consumption value of the power amplifier 37 detected by the current detection unit 43, which will be described later, and receives the phase command value φt. 31a adjusts the phase of the local signal input to the orthogonal demodulation unit 41 based on the phase command value φt.

The phase shift unit 31a is arranged on the quadrature modulation unit 34 side instead of being arranged on the quadrature demodulation unit 41 side in FIG. 1 so as to adjust the phase of the local signal input to the quadrature modulation unit 34. May be good.
The distortion compensation unit 32 includes error amplifiers 32a and 32b. In the error amplifier 32a, the in-phase component signal I of the baseband signal output from the baseband signal generation unit 21 and the feedback in-phase component signal I ′ output from the orthogonal demodulation unit 41 are input, and the in-phase component signal I and feedback are input. The in-phase component difference value ΔI (= I-I ′) with the in-phase component signal I ′ is added to the in-phase component signal I, the distortion component of the in-phase component signal I generated by the power amplifier 37 is canceled, and the distortion component is output to the low-pass filter 33a. ..

In the error amplifier 32b, the orthogonal component signal Q of the baseband signal output from the baseband signal generation unit 21 and the feedback orthogonal component signal Q'output from the orthogonal demodulator 41 are input, and the orthogonal component signal Q and feedback are input. The orthogonal component difference value ΔQ (= Q−Q ′) with the orthogonal component signal Q ′ is added to the orthogonal component signal Q, the distortion component of the orthogonal component signal Q generated by the power amplifier 37 is canceled, and the distortion component is output to the low pass filter 33b. ..

The low-pass filter 33a removes the noise component in the high frequency band of the in-phase component signal I output from the error amplifier 32a and outputs it to the quadrature modulation unit 34.
The low-pass filter 33b removes the noise component in the high frequency band of the quadrature component signal Q output from the error amplifier 32b and outputs it to the quadrature modulation unit 34.
The quadrature modulator 34 includes an in-phase component signal modulator 34a, an quadrature component signal modulator 34b, a 90 ° phase shifter 34c, and an adder 34d.

The in-phase component signal modulator 34a outputs an in-phase component modulated signal obtained by modulating a local signal output from the local oscillator 35 with the in-phase component signal I input from the low-pass filter 33a to the adder 34d.
The orthogonal component signal modulator 34b is an orthogonal component modulation in which a local signal is modulated by a 90 ° phase shifter 34c after being output from the local oscillator 35 by the orthogonal component signal Q input from the low-pass filter 33b. The signal is output to the adder 34d.

The adder 34d adds the in-phase component modulation signal output from the in-phase component signal modulator 34a and the orthogonal component modulation signal output from the orthogonal component signal modulator 34b, and outputs the modulation signal.
The upconverter 36 upconverts the modulated signal from the orthogonal modulation unit 34 to a predetermined frequency in the RF band (for example, 260 MHz to 270 MHz band), converts it into an RF signal SRF, and converts the RF signal SRF into a power amplifier 37 in the subsequent stage. Output.

The power amplifier 37 is configured to include, for example, an N-channel field effect transistor (FET), and a DC voltage is applied to the drain electrode from the power supply VDC. The power amplifier 37 amplifies the RF signal SRF output from the upconverter 36 to generate an output transmission signal SOT. Then, the power amplifier 37 outputs this output transmission signal SOT to the transmission antenna 50.

The directional coupler 38 takes out a part of the output transmission signal SOT output from the power amplifier 37 to the transmission antenna 50 as a feedback quadrature modulation signal and outputs it to the attenuator 39.
The attenuator 39 adjusts the level of the feedback quadrature modulation signal and outputs the feedback quadrature modulation signal after the level adjustment to the down converter 40.
The down converter 40 down-converts the feedback transmission signal in the RF band to a predetermined frequency and outputs it to the orthogonal demodulation unit 41.

The quadrature demodulation unit 41 demodulates the feedback quadrature modulation signal output from the down converter 40. The orthogonal demodulator 41 includes a demodulator for in-phase component signals 41a, a demodulator for orthogonal component signals 41b, and a 90 ° phase shifter 41c.
The demodulator 41a for the in-phase component signal demodulates the feedback orthogonal modulation signal output from the down converter 40 by the local signal whose phase is adjusted by the phase shift unit 31a, and the feedback in-phase component signal I ′ is obtained by the distortion compensation unit 32. Output to the error amplifier 32a.

The quadrature component signal demodulator 41b is phase-adjusted by the phase shift unit 31a, and the feedback obtained by demodulating the feedback quadrature modulation signal output from the down converter 40 by the local signal phase-shifted by 90 ° by the phase shifter 41c is obtained. The quadrature component signal Q'is output to the error amplifier 32b of the distortion compensation unit 32.
The feedback in-phase component signal I'and the feedback orthogonal component signal Q'are negatively fed back to the error amplifier 32a and the error amplifier 32b of the distortion compensation unit 32, respectively, to perform distortion compensation in the Cartesian loop system.

The current detection unit 43 is provided between the DC power supply terminal 37a of the power amplifier 37 and the power supply VDC, detects the current consumption Id of the power amplifier 37, and sets the detected current consumption detection value Id as the phase of the phase adjustment unit 31. Output to the control unit 31b.
The phase adjusting unit 31 adjusts the phase of the local signal input to the orthogonal demodulation unit 41 based on the current consumption detection value Id detected by the current detecting unit 43, and returns to the input baseband signal (I, Q). The phase difference from the baseband signal (I', Q') is set to be almost zero.

This phase adjustment will be described with reference to FIGS. 2 and 3.
When the relationship between the phase difference between the phase of the input baseband signal (I, Q) and the phase of the feedback baseband signal (I', Q') and the current consumption Id of the power amplifier 37 is actually measured, the solid line shown in FIG. As described above, it is represented by the inverted trapezoidal characteristic line L1. That is, when the phase of the feedback baseband signal (I', Q') is advanced from the phase difference of 0 ° and the phase difference is increased in the positive direction such as 3 °, 6 ° ..., The phase difference is 6 The current consumption Id almost maintains the minimum current up to °, but the current consumption Id increases slightly at a phase difference of 9 °, and then the current consumption increases sharply when the phase difference increases to 12 ° and 15 °.

Similarly, if the phase of the feedback baseband signal (I', Q') is delayed from the phase difference of 0 ° and the phase difference is increased in the negative direction such as -3 °, -6 °, etc., the phase difference is increased. Up to -9 °, the current consumption Id almost maintains the minimum current, but when the phase difference is increased to -12 ° and -15 ° in the negative direction, the current consumption increases sharply.
The range of the minimum current of the current consumption Id has a width with respect to the magnitude direction of the phase difference centering on zero phase difference. Therefore, when adjusting the phase difference based on the current consumption Id, the phase difference becomes almost zero by adjusting the phase so that the current consumption becomes the center value of the phase difference which is the minimum current.

The relationship between the phase difference (deg) and the spurious (dB) of the output transmission signal S0T due to the linear distortion of the power amplifier 37 is the consumption of the power amplifier 37 as shown by the characteristic line L2 shown by the dotted line in FIG. It correlates with the current Id. Here, assuming that the spurious standard is −60 dB or less with respect to the output transmission signal S0T, the standard can be satisfied if the phase difference is between −9 ° and 9 °.

Next, the phase control process of the phase control unit 31b of the phase adjustment unit 31 will be described with reference to the flowchart of FIG. The phase control unit 31b performs the phase control process of FIG. 2, calculates the phase command value φt, and outputs the phase command value φt to the phase shift unit 31a. The phase shift unit 31a adjusts the phase of the local signal based on the phase command value φt. As shown in FIG. 3, when the phase command value φt is increased or decreased by one step, the phase shift amount of the local signal is increased or decreased by 3 °. The amount of change in this phase shift can be changed as appropriate.
This phase control process is executed at the start of transmission, but may be repeatedly executed every time a predetermined time elapses after the start of transmission.

First, in step S1, when transmission is started, the current phase command value φt is set to the initial value φt (0), and then the process proceeds to step S2.
In step S2, the phase control unit 31b stores the current consumption detection value Id of the power amplifier 37 detected by the current detection unit 43 in a memory (not shown, hereinafter referred to as a memory) in the phase control unit 31b, and then in step S3. Transition.

In step S3, in order to advance the phase of the local signal by one step, a new phase command value φt (1) obtained by adding the phase command value φt to the current phase command value φt (0) and the phase shift amount Δφ (= 3 °). ), And then the process proceeds to step S4. Here, when the phase command values φt (0) and φt (1) are expressed as φt (n), n is an integer representing the number of steps of the command value, and the phase is used to advance the phase of the local signal step by step. It is incremented each time the shift amount Δφ (= 3 °) is added.

In this step S4, the phase command value φt set by the phase control unit 31b is output to the phase shift unit 31a, and the phase shift unit 31a adjusts the phase of the local signal based on the phase command value φt before proceeding to step S5. To do.
In step S5, the phase control unit 31b reads the current consumption detection value of the power amplifier 37 detected by the current detection unit 43 as Id, stores it in the memory, and then proceeds to step S6.

In this step S6, is the read current consumption detection value Id equal to or less than the set current Ids (for example, 4 [A] shown by the alternate long and short dash line in FIG. 3) in which the current value whose spurious exceeds the standard value is set as a threshold value in advance? It is determined whether or not, and when Id ≦ Ids, the process returns to step S3, and when Id> Ids, the process proceeds to step S7. In FIG. 3, since the current consumption detection value Id is equal to or less than the set current Ids when the phase command value φt is from φt (0) to φt (3), the process of returning from step S6 to step S3 is repeated, but the phase command value φt When becomes φt (4), the current consumption detection value Id exceeds the set current Ids.

In this step S7, the phase control unit sets the phase command value φt = φt (n-1) immediately before the current phase command value φt = φt (n) as the phase upper limit value φtmax (φt (3) in FIG. 3). After storing in the memory in 31b, the process proceeds to step S8.
In step S8, the phase command value φt is set to the initial value φt (0) again, and then the process proceeds to step S9 so that the phase of the local signal is delayed by one step from the current phase command value φt. , The phase command value φt is subtracted from the current phase command value φt (0) by the phase shift amount Δφ (= 3 °), and a new phase command value φt (-1) is set, and then the process proceeds to step S10. Note that n is decremented each time the phase shift amount Δφ (= 3 °) is subtracted in order to delay the phase of the local signal by one step.

In this step S10, the phase command value φt set by the phase control unit 31b is output to the phase shift unit 31a, and the phase shift unit 31a adjusts the phase of the local signal based on the phase command value φt before proceeding to step S11. To do.
In step S11, the phase control unit 31b reads the current consumption detection value Id of the power amplifier 37 detected by the current detection unit 43 and stores it in the memory, and then proceeds to step S12.

In this step S11, it is determined whether or not the read current consumption detection value Id is equal to or less than the preset current Ids, and when Id ≦ Ids, the process returns to step S8, and when Id> Ids, step S13. Move to. In FIG. 3, since the current consumption detection value Id is equal to or less than the set current Ids when the phase command value φt is from φt (0) to φt (-3), the process of returning from step S11 to step S8 is repeated, but the phase command value When φt becomes φt (-4), the current consumption detection value Id exceeds the set current Ids.
In this step S13, the phase control unit 31ab is set to the phase lower limit value φtmin (φt (-3) in FIG. 3) with the phase command value φt = φt (n + 1) immediately before the current phase command value φt = φt (n). After storing in the memory inside, the process proceeds to step S14.

In step S14, if all the current consumption detection values Id stored in the memory so far exceed the specified current Idr determined to be abnormal as the power amplifier if the spurious level by the power amplifier 37 is equal to or higher than the preset specified value. Judge whether or not. When the determination result is Id> Idr, the power amplifier 37 determines that it is abnormal and proceeds to step S15, and when Id ≦ Idr, the power amplifier 37 determines that it is normal and proceeds to step S16. The relationship between the specified current Idr and the set current Ids is such that the specified current Idr ≥ the set current Ids. When the specified current Idr = the set current Ids, the threshold value is a current value at which the set current Ids exceeds the spurious standard value. Further, when the specified current Idr> the set current Ids, the current value is set so that a margin can be taken so that the set current Ids does not exceed the spurious standard value.

Further, in step S15, since the power amplifier 37 is determined to be abnormal, the operation of the power amplifier is stopped to end the phase control process. In the case of a transmitter, an abnormality display indicating an abnormality may be displayed on a display unit (not shown).
In step S16, the upper limit value φtmax of the phase command value and the lower limit value φt (min) of the phase command value stored in the memory are read, and the upper limit value φt (max) of the phase command value and the lower limit value φtmin of the phase command value are set. After setting the average value as a new phase command value φt, the process proceeds to step S17.

In step S17, the set phase command value φt is output to the phase shift unit 31a, the phase shift unit 31a adjusts the phase of the local signal based on the phase command value φt, and then the phase control process ends.
Therefore, by adjusting the phase of the local signal with the phase shift unit 31a so that the current consumption of the power amplifier 37 is minimized, the phase of the input baseband signal (I, Q) and the feedback baseband signal (I', Q) are adjusted. The phase difference from the phase of ′) can be made almost zero.

In the present embodiment, the current consumption Id of the power amplifier 37 is set as a method of making the phase difference between the phase of the input baseband signal (I, Q) and the phase of the feedback baseband signal (I', Q') substantially zero. It is detected by the current detection unit 43, the phase control unit 31b sets the phase command value φt that minimizes the detected current consumption detection value Id, and the phase shift unit 31a sets the local signal based on the set phase command value φt. Adjust the phase. As a result, the phase difference between the phase of the input baseband signal (I, Q) and the phase of the feedback baseband signal (I', Q') in the Cartesian loop distortion compensation can be made almost zero, resulting in spurious emissions. Power can be suppressed.

Further, the upper limit value φtmax of the phase command value and the lower limit value φtmin of the phase command value are detected, and the average value of the upper limit value φtmax of the phase command value and the lower limit value φtmin of the phase command value is set as the phase command value φt. As shown in FIG. 3, it is possible to set the center value of the phase difference in which the current consumption detection value Id is the minimum current, that is, the phase command value φt in which the phase difference is almost zero.

In the above embodiment, the phase control when the minimum current of the current consumption detection value Id straddles a plurality of phase command values φt has been described, but the present invention is not limited to this, and is as shown in FIG. If the current consumption detection value Id is a characteristic that takes one minimum value, the phase command value φt is sequentially changed in the direction in which the current consumption detection value Id becomes smaller than the previous value when the phase is changed. When the detected current consumption value becomes larger than the previous value, the previous phase command value φt (n-1) may be set as the phase command value φt.

Although the present invention has been described above with reference to specific embodiments, it is not intended to limit the invention by these explanations. By reference to the description of the present invention, other embodiments of the present invention will be apparent to those skilled in the art as well as various variations of the disclosed embodiments. Therefore, it should be understood that the scope of claims also covers these modifications or embodiments included in the scope and gist of the present invention.

10 ... Transmitter, 30 ... Feedback Amplifier, 31 ... Phase Adjuster, 31a ... Phase Shift, 31b ... Phase Control, 32 ... Distortion Compensation, 32a, 32b ... Error Amplifier, 33a, 33b ... Low Pass Filter, 34 ... Orthogonal modulator, 34a ... In-phase component signal modulator, 34b ... Orthogonal component signal modulator, 34c ... 90 ° phase shifter, 34d ... Adder, 35 ... Local oscillator, 36 ... Upconverter, 37 ... Power amplifier , 37a DC power supply terminal, 38 ... Directional coupler, 39 ... Attenuator, 40 ... Down converter, 41 ... Orthogonal demodulator, 41a ... In-phase component signal demodulator, 41b ... Orthogonal component signal demodulator, 41c ... 90 ° Phase shifter, 43 ... Current detector

Claims (4)

A quadrature modulator that generates a quadrature modulation signal from a baseband signal and a local signal,
A power amplifier that amplifies the quadrature modulation signal output from the quadrature modulation unit and outputs it as a transmission signal.
A part of the transmission signal is used as a feedback signal, and the feedback signal and an orthogonal demodulation unit that generates an orthogonal demodulation signal from the local signal are provided.
In a feedback amplification device that compensates for non-linear distortion by negatively feeding the orthogonal demodulated signal to the baseband signal.
A current detector that detects the current consumption of the power amplifier,
A phase adjusting unit for adjusting the phase of the local signal input to the quadrature modulation unit or the quadrature demodulation unit in order to match the phase of the baseband signal and the quadrature demodulation signal is provided.
The phase adjusting unit is a feedback amplification device that adjusts the phase of the local signal so that the current consumption is minimized. The phase adjusting unit changes the phase of the local signal, detects an upper limit value of a phase and a lower limit value of the phase in which the current consumption becomes larger than a predetermined set current, and determines the phase of the local signal. The feedback amplification device according to claim 1, wherein the feedback amplification device is adjusted to an average value of an upper limit value and a lower limit value of. The phase adjusting unit determines whether or not the minimum value of the current consumption exceeds a predetermined specified current, and stops the operation of the power amplifier when the current consumption exceeds the specified current. The feedback amplification device according to claim 1 or 2, wherein the feedback amplification device is characterized. A baseband signal generator that generates a baseband signal,
The feedback amplification device according to any one of claims 1 to 3, wherein a baseband signal generated by the baseband signal generation unit is input.
A transmitter including a transmitting antenna for transmitting a transmission signal output from the feedback amplification device.

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