JP6812752B2 - Switching power supply - Google Patents

Description

The present invention relates to a switching power supply device capable of overcurrent protection.

In a switching power supply device, a plurality of converters may be connected in series in order to obtain a desired voltage. FIG. 9 shows a configuration example of a switching power supply device including a two-stage converter. The switching power supply device 101 includes a primary side circuit 25, a secondary side circuit 70, and a transformer 60, converts the power input from the input terminal IN, and outputs the power to the output terminal OUT. The primary circuit 25 is provided with a two-stage converter, that is, a front-stage first converter 10 and a rear-stage second converter 15. The second converter 15 is a so-called LLC series resonance converter, and has a switching circuit 30 and a series resonance circuit 50.

If a short circuit occurs in the secondary circuit 70 or the output terminal OUT side, the secondary circuit 70 may have a low resistance and a large current may flow. In that case, even in the primary circuit 25 coupled by the transformer 60, the electric charge accumulated in the smoothing capacitor 20 is released at once, and a large current flows through the switch elements Q1 to Q4 constituting the switching circuit 30, damaging the switch element. It is possible. In order to avoid such a failure due to overcurrent, the control unit 90 of the switching power supply device 101 provides a current detection circuit 45 in the second converter 15 in the primary side circuit 25, and performs a process as shown in FIG. I'm doing it.

FIG. 10 is a flowchart for overcurrent protection. The current detection circuit 45 including the current transformer 40 detects the current flowing through the second converter 15 (step T10) and detects the average current (step T20). After that, the process proceeds to step T30, and the control unit 90 determines whether or not the average current is equal to or greater than a predetermined threshold value. If the average current is less than a predetermined threshold (NO), the process returns to the beginning (step T10). On the other hand, when the average current is equal to or higher than a predetermined threshold value (YES), the control unit 90 controls the switch element of the first converter 10 to lower the output voltage of the first converter 10. The value of the current flowing through the converter 15 is reduced (step T40). Further, the control unit 90 controls the switch element of the second converter 15 by PFM (Pulse Frequency Modulation) to raise the switching frequency, reduce the transmission efficiency, and suppress the current (step T50).

Japanese Unexamined Patent Publication No. 2013-240168 International Publication No. 2013/114758

However, the responsiveness of the switching control of the switch element in the first converter in the previous stage is poor, and it may take time to reduce the output voltage in the previous stage. In such a case, the second converter in the second stage is overcurrent. There was a problem that the flow was transient. Further, when the second converter is an LLC series resonance converter as shown in FIG. 9, there is a drawback that the control circuit for controlling the switching frequency by PFM becomes complicated.

In view of such circumstances, the present invention intends to provide a switching power supply device capable of overcurrent protection by PWM (Pulse Width Modulation) control having a simple configuration.

(1) In the present invention, a primary side circuit having a converter including at least two switch elements, a secondary side circuit having a rectifying smoothing circuit, a transformer connecting the primary side circuit and the secondary side circuit, and the above. The primary side circuit or the secondary side circuit has a current detection circuit that detects the current value of the current flowing through the circuit, and the drive pulse of the switch element is based on the current value detected by the current detection circuit. Provided is a switching power supply device including a control unit for controlling the conduction width of the above.

According to the invention described in (1) above, since the primary side circuit or the secondary side circuit has a current detection circuit for detecting the current value of the current flowing through the circuit, it may cause a short circuit accident or other causes. By controlling the pulse width of the drive pulse that drives the switch element of the converter when it is detected that an inappropriate current may flow in the primary side circuit, the current flowing in the switch element is suppressed and the overcurrent is overcurrent. It has a remarkable effect that the failure of the switch element due to the current can be prevented.

(2) In the present invention, when the control unit defines the conduction width of the drive pulse for driving the switch element as the first conduction width in a steady operation state, the control unit defines the current as the current detected by the current detection circuit. When the value is equal to or greater than a predetermined first current value, the conduction width of the drive pulse for driving the switch element is controlled to a second conduction width narrower than the first conduction width (1). ) Is provided.

According to the invention described in (2) above, when the current value detected by the current detection circuit is equal to or greater than the predetermined first current value, the conduction width of the drive pulse that drives the switch element of the converter is set during normal operation. Since the second conduction width is controlled to be narrower than the first conduction width, it is possible to suppress the current value flowing through the converter, and overcurrent protection can be performed by a simple PWM (Pulse Width Modulation) control circuit. It has an excellent effect.

(3) In the present invention, when the current value detected by the control unit in the current detection circuit is equal to or less than a predetermined second current value smaller than the first current value, the conduction width is set to the first. The switching power supply device according to (1) or (2) above, which is characterized by controlling the conduction width.

According to the invention described in (3) above, when the overcurrent is not flowing, the control unit returns the conduction width of the drive pulse of the switch element to the first conduction width in the steady operation state. This has the effect of increasing the transmission efficiency of the converter.

(4) The present invention provides the switching power supply device according to (3) above, wherein the control unit continuously increases the conduction width from the second conduction width to the first conduction width. To do.

According to the invention described in (4) above, the conduction width of the drive pulse of the switch element is continuously increased from the second conduction width to the first conduction width while monitoring the flowing current value with the current detection circuit. It has a remarkable effect that it can be controlled up to a steady operation state without an overcurrent flowing.

(5) Of the above (1) to (4), the present invention is characterized in that at least one converter of the previous stage is further provided on the upstream side of the converter and a predetermined output voltage of the previous stage is output. The switching power supply device according to any one is provided.

According to the invention described in (5) above, when a plurality of converters are provided in the primary circuit,
When the current flowing through the primary side circuit or the secondary side circuit becomes an inappropriate magnitude, the switch elements that make up the converter in the previous stage are controlled based on that value to suppress the current flowing through the converter in the subsequent stage, and the latter stage. It has an excellent effect of being able to protect the switch element of the converter.

(6) In the present invention, when the current value detected by the current detection circuit is equal to or higher than a predetermined third current value, the control unit further controls the switch element of the converter in the previous stage to control the switch element in the previous stage. The switching power supply device according to (5) above, wherein the voltage output from the converter is lowered from the output voltage of the previous stage.

According to the invention described in (6) above, when the current detection circuit detects an overcurrent, the output voltage of the front-stage converter is throttled to suppress the current flowing through the rear-stage converter, and the switch element of the rear-stage converter It has the remarkable effect of being able to protect against overcurrent.

According to the present invention, since the primary side circuit or the secondary side circuit has a current detection circuit that detects the current value of the current flowing through the circuit, a short-circuit accident or other current that is not suitable for the secondary side circuit By controlling the pulse width of the drive pulse that drives the switch element of the converter when it is detected that there is a possibility that the current may flow, the current flowing through the switch element is suppressed and the switch element fails due to overcurrent. It has a remarkable effect of being able to.

It is a block diagram of the switching power supply device which concerns on 1st Embodiment of this invention. (A) It is a timing chart at the time of steady operation of the pulse which drives a switch element included in the 2nd converter which constitutes a switching power supply apparatus. (B) It is a timing chart of a switch element drive pulse when an overcurrent is detected by a current detection circuit. It is a flowchart about the process for controlling the switch element which constitutes the 2nd converter for the purpose of overcurrent protection. It is a block diagram of the switching power supply device which concerns on 2nd Embodiment of this invention. It is a circuit diagram of the circuit which provided the protection circuit which clamps the voltage of the resonance capacitor in the LLC converter which provided the full bridge as a switching circuit. It is a circuit diagram of the circuit which provided the protection circuit which clamps the voltage of the resonance capacitor by the LLC converter which provided the half bridge as a switching circuit. (A) It is a circuit diagram of the circuit which provided the protection circuit which clamps the voltage of a resonance inductor in the LLC converter which provided the full bridge as a switching circuit. (B) It is a circuit diagram of the circuit which provided the protection circuit which clamps the voltage of a resonance inductor in the LLC converter which provided the half bridge as a switching circuit. (A) It is a circuit diagram of the switching power supply device which provided the protection circuit which clamps the voltage of a resonance inductor in the LLC converter which provided the half bridge as a switching circuit. (B) It is a circuit diagram of the switching power supply device which provided the protection circuit which clamps the voltage of a resonance inductor in the LLC converter which provided the half bridge as a switching circuit. (C) It is a circuit diagram of the switching power supply device which provided the protection circuit which clamps the voltage of a resonance inductor in the LLC converter which provided the full bridge as a switching circuit. It is a block diagram of the conventional switching power supply device provided with an overcurrent protection circuit. It is a flowchart of the overcurrent protection processing in the conventional switching power supply device.

Hereinafter, embodiments of the present invention will be described with reference to the accompanying drawings.

1 to 8 are examples of embodiments in which the invention is carried out, and portions having the same reference numerals represent the same objects in the drawings.

FIG. 1 shows a circuit diagram of a switching power supply device according to a first embodiment of the present invention. The switching power supply 1 is a two-stage converter, which has a first converter 10 and a second converter 15, and the second converter 15 includes a primary circuit 25 and a secondary circuit 70 having a rectifying smoothing circuit. A transformer 60 for connecting the primary side circuit 25 and the secondary side circuit 70 is provided. In the steady state, the two-stage converter converts the input voltage input from the input terminal IN of the first-stage first converter 10 located on the upstream side into a predetermined front-stage output voltage and outputs it, and outputs the input voltage from the first-stage converter 10. The output pre-stage output voltage is converted into a predetermined post-stage output voltage of the rear-stage second converter 15. The second converter 15 determines the current value of the current flowing through the smoothing capacitor 20, the full bridge switching circuit 30 including the switch elements Q1 to Q4, and the circuit connecting the switching circuit 30 and the transformer 60 via the series resonance circuit 50. It has a current detection circuit 45 for detecting. The current detection circuit 45 includes a current transformer 40.

The switching power supply device 1 further includes a control unit 80 that controls the conduction width of the drive pulse that drives the switch element included in the switching circuit 30 of the second converter based on the current value detected by the current detection circuit 45. In other words, the control unit 80 is capable of PWM (Pulse Width Modulation) control.

When the current value detected by the current detection circuit 45 is at least a predetermined first current value or more, the control unit 80 controls the switch element of the first converter 10 in the previous stage and outputs the current value from the first converter 10. It also has the function of lowering the current voltage from the previous stage reference output voltage.

The second converter 15 is an LLC series resonance converter, and connects the first leg 32, which connects the first switch element Q1 and the second switch element Q2 in series, the third switch element Q3, and the fourth switch element Q4. It has a full bridge circuit in which the second leg 34 connected in series is connected in parallel between the input terminals IN.

In FIG. 1, diodes are reversely connected to each switch element Q1 to Q4. This diode represents a parasitic diode of the MOSFET (Metal Oxide Semiconductor Field Effect Transistor) itself, which is a switch element.

The connection point H of the first switch element Q1 and the second switch element Q2 in the first leg is of the primary side winding of the transformer 60 via a series resonance circuit 50 including a resonance inductor 52 and a resonance capacitor 54. Connected to one of the terminals. The connection point I of the third switch element Q3 and the fourth switch element Q4 in the second leg is connected to the other end of the primary winding of the transformer 60.

The protection circuit G is connected in parallel to the resonance capacitor 54 of the series resonance circuit 50. That is, a voltage clamp circuit that limits the voltage applied between both ends of the resonance capacitor 54 to a predetermined fluctuation range is provided in parallel with the resonance capacitor 54.

The protection circuit G includes at least two Zener diodes and is connected in series with a common cathode. When the protection circuit G having such a configuration is provided, the voltage related to the resonance capacitor 54 is clamped by the breakdown voltage of the Zener diode 100, so that the current flowing through the series resonance circuit 50 is limited and the second converter 15 is overcurrent. It has a protective effect.

In FIG. 1, the potentials of the connection point B and the connection point C, which are the connection points at both ends of the resonance capacitor 54, are defined as V (B) and V (C), respectively. Assuming that the breakdown voltage of the Zener diode 100 is V (ZD), the voltage applied between both ends of the resonance capacitor 54 is limited to −V (ZD) ≦ V (B) −V (C) ≦ V (ZD). To. In other words, since the absolute value of the voltage applied across the resonance capacitor 54 is limited up to V (ZD), the current flowing in the circuit is limited, and the switch element can be protected from overcurrent. become.

The protection circuit G further exerts the effect of protection from overcurrent by the synergistic effect with the switch element drive pulse width control of the second converter by the control unit 80, which will be described later.

In the present embodiment, the switch element is composed of a MOSFET (Metal Oxide Semiconductor Field Effect Transistor), but may be composed of another switch element, for example, an IGBT (Insulated Gate Bipolar Transistor).

Further, it is desirable that the control unit 80 is composed of a circuit using a semiconductor, a transistor, or the like for controlling a switch element. The resonance inductor 52 is a choke coil. The leakage inductance of the transformer 60 may be used as a resonance inductor instead of the choke coil.

FIG. 2A is a timing chart of the pulses for driving the switch elements Q1 to Q4 included in the second converter 15 included in the switching power supply device 1 during steady operation. This pulse control is performed by the control unit 80. Here, it is driven by a pulse having a duty ratio of about 50%. That is, when the period of the drive pulse is T, first, the first switch element Q1 and the fourth switch element Q4 are controlled so that the ON state lasts for a period of the first conduction width t1 = T / 2. At this time, the second switch element Q2 and the third switch element Q3 are controlled to the OFF state so that a through current does not occur. Next, the second switch element Q2 and the third switch element Q3 are controlled so that the ON state lasts for a period of the first conduction width t1 = T / 2. At this time, the first switch element Q1 and the fourth switch element Q4 are controlled to the OFF state so that a through current does not occur.

FIG. 2B shows a timing chart of the drive pulse that drives the switch element of the second converter 15 when the current detection circuit 45 detects an overcurrent. In the steady operation state, when the control unit 80 defines the conduction width of the drive pulse that drives the switch elements Q1 to Q4 of the second converter 15 as the first conduction width, the control unit 80 determines the current detected by the current detection circuit 45. When the value is equal to or greater than a predetermined first current value, the conduction width of the drive pulse for driving the switch elements Q1 to Q4 is controlled to be narrower than the first conduction width.

That is, in the case of the present embodiment, the conduction width of the drive pulse is changed from t1 = T / 2, which is the first conduction width, to t2 = T / 4, which is the second conduction width, and the duty ratio is changed to about 25%.

The first current value is set so that the current value detected by the current detection circuit 45 exceeds the first current value when a short circuit occurs in the primary side circuit 25 or the secondary side circuit 70.

Generally, in an LLC converter, it is difficult to follow a change in load by PWM control, so PFM control is performed. However, if the purpose is to protect against overcurrent by suppressing the current flowing through the circuit when an overcurrent is detected, changing the duty ratio is sufficient. Therefore, the control unit 80 may be configured not as a complicated PFM control circuit but as a simple PWM control circuit that drives the switch element so that the duty ratio becomes variable.

FIG. 3 shows a flowchart of a process for controlling a switch element constituting the second converter 15 for the purpose of overcurrent protection. The current detection circuit 45 detects the current flowing in the circuit of the second converter 15 (step S10). Then, the control unit 80 detects the average current (step S20), and in step 30, determines whether or not the average current is equal to or greater than the first current value. This first current value is set to a value that detects only a so-called overcurrent. Here, when a short circuit occurs in the primary side circuit 25 or the secondary side circuit 70, the current value detected by the current detection circuit 45 is equal to or higher than the first current value. It is desirable that the first current value is a current value at which the switch elements Q1 to Q4 of the second converter 15 do not fail. If the average current is equal to or greater than the first current value (YES), the control unit 80 sets the conduction width of the pulse driving the switch elements Q1 to Q4 of the second converter 15 from the first conduction width t1 in the steady operation state. , The second conduction width is changed to t2 (see FIG. 2), the duty ratio is lowered to about 25% (step S40), and the process returns to the beginning (step S10). If the average current is less than the first current value (NO), it is further determined whether the average current is equal to or less than the second current value (step S50). Here, the second current value is set smaller than the first current value. By providing a difference between the first current value and the second current value, chattering with a duty ratio of 25% and a duty ratio of 50% can be prevented at the time of overcurrent.

The second current value is set to the current value that flows through the circuit of the second converter 15 when the switching power supply device 1 is driven at a duty ratio of about 25% without any problem. When the average current is equal to or less than the second current value (YES), the control unit 80 continuously increases the drive pulse width of the switch element of the second converter 15 until the duty ratio becomes approximately 50%. That is, the control unit 80 continuously increases the conduction width from the second conduction width t2 to the first conduction width t1 (see FIG. 2). Even in this process, the current detection circuit 45 constantly monitors the current value flowing in the circuit of the second converter 15 (returning to step S10). When the average current is larger than the second current value (NO), the process returns to step S10. By the above processing, the switch elements Q1 to Q4 constituting the second converter 15 in the subsequent stage are protected from the overcurrent, and if there is no problem, the switch elements Q1 to Q4 return to the steady operation state (duty ratio of about 50%).

Although not described in this flowchart (see FIG. 3), at the same time as the processing shown in FIG. 3, the control unit 80 controls the switch element of the first converter 10 in the previous stage on the upstream side of the second converter 15. , The process of lowering the voltage output from the first converter 10 from the previous stage reference output voltage is performed.

Specifically, for example, when a third current value smaller than the first current value is set and the current value detected by the current detection circuit 45 becomes equal to or higher than the third current value, the control unit 80 switches the converter 10. Controls to raise the frequency. This control lowers the transmission efficiency of the converter 10, suppresses the current flowing through the second converter 15 in the subsequent stage, and has a remarkable effect that the second converter 15 can be protected more effectively. Play.

Needless to say, the above-mentioned processing for the converter 10 in the previous stage can be applied to other embodiments and other modified examples.

FIG. 4 is a switching power supply device 1 according to a second embodiment of the present invention. The primary side circuit 25 is a single-stage LLC current resonance converter, and the switching circuit 30 includes a full bridge circuit. Similar to the second converter 15 of the first embodiment, the switch elements constituting the full bridge circuit are driven at a duty ratio of about 50% during steady operation.

Similar to the first embodiment including the multi-stage converter shown in FIG. 1, when an overcurrent flows through any of the switch elements Q1 to Q4 constituting the switching circuit 30 of the converter of the primary side circuit 25, the current detection circuit The overcurrent is detected at 45, and the Duty ratio is controlled to approximately 25% by the control unit 80. The timing charts of the drive pulses of the switch elements Q1 to Q4 and the flow charts of the processes are the same as those in FIGS. 2 and 3, respectively, and are omitted. Further, protection circuits G for voltage clamping are provided at both ends of the current resonance capacitor 54.

The protection circuit G is connected in parallel to the resonance capacitor 54 of the series resonance circuit 50. That is, a voltage clamp circuit that limits the voltage applied between both ends of the resonance capacitor 54 to a predetermined fluctuation range is provided in parallel with the resonance capacitor 54.

The protection circuit G includes at least two Zener diodes 100 and is connected in series with a common cathode. When the protection circuit G having such a configuration is provided, the voltage applied between both ends of the resonance capacitor 54 is clamped by the breakdown voltage of the Zener diode 100, so that the current flowing through the series resonance circuit 50 is limited and the second converter At 15, the effect of overcurrent protection is achieved.

In FIG. 4, the potentials of the connection point B and the connection point C, which are the connection points at both ends of the resonance capacitor 54, are defined as V (B) and V (C), respectively. Assuming that the breakdown voltage of the Zener diode 100 is V (ZD), the voltage applied between both ends of the resonance capacitor 54 is limited to −V (ZD) ≦ V (B) −V (C) ≦ V (ZD). To. In other words, since the absolute value of the voltage applied across the resonance capacitor 54 is limited up to V (ZD), the current flowing in the circuit is limited, and the switch element can be protected from overcurrent. become.

The above configuration is sufficiently effective as overcurrent protection of the switch elements Q1 to Q4 of the primary side circuit 25 against a short circuit.

It should be noted that the switching power supply device of the present invention is not limited to the above-described embodiment, and it goes without saying that various modifications can be made without departing from the gist of the present invention.

For example, in FIGS. 5 to 8, as an application of the first embodiment and the second embodiment, another modified embodiment in which the voltage is clamped at the connection point in the circuit by another method and the overcurrent protection effect is obtained. Is shown.

In FIGS. 5 to 8, the control unit 80 controls for protecting the switch element from overcurrent based on the current detection circuit 45 that detects the current in the circuit and the detection result of the current detection circuit 45. , And the secondary side circuit is not shown. It is desirable that the control unit 80 protects the switch element by suppressing the current value of the pulse drive of the switch element by changing the duty ratio as in the first embodiment. The process is the same as in FIGS. 2 and 3 described above, and is therefore omitted.

FIG. 5 shows a circuit diagram of a circuit provided with a protection circuit for clamping the potential at one end of the terminal of the current resonance capacitor 54 in an LLC converter provided with a full bridge as a switching circuit.

In the switching power supply device 1 of FIG. 5A, the protection circuit E and the protection circuit F are connected to the connection point B for connecting the resonance capacitor 54 and the resonance inductor 52 in series. In this case, each protection circuit includes one diode and one Zener diode, and the cathode terminals of each are connected in series. Regarding the protection circuit E, the anode terminal of the diode is connected to the connection point B, and the anode terminal of the Zener diode is connected to the positive electrode side of the input terminal IN. Regarding the protection circuit F, the anode terminal of the Zener diode is connected to the connection point B, and the anode terminal of the diode is connected to the negative electrode side of the input terminal IN.

First, the potential of the positive electrode of the input terminal IN in the primary circuit is defined as V (A), and the potential of the negative electrode is set to 0. Further, the potential of each connection point B is represented by V (B), and the yield voltage of the Zener diode is represented by V (ZD). In the protection circuit E, when the potential V (B) of the connection point B becomes V (B) = V (A) + V (ZD), the connection point B is short-circuited to the positive electrode of the primary circuit, and V ( B) is V (B) ≤ V (A) + V (ZD) (assuming that the voltage drop of the diode 120 is sufficiently smaller than V (ZD)). Further, the protection circuit F short-circuits the connection point B to the negative electrode of the primary side circuit when the potential V (B) at the connection point B becomes V (B) = −V (ZD). That is, by connecting the protection circuit E and the protection circuit F to the connection point B, the potential V (B) at the connection point B can be changed to −V (ZD) ≦ V (B) ≦ V (A) + V (ZD). Limit within range. By clamping the potential at one end of the resonance capacitor 54 in this way, the voltage applied to the resonance capacitor 54 is limited, and the current flowing in the circuit is suppressed.

In the switching power supply device 1 shown in FIG. 5B, in the full bridge circuit, the connection point H between the first switch element Q1 and the second switch element Q2 in the first leg 32 is the terminal of the primary winding of the transformer 60. Connected to one. The connection point I between the third switch element Q3 and the fourth switch element Q4 of the second leg 34 is a terminal of the primary winding of the transformer 60 via a series resonance circuit 50 including a resonance capacitor 54 and a leakage inductor 130. Connected to the other of. In this case, the leakage inductor 130 in the figure is the leakage inductance of the transformer 60 and is used as a resonance inductor.

FIG. 6 is an LLC converter provided with a half bridge as a switching circuit. The potential at one end of the resonance capacitor 54 is clamped by the protection circuit E and the protection circuit F. That is, in each of FIGS. 6 (A) to 6 (C), by providing the protection circuit E and the protection circuit F, the potential V (B) at the connection point B, which is one end of the resonance capacitor 54, becomes −V. It is limited to the range of (ZD) ≤ V (B) ≤ V (A) + V (ZD) (assuming that the voltage drop of the diode 120 is sufficiently smaller than V (ZD)). By clamping the potential at one end of the resonance capacitor 54 in this way, the voltage applied to the resonance capacitor 54 is limited, and the current flowing in the circuit is suppressed. In FIGS. 6 (A) and 6 (B), the leakage inductor 130 in the figure represents one that uses the leakage inductance of the transformer 60 as a resonance inductor.

FIG. 7A shows a circuit in which an LLC converter equipped with a full bridge as a switching circuit is provided with a protection circuit for clamping the voltage of the resonance inductor 52. By connecting the protection circuit E and the protection circuit F to the connection point B of the resonance inductor 52 and the resonance capacitor 54, the potential V (B) of the connection point B can be changed to −V (ZD) ≦ V (B) ≦. It is limited to the range of V (A) + V (ZD) (assuming that the voltage drop of the diode 120 is sufficiently smaller than V (ZD)). The connection of the protection circuit is different from the cases of FIGS. 5 and 6 described above in that the connection of the protection circuit limits the potential at one end of the resonance inductor 52 instead of the resonance capacitor 54. That is, instead of limiting the voltage applied to the resonance capacitor 54, the series resonance of the resonance inductor 52, the resonance capacitor 54, the leakage inductor 130, and the excitation inductor (not shown) by the primary winding of the transformer 60 is performed. It changes the resonance mode itself. With this configuration, the current flowing through the circuit can be suppressed.

Specifically, even if a large current tries to flow from the smoothing capacitor to the primary circuit 25 due to a short circuit of the secondary circuit 70 or the like, the protection circuit E and the protection circuit F provide the potential V at the connection point B ( Since B) is limited to the range of −V (ZD) ≦ V (B) ≦ V (A) + V (ZD) (assuming that the drop voltage of the diode 120 is sufficiently smaller than V (ZD)). The inductor 52 does not accumulate energy by a magnetic field beyond a certain level. That is, the resonance frequency of the series resonance is mainly determined by the resonance capacitor 54 and the leakage inductor 130, and the resonance frequency rises. In other words, since the inductor 52 operates as a voltage source, the inductance component of the inductor 52 disappears, and resonance is performed at the resonance frequency of the resonance capacitor 54 and the leakage inductor 130. When the resonance frequency increases, the transmission efficiency decreases, so that the current flowing through the primary circuit 25 can be suppressed.

7 (B), 8 (A), and 8 (B) all show an LLC converter having a half bridge as a switching circuit, and protect the potential at one end of the resonance inductor 52. It is a circuit diagram of a circuit provided with a circuit. By providing the protection circuit E and the protection circuit F at the connection point B between the resonance inductor 52 and the resonance capacitor 54, the potential V (B) at the connection point B can be changed to −V (ZD) ≦ V (B) ≦ V ( A) Limit to the range of + V (ZD) (assuming that the voltage drop of the diode 120 is sufficiently smaller than V (ZD)). With this configuration, the current flowing through the circuit can be suppressed.

Further, the switching power supply 1 shown in FIG. 8C is an LLC converter provided with a full bridge as a switching circuit. The protection circuit G is provided so as to be connected to the inductor 52 in parallel. The protection circuit G includes at least two Zener diodes and is connected in series with a common cathode. When the protection circuit G having such a configuration is provided, the voltage applied to the inductor 52 is clamped by the yield voltage of the Zener diode 100. In this case as well, since the inductor 52 operates as a voltage source, the inductor 52 loses its property as an inductance, and resonance is performed at the resonance frequency of the resonance capacitor 54 and the leakage inductor 130. When the resonance frequency increases, the transmission efficiency decreases, so that the current flowing through the primary circuit 25 can be suppressed.

Although the two Zener diodes 100 are connected by a common cathode, they may be connected in series by a common anode. Further, in FIG. 8C, an LLC converter having a full bridge as a switching circuit is shown, but a half bridge may be provided.

In the modified embodiment of FIGS. 5 to 8, a mode in which one converter is provided in the primary circuit 25 has been described. However, the illustrated converter is used as a rear converter, and a multi-stage converter having a front converter on the upstream side thereof is provided. The primary side circuit 25 may be configured by the converter. In that case, as in the first embodiment, the control unit 80 controls the switch element of the converter in the previous stage based on the current value detected by the current detection circuit 45, and performs a process of reducing the output voltage of the converter in the previous stage. Is desirable.

1 Switching power supply 10 1st converter 15 2nd converter 20 Smoothing capacitor 25 Primary side circuit 30 Switching circuit 40 Current transformer 45 Current detection circuit 50 Series resonance circuit 52 Resonance inductor 54 Resonance capacitor 60 Transformer 70 Secondary side circuit 80 Control Part 90 Control part 100 Zener diode 120 Diode 130 Leakage inductor 140 Parasitic capacitance IN Input terminal IN
OUT output terminal OUT
E, F, G protection circuit Q1 1st switch element Q2 2nd switch element Q3 3rd switch element Q4 4th switch element

Claims (4)

A primary circuit with a converter containing at least two switch elements,
A secondary circuit with a rectifying smoothing circuit and
A transformer that connects the primary side circuit and the secondary side circuit,
The primary side circuit comprises a series circuit connected in series to the primary side winding of the transformer and comprising a capacitor and at least one inductor.
In the primary side circuit, the LLC series resonance converter is configured by the switch element, the transformer, and the series circuit, and further.
A current detection circuit that detects the current value of the current flowing through the primary side circuit or the secondary side circuit, and
A control unit for controlling the conduction width of the drive pulse of the switch element of the LLC series resonance converter based on the current value detected by the current detection circuit is provided .
When the conduction width of the drive pulse in which the control unit drives the switch element in the steady operation state is defined as the first conduction width,
When the current value detected by the current detection circuit is equal to or greater than the first current value meaning an overcurrent, the control unit sets the conduction width of the drive pulse for driving the switch element from the first conduction width. Is designed to switch to a narrow second conduction width, and further
At least one converter in the previous stage is provided on the upstream side of the LLC series resonance converter.
When the current value detected by the current detection circuit is equal to or greater than the third current value smaller than the first current value, the control unit controls the switch element of the converter in the previous stage from the converter in the previous stage. A switching power supply that lowers the output voltage . The switching power supply device according to claim 1, wherein the control unit controls the switching frequency of the switch element of the converter in the previous stage to reduce the voltage output from the converter in the previous stage . When the current value detected by the current detection circuit is equal to or less than a predetermined second current value smaller than the first current value, the control unit determines the conduction width of the drive pulse of the LLC series resonance converter. The switching power supply device according to claim 1 or 2, wherein the first conduction width is controlled. The switching power supply device according to claim 3 , wherein the control unit continuously increases the conduction width of the drive pulse of the LLC series resonance converter from the second conduction width to the first conduction width. ..

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