JP6833071B2 - Drive device, fluid utilization device and air conditioner - Google Patents

Description

The present invention relates to a drive device for driving two synchronous motors connected in parallel to one power converter, a fluid utilization device including the drive device, and an air conditioner including a fluid utilization device.

If the synchronous electric motor does not apply an appropriate voltage according to the rotation position by using the information about the rotor position which is the rotation position of the rotor, the torque cannot be generated properly and the synchronous motor can step out or stop the operation. There is sex. Therefore, when driving a plurality of synchronous motors, conventionally, the same number of power converters as the number of synchronous motors are used in order to apply a voltage corresponding to the rotation position of the rotor of each synchronous motor. Was there. However, in this configuration, as the number of synchronous motors increases, the number of power converters increases, so that an increase in cost becomes an issue. Therefore, in recent years, with the advancement of control technology for synchronous motors, attempts have been made to drive two synchronous motors with one power converter.

In Patent Document 1, of two synchronous motors connected to one power converter, one is a main side synchronous motor and the other is a sub side synchronous motor, and two synchronous motors are driven. A method of controlling by using the rotation speed of a synchronous motor of a stand is disclosed. In the method described in Patent Document 1, the drive device vector-controls the main-side synchronous motor, and the d-axis current of the main-side synchronous motor is used by using the difference in the rotational speed and the difference in the rotational position between the two synchronous motors. By deciding the command, the drive of the sub-side synchronous motor is stabilized. Vector control is a control method that decomposes the current flowing through a synchronous motor into a current component that generates torque and a current component that generates magnetic flux, and controls each current component independently.

U.S. Patent Application Publication No. 2015/0229245

The method described in Patent Document 1 assumes a case where the motor constants of the two synchronous motors are the same. However, even if the synchronous motors have the same specifications, the motor constants of the two synchronous motors do not always match. Therefore, in the method described in Patent Document 1, if the motor constant fluctuates, an erroneous determination of the compensation direction of the magnetic flux current may occur. As a result, speed unevenness (unevenness) occurs in the rotation speed of the sub-side synchronous motor. This speed unevenness is that the rotation speed of the rotor that tries to rotate at a constant speed fluctuates. As described above, since the method described in Patent Document 1 has low robustness to fluctuations in the motor constant, the vibration and noise of the sub-side synchronous motor increase due to the uneven rotation speed of the sub-side synchronous motor. In addition, there is a problem that the motor efficiency is lowered.

The present invention has been made in view of the above, and obtains a drive device capable of improving robustness against fluctuations in motor constants even when two synchronous motors are driven by using one power converter. With the goal.

In order to solve the above-mentioned problems and achieve the object, the drive device of the present invention is a first synchronization with a power converter that supplies power to a first synchronous motor and a second synchronous motor connected in parallel. It includes a first current detector that detects a first current flowing through the electric motor, and a second current detector that detects a second current flowing through the second synchronous motor. The drive device has a first magnetic pole position detecting unit that detects the first magnetic pole position of the rotor of the first synchronous motor, and a second magnetic pole position of the rotor of the second synchronous motor. A control unit that outputs a voltage command for driving a first synchronous motor using a torque current command, a magnetic flux current command, a first current, and a first magnetic pole position. Be prepared. The drive device is based on the difference in the motor constants of the subtractor for obtaining the first angle difference, which is the difference between the first magnetic pole position and the second magnetic pole position, and the first synchronous motor and the second synchronous motor. Or, based on the information indicating the electrical behavior caused by the difference in the motor constants, the angle difference correction unit that corrects the first angle difference and outputs the second angle difference that is the corrected angle difference, and the second It includes a pulsating component extraction unit that extracts at least one pulsating component of the pulsating component included in the torque current flowing through the synchronous motor 2 and the active power pulsating component contained in the active power consumed by the second synchronous motor. The drive device is characterized by including a magnetic flux current command determining unit that determines the compensation direction of the magnetic flux current command using the second angle difference and determines the magnetic flux current command using the determined compensation direction and the pulsating component. To do.

According to the present invention, even when two synchronous motors are driven by one power converter, the robustness against fluctuations in the motor constant can be improved.

The figure which shows the structure of the drive device which concerns on Embodiment 1 of this invention. The figure which shows the 1st structural example of the pulsating component extraction part shown in FIG. The figure which shows the 2nd structural example of the pulsating component extraction part shown in FIG. The figure which shows the 3rd structural example of the pulsating component extraction part shown in FIG. The first figure for demonstrating the behavior of the torque change of the sub-side synchronous motor shown in FIG. The second figure for demonstrating the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 3 for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 4 for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 5 for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 6 for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. 6 is a diagram showing the d-axis current on the main side shown in FIGS. 6, 7, 9 and 10 in association with the sign of the angle difference and the torque state of the sub-side synchronous motor. The figure which shows the structural example of the magnetic flux current command determination part shown in FIG. FIG. 1 for explaining a code determination process by the code determination device shown in FIG. FIG. 2 for explaining the code determination process by the code determination device shown in FIG. FIG. 3 for explaining the code determination process by the code determination device shown in FIG. The figure which shows the relationship between the angle difference correction amount calculated by the angle difference correction part shown in FIG. 1 and the rotation speed of a rotor. The figure which shows the 1st structural example of the angle difference correction part shown in FIG. The figure which shows the 2nd structural example of the angle difference correction part shown in FIG. Configuration diagram of the current control unit included in the drive device according to the second embodiment of the present invention. Configuration diagram of the angle difference correction unit included in the drive device according to the second embodiment of the present invention. The figure for demonstrating the torque current waveform of the sub side synchronous motor when the armature resistance of the main side synchronous motor shown in FIG. 1 is added with an error of + 30%. The figure for demonstrating the torque current waveform of the sub side synchronous motor when an error of -30% is added to the armature resistance of the main side synchronous motor shown in FIG. Configuration diagram of the angle difference correction unit included in the drive device according to the third embodiment of the present invention. The figure which shows the image of the automatic search of the angle difference correction amount by the mountain climbing method used for the automatic search part shown in FIG. Configuration diagram of the angle difference correction unit included in the drive device according to the fourth embodiment of the present invention. Configuration diagram of the fluid utilization device according to the fifth embodiment of the present invention. Configuration diagram of the air conditioner according to the sixth embodiment of the present invention.

Hereinafter, the drive device, the fluid utilization device, and the air conditioner according to the embodiment of the present invention will be described in detail with reference to the drawings. The present invention is not limited to this embodiment.

Embodiment 1.
FIG. 1 is a diagram showing a configuration of a drive device according to a first embodiment of the present invention. Synchronous motors utilize a permanent magnet field-type synchronous motor in which a permanent magnet is provided in the rotor, a winding field-type synchronous motor in which a field winding is wound around the rotor, and the salient polarity of the rotor. It is roughly divided into a relaxation type synchronous motor that obtains rotational torque. Permanent magnet field synchronous motors include surface magnet type synchronous AC motors in which permanent magnets are provided on the outer peripheral surface of the rotor core, and permanent magnet embedded motors in which permanent magnets are embedded inside the rotor core. It is roughly divided. The drive device 100 according to the first embodiment may use two synchronous motors of the same type among these types of synchronous motors, or two different types of synchronous motors. Further, in the first embodiment, it is described that each of the two synchronous motors is a three-phase synchronous motor, but the two synchronous motors are synchronized with a number of phases other than the three phases such as two-phase and five-phase. An electric motor may be used.

In the first embodiment, one of the two synchronous motors may be referred to as a "main side synchronous motor 1a" and the other may be referred to as a "sub-side synchronous motor 1b". The main side synchronous motor 1a is the first synchronous motor, and the sub side synchronous motor 1b is the second synchronous motor.

In the first embodiment, the motor constants of the two synchronous motors may be the same or different. However, even if two synchronous motors having the same specifications are used as the two synchronous motors, the motor constants may differ. Factors that cause different motor constants in synchronous motors with the same specifications include variations in the dimensions of the parts that make up each of the main-side synchronous motor 1a and sub-side synchronous motor 1b during manufacturing, and the main-side synchronous motor 1a and sub-side synchronous motor 1a. There is a difference in temperature generated by each drive of 1b. The factors that cause the motor constants to differ include the difference between the impedance of the wiring provided from the main side synchronous motor 1a to the power converter 2 and the impedance of the wiring provided from the sub side synchronous motor 1b to the power converter 2. There is. The factors that cause the motor constants to differ are the difference in load torque between the main side synchronous motor 1a and the sub side synchronous motor 1b, and the difference in the magnetic saturation points of the inductances of the main side synchronous motor 1a and the sub side synchronous motor 1b. and so on. The magnetic material is magnetized linearly with respect to the applied magnetic field in a weak magnetic field, but is not linearly magnetized when a magnetic field of a certain strength or higher is applied. Such a phenomenon of not being magnetized linearly is called magnetic saturation, and the portion where the magnetic flux is not linearly magnetized and becomes a constant value is the above-mentioned magnetic saturation point.

The drive device 100 detects the power converter 2 that supplies power to the main side synchronous motor 1a and the sub side synchronous motor 1b that are connected in parallel, the relay circuit 11, and the first current that flows through the main side synchronous motor 1a. It includes a current detection unit 4a and a current detection unit 4b that detects a second current flowing through the sub-side synchronous motor 1b.

The power converter 2 converts the DC power supplied from the DC voltage source 3 into AC power and outputs it to the main side synchronous motor 1a and the sub side synchronous motor 1b. In the first embodiment, a voltage inverter is used for the power converter 2. The voltage type inverter is a device that switches the DC voltage supplied from the DC voltage source 3 and converts it into an AC voltage. The power converter 2 is not limited to the voltage type inverter as long as it can output the AC power for driving the main side synchronous motor 1a and the sub side synchronous motor 1b, and the current type inverter and the AC power are oscillated. A circuit such as a matrix converter that converts AC power having different frequencies, or a multi-level converter in which the outputs of a plurality of converters are connected in series or in parallel may be used.

The relay circuit 11 is used when only one of the two synchronous motors is driven. In FIG. 1, it is provided in the wiring for connecting the sub-side synchronous motor 1b to the power converter 2. The relay circuit 11 is used when the number of synchronous motors driven by the power converter 2 is switched from two to one, or from one to two. The contacts of the relay circuit 11 are controlled by the presence or absence of a switching signal output from a switching unit (not shown). When two synchronous motors are driven at the same time, the contacts of the relay circuit 11 are closed, and the sub-side synchronous motor 1b is electrically connected to the power converter 2 via the relay circuit 11.

The relay circuit 11 may be provided in the wiring for connecting the main side synchronous motor 1a to the power converter 2 instead of being provided in the wiring for connecting the sub-side synchronous motor 1b to the power converter 2. Further, the relay circuit 11 may be composed of a mechanical relay or a semiconductor switch. Further, when the two synchronous motors are always driven, the drive device 100 may omit the relay circuit 11 and directly connect the main side synchronous motor 1a and the sub side synchronous motor 1b to the power converter 2.

The current detection unit 4a, which is the first current detector, detects the phase current flowing from the power converter 2 to the main synchronous motor 1a, and outputs current information indicating the value of the detected phase current. The current detection unit 4b, which is the second current detector, detects the phase current flowing from the power converter 2 to the sub-side synchronous motor 1b, and outputs current information indicating the value of the detected phase current.

The current detection units 4a and 4b may be a current sensor using a current transformer for an instrument called a CT (Curent Transformer), or a current sensor using a shunt resistor. Further, the current detection units 4a and 4b may be a combination of these. In the drive device 100 according to the first embodiment, the current is detected by the current detection units 4a and 4b provided near the synchronous motor. In the example shown in FIG. 1, the phase current flowing through the synchronous motor is directly detected, but the current detection method is not limited to the example of directly detecting as long as the current flowing through the synchronous motor can be calculated according to Kirchhoff's current law. For example, a one-shunt current detection method using a shunt resistor provided on the negative DC bus of the power converter 2, a lower arm shunt current detection method using a shunt resistor connected in series with the lower arm of the power converter 2, and the like. May be used to detect the phase current flowing through the synchronous motor. In the case of the three-phase power converter 2, the lower arm shunt current detection method is also called a three-shunt current detection method because it uses a shunt resistor connected in series to each of the three lower arms. However, in the 1-shunt current detection method or the 3-shunt current detection method, only the total value of the currents flowing through the main side synchronous motor 1a and the sub side synchronous motor 1b is measured, so that the main side synchronous motor 1a and the sub side synchronous motor 1a and the sub side synchronous motor 1b are measured. It is necessary to provide a current sensor near one of the synchronous motors in 1b. Needless to say, in the case of a three-phase synchronous electric motor, if a current sensor is provided in any two-phase wiring of the three-phase wiring connected to the synchronous electric motor, the current of the remaining one phase will be the Kirchhoff's current. Since it can be calculated by the current law, it is not necessary to provide a current sensor for all three-phase wiring. Various methods can be considered for the configuration and arrangement of the current detection unit 4a and the current detection unit 4b, but basically any method may be used.

Further, the drive device 100 includes a magnetic pole position detection unit 5a which is a first magnetic pole position detection unit, a magnetic pole position detection unit 5b which is a second magnetic pole position detection unit, and a current control which is a control unit which outputs a voltage command. A unit 6, a subtractor 8, a pulsating component extraction unit 7, an angle difference correction unit 10, and a magnetic flux current command determination unit 9 are provided.

The magnetic pole position detection unit 5a detects the magnetic pole position of the rotor of the main-side synchronous motor 1a and outputs a signal indicating the magnetic pole position. The magnetic pole position detection unit 5b detects the magnetic pole position of the rotor of the sub-side synchronous motor 1b and outputs a signal indicating the magnetic pole position.

Each of the magnetic pole position detecting unit 5a and the magnetic pole position detecting unit 5b is, for example, a rotary encoder, a resolver, a hall sensor, or the like. The rotary encoder converts the amount of mechanical displacement of the rotor into an electric signal, processes the converted signal, and outputs a signal indicating the magnetic pole position. The resolver is composed of a combination of an exciting coil and two detection coils orthogonal to each other. When a sinusoidal signal is input to the exciting coil, the voltage output from the detection coil changes in proportion to the rotation angle of the rotor. This change in voltage is output as a signal indicating the position of the magnetic pole of the rotor. The Hall sensor detects the magnetic field generated by the magnet or the magnetic field generated by the current flowing through the coil using the Hall effect, converts the detected magnetic field into an electric signal, and converts this electric signal into the magnetic pole position of the rotor. Output as a signal to indicate.

Each of the magnetic pole position detecting unit 5a and the magnetic pole position detecting unit 5b is not limited to the rotary encoder, the resolver, or the Hall sensor as long as the magnetic pole position of the rotor can be detected. For example, the magnetic pole position detection unit 5a estimates the magnetic pole position of the main side synchronous motor 1a by using the phase current flowing through the main side synchronous motor 1a and the voltage command output from the current control unit 6. May be good. Similarly, the magnetic pole position detecting unit 5b estimates the magnetic pole position of the sub-side synchronous motor 1b by using the phase current flowing through the sub-side synchronous motor 1b and the voltage command output from the current control unit 6. You may.

There are various methods for estimating the magnetic pole position, but in the medium and high speed range of the entire rotation speed of the rotor provided by the synchronous motor, the magnetic pole position is determined by using the information indicating the speed electromotive force of the synchronous motor. It is common to estimate. The velocity electromotive force is an induced power generated inside the synchronous motor due to the rotation of the rotor, and is proportional to the field generated between the rotor and the stator of the synchronous motor and the rotation speed of the rotor. There is also a method of estimating the magnetic pole position from the salient polarity of the inductance by investigating the resage locus of the high-frequency current generated when a high-frequency voltage is applied to the synchronous motor in the low-speed range within the entire rotation speed of the rotor. The Lissajous locus represents the time change of the magnetic flux density as a locus on a two-dimensional coordinate plane.

In order to control the current flowing through the main-side synchronous motor 1a, the current control unit 6 sets the direction of the magnetic flux generated by the permanent magnet of the rotor of the main-side synchronous motor 1a as the d-axis and the axis orthogonal to the d-axis as the q-axis. This is a vector controller that converts the current detected by the current detection unit 4a into the current command value of the dq coordinate system by vector control. In a general vector controller, current control is performed on the dq coordinates with reference to the magnetic pole of the rotor. This is because when the phase current is converted into a value on the dq coordinate, the amount of alternating current becomes the amount of direct current, which facilitates control. In a synchronous motor, the q-axis current is proportional to the magnet torque of the synchronous motor, so the q-axis is referred to as the "torque shaft" and the q-axis current is referred to as the "torque current". Since the d-axis current changes the magnetic flux generated in the stator and changes the amplitude of the output voltage of the synchronous motor with respect to the q-axis current, the d-axis is called the "magnetic flux axis" and the d-axis current is ". It is called "magnetic flux current" or "exciting current". In the above-mentioned permanent magnet embedded motor, the reluctance torque changes depending on the d-axis current, so not only the q-axis current acts on the torque, but the q-axis current is generally called the torque current. There are many.

The magnetic pole position detected by the magnetic pole position detection unit 5a is used for the coordinate conversion. In addition to the dq coordinate system in vector control, the current control unit 6 may use a polar coordinate system such as an αβ stator coordinate system or a γδ coordinate system. Further, direct torque control (Direct Torque Control: DTC) may be adopted for the current control unit 6 instead of the vector control. However, when DTC is adopted, it is necessary to convert the current command into the magnetic flux command and the torque command.

If the control is performed using the coordinate system based on the magnetic flux generated from the stator instead of the dq coordinate system, the torque current and the magnetic flux current can be calculated more accurately. This coordinate system is often referred to as an ft coordinate system, an nt coordinate system, or the like, but since it is known, detailed description thereof will be omitted. In the first embodiment, the q-axis current may be referred to as "torque current" and the d-axis current may be referred to as "magnetic flux current". However, when controlling using a coordinate system other than the dq coordinate system, magnet torque is the principle. This does not apply when using a relaxation synchronous motor that does not generate current.

The current control unit 6 is controlled so that the torque current flowing through the main side synchronous motor 1a matches the value of the torque current command, and the magnetic flux current flowing through the main side synchronous motor 1a matches the value of the magnetic flux current command. Is controlled. The specific implementation method of the current control unit 6 may be any method, but the current control unit 6 is generally composed of a proportional integration controller and a non-interfering controller. The torque / current command may be calculated as a result of speed control by the magnetic flux current command determination unit 9, or may be input from a higher-level controller. The details of the magnetic flux current command will be described later.

When the main side synchronous motor 1a is vector-controlled by the current control unit 6, the sub side synchronous motor 1b is driven around the main side synchronous motor 1a, so that the sub side synchronous motor 1b is in an open loop drive state. Become. Reference 1 below is a well-known paper on open-loop driving of synchronous motors.
(Reference 1) Junichi Ito, Jiro Toyosaki, Hiroshi Osawa, "High Performance of V / f Control of Permanent Magnet Synchronous Motors", IEEJ Transactions D, Vol. 122 (2002) No. 3, P253-259

According to Reference 1 above, it is stated that when the synchronous motor is driven in an open loop, the synchronous motor _{self-oscillates at an intrinsic angular frequency ω n} , and the control may become unstable. The natural angular frequency ω _n is represented by the approximate expression of the following equation (1). However, P _m is the number of pole pairs, [Phi _a is armature flux linkage number, L _a is the armature inductance, J represents a moment of inertia.

Since the electromechanical coupled vibration is sometimes called the electric spring resonance, the natural angular frequency ω _n represented by the above equation (1) is also called the electric spring resonance angle frequency. A stabilization compensator is added to the technique disclosed in Reference 1 in order to suppress the resonance of the electric spring, but the drive device 100 also requires the same stabilization compensation. Therefore, it is necessary to investigate how much the torque current flowing through the sub-side synchronous motor 1b shown in FIG. 1 vibrates due to the resonance of the electric spring.

A component due to acceleration / deceleration torque and a component due to load torque are superimposed on the torque current flowing through the sub-side synchronous motor 1b. The pulsating component extraction unit 7 shown in FIG. 1 extracts pulsating components near the electric spring resonance angular frequency included in the torque current of the sub-side synchronous motor 1b. There are two methods for the pulsation component extraction unit 7 to extract the pulsation component near the electric spring resonance angle frequency, a method using a high-pass filter and a method using a band-pass filter, and these will be described in order. ..

FIG. 2 is a diagram showing a first configuration example of the pulsating component extraction unit shown in FIG. FIG. 2 shows a configuration example of the pulsating component extraction unit 7A using a first-order high-pass filter, and its transfer function is represented by the following equation (2). However, s is the Laplace transform operator and ω _c is the cutoff angular frequency.

The transfer function when a first-order high-pass filter is used is shown in the above equation (2). However, when a steeper filter characteristic is desired, a high-pass filter having an order of n may be used. n is an integer of 2 or more. When a high-pass filter is used, it _{is preferable to set the cutoff angle frequency ω c} to a value of 1/3 or less of the electric spring resonance angle frequency, for example, 1/5 to 1/20 of the electric spring resonance angle frequency.

FIG. 3 is a diagram showing a second configuration example of the pulsating component extraction unit shown in FIG. FIG. 3 shows a configuration example of the pulsating component extraction unit 7B using a second-order bandpass filter, and its transfer function is represented by the following equation (3). However, s represents the Laplace transform operator and ω _p represents the peak angular frequency. q is a quality factor, a factor that determines the passband bandwidth of the filter.

The transfer function when a second-order bandpass filter is used is shown in the above equation (3), but if a steeper filter characteristic is desired, a bandpass filter having a degree of m may be used. m is an integer of 3 or more. When a bandpass filter is used, the pulsating component extraction unit 7B _{matches the peak angle frequency ω p} with the electric spring resonance angle frequency. However, although not mentioned in Reference 1 above, the electric spring resonance angular frequency has a property of fluctuating depending on the driving conditions. Therefore, it is necessary to design the pass band width of the bandpass filter to be wide in anticipation of the fluctuation of the electric spring resonance angular frequency. It should be noted that the electric spring resonance angle frequency may be actually measured and the peak angle frequency ω _p may be tracked to the electric spring resonance angle frequency. In that case, the passing bandwidth can be narrowed.

Instead of performing the calculation of the above equation (3), a bandpass filter using Fourier series expansion may be used as shown in FIG. FIG. 4 is a diagram showing a third configuration example of the pulsating component extraction unit shown in FIG. The pulsation component extraction unit 7C shown in FIG. 4 includes a pulsation frequency measurement unit 71, a cosine wave generator 72, a sine wave generator 73, a Fourier cosine coefficient calculation unit 74, a Fourier sine coefficient calculation unit 75, and an AC restorer 76.

The pulsation frequency included in the input signal, which is the current detected by the current detection unit 4b, that is, the pulsation frequency included in the current detected by the current detection unit 4b is measured by the pulsation frequency measurement unit 71. The cosine wave generator 72 generates a cosine wave signal that oscillates at a pulsating frequency, and the sine wave generator 73 generates a sine wave signal that oscillates at a pulsating frequency.

The Fourier chord coefficient calculation unit 74 uses the chord wave signal from the chord wave generator 72 to perform Fourier series expansion of the input signal which is the current detected by the current detection unit 4b, and the specific frequency included in the input signal. Convert the components to DC and calculate the Fourier cosine coefficient. The Fourier cosine coefficient is a coefficient obtained by expanding an even function having an arbitrary period into a series of cos. The Fourier sine coefficient calculation unit 75 calculates the Fourier sine coefficient by expanding the Fourier series of the input signal using the sine wave signal from the sine wave generator 73 and converting the specific frequency component of the input signal into DC. .. The Fourier sine coefficient is the coefficient when an odd function with an arbitrary period is expanded into a series of sin.

The AC restorer 76 includes a cosine wave signal from the cosine wave generator 72, a sine wave signal from the sine wave generator 73, a Fourier cosine coefficient obtained by Fourier series expansion, and Fourier obtained by Fourier series expansion. Restore the AC using the sinusoidal coefficient. According to the pulsating component extraction unit 7C shown in FIG. 4, the characteristics of the bandpass filter can be obtained by the Fourier series expansion and the inverse transform.

If the above equation (3) is discretized, the calculation accuracy tends to decrease, but the decrease in the calculation accuracy can be suppressed by using the Fourier series expansion. Therefore, the method of extracting the pulsating component near the electric spring resonance angular frequency included in the torque current of the sub-side synchronous motor 1b by using the Fourier series expansion is excellent in terms of mounting. From this, the method of extracting the pulsating component using the Fourier series expansion is considered to be useful when changing the peak frequency of the bandpass filter, but when the calculation accuracy can be ensured, the pulsating frequency shown in FIG. 4 is considered. A bandpass filter that combines the measuring unit 71 and the above equation (3) may be used as the pulsating component extraction unit 7 shown in FIG.

As described above, if the pulsating component extraction unit 7 can extract the pulsating component near the electric spring resonance angular frequency included in the torque current of the sub-side synchronous motor 1b, it may be configured by any of the filters shown in FIGS. 2 to 4. Good. Since the high-pass filter is easier to design and mount on the circuit than the band-pass filter, it is preferable to select the high-pass filter when the ease of design and mounting on the circuit is important. If you want to obtain sharp cutoff characteristics, you should select a bandpass filter.

In the technique disclosed in Patent Document 1, a speed difference, which is a difference in the rotation speeds of the rotors of the main side synchronous motor and the sub side synchronous motor, is obtained, and the speed difference is stabilized by using this speed difference. Compensation is being made. Since the main side synchronous motor is stably controlled by this, it can be said that the technique disclosed in Patent Document 1 obtains the velocity pulsation component of the sub side synchronous motor and performs stabilization compensation. The pulsating component extraction method other than the technique disclosed in Patent Document 1 uses the pulsating component extracted in the torque current of the sub-side synchronous motor 1b as in the first embodiment and the active power of the sub-side synchronous motor 1b. A method of extracting the contained pulsating component can be considered. As a result of the investigation by the inventor of the present application, when the magnetic pole detection accuracy by the magnetic pole position detection units 5a and 5b is high, the pulsating component extraction method disclosed in Patent Document 1 is effective, and the magnetic poles by the magnetic pole position detection units 5a and 5b are effective. When the position detection accuracy is low, it has been found that the method of extracting the pulsating component contained in the torque current or the active power is effective.

For example, in a fluid utilization device such as a fan or a blower having a large moment of inertia of a mechanical system, the speed pulsation appearing in a speed signal may be very small even if the torque pulsation is large. In this case, the torque pulsation has a higher S / N ratio (Signal to Noise Radio) than the velocity pulsation, and the magnetic pole position can be easily detected. Therefore, it is rational to use the pulsating component of the torque current as in the first embodiment instead of the speed difference in detecting the magnetic pole position. Further, when the moment of inertia of the mechanical system is large but the speed pulsation is small, the pulsation of active power can be treated equivalent to the pulsation of torque. Therefore, the pulsation component extraction unit 7 may be configured to detect the pulsation of the active power.

Further, in the first embodiment, since the main side synchronous motor 1a is vector-controlled, the main side synchronous motor 1a should be driven following the torque current command value. However, in reality, weak vibration may occur in the main side synchronous motor 1a due to various disturbance factors. Disturbance factors include the short-circuit prevention time of the semiconductor elements of the upper and lower arms in series constituting the power converter 2, the offset of the current sensor, the gain imbalance of the current sensor, and the distortion of the magnetic flux generated from the magnet provided in the rotor. Conceivable. The drive device 100 according to the first embodiment has the main side by subtracting the pulsating component of the main side synchronous motor 1a from the pulsating component of the sub side synchronous motor 1b in consideration of the case where such a weak vibration occurs. The configuration may be such that the influence of the pulsating component of the synchronous motor 1a is removed.

Further, when a sensor such as an acceleration sensor or a torque sensor can be attached to the sub-side synchronous motor 1b in addition to the magnetic pole position detecting units 5a and 5b, the pulsating component extraction unit 7 outputs the sensor such as the acceleration sensor and the torque sensor. The pulsating component of the sub-side synchronous motor 1b may be extracted from the value.

The subtractor 8 shown in FIG. 1 obtains a first angle difference which is a difference in the magnetic pole positions of the rotors of the main side synchronous motor 1a and the sub side synchronous motor 1b. The magnetic pole position is equal to the rotation position of each rotor of the main side synchronous motor 1a and the sub side synchronous motor 1b, or equal to the rotation angle of each rotor of the main side synchronous motor 1a and the sub side synchronous motor 1b. Hereinafter, the first angle difference may be simply referred to as "angle difference". In order to explain the reason for obtaining the angle difference, the voltage equation and the torque equation in the steady state of the permanent magnet synchronous motor are shown below.

The voltage equation is expressed as the following equation (4). The torque equation is expressed as the following equation (5). The first term on the right side of the following equation (5) represents the magnet torque, and the second term represents the reluctance torque. The magnet torque is proportional to the q-axis current, and the reluctance torque is proportional to the product of the d-axis current and the q-axis current.

In the above equations (4) and (5), _Ra is the armature resistance, L _d is the d-axis inductance, L _q is the q-axis inductance, P _m is the number of pole pairs, and Φ _a is the number of armature interlinkage magnetic fluxes, ω. _e is the angular velocity, _id is the d-axis current, i _q is the q-axis current, v _d is the d-axis voltage, v _q is the q-axis voltage, and t is the generated torque. _{The subscript "x} " of each of these coefficients is for identifying whether the synchronous motor is on the main side or the sub side. For example, if it is not necessary to distinguish between the main side and the sub side, the subscript " _x " is added or the subscript " _x " is omitted. If the subscript has " _{m " instead of "x} ", it represents the main side, and if the subscript _{has "s} " instead of " _x ", it represents the sub side.

Next, the behavior of the torque change of the sub-side synchronous motor 1b due to the magnetic flux current compensation will be described with reference to FIGS. 5 to 11. FIG. 5 is a first diagram for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 6 is a second diagram for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 7 is a third diagram for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 8 is a fourth diagram for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 9 is a fifth diagram for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 10 is a sixth diagram for explaining the behavior of the torque change of the sub-side synchronous motor shown in FIG. FIG. 11 is a diagram showing the d-axis current on the main side shown in FIGS. 6, 7, 9 and 10 in association with the sign of the angle difference and the torque state of the sub-side synchronous motor.

5 to 10 show the behavior of the torque change of the sub-side synchronous motor 1b due to the magnetic flux current compensation, and the contents of FIGS. 5 to 10 are disclosed in Patent Document 1. In addition, in FIGS. 5 to 10, it is assumed that the motor constants of the main side synchronous motor 1a and the sub side synchronous motor 1b are equal.

First, a case will be described in which the d-axis of the main-side synchronous motor 1a has a lagging phase from the d-axis of the sub-side synchronous motor 1b with reference to FIG. In FIG. 5, when the magnetic flux current i _dm of the main side synchronous motor 1a is zero and the torque current i _{qm of the} main side synchronous motor 1a is flowing in the positive direction, the voltage command vector v ^→ _dq ^* is the direction of the second quadrant. Occurs in. When different load torques are generated in the two synchronous motors, if the motor constants of the two synchronous motors are the same, the phases of the heavy load synchronous motors are delayed. Therefore, in the case of FIG. 5, it can be said that the load of the main side synchronous motor 1a is larger than the load of the sub side synchronous motor 1b. That is, the main side synchronous motor 1a has a heavier load. The drive device 100 applies the same voltage to the two synchronous motors, but when the main side synchronous motor 1a has a heavier load, the magnetic flux current of the sub side synchronous motor 1b flows in the positive direction. This is clear by solving the above equation (4).

Here, consider the case where a positive magnetic flux current flows through the main-side synchronous motor 1a as shown in FIG. In this case, q-axis voltage of the main synchronous motor 1a is by increasing in the positive direction, the voltage command vector v ^→ changes from _dq ^* into v ^→ _dq ^**. As the q-axis voltage of the main-side synchronous motor 1a changes in this way, the d-axis voltage of the sub-side synchronous motor 1b decreases, and the q-axis voltage of the sub-side synchronous motor 1b increases. When the d-axis voltage of the sub-side synchronous motor 1b decreases, _{ω es} L _qs i _qs, which is the armature reaction of the sub-side synchronous motor 1b on the q-axis, decreases. Therefore, the q-axis current of the sub-side synchronous motor 1b is reduced. Further, as the q-axis voltage of the sub-side synchronous motor 1b increases, the d-axis current of the sub-side synchronous motor 1b increases. _{By changing the d-axis current which is the magnetic flux current idm} of the main side synchronous motor 1a in this way, the current of the sub side synchronous motor 1b changes. Due to the change in the current of the sub-side synchronous motor 1b, the torque of the sub-side synchronous motor 1b changes as compared with the torque of the sub-side synchronous motor 1b shown in FIG. Here, for the sake of simplicity, it is assumed that the synchronous motor is a surface magnet type synchronous AC motor and has no reluctance torque. In this case, the torque of the sub-side synchronous motor 1b when the current of the sub-side synchronous motor 1b changes is smaller than that in the state of FIG.

Contrary to the case of FIG. 6, FIG. 7 shows the torque state of the sub-side synchronous motor 1b when a negative magnetic flux current flows through the main-side synchronous motor 1a. In this case, by the q-axis voltage of the main synchronous motor 1a is decreased, the voltage command vector v ^→ changes from _dq ^* into v ^→ _dq ^**. As a result, the d-axis voltage of the sub-side synchronous motor 1b increases, and the q-axis voltage of the sub-side synchronous motor 1b decreases. As the d-axis voltage of the sub-side synchronous motor 1b increases, the q-axis current of the sub-side synchronous motor 1b increases, and as the q-axis voltage of the sub-side synchronous motor 1b decreases, the d of the sub-side synchronous motor 1b The shaft current decreases. In this case, the torque of the sub-side synchronous motor 1b increases as compared with the state of FIG.

Next, a case will be described in which the d-axis of the main-side synchronous motor 1a is in a leading phase with respect to the d-axis of the sub-side synchronous motor 1b with reference to FIG. In FIG. 8, the d-axis current of the main side synchronous motor 1a is zero, and the load of the main side synchronous motor 1a is larger than the load of the sub side synchronous motor 1b, that is, the main side synchronous motor 1a has a heavier load. It has become. Since the same voltage is applied to the main side synchronous motor 1a and the sub side synchronous motor 1b, the d-axis current of the sub side synchronous motor 1b flows in the negative direction when the sub side synchronous motor 1b has a heavier load. ..

Here, consider the case where a positive d-axis current flows through the main-side synchronous motor 1a as shown in FIG. In this case, by the q-axis voltage of the main synchronous motor 1a is increased, the voltage command vector v ^→ changes from _dq ^* into v ^→ _dq ^**. As the q-axis voltage of the main-side synchronous motor 1a changes in this way, the d-axis voltage of the sub-side synchronous motor 1b increases, and the q-axis voltage of the sub-side synchronous motor 1b also increases. As the d-axis voltage of the sub-side synchronous motor 1b increases, the q-axis current of the sub-side synchronous motor 1b increases. Further, as the q-axis voltage of the sub-side synchronous motor 1b increases, the d-axis current of the sub-side synchronous motor 1b decreases. In this case, the torque of the sub-side synchronous motor 1b increases as compared with the state of FIG.

FIG. 10 shows the torque state of the sub-side synchronous motor 1b when a negative magnetic flux current flows through the main-side synchronous motor 1a, contrary to the case of FIG. 9. In this case, the q-axis current of the sub-side synchronous motor 1b decreases. Therefore, the torque of the sub-side synchronous motor 1b is reduced as compared with the state of FIG.

FIG. 11 shows the d-axis current on the main side shown in FIGS. 6, 7, 9 and 10 in association with the sign of the angle difference and the torque state of the sub-side synchronous motor 1b. When the angle difference λ of the two synchronous motors is determined as in equation (6) below, if the angle difference λ is positive when the d-axis current of the main side synchronous motor 1a is increased, the sub side synchronous motor The torque of 1b decreases, and if the angle difference λ is negative, the torque of the sub-side synchronous motor 1b increases. _{However, θ es} in the following equation (6) represents the magnetic pole position of the sub-side synchronous motor 1b by the electric angle, and θ _em represents the magnetic pole position of the main side synchronous motor 1a by the electric angle. is there. On the other hand, when the d-axis current of the main side synchronous motor 1a is reduced, the torque of the sub side synchronous motor 1b increases if the angle difference λ is positive, and the torque of the sub side synchronous motor 1b increases if the angle difference λ is negative. Torque is reduced. That is, when the torque of the sub-side synchronous motor 1b is changed by changing the d-axis current of the main-side synchronous motor 1a to stabilize the drive of the sub-side synchronous motor 1b, the angle difference λ is positive. It is necessary to determine the compensation direction of the d-axis current depending on whether it is negative or negative. For this reason, the drive device 100 uses the subtractor 8 to obtain the angle difference λ.

However, since the above description assumes that the motor constant of the main side synchronous motor 1a is equal to the motor constant of the sub side synchronous motor 1b, the motor constant of the main side synchronous motor 1a is the same as the motor constant of the sub side synchronous motor 1b. If they are different, the intended torque change may not be obtained. The problem that the intended torque change cannot be obtained has been discovered by the study of the inventor of the present application. The reason why the intended torque change cannot be obtained will be described later, and the function of the magnetic flux current command determination unit 9 shown in FIG. 1 will be described first.

The magnetic flux current command determination unit 9 determines the magnetic flux current command for stabilizing the drive of the sub-side synchronous motor 1b. As already described, the torque of the sub-side synchronous motor 1b can be changed by changing the magnetic flux current.

FIG. 12 is a diagram showing a configuration example of the magnetic flux current command determination unit shown in FIG. FIG. 13 is a first diagram for explaining a code determination process by the code determination device shown in FIG. FIG. 14 is a second diagram for explaining the code determination process by the code determination device shown in FIG. FIG. 15 is a third diagram for explaining a code determination process by the code determination device shown in FIG.

The magnetic flux current command determination unit 9 shown in FIG. 12 includes a pulsation suppression control unit 91 and a compensation direction determination unit 92. The magnetic flux current command determination unit 9 receives the pulsating component of the torque current of the sub-side synchronous motor 1b as an input, and determines the magnetic flux current command using the pulsation suppression control unit 91 and the compensation direction determination unit 92. The pulsation suppression control unit 91 is composed of a gain multiplication unit 911 and a phase adjustment unit 912. In the technique of Patent Document 1, a speed difference is used for the input of the magnetic flux current command determination unit 9, but the pulsating component of the torque current is used in the magnetic flux current command determination unit 9 of the first embodiment. There is. The input to the magnetic flux current command determination unit 9 is not limited to the pulsating component of the torque current, but may be the pulsating component included in the active power.

The gain multiplication unit 911 adjusts the gain of the pulsating component which is an input signal. The phase adjusting unit 912 adjusts the phase of the pulsating component which is an input signal, and outputs the pulsating component whose amplitude is adjusted. If the stability of the system can be ensured by only one of the gain multiplication unit 911 and the phase adjustment unit 912, the pulsation suppression control unit 91 must necessarily include both the gain multiplication unit 911 and the phase adjustment unit 912. There is no.

The gain multiplying unit 911 multiplies a pulsating component which is an input signal by a specific gain and outputs the gain, and has a function of adjusting the stability and responsiveness of the system. The gain may be changed according to the operating conditions. The operating conditions are equal to the angular velocity commands of the main side synchronous motor 1a and the sub side synchronous motor 1b. For example, the gain in the gain multiplication unit 911 may be increased in the low speed region, and the gain in the gain multiplication unit 911 may be decreased in the high speed region. The reason for lowering the gain in the gain multiplication unit 911 in the high speed range is that in high speed operation, the rotation of the synchronous motor is stable even if the angle difference is the same as in low speed operation or the gain is lower than in low speed operation, and high speed operation is performed. This is because if the gain is increased as in the case of low-speed operation, compensation will be excessive and the rotation of the synchronous motor will become unstable. The phase adjustment unit 912 is composed of, for example, a phase delay compensator, a low-pass filter, an integration controller, and the like. The phase lag compensator is generally used in the industrial world because it stabilizes the gain by lowering the gain by a certain value in the high frequency range. Since the low-pass filter and the integral controller also have the property of changing the signal phase in the high frequency region, the low-pass filter or the integral controller can be used in the same manner as the phase delay compensator.

When an approximate integrator with a first-order low-pass filter is used as the phase adjustment unit 912, the cutoff angular frequency thereof may be set to 1/3 or less of the electric spring resonance angular frequency. If possible, use a value from 1/10 to 1/20 of the electric spring resonance angle frequency. With this setting, the phase can be delayed by about 90 degrees near the resonance angle frequency of the electric spring, and the control stability is improved.

Although not shown in FIG. 12, a dead zone may be provided in any of the input / output of the pulsation suppression control unit 91. This dead zone is useful for removing frequency components other than the electric spring resonance that could not be completely removed by the pulsating component extraction unit 7 described above.

The compensation direction determination unit 92 is composed of a code determining device 921 and a multiplier 922, and determines the compensation direction of the magnetic flux current command from the angle difference according to the operating principle described with reference to FIGS. 5 to 11. The code determination device 921 performs the code determination process shown in FIGS. 13 to 15. The horizontal axis of FIGS. 13 to 15 represents the angle difference which is the input of the sign determination device 921. The angle difference shows a positive or negative value as shown in FIG. The vertical axis of FIGS. 13 to 15 shows the output value of the sign determining device 921.

The most basic code determination process is the method shown in FIG. The sign determination device 921 outputs "1" when the angle difference indicates positive, and outputs "-1" when the angle difference indicates negative. However, in the method of FIG. 13, chattering may occur when the angle difference is close to zero. Therefore, in a region where the angle difference is close to zero as shown in FIG. 14, the output of the code determination device 921 is gradually switched from "1" to "-1", or the output of the code determination device 921 is changed from "-1" to "-1". It may be configured to gradually switch to "1".

Further, under the condition where the angle difference is large, the influence of the change amount of the magnetic flux current of the main side synchronous motor 1a on the torque change amount of the sub side synchronous motor 1b is larger than that under the condition where the angle difference is small. Therefore, in a region where the angle difference is large as shown in FIG. 15, the output value of the code determining device 921 may be gradually lowered as the angle difference becomes large.

The multiplier 922 multiplies the output of the sign determination device 921 and the output of the pulsation suppression control unit 91 to generate a magnetic flux current command. That is, the magnetic flux current command determination unit 9 determines the magnetic flux current command based on the pulsation component suppressed by the pulsation suppression control unit 91 and the compensation direction determined by the sign determination device 921 of the compensation direction determination unit 92.

In Patent Document 1, the synchronous motor is controlled based on the compensation direction determined from the angle difference, and the compensation direction of the magnetic flux current is switched at the point where the angle difference becomes zero. However, when the motor constants of the two synchronous motors are different, the control method of Patent Document 1 causes an erroneous determination of the compensation direction. The reason is as follows.

In the technique of Patent Document 1, the d-axis current of the main side synchronous motor 1a is operated. In the following, it will be mathematically clarified how the operation of the d-axis current of the main side synchronous motor 1a affects the sub side synchronous motor 1b. Here, not only the case where the motor constants of the two synchronous motors are the same, but also the case where the motor constants of the two synchronous motors are different will be described.

In the voltage equation in the steady state of the main-side synchronous motor 1a, when a minute perturbation is applied to the d-axis current, the following equation (7) is obtained.

In the above equation (7), v _dm is the d-axis voltage _{of the main side synchronous motor 1a, and v qm} is the q-axis voltage of the main side synchronous motor 1a. Δv _dm is a minute perturbation of the d-axis voltage of the main-side synchronous motor 1a caused by a minute perturbation of the d-axis current of the main-side synchronous motor 1a. Δv _qm is a minute perturbation of the q-axis voltage of the main side synchronous motor 1a caused by a minute perturbation of the d-axis current of the main side synchronous motor 1a. _Ram is the armature resistance of the main synchronous motor 1a, ω _em is the angular speed of the main synchronous motor 1a, L _dm is the inductance of the d-axis of the _{main synchronous motor 1a, and L qm} is the q-axis of the main synchronous motor 1a. It is an inductance. i _dm is the d-axis current of the main side synchronous motor 1a, i _qm is the q-axis current _{of the main side synchronous motor 1a, and Δi dm} is a minute perturbation of the d-axis current of the main side synchronous motor 1a. Φ _am is the number of armature interlinkage magnetic fluxes of the main side synchronous motor 1a.

By taking the difference between the above equation (7) and the above equation (4), the minute perturbation of the dq axis voltage of the main side synchronous motor 1a can be obtained as in the following equation (8).

When the angle difference is represented by λ described in the above equation (6), the dq-axis voltage perturbation of the main side synchronous motor 1a is coordinate-converted on the dq plane of the sub-side synchronous motor 1b as shown in the following equation (9). can do. In (9) below, v _ds is the d-axis voltage _{of the sub-side synchronous motor 1b, and v qs} is the q-axis voltage of the sub-side synchronous motor 1b.

By substituting the above equation (8) into the above equation (9) and rearranging, the following equation (10) can be obtained. In the following equation (10), Δv _ds is a minute perturbation of the d-axis voltage of the sub-side synchronous motor 1b caused by a minute perturbation of the d-axis current of the main side synchronous motor 1a. Δv _qs is a minute perturbation of the q-axis voltage of the sub-side synchronous motor 1b caused by a minute perturbation of the d-axis current of the main side synchronous motor 1a.

Next, the following equation (11) can be obtained by solving the voltage equation in the steady state of the sub-side synchronous motor 1b for the current. In the following equation (11), i _ds is the d-axis current _{of the sub-side synchronous motor 1b, and i qs} is the q-axis current of the sub-side synchronous motor 1b. R _as is the armature resistance of the sub-side synchronous motor 1b, ω _es is the angular velocity of the sub-side synchronous motor 1b, L _ds is the inductance of the d-axis of the _{sub-side synchronous motor 1b, and L qs} is the q-axis of the sub-side synchronous motor 1b. It is an inductance. Φ _as is the number of armature interlinkage magnetic fluxes of the sub-side synchronous motor 1b.

Here, the dq axis of the sub-side synchronous motor 1b is generated by adding the minute perturbation of the above equation (10) to the dq-axis voltage of the sub-side synchronous motor 1b shown in the last item on the right side of the above equation (11). When the change in current is added to the left side of the above equation (11) as _{Δi ds} and Δi _{qs, the following equation (12) is obtained.}

By subtracting the above equation (11) from the above equation (12), a minute perturbation component of the dq axis current of the sub-side synchronous motor 1b can be extracted. The minute perturbation of the dq-axis current of the sub-side synchronous motor 1b can be expressed by the following equation (13).

Substituting the above equation (10) into the above equation (13), the change in the dq-axis current of the sub-side synchronous motor 1b due to the d-axis current perturbation of the main side synchronous motor 1a is as shown in the following equation (14). Can be represented.

Since the coefficient of the above equation (14) is complicated, it is arranged by replacing it with a simple coefficient as shown in the following equation (15). In the following (15), A is obtained by A = R _{as Ram} + ω _em ω _es L _dm L _qs . B is calculated by _{B = ω em L dm R as} -ω es L qs R am. C is calculated by _{C = ω em L dm R as} -ω es L ds R am. D is obtained by D = ω _em ω _es L _dm L _ds + _Ram R _as . E is obtained by E = Δi _dm / (R _as ² + ω _es ² L _ds L _qs ).

On the other hand, the torque t _s of the sub-side synchronous motor 1b is expressed by the following equation (16).

In the above equation (16), the following equation (17) is obtained by adding a minute perturbation to the dq-axis current of the sub-side synchronous motor 1b. _{The subscript "0} " in Eq. (17) below represents the value of the operating point.

By extracting the portion related to the dq-axis current perturbation of the sub-side synchronous motor 1b from the above equation (17), the following equation (18) is obtained. The first term on the right side of the following equation (18) is the magnet torque perturbation of the sub-side motor, and the second term is the reluctance torque perturbation.

Here, in order to make the explanation easy to understand, it is first assumed that the reluctance torque perturbation is very small and can be ignored. At this time, the relationship between the magnet torque perturbation and the angle difference λ can be expressed by the following equation (19). In the following equation (19), C is obtained by _{C = ω em L dm R as} -ω es L ds R am. D is obtained by D = ω _em ω _es L _dm L _ds + _Ram R _as . E is obtained by E = Δi _dm / (R _as ² + ω _es ² L _ds L _qs ).

By ignoring the reluctance torque perturbation, it is possible to calculate when the reluctance torque perturbation is always zero when the angle difference λ is. By substituting zero for the left side of the above equation (19) and rearranging the equation for the angle difference λ, the following equation (20) can be obtained. In the following equation (20), C is obtained by _{C = ω em L dm R as} -ω es L ds R am. D is obtained by D = ω _em ω _es L _dm L _ds + _Ram R _as .

The above equation (20) includes the d-axis inductance of each of the two synchronous motors, the armature resistance of each of the two synchronous motors, and the angular velocity of each of the two synchronous motors.

If the parallel drive is performed properly to some extent, the angular velocities of the two synchronous motors are considered to be equal to each other. Therefore, here, assuming that there is no difference in the angular velocities of the two synchronous motors, 2 Consider the difference in the motor constants of each of the synchronous motors, that is, the difference in the d-axis inductance and the difference in the armature resistance. When the motor constants of the two synchronous motors are equal to each other, the numerator of the fraction on the right side of the above equation (20) becomes zero, so the d of the main synchronous motor 1a is defined by the point where the angle difference becomes zero. The increase / decrease relationship between the shaft current and the torque of the sub-side synchronous motor 1b changes. For this reason, in the control method disclosed in Patent Document 1, a process of switching the compensation direction of the magnetic flux current is performed at the point where the angle difference becomes zero.

When the motor constants of the two synchronous motors are different from each other, the numerator of the fraction on the right side of the above equation (20) does not become zero. In this case, the point at which the increase / decrease relationship between the d-axis current of the main side synchronous motor 1a and the torque of the sub side synchronous motor 1b changes is not the point where the angle difference becomes zero. Therefore, the control method disclosed in Patent Document 1 has a drawback that when the motor constants of the two synchronous motors are different from each other and the angle difference becomes close to zero, the compensation direction of the magnetic flux current is erroneously determined. ..

If the compensation direction of the magnetic flux current is erroneously determined, an unintended torque change occurs in the sub-side synchronous motor 1b, so that the sub-side synchronous motor 1b may continuously vibrate. The process of generating continuous vibration of the sub-side synchronous motor 1b is as follows. First, the compensation direction of the magnetic flux current is erroneously determined, so that an unintended torque change occurs in the sub-side synchronous motor 1b. This increases the angle difference. When the angle difference exceeds a certain value, the judgment of the compensation direction returns to the normal state. This reduces the angle difference. However, when the angle difference decreases and becomes close to zero, the compensation direction is erroneously determined again. By repeating such a vicious cycle, continuous vibration is generated in the sub-side synchronous motor 1b.

From this, it can be said that when the motor constants of the two synchronous motors are different from each other, it is necessary to determine the compensation direction of the magnetic flux current in consideration of the difference in the motor constants. The angle difference correction unit 10 according to the first embodiment is for correcting the compensation direction of the magnetic flux current. For example, when the motor constants of the two synchronous motors are known, the angle difference signal may be modified as in the following equation (21). In the following equation (21), " ^~ " means "after correction". θ _es represents the magnetic pole position of the sub-side synchronous motor 1b by an electric angle, and θ _em represents the magnetic pole position of the main side synchronous motor 1a by an electric angle. C is calculated by _{C = ω em L dm R as} -ω es L ds R am. D is obtained by D = ω _em ω _es L _dm L _ds + _Ram R _as . L _dm , L _ds , R _as and _Ram correspond to the motor constants.

In the above equation (21), the second term including the arc tangent represents the angle difference correction amount which is the correction amount of the first angle difference. The angle difference correction amount may be calculated in the angle difference correction unit 10 each time the rotation speed of the rotor changes by using the equation of the second term, but the value calculated in advance is used for the angle difference correction. It is stored as a data table in the unit 10, and the angle difference correction unit 10 reads out the angle difference correction amount corresponding to the rotation speed of the rotor by referring to the data table, and reads the read angle difference correction amount. , May be used instead of the angle difference correction amount obtained by the formula of the second term above. A specific example of a method of calculating the angle difference correction amount using the data table will be described with reference to FIG.

FIG. 16 is a diagram showing the relationship between the angle difference correction amount calculated by the angle difference correction unit shown in FIG. 1 and the rotation speed of the rotor. In FIG. 16, the vertical axis shows the angle difference correction amount, and the horizontal axis shows the rotation speed of the rotor. In FIG. 16, the rotation speed of the rotor is referred to as "speed". According to FIG. 16, it can be seen that the angle difference correction amount changes according to the rotation speed. The angle difference correction amount corresponding to such a rotation speed is calculated in advance, and the calculated angle difference correction amount is stored in the above-mentioned data table. A simple approximation formula is derived from such a transition of the angle difference correction amount, and the angle difference correction unit 10 uses the derived approximation instead of the second term including the arc tangent of the above formula (21). The angle difference correction amount may be calculated using the formula.

Next, a configuration example of the angle difference correction unit 10 shown in FIG. 1 will be described with reference to FIGS. 17 and 18. FIG. 17 is a diagram showing a first configuration example of the angle difference correction unit shown in FIG. FIG. 17 shows a configuration example of the angle difference correction unit 10A for obtaining the second angle difference, which is the angle difference after correction, using the above equation (21). Hereinafter, the second angular velocity will be referred to as "corrected angular velocity". The angle difference correction unit 10A includes a correction amount calculation unit 10a and a subtractor 10b. The correction amount calculation unit 10a performs the calculation of the second term including the arc tangent of the above equation (21) by using the motor constant and the speed command. If the rotation speeds of the two synchronous motors can be accurately detected, the calculation of the second term including the arc tangent of the above equation (21) may be performed using the rotation speed instead of the speed command. The subtractor 10b calculates the corrected angle difference by subtracting the angle difference correction amount calculated by the correction amount calculation unit 10a from the angle difference according to the above equation (21).

FIG. 18 is a diagram showing a second configuration example of the angle difference correction unit shown in FIG. FIG. 18 shows a configuration example of the angle difference correction unit 10B for obtaining the angle difference after correction using the data table 10c. The angle difference correction unit 10B includes a correction amount calculation unit 10d and a data table 10c instead of the correction amount calculation unit 10a shown in FIG. It is assumed that the data table 10c of FIG. 18 stores a plurality of angle difference correction amounts that change according to the motor constant and the rotation speed. The angle difference correction unit 10B reads out the angle difference correction amount corresponding to the motor constant and the speed command by referring to the data table 10c, and outputs the read angle difference correction amount. In addition, in order to reduce the amount of data stored in the data table 10c, the data table 10c may have an interpolation processing means (not shown).

In the first embodiment, it is assumed that the synchronous motor is a surface magnet type synchronous AC motor, and a configuration example for correcting the angle difference is described assuming that there is no reluctance torque. However, even if there is reluctance torque, the angle is described. Assuming that the difference is close to zero, it is relatively easy to derive the condition that the magnet torque perturbation on the sub-side synchronous motor 1b side due to the d-axis current perturbation on the main side synchronous motor 1a side becomes zero. Can be done. Since the first embodiment targets a phenomenon in which the compensation direction of the magnetic flux current is erroneously determined when the angle difference is close to zero, the above approximation is valid. Equation (22) below represents a condition in which the sum of the magnet torque perturbation on the sub-side synchronous motor 1b side due to the d-axis current perturbation on the main side synchronous motor 1a side and the reluctance perturbation is zero. In the following (22), A is obtained by A = R _{as Ram} + ω _em ω _es L _dm L _qs . B is calculated by _{B = ω em L dm R as} -ω es L qs R am. C is calculated by _{C = ω em L dm R as} -ω es L ds R am. D is obtained by D = ω _em ω _es L _dm L _ds + _Ram R _as . E is obtained by E = Δi _dm / (R _as ² + ω _es ² L _ds L _qs ). F is .G sought _{F = P m (Φ as +} (L ds -L qs) i ds0 _is calculated by _{G = P m (L ds -L} qs) i qs0 .H is H = P _{It is calculated by} m (L _ds −L qs).

In the case of a synchronous motor without reluctance torque, the perturbation term of the d-axis current on the main side synchronous motor 1a side is erased in the process of formula transformation, whereas in the case of a synchronous motor with reluctance torque, the perturbation term is not erased. However, when determining the angle difference correction amount, the calculation becomes too complicated if the perturbation term of the d-axis current on the main side synchronous motor 1a side is taken into consideration. Therefore, in practice, it is better to eliminate the perturbation term of the d-axis current on the main side synchronous motor 1a side, assuming that it is minute. As a result, the theoretical formula of the angle difference correction value in the synchronous motor having a large reluctance torque is expressed as the following formula (23). By calculating the following equation (23), even a synchronous motor having a large reluctance torque can perform stabilization compensation similar to that of a surface magnet type synchronous motor. However, there are many parameters required to calculate the following equation (23), and the calculation is complicated. Countermeasures for such problems will be described in the second and subsequent embodiments.

When two synchronous motors with different motor constants are driven in parallel and the sub-side synchronous motor 1b is stabilized by changing the magnetic flux current, the compensation direction of the magnetic flux current should be adjusted in consideration of the difference in the motor constants. If it is not determined, a misjudgment of the compensation direction may occur. In the method described in Patent Document 1, even when two synchronous motors having the same specifications are driven in parallel, if the motor constants of the two synchronous motors are different, the compensation direction may be erroneously determined. is there. Since this erroneous determination causes continuous vibration in the sub-side synchronous motor 1b, a large speed unevenness occurs in the sub-side synchronous motor 1b. Along with this, there is a concern that the vibration and noise of the sub-side synchronous motor 1b will increase and the motor efficiency will decrease.

Since the drive device 100 according to the first embodiment is configured to determine the compensation direction of the magnetic flux current in consideration of the difference in the motor constants, two synchronous motors are used by using one power converter. Even when the motor is driven, the robustness against fluctuations in the motor constant is improved, and it is possible to prevent the magnetic flux current command determining unit 9 from erroneously determining the compensation direction of the magnetic flux current. Therefore, the occurrence of speed unevenness due to erroneous determination of the compensation direction of the magnetic flux current is suppressed. Further, since the occurrence of speed unevenness is suppressed, the increase in vibration and noise of the sub-side synchronous motor 1b can be suppressed, and the decrease in motor efficiency can be further suppressed.

Embodiment 2.
In the first embodiment, the motor constants of the two synchronous motors are required in order to correct the angle difference. However, these motor constants change due to various factors. In the second embodiment, the drive device 100 having a motor constant measurement function for correcting the angle difference will be described. FIG. 19 is a configuration diagram of a current control unit included in the drive device according to the second embodiment of the present invention. FIG. 20 is a configuration diagram of an angle difference correction unit included in the drive device according to the second embodiment of the present invention. The drive device 100 of the second embodiment includes a current control unit 6A and an angle difference correction unit 10C instead of the current control unit 6 and the angle difference correction unit 10 of the first embodiment. Other configurations are the same as or equivalent to the configuration of the first embodiment, and the same or equivalent components are designated by the same reference numerals, and redundant description will be omitted.

The current control unit 6A shown in FIG. 19 is a function for controlling the torque current and the magnetic flux current, respectively. The current control unit 6A includes an adder 61a, an adder 61b, an adder 61c, a subtractor 61d, an adder 61e, an adder 61f, an adder 61g, an adder 61h, and proportional integration differentiation ( It includes a Proportional Adder Daily (PID) controller 62a and a PID controller 62b. Further, the current control unit 6A includes a non-interference controller 63 and a test signal generation unit 64 that generates a test signal for measuring the motor constant. In addition to these functions, the current control unit 6A reverses the phase current of the main synchronous motor 1a to the current of the desired coordinate system and the voltage command to the three-phase coordinate system. It includes a second coordinate converter for conversion. In FIG. 19, the illustration of the first coordinate converter and the second coordinate converter is omitted. As the current control unit 6A of the second embodiment, it is preferable to use a vector controller on the dq rotor coordinates as in the current control unit 6 of the first embodiment, but dq in the vector control In addition to the coordinate system, a polar coordinate system such as an αβ stator coordinate system or a γδ coordinate system may be used. Further, DTC may be adopted for the current control unit 6A instead of the vector control.

In order to measure the motor constant, the test signal generator 64 sends a test signal of DC voltage, DC current, AC voltage or AC current to the main side synchronous motor 1a when the main side synchronous motor 1a and the sub side synchronous motor 1b are stopped. And the sub-side synchronous motor 1b. The test signal is a signal for changing the torque current command, the magnetic flux current command, the q-axis voltage command, and the d-axis voltage command. The details of the test signal generated by the test signal generation unit 64 will be described later.

The adder 61a adds the torque current command and the test signal for changing the torque current command output from the test signal generation unit 64. The subtractor 61c calculates the torque current deviation by subtracting the torque current from the torque current command and test signal added by the adder 61a. The PID controller 62a determines the control amount by performing proportional calculation, integral calculation and differential calculation with respect to the torque current deviation.

The adder 61b adds the magnetic flux current command and the test signal for changing the magnetic flux current command output from the test signal generation unit 64. The subtractor 61d calculates the magnetic flux current deviation by subtracting the magnetic flux current from the magnetic flux current command and the test signal added by the adder 61b. The PID controller 62b determines the control amount by performing proportional calculation, integration calculation and differential calculation with respect to the magnetic flux current deviation.

The non-interference controller 63 uses feed-forward control using the torque current command and test signal added by the adder 61a, the speed command, and the magnetic flux current command and test signal added by the adder 61b. Generate a d-axis voltage and a q-axis voltage to eliminate control interference between the dq axes. Specifically, the decoupling controller 63 cancels the voltage that interferes with the q-axis voltage generated by the d-axis current by using the speed command, the magnetic flux current command added by the adder 61b, and the test signal. The voltage for this is generated and output to the adder 61e. The non-interference controller 63 uses the torque current command and test signal added by the adder 61a and the speed command to generate a voltage for canceling a voltage that interferes with the d-axis voltage generated by the q-axis current. Then, it is output to the subtractor 61f.

The adder 61e adds the output of the non-interfering controller 63 to the output of the PID controller 62a and outputs the adder 61e. The adder 61g outputs a q-axis voltage command by adding a test signal for changing the q-axis voltage command output from the test signal generation unit 64 to the output of the adder 61e.

The subtractor 61f subtracts the output of the non-interfering controller 63 from the output of the PID controller 62b and outputs the subtractor 61f. The adder 61h outputs a d-axis voltage command by adding a test signal for changing the d-axis voltage command output from the test signal generation unit 64 to the output of the subtractor 61f.

The angle difference correction unit 10C shown in FIG. 20 includes a correction amount calculation unit 10d and a data table 10c shown in FIG. 18 in order to correct the angle difference using the analysis result of the voltage change or the current change by the current control unit 6A. Along with this, a test signal analysis unit 10e is provided. The main side phase current, which is the phase current on the main side synchronous motor 1a side, the sub side phase current, which is the phase current on the sub side synchronous motor 1b side, and the voltage command are input to the test signal analysis unit 10e. By analyzing these, the test signal analysis unit 10e measures and outputs the motor constants, and further analyzes the pulsating component of the sub-side torque current, which is the electrical behavior caused by the difference in the motor constants. That is, the test signal analysis unit 10e analyzes the voltage change or current change caused by the test signal. If the d-axis inductance and armature resistance of the main side synchronous motor 1a and the sub side synchronous motor 1b are known as a result of the measurement, the angle difference correction amount can be calculated by the correction amount calculation unit 10d. The angle difference correction unit 10C may calculate the angle difference correction amount by using the correction amount calculation unit 10a shown in FIG. 17 instead of the correction amount calculation unit 10d and the data table 10c.

Various measurement methods have been proposed for measuring the motor constant, but since it is not possible to exemplify all of these measurement methods, an example will be described in the second embodiment.

For example, the test signal generation unit 64 gives a test signal of AC voltage or AC current to the main side synchronous motor 1a and the sub side synchronous motor 1b when the main side synchronous motor 1a and the sub side synchronous motor 1b are stopped. Measure the phase current of each of the synchronous motors. Then, the test signal analysis unit 10e can measure the armature resistance of each of the two synchronous motors by measuring this phase current. However, in general, in the power converter 2, an error is included in the output voltage due to the influence of the short-circuit prevention time of the upper and lower arms, and therefore, if the compensation accuracy of the voltage error is low, a measurement error occurs in the armature resistance. In such a case, the test signal analysis unit 10e measures the ratio of the phase currents flowing through each of the two synchronous motors. Since the nominal value of the armature resistance is generally available, the test signal analysis unit 10e may compare the nominal value of the armature resistance with the measured phase current ratio value. In the method of measuring the phase current ratio, it is not known which motor constant of the main side synchronous motor 1a or the sub side synchronous motor 1b has changed, but in practice, there is no problem in performing the calculation of the above equation (21). Absent.

Further, the test signal generation unit 64 sends a test signal of AC voltage or AC current in the d-axis direction to the main side synchronous motor 1a and the sub side synchronous motor 1b when the main side synchronous motor 1a and the sub side synchronous motor 1b are stopped. Then, the test signal analysis unit 10e can measure the d-axis inductance from the value of the current flowing through each of the main side synchronous motor 1a and the sub side synchronous motor 1b. When the voltage error cannot be ignored, the test signal analysis unit 10e extracts the AC amplitude of the phase current flowing through each of the main side synchronous motor 1a and the sub side synchronous motor 1b, measures the ratio, and uses this ratio as described above ( 21) Equation may be calculated.

In this way, if the armature resistance and the d-axis inductance of the two synchronous motors are measured while the synchronous motor is stopped, the calculation of the above equation (21) can be easily performed. However, the armature resistance changes due to the temperature change of the synchronous motor during operation. Therefore, consider a method of measuring the change in armature resistance while the synchronous motor is operating. One way, in the first row R _am on the right side of equation (8), modified as follows (24), the d-axis current of the main synchronous motor 1a during steady operation, the direct current test A method of measuring the change in the d-axis voltage before and after superimposition by superimposing the signals can be considered. This measuring method requires high accuracy with respect to the output voltage of the power converter 2, but has an advantage that the armature resistance can be measured during operation. Although the above equation (8) is a mathematical expression relating to the main side synchronous motor 1a, the armature resistance can be calculated by the same mathematical expression for the sub side synchronous motor 1b. If a temperature sensor that measures the temperature change of the winding can be used, the temperature change generated by the synchronous motor may be measured by the temperature sensor and the amount of change in resistance from the nominal value may be calculated.

In order to perform the calculation of the above equation (21), it is sufficient that the armature resistance and the d-axis inductance of each of the two synchronous motors are known, but to perform the calculation of the above equation (23), the sub-side synchronous motor The armature interlinkage magnetic flux number Φ _{as of} 1b and the q-axis inductance L _qs are also required. The armature interlinkage magnetic flux number Φ _as changes due to factors such as temperature change and overcurrent demagnetization, and the q-axis inductance L _qs changes due to magnetic saturation when a large current flows. It is necessary to measure these constants in order to perform the calculation of the above equation (23) with high accuracy.

As a method for measuring the inductance, for example, the method of Japanese Patent No. 5634620 (hereinafter referred to as Reference 2) is known. In Reference 2, a plurality of constant DC voltage voltage commands are applied to the rotator, and the measurement voltage command arbitrarily selected from the applied voltages and the rotator current before and after the application of the measurement voltage command are obtained. A method for calculating the inductance has been devised using this method.

When the inductance calculation method disclosed in Reference 2 is applied to the second embodiment, the test signal generation unit 64 shown in FIG. 19 generates a plurality of constant DC voltage voltage commands as test signals, and analyzes the test signals. The unit 10e calculates the inductance by using the measurement voltage command arbitrarily selected from the voltage commands generated by the test signal generation unit 64 and the rotating machine current before and after the measurement voltage command is generated. Here, the test signal becomes an impulse-like voltage. Since the principle of calculating the inductance is described in detail in Reference 2, the description thereof is omitted here. By using this inductance measurement method, it is possible to measure the q-axis inductance even in the drive device 100 according to the second embodiment. It is also possible to measure the d-axis inductance by applying the inductance method disclosed in Reference 2.

As a method for measuring the number of armature interlinkage magnetic fluxes, for example, a magnetic flux observer is known. An example of utilization of the magnetic flux observer is disclosed in Japanese Patent Application Laid-Open No. 2003-302413 (hereinafter referred to as Reference 3). The magnetic flux observer estimates the stator magnetic flux and the rotor magnetic flux by using the voltage applied to the motor and the current flowing through the motor. In Reference 3, the speed of the synchronous motor is estimated by combining the magnetic flux observer and the adaptive control, and the rotor d-axis magnetic flux, that is, the number of armature interlinkage magnetic fluxes is estimated in the calculation process. There is. Velocity information is required for estimating the rotor d-axis magnetic flux, and the velocity information may be a velocity estimation value calculated by adaptive control or a velocity true value detected by sensors.

When the method for measuring the number of armature interlinkage magnetic fluxes disclosed in Reference 3 is applied to the second embodiment, a magnetic flux observer may be provided inside the test signal analysis unit 10e. In this case, the test signal generation unit 64 does not need to generate the test signal, so the test signal is set to zero. A method for measuring the number of armature interlinkage magnetic fluxes in this way is also known.

By the above method, in the second embodiment, the armature resistance, the d-axis inductance, the q-axis inductance, and the number of armature interlinkage magnetic fluxes can be measured. Then, if the motor constant is known, it is easy to calculate the above equations (21) and (23). According to the second embodiment, by providing the drive device 100 with a motor constant measurement function, the angle difference between the two synchronous motors can be accurately corrected, and an erroneous determination in the compensation direction of the magnetic flux current can be prevented.

Embodiment 3.
In the third embodiment, a configuration example in which the angle difference is corrected by examining the characteristics of the pulsating component of the sub-side synchronous motor 1b when the compensation direction of the magnetic flux current is erroneously determined will be described. Before explaining a specific configuration example of the third embodiment, what happens when the compensation direction of the magnetic flux current is erroneously determined will be described with reference to FIGS. 21 and 22.

FIG. 21 is a diagram for explaining a torque current waveform of the sub-side synchronous motor when an error of + 30% is added to the armature resistance of the main-side synchronous motor shown in FIG. FIG. 22 is a diagram for explaining a torque current waveform of the sub-side synchronous motor when an error of −30% is added to the armature resistance of the main-side synchronous motor shown in FIG. In FIGS. 21 and 22, it is assumed that a difference occurs in the motor constants of the two synchronous motors because only one of the two synchronous motors having the same motor constants is driven during parallel driving. ing. Only one synchronous motor is driven when the contact of the relay circuit 11 shown in FIG. 1 is opened.

The sub-side torque current, which is the torque current of the sub-side synchronous motor 1b, vibrates at the natural angular frequency described by the approximate expression of the above equation (1) due to the electric spring resonance described above. The upper view of FIG. 21 shows the original waveform of the sub-side torque current and the fundamental wave component of the natural angular frequency. The original waveform is shown by a solid line, and the fundamental wave component is shown by a broken line. The horizontal axis is time. The fundamental wave component is an intrinsic angular frequency component included in the original waveform of the sub-side torque current. In the upper diagram of FIG. 21, the original waveform and the fundamental wave component are shown together in order to clarify how much the original angular frequency component is included in the original waveform of the sub-side torque current.

The magnetic flux current command determining unit 9 described above appropriately changes the magnetic flux current in order to suppress the resonance of the electric spring, but in FIG. 21, the magnetic flux current command is given because the angle difference correction of the above equation (21) is not performed. It is assumed that the determination unit 9 erroneously determines the compensation direction of the magnetic flux current, and that continuous vibration is generated in the sub-side synchronous motor 1b. As described above, it can be seen that the waveform of the sub-side torque current when the sub-side synchronous motor 1b is continuously vibrated is distorted.

The lower view of FIG. 21 shows the pulsating component of the sub-side torque current caused by the erroneous determination. The pulsating component includes a waveform component obtained by subtracting the fundamental wave component from the original waveform and a second harmonic component of the electric spring resonance. The waveform component obtained by subtracting the fundamental wave component from the original waveform is shown by a solid line, and the second harmonic component of the electric spring resonance is shown by a broken line. The horizontal axis is time.

When the erroneous determination of the compensation direction occurs constantly, the erroneous determination occurs twice in one cycle of the electric spring resonance. Therefore, when the sub-side torque current is frequency-analyzed, the even-order harmonic components of the electric spring resonance tend to increase. In order to clarify that the even-order harmonic components of the electric spring resonance are increasing, the lower figure of FIG. 21 shows the waveform component obtained by subtracting the fundamental wave component from the original waveform and the second order of the electric spring resonance. Harmonic components are also listed.

The upper view of FIG. 22 shows the original waveform of the sub-side torque current and the fundamental wave component of the natural angular frequency. The original waveform is shown by a solid line, and the fundamental wave component is shown by a broken line. The horizontal axis is time. The lower view of FIG. 22 shows the pulsating component of the sub-side torque current caused by the erroneous determination. According to FIG. 22, as in FIG. 21, it can be seen that pulsation due to erroneous determination occurs, and that even-order harmonic components of the electric spring resonance are generated.

However, the waveform of the sub-side torque current shown in FIG. 22 is different from the waveform of the sub-side torque current of FIG. The reason why the waveforms are different is that the ideal value of the angle difference correction shown by the above equation (21) changes due to the change of the armature resistance from the state shown in FIG. 21, and the ideal value of the angle difference correction changes. This is because the timing of occurrence of the erroneous determination has changed, and the direction of the torque shock caused by the erroneous determination has also changed. In FIG. 21, the peak of the torque shock is in the negative direction, but in FIG. 22, the peak of the torque shock is in the positive direction. Further, the phases of the second harmonic components are also different in FIGS. 21 and 22. The calculation result of the ideal value of the angle difference correction represented by the above equation (20) is a negative value in the case of FIG. 21 and a positive value in the case of FIG. 22.

These results suggest that even-numbered vibrations of the electric spring resonance occur in the sub-side synchronous motor 1b when the compensation direction of the magnetic flux current is erroneously determined. It also suggests that the direction of torque shock peak generation and the phase of even-order harmonic components change depending on whether the optimum value for angle difference correction is positive or negative.

As a result of the investigation by the inventor of the present application, such a fact was found. Therefore, the inventor of the present application examined a method of analyzing the waveform of the torque current to detect an erroneous determination in the compensation direction. In addition, in FIGS. 21 and 22, the waveform of the torque current is shown as the most easily understood example, but instead of the torque current, the phenomenon as shown in FIGS. 21 and 22 is observed from the pulsation of the active power and the pulsation of the speed. You can also do it.

FIG. 23 is a configuration diagram of an angle difference correction unit included in the drive device according to the third embodiment of the present invention. The drive device 100 of the third embodiment includes an angle difference correction unit 10D instead of the angle difference correction unit 10 of the first embodiment. The angle difference correction unit 10D includes a pulsation component analysis unit 10f and an offset amount control unit 10g. Other configurations are the same as or equivalent to the configuration of the first embodiment, and the same or equivalent components are designated by the same reference numerals, and redundant description will be omitted.

The pulsation component analysis unit 10f has a function of frequency-analyzing the pulsation component included in the sub-side torque current flowing through the sub-side synchronous motor 1b. The sub-side torque current is the current value in the three-phase coordinate system detected by the current detection unit 4b, which is converted into the dq-axis current in the rotational Cartesian coordinates (dq-axis) by a coordinate converter (not shown). Indicates the q-axis current. Since the electric spring resonance angle frequency represented by the above equation (1) includes an approximation error, the pulsation component analysis unit 10f has a resonance angle frequency measurement unit 10h for measuring the resonance angle frequency included in the sub-side torque current. To be equipped with. It is desirable that the resonance angular frequency measuring unit 10h is provided in the pulsating component analysis unit 10f not only for measuring the resonance angular frequency but also in consideration of the fact that the motor constant changes when the synchronous motor is driven.

Further, the pulsation component analysis unit 10f includes a filter unit 10i that extracts a specific harmonic component by performing an analysis process on the pulsation component of the sub-side torque current. For the filter unit 10i, for example, a bandpass filter that extracts the second harmonic component of the electric spring resonance angular frequency from the pulsating component of the sub-side torque current can be used. Further, the filter unit 10i may be a combination of a high-pass filter that removes the DC component of the pulsating component of the sub-side torque current and a notch filter that removes the fundamental wave component of the electric spring resonance angular frequency.

The offset amount control unit 10g is a function of adding a correction amount to the angle difference so that the harmonic component of the pulsation component is reduced. The details of the offset amount control unit 10g will be described later. The subtractor 10b subtracts the output of the offset amount control unit 10g from the angle difference to determine the corrected angle difference.

As the simplest configuration example of the offset amount control unit 10g, a method using a search algorithm can be considered. There are various search algorithms, but here we will take one of the most famous search algorithms, the hill climbing method, as an example. The mountain climbing method was selected only in consideration of the ease of illustration, and the configuration of the third embodiment is not limited to this.

The offset amount control unit 10g includes an automatic search unit 10j. The automatic search unit 10j automatically searches for the optimum value of the angle difference correction amount by the mountain climbing method by investigating the increase / decrease of the harmonic component while changing the angle difference correction amount with respect to the harmonic component from the filter unit 10i. To do. FIG. 24 is a diagram showing an image of automatic search for the amount of angle difference correction by the mountain climbing method used in the automatic search unit shown in FIG. 23. The vertical axis represents the second harmonic component of the resonance angular frequency, and the horizontal axis represents the angle difference correction amount.

Since the cause of the generation of the second harmonic component of the pulsation component is an erroneous determination of the compensation direction of the magnetic flux current, it is expected that the second harmonic component will be minimized when the angle difference correction amount is optimally adjusted. At the operating point X under the initial condition, the angle difference correction amount is zero, and the compensation direction of the magnetic flux current is erroneously determined, so that the second harmonic component is observed to some extent. Here, it is assumed that the angle difference correction amount is increased and the operating point is changed from X to Y. In this case, since the operating point moves away from the direction of the optimum point, the frequency of erroneous determination increases and the second harmonic component increases. Since this is not desirable, consider reducing the amount of angle difference correction. When the angle difference correction amount is reduced and the operating point is changed from X to Z, the operating point approaches the direction of the optimum point, so that the frequency of erroneous determination is reduced and the second harmonic component is reduced. In this way, by using the hill climbing method, the optimum point is finally reached by searching for the point where the second harmonic component of the pulsation component is minimized while gradually changing the angle difference correction amount. When the angle difference signal is appropriately corrected, not only the second harmonic component of the electric spring resonance but also the fundamental wave component of the electric spring resonance gradually weakens. Therefore, even if the point where the second harmonic component is minimized is not found, the search may be stopped when the vibration weakens to some extent.

As another configuration example of the offset amount control unit 10g, a method of using PID control instead of using the search algorithm by the automatic search unit 10j can be considered. In the PID control of the offset amount control unit 10 g, for example, a portion where a sharp peak is generated is extracted from the sub-side torque current pulsation shown in FIG. 21 or FIG. 22, and the extracted peak portion is integrated. Therefore, an appropriate correction amount can be obtained.

The method based on the automatic search is excellent in that the search does not require a motor constant, although the search requires a certain amount of time. In the case of a motor having a large reluctance torque, many motor constants are required to solve the above equation (23). Therefore, a large number of motor constant measuring means as described in the second embodiment, for example, a test signal generator 64. , Test signal analysis unit 10e and the like are required. On the other hand, the method based on the automatic search of the third embodiment can obtain the effect that the control configuration can be simplified. Further, since the method based on the automatic search of the third embodiment does not require a motor constant, it is possible to correct the angle difference online, and it is possible to deal with the constant change when the synchronous motor is operated for a long time.

Embodiment 4.
FIG. 25 is a configuration diagram of an angle difference correction unit included in the drive device according to the fourth embodiment of the present invention. The angle difference correction unit 10E according to the fourth embodiment includes the test signal analysis unit 10e, the correction amount calculation unit 10d, and the data table 10c of the second embodiment, and further includes the pulsating component analysis unit 10f and the offset amount of the third embodiment. A control unit (10 g) is provided.

In the angle difference correction unit 10E, the corrected angle difference is calculated by subtracting the angle difference correction amount calculated based on the value of the motor constant from the angle difference, or the even-order harmonics generated by the erroneous determination of the compensation direction are generated. The corrected angle difference is calculated by subtracting the angle difference correction amount calculated based on the angle difference from the angle difference. In this way, the angle difference correction unit 10E can individually calculate the angle difference correction amount calculated based on the value of the motor constant and the angle difference correction amount calculated based on the even-order harmonics. The angle difference correction unit 10E may be configured to subtract both the angle difference correction amount calculated based on the value of the motor constant and the angle difference correction amount calculated based on the even-order harmonics from the angle difference. ..

According to the fourth embodiment, by using the angle difference correction amount calculated based on the value of the motor constant, it is possible to prevent erroneous determination of the compensation direction of the magnetic flux current when a motor having a large reluctance torque is used. Even when the motor constant is not required, it is possible to obtain the corrected angular difference by using the angular difference correction amount calculated based on the even-order harmonics.

Embodiment 5.
In the fifth embodiment, a configuration example of a fluid utilization device using the drive device 100 according to the first, second, third, and fourth embodiments will be described. FIG. 26 is a block diagram of the fluid utilization device according to the fifth embodiment of the present invention. In the fifth embodiment, the fluid utilization device 300 in which the propeller fan 300a is provided on the rotating shaft of the main-side synchronous motor 1a and the propeller fan 300b is provided on the rotating shaft of the sub-side synchronous motor 1b will be described.

The fluid utilization device 300 shown in FIG. 26 includes the drive device 100 of the first embodiment, and the drive device 100 includes a power converter drive device 200. The power converter drive device 200 includes a processor 201 and a memory 202. Each function shown in FIG. 1, that is, the current control unit 6, the pulsating component extraction unit 7, the subtractor 8, the angle difference correction unit 10, and the magnetic flux current command determination unit 9 is realized by using the processor 201 and the memory 202. To.

When the processor 201 and the memory 202 are used as shown in FIG. 26, each of the above functions is realized by software, firmware, or a combination thereof. The software or firmware is written as a program and stored in memory 202. The processor 201 reads and executes the program stored in the memory 202. It can also be said that these programs cause the computer to execute the procedures and methods to be executed by each of the above functions. The memory 202 is a RAM (Random Access Memory), a ROM (Read Only Memory), a flash memory, an EPROM (Erasable Programmable ReadOnly Memory), or an EEPROM (Electrically Memory) Memory (Electrically .. The semiconductor memory may be a non-volatile memory or a volatile memory. In addition to the semiconductor memory, the memory 202 corresponds to a magnetic disk, a flexible disk, an optical disk, a compact disk, a mini disk, or a DVD (Digital Versaille Disc). The processor 201 may output or store data such as a calculation result to the memory 202, or may store the data in an auxiliary storage device (not shown) via the memory 202.

The fluid utilization device 300 may include the drive device 100 of the second, third, or fourth embodiment instead of the drive device 100 of the first embodiment. In this case, the current control unit 6A shown in FIG. 19, the angle difference correction unit 10C shown in FIG. 20, the angle difference correction unit 10D shown in FIG. 23, and the angle difference correction unit 10E shown in FIG. 25 are the processor 201 and The function is realized by using the memory 202.

As described in the first embodiment, the power converter 2 may have basically any circuit configuration as long as it can supply arbitrary AC power to the main side synchronous motor 1a and the sub side synchronous motor 1b. .. The information detected by the current detection units 4a and 4b and the magnetic pole position detection units 5a and 5b is transmitted to the processor 201.

Although the relay circuit 11 is not an indispensable configuration requirement, the number of operating synchronous motors can be changed by using the relay circuit 11. Further, when switching to parallel operation of the two synchronous motors after operating one of the two synchronous motors for a long time, there is a temperature difference between the temperatures of the two synchronous motors. , The stability of the sub-side synchronous motor 1b may decrease due to the influence of the temperature difference. However, even when such switching is performed, it is possible to stably operate two synchronous motors in parallel by using the methods described in the first, second, third, and fourth embodiments.

The two propeller fans 300a and 300b may have the same shape or different shapes. Further, the air flow paths of the two propeller fans 300a and 300b do not necessarily have to be the same. For example, when the fluid utilization device 300 is an air conditioner, the two propeller fans 300a and 300b correspond to two blower fans provided in the blower chamber in the outdoor unit of the air conditioner, and the above air flow path is Corresponds to the air conditioner. The blower chamber is a space formed by being surrounded by side plates, ceiling plates, bottom plates, heat exchangers, etc. of the outdoor unit. An air flow is formed in the blower chamber by rotating the propeller fans 300a and 300b.

Since it is easier to stably drive in parallel if the characteristics of the rotation speed and the load torque of the two propeller fans 300a and 300b are different, fans of different shapes may be provided to the two synchronous motors, or one of them. The cross-sectional area of the flow path provided with the fan may be smaller than the cross-sectional area of the flow path provided with the other fan. Further, a fluid utilization device having different specifications may be driven, such as driving a propeller fan with one synchronous motor and driving a pump with the other synchronous motor.

Although not shown in FIG. 26, the fluid utilization device 300 includes a voltage detection unit that detects the voltage output by the power converter 2, and the voltage information detected by the voltage detection unit is input to the processor 201. It may be configured as follows. Further, although not shown in FIG. 26, the fluid utilization device 300 may include a wind speed sensor for measuring the wind speed of the fan, and may be configured so that the wind speed information detected by the wind speed sensor is input to the processor 201. .. Further, although not shown in FIG. 26, a temperature sensor for detecting the temperature of the object cooled by the fan may be provided, and the temperature information detected by the temperature sensor may be input to the processor 201. ..

The fluid load of the fluid utilization device 300 has a damper characteristic, and the damper characteristic stabilizes the drive of an open-loop driven synchronous motor in a high rotation speed region. However, in the low rotation range, the damper characteristics are weakened and the drive of the synchronous motor becomes unstable. Therefore, the fluid utilization device 300 aims to stabilize the drive of the synchronous motor by changing the magnetic flux current. However, when the magnetic flux current control method disclosed in Patent Document 1 is applied to the power converter drive device 200 of the fluid utilization device 300, when a difference occurs in the motor constants of the two synchronous motors, the angle is increased. In the region where the difference is close to zero, an erroneous determination of the compensation direction of the magnetic flux current occurs. Since the fluid utilization device 300 has a low load in the low rotation speed region, the angle difference approaches zero in the low rotation speed region, and erroneous determination occurs frequently. In this way, if erroneous determination of the compensation direction occurs frequently in the low rotation range where stability is low, there is a risk of step-out. Further, there is a concern that the vibration and noise of the synchronous motor will increase and the motor efficiency will decrease due to the continuous vibration generated due to the erroneous determination. For these reasons, using the magnetic flux current control method disclosed in Patent Document 1 for the power converter drive device 200 of the fluid utilization device 300 has a problem in terms of low-speed drive characteristics.

By using the angle difference correction method or the constant measurement method described in the first, second, third, and fourth embodiments in the power converter drive device 200 of the fluid utilization device 300 shown in FIG. 26, a misjudgment of the compensation direction in a low rotation range Can be prevented from occurring. As a result, in the fluid utilization device 300 of the fifth embodiment, parallel driving of the synchronous motor can be realized in a wide speed range. Further, the fluid utilization device 300 of the fifth embodiment can realize an existing synchronous motor drive device that drives one synchronous motor with one power converter by rewriting software or the like, so that the cost increases. It is possible to obtain a fluid utilization device 300 capable of driving two propeller fans 300a and 300b while suppressing the above.

Embodiment 6.
In the sixth embodiment, a configuration example of an air conditioner using the fluid utilization device 300 according to the fourth embodiment will be described. FIG. 27 is a configuration diagram of an air conditioner according to a sixth embodiment of the present invention. The air conditioner 400 according to the sixth embodiment includes a fluid utilization device 300, a refrigerant compressor 401, a condenser 403, a liquid receiver 404, an expansion valve 405, and an evaporator 406. The refrigerant compressor 401 and the condenser 403 are connected by a pipe. Similarly, the condenser 403 and the receiver 404 are connected by a pipe, the receiver 404 and the expansion valve 405 are connected by a pipe, and the expansion valve 405 and the evaporator 406 are connected by a pipe. It is connected, and the evaporator 406 and the refrigerant compressor 401 are connected by a pipe. As a result, the refrigerant circulates in the refrigerant compressor 401, the condenser 403, the liquid receiver 404, the expansion valve 405, and the evaporator 406. Although not shown in FIG. 27, the fluid utilization device 300 includes current detection units 4a and 4b and magnetic pole position detection units 5a and 5c shown in FIG. 1 and the like.

In the air conditioner 400, the steps of evaporating, compressing, condensing, and expanding the refrigerant are repeated, so that the refrigerant changes from liquid to gas, and further changes from gas to liquid, so that between the refrigerant and the outside air. Heat exchange takes place.

The evaporator 406 exerts a cooling action by evaporating the refrigerant liquid in a low pressure state and removing heat from the air around the evaporator 406. The refrigerant compressor 401 compresses the refrigerant gas gasified by the evaporator 406 in order to condense the refrigerant into a high-pressure gas. The condenser 403 discharges the heat of the refrigerant gas that has become hot in the refrigerant compressor 401 to condense the high-pressure refrigerant gas and convert it into a refrigerant liquid. The fluid utilization device 300 generates wind by rotating the propeller fans 300a and 300b, and cools the condenser 403 by passing the wind through the condenser 403. The expansion valve 405 throttles and expands the refrigerant liquid in order to evaporate the refrigerant, and converts the refrigerant liquid into a low-pressure liquid. The receiver 404 is provided for adjusting the amount of circulating refrigerant, and may be omitted in a small device.

As the size of the condenser 403 increases with the increase in the output of the air conditioner 400, it becomes necessary to increase the cooling performance of the fluid utilization device 300 that functions as a cooling device for cooling the condenser 403. However, it is complicated to change the specifications of the fluid utilization device 300 that functions as a cooling device in accordance with the increase in the size of the condenser 403. Further, in order to increase the cooling performance of the fluid utilization device 300, in order to increase the output of the fluid utilization device 300, it may be necessary to change the production line for mass production of the fluid utilization device 300. The initial investment to build the line is high. Therefore, in the large air conditioner 400, the cooling performance is improved by using the fluid utilization device 300 provided with a plurality of cooling fans.

Further, the air conditioner 400 is required to reduce the cost, and on the other hand, the energy saving regulation is tightened year by year, so that the efficiency is also required to be improved. In recent energy conservation regulations, not only the rated operating point but also the driving efficiency at the operating point of low output drive is emphasized. Therefore, it is necessary to lower the lower limit of the operating rotation speed of the cooling fan as much as possible.

As described above, the parallel drive device using the technique disclosed in Patent Document 1 is very excellent in terms of cost, but the technique disclosed in Patent Document 1 has a difference in motor constants. , An erroneous determination of the compensation direction of the magnetic flux current occurs, and continuous vibration occurs due to the erroneous determination. When the fluid utilization device 300 is used as a cooling fan for the air conditioner 400, the control on the low speed side tends to be unstable due to the characteristics of the fluid load. Therefore, when the command value of the air volume is a low-speed command, the number of operating synchronous motors may be reduced from two to one by using the relay circuit 11 shown in FIG. 1. In such a case, two synchronous motors may be operated. There may be a temperature difference in the temperature generated in each of the synchronous motors. When such a temperature difference occurs, the motor constants differ due to the temperature difference. Therefore, in the drive device using the magnetic flux current control method disclosed in Patent Document 1, the erroneous determination of the magnetic flux current compensation direction is made. May occur. For this reason, the drive device using the magnetic flux current control method disclosed in Patent Document 1 cannot lower the lower limit of the operating rotation speed of the cooling fan to a certain value or less, and includes the drive device. There was a problem that the operating range of the air conditioner was narrowed. Therefore, it is difficult for a drive device using the magnetic flux current control method disclosed in Patent Document 1 to achieve both low cost and high cooling performance required for a cooling fan of an air conditioner.

Since the air conditioner 400 according to the sixth embodiment uses the parallel drive method described in the first to fourth embodiments, the drive in the low speed range is not unstable and the driveable range can be expanded. .. Further, the air conditioner 400 according to the sixth embodiment does not require the addition of sensors for realizing parallel drive as compared with the magnetic flux current control method disclosed in Reference 1. Therefore, in the sixth embodiment, it is possible to achieve both low cost and high cooling performance required for the cooling fan of the air conditioner 400. Further, in the sixth embodiment, the reliability against fluctuations in the motor constant can be significantly improved as compared with the technique disclosed in the cited document 1.

Since the drive device 100 according to the first to fourth embodiments can stably drive the synchronous motors even when the motor constants of the two synchronous motors are completely different, other than the fluid utilization device 300 and the air conditioner 400. It can be applied to all kinds of equipment and is useful for industrial development.

The configuration shown in the above-described embodiment shows an example of the content of the present invention, can be combined with another known technique, and is one of the configurations without departing from the gist of the present invention. It is also possible to omit or change the part.

1a Main side synchronous motor, 1b Sub side synchronous motor, 2 Power converter, 3 DC voltage source, 4a, 4b Current detector, 5a, 5b, 5c Magnetic pole position detector, 6,6A Current control unit, 7,7A, 7B, 7C Pulsating component extraction unit, 8,10b, 61c, 61d, 61f subtractor, 9 Magnetic current command determination unit, 10,10A, 10B, 10C, 10D, 10E Angle difference correction unit, 10a, 10d correction amount calculation unit 10c data table, 10e test signal analysis unit, 10f pulsation component analysis unit, 10g offset amount control unit, 10h resonance angle frequency measurement unit, 10i filter unit, 10j automatic search unit, 11 relay circuit, 61a, 61b, 61e, 61g , 61h adder, 62a, 62b PID controller, 63 non-interfering controller, 64 test signal generator, 71 pulsating frequency measurement unit, 72 chord wave generator, 73 sine wave generator, 74 Fourier chord coefficient calculation unit, 75 Fourier sine coefficient calculation unit, 76 AC restorer, 91 pulsation suppression control unit, 92 compensation direction determination unit, 100 drive unit, 200 power converter drive unit, 201 processor, 202 memory, 300 fluid utilization device, 300a, 300b propeller Fan, 400 air conditioner, 401 refrigerant compressor, 403 condenser, 404 receiver, 405 expansion valve, 406 evaporator, 911 gain multiplier, 912 phase adjuster, 921 code determiner, 922 multiplier.

Claims (12)

A power converter that supplies power to the first synchronous motor and the second synchronous motor that are connected in parallel,
A first current detector that detects the first current flowing through the first synchronous motor, and
A second current detector that detects the second current flowing through the second synchronous motor, and
A first magnetic pole position detecting unit that detects the first magnetic pole position of the rotor of the first synchronous motor, and
A second magnetic pole position detecting unit that detects the second magnetic pole position of the rotor of the second synchronous motor, and
A control unit that outputs a voltage command for driving the first synchronous motor by using a torque current command, a magnetic flux current command, the first current, and the first magnetic pole position.
A subtractor for obtaining the first angle difference, which is the difference between the first magnetic pole position and the second magnetic pole position,
The first angular difference is corrected based on the difference in the motor constants of the first synchronous motor and the second synchronous motor, or based on the information indicating the electrical behavior caused by the difference in the motor constants. Then, the angle difference correction unit that outputs the second angle difference, which is the corrected angle difference,
A pulsating component extraction unit that extracts at least one pulsating component of a pulsating component included in the torque current flowing through the second synchronous motor and an active power pulsating component contained in the active power consumed by the second synchronous motor. ,
A magnetic flux current command determining unit that determines the compensation direction of the magnetic flux current command using the second angle difference and determines the magnetic flux current command using the determined compensation direction and the pulsating component.
A drive device characterized by being provided with. The angle difference correction unit calculates and calculates the correction amount of the first angle difference from the operating conditions of the first synchronous motor and the second synchronous motor and the difference in the motor constants. The driving device according to claim 1, wherein the first angle difference is corrected by using a correction amount. The angle difference correction unit
It is provided with a data table in which the correction amount of the first angle difference, the rotation speed of the rotor of the synchronous motor, and the difference of the motor constant are associated with each other.
With reference to the data table, the correction amount of the first angle difference corresponding to the difference of the motor constant and the speed command is calculated, and the calculated correction amount is used to correct the first angle difference. The drive device according to claim 1. The control unit includes a test signal generation unit that generates a test signal for measuring the difference in the motor constants.
The angle difference correction unit
A test signal analysis unit that analyzes voltage changes or current changes caused by the test signal is provided.
The driving device according to claim 1, wherein the first angle difference is corrected by using the analysis result of the voltage change or the current change. The angle difference correction unit
A pulsating component analysis unit that frequency-analyzes the pulsating component of the second synchronous motor,
An offset amount control unit that adds a correction amount to the first angle difference so that the harmonic component of the pulsation component is reduced.
With
The drive device according to claim 1, wherein the offset amount control unit corrects the first angle difference by using the harmonic component. The offset amount control unit is characterized in that it automatically searches for a correction amount in which the harmonic component decreases by investigating an increase or decrease in the harmonic component while changing the correction amount of the first angle difference. The drive device according to claim 5. When the first synchronous motor and the second synchronous motor are stopped, the test signal generator transmits the test signal of DC voltage or DC current to the first synchronous motor and the second synchronous motor. give,
The test signal analysis unit determines the electric motors of the first synchronous motor and the second synchronous motor based on the value of the current flowing through each of the first synchronous motor and the second synchronous motor or the ratio of the currents. Measure the child resistance and
The driving device according to claim 4, wherein the angle difference correction unit determines a correction amount of the first angle difference based on the armature resistance. When the first synchronous motor and the second synchronous motor are stopped, the test signal generation unit refers to the first synchronous motor and the second synchronous motor with respect to an AC voltage or an AC current in the d-axis direction. Given the test signal of
The test signal analysis unit measures the d-axis inductance from the value of the current flowing through each of the first synchronous motor and the second synchronous motor or the ratio of the currents.
The driving device according to claim 4, wherein the angle difference correction unit determines a correction amount of the first angle difference based on the d-axis inductance. When the first synchronous motor and the second synchronous motor are stopped, the test signal generator has at least the d-axis direction and the q-axis direction with respect to the first synchronous motor and the second synchronous motor. The test signal of the impulse-like voltage is given to one side,
The test signal analysis unit measures at least one of the d-axis inductance and the q-axis inductance from minute changes in the dq current flowing through each of the first synchronous motor and the second synchronous motor.
The driving device according to claim 4, wherein the angle difference correction unit determines a correction amount of the first angle difference based on at least one of a d-axis inductance and a q-axis inductance. The test signal generation unit changes the DC component of the magnetic flux current command when driving the first synchronous motor and the second synchronous motor.
The test signal analysis unit calculates the armature resistance from the current values flowing in each of the first synchronous motor and the second synchronous motor before and after changing the DC component of the magnetic flux current command.
The driving device according to claim 4, wherein the angle difference correction unit determines a correction amount of the first angle difference based on the armature resistance. A fluid utilization device comprising the drive device according to any one of claims 1 to 10. An air conditioner comprising the fluid utilization device according to claim 11.

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