JP6944628B2 - Double three-phase winding permanent magnet synchronous motor drive system - Google Patents

Description

The present invention is a fixed having a rotor having a permanent magnet and two three-phase windings (first three-phase winding and second three-phase winding, or own three-phase winding and other three-phase winding). A dual three-phase winding permanent magnet synchronous motor equipped with at least a dual three-phase winding permanent magnet synchronous motor consisting of children and a power conversion device capable of simultaneously supplying a three-phase current to two three-phase windings. Regarding the drive system of. In the present invention, the two three-phase windings may be referred to as a self-three-phase winding and another three-phase winding, and the distinction between the self and the other is removed, and the first three-phase winding and the second three-phase winding are simply used. It is also called a three-phase winding. When the first three-phase winding is a self-three-phase winding, the second three-phase winding is another three-phase winding. Similarly, when the second three-phase winding is a self-three-phase winding, the first three-phase winding is another three-phase winding. Hereinafter, in order to ensure the clarity of the description, the invention will be described with the first three-phase winding as its own three-phase winding and the second three-phase winding as the other three-phase winding unless otherwise specified. .. Thereby, the generality of the invention is not lost.

Further, in the following description, for the sake of simplicity, "winding" is used synonymously with "three-phase winding". The above-mentioned double Sanwa winding permanent magnet synchronous motor is abbreviated as a double synchronous motor for the sake of simplicity. Furthermore, for the same reason, the drive system is abbreviated as a dual synchronous motor drive system. The applications of the dual synchronous motor drive system of the present invention include main drive of battery electric vehicles, fuel cell electric vehicles, hybrid electric vehicles, etc., applications of home appliances that require a wide range of efficient drive, fault resistance, and functional safety. Is the required application.

In the present invention, the portion of the dual synchronous motor to which the three-phase winding is applied is referred to as a "stator". The "stator" in the present invention is synonymous with "armature". There are Y-type and Δ-type in the three-phase winding applied to the stator. As is well known to those skilled in the art, when evaluated from a three-phase terminal, the characteristics of the Y-shaped winding and the characteristics of the Δ-shaped winding are equivalently converted to each other. In order to ensure the simplicity of the description, the technical description in this specification assumes a Y-shaped connection. As evidenced by the existence of the equivalent transformation, this does not lose the generality of the invention.

In the present invention, a two-dimensional plane is grasped in polar coordinates, and the three terms of angle, spatial position, and spatial phase are used synonymously. These units are "radians" or "degrees". The angle, the spatial position, and the positive direction of the spatial phase in the present invention may be defined as either left-handed (counterclockwise) or right-handed (clockwise). However, in the present specification, in order to maintain the simplicity of the description, the positive direction of the angle, the spatial position, and the spatial phase is defined as left-handed (counterclockwise), and the present invention will be described. Thereby, the generality of the present invention is not lost.

In the present invention, a device that supplies AC power to a dual synchronous motor is referred to as a power conversion device. Inverters, matrix converters, and the like have been put into practical use as power converters, which are the main devices of power converters. A single / six-phase power converter, a two-phase / three-phase power converter, a six-phase / single-phase power converter, or the like can constitute the power converter of the present invention.

In the present invention, as a general rule, parameters, physical quantities, etc. related to the first winding and the second winding are specified by adding legs 1 and 2, respectively. In addition, as a general rule, in the dq synchronous coordinate system (detailed later with reference to FIG. 5) consisting of two orthogonal axes of the d-axis and the q-axis, the legs d and q are used for parameters, physical quantities, etc. related to each axis. Attach to clearly indicate the relationship with each axis. Furthermore, as a general rule, the signal corresponding to the command value of the response signal is clearly indicated to be the command value of the corresponding signal by adding a prefix * to the same symbol as the response signal.

漏れインダクタンスを次の（２）式に従い定義する。

本発明における「漏れインダクタンスの相当値」は、真の漏れインダクタスσＬ１、σＬ２概略値を意味する。この場合の概略値は、真値の数倍値をも含む（後掲の（２２）式参照）。When the self-inductance of the own winding is L1, the self-inductance of the other winding is L2, and the mutual inductance between both windings is M, the leakage coefficient σ is defined according to the following equation (1).

Leakage inductance is defined according to the following equation (2).

The "equivalent value of leakage inductance" in the present invention means the approximate values of true leakage inductors σL1 and σL2. The approximate value in this case also includes a value several times the true value (see equation (22) below).

なお、（３）式中辺は自己インダクタンスと漏れインダクタンスの差を示している。また、同式右辺の近似は、漏れ係数σが微小な場合には問題なく成立する。当然のことながら、相互インダクタス相当値は非ゼロである。The "equivalent value of mutual inductance" in the present invention means a general term for a wide range of approximate values of the true value of mutual inductance M. This approximate value includes a value expressed by the following equation (see equation (25) below).

The middle side of Eq. (3) shows the difference between self-inductance and leakage inductance. Further, the approximation of the right side of the same equation holds without any problem when the leakage coefficient σ is small. As a matter of course, the mutual inductors equivalent value is non-zero.

In the present invention, the "current equivalent value" is used as a general term for the true value, command value, estimated value, approximate value, etc. of the stator current.

Similarly, in the present invention, a two-axis orthogonal rotating coordinate system that rotates the "velocity equivalent value" of the rotor at the true value, command value, estimated value, approximate value of the rotor speed, and the same average speed as the rotor. It is used as a general term for the true value and approximate value of the speed of. As is well known to those skilled in the art, there are electric speed and mechanical speed in rotor speed, but there is a strict one-to-one relationship between both speeds, from electric speed to mechanical speed and from mechanical speed to electric speed. Unique conversion to is possible. In the present invention, considering the familiarity among those skilled in the art, unless the clarity of the description is lost, the rotor speed is used as meaning the electric speed. In the present invention, "speed multiplication equivalent processing" is used as a general term for processing involving multiplication of speed equivalent values.

In the present invention, "differential equivalent processing" is used as a general term for processing centered on differential processing, such as pure differential processing, approximate differential processing, and linear differential processing with non-linear processing such as limiter processing added.

Prior inventions relating to a dual synchronous motor to be driven by the dual synchronous motor drive system of the present invention include, for example, Patent Documents 1 and 2 and Non-Patent Documents 1 to 5. The previously reported double synchronous motors include a three-phase simple synchronous motor (Non-Patent Document 1) and a six-phase synchronous motor (Patent Document 2, Non-Patent Documents 2 and 3) from the viewpoint of arranging the double three-phase winding of the stator. ), And three-phase reverse synchronous motors (Non-Patent Documents 4 to 5) are roughly classified into three types.

With reference to Non-Patent Document 1, the outline of the conventional double synchronous motor (three-phase simple synchronous motor) is shown in FIG. 1 by taking the case where the pole logarithm Np is Np = 1. 1 is a double synchronous motor (including a rotor and a stator), 11 is a rotor of a double synchronous motor, 121 is a first winding of a stator of a double synchronous motor, and 122 is a double synchronous motor. The second winding of the stator of is shown respectively. In the figure, in order to clarify the distinction between the first winding and the second winding of the stator, the first winding is indicated by a solid line and the second winding is indicated by a broken line. Further, since the second winding overlaps with the first winding due to the winding arrangement, the second winding is intentionally shifted to the right for drawing in order to avoid duplication in drawing.

The double synchronous motor (three-phase simple synchronous motor) with this arrangement has the following features. (A) For both the first winding and the second winding, the u-phase winding, the v-phase winding, and the w-phase winding are sequentially 2π / 3 [rad] spatial in the space based on the one-pole logarithm. It is arranged at the phase advance position. (B) In principle, the first winding and the second winding are arranged in space without any phase difference. (C) In principle, the pole logarithm can be any integer. That is, an odd or even number of pole pairs can be adopted. (D) When the first winding and the second winding are energized at the same time, or when only one of the windings is energized, the number of phases remains unchanged at three phases. (E) In principle, in the case of simultaneous energization of the first winding and the second winding, the currents of the first winding and the second winding need to be synchronized without a phase difference.

As a second example (example of a six-phase synchronous motor) of the stator winding arrangement of the double synchronous motor, an example in which the pole logarithm Np is set to Np = 1 with reference to Patent Document 2 and Non-Patent Documents 2 and 3. 2 is shown schematically together with the rotor (drawing of winding resistance is omitted). The meanings of the drawn wire numbers 1, 11, 121, and 122 are the same as those in FIG. However, it differs from the example of FIG. 1 in that the arrangement of the second winding is spatially shifted by θ12 = π / 6 [rad] with respect to the first winding in a space based on the logarithm of one pole. ing.

The double synchronous motor (six-phase synchronous motor) according to this arrangement has the following features as compared with the three-phase simple synchronous motor (FIG. 1). (A) Same as item (a) of Example 1. (B) In principle, the first winding and the second winding are arranged so as to have a spatial phase difference of π / 6 [rad] in a space based on a one-pole logarithm. (C) Same as item (c) of Example 1. (D) When the first winding and the second winding are energized at the same time, it operates as a six-phase motor, and when only one of the windings is energized, it operates as a three-phase motor. (E) In principle, in the case of simultaneous energization of the first winding and the second winding, the currents of the first winding and the second winding have a phase difference corresponding to the spatial phase difference. Synchronization is required.

As a third example of the stator winding arrangement of the double synchronous motor (example of the three-phase reverse synchronous motor), FIG. 3 shows an example in which the pole pair number Np is Np = 2 with reference to Non-Patent Documents 4 to 5. (Drawing of winding resistance is omitted). The meanings of the drawn wire numbers 1, 11, 121, and 122 are the same as those in FIG.

The double synchronous motor (three-phase reverse synchronous motor) with this winding arrangement has the following features as compared with Examples 1 and 2. (A) For both the first winding and the second winding, the u-phase winding, the v-phase winding, and the w-phase winding are sequentially 2π / 3 [rad] spatial in the space based on the 2-pole logarithm. It is located at a phase lag position. (B) In principle, the first winding and the second winding are arranged so as to have a phase difference of ± π [rad] in a space based on a two-pole logarithm. (C) Only an even number of pole logarithms can be taken. That is, an odd number of pole pairs cannot be adopted. (D) Same as item (d) of Example 1. (E) Same as item (e) of Example 1.

In the dual synchronous motor illustrated in FIGS. 1 to 3, the first winding and the second winding do not necessarily have to have the same characteristics. Both windings are configured to have the same characteristics as shown in Patent Documents 1 and 2 and Non-Patent Documents 1 to 4, and are also different from each other as shown in Non-Patent Document 5. It can also be configured to have characteristics.

In the dual synchronous motor illustrated in FIGS. 1 to 3, the neutral point of the first winding and the neutral point of the second winding are not connected. In the dual synchronous motor targeted by the present invention, in general, the neutral point of the first winding and the neutral point of the second winding can be disconnected or connected.

Next, we will introduce the conventional technology related to the dual synchronous motor drive system, that is, the drive system for the dual synchronous motor. The present invention relates to a current control device, which is one of the main components of a dual synchronous motor drive system. Based on this point, we will introduce the conventional technology related to the current control device for the dual synchronous motor drive system. FIG. 9 cites a current control device using an interwinding non-interfering device for a dual synchronous motor drive system proposed in Non-Patent Document 1 (Patent Document 1 is also substantially the same by the same inventor. The same interwinding non-interferer is shown). In addition, Non-Patent Document 1 targets the three-phase simple synchronous motor of FIG. 1 as a double synchronous motor, and the double synchronous motor at this time is non-polar.

（４）式では、第１巻線用のフィードバック電流制御器の出力信号（図９では「電圧指令−１」と記載）をｖ１１＊で表現し、第２巻線用のフィードバック電流制御器の出力信号（図９では「電圧指令−２」と記載）をｖ２２＊で表現している。フィードバック電流制御器からのこれら２出力信号（２電圧指令値）ｖ１１＊、ｖ２２＊が、（４）式で数式表現された非干渉化部の入力信号となっている。（４）式における係数（Ｍ／Ｌ）は、非突極二重同期電動機のインダクタンスより定まる定数である。非干渉化部の出力信号であるｖ１２＊、ｖ２１＊は、（４）式に従い生成されている。In FIG. 9, the current control device is composed of the first and second "current control systems" arranged at the left end and the "non-interfering unit" arranged at the center. The "current control system" in FIG. 9 means "feedback current controller" in the correct academic term (technical term). The configuration of the non-interfering portion in FIG. 9 is described by the following mathematical formula.

In equation (4), the output signal of the feedback current controller for the first winding (described as "voltage command-1" in FIG. 9) is expressed by v11 *, and the feedback current controller for the second winding is represented by v11 *. The output signal (described as "voltage command-2" in FIG. 9) is represented by v22 *. These two output signals (two voltage command values) v11 * and v22 * from the feedback current controller are input signals of the non-interfering unit expressed mathematically by the equation (4). The coefficient (M / L) in the equation (4) is a constant determined by the inductance of the non-slip pole double synchronous motor. The output signals v12 * and v21 * of the non-interfering unit are generated according to the equation (4).

ここに、ｖ１＊、ｖ２＊は各々第１巻線用の最終電圧指令値、第２巻線用の最終電圧指令値を意味する。なお、簡略図である図９においては、インバータ等の電力変換装置は省略され記載されていない。The output signal v12 * is a decoupling signal to the side of the first three-phase winding, and the output signal v21 * is a decoupling signal to the side of the second three-phase winding. Is added to the output signal (voltage command value) of the above, and the final voltage command values v1 * and v2 * are generated. The generation of this final voltage command value is described by the following equation.

Here, v1 * and v2 * mean the final voltage command value for the first winding and the final voltage command value for the second winding, respectively. In FIG. 9, which is a simplified diagram, a power conversion device such as an inverter is omitted and not described.

As is clear from the equations (4) and (5), the conventional interwinding non-interfering device mutually weights the output signal (voltage command value) of the feedback current controller for each winding in a constant linear relationship. The final voltage command value for each winding is combined. That is, the input signal of the conventional interwinding non-interfering device is a voltage command value, and the conventional interwinding non-interfering device should also be called a "voltage type". Further, the current control device with the voltage type interwinding non-interfering device is constructed for a non-polar double synchronous motor without performing rigorous mathematical analysis. When mutual induction between the first winding and the second winding (in the present invention, mutual induction and magnetic coupling are used synonymously) is strong, current control is performed when a current control device as in Patent Document 1 is used. A system destabilization phenomenon or a vibration phenomenon similar thereto easily occurs (see, for example, Non-Patent Document 2), and the desired current control performance cannot be exhibited at all.

Akira Satake and Shigeki Mizuno: "Control device for multi-winding motor", Japanese Patent Application Laid-Open No. 2001-341135 (2001-11-6) Keiichiro Ban, Kazuyoshi Obayashi: "Electric Drive Device for Automobiles", Japanese Patent Application Laid-Open No. 2000-41392 (1998-7-23)

Akira Satake, Satoshi Kato, Akira Imanaka: "Modeling of Multi-Wound Permanent Magnet Motors and Non-Interference Control Methods", Proceedings of the IEEJ Industrial Application Division, I, pp. 199-202 (2005) Takafumi Imai, Fumiaki Osawa, Yasushi Yamada, Yukio Inakuma: "Possibility of Standardization of EV / HEV Electric Drive Systems (Current Ripple Suppression of Multi-Phase Motors)", Proceedings of the National Conference of the Institute of Electrical Engineers of Japan, 4, pp. 361-362 (2016) Tatsuya Mori and Akira Furukawa: "Pulse pattern to reduce torque ripple in double three-phase PMSM drive 1 shunt current detection double inverter", Proceedings of the IEEJ Industrial Application Division, III, pp. 159-164 (2016) Shinji Shinnaka: "Three-Phase Permanent Magnet Synchronous Motor with Inverse Double Three-Phase Winding with 180 Degree Spatial Phase Difference (Double Winding Arrangement, Dynamic Mathematical Model, Vector Simulator)", 2016 Institute of Electrical Engineers of Japan Industry Proceedings of the Applied Division Conference, III, pp. 285-290 (2016) Shinji Shinnaka: "Three-Phase Permanent Magnet Synchronous Motor with Inverse Double Three-Phase Winding with 180 Degree Spatial Phase Difference (Double Winding Arrangement, Dynamic Mathematical Model, Vector Simulator)", Journal of the Institute of Electrical Engineers of Japan D, Vol. 137, No. 2, pp. 75-86 (2017)

The present invention has been made under the above background. It is also applicable when the dual synchronous motor has strong mutual induction between the first and second windings, and thus high quality with high stability of the first and second windings. The purpose is to provide a new current control device for a dual synchronous motor drive system that enables current control.

In order to achieve the above object, the invention of claim 1 is permanently composed of a rotor having a permanent magnet and a stator having two three-phase windings (own three-phase winding and other three-phase winding). It includes a magnet-synchronized electric motor, a power conversion device that can simultaneously supply current to two three-phase windings, and a current control device that controls the current flowing through the two three-phase windings via the power conversion device. A permanent magnet synchronous motor drive system with a controller coefficient determined by at least utilizing the equivalent value of the leakage inductance caused by the mutual induction of the two three-phase windings, and of the two three-phase windings. Equivalent to the mutual inductance caused by the mutual induction of the two three-phase windings and the high-speed mode current controller for each three-phase winding that generates the voltage command value for controlling the high-speed mode current generated by mutual induction. Using at least the value and the current-equal value of the own three-phase winding, generate the voltage command value for the own three-phase winding for canceling the low-speed mode current generated by the mutual induction of the two three-phase windings. The current control device is configured by using at least one of two different devices (high-speed mode current controller and low-speed mode current canceller) with a low-speed mode current canceller for each three-phase winding. do.

The invention of claim 2 is the permanent magnet synchronous motor drive system according to claim 1, wherein the phase of the north pole of the rotor permanent magnet is the phase of the d-axis, and π / 2 [rad] with respect to the d-axis. ] When the 2-axis orthogonal coordinate system having the q-axis in the phase advance is the dq synchronous coordinate system and the coordinate system based on the dq synchronous coordinate system is the γδ quasi-synchronous coordinate system, the dq synchronous coordinate system or the γδ quasi-synchronous coordinate The current deviation of each three-phase winding, which is the difference between the current command value of each three-phase winding on the system and the current response value of each three-phase winding, is processed in a feedback-controlled manner on the dq synchronous coordinate system or γδ. It is characterized in that the high-speed mode current controller for each three-phase winding is configured so as to generate the voltage command value of each three-phase winding on the quasi-synchronous coordinate system.

生成の自三相巻線用の中間信号に対して、微分相当処理、速度乗算相当処理の少なくともいずれかの処理を施して、自三相巻線用電圧指令値を生成するように、該各三相巻線用低速モード電流キャンセラを構成したことを特徴とする。The invention of claim 3 is the permanent magnet synchronous motor drive system according to claim 1, wherein the phase of the north pole of the rotor permanent magnet is the phase of the d-axis, and π / 2 [rad] with respect to the d-axis. ] When the two-axis orthogonal coordinate system having the q-axis in the phase advance is the dq synchronous coordinate system and the coordinate system based on the dq synchronous coordinate system is the γδ quasi-synchronous coordinate system, further, on the dq synchronous coordinate system or γδ quasi. The current equivalent value of the self-three-phase winding on the synchronous coordinate system is i1 ^~ , the current equivalent value of the other three-phase winding is i2 ^~, and mutual induction of two (self and other) three-phase windings. When the equivalent values of the mutual inductance caused by the above are M1d ^~ , M1q ^~ , M2d ^~ , M2q ^{~ , an intermediate signal φ1M ~} for the self-three-phase winding is generated according to the following equation.

Each of the generated intermediate signals for the three-phase winding is subjected to at least one of the differential equivalent processing and the speed multiplication equivalent processing to generate the voltage command value for the own three-phase winding. It is characterized by configuring a low-speed mode current canceller for three-phase winding.

The effect of the present invention will be described. The present invention is based on the fact recognition that "currents in high-speed mode and low-speed mode appear due to mutual induction between the own winding (first winding) and the other winding (second winding)", and the high-speed mode current The high-speed mode current controller controls this, and the low-speed mode current canceller cancels and suppresses this with the low-speed mode current canceller, and flows to the two windings, the own winding and the other winding. It is intended to control the current in a stable manner. The fact that the high-speed mode and the low-speed mode are generated by mutual induction, which is one of the keys to the implementation of the present invention, will be clarified.

上式における漏れ係数σの定義は（１）式の通りである。Consider FIG. 4 (a). The figure schematically shows how two single-phase LR circuits are magnetically coupled (mutually induced). At this time, the inductance and resistance on the primary side and the secondary side are L ₁ , L ₂ , R ₁ , and R ₂ , respectively, and the mutual inductance is M. The characteristic equation that governs the characteristics of the circuit is given by the following equation.

The definition of the leakage coefficient σ in the above equation is as in equation (1).

一方、１次側と２次側との間に強い相互誘導が存在し、下の（９ａ）式が成立する場合には、２個の特性根ｓ１、ｓ２は、概ね（９ｂ）式の２実根となる。

Ｒ１／Ｌ１＞＝Ｒ２／Ｌ２とするとき、（８）、（９）式は、相互誘導の向上σ→０に応じて、２個の特性根は高速モード対応の根（以下、高速根）ｓ１と低速モード対応の根（以下、低速根）ｓ２に変化することを意味する。すなわち、

相互誘導の向上σ→０に応じた低速根ｓ２の変化は緩慢・微小であるが、高速根ｓ１の変化は、逆比１／σの発散的増大を示す。When σ = 1 in which mutual induction between the primary side and the secondary side does not exist, the characteristic roots s1 and s2 that determine the circuit mode become the following two real roots.

On the other hand, when there is strong mutual induction between the primary side and the secondary side and the following equation (9a) holds, the two characteristic roots s1 and s2 are approximately 2 in equation (9b). It becomes a real root.

When R1 / L1> = R2 / L2, in equations (8) and (9), the two characteristic roots correspond to the high-speed mode (hereinafter referred to as high-speed roots) according to the improvement of mutual induction σ → 0. It means that the root changes to s1 and the root corresponding to the low speed mode (hereinafter, low speed root) s2. That is,

Improvement of mutual induction The change of slow root s2 according to σ → 0 is slow and minute, but the change of fast root s1 shows a divergent increase of inverse ratio 1 / σ.

（１１）式の中辺は１次側端子から見た表現であり、右辺は２次側端子から見た表現である。（１１）式の第１式の分母における（σＬ１）、（σＬ２）は、（２）式の通り展開され、これらは、各々１次側端子、２次側端子からみた漏れインダクタンスを意味している。The high-speed root s1 and low-speed root s2 at the right end of the equation (10) can be rewritten as in the following equation (11).

The middle side of the equation (11) is the expression seen from the primary side terminal, and the right side is the expression seen from the secondary side terminal. (ΣL1) and (σL2) in the denominator of the first equation of the equation (11) are expanded as in the equation (2), and these mean the leakage inductance seen from the primary side terminal and the secondary side terminal, respectively. There is.

In the configuration of the feedback current control system that targets a circuit with strong mutual induction, only the two low-speed roots in Eq. (8) are assumed, and the existence of the high-speed root s1 in Eq. (9b) and Eq. 1 is ignored. In that case, the current control system becomes unstable (generation of high-frequency current ripple, current divergence, etc.).

In order to simplify the explanation of the effect of the present invention, the coordinate system will be described before starting the full-scale explanation. Consider FIG. FIG. 5 shows an αβ fixed coordinate system, a dq synchronous coordinate system, and a γδ general coordinate system. The αβ fixed coordinate system is a coordinate system corresponding to the stator, and in general, the α axis is taken at the center of the u-phase winding of the stator first winding (of the u-phase winding of the stator second winding). There is no essential difference for the center). The dq synchronous coordinate system is one of the rotating coordinate systems, and in particular, the d-axis is a coordinate system synchronized with the north pole of the rotor. That is, in the dq synchronous coordinate system, the phase of the d-axis is the same as the phase of the rotor magnetic flux. The velocity of the dq synchronous coordinate system is instantly the same as the rotor velocity ωn. The γδ general coordinate system is a generalized coordinate system having an arbitrary coordinate system velocity ωγ. The γδ general coordinate system includes the αβ fixed coordinate system and the dq synchronous coordinate system as a special case. It also includes a γδ quasi-synchronous coordinate system, which can have a small phase difference with respect to the dq synchronous coordinate system, as a special case. Regarding the phase of the coordinate system, the phase of the d-axis seen from the α-axis is θα, the phase of the γ-axis seen from the α-axis is θαγ, and the phase of the d-axis seen from the γ-axis is θγ.

The three types of dual synchronous motors shown in FIGS. 1 to 3 have clearly different winding arrangements, but on the dq synchronous coordinate system, these mathematical models (circuit equations) commonly have the following equations. (See Non-Patent Document 5).

The meanings of the legs 1, 2 and d, q in the mathematical model have already been explained. The symbol s means the differential operator d / dt. The physical quantity used in the mathematical model will be described using the first winding as an example. The 2 × 1 vectors v1 and i1 defined on the dq synchronous coordinate system mean the voltage and current of the stator, respectively. I is a 2 × 2 identity matrix. R1 is the resistance of the stator winding, and Φ1 is the rotor magnetic flux strength (electromotive force constant) seen from the stator winding. L1d and L1q are d-axis (self) inductance and q-axis (self) inductance. Further, Md and Mq are d-axis mutual inductance and q-axis mutual inductance, respectively.

（１３）式が明瞭に示しているように、同相インダクタンスＬ１ｉは、対応のｄ軸、ｑ軸インダクタンスの平均値を意味している。In the present invention, the in-phase (self) inductance L1i and the mirror phase (self) inductance L1m are defined by the following equation (13) using the d-axis (self) inductance L1d and the q-axis (self) inductance L1q. And use it.

As the equation (13) clearly shows, the common-mode inductance L1i means the average value of the corresponding d-axis and q-axis inductances.

In the dual synchronous motor targeted by the present invention, the motor parameters (winding resistance R1, inductance L1d, L1q, etc.) caused by the two three-phase windings may be the same or different. .. The present invention does not depend on the difference in motor parameters due to windings.

特にゼロ速度ωｎ＝０では、ｄ軸、ｑ軸間における軸間干渉は消滅し、仮想ベクトル回路は、「図４（ａ）の単相用ＬＲ回路（相互誘導回路）を、軸ごとのパラメータに置換した上で、軸ごとに独立・並列に配した二相回路」に帰着する。ゼロ速度での同一性より理解されるように、二重同期電動機は、ｄ軸、ｑ軸の漏れ係数σｄ、σｑの逆数におおむね比例した高速モードを有する。併せて、低速モードを有する。なお、各軸の漏れ係数σｄ、σｑは、（１）式と同様の次式に従い定義されている。

The relationship of equation (12) is drawn as shown in FIG. 4 (b) as a virtual vector circuit on the dq synchronous coordinate system that rotates at ωn. All physical quantities such as voltage and current in the figure are 2 × 1 vectors defined on the dq synchronous coordinate system. The 2 × 2 inductance matrix in the figure is defined as follows.

In particular, at zero speed ωn = 0, the inter-axis interference between the d-axis and the q-axis disappears, and the virtual vector circuit uses the single-phase LR circuit (mutual induction circuit) shown in FIG. 4A as a parameter for each axis. After replacing with, it comes down to "a two-phase circuit arranged independently and in parallel for each axis". As can be understood from the identity at zero speed, the dual synchronous motor has a high speed mode that is roughly proportional to the reciprocals of the d-axis and q-axis leakage coefficients σd and σq. In addition, it has a low speed mode. The leakage coefficients σd and σq of each axis are defined according to the following equation similar to equation (1).

As can be understood from the zero-speed equivalence of the single-phase LR circuit (mutual induction circuit) in FIG. 4 and the virtual vector circuit of the dual synchronous motor, the first winding and the second winding have strong mutual induction. In the configuration of the feedback current control system for the dual synchronous motor, when only the low speed mode current caused by the low speed root is assumed, the feedback current control system is easily destabilized (generation of high frequency current ripple, current divergence). Etc.).

According to the first aspect of the present invention, a double synchronous motor having a strong mutual induction between the first winding and the second winding operates in two different modes, a high-speed mode and a low-speed mode (high-speed mode current, low-speed mode current, low speed). It is intended to control these separately after deeply recognizing that they have a mode current). In particular, according to the invention of claim 1, the two three-phase windings have a controller coefficient determined by at least utilizing the equivalent value of the leakage inductance caused by mutual induction of the two three-phase windings. The high-speed mode current is controlled by utilizing the high-speed mode current controller for each three-phase winding that generates a voltage command value for controlling the high-speed mode current generated by mutual induction. As has already been analytically clarified, since the high-speed mode current is dominated by the leakage inductance, the effect that the high-speed mode current can be effectively controlled can be obtained according to the invention of claim 1. As a result, instability due to high-speed mode current can be prevented. That is, the effect of ensuring the stability of the feedback current control system can be obtained.

Successful control of high-speed mode current does not mean successful control of low-speed mode current. Even when the high-speed mode current is properly controlled, the low-speed mode current causes low-frequency pulsation in the winding current. As can be understood from the virtual vector circuit of FIG. 4B, the low-speed mode current is the mutual inductance matrix M (see equation (14)) and the self-three-phase when the self-three-phase winding is the first winding. It can be seen that the winding current (that is, the first winding current) i1 is contained in the form of a product of these. Similarly, the low-speed mode current is the mutual inductance matrix M (see equation (14)) and the current of the self-three-phase winding (that is, the second winding current) when the self-three-phase winding is the second winding. It can be seen that i2 is contained in the form of a product of these. According to the invention of claim 1, the two three-phase windings use at least the equivalent value of the mutual inductance caused by the mutual induction of the two three-phase windings and the current phase equivalent value of the own three-phase windings. It constitutes a low speed mode current canceller for each three-phase winding that generates a voltage command value for canceling the low speed mode current generated by mutual induction. That is, according to the invention of claim 1, the low-speed mode current can be offset via the voltage command value generated by the low-speed mode current canceller. As a result, the effect of suppressing the low-frequency pulsation of the winding current caused by the low-speed mode current can be obtained. In particular, when used in combination with a high-speed mode current controller that contributes to ensuring stability, it is possible to obtain the effect of bringing about a high-quality current control effect that suppresses low-frequency pulsation. The invention of claim 1 does not impose any restrictions on the salient polarity of the double synchronous motor, and as a matter of course, it can be applied to the double synchronous motor having a strong salient polarity.

例えば、（１６）式が示す第１巻線の電圧ｖ１に関しては、（１６ａ）式の左辺が電圧ｖ１そのものを示しており、同式の右辺がモード対応の電圧を示している。特に、ｖ１^〜は高速モード電流

の詳細は、各々（１６ｂ）式、（１６ｃ）式に示されている。（１６ｂ）式、（１６ｃ）式より再確認されるように、高速モード電流はｄ軸、ｑ軸の漏れインダクタンス（σｄ・Ｌ１ｄ）、（σｑ・Ｌ１ｑ）によって、低速モード電流はｄ軸、ｑ軸の相互インダクタンス相当値（（１−σｄ）Ｌ１ｄ，Ｍｄ）、（（１−σｑ）Ｌ１ｑ，Ｍｑ）によって支配れている。（１７）式が示す第２巻線の電圧ｖ２に関しても同様である。The effects of the high-speed mode current controller and low-speed mode current canceller described above focus on the leakage inductance directly related to the high-speed mode current and the mutual inductors directly related to the low-speed mode current, and equation (12). It can also be confirmed by re-expanding the circuit equation of.

For example, regarding the voltage v1 of the first winding represented by the equation (16), the left side of the equation (16a) indicates the voltage v1 itself, and the right side of the equation indicates the voltage corresponding to the mode. In particular, v1 ^~ is the high-speed mode current.

Details of the above are shown in Eqs. (16b) and (16c), respectively. As reconfirmed from the equations (16b) and (16c), the high-speed mode current has the d-axis and q-axis leakage inductances (σd · L1d) and (σq · L1q), and the low-speed mode current has the d-axis and q. It is governed by the mutual inductance equivalent values of the shafts ((1-σd) L1d, Md) and ((1-σq) L1q, Mq). The same applies to the voltage v2 of the second winding represented by the equation (17).

Next, the effect of the invention of claim 2 will be described. The high-speed mode current controller according to the invention of claim 1 can be constructed on any of the dq synchronous coordinate system, the γδ quasi-synchronous coordinate system, and the αβ fixed coordinate system. However, the difficulty of construction is not the same. Construction on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system is the easiest and is expected to be effective. According to the invention of claim 2, each three-phase winding is the difference between the current command value of each three-phase winding on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system and the current response value of each three-phase winding. High-speed mode current control for each three-phase winding so that the current deviation of the line is processed in a feedback control manner and the voltage command value of each three-phase winding on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system is generated. Can configure a vessel. As a result, according to the invention of claim 2, the effect that the high-speed mode current controller can be easily constructed can be obtained. As a result, according to the invention of claim 2, the effect that the effect of the invention of claim 1 can be enhanced can be obtained.

（１８）式における頭符＊の意味は、すでに説明した通りである。請求項３の発明に基づき構成された低速モード電流キャンセラによれば、第１巻線を自巻線とし、第２巻線を他巻線とする場合には、（１８ｂ）式を包含する（６）式に従って自三相巻線用の中間信号φ１Ｍ^〜を生成し、生成の自三相巻線用の中間信号に対して、微分相当処理、速度乗算相当処理の少なく

発明は（１８）式の構成を包含しており、請求項３の発明によれば、低速モード電流キャンセラは理想的な構造の１つを採用できるようになると言う効果が得られる。ひいては、低速モード電流キャンセラは好ましい性能を発揮できるようになると言う効果が得られる。Subsequently, the effect of the invention of claim 3 will be described. When the first winding is the self-winding coil and the second winding is the other winding, one of the ideal configurations of the low-speed mode current canceller for the self-winding coil is particularly on the dq synchronous coordinate system. One of the configurations is described by the following equation from the equation (16c).

The meaning of the acronym * in Eq. (18) is as described above. According to the low-speed mode current canceller configured based on the invention of claim 3, when the first winding is the own winding and the second winding is the other winding, the equation (18b) is included (18b). ^{An intermediate signal φ1M ~} for the own three-phase winding is generated according to the equation 6), and there is less differential equivalent processing and speed multiplication equivalent processing for the generated intermediate signal for the own three-phase winding.

The invention includes the configuration of the equation (18), and according to the invention of claim 3, the effect that the low-speed mode current canceller can adopt one of the ideal structures can be obtained. As a result, the low-speed mode current canceller has the effect of being able to exhibit preferable performance.

When the second winding is the self-winding coil and the first winding is the other winding, one of the ideal configurations of the low-speed mode current canceller for the self-winding coil is, in particular, the dq synchronous coordinate system. The above configuration is described by the following equation from the equation (17c).

"Figure showing example of winding arrangement (three-phase simple synchronous motor) of double three-phase winding permanent magnet synchronous motor" "Figure showing example of winding arrangement (six-phase synchronous motor) of double three-phase winding permanent magnet synchronous motor" "Figure showing example of winding arrangement (three-phase reverse synchronous motor) of double three-phase winding permanent magnet synchronous motor" "Figure showing one example each of single-phase mutual induction circuit and double three-phase winding permanent magnet synchronous motor virtual vector circuit" "Figure showing the relationship between three types of two-axis Cartesian coordinate systems" "A diagram showing an example of a configuration of a dual synchronous motor drive system using a high-speed mode current controller on a dq synchronous coordinate system and a low-speed mode current canceller according to the present invention." "A diagram showing an example of the configuration of a low-speed mode current canceller on a dq synchronous coordinate system according to the present invention." "A diagram showing an example of a configuration of a dual synchronous motor drive system using a low-speed mode current canceller on an αβ fixed coordinate system according to the present invention." "A diagram showing the configuration of a current control device using a conventional voltage-type interwinding non-interferer"

Hereinafter, preferred embodiments of the present invention will be specifically described with reference to the drawings.

FIG. 6 shows an example of an embodiment of a dual synchronous motor drive system using all the inventions of claims 1 to 3 with respect to the dual synchronous motor. The drive system is largely composed of a double synchronous motor (including a rotor and a stator) 1, a power conversion device 2 (broken line block display), and a current control device 3 (broken line block display). The internal configuration of the power conversion device and the internal configuration of the current control device are basically the same for the first winding and the second winding. Based on this point, these are basically described mainly for the first winding, and only when there is a difference between the first winding and the second winding, they will be described individually.

The power converter includes a power converter 21 for the first and second windings and a current detector 22. The current control device 3 is largely composed of a signal conversion unit 32 (displayed with a broken line block) responsible for converting the stator current and the stator voltage command value of the first and second windings, and a high-speed mode for controlling the high-speed mode current. It is composed of a current controller 33 (displayed as a broken line block) and a low-speed mode current canceller 34 that cancels and suppresses the low-speed mode current. Auxiliary includes a phase detector 311 for detecting the rotor phase used in the signal conversion unit 32, and a speed detector 312 for detecting the rotor speed used in the low-speed mode current canceller. The phase detector 311 and the speed detector 312 are shared by current control of both windings and the like. In the signal conversion unit 32, a three-phase two-phase converter 321a, a two-phase three-phase converter 321b, and a vector rotator are used for each winding so that the current control of the first and second windings can be performed independently. 322a and 322b are configured.

Regarding the 3-phase 2-phase converter (2 x 3 matrix) and 2-phase 3-phase converter (3 x 2 matrix) for the second winding, depending on the difference in the winding arrangement of the dual synchronous motor, Some changes are needed. That is, with respect to SR (・) in FIG. 6, the three-phase simple synchronous motor (see FIG. 1) and the three-phase reverse synchronous motor (see FIG. 3) use the following equation (20a), and the six-phase synchronous motor is used. In the case of (see FIG. 2), the following equation (20b) is used.

In the figure, for the sake of simplicity, a plurality of scalar signals are regarded as one vector signal, and a plurality of scalar signal lines are represented by one thick signal line. The voltage and current vector signals r, s, and t are signals on the dq synchronous coordinate system, signals on the αβ fixed coordinate system, and signals on the uvw coordinate system (three-phase signals), respectively. Is shown.

The signal conversion unit 32 constituting the current control device 3 is basically the same as the conventional one. Since this device is well known to those skilled in the art, further description thereof will be omitted. The core of the present invention lies in the high-speed mode current controller 33 and the low-speed mode current canceller 34 that constitute the current control device 3. Hereinafter, the high-speed mode current controller 33 and the low-speed mode current canceller 34 of FIG. 6 will be described.

The high-speed mode current controller 33 is composed of a high-speed mode current controller 331 for the first winding and a high-speed mode current controller 332 for the second winding, and voltage command values v1 ^to * for high-speed mode current control of each winding. V2 ^~ * are generated. The high-speed mode current controller 33 in FIG. 6 is configured according to the inventions of claims 1 and 2. This configuration is based on the equations (16b) and (17b) presented in the present invention. The equation shows the dynamic characteristics of the high-speed mode current dominated by the stator resistance and the leakage inductance. Basically, any structure can be adopted for the high-speed mode current controllers 331 and 332 of each winding. When a simple PI structure is adopted, the high-speed mode current controllers 331 and 332 of each winding are described as the following equations (21a) and (21b), respectively.

制御器係数を具体的に定める（２２ａ）、（２２ｂ）式には、「漏れインダクタンスの相当値」として、（σｄ^〜・Ｌ１ｄ）、（σｄ^〜・Ｌ１ｉ）を用いている点に、注意されたい。ｄ軸に関する（σｄ^〜）の選定範囲は、（２２ｃ）式が明示している。（２２ｃ）式のｄ軸の（σｄ^〜）の基準値は、ｄ軸の漏れ係数（σｄ）自体である。また、同式右辺の上限は、採用の制御器設計法と密接に関連しており、一応の目安である。（σｄ^〜）は「漏れ係数相当値」とも言うべきものである。以上の説明より明らかなように、（２２）式の制御器係数の設定に用いた漏れインダクタンス相当値（σｄ^〜・Ｌ１ｄ）、（σｄ^〜・Ｌ１ｉ）は、（２）式を用いて説明した「漏れインダクタンスの相当値」の定義にも従っている。なお、第１巻線の同相インダクタンスＬｉ１の定義は、（１３）式の通りである。第２巻線の同相インダクタンスＬｉ２も同様に定義される。In order to achieve the desired purpose "control of high-speed mode current", the controller coefficient (PI gain) in the equation (21) is the leakage inductance caused by the mutual induction of the three-phase windings according to the invention of claim 1. It will be determined using the equivalent value of. An example of the control coefficient design method using the equivalent value of the leakage inductance is shown below. The controller coefficients (PI gain) d1d1 and d1d0 of the first winding d-axis of the equation (21a) are designed as follows, for example, when the expected bandwidth of the current control system is ωic and the design parameter is w1. Will be done.

Defining a controller coefficient specifically (22a), the (22b) type, the "equivalent value of the leakage inductance", in that it uses the ^{(σd ~ · L1d), (} σd ~ · L1i), be noted sea bream. ^{The selection range of (σd to} ) with respect to the d-axis is specified by Eq. (22c). ^{The reference value of (σd to} ) on the d-axis of the equation (22c) is the leakage coefficient (σd) on the d-axis itself. In addition, the upper limit on the right side of the same formula is closely related to the adopted controller design method and is a tentative guide. (Σd ^~ ) should be called "leakage coefficient equivalent value". As apparent from the above description, (22) the leakage inductance equivalent value used for setting of the controller coefficients ^{(σd ~ · L1d), (} σd ~ · L1i) is described with reference to (2) It also follows the definition of "equivalent value of leakage inductance". The definition of the common-mode inductance Li1 of the first winding is as in Eq. (13). The common mode inductance Li2 of the second winding is also defined in the same manner.

The example of the controller coefficient design method according to the equation (22) is intended for the first equation of the equation (21a), that is, the controller coefficient of the high-speed mode current controller for the first winding d-axis. The method of designing the controller coefficient of the high-speed mode current controller for the first winding q-axis can be obtained only by changing the axis-corresponding leg to "d → q" in the equation (22). Similarly, the method of designing the controller coefficient of the high-speed mode current controller for the second winding d-axis can be obtained only by changing the winding-corresponding leg to "1 → 2" in the equation (22). The method of designing the controller coefficient of the high-speed mode current controller for the second winding q-axis is as follows: in equation (22), the winding-corresponding leg is "1 → 2" and the axis-corresponding leg is "d → q". You can get it just by changing it.

It should be pointed out that a structure other than the PI structure may be adopted as the structure of the high-speed mode current controllers 331 and 332 of each winding. In addition, it should be pointed out that there are various methods for designing the controller coefficient using the "equivalent value of leakage inductance" other than the equation (22). In the embodiment of FIG. 6, the high-speed mode current controller 33 is configured on the dq synchronous coordinate system according to the invention of claim 2. Instead, it should be pointed out that the high-speed mode current controller can be configured on the γδ quasi-synchronous coordinate system or on the αβ fixed coordinate system. The present invention does not preclude these configurations.

請求項３の発明に従い構成されている。本構成は、本発明提示の（１６ｃ）式、（１７ｃ）式に立

した。実際的には、相互インダクタンス相当値、巻線電流相当値を用いて、この生成を以下のように変更することが望まれる。

It is configured according to the invention of claim 3. This configuration is based on the equations (16c) and (17c) presented in the present invention.

bottom. In practice, it is desirable to change this generation as follows using the mutual inductance equivalent value and the winding current equivalent value.

The detailed configuration of the low-speed mode current canceller 34 in FIG. 6 is shown in FIG. 7. The low-speed mode current canceller 341 for the first winding in FIG. 7 is configured based on the equation (23), and the low-speed mode current canceller 342 for the second winding is configured based on the equation (24). However, (23), (24) a current equivalent value ⁱ¹ ~ used for expression, i2 as the ^~, has become those utilizing current true value. The definitions of the current equivalent values i1 ^to i2 ^to are as described above.

（２５）式では、相互インダクタンス相当値の上限を、対応の自己インダクタンスとしている。「相互誘導（磁気的結合）が強い場合には、漏れ係数が微小、漏れインダクタンスが微小となり、ひいては相互インダクタンスが自己インダクタンスに概ね等しくなる」と言う事実に基づいている。（２３）式の第１巻線用低速モード電流キャンセラ３４１と、（２４）式の第２巻線用低速モード電流キャンセラ３４２とにおいて、同一の相互インダクタンス相当値Ｍ１ｄ^〜、Ｍ１ｑ^〜、Ｍ２ｄ^〜、Ｍ２ｑ^〜を使用する必要はなく、各巻線用低速モード電流キャンセラごとに異なった値を利用してよい。Ｍ１ｄ^〜を例にとり、より具体的に言えば、第１巻線用低速モード電流キャンセラ３４１・（２３）式と第２巻線用低速モード電流キャンセラ３４１・（２４）式とで、Ｍ１ｄ^〜に関し異なる値を利用してよい。図７は、各巻線用低速モード電流キャンセラ３４１、３４２において、用いるべき相互インダクタンス相当値として、各巻線用低速モード電流キャンセラごとに異なった値を利用することを想定した構成としている。すなわち、図７においては、２個の中間信号φ１Ｍ^〜とφ２Ｍ^〜は必ずしも等しくないことを想定したものとなっている。２個の中間信号φ１Ｍ^〜とφ２Ｍ^〜とが等しくなるように相互インダクタンス相当値を選定する場合には、各巻線の低速モード電流相殺用の電圧

できる。 ^{The range of mutual inductance equivalent values M1d ~} , M1q ^~ , M2d ^~ , and M2q ^~ used for the low-speed mode current cancellers 341 and 342 of each winding is as follows.

In equation (25), the upper limit of the mutual inductance equivalent value is the corresponding self-inductance. It is based on the fact that "when mutual induction (magnetic coupling) is strong, the leakage coefficient is small, the leakage inductance is small, and the mutual inductance is almost equal to the self-inductance". In the low-speed mode current canceller 341 for the first winding of the formula (23) and the low-speed mode current canceller 342 for the second winding of the formula (24), the same mutual inductance equivalent values M1d ^~ , M1q ^~ , M2d ^~ , It ^{is not necessary to use M2q ~} , and different values may be used for each low speed mode current canceller for each winding. Taking M1d ^~ as an example, more specifically, the low-speed mode current canceller 341 (23) for the first winding and the low-speed mode current canceller 341 (24) for the second winding are related ^{to M1d ~.} Different values may be used. FIG. 7 has a configuration in which it is assumed that different values are used for the low-speed mode current cancellers for each winding as the mutual inductance equivalent values to be used in the low-speed mode current cancellers 341 and 342 for each winding. That is, in FIG. 7, it is assumed that the ^{two intermediate signals φ1M to} and φ2M ^{to are not necessarily equal.} When selecting the mutual inductance equivalent value so that the two intermediate signals φ1M ^~ and φ2M ^~ are equal, the voltage for canceling the low-speed mode current of each winding.

can.

近似微分器の帯域幅でもある設計パラメータωａｄは、高速モード電流制御器によって構成された電流制御系の帯域幅ωｉｃと同等あるいはそれ以上に選定することになる。当然のことながら、（２６）式以外の他の近似微分処理を採用してもよい。Fad (s) in the equations (23) and (24) is an approximate differentiator introduced to eliminate the pure differentiation, and may be simply as follows.

The design parameter ωad, which is also the bandwidth of the approximate differentiator, is selected to be equal to or greater than the bandwidth ωic of the current control system configured by the high-speed mode current controller. As a matter of course, other approximate differential processing other than the equation (26) may be adopted.

In the embodiment using FIG. 7, the same true value (current detection value) is used as the equivalent value of the currents of the first winding and the second winding, and the same true value (speed detection value) is used as the speed equivalent value. It is an example. It should be pointed out again that both signals may be replaced with equivalent values. The definitions of current equivalent value and velocity equivalent value are as already given.

In the example of the embodiment of FIG. 7, it is an example of performing both the differentiation equivalent processing and the speed multiplication equivalent processing. It should be pointed out that the speed multiplication equivalent process becomes more important at high speed rotation, and the differential equivalent process can be omitted. On the contrary, it should be pointed out that the speed multiplication equivalent processing becomes relatively less important at low speed rotation, and thus the speed multiplication equivalent processing can be omitted.

In the embodiment of FIG. 7, the low-speed mode current canceller 34 is configured on the dq synchronous coordinate system. It should be pointed out that a low-speed mode current canceller substantially the same as in FIG. 7 can be configured on the γδ quasi-synchronous coordinate system.

It should also be pointed out that the low speed mode current canceller 34 can also be configured on the αβ fixed coordinate system. FIG. 8 shows this situation. When configuring the low-speed mode current canceller on the αβ fixed coordinate system, it is generally necessary to slightly change the configuration of FIG. 7. However, when the mutual inductance equivalent values of the d-axis and the q-axis are selected equally, no additional change is necessary, and the one of FIG. 7 in which the speed multiplication equivalent process is simply removed can be used.

The present invention relates to a dual synchronous motor for a wide range of applications requiring efficient driving, such as a battery electric vehicle, a fuel cell electric vehicle, a main drive motor of a hybrid electric vehicle, and a high-speed motor for home appliances. It is suitable for driving systems of dual synchronous motors in applications that require functional safety.

1 Double synchronous motor 11 Rotor of double synchronous motor 121 First winding of stator of double synchronous motor 122 Second winding of stator of double synchronous motor 2 Power converter 21 Power converter 22 Current detection Instrument 3 Current controller 311 Phase detector 312 Speed detector 32 Signal converter 321a Three-phase two-phase converter 321b Two-phase three-phase converter 322a Vector rotator 322b Vector rotator 33 High-speed mode current controller 331 First winding High-speed mode current controller 332 High-speed mode current controller for 2nd winding 34 Low-speed mode current canceller 341 Low-speed mode current canceller for 1st winding 342 Low-speed mode current canceller for 2nd winding

Claims (3)

A permanent magnet synchronous motor consisting of a rotor with a permanent magnet and a stator with two three-phase windings (self-three-phase winding and other three-phase winding),
A power converter that can supply current to two three-phase windings at the same time,
A permanent magnet synchronous motor drive system including a current control device that controls a current flowing through two three-phase windings via a power conversion device.
It has a controller coefficient determined by using at least the equivalent value of the leakage inductance caused by the mutual induction of the two three-phase windings, and controls the high-speed mode current generated by the mutual induction of the two three-phase windings. High-speed mode current controller for each three-phase winding, which produces a voltage command value for
The low-speed mode current generated by the mutual induction of the two three-phase windings using at least the equivalent value of the mutual inductance caused by the mutual induction of the two three-phase windings and the current-equal value of the own three-phase windings. With a low speed mode current canceller for each three-phase winding that generates a voltage command value for its own three-phase winding for offsetting
A permanent magnet synchronous motor drive system characterized in that the current control device is configured by using at least one of two different devices (high-speed mode current controller and low-speed mode current canceller). The phase of the N pole of the permanent magnet of the rotor is defined as the phase of the d-axis, and the 2-axis Cartesian coordinate system having the q-axis with a phase advance of π / 2 [rad] with respect to the d-axis is defined as the dq synchronous coordinate system. When the coordinate system based on the synchronous coordinate system is the γδ quasi-synchronous coordinate system,
The current deviation of each three-phase winding, which is the difference between the current command value of each three-phase winding on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system and the current response value of each three-phase winding, is used as feedback control. The feature is that the high-speed mode current controller for each three-phase winding is configured so as to process and generate the voltage command value of each three-phase winding on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system. The permanent magnet synchronous motor drive system according to claim 1. The phase of the N pole of the permanent magnet of the rotor is defined as the phase of the d-axis, and the 2-axis Cartesian coordinate system having the q-axis with a phase advance of π / 2 [rad] with respect to the d-axis is defined as the dq synchronous coordinate system. When the coordinate system based on the synchronous coordinate system is the γδ quasi-synchronous coordinate system,
Further, the current equivalent value of the self-three-phase winding on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system is i1 ^~, and the current equivalent value of the other three-phase winding is i2 ^~, and two (self and). When the equivalent values of the mutual inductance caused by the mutual induction of the three-phase windings of (others) are M1d ^~ , M1q ^~ , M2d ^~ , M2q ^~
Generates ^{an intermediate signal φ1M ~} for own three-phase winding according to the following equation.

The generated intermediate signal for the three-phase winding is subjected to at least one of the differentiation equivalent processing and the speed multiplication equivalent processing to generate the voltage command value for the own three-phase winding.
The permanent magnet synchronous motor drive system according to claim 1, wherein a low-speed mode current canceller for each of the three-phase windings is configured.

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