JP7045962B2 - AC motor control device and its control method - Google Patents

Description

The present invention relates to a vector control device and an AC motor control system that control an AC motor based on a harmonic current generated by superimposing a harmonic voltage on the AC motor.

The drive system for AC motors (hereinafter referred to as "motors") is used for industrial applications such as machine tools, fans, pumps (hydraulic pumps, water pumps), compressors, and heating and cooling equipment, as well as AC motors for electric vehicles and railway vehicles. It is used for various purposes such as, and among them, electric motor drive devices using permanent magnet synchronous motors, which have the features of small size and high efficiency, are expanding.

In order to drive a permanent magnet synchronous motor, it is necessary to accurately detect the rotor position and speed of the motor. In recent motor drive devices, sensorless control that estimates the rotor position and speed from the counter electromotive voltage information generated in the motor without using a position sensor or speed sensor is adopted. However, when estimating the rotor position from the counter electromotive voltage information, it is difficult to apply because the counter electromotive voltage becomes small near the extremely low speed. Therefore, as a method of estimating the rotor position at low speed, there is a method using the salient pole ratio of the motor.

Patent Document 1 and Patent Document 2 are background techniques in this technical field. Patent Document 1 discloses a conventional technique for estimating a rotor position by using a salient pole ratio of an electric motor. Here, in order to improve the estimation accuracy of the rotor position, a technique for reducing the influence of the detection error of the harmonic current of the motor is disclosed. The accuracy is ensured by setting the period of the harmonic current longer than the current sampling period and increasing the number of detections in one cycle of the harmonic current.

Further, in the prior art of Patent Document 2, in order to accurately estimate the stop / initial position of the permanent magnet synchronous motor, the harmonic current of the electric motor is set to at least 2 of the harmonic current for each of the positive side and the negative side. Two current values are detected, and the initial position of the rotor is estimated from the amount of change in the harmonic current.

Japanese Unexamined Patent Publication No. 2012-161143 Japanese Unexamined Patent Publication No. 2005-02018

In Patent Document 1, the period of the harmonic current is set sufficiently longer than the number of samplings to increase the number of detections of the harmonic current and improve the estimation accuracy of the rotor position. However, since the detection timing of the harmonic current is synchronized with the triangular wave carrier for performing PWM (pulse width modulation) of the controller, the number of detections is limited by the switching frequency of the power semiconductor element. Therefore, in order to realize this method, it is necessary to set the frequency of the harmonics superimposed on the switching frequency sufficiently low. Superimposing low-frequency harmonics may cause jarring electromagnetic noise, torque pulsation, and associated mechanical vibration, which is not realistic. Further, by lowering the frequency of the harmonics, the responsiveness of the essential sensorless portion is lowered, and it may be difficult to increase the responsiveness as a system.

Furthermore, in recent years, there has been an increasing tendency in the industrial field to apply low-speed position sensorless control to inexpensive motors. Since many inexpensive motors have a low salient pole ratio, there is a problem that the estimation accuracy of the rotor position is lowered. In order to drive a motor with a low salient pole ratio sensorlessly by the method of Patent Document 1, it is necessary to further lower the frequency of harmonics or increase the amplitude to improve the sensitivity, and electromagnetic noise and torque pulsation are increasing. Problem occurs. In addition, it becomes difficult to improve the control response.

Since the method of Patent Document 2 is basically a task process of initial position estimation before execution of vector control, position estimation in a state where an electric motor is driven, that is, a state at the time of vector control is not disclosed. However, as a method for detecting the amount of change in the harmonic current, a method for setting the peak value and the intermediate value of the triangular wave carrier for performing PWM (pulse width modulation) as the detection timing is disclosed. However, it is difficult to significantly improve the detection accuracy because it is easily affected by noise and error at the two timings of the peak and the intermediate value of the triangular wave carrier.

The present invention has been made in view of such a problem, and is a control device for an AC motor capable of accurately, quietly, and versatilely estimating the rotor position in the rotor position sensorless control by harmonic superimposition. The purpose is to provide.

In view of the above background technology and problems, the present invention, for example, uses a power conversion device for driving an AC motor and a power conversion device to apply a harmonic component to the AC motor to move the AC motor without a position sensor. It is a control method of an AC motor equipped with a controller that performs vector control. The processing of vector control and the processing of current detection of the AC motor are started with different trigger signals, and are included in the current detected by the current detection processing. Vector control is performed without a position sensor using the harmonic component.

INDUSTRIAL APPLICABILITY According to the present invention, it is possible to provide a control device for an AC motor and a control method thereof that can estimate the rotor position accurately, quietly, and universally in the position sensorless control of the rotor by superimposing harmonics.

It is a block diagram of the control device of the AC motor in Example 1. FIG. It is a block diagram of the vector control processor in Example 1. FIG. It is a block diagram of the current detection processing apparatus in Example 1. FIG. It is a figure which shows the relationship between the actual rotation coordinate axis (d-q axis) of an AC motor, the rotation coordinate axis (dc-qc axis) inside a controller, and the axis error Δθc. It is explanatory drawing of the operation of the controller in Example 1. FIG. It is a block diagram of the filter processing apparatus in the current detection processing apparatus in Example 1. FIG. It is a block diagram of the ΔIq arithmetic unit in the current detection processing apparatus in Example 1. FIG. It is explanatory drawing of the operation of the interrupt signal generator in Example 2. FIG. It is a block diagram of the interrupt signal generator in Example 2. FIG. It is a block diagram of the ΔIq arithmetic unit in the current detection processing apparatus in Example 3. FIG. It is explanatory drawing of the operation of the ΔIq arithmetic unit in Example 3. FIG. It is a block diagram of the current detection processing apparatus in Example 4. FIG. It is a block diagram of the polarity discriminator in the current detection processing apparatus in Example 4. FIG. It is explanatory drawing of the operation of the polarity discriminator in Example 4. FIG. It is a block diagram of the control device of the AC motor in Example 5. FIG.

Hereinafter, examples of the present invention will be described with reference to the drawings.

The control device of the AC motor in this embodiment will be described with reference to FIGS. 1 to 7.

FIG. 1 is a configuration diagram of a control device for an AC motor in this embodiment. In FIG. 1, the control device 20 is composed of a power converter 3 and a controller 4. The power supply 1 is a DC power supply that supplies electric power to a power converter 3 for driving a permanent magnet synchronous motor 2 (hereinafter, abbreviated as PM motor). The power converter 3 includes an inverter main circuit unit 31 composed of six power semiconductor elements, a gate driver 32 that generates on / off signals for each power semiconductor element of the inverter main circuit unit 31, and a filter capacitor 33. , The current sensors 8a and 8b for detecting the phase current flowing through the PM motor 2, and the power converter 3 is controlled based on the PWM signal (pulse width modulation signal) output by the controller 4.

In the controller 4, the vector control processor 6 starts the control process by the trigger signal TRG1 output from the interrupt signal generator 5, and the phase currents Iu and Iw detected by the current sensors 8a and 8b become desired values. As described above, the output voltage command of the power converter 3 is calculated. Finally, a PWM signal is generated and output to the gate driver 32.

On the other hand, in the current detection processor 7, arithmetic processing is started by the trigger signal TRG2 output by the interrupt signal generator 5, Iu and Iw detected by the current sensors 8a and 8b are sampled, and the vector control processor 6 is used. The required currents Idm and Iqm are calculated, and the axis error Δθc required for driving the PM motor 2 without a rotation position sensor is performed.

Next, the operation of the vector control processor 6 will be described with reference to FIG. As shown in FIG. 2, the vector control processor 6 starts the control process by the trigger signal TRG1, and the speed command generator 61 that generates the rotation speed command of the PM motor 2 generates the speed command ωr *. The speed controller 62 operates so that the rotation speed of the PM motor 2 (here, sensorless control is assumed and is an estimated rotation speed) ωrc ^ matches, and the current command value id *, on the dq conversion. iq * is calculated and output. In the vector control calculator 63, the voltage commands vdc * and vqc * are calculated so that there is no deviation between the current command values id * and iq * and the currents Idm and Iqm of the PM motor 2. The Δvdc * and Δvqc * output by the harmonic voltage generator 67 are added to these values, and the dq inverse converter 64 converts them into three-phase AC voltage commands vu *, vv *, vw *. These three-phase AC voltage commands are pulse-width modulated by the PWM signal generator 66 and output as PWM signals Pu, Pv, Pw. The harmonic voltage generator 67 generates a harmonic voltage required for position sensorless control of the PM motor 2, and at the same time generates a code signal Pvh necessary for position estimation, and is a current detection processor. It is output to 7.

Since this system is premised on rotor position sensorless control of PM motor 2, there is an error between the actual rotor position θd and the rotor position θdc assumed inside the controller. It is defined as the axis error Δθc.
Δθc = θdc－θd… (1)

This Δθc is calculated by the current detection processor 7 and output to the vector control processor 6. In the vector control processor 6, in the speed estimator 68, the rotation speed ωrc ^ is modified so that this Δθc becomes zero, so that both speed estimation and position control are compatible. The phase calculation is performed using ωrc ^, and the phase θdc is used as the phase of the dq inverse converter 64. Further, the obtained θdc is also used for coordinate conversion in the current detection processor 7.

Next, the current detection processor 7, which is a feature of the present invention, will be described in detail with reference to FIGS. 3 to 7. FIG. 3 is a block diagram of the current detection processor in this embodiment. As shown in FIG. 3, the current detection processor 7 starts arithmetic processing by the trigger signal TRG2, and uses the current detection values Iu and Iw of the PM motor 2 to dq with respect to the coordinates of θdc by the dq coordinate converter 71. Coordinate conversion is performed, and the phase currents Iu and Iw are converted into the currents Idc and Iqc on the dq axis. Idc and Iqc contain a large amount of harmonic components generated by the harmonic voltage generator 67 of the vector control processor 6, and are filtered by the filter processor 72 in order to cut them. Output Idm and Iqm.

The ΔIq calculator 9 calculates ΔIq, which is a component that changes according to the amount of occurrence of the axis error Δθc, from the harmonic components contained in Iqc. The axis error calculator 73 calculates and outputs the axis error Δθc by multiplying this ΔIq by a coefficient.

Here, the relationship between the axis error Δθc and ΔIq will be described with reference to FIG. This part is a known technique.

FIGS. 4A to 4C show the actual dq coordinate axis dq axis of the PM motor 2 and the dc-qc axis which is the internal coordinate axis of the controller 4, respectively, both of which have an axis error Δθc. This is an example.

As shown in FIG. 4A, a harmonic voltage Vh is applied to the dc axis, which is the coordinate axis on the control side. That is, in the harmonic voltage generator 67 in FIG. 2,
Δvdc * = Vh, Δvqc * = 0… (2)
It is supposed to be. The harmonic voltage Vh on the dc axis is vector-decomposed into the actual dq coordinate axes of the PM motor 2, and harmonic currents Ihd and Ihq are generated in each axis.

At this time, as shown in FIG. 4B, when the d-axis inductance Ld and the q-axis inductance Lq are equal, Ihd and Ihq are generated according to the vector ratio of the harmonic voltage. Under such conditions, the axis error Δθc cannot be estimated. However, when the inductance of the dq axis is different, the amount of current generated is different. As shown in FIG. 4C, a large amount of current flows in the d-axis having a small inductance, and conversely, a large amount of current does not flow in the q-axis having a large inductance. As a result, the harmonic current Ih tilts in the direction in which the d-axis exists, and the harmonic component Iqhc is generated on the qc axis. This amount of generation is related to the axis error Δθc, and if the phase is corrected so that this Iqch becomes zero, position sensorless control is established.

These operations will be described using a waveform diagram. In FIG. 5, FIG. 5A shows a triangular wave carrier for PWM generation used in the PWM signal generator 66 and a phase voltage command. A PWM signal is generated by comparing the triangular wave carrier wave with the voltage command. At that time, the interrupt signal TRG1 for the vector control processor 6 is generated at the peak time of the peak and valley of the triangular wave carrier. Vector control processing is performed using this signal TRG1 as a trigger (FIG. 5 (d)). Further, as shown in FIG. 3E, the harmonic voltage is applied as a waveform synchronized with the peaks and valleys of the triangular wave carrier. In the PWM method using a triangular wave carrier, it is not possible to apply a signal higher than the triangular wave carrier frequency, so the application method as shown in FIG. 5 (e) has the maximum superimposed frequency. A code signal Pvh of the harmonic voltage is also generated in synchronization with this superimposed voltage waveform and is given to the current detection process 7.

By applying harmonics, the q-axis current Iqc includes Iqhc, which is a harmonic component (FIG. 5 (g)). FIG. 5 (g) shows the timing of conventional current sampling, sampling is performed at the timing of carrier waves and peaks and valleys, and sensorless control is performed based on the difference (ΔIq) between them. However, the timing of current detection is only at the moment of the peak or valley of the carrier, and it is easily affected by noise and detection error. Considering the S / N ratio of detection, it is necessary to apply a harmonic voltage with a large amplitude in order to obtain ΔIq with sufficient accuracy, in which case the electromagnetic noise increases and the loss due to the harmonics increases. There is also concern about the change.

In this embodiment, the current detection processor 7 is operated with a shorter cycle than the vector control processor 6 to solve this problem. FIG. 5 (h) is the trigger signal TRG2 in the high-speed sampling, and the current waveform is sampled as shown in FIG. 5 (i).

As shown in FIG. 3, the high-speed sampled phase current is subjected to arithmetic processing by the filter processor 72 and the ΔIq arithmetic unit 9 after the coordinate conversion.

FIG. 6 is a configuration diagram of the filter processor 72 in FIG. 3, where the delay device 10, the adder 11, and the calculation gain 12 are used to perform filtering by moving average processing. The delay device 10 uses "2N-1" pieces, and performs moving average processing for 2N pieces. Here, N is the number of samples during the half cycle of the harmonic voltage Vh (here, it corresponds to T1). By taking 2N moving averages, the influence of harmonic components can be eliminated.

FIG. 7 is a detailed block of the ΔIq calculator 9 in FIG. In FIG. 7, the filter processor 91 removes the noise contained in Iqc, and then the difference calculator 92 calculates the first-order difference of the waveform. This difference calculation is a calculation corresponding to the "difference calculation at the timing of the peak and valley of the carrier wave" shown in FIG. 5 (g), and this result and the code signal Pvh synchronized with the polarity of the harmonic voltage are combined. By multiplying by the multiplier 13, ΔIq is calculated. At this time, in consideration of the signal delay in the filter processor 91, the delay device 93 is inserted with respect to the code signal Pvh. Further, since there is a possibility that a fluctuation amount remains in ΔIq, the calculation result is smoothed by the low-pass filter (LPF) 14.

The filter processor 91 in FIG. 7 performs moving average processing for K pieces, but if the value of K is too large, the harmonic component disappears completely, so movements for several times for noise reduction are performed. It is desirable to use an average. Alternatively, there is no problem even if it is a bandpass filter by digital filtering.

In this embodiment, in position sensorless control by applying harmonics, it is possible to introduce a short sample period different from vector control processing to detect current and perform position estimation, which is more resistant to noise and more resistant to position than before. Harmonic voltage can be reduced.

In this embodiment, the harmonic voltage is applied only to the dc axis, but there is no problem even if it is applied only to the qc axis or to both axes. Further, although a harmonic of a frequency equal to the triangular wave carrier frequency is applied, this can also be realized if it is below the carrier frequency, and in that case, a harmonic of a sine wave can be applied instead of a square wave. ..

As described above, according to the present embodiment, in the control device of the AC motor, even a minute current can be detected by detecting the current in a cycle finer than the vector control cycle. Therefore, it is possible to reduce the amount of superimposed voltage and reduce the electromagnetic noise and torque pulsation generated in the motor. Further, by applying the present invention to an electric motor having a low salient pole ratio, the rotor position can be estimated accurately and position sensorless control can be performed from a low speed. Further, since the response of the control system can be set to the same frequency as the frequency of the harmonic current and the frequency of the PWM triangular wave carrier, it is possible to increase the response of the control system.

The control device for the AC motor in this embodiment will be described with reference to FIGS. 8 and 9.

FIG. 8 is an explanatory diagram of the operation of the interrupt signal generator in this embodiment. In FIG. 8, TRG1, TRG2, and Iu are the trigger signal and the phase current in FIG. The trigger signal TRG1 is a signal synchronized with the triangular wave carrier as described above. Therefore, T1 which is the cycle of the trigger signal TRG1 is the cycle in which the inverter main circuit operates for switching, and is accompanied by at least one on or off operation within the period of T1. As a result, the pulsating component contained in the phase current Iu becomes a component of the cycle with reference to this T1.

Since the pulsating component due to the harmonic voltage shown in Example 1 is also added to this pulsating component, it is desired to detect it as accurately as possible. For that purpose, it is desirable to synchronize the period T1 so that the period T2 that generates the trigger signal TRG2 is always an integer number. If this synchronization is not achieved, there is a risk of "waviness (beat)" due to sampling. Therefore, in order to prevent a detection error due to undulation, the trigger signals TRG1 and TRG2 are generated so that an integral multiple of T2 becomes T1.

FIG. 9 shows the configuration of the interrupt signal generator 5 that generates the trigger signal under such conditions. In FIG. 9, the transmitter 51 generates the signal of TRG2 in the period T2, and the signal is counted by the counter 52 to generate the signal of the period T1. By setting the counter 52, it is possible to set how many times T1 is multiplied by T2.

As described above, according to the present embodiment, it is possible to realize a control device for an AC motor capable of sensing the rotor position with higher accuracy.

In this embodiment, the rotation position sensorless control is performed by extracting a frequency component preset with respect to the detected value of the harmonic current and extracting the amplitude of the harmonic component.

In Example 1, a means for reducing noise by filtering by increasing the number of times of current sampling and improving the position estimation accuracy in a position sensorless manner has been described. However, the position estimation calculation requires calculation of the "difference" with respect to the current value after noise removal, and the resolution at the time of current detection is important. That is, there is a problem that accuracy cannot be obtained unless a high-resolution AD converter is used as the bit number of the AD converter used for the controller 4. In order to solve this problem, a method that does not use the difference operation will be described in this embodiment.

The control device for the AC motor in this embodiment will be described with reference to FIGS. 10 and 11.

This embodiment can be realized by replacing the ΔIq calculator 9 shown in FIG. 7 in the first embodiment with the ΔIq calculator 9B shown in FIG.

FIG. 10 shows the configuration of the ΔIq calculator 9B in this embodiment. The ΔIq calculator 9B includes a sine wave generator 94 that generates a sine wave synchronized with the harmonic voltage code Pvh, a multiplier 13 and a filter processor 91B. Here, the q-axis current Iqc is input, and the amplitude component ΔIq2 of the harmonic current Iqhc included in the q-axis current Iqc is output.

FIG. 11 shows the operation of the ΔIq calculator 9B of FIG. Synchronized with the harmonic voltage code Pvh in FIG. 11 (a), the sine wave generator 94 generates a sine wave function S (FIG. 11 (b)) equal to the frequency of the harmonic, and the harmonic current Iqhc is multiplied by the multiplier 14. Multiply by. Here, by setting the magnitude of the sine wave function S to 1 and setting the phase so as to be in phase with the harmonic current Iqhc, the amplitude component of the harmonic current Iqhc can be extracted. This phase may be set so that it becomes the peak of the sine wave at the falling timing of the harmonic voltage code Pvh (the PM motor 2 can be regarded as an inductance with respect to the harmonic voltage, and its current). Because the phase is delayed by 90 degrees).

The multiplication result of the sine wave function S and the harmonic current Iqhc is shown in the waveform of FIG. 11 (d). It can be seen that the amplitude component of the harmonic current Iqhc is extracted from the waveform shown in FIG. 11 (d), but since the frequency component twice the harmonic is included, the frequency component doubled by the filter processor 91B. To remove. In the filter processor 91B, the number of moving averages that can sufficiently remove the double frequency component is performed, and the amplitude component ΔIq2 of the harmonic current Iqhc corresponding to the axis error Δθc is calculated. The moving average period of the filter processor 91B is matched with the half cycle of the harmonic voltage Vh (the cycle of T1).

As described above, according to this embodiment, it is possible to extract the frequency component preset with respect to the detected value of the harmonic current to obtain the amplitude of the harmonic component, and further, the difference calculation is used. By not doing so, it is possible to perform highly accurate rotation position sensorless control even when an AD converter having a low resolution is used.

In this embodiment, for the detected value of the harmonic current, the frequency component that is the main component of the harmonic current is extracted and the rotation position sensorless control is performed, and the frequency component that is twice that is extracted. The polarity determination process of the rotor position of the PM motor is performed.

In the estimation calculation of the axis error Δθc in the first embodiment, the position can be estimated only in the range where the axis error Δθc is ± 90 degrees. However, in the state of the initial position of the rotor, the axis error Δθc is not always within the range of ± 90 degrees, and it is conceivable that the axis error Δθc converges to either 0 degrees or 180 degrees. That is, to explain with the coordinate axes shown in FIG. 4, when the axis error Δθc is 0 degrees, it means that the dc axis coincides with the d axis. On the other hand, when the axis error Δθc is 180 degrees, it means that the dc axis is inverted 180 degrees from the d axis. A polarity discrimination process is performed to discriminate between these two points. The frequency component extraction operation in this embodiment is performed in the same manner as in the third embodiment. Hereinafter, the operation of the polarity discrimination process will be described.

The control device for the AC motor in this embodiment will be described with reference to FIGS. 12, 13, and 14.

This embodiment can be realized by replacing the current detection processor 7 shown in FIG. 3 in the first embodiment with the current detection processor 7C shown in FIG.

FIG. 12 shows the configuration of the current detection processor 7C including the polarity discrimination process in this embodiment. The current detection processor 7C is composed of a dq coordinate converter 71, a filter processor 72, a ΔIq calculator 9, an axis error calculator 73, and a polarity discriminator 15. The current detection processor 7C has the same configuration and operation as that of the first embodiment except for the polarity discriminator 15.

FIG. 13 shows the configuration of the polarity discriminator 15. The polarity discriminator 15 is composed of a secondary harmonic generator 95, a multiplier 13, and a filter processor 91C for moving average. Here, the d-axis current Idc is used as an input, the polarity is discriminated using the Idc, and the polarity discrimination signal NS is output.

FIG. 14 shows the operation of the polarity discriminator 15 of FIG. Synchronized with the harmonic voltage code Pvh of FIG. 14 (a), the secondary wave generator 95 generates a sinusoidal function S2 (FIG. 14 (b)) having a frequency twice the harmonic. The peak value of the sine wave function S2 is synchronized with the falling timing of the harmonic voltage sign Pvh. The sine wave function S2 is multiplied by the harmonic current Idhc of the d-axis current with the multiplier 14 to obtain the waveform shown in FIG. 14 (d). The waveform shown in FIG. 14 (d) is subjected to the number of moving averages capable of sufficiently removing the harmonic components by the filter processor 91C, the average value shown in FIG. 14 (e) is calculated as the discrimination signal NS, and the average value is calculated. The polarity is determined by the sign of. The period of the moving average here is one cycle of Vh (twice T1).

The principle of polarity discrimination performed here is known, but the principle will be briefly explained as follows. The polarity is discriminated by using the magnetic saturation characteristic of the magnetic flux of the magnet of the PM motor. When a harmonic voltage Vh is applied to the dc axis, which is the coordinate axis on the control side, the harmonic current Idhc of the d-axis current Idc is affected by the magnetic saturation of the magnetic flux, and the positive and negative are asymmetric as shown in FIG. 14 (c). It becomes a waveform. Polarity discrimination is performed using the second harmonic component included in this asymmetric waveform. In order to extract the component of the second harmonic of Idhc, it is multiplied by the sine wave function S2 having a frequency twice the harmonic to obtain the waveform shown in FIG. 14 (d). By moving averaging the waveform of FIG. 14 (d), the average value of the required frequency can be extracted, and the discrimination signal NS shown in FIG. 14 (e) is obtained. Here, if the discrimination signal NS is positive, the axis error Δθc is 0 degrees (the dc axis coincides with the d axis), and if the discrimination signal NS is negative, the axis error Δθc is 180 degrees (the dc axis is). It is inverted 180 degrees from the d-axis).

As described above, according to this embodiment, the frequency component that is the main component of the harmonic current is extracted and the frequency component of the second harmonic of the harmonic current is extracted from the detected value of the harmonic current. This makes it possible to realize a control device for an AC electric motor capable of controlling the rotor position sensorlessly and discriminating the polarity.

In this embodiment, the function of freely setting the cycle T1 for performing vector control and the processing cycle T2 for current detection in the first embodiment is applied.

FIG. 15 is a configuration diagram of a control device for an AC motor in this embodiment. This embodiment is applicable to Examples 1 to 4, and the configuration thereof is realized by replacing the controller 4 in FIG. 1 with the controller 4b shown in FIG. 15 and using the same components in other cases. can.

In FIG. 15, the difference from FIG. 1 is that the controller 4b adds the communication means 16 so that the controller 4b can communicate with other devices connected to the controller 4 of the first to fourth embodiments. Here, the processing cycle T1 for vector control and the processing cycle T2 for current detection in this embodiment can be set via the communication means 16.

The communication means 16 is connected to, for example, a local area network on a personal computer 17 or a fieldbus of another connected device to perform wired communication, and can freely set the processing cycle T1 and the processing cycle T2. It may be set. Alternatively, for example, the processing cycle T1 and the processing cycle T2 may be set by communicating with another device connected wirelessly.

As described above, according to the present embodiment, the processing cycle T1 for vector control and the processing cycle T2 for current detection can be freely set by wired communication or wireless communication.

Although the examples have been described above, the present invention is not limited to the above-mentioned examples, and includes various modifications. For example, the above-described embodiment has been described in detail in order to explain the present invention in an easy-to-understand manner, and is not necessarily limited to the one including all the described configurations. Further, it is possible to replace a part of the configuration of one embodiment with the configuration of another embodiment, and it is also possible to add the configuration of another embodiment to the configuration of one embodiment. It is also possible to add, delete, or replace a part of the configuration of each embodiment with another configuration. Further, the controller may realize a part or all of them by hardware, for example, by designing an integrated circuit, or the processor interprets and executes a program that realizes each function. It may be realized by software by.

1: Power supply, 2: Permanent magnet synchronous motor (PM motor), 3: Power converter, 4, 4b: Controller, 5: Interruption signal generator, 6: Vector control processor, 7, 7c: Current detection processor , 8a, 8b: Current sensor, 9: ΔIq calculator, 10: Delayer, 11: Adder, 12: Computation gain, 13: Multiplier, 14: Low-pass filter, 15: Polarity discriminator, 16: Communication means, 17: Personal computer, 20: Control device, 31: Inverter main circuit section, 32: Gate driver, 33: Filter capacitor, 51: Transmitter, 52: Counter, 71: dq coordinate converter, 72: Filter processor 73: Axis error calculator, 721a: dc axis current Idc filter processor, 721b: qc axis current Iqc filter processor, 9B: ΔIq calculator, 91B: filter processor (Embodiment 3), 91C: filter Processor (inside polarity discriminator), 94: sine wave generator, 95: secondary wave control generator

Claims (8)

A control device for an AC motor equipped with a power conversion device that drives the AC motor and a controller that applies a harmonic component to the AC motor using the power conversion device and controls the AC motor in a vector without a position sensor. There,
The controller includes a vector control processor that performs vector control processing, a current detection processing device that performs current detection processing of the AC motor, a trigger TRG1 of the vector control processing cycle T1, and a current of the AC motor. It has an interrupt signal generator that outputs different trigger signals for the trigger TRG2 of the detection processing cycle T2 .
The processing cycle T1 of the vector control is a cycle synchronized with a cycle of outputting a switching command of the power semiconductor element of the power conversion device.
The current detection process of the process cycle T2 is executed twice or more within the process cycle T1 period, and the current detection process is executed twice or more.
A control device for an AC motor, which performs vector control by the position sensorless using the harmonic component included in the current detected by the processing of the current detection processor. The control device for an AC motor according to claim 1.
The AC motor control device, wherein the vector control processing cycle T1 is an integral multiple of the current detection processing cycle T2 of the AC motor. The control device for an AC motor according to claim 1 .
The current detection processor is a control device for an AC motor, characterized in that the detected value of the harmonic component is subjected to processing including a difference from a past detected value and filtering . A control device for an AC motor equipped with a power conversion device that drives the AC motor and a controller that applies a harmonic component to the AC motor using the power conversion device and controls the AC motor in a vector without a position sensor . There,
The controller starts processing of the vector control processor that performs the vector control processing, the current detection processing device that performs the current detection processing of the AC motor, and the vector control processing device and the current detection processing device. Each has an interrupt signal generator that outputs different trigger signals.
Using the harmonic component included in the current detected by the processing of the current detection processor, vector control is performed without the position sensor.
The current detection processor is a control device for an AC motor, characterized in that a peak value calculation or a preset frequency component extraction calculation is performed on the detected value of the harmonic component . A control device for an AC motor equipped with a power conversion device that drives the AC motor and a controller that applies a harmonic component to the AC motor using the power conversion device and controls the AC motor in a vector without a position sensor . There,
The controller starts processing of the vector control processor that performs the vector control processing, the current detection processing device that performs the current detection processing of the AC motor, and the vector control processing device and the current detection processing device. Each has an interrupt signal generator that outputs a different trigger signal.
Using the harmonic component included in the current detected by the processing of the current detection processor, vector control is performed without the position sensor.
The current detection processor is characterized in that it performs an extraction calculation of a frequency component which is a main component of the harmonic component and an extraction calculation process of twice the frequency component of the detected value of the harmonic component. AC motor control device. The control device for an AC motor according to claim 1.
A control device for an AC motor, characterized by having a function of freely setting the processing cycle T1 and the processing cycle T2 . The control device for an AC motor according to claim 6 .
A control device for an AC motor, wherein the processing cycle T1 and the processing cycle T2 are set by another device connected to the control device of the AC motor by wired communication or wireless communication . A control method for an AC motor including a power converter that drives the AC motor and a controller that applies harmonic components to the AC motor using the power converter and controls the AC motor in a vector without a position sensor. There,
The vector control processing and the current detection processing of the AC motor are processed by different trigger signals of the trigger TRG1 of the vector control processing cycle T1 and the trigger TRG2 of the current detection processing cycle T2 of the AC motor.
The processing cycle T1 of the vector control is a cycle synchronized with a cycle of outputting a switching command of the power semiconductor element of the power conversion device.
The current detection process of the process cycle T2 is executed twice or more within the process cycle T1 period, and the current detection process is executed twice or more.
A control method for an AC motor, which comprises performing vector control by the position sensorless using the harmonic component included in the current detected by the current detection process .

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