JP7308949B2 - Motor drive device and refrigeration cycle application equipment - Google Patents

Description

TECHNICAL FIELD The present invention relates to an electric motor drive device and a refrigerating cycle application device provided with the same.

Refrigerating cycle equipment includes a compressor for compressing a refrigerant. The compressor is driven by an electric motor.
Metal refrigerant pipes are connected to the compressor for sucking in and discharging refrigerant. Mechanical vibration occurs when the operation of the compressor is stopped. When vibration occurs in the compressor, the vibration is transmitted to the pipes, which may cause metal fatigue and break the pipes. Therefore, conventionally, there has been proposed a control for suppressing the vibration when the operation of the compressor is stopped. For example, Patent Literature 1 describes turning off the output of the inverter and turning off the output torque of the electric motor in a phase in which the vibration when the compressor is stopped is reduced.

Patent No. 5094256 (claim 2, paragraph 0024)

Although Patent Document 1 exemplifies a crank angle of 60 to 300 degrees as a phase at which vibration decreases, there are cases where vibration is not sufficiently suppressed even if the phase is stopped within the exemplified range.
A similar problem occurs when the electric motor drives something other than the compressor.

SUMMARY OF THE INVENTION It is an object of the present invention to provide an electric motor drive device capable of sufficiently suppressing vibration of a load element when stopped.

An electric motor drive device according to the present invention includes:
an electric motor that drives a load element whose load torque varies periodically;
an inverter that applies an alternating voltage with a variable frequency and voltage value to the electric motor;
and a control device that controls the inverter,
The control device is
controlling the inverter so that the output torque of the electric motor follows periodic fluctuations of the load torque;
When stopping the electric motor, stopping the inverter during a period including a torque minimum phase in which the load torque becomes a minimum value or a value close to the minimum value,
The control device generates a torque current compensation value for causing the output torque of the electric motor to follow periodic fluctuations in the load torque, corrects the torque current command value based on the generated torque current compensation value, and corrects generating a voltage command value based on the generated torque current command value, controlling the inverter based on the generated voltage command value;
The control device calculates the output torque of the electric motor, uses the calculated output torque as an estimated value of the load torque, and detects the torque minimum phase based on the estimated value ,
The control device extracts a frequency component n times the rotational frequency of the electric motor from the calculated output torque, and sets a phase at which the extracted n times frequency component has a minimum value as the minimum torque phase. and detect as
The control device is
Obtaining the cosine and sine of the phase that varies at the n-fold frequency;
Obtaining a cosine component of the output torque by multiplying the output torque by the cosine;
Obtaining a sine component of the output torque by multiplying the output torque by the sine,
obtaining a low frequency component of the cosine component of the output torque by removing a frequency component higher than the n-fold frequency from the cosine component of the output torque;
obtaining a low frequency component of the sine component of the output torque by removing frequency components higher than the n-fold frequency from the sine component of the output torque;
The torque minimum phase is obtained by summing the arctangent of the ratio of the low frequency component of the sine component of the output torque to the low frequency component of the cosine component of the output torque and π.

ADVANTAGE OF THE INVENTION According to this invention, the vibration of a load element at the time of a stop can be suppressed sufficiently.

It is a schematic diagram showing an example of a refrigerating cycle of an air conditioner. (a) is a schematic longitudinal sectional view showing an example of a compressor, and (b) is a sectional view taken along line 2B-2B of FIG. 2(a). It is a figure which shows the electric motor drive device of embodiment of this invention. 4 is a diagram showing a configuration example of an inverter in FIG. 3; FIG. It is a functional block diagram showing an example of a control device used in an embodiment of the invention. FIG. 6 is a functional block diagram showing a configuration example of a torque minimum phase calculator in FIG. 5; FIG. 6 is a functional block diagram showing a configuration example of a voltage command value calculation unit shown in FIG. 5; FIG. 8 is a functional block diagram showing a configuration example of a compensating torque current command generation unit shown in FIG. 7; FIG. 4 is a model diagram for analyzing mechanical vibration of a compressor; (a) to (e) are graphs showing an example of changes in rotation speed, torque, etc. when the inverter is stopped without imposing restrictions on the stop phase. (a) to (e) are graphs showing an example of changes in rotational speed, torque, etc. when the inverter is stopped by imposing restrictions on the stop phase. An example of the magnitude of the angular acceleration of the compressor shell when the inverter is stopped at each phase is shown. It is a figure which shows an example of the relationship with the period to maintain. FIG. 6 is a functional block diagram showing a modification of the torque minimum phase calculator of FIG. 5; FIG. 14 is a diagram showing an example of a history of torque calculation values calculated by the torque calculation unit of FIG. 13;

BEST MODE FOR CARRYING OUT THE INVENTION An electric motor drive device according to an embodiment of the present invention and a refrigerating cycle application device including the same will be described below with reference to the accompanying drawings. It should be noted that the present invention is not limited by the embodiments shown below.

An example of a refrigeration cycle application device is an air conditioner, and the following embodiments apply the present invention to a drive device for an electric motor that drives a compressor of the air conditioner.

First, a refrigerating cycle in an example of an air conditioner will be described with reference to FIG.
The refrigeration cycle 900 of FIG. 1 can perform heating operation or cooling operation by switching operation of the four-way valve 902 .

During heating operation, as indicated by solid arrows, the refrigerant is pressurized by the compressor 904 and sent through the four-way valve 902, the indoor heat exchanger 906, the expansion valve 908, the outdoor heat exchanger 910, and the four-way valve 902. Return to compressor 904 .
During cooling operation, as indicated by the dashed arrow, the refrigerant is pressurized by the compressor 904 and sent through the four-way valve 902, the outdoor heat exchanger 910, the expansion valve 908, the indoor heat exchanger 906, and the four-way valve 902. Return to compressor 904 .

During heating operation, the indoor heat exchanger 906 acts as a condenser to release heat, and the outdoor heat exchanger 910 acts as an evaporator to absorb heat. During cooling operation, the outdoor heat exchanger 910 acts as a condenser to release heat, and the indoor heat exchanger 906 acts as an evaporator to absorb heat. The expansion valve 908 reduces the pressure of the refrigerant to expand it.
Compressor 904 is driven by electric motor 7 whose speed is controlled.

An example of the compressor 904 is shown in FIGS. 2(a) and (b). The illustrated compressor 904 is a closed rotary compressor, and includes a compressor shell 922 forming a closed container and a compression mechanism 924 arranged within the compressor shell 922 . Refrigerant is introduced into the compression mechanism 924 from the suction pipe 926 and discharged from the discharge pipe 928 .
Compressor shell 922 is supported by support member 930 .
The compression mechanism 924 has a cylinder 932 and a rotary piston 934 arranged within the cylinder 932 .

The electric motor 7 is disposed within the compressor shell 922 and has a rotor 7 a and a stator 7 b rotatably retaining the rotor 7 a , the rotor 7 a being coupled to the shaft 936 . The shaft 936 is rotatably held by bearings (not shown) to a frame (not shown), which is fixed to the compressor shell 922 .
Shaft 936 is coupled to crankshaft 938 , which in turn is coupled to rotary piston 934 .
Rotation of the rotor 7 a of the electric motor 7 is transmitted to the rotary piston 934 via the shaft 936 and crankshaft 938 .

A suction port 942 and a discharge port 944 are formed in the cylinder 932 , and vanes 946 are provided inside the cylinder 932 . The suction port 942 is connected to the suction pipe 926 and the discharge port 944 is connected to the discharge pipe 928 . Note that the suction port 942 and the discharge port 944 are conceptually illustrated, and their positions in FIG. 2(b) do not necessarily represent their actual positions accurately.
The vanes 946 are biased toward the center of the cylinder 932 so that they can move in the radial direction of the cylinder 932 while sliding on the peripheral surface of the rotary piston 934 .

When the shaft 936 rotates, the crankshaft 938 and the rotary piston 934 rotate in the direction indicated by the arrow RP. is discharged from
In the series of steps of suction, compression, and discharge as described above, the pressure applied to the rotary piston 934 changes, and this pressure change changes the load torque _Tl .

Since the electric motor 7 is arranged inside the compressor shell 922 as described above, the electric motor 7 is a part of the compressor 904, and the electric motor 7 drives the compression mechanism 924 of the compressor 904. You can also see that there is.

As will be described in detail below, the electric motor driving device of the present invention drives the electric motor 7, and is capable of reducing vibrations when stopping the electric motor 7, particularly vibrations of the compressor shell 922 and the pipes 926 and 928. It is possible.

FIG. 3 is a schematic wiring diagram showing the electric motor driving device 2 of the embodiment of the present invention together with the electric motor 7. As shown in FIG.
The illustrated electric motor drive device 2 is for driving the electric motor 7, and includes a reactor 4, a rectifier circuit 10, a smoothing capacitor 20, an inverter 30, a bus voltage detector 82, and a bus current detector 84. , a control power generation circuit 90 and a control device 100 .

The rectifier circuit 10 rectifies the AC voltage supplied from the AC power supply 1 . In the illustrated example, the rectifier circuit 10 is composed of a diode bridge. The input terminal of the diode bridge is connected to the AC power supply 1 via the reactor 4 and the output terminal is connected to the smoothing capacitor 20 .

Smoothing capacitor 20 smoothes the output voltage of rectifier circuit 10 .
One electrode of the smoothing capacitor 20 is connected to the first output terminal of the rectifier circuit 10 and the high potential side (positive side) DC bus 22a.
The other electrode of the smoothing capacitor 20 is connected to the second output terminal of the rectifier circuit 10 and the low potential side (negative side) DC bus 22b.
The voltage smoothed by the smoothing capacitor 20 is called a "bus voltage".

The inverter 30 receives the voltage across the smoothing capacitor 20, that is, the bus voltage, generates a frequency-variable AC voltage, and supplies the AC voltage to the electric motor 7 via output lines 331-333.

The electric motor 7 is a three-phase permanent magnet synchronous motor. In this case, inverter 30 generates a three-phase AC voltage and supplies it to electric motor 7 .

Bus voltage detector 82 detects voltage _Vdc between buses 22a and 22b as a bus voltage. The bus voltage detection unit 82 includes, for example, a circuit that divides the bus voltage _Vdc with series-connected resistors, and converts it into a voltage suitable for processing by the microcomputer in the control device 100, for example, a voltage of 5 V or less. Output. This signal (voltage detection signal) is converted into a digital signal by an A/D converter (not shown) in the control device 100 and used for processing inside the control device 100 .

A bus current detector 84 detects a bus current, that is, an input current _Idc of the inverter 30 . The bus current detector 84 includes a shunt resistor inserted in the DC bus 22b, and supplies an analog signal indicating the detection result to the controller 100. FIG. This signal (current detection signal) is converted into a digital signal by an A/D converter (not shown) in the control device 100 and used for processing inside the control device 100 .

The control power generation circuit 90 receives the voltage between both electrodes of the capacitor 20 , that is, the bus voltage _Vdc , and steps down the voltage to generate the control power voltage V100 and supply the control power voltage V100 to the control device 100 .

Control device 100 controls the operation of inverter 30 .
In order to operate the inverter 30 , the control device 100 generates PWM signals Sm 1 to Sm 6 and supplies them to the inverter 30 .

As shown in FIG. 4, the inverter 30 has an inverter main circuit 310 and a drive circuit 350, and input terminals of the inverter main circuit 310 are connected to the DC buses 22a and 22b.

The inverter main circuit 310 has six arms each including switching elements 311-316. Rectifying elements 321 to 326 for freewheeling are connected in antiparallel to the switching elements 311 to 316, respectively.

The drive circuit 350 generates the drive signals Sr1 to Sr6 based on the PWM signals Sm1 to Sm6, and controls the switching elements 311 to 316 to be turned on and off by the drive signals Sr1 to Sr6. are applied to the electric motor 7 via the output lines 331-333.

The PWM signals Sm1-Sm6 are of logic circuit signal level magnitude (0-5V), while the drive signals Sr1-Sr6 are of voltage levels required to control the switching elements 311-316, for example It is a signal having a magnitude of +15V to -15V. Further, while the PWM signals Sm1 to Sm6 use the ground potential of the control device 100 as a reference potential, the drive signals Sr1 to Sr6 are the potentials of the negative terminals (emitter terminals) of the corresponding switching elements. is used as the reference potential.

Control device 100 controls inverter 30 as described above. Control of inverter 30 includes control for operating inverter 30 and control for stopping inverter 30 .

When operating the inverter 30, the control device 100 supplies PWM signals Sm1 to Sm6 corresponding to frequencies and voltage values other than zero to the inverter 30, thereby controlling the frequency and voltage value of the output voltage of the inverter 30. Control.
When stopping the inverter 30, the control device 100 supplies the inverter 30 with the PWM signals Sm1 to Sm6 corresponding to the output voltage value of the inverter 30 being zero.

本書では、回転角速度を単に回転速度と言い、角周波数を単に周波数と言うことがある。The angular frequency ω of the output voltage of the inverter 30 determines the rotational angular velocity of the electric motor 7 in electrical angle (represented by the same symbol ω as the angular frequency of the output voltage), and is the rotational angular velocity of the electric motor 7 in mechanical angle. ω _m is equal to the rotational angular velocity ω of the electric motor 7 divided by the pole logarithm P _m . Therefore, there is a relationship represented by the following equation (1) between the rotational angular velocity ω _m of the electric motor 7 in mechanical angle and the angular frequency ω of the output voltage of the inverter 30 .

In this document, the rotational angular velocity is sometimes simply referred to as rotational speed, and the angular frequency is simply referred to as frequency.

The control device 100 generates an excitation current command value _Iγ ^* based on the currents _Iu , _Iv , and _Iw flowing through the electric motor 7, and generates a γ-axis voltage command value _Vγ ^* based on the excitation current command value _Iγ ^* . is calculated to match the estimated speed ω _est of the electric motor 7 with the speed command value ω ^{* ,} and based on the calculated torque current command value _{I δ *} ^, the δ-axis voltage A command value V _δ ^* is generated, and the inverter 30 is controlled based on the γ-axis voltage command value V _γ ^* and the δ-axis voltage command value V _δ ^* .

When the electric motor 7 drives a load element whose load torque varies periodically, the control device 100 controls the inverter so that the output torque _Tm of the electric motor 7 follows the periodic variation (pulsation) of the load torque _Tl . 30 is desirable.
The control device 100 may generate a torque current compensation value for the above tracking. The generated torque current compensation value is used to correct the torque current command value I _δ ^* .

When stopping the compressor 904 , the control device 100 stops the driving of the electric motor 7 by the inverter 30 by stopping the inverter 30 , thereby stopping the electric motor 7 .
Specifically, the control device 100 detects the phase (torque minimum phase) at which the load torque _Tl becomes the minimum (minimum value), and stops the inverter 30 during the period including the torque minimum phase (the AC output from the inverter 30 is voltage is not output), and the driving of the electric motor 7 is stopped.

When the control device 100 performs control for causing the output torque _Tm of the electric motor to follow the pulsation of the load torque _Tl , it can be considered that the output torque _Tm matches the load torque _Tl .

Therefore, the control device 100 may calculate the output torque _Tm of the electric motor 7, use the calculated output torque _Tm as an estimated value of the load torque, and detect the torque minimum phase based on the estimated value. .

Among the pulsating components of the load torque _Tl , when the frequency component (n-th order pulsation component) that is n times the rotation frequency of the electric motor 7 is the largest, the torque current compensation value is mainly based on the n-th order pulsation component. It may be for compensation.

In this case, the control device 100 may detect the phase that minimizes the n-order pulsation component of the load torque _Tl .

Then, the control device 100 calculates the output torque _Tm of the electric motor 7, extracts a frequency component n times the rotational frequency of the electric motor 7 from the calculated output torque _Tm , and minimizes the extracted component. The phase may be detected as the phase that minimizes the n-th order pulsation component of the load torque _Tl .

Control device 100 is realized by a microprocessor. A microprocessor may be a processor or processing device called a CPU (Central Processing Unit), a microcomputer, or a DSP (Digital Signal Processor).

FIG. 5 is a functional block diagram showing an example of the control device 100. As shown in FIG. As illustrated, the control device 100 has an operation control section 102 , a minimum torque phase calculation section 104 and an inverter control section 110 .

The operation control unit 102 outputs a frequency command value ω ^* . The operation control unit 102 also controls the pole pair number P _m of the electric motor 7, the flux linkage Φ _f of the permanent magnet, the γ-axis inductance L _γ , the δ-axis inductance L _δ , and the time constant T _f of the low-pass filter described later (that is, these information) is output.

The frequency command value ω ^* is obtained by multiplying the rotational speed command value (rotational angular velocity command value) _ωm ^* of the electric motor 7 by the number of pole pairs _Pm , as shown in the following equation (2).

Frequency command value ω ^* is supplied to inverter control unit 110 .
The number of pole pairs P _m , the flux linkage Φ _f , the γ-axis inductance L _γ and the δ-axis inductance L _δ are supplied to the torque minimum phase calculator 104 .
The time constant _Tf is supplied to the torque minimum phase calculator 104 and the inverter controller 110 .

The operation control unit 102 receives information indicating the room temperature (temperature of the air-conditioned space) detected by a temperature sensor (not shown), receives instructions from an operation unit (not shown) such as a remote controller, and controls the operation of each part of the air conditioner. do. Instructions from the operation unit include information indicating the set temperature, selection of operation modes (heating, cooling, dehumidification, etc.), and instructions for starting and ending operation.

A torque minimum phase calculator 104 obtains the phase that minimizes the load torque _Tl .
As described above, when the output torque _Tm is controlled to follow the pulsation of the load torque _Tl , it can be considered that the output torque _Tm matches the load torque _Tl .
Therefore, in the example shown below, the output torque T _m is calculated from the current flowing through the electric motor 7, and the calculated value (calculated output torque) is used as the estimated value of the load torque, and the estimated value of the load torque (estimated load torque) Detect the phase that minimizes

Also, assuming that the n-th order pulsation component is the largest among the pulsation components of the load torque _Tl , the phase θ _{mn_min} that minimizes the n-th order pulsation component is detected in the following example.

The phase θ _{mn_min} at which the n-th order pulsation component is minimized is the phase angle position at which the fluctuation of the n-th order pulsation component is minimized in each cycle.
On the other hand, the phase at which the load torque is minimized is the phase angle position at which the load torque Tl is minimized in each period of the fundamental wave of the variation of the load torque _Tl .

For example, when the load element of the electric motor 7 is a single rotary compressor, the frequency of the largest pulsation component is the same as the rotation frequency of the electric motor 7, so n=1.
When the load element of the electric motor 7 is a twin-rotary compressor, the frequency of the largest pulsation component is twice the rotation frequency of the electric motor 7, so n=2.
Here, it is assumed that no transmission mechanism is interposed between the electric motor 7 and the compressor. If a transmission mechanism is interposed, it is necessary to consider the transmission ratio as well.

In calculating the torque minimum phase θ _{mn_min} , in addition to the excitation current I _γ and the torque current I _δ output from the inverter control unit 110, the number of pole pairs P _m , the interlinkage magnetic flux Φ _f , the γ-axis inductance L _γ and the δ-axis inductance L _delta is used.
Here, it is assumed that the number of pole pairs P _m , the interlinkage magnetic flux Φ _f , the γ-axis inductance L _γ and the δ-axis inductance L _δ are notified from the operation control unit 102 .
As the flux linkage Φ _f , the γ-axis inductance L _γ and the δ-axis inductance L _δ , predetermined and held values may be used.

A torque minimum phase calculator 104 calculates the output torque _Tm , extracts the n-th order pulsation component _Tmn from the calculated output torque _Tm , and determines the phase at which the n-th order pulsation component _Tmn is minimized. demand.

FIG. 6 shows a configuration example of the torque minimum phase calculator 104 of FIG.
The illustrated torque minimum phase calculator 104 includes a torque calculator 401, a divider 402, a multiplier 403, a cosine calculator 405, a sine calculator 406, multipliers 407 and 408, and low-pass filters 409 and 410. , an arctangent calculator 411 , and an adder 412 .

Torque calculation unit 401 receives excitation current I _γ and torque current I _δ from inverter control unit 110, and receives pole pair number P _m , interlinkage magnetic flux Φ _f , γ-axis inductance L _γ and δ-axis inductance L from operation control unit 102. _δ is received, and the output torque _Tm is determined from these. Calculation of the output torque _Tm can be performed by the following equation (3).

The output torque _Tm calculated by Equation (3) includes a DC component and an AC component. The AC component is a component that periodically changes with time.

A division unit 402 divides the electrical angle _θe by the number of pole pairs _Pm to calculate the rotational phase (mechanical angle) _θm .
The multiplier 403 multiplies the rotational phase _θm by n to obtain a phase (phase angle) _θmn that changes at a frequency that is n times the rotational frequency of the electric motor 7 .

A cosine calculator 405 receives the phase θ _mn and outputs a cosine cos θ _mn .
A sine calculator 406 receives the phase θ _mn and outputs a sine sin θ _mn .

A multiplier 407 multiplies the output torque _Tm by cos θ _mn to obtain a cosine component T _m ·cos θ _mn of the output torque _Tm . A multiplier 408 multiplies the output torque _Tm by sin θ _mn to obtain the sine component T _m ·sin θ _mn of the output torque T _m .

The cosine component T _m ·cos θ _mn and the sine component T _m ·sin θ _mn include a pulsation component with a frequency of ω _mn and a pulsation component with a higher frequency (harmonic component).

Here, the frequency ω _mn is n times the frequency ω _m , and the relationship between ω _mn and θ _mn is expressed by the following equation (4).

Low-pass filters 409 and 410 are first-order lag filters whose transfer function is expressed as 1/(1+sT _f ).
where s is the Laplacian operator. T _f is a time constant, and is determined to remove (sufficiently attenuate) pulsation components with frequencies higher than the frequency ω _mn .

式（５）で、ｔは時間を表す。The time constant T _f is determined by the operation control unit 102 using the estimated value ω _est of the frequency ω so as to remove the pulsation component with a frequency higher than the estimated value ω _{mn_est} calculated by the following equation (5). .

In equation (5), t represents time.

Estimated value ω _est is calculated in inverter control unit 110 as described later.
As described above, the estimated value ω _{mn_est} is used instead of the frequency ω _mn to determine the time constant T _f , but here ω _{mn_est} is assumed to be equal to ω _mn .
The time constant _Tf thus determined is notified from the operation control unit 102 to the low-pass filters 409 and 410 .

A low-pass filter 409 performs low-pass filtering on the cosine component T _m ·cos θ _mn to remove pulsation components with a frequency higher than the frequency ω _mn , and outputs a low frequency component T _mcos .
The low-frequency component _Tmcos is a DC quantity representing a cosine component with a frequency _ωmn among the pulsating components of the torque _Tm .

The low-pass filter 410 performs low-pass filtering on the sine component T _m ·sin θ _mn to remove pulsation components with a frequency higher than the frequency ω _mn , and outputs a low frequency component T _msin .
The low-frequency component _Tmsin is a DC quantity representing a sinusoidal component with a frequency _ωmn among the pulsating components of the torque _Tm .

The arctangent calculator 411 receives _Tmcos and _Tmsin and calculates the arctangent represented by the following equation (6).

The addition unit 412 obtains the phase θ _{mn_min} by adding the arc tangent represented by Equation (6) and π as represented by Equation (7) below.

The obtained phase θ _{mn_min} is the phase at which the n-th order pulsation component of the output torque T _m is minimized. As described above, when the pulsation is compensated, the phase θ _{mn_min} can be regarded as the phase at which the n-order pulsation component of the load torque _Tl is minimized.

The torque minimum phase θ _{mn_min} obtained in this manner is notified to the operation control unit 102 . When there is a request to stop the electric motor 7, the operation control unit 102 stops the inverter 30 during the period Ty including the torque minimum phase _{θmn_min} .

Specifically, the operation control unit 102 sets the speed command value ω ^* to zero. Assuming that the speed command value ω ^* is 0, the PWM signals Sm1 to Sm6 correspond to a voltage value of zero.
Alternatively, the PWM signals Sm 1 to Sm 6 output from the inverter control section 110 may not be supplied to the inverter 30 . A switch (not shown) is provided between the output of the inverter control unit 110 and the drive circuit 350 to prevent the supply of the PWM signals Sm1 to Sm6. The switch may be closed and opened when blocking the supply of the PWM signals Sm1 to Sm6. The opening and closing of such switches may be controlled by the operation control unit 102 .

By limiting the timing of stopping within the period Ty, it is possible to reduce the vibration of the pipes, for example, the suction pipe 926 and the discharge pipe 928, at the time of stopping.

Inverter control unit 110 detects bus voltage _Vdc detected by bus voltage detection unit 82, bus current _Idc detected by bus current detection unit 84, frequency command value ω ^* supplied from operation control unit 102, and Based on the time constant _Tf , PWM signals Sm1 to Sm6 are generated and supplied to the inverter 30 to cause the inverter 30 to output an AC voltage with variable frequency and voltage value.

The inverter control unit 110 includes a current restoration unit 111, a three-phase to two-phase conversion unit 112, an excitation current command value generation unit 113, a voltage command value calculation unit 115, an electric phase calculation unit 116, and a two-to-three phase conversion unit. It has a section 117 and a PWM signal generation section 118 .

A current restoration unit 111 restores the phase currents I _u , I _v , and I _w flowing through the electric motor 7 based on the current value I _dc detected by the bus current detection unit 84 . The current restoration unit 111 restores the phase current by sampling the DC current _Idc detected by the bus current detection unit 84 at timing determined based on the PWM signals Sm1 to Sm6 from the PWM signal generation unit 118. .

The three-to-two phase conversion unit 112 converts the current values _Iu , _Iv , and _Iw restored by the current restoration unit 111 into an excitation _current (γ axis current) I _γ and torque current (δ axis current) I _δ , that is, the current value of the γ-δ axis.

Based on the torque current I _δ , the excitation current command value generation unit 113 obtains an optimum excitation current command value I _γ ^* for driving the electric motor 7 with the highest efficiency.
_In FIG ^. 5, _the excitation current command value _Iγ ^* is obtained based on the torque current _Iδ ^. The same effect can be obtained even if

Based on the torque current I _δ (or the excitation current I _γ and the frequency command value ω ^* ), the excitation current command value generation unit 113 determines that the output torque is a predetermined value or more (or maximum), that is, the current value is a predetermined value or less (or output the excitation current command value I _γ ^* that becomes the current phase β _m (not shown) that becomes the minimum).

The voltage command value calculation unit 115 generates the excitation current I _γ and the torque current I _δ obtained from the three-phase two-phase conversion unit 112, the frequency command value ω ^* output from the operation control unit 102, and the excitation current command value. The exciting current command value I _γ ^* obtained from the unit 113 is input, and based on these, the voltage command values V _γ ^* and V _δ ^* are generated and output.
The voltage command value calculator 115 further estimates and outputs an estimated frequency value ω _est from the voltage command values V _γ ^* and V _δ ^* , the excitation current I _γ and the torque current I _δ .

The electrical phase calculator 116 calculates the electrical phase θ _e by integrating the estimated value ω _est of the frequency output from the voltage command value calculator 115 .

The two-to-three phase conversion unit 117 performs electric phase calculation on the γ-axis voltage command value V _γ ^* and the δ-axis voltage command value V _δ ^* (voltage command value in the two-phase coordinate system) obtained by the voltage command value calculation unit 115. The electrical phase θ _e obtained by the unit 116 is used to convert the output voltage command values (three-phase voltage command values) V _u ^* , V _v ^* , V _w ^* in the three-phase coordinate system and output them.

PWM signal generator 118 converts bus voltage V _dc detected by bus voltage detector 82 and three-phase voltage command values V _u ^* , V _v ^* , and V _w ^* obtained by two-to-three-phase converter 117 into Based on this, PWM signals Sm1 to Sm6 are generated and output.
The PWM signals Sm1-Sm6 control the on/off timing of the switching elements 311-316 of each arm of the inverter 30 so that the output voltage of the inverter 30 matches the three-phase voltage command values Vu ^* , Vv ^* , Vw ^* . is a signal.

For example, as shown in FIG. 7, the voltage command value calculation unit 115 includes a frequency estimation unit 501, a subtraction unit 502, a speed control unit 503, a compensation value calculation unit 504, an addition unit 505, a subtraction unit 509, and a subtraction unit 509. 510 , an excitation current control unit 511 , and a torque current control unit 512 .

The frequency estimator 501 receives the excitation current I _γ and the torque current I _δ and the voltage command values V _γ ^* and V _δ ^* as inputs, estimates the frequency of the voltage applied to the electric motor 7 based on these, and estimates Output the value ω _est .

The subtraction unit 502 calculates the difference (ω ^* _−ωest ) between the frequency estimation value ω _est generated by the frequency estimation unit 501 and the frequency command value ω ^* .

Speed control unit 503 performs proportional integral (PI) calculation on the difference (ω ^* −ω _est ) calculated by subtraction unit 502 to obtain torque current command value I _δ ^* that brings the difference closer to zero. By generating the torque current command value I _δ ^* in this manner, control is performed to match the frequency estimated value ω _est with the frequency command value ω ^* .

Compensation value calculation section 504 outputs torque current compensation value I _{δ_trq} based on frequency estimation value ω _est output from frequency estimation section 501 .
The torque current compensation value I _{δ_trq} is for suppressing the pulsation component with the frequency ω _est , especially the pulsation component with the frequency ω _mn . Here, “the pulsation component with the frequency ω _est , especially the component with the frequency ω _mn ” means the pulsation component with the value (direct current quantity) representing the frequency ω _est , especially the pulsation component with the pulsation frequency ω _mn . .

Addition unit 505 adds output I _δ ^* of speed control unit 503 and torque current compensation value I _{δ_trq} to generate corrected torque current command value I _δ ^** .
By correcting the torque current command value I _δ ^* with the torque current compensation value I _{δ_trq} , it is possible to suppress the speed pulsation caused by the load torque pulsation.

A subtraction unit 509 obtains a difference (I _γ ^* −I _γ ) between the excitation current command value I _γ ^* generated by the excitation current command value generation unit 113 and I _γ .

An excitation current control unit 511 performs a proportional integral (PI) operation on the difference (I _γ ^* −I _γ ) obtained by the subtraction unit 509, and the γ-axis voltage command value V _γ ^* that makes the difference approach zero. to generate By generating the γ-axis voltage command value V _γ ^* in this manner, control is performed to match I _γ with I _γ ^* .

Subtraction unit 510 obtains the difference (I _δ ^** - I _δ ) between torque current command value I _δ ^** generated by addition unit 505 and I _δ .

Torque current control unit 512 performs proportional integral (PI) calculation on the difference (I _δ ^** -I _δ ) obtained by subtraction unit 510, and δ-axis voltage command value V _δ that brings the difference closer to zero. ^* is generated. By generating the δ-axis voltage command value V _δ ^* in this manner, control is performed to match I _δ with I _δ ^** .

The compensation value calculator 504 is configured as shown in FIG. 8, for example. The illustrated compensation value calculator 504 includes a cosine calculator 551, a sine calculator 552, multipliers 553 and 554, low-pass filters 555 and 556, subtractors 557 and 558, frequency controllers 559 and 560, It has multipliers 561 and 562 and an adder 563 .

The cosine calculator 551 receives θ _mn and calculates its cosine cos θ _mn . The sine calculator 552 receives θ _mn and calculates its sine sin θ _mn .

The multiplier 553 multiplies the estimated value ω _est by cos θ _mn to obtain the cosine component ω _est ·cos θ _mn of the estimated value ω _est .

The multiplier 554 multiplies the estimated value ω _est by sin θ _mn to obtain the sine component ω _est ·sin θ _mn of the estimated value ω _est .

The cosine component ω _est ·cos θ _mn and the sine component ω _est ·sin θ _mn calculated by the multipliers 553 and 554 include a pulsation component with a frequency of ω _mn and a pulsation component with a higher frequency (harmonic component). It is included.

The low-pass filters 555 and 556 are, like the low-pass filters 409 and 410, first-order lag filters whose transfer functions are represented by 1/(1+sT _f ).
where s is the Laplacian operator. T _f is a time constant, and is determined to remove (sufficiently attenuate) pulsation components with frequencies higher than the frequency ω _mn .

The time constant T _f is determined by the operation control unit 102 and notified to the low-pass filters 555 and 556 as described for the low-pass filters 409 and 410 .

A low-pass filter 555 performs low-pass filtering on the cosine component ω _est ·cos θ _mn to remove pulsation components with a frequency higher than the frequency ω _mn , and outputs a low frequency component ω _{est_cos} . The low-frequency component ω _{est_cos} is a DC quantity representing a cosine component with a frequency ω _mn among the pulsating components of the estimated value ω _est .

A low-pass filter 556 performs low-pass filtering on the sine component ω _est ·sin θ _mn to remove pulsation components with a frequency higher than the frequency ω _mn and outputs a low frequency component ω _{est_sin} . The low-frequency component ω _{est_sin} is a DC quantity representing a sinusoidal component with a frequency ω _mn among the pulsating components of the estimated value ω _est .

A subtraction unit 557 obtains the difference between the output ω _{est_cos} of the low-pass filter 555 and zero.
A subtraction unit 558 obtains the difference between the output ω _{est_sin} of the low-pass filter 556 and zero.

The frequency control unit 559 performs a proportional integral (PI) operation on the difference (ω _{est_cos} −0) obtained by the subtraction unit 557 to obtain the cosine component I _{δ_trq_cos} of the current command value that brings the difference closer to zero. By generating the cosine component I _{δ_trq_cos} in this way, control is performed to match the low frequency component ω _{est_cos} to zero.

Frequency control unit 560 performs proportional integral (PI) calculation on the difference (ω _{est_sin} −0) obtained by subtraction unit 558 to obtain the sine component I _{δ_trq_sin} of the current command value that brings the difference closer to zero. By generating the sine component I _{δ_trq_sin} in this manner, control is performed to match the low frequency component ω _{est_sin} to zero.

The multiplier 561 multiplies the output I _{δ_trq_cos} of the frequency control unit 559 by cos θ _mn to generate I _{δ_trq_cos} ·cos θ _mn . I _{δ_trq_cos} ·cos θ _mn is the AC component with frequency n·ω _est .

The multiplier 562 multiplies the output I _{δ_trq_sin} of the frequency control unit 560 by sin θ _mn to generate I _{δ_trq_sin} ·sin θ _mn . I _{δ_trq_sin} ·sin θ _mn is the AC component with frequency n·ω _est .

The adder 563 obtains the sum of the output I _{δ_trq_cos} ·cos θ _mn from the multiplier 561 and the output I _{δ_trq_sin} ·sin θ _mn from the multiplier 562 .
The output of the adder 563 is output as the torque current compensation value I _{δ_trq} .

The torque current compensation value I _{δ_trq} obtained in this manner is added by the addition unit 505 in FIG. 7, and the addition result is used as the corrected torque current command value I _δ ^** , thereby suppressing the ripple component. can be done.

In the state where the pulsation component is suppressed, it can be seen that the output torque _Tm and the load torque _Tl match as described above. Therefore, the drive of the electric motor 7 by the inverter 30 is stopped during the period including the phase in which the output torque _Tm is minimized, and the compressor 904 is stopped.
The advantages of stopping the inverter 30 during a period including a phase in which the output torque _Tm is minimized will be described below.

FIG. 9 is a model diagram for analyzing mechanical vibration of the compressor 904. As shown in FIG. The compressor 904 is mechanically divided into a rotating portion 951 and a fixed portion 952 that rotatably supports the rotating portion 941 , and the fixed portion 952 is supported by a support member 930 .
The rotating part 951 includes the rotor 7 a of the electric motor, the rotary piston 934 , the shaft 936 and the crankshaft 938 .
The fixed part 952 includes the motor stator 7 b , the compressor shell 922 and the cylinder 932 .

Jsh is the inertia of the fixed portion 952, ksh is the spring constant of the support member 930, Dsh is the damping constant, _θs is the deflection angle of the fixed portion 952, _Tm is the output torque of the electric motor 7, and _{Tl is} the load torque of the compression mechanism 924. Then, the equation of motion regarding the vibration of the fixed part 952 is given by the following equation (8).
Since the compressor shell 922 is part of the fixed portion 952 , the deflection angle and vibration of the compressor shell 922 can be viewed as equal to the deflection angle and vibration of the fixed portion 952 .

From equation (8), it can be seen that the smaller T _l - T _m = ΔT (torque difference), the smaller the angular acceleration (d ² θ _s /dt ² ) and the smaller the deflection angle θ _s .

In order to reduce the vibration at the time of stopping, the change in the torque difference ΔT at the time of stopping should be reduced. The torque difference ΔT before stopping is equal to T ₁ −T _m , while the torque difference ΔT after stopping is equal to T ₁ . This is because the output torque _Tm of the electric motor 7 becomes zero when the inverter 30 stops.
Therefore, it can be seen that the inverter 30 should be stopped in a phase where the load torque _Tl is small in order to reduce the change in the torque difference ΔT at the time of stopping.

Assuming the load torque of a single rotary compressor as an example, changes in torque, changes in angular acceleration, etc. when the inverter 30 is stopped are shown in FIGS. Description will be made with reference to (e). The horizontal axis in FIGS. 10(a) to (e) and FIGS. 11(a) to (e) is time (seconds). In FIGS. 10(a) to (e) and FIGS. 11(a) to (e), the timing of stopping the inverter 30 is indicated by symbol Ts.

FIGS. 10A to 10E show the case where the inverter 30 is stopped without any particular timing control, and FIGS. 11A to 11E show the case where the inverter 30 is stopped.

In FIGS. 10(a) and 11(a), the dashed line indicates the rotation speed command value f _m ^* , the solid line indicates the actual speed f _m in the rotation speed per second, and the dotted line indicates the estimated value f _{m_est} of the rotation speed per second. indicates
These have the relationships of the following equations (9a) to (9c) with respect to the command value ω ^* in electrical angular velocity, the actual speed ω, and the estimated value ω _est .

10(b) and 11(b), the solid line indicates the U-phase current _Iu , the dotted line indicates the V-phase current _Iv , and the dashed line indicates the W-phase current _Iw .

10(c) and 11(c), the solid line indicates the output torque _Tm , and the dotted line indicates the load torque _Tl .
In FIGS. 10(d) and 11(d), the dashed line indicates the target value I _δ ^** of the torque current, and the solid line indicates the actual torque current I _δ .
10(e) and 11(e) show the angular acceleration Aa of the compressor shell 922 in solid lines.

As can be seen from FIGS. 10(c) and 11(c), it is assumed that the pulsation of the load torque _Tl has a period of 0.05 seconds and therefore a frequency of 20 Hz.

If the timing of stopping is not restricted, the electric motor 7 is operated at a phase other than the phase estimated to minimize the load torque _Tl (estimated minimum torque phase), as shown in FIG. 10C, for example. It may cause you to stop. In the case of FIG. 10(e), the angular acceleration Aa of the compressor shell 922 after stopping may increase, which is not preferable.

On the other hand, when the inverter 30 is stopped during a period including the phase estimated to minimize the load torque _Tl (estimated minimum torque phase), as shown in FIG. It can be seen that the angular acceleration Aa of the shell 922 has a small value, which is preferable.

FIG. 12 shows an example of the load torque in each phase and the magnitude Ap of the angular acceleration of the compressor shell 922 after stopping when the inverter 30 is stopped in each phase.
While the angular acceleration Aa in FIGS. 10(e) and 11(e) indicates an instantaneous value at each time point, the magnitude of the angular acceleration Ap in FIG. , represents the pp value (peak-to-peak value) of the angular acceleration applied to the compressor shell 922 immediately after that, more specifically, the maximum value of the pp value during the period until the vibration subsides after stopping.

As shown in FIG. 12, the smaller the load torque in the stop phase, the smaller the post-stop angular acceleration (pp value) Ap. can be minimized and therefore less vibration is required.

In the present embodiment, as described above, torque minimum phase calculation section 104 obtains output torque _Tm , obtains low frequency components _Tmcos and _Tmsin from _Tm , and obtains _Tmcos and _Tmsin from equation (7). ), the phase θ _{mn_min} that minimizes the n-th pulsation component of the estimated load torque is obtained.
The reason why the above method can obtain the phase θ _{mn_min} that minimizes the n-th pulsation component of the estimated load torque will be described below.

In a situation where speed pulsation suppression control (pulsation suppression control) is ideally performed (a state in which speed fluctuations are suppressed), the output torque _Tm has the same phase and amplitude as the load torque _Tl , The phase at which the output torque _Tm is minimized coincides with the phase at which the load torque _Tl is minimized.

Therefore, if the pulsation suppression control is performed and the output torque _Tm is in the same phase and the same amplitude as the load torque _Tl , in this state, the load torque _Tl is reduced from the phase at which the output torque _Tm is minimized. It is possible to estimate the minimum phase.

In the case of the electric motor 7 having a small reluctance torque, the phase at which the torque current is minimized and the phase at which the load torque is minimized are substantially the same. Therefore, when the output torque is minimum, the torque current also becomes the minimum value or a value close to it, and the phase current also becomes a minimum value or a value close to this. Therefore, the phase current may be detected, and the phase at which the detected phase current becomes the minimum value may be detected as the minimum torque phase.

The output torque _Tm is given by equation (3) above.
For example, when pulsation suppression control is performed to compensate only for the pulsation component with a frequency n times as high as the rotation frequency of the electric motor 7, the DC _component and the AC component (n-order pulsation component), it is possible to transform equation (3) into equation (10) below.

_TmDC is the DC component of the torque _Tm ;
_Tmcos is the amplitude of the cosine component of the n-th order pulsation component of the torque _Tm ;
_Tmsin is the amplitude of the sinusoidal component of the n-order pulsation component of the torque _Tm .

Equation (10) can be transformed into Equation (11) below.

T _mcos and T _msin are direct current quantities, and although they may change gently over time, they do not change with the same periodicity as the change in θ _m . Therefore, the output torque _Tm given by equation (11) is minimized when the cosine function represented by equation (12) below is minimized.

The cosine function represented by Equation (12) is minimized when its phase is π. That is, the cosine function represented by Equation (12) is minimized when Equation (13) below is satisfied.

Denoting θ _mn when equation (13) is satisfied by θ _{mn_min} yields equation (7) above.
That is, it can be seen that the phase θ _{mn_min} obtained by the above equation (7) is the phase that minimizes the n-th order pulsation component of the estimated load torque.

Next, the relationship between the phase θ _{mn_min} at which the n-th order pulsation component is minimized and the period Ty during which the torque maintains the minimum value or a value close thereto will be described.
Depending on the characteristics of the load element, the load torque may include not only the n-th order pulsation component but also components other than the n-th order pulsation component, particularly harmonic components. The length of time that the torque remains close to the minimum depends on the harmonics involved.

FIG. 12 shows not only the load torque T _l but also its nth order pulsation component T _ln . The horizontal axis is the phase θ _mn where one cycle of the n-th order pulsation component T _ln is 360 degrees.

In the illustrated example, the period Ty from before the angle _{θmnya to after the angle θmnyb with} respect to the phase _{θmn_min} at which the n-th order pulsation component _Tln is minimized, and the torque _Tl is at a minimum or a value close to the minimum. be.

Specifically, the phase θ _{mn_min} at which the n-th order pulsation component T _ln is minimized is 62 degrees, and the phase at which the torque T _l is minimized is the phase θ _{mn_min} at which the n-th order pulsation component T _ln is minimized. is consistent with
During the period Ty from 2 degrees to 120 degrees, the torque _Tl is a negative value and maintains the minimum value or a value close thereto. That is, the period Ty from before the angle θ _mnya =60 degrees to after the angle θ _mnyb =58 degrees with respect to the phase θ _{mn_min} (=62 degrees) at which the n-th order pulsation component T _ln is minimized, and the torque T _l is the minimum value or a value close to the minimum value.

In such a case, the pp value Ap of the angular acceleration of the compressor shell 922 after stopping is the same in any phase of the period Ty, and the vibration of the piping at the time of stopping becomes small.

The angles θ _mnya and θ _mnyb that define the period Ty are angles that make one cycle of the nth pulsation component 360 degrees. The example shown in FIG. 12 is for n=1, and the angles θ _mnya and θ _mnyb are equal to the mechanical angles θ _mya and θ _myb .
When n is other than 1, the following equations (14a) and (14b) are established between θ _mnya , θ _mnyb and θ _mya , θ _myb .

Therefore, when the period Ty is expressed in mechanical angles, it is necessary to use _θmya /n and _θmyb /n instead of _θmnya and _θmnyb .
For example, the above first period is a value obtained by dividing the first mechanical angle θ _mya by n, and the above second period is a value obtained by dividing the second mechanical angle θ _myb by n. is the value obtained.

In the above example, the torque minimum phase calculator 104 calculates the phase at which the torque is minimized (the phase at which the n-th pulsation component of the torque is minimized) by the calculation represented by Equation (7). Alternatively, the phase that minimizes the load torque may be estimated from the history of changes in the output torque _Tm .

A configuration example in this case is shown in FIG.
The illustrated torque minimum phase calculator 104 b has a torque calculator 401 , a calculated value history storage unit 421 , and a calculated value history analysis unit 422 .

The torque calculation unit 401 calculates the output torque _Tm by performing the calculation represented by the equation (3), for example, in the same manner as the torque calculation unit 401 in FIG. The calculation of the output torque _Tm is performed at each sampling timing of a predetermined cycle.

The calculated value history storage unit 421 accumulates the values (calculated values) of the output torque _Tm calculated at each sampling timing by the torque calculation unit 401, and generates a time series of the accumulated values. For example, the values up to a predetermined number of cycles before are stored. The “period” referred to here is the period of the n-th order pulsation component of the torque. FIG. 14 shows an example of waveforms representing the accumulated time series. In FIG. 14, the white "O" indicates the sampled output torque value. In the illustrated example, a time series is formed by values for three cycles.

Calculated value history analysis unit 422 obtains the phase (torque minimum phase) at which the value of output torque _Tm is the minimum in each period from the above time series.
Each sampling timing is associated with, for example, the phase _{θ e} calculated by the electrical phase calculator 116 in _FIG . There is

Therefore, for example, θ _{mn_min} can be obtained by multiplying the phase θ _e corresponding to the sampling timing at which the value of the output torque T _m calculated in each cycle is determined to be the minimum value by n/P _m . .
The length of each cycle can be determined based on the calculated change in output torque value. For example, it can be calculated based on one or both of the time from the occurrence of the minimum value to the occurrence of the next minimum value and the time from the occurrence of the maximum value to the occurrence of the next maximum value.

In obtaining the torque minimum phase in each cycle, correction may be performed based on the torque minimum phase obtained in one or more cycles before that. For example, for each cycle, the torque minimum phase determined based only on the calculated value of the cycle (torque value calculated at the sampling timing within the cycle), and each of the preceding one or two or more cycles , and the minimum torque phase obtained based only on the calculated values for each period may be used as the minimum torque phase for each period.

In weighting, the torque minimum phase obtained based only on the calculated value of each period may be given the largest weight. Also, with respect to cycles other than the respective cycles, a larger weight may be assigned to a cycle closer to the respective cycles.

In the above example, the phase that minimizes the output torque is calculated. Alternatively, the phase that minimizes the torque current may be calculated, and the calculated phase may be used as the phase that minimizes the load torque.

In the above example, the inverter 30 is stopped during a period including a phase in which the load torque becomes the minimum (minimum value) in each cycle. Alternatively, the inverter 30 may be stopped during a period including a phase in which the load torque is close to the minimum value in each cycle.
In short, the inverter 30 should be stopped during a period including a phase in which the load torque becomes the minimum value or a value close to the minimum value.

In addition, if there is a period during which the load torque is negative, all or part of the period during which the load torque is negative is defined as a period including a phase in which the load torque is the minimum value or a value close to the minimum value. You can use it.

In the above example, the phase currents I _u , I _v , and I _w are restored from the DC current I _dc on the input side of the inverter 30 . Alternatively, current detectors may be provided on the output lines 331, 332, and 333 of the inverter 30 to detect the phase currents. In that case, the current detected by the detector may be used instead of the current restored by the current restoration section 111 .

The switching elements 311 to 316 of the inverter main circuit 310 are assumed to be IGBTs (Insulated Gate Bipolar Transistors) or MOSFETs, but any element capable of switching may be used. . In the case of a MOSFET, since it has a parasitic diode due to its structure, the same effect can be obtained without connecting anti-parallel rectifying elements (321 to 326) for freewheeling.

As for the materials constituting the switching elements 311 to 316, not only silicon (Si) but also wide bandgap semiconductors such as silicon carbide (SiC), gallium nitride (GaN), and diamond are used. Loss can be reduced.

According to the above embodiment, it is possible to reliably and sufficiently suppress the vibration of the load element at the time of stopping. When the load element is a compressor, vibration of the compressor shell can be suppressed, and damage to piping connected to the compressor shell can be prevented. Conversely, it is possible to use a pipe having a relatively low metal fatigue resistance, thereby reducing the cost of the pipe.

The configuration of the above-described embodiment is an example of the configuration of the present invention, and can be combined with another known technique. It is also possible to configure

INDUSTRIAL APPLICABILITY As described above, the present invention is suitable for an electric motor drive device and a refrigeration cycle application device including the same.
Although an air conditioner is given as an example of a refrigeration cycle application device, the present invention is not limited to this, and can be applied to refrigerators, freezers, heat pump water heaters, and the like.

1 AC power supply 2 Electric motor drive device 4 Reactor 7 Electric motor 20 Smoothing capacitor 30 Inverter 82 Bus voltage detection unit 84 Bus current detection unit 90 Control power generation circuit 100 Control device 102 Operation control unit 104 torque minimum phase calculator 110 inverter controller 111 current restorer 112 three-phase two-phase converter 113 excitation current command value generator 115 voltage command value calculator 116 electric phase calculator 117 two-phase three-phase conversion unit 118 PWM signal generation unit 401 torque calculation unit 402 division unit 403 multiplication unit 405 cosine calculation unit 406 sine calculation unit 407, 408 multiplication unit 409, 410 low-pass filter 411 arctangent calculation unit, 412 addition unit 421 calculated value history storage unit 422 calculated value history analysis unit 501 frequency estimation unit 502 subtraction unit 503 speed control unit 504 compensation value calculation unit 505 addition unit 509, 510 subtraction unit 511 excitation Current control section 512 Torque current control section 551 Cosine calculation section 552 Sine calculation section 553,554 Multiplication section 555,556 Low-pass filter 557,558 Subtraction section 559,560 Frequency control section 561,562 Multiplication section , 563 adder 900 refrigeration cycle 902 four-way valve 904 compressor 906 indoor heat exchanger 908 expansion valve 910 outdoor heat exchanger 922 compressor shell 924 compression mechanism 926 suction pipe 928 discharge pipe.

Claims (7)

an electric motor that drives a load element whose load torque varies periodically;
an inverter that applies an alternating voltage with a variable frequency and voltage value to the electric motor;
and a control device that controls the inverter,
The control device is
controlling the inverter so that the output torque of the electric motor follows periodic fluctuations of the load torque;
When stopping the electric motor, stopping the inverter during a period including a torque minimum phase in which the load torque becomes a minimum value or a value close to the minimum value,
The control device generates a torque current compensation value for causing the output torque of the electric motor to follow periodic fluctuations in the load torque, corrects the torque current command value based on the generated torque current compensation value, and corrects generating a voltage command value based on the generated torque current command value, controlling the inverter based on the generated voltage command value;
The control device calculates the output torque of the electric motor, uses the calculated output torque as an estimated value of the load torque, and detects the torque minimum phase based on the estimated value ,
The control device extracts a frequency component n times the rotational frequency of the electric motor from the calculated output torque, and sets a phase at which the extracted n times frequency component has a minimum value as the minimum torque phase. and detect as
The control device is
Obtaining the cosine and sine of the phase that varies at the n-fold frequency;
Obtaining a cosine component of the output torque by multiplying the output torque by the cosine;
Obtaining a sine component of the output torque by multiplying the output torque by the sine,
obtaining a low frequency component of the cosine component of the output torque by removing a frequency component higher than the n-fold frequency from the cosine component of the output torque;
obtaining a low frequency component of the sine component of the output torque by removing frequency components higher than the n-fold frequency from the sine component of the output torque;
The torque minimum phase is obtained by obtaining the sum of the arctangent of the ratio of the low frequency component of the sine component of the output torque to the low frequency component of the cosine component of the output torque and π.
electric motor drive. the electric motor is a permanent magnet electric motor,
The control device obtains an exciting current and a torque current of the electric motor,
The number of pole pairs of the motor is P _m , the flux linkage of the permanent magnet is Φ _f , the γ-axis inductance is L _γ , the δ-axis inductance is L _δ , the excitation current I _γ , the torque current I _δ , the output torque is T _m , the control device has the formula

Find the output torque by the calculation represented by
The electric motor drive device according to claim 1. The torque current compensation value is for suppressing the n-fold frequency component,
The control device is
Multiplying the estimated value of the frequency of the voltage applied to the electric motor by the cosine to obtain a cosine component of the estimated value,
obtaining a sine component of the estimated value by multiplying the estimated value of the frequency by the sine;
Obtaining a low frequency component of the cosine component of the estimated value by removing frequency components higher than the n-fold frequency of the cosine component of the estimated value;
Obtaining a low frequency component of the sine component of the estimated value by removing frequency components higher than the n-fold frequency of the sine component of the estimated value;
obtaining the cosine component of the current command value by performing control so that the low frequency component of the cosine component of the estimated value approaches 0;
obtaining the sine component of the current command value by performing control so that the low frequency component of the sine component of the estimated value approaches 0;
The torque current compensation value is obtained by adding the product of the cosine component of the current command value and the cosine and the product of the sine component of the current command value and the sine.
3. The electric motor driving device according to claim 1 or 2. The control device obtains the torque current command value by performing control such that the estimated value of the frequency of the voltage applied to the electric motor matches the frequency command value.
The electric motor drive device according to claim 3. The period including the minimum torque phase is a period from a first period before to a second period after the phase in which the n-fold frequency component has a minimum value. 5. The electric motor drive device according to any one of 1 to 4 . The first period is an angle obtained by dividing a predetermined first mechanical angle by n,
The electric motor drive device according to claim 5, wherein the second period is an angle obtained by dividing a predetermined second mechanical angle by n. A refrigeration cycle application device comprising the electric motor drive device according to any one of claims 1 to 6.

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