JP7348667B2 - switching power supply - Google Patents

Description

The present invention relates to a switching power supply, and more particularly or to a power supply capable of outputting pulses with a fast rise time for use, for example, in driving semiconductor lasers.

Semiconductor lasers have come into widespread use in recent years. Semiconductor lasers have the advantage of being easier to install and maintain than conventional solid-state lasers and carbon dioxide lasers, and are expected to become more widely used. Furthermore, as semiconductor lasers are becoming more powerful, there is a demand for pulsed driving instead of the conventional continuous driving, and high-speed pulsed driving is also becoming necessary. Conventionally, a dropper power supply is often used as a power supply to drive a semiconductor laser, but dropper power supplies are inefficient, generate a lot of heat, and tend to be large in size. In other words, most laser processing machines used for welding and cutting are equipped with power supply equipment that weighs about 1 cubic meter and weighs about 50 kg, and are water-cooled to cope with the large amount of heat generated, making them difficult to maintain and use. is the current situation.

From the viewpoint of efficiency and miniaturization, it is desirable to use a switching power supply instead of a conventional dropper power supply as the power supply for driving the semiconductor laser. However, in a switching power supply, the power supply current is turned on and off by a switching element, and the resulting pulse waveform is smoothed by a coil and a capacitor to reduce ripples in the output current. Therefore, in order to obtain a sufficiently clean power output, it is necessary to increase the values of the coil inductance and capacitor capacitance, and because of the delay caused by the coil and capacitor, it is difficult to obtain a high-speed pulse waveform power output. It is said. A general-purpose switching power supply is sometimes used, but while it is efficient and can be made compact, the power supply startup time is slow at several milliseconds or more, making it unsuitable for the high-speed pulse drive required for laser processing. . Furthermore, a multi-phase power supply using a plurality of switching elements and coils constituting a switching power supply has also been proposed, as shown in, for example, Patent Document 1 or 2 below.

JP2012-050207 JP2014-171351

Even the multiphase type described above is not suitable for high-speed pulse driving.

The present invention was made in order to solve these conventional problems. For example, by using a coil and capacitor whose inductance and capacitance are as small as possible, the resulting unsmoothed output waveform with many ripples can be corrected by adjusting the power supply voltage appropriately. The present invention provides a method for obtaining a sufficiently clean power output with a fast rise time, for example, by combining a plurality of switching elements that are set or selected and turned on and off in a multiphase manner.

To achieve this objective, the high-speed switching power supply of the present invention includes a primary power supply, a plurality of switching elements (multiple phases) connected to the primary power supply, a plurality of coils (multiple phases) connected to the plurality of switching elements, and a plurality of coils. It is composed of one connected capacitor (including a set of capacitors), and by setting or selecting the power supply voltage or primary power supply voltage according to the number of coils (number of coil phases), it is possible to calculate the sum of the current flowing through multiple coils. It has means to keep the output current (output current: current flowing through the load) constant. For example, switching elements are arranged in parallel, coils are connected in series to each switching element, and the plurality of coils are arranged in parallel. Then, for example, a load is connected in parallel with the capacitor so that the currents flowing through the multiple coils are combined or combined and flow to the load (for example, a laser or a laser diode).

Here, for example, each of the plurality of switching elements (multiple phases) has a function of changing or controlling the current of the coil connected to the switching element, and a drive-side switching element that increases the current of the coil. It is composed of a switching element on the freewheeling side that reduces the current in the coil. For example, when the switching element on the driving side is ON and the switching element on the freewheeling side is OFF, current flows from the primary power source to the coil via the switching element on the driving side, energy is stored in the coil, and the switching element on the driving side is turned off. When the switching element on the freewheeling side is OFF and the switching element on the freewheeling side is ON, the current may continue to flow from the switching element on the freewheeling side through the coil while releasing energy. Furthermore, a diode can be used instead of the switching element on the freewheeling side.

In the switching power supply of the present invention, the number of coils (number of phases) is N, the waveforms of the currents flowing through the N coils are the same, the phases are shifted by 2π/N, and the power supply voltage is output voltage x ( 1+(number of coils where current decreases)/(number of coils where current increases)). Output voltage is the voltage applied to the load. The current flowing through the N coils is controlled so that the number of coils in which the current decreases and the number of coils in which the current increases are the same at any timing. Alternatively, the current flowing through N coils increases in the first half NON/N of the coil current increase/decrease cycle for NON and NOFF satisfying N=NON+NOFF (NON and NOFF are natural numbers), and decreases in the second half NOFF/N. controlled. Here, at any timing, the number of coils whose current increases is NON, and the number of coils whose current decreases is NOFF.

Furthermore, the switching power supply of the present invention can include means for controlling the sum of currents flowing through a plurality of coils (output current: current flowing through a load) to a target value by controlling a plurality of switching elements.

Further, by changing the power supply voltage or the primary power supply voltage, it is possible to have means for controlling the sum of the currents flowing through the plurality of coils to a target value by changing the rate of increase of the current flowing through the coils, for example.

Furthermore, by changing the resistance value (resistance value of the resistor) inserted or connected (for example, connected in series) to multiple switching elements, for example, the rate of decrease of the current flowing through the coils can be changed, and the current flowing through the multiple coils can be changed. It is possible to have means for controlling the total sum to a target value.

Next, it has a means to speed up the startup by changing the power supply voltage or the primary power supply voltage, and further has a means to speed up the startup by changing the resistance value inserted or connected (for example, connected in series) to multiple switching elements. It is possible to have a means to speed up the process.

For example, by controlling the plurality of switching elements, changing the power supply voltage, and changing the inserted resistance value, when the sum of the currents flowing through the plurality of coils is less than the target value, a larger current flows through the coil. When the sum of the currents flowing through the coils is larger than the target value, the coils are operated so that less current flows, and the sum of the currents flowing through the plurality of coils is controlled to the target value.

Furthermore, the switching power supply of the present invention to achieve this object has the characteristic that the resonance frequency determined by the coil and the capacitor is 0.1 times or more of the switching frequency, and the number of phases (operation It has the feature of increasing or decreasing the number of phases), the feature of changing the clock cycle according to the output current, the feature of matching the current value flowing through multiple coils, and the feature of realizing phase compensation by digital control. can have.

Then, it is possible to realize a laser device characterized in that a laser is driven by a switching power supply having the above means. The load, laser or laser diode, for example, is connected in series with the plurality of coils and in parallel with the capacitor.

According to the present invention, high-speed pulse drive using a switching power supply is possible for a high-output semiconductor laser device, and precise laser processing can be realized. Furthermore, by employing a switching power supply, efficiency is increased, and miniaturization and cost reduction are possible.

1 is a diagram showing a circuit of a switching power supply or a laser device in a first embodiment of the present invention. FIG. FIG. 3 is a diagram showing the operation timing of the switching power supply in the first embodiment of the present invention. FIG. 3 is a diagram showing current composition of the switching power supply in the first embodiment of the present invention. FIG. 7 is a diagram showing a circuit of a switching power supply or a laser device in a second embodiment of the present invention. FIG. 7 is a diagram showing one voltage-PWM conversion circuit in a second embodiment of the present invention. FIG. 7 is a diagram showing the operation timing of a switching power supply in a second embodiment of the present invention. It is a figure which shows the circuit of the switching power supply or laser device in the 3rd Embodiment of this invention. FIG. 7 is a diagram showing one second voltage-PWM conversion circuit in a third embodiment of the present invention. FIG. 7 is a diagram showing the operation timing of a switching power supply in a third embodiment of the present invention. FIG. 7 is a diagram showing other operation timings of the switching power supply in the third embodiment of the present invention. It is a figure showing the circuit of a switching power supply or a laser device in a 4th embodiment of the present invention. FIG. 7 is a diagram showing a third voltage-PWM conversion circuit in the fourth embodiment of the present invention. It is a figure which shows the operation timing of the switching power supply in the 4th Embodiment of this invention. It is a figure showing other operation timings of a switching power supply in a 4th embodiment of the present invention. FIG. 3 is a diagram showing how the switching power supply according to the present invention starts up faster. FIG. 3 is a diagram for explaining a reduction in the number of phases and a reduction in clock cycle in a low current region of the switching power supply of the present invention. FIG. 3 is a diagram showing a method of aligning currents in a switching power supply according to the present invention. FIG. 2 is a diagram showing a configuration of phase compensation of a switching power supply according to the present invention.

Hereinafter, a switching power supply or a laser device according to a first embodiment of the present invention will be described with reference to FIGS. 1 to 3.

The switching power supply of the first embodiment includes a primary power supply, a plurality of switching elements connected to the primary power supply, a plurality of coils connected to the plurality of switching elements, and one capacitor connected to the plurality of coils. This device has a structure in which the total sum of currents flowing through a plurality of coils is made constant by means of setting or selecting a voltage.

First, the configuration of the switching power supply or switching power supply device of the first embodiment will be described using FIG. 1.

In FIG. 1, 10 is a primary power source. The primary power source 10 is composed of a power transformer 11, which has a plurality of taps drawn out on the secondary side, and has a structure in which the output voltage of the secondary side can be selected. Reference numeral 12 denotes a switch which, as will be described later, allows setting or selection of the power supply voltage to be applied to the switching power supply or the main body of the switching power supply by selecting a plurality of taps prepared on the secondary side of the power transformer 11. be. 13, 14, 15, and 16 are rectifier diodes, which perform bridge type rectification. A capacitor 17 converts the pulsating current obtained by the rectifying diodes 13, 14, 15, and 16 into direct current to obtain a primary power supply voltage to be applied to the switching power supply or the main body of the switching power supply.

18 is an FPGA (field programmable gate array), which has a function of controlling the entire system. A timing generator 19 has a function of generating timing pulses having different phases at a period of about 1 μs to 3 μs.

20, 21, 22, and 23 are level conversion circuits that convert timing pulses given from the timing generator 19 into pulses for driving switching elements. Reference numerals 24, 25, 26, and 27 are switching elements on the driving side, which are P-channel FETs, and whose source terminals are connected to the primary power supply voltage. Reference numerals 28, 29, 30, and 31 are switching elements on the freewheeling side, and N-channel FETs are used, the drain terminals of which are connected to the drain terminals of the switching elements 24, 25, 26, and 27 on the driving side. Diodes 32, 33, 34, and 35 are on the freewheeling side, and are connected in parallel to switching elements 28, 29, 30, and 31 on the freewheeling side, which realize high-speed current switching using Schottky diodes. Although the diode on the freewheeling side operates integrally with the switching element on the freewheeling side, it is also possible to omit the switching element on the freewheeling side and give that role only to the diode on the freewheeling side. 36, 37, 38, and 39 are coils having multiple windings around the core, each having the same appropriate inductance value, and connected to the drain terminals of the switching elements 24, 25, 26, and 27 on the driving side or to the reflux. It has the role of changing the current flowing from the drain terminals of the side switching elements 28, 29, 30, and 31 with a gentle slope. A capacitor 40 has the role of receiving current flowing from the coils 36, 37, 38, and 39 to maintain a constant voltage. It goes without saying that a plurality of capacitors can be arranged in series or in parallel as long as they have substantially the same function as capacitor 40. 41 is a semiconductor laser diode serving as a load, which is connected in series with the coils 36, 37, 38, and 39 and in parallel with the capacitor 40; It has a property that the voltage at both ends gradually increases from about 1.8V to about 2.1V. In order to obtain larger optical output, multiple semiconductor laser diodes may be connected in series, and in that case, a larger applied voltage is required depending on the number of connections. The switching element 24 on the drive side and the switching element 28 on the freewheeling side constitute one, one set, or one phase switching element, and the switching element 25 on the drive side and the switching element 29 on the freewheeling side constitute one or one set. Alternatively, a one-phase switching element is configured, and the driving-side switching element 26 and the free-wheeling side switching element 30 constitute one, one set, or one-phase switching element, and the driving-side switching element 27 and the free-wheeling side switching element 27 are connected to the free-wheeling side. Together with the switching element 31, one, one set, or one phase of switching element is configured.

Next, the operation of the switching power supply of the first embodiment will be explained using FIG. 2.

In FIG. 2, 42 is a clock having a period of about 10 ns to 100 ns, which is output from the timing generator 19, and is the basis of the operation of the high-speed switching power supply. 43 is a timing pulse outputted from the timing generator 19, which is applied to the driving side switching element 24 through the level conversion circuit 20. When this timing pulse is ON, the driving side switching element 24 is turned ON, applying the primary power supply voltage to the coil 36, and increasing the coil current i1. 44 is a timing pulse output from the timing generator 19, which is applied to the freewheeling side switching element 28 through the level conversion circuit 20. When this timing pulse is ON, the freewheeling side switching element 28 is turned ON and absorbs the coil current i1 flowing through the coil 36. 45 indicates a change in the current i1 of the coil 36, and the current increases and decreases repeatedly due to the operation of the switching element 24 on the drive side and the switching element 28 on the return side. When the switching element 24 on the driving side is ON, the switching element 28 on the freewheeling side is OFF, and when the switching element 24 on the driving side is OFF, the switching element 28 on the freewheeling side is ON; When the switching element 24 and the switching element 28 on the freewheeling side are turned on at the same time, a breakthrough current is generated in the switching element 24 on the driving side and the switching element 28 on the freewheeling side. Therefore, the timing 44 rises later than the fall of the timing 43, and the timing 44 It is desired that the timing 43 falls earlier than the rising edge.

46 is a timing pulse outputted from the timing generator 19, and is applied to the driving side switching element 25 through the level conversion circuit 21. When this timing pulse is ON, the driving side switching element 25 is turned ON, applying the primary power supply voltage to the coil 37, and increasing the coil current i2. Reference numeral 47 denotes a timing pulse output from the timing generator 19, which is applied to the freewheeling side switching element 29 through the level conversion circuit 21. When this timing pulse is ON, the freewheeling side switching element 29 is turned ON and absorbs the coil current i2 flowing through the coil 37. 48 indicates a change in the current i2 of the coil 37, and the current increases and decreases repeatedly due to the operation of the switching element 25 on the driving side and the switching element 29 on the freewheeling side. Timing 46 and timing 47 are operated 1/4 cycle (phase is 2π/4) later than timing 43 and timing 44.

49 is a timing pulse outputted from the timing generator 19, and is applied to the driving side switching element 26 through the level conversion circuit 22. When this timing pulse is ON, the driving side switching element 26 is turned ON, applying the primary power supply voltage to the coil 38, and increasing the coil current i3. Reference numeral 50 denotes a timing pulse output from the timing generator 19, which is applied to the freewheeling side switching element 30 through the level conversion circuit 22. When this timing pulse is ON, the freewheeling side switching element 30 is turned ON and absorbs the coil current i3 flowing through the coil 38. 51 indicates a change in the current i3 of the coil 38, and the current increases and decreases repeatedly due to the operation of the switching element 26 on the drive side and the switching element 30 on the freewheeling side. Timing 49 and timing 50 are operated 1/2 cycle (phase is 2π/2) later than timing 43 and timing 44.

52 is a timing pulse output from the timing generator 19, and is applied to the driving side switching element 27 through the level conversion circuit 23. When this timing pulse is ON, the driving side switching element 27 is turned ON, applying the primary power supply voltage to the coil 39, and increasing the coil current i4. A timing pulse 53 is output from the timing generator 19 and is applied to the freewheeling side switching element 31 through the level conversion circuit 23. When this timing pulse is ON, the freewheeling side switching element 31 is turned ON and absorbs the coil current i4 flowing through the coil 39. 54 indicates a change in the current i4 of the coil 39, and the current increases and decreases repeatedly due to the operation of the switching element 27 on the drive side and the switching element 31 on the freewheeling side. Timing 52 and timing 53 are operated 3/4 cycle (phase is 2π×3/4) later than timing 43 and timing 44.

In the above description, it has been explained that the number of coils N=4, and the waveforms of the currents flowing through N coils or N phase coils are the same, and the phases are shifted by 2π/N (2π/4).

In addition to the above-described operations, the switching power supply of the first embodiment also sets or selects the power supply voltage as a means for keeping the sum of the currents flowing through the plurality of coils constant. In other words, set or select the power supply voltage VS so that VS=VO×(1+(NOFF/NON)), where the output voltage is VO, the number of coils where the current increases is NON, and the number of coils where the current decreases is NOFF. However, this principle will be explained using FIGS. 2 and 3.

In FIG. 1, a primary power supply 10 supplies a primary power supply voltage VS to a switching power supply or a main body of a switching power supply. When the driving side switching element 24 is ON, the power supply voltage of the primary power source is VS, the laser power output (load or laser diode 41 output voltage, voltage applied to the load or laser diode 41) is VO, and the inductance value of the coil 36 is L. The current i1 flowing through the coil 36 shown at 45 in FIG. 2 becomes an increase amount i1ON of the following equation after T seconds.

i1ON=(VS-VO)×T/L

When the driving side switching element 24 is OFF, the current i1 flowing through the coil 36 shown at 45 in FIG. 2 becomes a decrease amount i1OFF expressed by the following equation after T seconds.

i1OFF=-(VO×T/L)

Since the same primary power supply voltage VS is applied to the current i2 flowing through the coil 37 shown at 48 in FIG. 2, the current i3 flowing through the coil 38 shown at 51, and the current i4 flowing through the coil 39 shown at 54, Increase in current i2 after T seconds i2ON, decrease in current i2 after T seconds i2OFF, increase in current i3 after T seconds i3ON, decrease in current i3 after T seconds i3OFF, increase in current i4 after T seconds i4ON , the amount of decrease i4OFF of the current i4 after T seconds is the same as i1ON and i1OFF in the case of i1. That is, in any coil 36, 37, 38, 39, the amount of increase iLON after T seconds is (VS - VO) x T/L, and the amount of decrease iLOFF after T seconds is -(VO x T/L). . The laser output current is the sum of i1, i2, i3, and i4, but in order to maintain a constant value, the current flowing through each coil shown at 45, 48, 51, and 54 in FIG. The sum (iLON×NON+iLOFF×NOFF) of the total amount of current increase in the current increase coil and the total amount of current decrease in the current decrease coil must be 0. That is, the following equation needs to hold true.

iLON ×NON+iLOFF×NOFF=(VS-VO)×T/L×NON-VO×T/L×NOFF=0

From this, VS=VO×(1+(NOFF/NON)) is obtained.

In the switching power supply of the first embodiment, the timings of the coil currents i1, i2, i3, and i4 shown at 45, 48, 51, and 54 in FIG. 2 are repeated such that the first half is increased and the second half is decreased. Since it is considered to have a cycle, or a repeating cycle in which the first half is increased and the second half is decreased, NON=NOFF=2, so VS becomes the following equation.

VS=2×VO

FIG. 3 shows changes in coil currents i1, i2, i3, and i4 shown at 45, 48, 51, and 54 in FIG. 2 in the switching power supply of the first embodiment. As shown in FIG. 3, the currents i1, i2, i3, and i4 flowing through the coils 36, 37, 38, and 39 become triangular waves with equal up and down slopes, that is, they become triangular waves with equal waveforms, and the sum of the currents flowing into the capacitor 40 The laser output current i is i1+i2+i3+i4, and takes a flat constant value in the section a-b.

The supply voltage VS can be set or selected differently, if NON=3, NOFF=1, i.e. the coil current is equal with repeated cycles increasing in the first half and decreasing in the second half. If the waveform is a triangular wave, VS becomes the following equation, and the laser output current can be kept at a flat constant value.

VS=4/3×V0

In other words, if the power supply voltage is 4/3 times the output voltage VO, and the current has the same waveform with the phase increasing by 3/4 in the first half of the repetition cycle and decreasing in the second half by π/2, the same waveform will be obtained. A constant value can be obtained for the laser output current.

Since there are some resistance values in the drive side switching elements 24, 25, 26, 27, the freewheeling side switching elements 28, 29, 30, 31, and the coils 36, 37, 38, 39 shown in FIG. It may be corrected to the power supply voltage. Furthermore, if the output voltage required by the semiconductor laser diode has a fixed value and the waveform of the current flowing through the coil is one type, it is sufficient to have a corresponding fixed primary power supply voltage. tap and switch 12 are not required. However, if the output voltage required by the semiconductor laser diode changes, or if the waveform of the current flowing through the coil has multiple types and can be changed, the primary power supply voltage should be adjusted so that multiple corresponding primary power supply voltages can be obtained. (tap) must be configured.

The number of coils can be selected depending on the configuration of the semiconductor laser diode serving as the load and the amount of tolerable ripple. The power supply voltage according to the combination of NON and NOFF and the combination of increase time and decrease time with respect to the number of coils is given by, for example, Table 1 below. In Table 1, the current waveforms flowing through each coil are the same, and the phases are shifted by 2π/number of coils.

In a conventional switching power supply having a plurality of coils, the current flowing through the coils is increased by increasing the inductance of the coils corresponding to the coils 36, 37, 38, and 39 of the present invention and the capacitance of the capacitor corresponding to the capacitor 40. The sum is kept constant at the average value.

The fast switching power supply according to the invention has the different feature that the sum of the currents at each moment is kept at a constant value. Therefore, the inductance of the coils 36, 37, 38, and 39 and the capacitance of the capacitor 40 of the present invention can be given minute values. This feature makes it possible to speed up the rise time of the laser power output, making it possible to realize a high-speed switching power supply.

A switching power supply or laser device according to a second embodiment of the present invention will be described below with reference to FIGS. 4 to 6.

The switching power supply of the second embodiment includes a primary power supply, a plurality of switching elements connected to the primary power supply, a plurality of coils connected to the plurality of switching elements, and one capacitor connected to the plurality of coils. A device with a structure that uses means for setting or selecting the power supply voltage and means for controlling the switching elements to keep the sum of the currents flowing through multiple coils constant, that is, to reduce the amount of ripple, and to control it to a target value. It is.

First, the configuration of the switching power supply according to the second embodiment will be described using FIG. 4.

In FIG. 4, 110 is a primary power source. The primary power supply 110 is composed of a power transformer 111, which has a plurality of taps drawn out on the secondary side, and has a structure in which the output voltage of the secondary side can be selected. 112 is a switch which, as will be described later, allows setting or selection of the power supply voltage to be applied to the switching power supply or the main body of the switching power supply by selecting a plurality of taps prepared on the secondary side of the power transformer 111. be. 113, 114, 115, and 116 are rectifier diodes, which perform bridge type rectification. A capacitor 117 converts the ripple current obtained by the rectifying diodes 113, 114, 115, and 116 into direct current to obtain a primary power supply voltage to be applied to the switching power supply or the main body of the switching power supply.

Reference numeral 118 denotes an FPGA (field programmable gate array), which has a function of controlling the entire system. A timing generator 119 has a function of generating timing pulses having different phases at a period of about 1 μs to 3 μs. Reference numerals 120, 121, and 122 are voltage-PWM (Pulse Width Modulation) conversion circuits, which have a function of converting an output difference voltage value into a pulse time width. Level conversion circuits 123, 124, and 125 convert timing pulses given from the voltage-PWM conversion circuits 120, 121, and 122 into pulses for driving switching elements. Reference numerals 126, 127, and 128 are switching elements on the driving side, which are P-channel FETs, and whose source terminals are connected to the primary power supply voltage. Reference numerals 129, 130, and 131 are switching elements on the freewheeling side, and N-channel FETs are used, the drain terminals of which are connected to the drain terminals of the switching elements 126, 127, and 128 on the driving side. Diodes 132, 133, and 134 are on the freewheeling side, and are connected in parallel to switching elements 129, 130, and 131 on the freewheeling side, which realize high-speed current switching using Schottky diodes. 135, 136, and 137 are coils having multiple windings around the core, each having the same appropriate inductance value, and connected to the drain terminals of the switching elements 126, 127, and 128 on the drive side or the switching elements on the free circulation side. It has the role of changing the current flowing from the drain terminals 129, 130, and 131 with a gentle slope. A capacitor 138 has the role of receiving current flowing from the coils 135, 136, and 137 to maintain a constant voltage. 139 is a semiconductor laser diode serving as a load, which is connected in series with the coils 135, 136, and 137 and in parallel with the capacitor 138, and as the current increases, as the current is applied, 1. It has the property of gradually rising from a voltage across both ends of about 8V to about 2.1V. Reference numeral 140 denotes a current detection resistor for measuring the current flowing through the semiconductor laser diode 139, and a resistor capable of passing a large current of approximately several tens of mΩ is used. In order to obtain larger optical output, multiple semiconductor laser diodes may be connected in series, and in that case, a larger applied voltage is required depending on the number of connections. The switching element 126 on the drive side and the switching element 129 on the freewheeling side constitute one, one set, or one phase switching element, and the switching element 127 on the drive side and the switching element 130 on the freewheeling side constitute one or one set. Alternatively, a one-phase switching element is configured, and the driving side switching element 128 and the freewheeling side switching element 131 constitute one, one set, or one phase switching element.

141 is an A/D converter that converts the voltage across the current detection resistor 140 into a numerical value, and converts the current flowing through the semiconductor laser diode 139 into a numerical value. A phase compensation circuit 142 is composed of a digital filter, and has the role of performing phase compensation so that the control loop does not oscillate even if the current value flowing through the semiconductor laser diode 139 is strongly controlled. A laser current register 143 holds the output of the phase compensation circuit 142. Reference numeral 144 denotes an output value register that holds the current value to be passed through the target semiconductor laser diode 139. 145 is a subtraction circuit that compares the value of the laser current register 143 with the value of the output value register 144 and outputs the difference voltage value between the target current value flowing through the semiconductor laser diode 139 and the actual current value flowing through the semiconductor laser diode 139. .

Next, the configuration of the voltage-PWM conversion circuit for the switching power supply of the second embodiment will be explained using FIG. 5.

In FIG. 5, the FPGA 118, timing generator 119, phase compensation circuit 142, laser current register 143, output value register 144, subtraction circuit 145, and level conversion circuit 123 are those shown in FIG. FIG. 5 shows the configuration of the voltage-PWM conversion circuit 120. Although the voltage-PWM conversion circuits 121 and 122 are not shown, the voltage-PWM conversion circuits 121 and 122 are similar to the voltage-PWM conversion circuit 120. It has the same or the same configuration as .

The voltage-PWM conversion circuits 120, 121, and 122 have a function of converting the output difference voltage value into a pulse time width. A counter 146 that constitutes the voltage-PWM conversion circuits 120, 121, and 123 receives a reference pulse from the timing generator 119, and outputs a numerical value that increases with time to generate drive and circulation timing. A subtraction circuit 147 compares the output from the counter 146 and the difference voltage value output from the subtraction circuit 145 by subtraction, and outputs a pulse when they become equal.

148 is a flip-flop (F.F.) that provides drive timing; it receives the reference pulse from the timing generator 119 and the output of the subtraction circuit 147, and if the difference voltage value is small, it generates a short width pulse; It has a function of shortening the driving time of the switching elements 126, 127, 128, generating a pulse with a longer width if the difference voltage value is large, and extending the driving time of the driving side switching elements 126, 127, 128. Reference numeral 149 denotes a flip-flop (F.F.) that provides reflux timing, which receives the reference pulse from the timing generator 119 and the output of the subtraction circuit 147, and generates a long width pulse if the difference voltage value is small. It has a function of extending the driving time of the switching elements 129, 130, 131, and generating short width pulses if the difference voltage value is large, thereby shortening the driving time of the switching elements 129, 130, 131 on the free circulation side.

Next, the operation of the switching power supply according to the second embodiment will be explained using FIG.

First, the operation related to the voltage-PWM conversion circuit 120 in FIG. 4 will be explained. In FIG. 6, 150 is a clock having a period of about 10 ns to 100 ns, which is output from the timing generator 119, and is the basis of the operation of the switching power supply. Reference numeral 151 indicates the timing of resetting the counter 146 and setting the flip-flop 148 in FIG. 5, and in this embodiment, pulses are regularly given from the timing generator 119 every 3 μs. Reference numeral 152 indicates the reset timing of the flip-flop 148, and the generation position varies depending on the pulse output from the subtraction circuit 147. Reference numeral 153 indicates the input of the output signal of the flip-flop 148 to the driving side switching element 126 through the level conversion circuit 123, and it is repeatedly turned on and off depending on the set timing and reset timing of the flip-flop 148, and when this input is on, The driving side switching element 126 is driven. Reference numeral 154 indicates the timing of setting the flip-flop 149, and in this embodiment, the generation position varies depending on the pulse output from the subtraction circuit 147. Reference numeral 155 indicates the reset timing of the flip-flop 149, and a pulse is regularly given from the timing generator 119 every 3 μs. Reference numeral 156 indicates the input of the output signal of the flip-flop 149 to the switching element 129 on the freewheeling side through the level conversion circuit 123, and it is repeatedly turned on and off depending on the set timing and reset timing of the flip-flop 149, and when this input is on, The switching element 129 on the reflux side is driven. Reference numeral 157 indicates a change in the current i1 of the coil 135, and the current increases and decreases repeatedly due to the operation of the switching element 126 on the drive side and the switching element 129 on the freewheeling side. When the switching element 126 on the driving side is ON, the switching element 129 on the freewheeling side is OFF, and when the switching element 126 on the driving side is OFF, the switching element 129 on the freewheeling side is ON; When the switching element 126 on the drive side and the switching element 129 on the freewheeling side are turned on at the same time, a breakthrough current is generated in the switching element 126 on the drive side and the switching element 129 on the freewheeling side. It is desirable that the reset timing 155 of the flip-flop 149 be generated slightly later than the timing 152 and the reset timing 155 of the flip-flop 149 be generated slightly earlier than the set timing 151 of the flip-flop 148.

Next, the operation related to the voltage-PWM conversion circuit 121 in FIG. 4 will be explained. Like the voltage-PWM conversion circuit 120, the voltage-PWM conversion circuit 121 includes a counter 146, a subtraction circuit 147, and flip-flops 148 and 149. It operates similarly to the PWM conversion circuit 120. In FIG. 6, the set-reset timing of the flip-flop 148 of the voltage-PWM conversion circuit 121 and the set-reset timing of the flip-flop 149 are basically the set-reset timing of the flip-flop 148 of the voltage-PWM conversion circuit 120, and the set-reset timing of the flip-flop 148 of the voltage-PWM conversion circuit 120. The timing is delayed by 1/3 cycle compared to the set/reset timing of the flip-flop 149. The timing at which the flip-flop 148 is reset and the timing at which the flip-flop 149 is set will vary in generation position depending on the pulse output from the subtraction circuit 147, but illustration thereof is omitted. Reference numeral 158 in FIG. 6 indicates an input signal to the driving side switching element 127, and when this input is ON from setting to resetting the flip-flop 148, the driving side switching element 127 is driven. Reference numeral 159 indicates an input signal to the switching element 130 on the freewheeling side, and when this input is ON from the setting to the reset of the flip-flop 149, the switching element 130 on the freewheeling side is driven. Reference numeral 160 indicates a change in the current i2 of the coil 136, and the current increases and decreases repeatedly due to the operation of the switching element 127 on the driving side and the switching element 130 on the freewheeling side. These pulse signals 158 and 159 basically operate at timings that are 1/3 cycle later than the pulse signals shown at timing 153 and timing 156 (signals shown by solid lines).

Next, the operation related to the voltage-PWM conversion circuit 122 in FIG. 4 will be explained. Like the voltage-PWM conversion circuit 120, the voltage-PWM conversion circuit 122 includes a counter 146, a subtraction circuit 147, and flip-flops 148 and 149. It operates similarly to the PWM conversion circuit 120. In FIG. 6, the set-reset timing of the flip-flop 148 of the voltage-PWM conversion circuit 122 and the set-reset timing of the flip-flop 149 are basically the set-reset timing of the flip-flop 148 of the voltage-PWM conversion circuit 120, and the set-reset timing of the flip-flop 148 of the voltage-PWM conversion circuit 120. The timing is delayed by 2/3 period with respect to the set/reset timing of the flip-flop 149. The timing at which the flip-flop 148 is reset and the timing at which the flip-flop 149 is set will vary in generation position depending on the pulse output from the subtraction circuit 147, but illustration thereof is omitted. Reference numeral 161 in FIG. 6 indicates an input signal to the driving side switching element 128, and when this input is ON corresponding to the setting to reset of the flip-flop 148, the driving side switching element 128 is driven. Reference numeral 162 indicates an input signal to the freewheeling side switching element 131, and when this input is ON from the setting to the reset of the flip-flop 149, the freewheeling side switching element 131 is driven. Reference numeral 163 indicates a change in the current i3 of the coil 137, and the current increases and decreases repeatedly due to the operation of the switching element 128 on the drive side and the switching element 131 on the freewheeling side. These pulse signals 161 and 162 basically operate at timings that are 2/3 cycles later than the pulse signals shown at timing 153 and timing 156 (signals shown by solid lines).

The configuration in FIG. 4 has three phases, and the currents i1, i2, i3 flowing through the coils 135, 136, 137 (coil currents 157, 160, 163 in FIG. 6) are basically the current target. When value control or feedback control is not performed, the waveforms are the same, increasing or rising by 2/3 in the first half and decreasing or falling by 1/3 in the latter half, but the phase is shifted by 2π/3 (as shown by the solid line in Figure 6). coil current), select the number of phases 3, NON2, NOFF1 from Table 1 of the power supply voltage and increase time and decrease time according to the number of coils, and calculate the voltage of the primary power supply 110 by (3/2) × output It is assumed that the voltage is VO (the voltage drop due to the current detection resistor 140 is negligibly small). As described above, if the circuits shown in FIGS. 4 and 5 operate at the timing shown in FIG. , 137, and the capacitor 138 receives the currents i1, i2, and i3 flowing from the coils 135, 136, and 137 to maintain a constant voltage, allowing a ripple-free current to flow into the semiconductor laser diode 139.

On the other hand, a function is required to control this current so that it reaches a target value. In order to realize this function, a laser current register 143 is prepared through a current detection resistor 140, an A/D converter 141, and a phase compensation 142, and a value proportional to the current flowing through the laser diode 139 is set in the laser current register 143. The value of the laser current register 143 is compared with the target laser current preset in the output value register 144 in a subtraction circuit 145 to generate a difference voltage value.

When the actual current value flowing through the laser diode 139 is larger than the target current value, the large value is held in the laser current register 143, and is compared with the value in the output value register 144 in the subtraction circuit 145, and a small difference voltage value is output. . In each of the voltage-PWM conversion circuits 120, 121, and 122, the output given from the counter 146 and the difference voltage value of the subtraction circuit 145 are compared by subtraction in the subtraction circuit 147, and when they become equal, a pulse is output. In the voltage-PWM conversion circuit 120, specifically, it is a reset pulse to the flip-flop 148 at the early position a shown at 152 in FIG. 6, and a short width pulse c to the input of the driving side switching element 126 shown at 153. Become. Furthermore, it becomes a set pulse to the flip-flop 149 at an early position of e shown at 154, and a long width pulse of g to the input of the freewheeling side switching element 129 shown at 156. In other words, if the timing of the output from the subtraction circuit 147 in the voltage-PWM conversion circuits 120, 121, 122 is earlier, the time width during which the drive side switching elements 126, 127, 128 are turned on becomes narrower, and the freewheeling side switching elements 129, 130, 131 are turned on. As a result, the current i1 of the coil 135 shown at 157 becomes a small value of k.

When the actual current value flowing through the laser diode 139 is smaller than the target current value, the small value is held in the laser current register 143, and is compared with the value in the output value register 144 in the subtraction circuit 145 to output a large difference voltage value. . In each of the voltage-PWM conversion circuits 120, 121, and 122, the output given from the counter 146 and the difference voltage value of the subtraction circuit 145 are compared by subtraction in the subtraction circuit 147, and when they become equal, a pulse is output. In the voltage-PWM conversion circuit 120, specifically, it becomes a reset pulse to the flip-flop 148 at the slow position b shown at 152 in FIG. 6, and a long width pulse d to the input of the driving side switching element 126 shown at 153. Become. Further, it becomes a set pulse shown at 154 to the flip-flop 149 at a slow position of f, and becomes a short width pulse of h to the input of the freewheeling side switching element 129 shown at 156. That is, if the timing of the output from the subtraction circuit 147 in the voltage-PWM conversion circuits 120, 121, 122 is delayed, the time width during which the drive side switching elements 126, 127, 128 are turned on becomes wider, and the freewheeling side switching elements 129, 130, 131 are turned on. As a result, the current i1 of the coil 135 shown at 157 has a large value of j.

In this way, by controlling the operation timings of the driving side switching elements 126, 127, 128 and the freewheeling side switching elements 129, 130, 131, the currents i1, i2, i3 flowing through the plurality of coils 135, 136, 137 are controlled. The total sum can be controlled to the target value.

Incidentally, the voltage applied to the semiconductor laser diode serving as a load has a property of gradually rising from about 1.8V to about 2.1V as the applied current to the semiconductor laser diode increases. Therefore, as the current applied to the semiconductor laser diode increases, the primary power supply voltage of the switching power supply in Figure 4 increases from (3/2) x 1.8V = 2.7V to (3/2) x 2.1V = 3.15V. Change is desirable. However, since this amount of change is small, the primary power supply voltage can be set to (3/2)×1.95V=2.925V by setting the output voltage to 1.95V, which is the average value of the voltage applied to the semiconductor laser diode. Furthermore, when the target applied current to the semiconductor laser diode 139 is changed, the current flowing through the semiconductor laser diode 139 is controlled to be maintained at the target laser current, so when the target laser current is low, the increase time shown in Table 1 above is The increase time is shorter and the decrease time is longer, and when the target laser current is large, the increase time is longer and the decrease time is shorter than the increase time in Table 1 above. Therefore, the sum of the currents flowing through the plurality of coils is not constant, and some ripples occur. However, the influence of the difference between the increase time shown in Table 1 and the controlled increase time is small, and the ripple can be maintained at a value of about 1%, so no problem occurs in driving the semiconductor laser diode. If the number of phases is 6 or more, the ripple can be further reduced to 0.1% or less.

To determine the primary power supply voltage of various switching power supplies using Table 1, the average value of the applied voltage of the semiconductor laser diode can be adopted as the output voltage VO as described above. No problem will occur even if any value between the lowest and highest values of is adopted as the output voltage VO.

A switching power supply or laser device according to a third embodiment of the present invention will be described below with reference to FIGS. 7 to 10.

The switching power supply of the third embodiment includes a primary power supply, a plurality of switching elements connected to the primary power supply, a plurality of coils connected to the plurality of switching elements, and one capacitor connected to the plurality of coils. A device that uses means for setting or selecting the power supply voltage and means for changing the power supply voltage to keep the sum of the currents flowing through multiple coils constant, that is, to reduce the amount of ripple, and has a structure that controls the current. It is.

First, the configuration of the switching power supply according to the third embodiment will be described using FIG. 7. In FIG. 7, 210 and 211 are primary power sources. As components of the primary power source 210 and the primary power source 211, reference numerals 212 and 213 are power transformers, and a plurality of taps are drawn out on the secondary side, so that the output voltage of the secondary side can be selected. . Reference numerals 214 and 215 are switches, and as described later, by selecting a plurality of taps prepared on the secondary side of the power transformers 212 and 213, the power supply voltage to be applied to the switching power supply or the main body of the switching power supply can be set or selected. It is meant to be. 216, 217, 218, 219, 220, 221, 222, and 223 are rectifier diodes, which perform bridge type rectification. 224 and 225 are capacitors that convert the ripple current obtained by the rectifier diodes 216, 217, 218, 219, 220, 221, 222, and 223 into direct current to obtain the primary power supply voltage to be applied to the switching power supply or the main body of the switching power supply. It is. The power transformer 213 of the primary power source 211 has a tap with a higher voltage than the tap of the power transformer 212 of the primary power source 210, and the primary power source 211 supplies a higher primary power voltage than the primary power source 210. 226, 227, and 228 are P-channel FETs, which are connected to the primary power source 210. 229, 230, and 231 are P-channel FETs, which are connected to the primary power source 211. The switching power supply is configured such that the voltage of the primary power supply 210 can be applied normally, that is, when target value control or feedback control of the current is not performed.

232 is an FPGA (field programmable gate array), which has a function of controlling the entire system. A timing generator 233 has a function of generating timing pulses having different phases at a period of about 1 μs to 3 μs. 234, 235, and 236 are second voltage-PWM (Pulse Width Modulation) conversion circuits, which have a function of converting the output difference voltage value into a pulse time width. 237, 238, and 239 are N-channel FETs that drive FETs 226, 227, and 228 upon receiving the outputs of second voltage-PWM conversion circuits 234, 235, and 236. 240, 241, and 242 are N-channel FETs that drive FETs 229, 230, and 231 upon receiving the outputs of second voltage-PWM conversion circuits 234, 235, and 236.

Level conversion circuits 243, 244, and 245 convert timing pulses given from the timing generator 233 into pulses for driving switching elements. 246, 247, and 248 are switching elements on the drive side, and P-channel FETs are used, and the source terminals are connected to the primary power supply voltages 210, 211 via the FETs 226, 227, and 228, or FETs 229, 230, and 231. . Reference numerals 249, 250, and 251 are switching elements on the free circulation side, and N-channel FETs are used, and the drain terminals are connected to the drain terminals of the switching elements 246, 247, and 248 on the driving side. Diodes 252, 253, and 254 are on the freewheeling side, and are connected in parallel to switching elements 249, 250, and 251 on the freewheeling side, which realize high-speed current switching using Schottky diodes. 255, 256, and 257 are coils having multiple windings around the core, each having the same appropriate inductance value, and connected to the drain terminals of the switching elements 246, 247, and 248 on the drive side or the switching elements on the return side. It has the role of changing the current flowing from the drain terminals of 249, 250, and 251 with a gentle slope. A capacitor 258 has the role of receiving current flowing from the coils 255, 256, and 257 to maintain a constant voltage. 259 is a semiconductor laser diode serving as a load, which is connected in series with the coils 255, 256, and 257 and in parallel with the capacitor 218, and as the current increases, 1. It has the property of gradually rising from a voltage across both ends of about 8V to about 2.1V. 260 is a current detection resistor that measures the current flowing through the semiconductor laser diode 259, and a resistor that can flow a large current of about several tens of mΩ is used. In order to obtain larger optical output, multiple semiconductor laser diodes may be connected in series, and in that case, a larger applied voltage is required depending on the number of connections. The switching element 246 on the drive side and the switching element 249 on the freewheeling side constitute one, one set, or one phase switching element, and the switching element 247 on the drive side and the switching element 250 on the freewheeling side constitute one or one set. Alternatively, a one-phase switching element is configured, and the driving side switching element 248 and the freewheeling side switching element 251 constitute one, one set, or one phase switching element.

261 is an A/D converter that converts the voltage across the current detection resistor 260 into a numerical value, and converts the current flowing through the semiconductor laser diode 259 into a numerical value. A phase compensation circuit 262 is composed of a digital filter, and has the role of performing phase compensation so that the control loop does not oscillate even if the current value flowing through the semiconductor laser diode 259 is strongly controlled. A laser current register 263 holds the output of the phase compensation circuit 262. Reference numeral 264 is an output value register that holds the current value to be passed through the target semiconductor laser diode 259. 265 is a subtraction circuit that compares the value of the laser current register 263 and the value of the output value register 264 and outputs the difference voltage value between the target current value flowing through the semiconductor laser diode 259 and the actual current value flowing through the semiconductor laser diode 259. .

Next, the configuration of the second voltage-PWM conversion circuit for the switching power supply of the third embodiment will be explained using FIG. 8.

In FIG. 8, the FPGA 232, timing generator 233, phase compensation circuit 262, laser current register 263, output value register 264, subtraction circuit 265, and FETs 237 and 240 are those shown in FIG. FIG. 8 shows the configuration of the second voltage-PWM conversion circuit 234, and although the second voltage-PWM conversion circuit 235 and 236 portions are not shown, the second voltage-PWM conversion circuit 235 and 236 have the same or the same configuration as the second voltage-PWM conversion circuit 234.

The second voltage-PWM conversion circuits 234, 235, and 236 have a function of converting the output difference voltage value into a pulse time width. The second voltage-PWM conversion circuits 234, 235, and 236 are configured by a counter 266 that receives the reference pulse from the timing generator 233 and outputs a numerical value that increases with time to generate timing for drive and circulation. . 267 is a subtraction circuit that compares the output from the counter 266 and the difference voltage value output from the subtraction circuit 265 by subtraction, and outputs a pulse when they become equal.

268 is a flip-flop (F.F.) that provides voltage change timing, and receives the reference pulse from the timing generator 233 and the pulse from the subtraction circuit 267, and if the difference voltage value is small, generates a short width pulse. It outputs to FETs 237, 238, and 239, and also generates a long width pulse and outputs it to FETs 240, 241, and 242. If the difference voltage value is large, it generates a long width pulse and outputs it to FETs 237, 238, and 239. It also has the function of generating short pulses and outputting them to FETs 240, 241, and 242. The outputs of the flip-flop 268 include outputs connected to FETs 237, 238, and 239 and outputs connected to FETs 240, 241, and 242, and these have opposite logics.

As a first application example of the switching power supply of the third embodiment, which sets or selects the power supply voltage and changes the increasing speed of the current flowing through the coils by means of changing the power supply voltage, the sum of the currents flowing through a plurality of coils is calculated. The operation of the switching power supply that is controlled to the target value will be explained using FIG. 9.

In FIG. 9, 269 is a clock output from the timing generator 233 and has a period of about 10 ns to 100 ns, and is the basis of the operation of the switching power supply. Reference numeral 270 indicates a signal output by the timing generator 233 at a constant timing, and when this signal is ON, the driving side switching element 246 is driven through the level conversion circuit 243. Reference numeral 271 indicates a signal output by the timing generator 233 at a constant timing, and when this signal is ON, the switching element 249 on the freewheeling side is driven through the level conversion circuit 243. Here, the operation of the second voltage-PWM conversion circuit 234 in FIG. 8 will be explained. In FIG. 9, reference numeral 272 indicates the timing of resetting the counter 266 and setting the flip-flop 268 in FIG. 8, and in this embodiment, pulses are regularly given from the timing generator 233 every 3 μs. Reference numeral 273 indicates the reset timing of the flip-flop 268, and the generation position varies depending on the pulse output from the subtraction circuit 267. Reference numeral 274 indicates an input to the FET 237 based on the output of the flip-flop 268, which repeats ON and OFF depending on the set timing and reset timing of the flip-flop 268, and when this output is ON, the FET 226 is turned ON via the FET 237. 275 indicates the input of the FET 240 by the output of the flip-flop 268, which is repeatedly turned ON and OFF in the opposite logic to the input of the FET 237 shown at 274 according to the set timing and reset timing of the flip-flop 268, and when this output is ON, the input is input to the FET 240 via the FET 240. to turn on FET229.

Reference numeral 276 indicates a change in the current i1 of the coil 255, and the current increases and decreases repeatedly due to the operation of the switching element 246 on the driving side and the switching element 249 on the freewheeling side. When the switching element 246 on the driving side is ON, the switching element 249 on the freewheeling side is OFF, and when the switching element 246 on the driving side is OFF, the switching element 249 on the freewheeling side is ON; When the switching element 246 on the drive side and the switching element 249 on the freewheeling side are turned on at the same time, a breakthrough current is generated in the switching element 246 on the drive side and the switching element 249 on the freewheeling side. It is desired that the timing 271 falls earlier than the rising edge.

Reference numeral 277 indicates a signal outputted by the timing generator 233 at a constant timing, and when this signal is ON, the driving side switching element 247 is driven through the level conversion circuit 244. Reference numeral 278 indicates a signal output by the timing generator 233 at a constant timing, and when this signal is ON, it drives the switching element 250 on the freewheeling side through the level conversion circuit 244. Timing 277 and timing 278 are operated 1/3 cycle (phase is 2π/3) later than timing 270 and timing 271.

The second voltage-PWM conversion circuit 235 in FIG. 7 receives the timing generation 233 and the difference voltage value that is the output of the subtraction circuit 265, and calculates the difference voltage value between the output from the counter 266 and the subtraction circuit 265 in the subtraction circuit 267. Compare by subtraction and output a pulse when they are equal. The timing of setting and resetting the flip-flop 268 of the second voltage-PWM conversion circuit 235 is basically 1/3 cycle later than the timing of setting and resetting the flip-flop 268 of the voltage-PWM conversion circuit 234, and The operation timing of FET 241 is delayed by 1/3 cycle with respect to the operation timing of FET 237 and FET 240. The timing of resetting the flip-flop 268 of the second voltage-PWM conversion circuit 235 varies in its generation position depending on the pulse output from the subtraction circuit 267, but this is not shown. 279 indicates a change in the current i2 of the coil 256, and the current repeatedly increases and decreases due to the operation of the driving side switching element 247 and the freewheeling side switching element 250.

Reference numeral 280 indicates a signal output by the timing generator 233 at a constant timing, and when this signal is ON, the driving side switching element 248 is driven through the level conversion circuit 245. Reference numeral 281 indicates a signal output by the timing generator 233 at a constant timing, and when this signal is ON, the switching element 251 on the freewheeling side is driven through the level conversion circuit 245. Timing 280 and timing 281 are operated 2/3 cycle (phase is 4π/3) later than timing 270 and timing 271.

The second voltage-PWM conversion circuit 236 in FIG. 7 receives the timing generation 233 and the difference voltage value that is the output of the subtraction circuit 265, and the difference voltage value of the output given from the counter 266 and the subtraction circuit 265 is calculated in the subtraction circuit 267. Compare by subtraction and output a pulse when they are equal. Basically, the set/reset timing of the flip-flop 268 of the second voltage-PWM conversion circuit 236 is delayed by 2/3 period with respect to the set/reset timing of the flip-flop 268 of the voltage-PWM conversion circuit 234, and the FET 239 And the operation timing of FET 242 is delayed by 2/3 period with respect to the operation timing of FET 237 and FET 240. The timing of resetting the flip-flop 268 of the second voltage-PWM conversion circuit 236 varies in its generation position depending on the pulse output from the subtraction circuit 267, but this is not shown. Reference numeral 282 indicates a change in the current i3 of the coil 257, and the current increases and decreases repeatedly due to the operation of the driving side switching element 248 and the freewheeling side switching element 251.

The configuration of FIG. 7 has three phases, and the currents i1, i2, and i3 flowing through the coils 255, 256, and 257 are basically as follows: The waveforms are the same, with the first half increasing or rising by 2/3 and the second half decreasing or falling by 1/3, but the phases are shifted by 2π/3 (coil currents 276, 279, and 282 in Figure 9), so the number of coils Select the number of phases 3, NON2, and NOFF1 from Table 1 of the power supply voltage, increase time, and decrease time according to (the drop is negligible). When the circuits shown in FIGS. 7 and 8 operate at the timing shown in FIG. , the capacitor 258 receives the currents i1, i2, and i3 flowing from the coils 255, 256, and 257, and maintains a constant voltage, thereby allowing a ripple-free current to flow into the semiconductor laser diode 259.

On the other hand, a function is required to control this current so that it reaches a target value. In order to realize this function, a laser current register 263 is prepared through a current detection resistor 260, an A/D converter 261, and a phase compensation 262, and a value proportional to the current flowing through the semiconductor laser diode 259 is set in the laser current register 263. The value of the laser current register 263 is compared with the target laser current preset in the output value register 264 in a subtraction circuit 265 to generate a difference voltage value.

When the actual current value flowing through the semiconductor laser diode 259 is larger than the target current value, a large value is held in the laser current register 263, and the value is compared with the value in the output value register 264 in the subtraction circuit 265, and a large difference voltage value is output. do. In each of the second voltage-PWM conversion circuits 234, 235, and 236, the output given from the counter 266 and the difference voltage value of the subtraction circuit 265 are compared by subtraction in the subtraction circuit 267, and when they become equal, a pulse is output. In the second voltage-PWM conversion circuit 234, specifically, it becomes a reset pulse to the flip-flop 268 at the slow position a shown at 273 in FIG. 9, and becomes an input pulse to the long width FET 237 shown at 274 c. . 275 is the inverse logic of 274 and becomes an input pulse to the short width FET 240 of e. In other words, if the timing of the output from the subtraction circuit 267 in the second voltage-PWM conversion circuits 234, 235, 236 is delayed, the low primary voltage that turns on the FETs 226, 227, 228 via the FETs 237, 238, 239 will drive switching for a long time. The high primary voltage that is applied to the coils 255, 256, 257 through the elements 246, 247, 248, and then turns on the FETs 229, 230, 231 through the FETs 240, 241, 242 is applied to the coils through the driving switching elements 246, 247, 248. 255, 256, and 257, resulting in a current i1 of the coil 255 having a small value of g shown at 276, for example.

When the actual current value flowing through the semiconductor laser diode 259 is smaller than the target current value, a small value is held in the laser current register 263, which is compared with the value in the output value register 264 in the subtraction circuit 265, and a small difference voltage value is output. do. In each of the second voltage-PWM conversion circuits 234, 235, and 236, the output given from the counter 266 and the difference voltage value of the subtraction circuit 265 are compared by subtraction in the subtraction circuit 267, and when they become equal, a pulse is output. In the second voltage-PWM conversion circuit 234, specifically, it becomes a reset pulse to the flip-flop 268 at an early position b shown at 273, and becomes an input pulse to the short width FET 237 shown at d shown at 274. 275 is the inverse logic of 274 and becomes an input pulse to the FET 240 with a long width of f. In other words, if the second voltage-PWM conversion circuit 234, 235, 236 outputs from the subtraction circuit 267 at an early timing, the low primary voltage that turns on the FETs 226, 227, 228 via the FETs 237, 238, 239 will shorten the drive switching. A high primary voltage is applied to the coils 255, 256, 257 through the elements 246, 247, 248, and then the FETs 229, 230, 231 are turned on through the FETs 240, 241, 242. 255, 256, and 257, resulting in a current i1 of the coil 255 having a large value of h shown at 276, for example.

That is, in FIG. 7, when the driving side switching elements 246, 247, and 248 are ON, the power supply voltage of the primary power source is VS, the laser power output is VO, and the inductance value of the coils 255, 256, and 257 is L. , 257 respectively have an increase amount iLON expressed by the following equation after T seconds.

iLON=(VS-VO)×T/L

According to this formula, the coil current increases in proportion to the difference between the power supply voltage VS and the laser output voltage VO, so the coil current can be controlled by means of increasing or decreasing VS as appropriate.

When trying to strongly increase the coil current, increase the coil current by increasing the time to apply the high primary voltage through FETs 229, 230, and 231 compared to the time to apply the low primary voltage through FETs 226, 227, and 228. increase. When trying to increase the coil current weakly, the coil current can be increased by shortening the time for applying a high primary voltage through FETs 229, 230, and 231 compared to the time for applying a low primary voltage through FETs 226, 227, and 228. Reduce the increase in

That is, if the output current is small, the power supply voltage is increased, and if the output current is large, the power supply voltage is controlled to be low. By this means, the sum of the currents flowing through the plurality of coils can be controlled to the target value. In addition, the influence of changes in the voltage applied to the semiconductor laser diode 259 and changes in the current waveform due to control of the voltage of the primary power supply is small, and the ripple can still be maintained at a value of about 1%, so there is no problem in driving the semiconductor laser diode. does not occur.

As a second application example of the switching power supply of the third embodiment in which the rate of increase of the current flowing through the coil is changed by means of changing the power supply voltage, a switching power supply characterized by quick start-up is shown in FIGS. This will be explained below using FIG.

When attempting to raise the laser output current of the switching power supply more rapidly, a large value is set in the output value register 264 in FIG. 7 before starting. Next, the operation of the second voltage-PWM conversion circuit 234 will be explained. The timing generator 233 is operated to drive the drive side switching element 246 as shown at 270 in FIG. 9, and further to drive the freewheeling side switching element 249 as shown at 271. Furthermore, the timing generator 233 is operated to set the counter 266 and flip-flop 268 of the second voltage-PWM conversion circuit 234 in FIG. It will be done. Subsequently, since a large value is set in the output value register 264, a reset pulse for the flip-flop 268 is generated at an early timing of a, as shown at 284 in FIG. Therefore, the flip-flop 268 immediately after startup inputs a short pulse to the FET 237 as shown in 285b, so that the FET 226 remains OFF for a long time. At the same time, as shown at 286, a long pulse c is input to the FET 240, so that the FET 229 is turned on for a long time, and a high primary power supply voltage is applied to the drive switching element 246. Therefore, the current flowing through the coil 255 increases rapidly as shown in d of 287, and the output current also increases.

Further, as shown at 277 in FIG. 9, the drive side switching element 247 is driven, and furthermore, as shown at 278, the freewheeling side switching element 250 is driven. The second voltage-PWM conversion circuit 235 in FIG. 7 operates in the same manner as the second voltage-PWM conversion circuit 234, and inputs the input to the FET 238 according to the short pulse 288 in FIG. . At the same time, the long pulse 289 is input to the FET 241, so that the FET 230 is turned on for a long time, and a high primary power supply voltage is applied to the drive switching element 247. Therefore, the current i2 flowing through the coil 256 rapidly increases as shown at 290, and the output current also increases. These pulse signals 288 and 289 operate at timings delayed by 1/3 period from the pulse signals shown at timings 285 and 286.

Furthermore, as shown at 280 in FIG. 9, the drive side switching element 248 is driven, and furthermore, as shown at 281, the freewheeling side switching element 251 is driven. The second voltage-PWM conversion circuit 236 in FIG. 7 operates in the same manner as the second voltage-PWM conversion circuit 234, and inputs the input to the FET 239 according to the short pulse 291 in FIG. . At the same time, a long pulse 292 is input to the FET 242, so that the FET 231 is turned on for a long time, and a high primary power supply voltage is applied to the drive switching element 248. Therefore, the current i3 flowing through the coil 257 rapidly increases as shown at 293, and the output current also increases. These pulse signals 291 and 292 operate at timings that are 2/3 cycles later than the pulse signals shown at timings 285 and 286.

In this way, the currents i1, i2, i3 of the coils 255, 256, 257 rise rapidly. As the currents in the coils 255, 256, and 257 increase, the value of the output value register 264 is sequentially lowered from the largest value, and the FET 237 input is sequentially pulsed with a wider width as shown in 285 e, f, and g in FIG. As a result, the increase in coil current is reduced, and a target current value to be passed through the laser diode 259 is set at the end of startup. After the start-up is completed, the increasing speed of the currents i1, i2, i3 flowing through the coils 255, 256, 257 is changed by means of continuously changing the power supply voltage, and the sum of the currents flowing through the plurality of coils 255, 256, 257 is calculated. is controlled to the target value.

A switching power supply or laser device according to a fourth embodiment of the present invention will be described below with reference to FIGS. 11 to 14.

The high-speed switching power supply of the fourth embodiment includes a primary power supply, a plurality of switching elements connected to the primary power supply, a plurality of coils connected to the plurality of switching elements, and one capacitor connected to the plurality of coils, and the power supply voltage, i.e. A means for setting or selecting the primary power supply voltage, and a means for changing the value of a resistance inserted or connected to a switching element (including cases where the mode of change is only to enable or short-circuit the resistance). It has a structure that keeps the total sum of current flowing through the coils constant, that is, reduces the amount of ripple, and controls it to a target value.

First, the configuration of the switching power supply according to the fourth embodiment will be described using FIG. 11.

In FIG. 11, 310 is a primary power source. The primary power supply 310 includes a power transformer 311, which has a plurality of taps drawn out on the secondary side, and has a structure that allows the output voltage of the secondary side to be selected. Reference numeral 312 denotes a switch which, as will be described later, enables setting or selection of the power supply voltage applied to the switching power supply or the main body of the switching power supply by selecting a plurality of taps prepared on the secondary side of the power transformer 311. be. 313, 314, 315, and 316 are rectifier diodes, which perform bridge type rectification. A capacitor 317 converts the ripple current obtained by the rectifying diodes 313, 314, 315, and 316 into direct current to obtain a primary power supply voltage to be applied to the switching power supply or the main body of the switching power supply.

318 is an FPGA (field programmable gate array) and has a function of controlling the entire system. A timing generator 319 has a function of generating timing pulses with different phases at a period of about 1 μs to 3 μs. Level conversion circuits 320, 321, and 322 convert timing pulses given from timing generator 319 into pulses for driving switching elements. Reference numerals 323, 324, and 325 are switching elements on the driving side, which are P-channel FETs, and whose source terminals are connected to the primary power supply voltage. Reference numerals 326, 327, and 328 are switching elements on the free circulation side, and N-channel FETs are used, the drain terminals of which are connected to the drain terminals of the switching elements 323, 324, and 325 on the driving side. Diodes 329, 330, and 331 are on the freewheeling side, and are connected in parallel to switching elements 326, 327, and 328 on the freewheeling side, which realize high-speed current switching using Schottky diodes. Attenuation resistors 332, 333, and 334 are inserted in series with the switching elements 326, 327, and 328 on the freewheeling side and the diodes 329, 330, and 331 on the freewheeling side. 335, 336, and 337 are N-channel FETs, which are connected in parallel to attenuation resistors 332, 333, and 334. The switching element 323 on the driving side and the switching element 326 on the freewheeling side constitute one, one set, or one phase switching element, and the switching element 324 on the driving side and the switching element 327 on the freewheeling side constitute one or one set. Alternatively, a one-phase switching element is configured, and the driving side switching element 325 and the freewheeling side switching element 328 constitute one, one set, or one phase switching element.

338, 339, and 340 are third voltage-PWM (Pulse Width Modulation) conversion circuits, which have a function of converting an output difference voltage value into a pulse time width. Normally, a voltage is applied from the third voltage-PWM conversion circuit 338, 339, 340 to the gate of FET 335, the gate of FET 336, and the gate of FET 337, and FET 335, FET 336, and FET 337 are in the ON state and resist the attenuation resistance. 332, 333, and 334 are shorted.

341, 342, and 343 are coils having multiple windings around the core, each having the same appropriate inductance value, and connected to the drain terminals of the switching elements 323, 324, and 325 on the drive side or the switching elements on the free circulation side. It has the role of changing the current flowing from the drain terminals of 326, 327, and 328 with a gentle slope. A capacitor 344 has the role of receiving current flowing from the coils 341, 342, and 343 to maintain a constant voltage. 345 is a semiconductor laser diode serving as a load, which is connected in series to the coils 341, 342, and 343 and in parallel to the capacitor 344, and as the current increases, as the current is applied, 1. It has the property of gradually rising from a voltage across both ends of about 8V to about 2.1V. Reference numeral 346 is a current detection resistor for measuring the current flowing through the semiconductor laser diode 345, and a resistor capable of passing a large current of approximately several tens of mΩ is used. In order to obtain larger optical output, multiple semiconductor laser diodes may be connected in series, and in that case, a larger applied voltage is required depending on the number of connections.

347 is an A/D converter that converts the voltage across the current detection resistor 346 into a numerical value, and converts the current flowing through the semiconductor laser diode 345 into a numerical value. A phase compensation circuit 348 is composed of a digital filter, and has the role of performing phase compensation so that the control loop does not oscillate even if the current value flowing through the semiconductor laser diode 345 is strongly controlled. A laser current register 349 holds the output of the phase compensation circuit 348. 350 is an output value register that holds the current value to be passed through the target semiconductor laser diode 345. 351 is a subtraction circuit that compares the value of the laser current register 349 and the value of the output value register 350 and outputs the difference voltage value between the target current value flowing through the semiconductor laser diode 345 and the actual current value flowing through the semiconductor laser diode 345. .

Next, the configuration of the third voltage-PWM conversion circuit for the switching power supply of the fourth embodiment will be explained using FIG. 12.

In FIG. 12, the FPGA 318, timing generator 319, phase compensation circuit 348, laser current register 349, output value register 350, subtraction circuit 351, FET 335, and attenuation resistor 332 are those shown in FIG. FIG. 12 shows the configuration of the voltage-PWM conversion circuit 338, and although the voltage-PWM conversion circuits 339 and 340 are not shown, the voltage-PWM conversion circuits 339 and 340 are similar to the voltage-PWM conversion circuit 338. It has the same or the same configuration as .

The third voltage-PWM conversion circuits 338, 339, and 340 have a function of converting the output difference voltage value into a pulse time width. As a component of the third voltage-PWM conversion circuits 338, 339, and 340, 352 is a counter that receives the reference pulse from the timing generator 319 and outputs a numerical value that increases with time to generate timing for controlling the attenuation resistance. . A subtraction circuit 353 compares the output from the counter 352 and the difference voltage value output from the subtraction circuit 351 by subtraction, and outputs a pulse when they become equal. 354 is a flip-flop (F.F.) that provides timing for attenuation resistance control, which receives the reference pulse from the timing generator 319 and the pulse from the subtraction circuit 353, and generates a short width pulse if the difference voltage value is small; If the difference voltage value is large, it has the function of generating a pulse with a long width. This output is applied to the gates of FETs 335, 336, and 337 to change the resistance values of attenuation resistors 332, 333, and 334 to enable or short circuit.

As a first application example of the switching power supply of the fourth embodiment, a plurality of coils are used as a first application example of the switching power supply of the fourth embodiment, in which the speed of decrease of the current flowing through the coil is changed by means of setting or selecting the power supply voltage and changing the resistance value inserted in the freewheeling circuit. The operation of the switching power supply that controls the sum of currents flowing through the circuit to a target value will be explained using FIG. 13.

In FIG. 13, 355 is a clock having a period of about 10 ns to 100 ns, which is output from the timing generator 319, and serves as the basis for the operation of the switching power supply. Reference numeral 356 indicates a signal output by the timing generator 319 at a constant timing, and when this signal is ON, the driving side switching element 323 is driven through the level conversion circuit 320. Reference numeral 357 indicates a signal output by the timing generator 319 at a constant timing, and when this signal is ON, the switching element 326 on the freewheeling side is driven through the level conversion circuit 320. Here, the operation of the third voltage-PWM conversion circuit 338 in FIG. 12 will be explained. In FIG. 13, reference numeral 358 indicates the timing of resetting the counter 352 and setting the flip-flop 354 in FIG. 12, and in this embodiment, the timing generator 319 regularly applies pulses every 3 μs. Reference numeral 359 indicates the reset timing of the flip-flop 354, and the generation position varies depending on the pulse output from the subtraction circuit 353. Reference numeral 360 indicates an input to the FET 335 based on the output of the flip-flop 354, which is repeatedly turned on and off depending on the set timing and reset timing of the flip-flop 354, and when this output is on, the attenuation resistor 332 is short-circuited via the FET 335.

361 indicates a change in the current i1 of the coil 341, and the current increases and decreases repeatedly due to the operation of the switching element 323 on the driving side and the switching element 326 on the freewheeling side. When the switching element 323 on the driving side is ON, the switching element 326 on the freewheeling side is OFF, and when the switching element 323 on the driving side is OFF, the switching element 326 on the freewheeling side is ON; When the switching element 323 and the switching element 326 on the freewheeling side are turned on at the same time, a breakthrough current is generated in the switching element 323 on the driving side and the switching element 326 on the freewheeling side. Therefore, the timing 356 rises later than the fall of the timing 357, and the timing 356 It is desired that the timing 357 falls earlier than the rising edge.

Reference numeral 362 indicates a signal outputted by the timing generator 319 at a constant timing, and when this signal is ON, the driving side switching element 324 is driven through the level conversion circuit 321. Reference numeral 363 indicates a signal outputted by the timing generator 319 at a constant timing, and when this signal is ON, the switching element 327 on the freewheeling side is driven through the level conversion circuit 321. Timing 362 and timing 363 are operated 1/3 cycle (phase is 2π/3) later than timing 356 and timing 357.

The third voltage-PWM conversion circuit 339 in FIG. 11 receives the timing generation 319 and the difference voltage value that is the output of the subtraction circuit 351, and calculates the difference voltage value between the output from the counter 352 and the subtraction circuit 351 in the subtraction circuit 353. Compare by subtraction and output a pulse when they are equal. The timing of setting and resetting the flip-flop 354 of the third voltage-PWM conversion circuit 339 is basically 1/3 cycle later than the timing of setting and resetting the flip-flop 354 of the voltage-PWM conversion circuit 338. The operation timing of the FET 335 is delayed by 1/3 cycle compared to the operation timing of the FET 335. The reset timing of the flip-flop 354 of the third voltage-PWM conversion circuit 339 varies in generation position depending on the pulse output from the subtraction circuit 353, but this is not shown. 364 indicates a change in the current i2 of the coil 342, and the current increases and decreases repeatedly due to the operation of the driving side switching element 324 and the freewheeling side switching element 327.

Reference numeral 365 indicates a signal outputted by the timing generator 319 at a constant timing, and when this signal is ON, the driving side switching element 325 is driven through the level conversion circuit 322. Reference numeral 366 indicates a signal outputted by the timing generator 319 at a constant timing, and when this signal is ON, the switching element 328 on the freewheeling side is driven through the level conversion circuit 322. Timing 365 and timing 366 are operated 2/3 cycle (phase is 4π/3) later than timing 356 and timing 357.

The third voltage-PWM conversion circuit 340 in FIG. 11 receives the timing generation 319 and the difference voltage value that is the output of the subtraction circuit 351, and calculates the difference voltage value between the output from the counter 352 and the subtraction circuit 351 in the subtraction circuit 353. Compare by subtraction and output a pulse when they are equal. The timing of setting and resetting the flip-flop 354 of the third voltage-PWM conversion circuit 340 is basically 2/3 cycle later than the timing of setting and resetting the flip-flop 354 of the voltage-PWM conversion circuit 338. The operation timing of the FET 335 is delayed by 2/3 cycles compared to the operation timing of the FET 335. The reset timing of the flip-flop 354 of the third voltage-PWM conversion circuit 340 varies in generation position depending on the pulse output from the subtraction circuit 353, but this is not shown. 367 indicates a change in the current i3 of the coil 343, and the current increases and decreases repeatedly due to the operation of the driving side switching element 325 and the freewheeling side switching element 328.

The configuration of FIG. 11 has three phases, and the currents i1, i2, and i3 flowing through the coils 341, 342, and 343 are basically as follows: The waveforms are the same, with the first half increasing or rising by 2/3 and the second half decreasing or falling by 1/3, but the phases are shifted by 2π/3 (coil currents 361, 364, and 367 in Figure 13), so the number of coils Select the number of phases 3, NON2, and NOFF1 from Table 1 of the power supply voltage, increase time, and decrease time according to (the drop is negligible). When the circuits shown in FIGS. 11 and 12 operate at the timing shown in FIG. 13, currents from the drive side switching elements 323, 324, 325 and the freewheeling side switching elements 326, 327, 328 flow to the coils 341, 342, 343. , the capacitor 344 receives the currents i1, i2, and i3 flowing from the coils 341, 342, and 343, maintains a constant voltage, and can flow ripple-free current into the semiconductor laser diode 345.

On the other hand, a function is required to control this current so that it reaches a target value. In order to realize this function, a laser current register 349 is prepared through a current detection resistor 346, an A/D converter 347, and a phase compensation 348, and a value proportional to the current flowing through the laser diode 345 is set in the laser current register 349. The value of the laser current register 349 is compared with the target laser current preset in the output value register 350 in a subtraction circuit 351 to generate a difference voltage value.

When the actual current value flowing through the semiconductor laser diode 345 is larger than the target current value, a large value is held in the laser current register 349, and is compared with the value in the output value register 350 in the subtraction circuit 351 to output a small difference voltage value. do. In each of the third voltage-PWM conversion circuits 338, 339, and 340, the output given from the counter 352 and the difference voltage value of the subtraction circuit 351 are compared by subtraction in the subtraction circuit 353, and when they become equal, a pulse is output. In the third voltage-PWM conversion circuit 338, specifically, it becomes a reset pulse to the flip-flop 354 in the early position a shown at 359 in FIG. 13, and becomes an input pulse to the short width FET 335 shown at 360 c. The short circuit of the damping resistor 332 is quickly released. That is, if the timing of the output from the subtraction circuit 353 in the third voltage-PWM conversion circuits 338, 339, 340 is early, the attenuation resistors 332, 333, 334 will be effective for a long time via FETs 335, 336, 337, and, for example, in 361. This results in a coil current that decays quickly as shown in e.

When the actual current value flowing through the semiconductor laser diode 345 is smaller than the target current value, the small value is held in the laser current register 349, and is compared with the value in the output value register 350 in the subtraction circuit 351, and a large difference voltage value is output. do. In each of the third voltage-PWM conversion circuits 338, 339, and 340, the output given from the counter 352 and the difference voltage value of the subtraction circuit 351 are compared by subtraction in the subtraction circuit 353, and when they become equal, a pulse is output. In the third voltage-PWM conversion circuit 338, specifically, it becomes a reset pulse to the flip-flop 354 at the slow position b shown at 359, and becomes an input pulse to the long width FET 335 shown at d shown at 360, and the attenuation resistor Continue shorting 332. In other words, if the timing of the output from the subtraction circuit 353 in the third voltage-PWM conversion circuits 338, 339, 340 is delayed, the attenuation resistors 332, 333, 334 will be disabled for a long time via the FETs 335, 336, 337, as shown in 361, for example. As shown in f, the coil current does not attenuate significantly.

In FIG. 11, when the freewheeling side switching elements 326, 327, and 328 are ON, the laser output voltage is VO, the resistance value of the attenuation resistors 332, 333, and 334 is Rd (0 in case of short circuit), and the resistance value of the coils 341, 342, and 343 is set to VO. Each of the currents iL (i1, i2, i3) flowing through the coils 341, 342, and 343 with an inductance of L becomes a reduction amount iLOFF expressed by the following equation after T seconds.

iLOFF=-(VO+Rd×iL)×T/L

As a result, if the value of Rd is large, the coil current is greatly attenuated, so the coil current can be controlled by means of appropriately increasing or decreasing the value of Rd. When maintaining the coil current, the decrease in the coil current can be reduced by lengthening the time during which the values of the damping resistors 332, 333, and 334 are 0 or low. When attempting to reduce the coil current, the reduction in the coil current can be increased by lengthening the time during which the values of the damping resistors 332, 333, and 334 are enabled or increased.

That is, if the output current is small, the value of the attenuation resistor is short-circuited or lowered, and if the output current is large, the value of the attenuation resistor is enabled or increased.The total sum of the currents flowing through the plurality of coils can be controlled to the target value.

The influence of fluctuations in the voltage applied to the semiconductor laser diode 345 and changes in the current waveform controlled by the attenuation resistor is small, and the ripple can be maintained at a value of about 1%, so there is no problem in driving the semiconductor laser diode.

A second application example of the switching power supply of the fourth embodiment, in which the rate of decrease of the current flowing through the coil is changed by means of changing the resistance value inserted in the switching element, is a switching device characterized in that the fall is made faster. The power supply will be explained below using FIGS. 11 to 14.

When cutting off the laser output current of the switching power supply, the laser output current does not decrease rapidly due to residual energy stored in the coil and capacitor. In order to more rapidly reduce the laser output current, it is effective to consume the energy stored in the coil by inserting a damping resistor into the circulation circuit.

A method of rapidly cutting off the laser output current from the supply state will be described using FIG. 14.

When the laser output current is supplied, as shown in the first half of 368 and 369, 374 and 375, 378 and 379 in FIG. The laser diode 345 is driven by operating the switching element 324, the switching element 327 on the return side, the switching element 325 on the drive side, and the switching element 328 on the return side. Further, the third voltage-PWM converters 338, 339, and 340 are operated to control the sum of currents flowing through the plurality of coils 341, 342, and 343 to a target value at the timing shown in FIG. At this time, the attenuation resistors 332, 333, and 334 remain short-circuited for a long time, and the attenuation of the coil current is small.

Next, when cutting off the laser output current of the switching power supply, the operation of the driving side switching element 323 is stopped from the timing generation 319 as shown in the second half of 368 in FIG. The side switching element 326 is operated. Furthermore, the third voltage-PWM conversion 338 is operated, the counter 352 and the flip-flop 354 are set at the timing shown at 370 in FIG. 14, and the flip-flop 354 is reset at the early timing of the position a at the timing shown at 371. . In this way, the input to the FET 335 is turned off for a long time as shown in 372b, thereby making the attenuation resistor 332 effective. That is, the attenuation resistor 332 has a high resistance value for a long time, and as shown in 373c, the attenuation of the current i1 flowing through the coil 341 is large.

Similarly, as shown in the second half of 374 in FIG. 14, the driving side switching element 324 stops operating from timing generation 319, and as shown in the second half of 375, the freewheeling side switching element 327 is activated. Furthermore, the third voltage-PWM conversion 339 is operated in the same manner as the third voltage-PWM conversion 338, and the input of the FET 336 is turned OFF for a long time as shown in 376d to enable the attenuation resistor 333. That is, the attenuation resistor 333 has a high resistance value for a long time, and as shown in 377e, the attenuation of the current i2 flowing through the coil 342 is large.

Similarly, as shown in the second half of 378 in FIG. 14, the operation of the switching element 325 on the drive side is stopped from timing generation 319, and the switching element 328 on the free circulation side is started as shown in the second half of 379. Further, the third voltage-PWM conversion 340 is operated in the same manner as the third voltage-PWM conversion 338, and the input of the FET 337 is turned off for a long time as shown in 380f, thereby making the attenuation resistor 334 effective. That is, the attenuation resistor 334 has a high resistance value for a long time, and as shown in g of 381, the attenuation of the current i3 flowing through the coil 343 is large.

That is, when cutting off the laser current, the currents in the coils 341, 342, 343 can be reduced by activating the attenuation resistors 332, 333, 334, and the laser output current can be rapidly reduced.

Here, only one set of attenuation resistors and FETs for controlling the attenuation resistors are provided for each coil or switching element. For example, by preparing multiple attenuation resistors 332 and FETs 335 to control the attenuation resistors, a wide range of control is possible by varying the value of the attenuation resistors, or by preparing means for continuously changing the attenuation resistors. Needless to say, it will become.

A high-speed switching power supply characterized in that the resonance frequency determined by the coil and the capacitor is 0.1 times or more the switching frequency will be described below with reference to FIGS. 1 to 3 of the first embodiment of the present invention.

As shown by the coil current 45 in FIG. 2, the current flowing through the coil increases and decreases in a triangular waveform having a switching frequency determined by 43. This increase or decrease in current causes large ripples in the output voltage and output current, and the output voltage and output current do not become constant. In order to avoid this, in the conventional switching power supply, the values of the inductance L of the coil 36 and the capacitance C of the capacitor 40 in FIG. In other words, it is a value that satisfies the following equation. Specifically, the LC resonance frequency is set to 1 kHz or less with respect to the switching frequency of 300 kHz.

LC resonance frequency = 1/(2π√(LC)) <<0.1× switching frequency

However, in the first embodiment of the present invention, the value is set to satisfy the following equation. Specifically, it is preferable to set the LC resonance frequency to about 300 kHz, which is equivalent to the switching frequency of 300 kHz.

LC resonance frequency = 1/(2π√(LC))≧0.1×switching frequency

In this way, the high-speed switching power supply according to the present invention makes the inductance of the coil and the capacitance of the capacitor extremely small, but by setting or selecting the power supply voltage as shown in Fig. 3, the total amount of current flowing through multiple coils can be increased or decreased. Ripple can be eliminated by making the total amount equal. Furthermore, by extremely reducing the inductance of the coil and the capacitance of the capacitor, it is possible to rapidly increase the output voltage and output current, which is considered to be a major feature not found in conventional switching power supplies.

A high-speed switching power supply characterized by increasing or decreasing the number of phases at startup or in accordance with the output current will be described below.

As a first application example of increasing/decreasing the number of phases, a switching power supply characterized in that a plurality of switching elements are simultaneously turned on at the time of startup to speed up the startup will be described below with reference to FIGS. 3 and 15. In FIG. 3, the driving side switching elements 24, 25, 26, 27 and the freewheeling side switching elements 28, 29, 30, 31 in FIG. 1 are operated in sequence to generate a laser output current. On the other hand, in the part a in FIG. 15, the drive-side switching elements 24, 25, 26, and 27 in FIG. That is, in this portion, a current i four times as large flows into the capacitor 40, and a rapidly rising laser output current is obtained. Therefore, the rise time in FIG. 15 is reduced to 1/4 compared to FIG. 3. By means of returning to the control method of FIG. 3 when the elapsed time or the voltage across the capacitor 40 exceeds a predetermined value during the rise, it is possible to obtain a rapid rise time of the laser output current without overshooting.

As a second application example of increasing/decreasing the number of phases, a switching power supply characterized by increasing/decreasing the number of phases according to the output current will be described below with reference to FIG. 16. FIG. 16(a) shows the sum of solid line coil currents i1, i2, i3, and i4 as laser output current i in the switching power supply using the four-phase coil shown in FIG. On the other hand, when the value of the output laser current i is decreased as shown by the broken line, the entire triangular wave shifts to a lower value as shown by the broken line for the coil currents i1, i2, i3, and i4, and the lower part of the triangular wave becomes 0. When the value of the laser current i to be output is further reduced, a time occurs when the coil currents i1, i2, i3, and i4 are 0. A switching power supply has a property of causing a discontinuous change in the output voltage when the output current decreases and the current flowing through the coils 36, 37, 38, and 39 continues to have a value of 0, and the ripple increases. As a method to avoid such a situation, the timing generation 19 in the first embodiment is controlled, and as shown in FIG. If the laser current is supplied using, for example, the current is halved, so it is possible to maintain the same triangular wave value as the coil current. When the output current is reduced in this way, it is effective to reduce the number of phases.

A high-speed switching power supply characterized by changing the clock period according to the output current will be described below with reference to FIG. 16. FIG. 16(a) shows the sum of solid line coil currents i1, i2, i3, and i4 as laser output current i in the switching power supply using the four-phase coil shown in FIG. On the other hand, when the value of the output laser current i is decreased as shown by the broken line, the entire triangular wave shifts to a lower value as shown by the broken line for the coil currents i1, i2, i3, and i4, and the lower part of the triangular wave becomes 0. When the value of the laser current i to be output is further reduced, a time occurs when the coil currents i1, i2, i3, and i4 are 0. A switching power supply has a property of causing a discontinuous change in the output voltage when the output current decreases and the current flowing through the coil continues to have a value of 0, resulting in an increase in ripple. As a method to avoid such a situation, the timing generation 19 in the first embodiment is controlled to reduce the clock period and reduce the current i1 flowing through the coils 36, 37, 38, 39, as shown in FIG. 16(c). Decrease the amplitudes of i2, i3, and i4. In FIG. 16(c), if the clock period is halved and the laser current i is supplied, for example, the current is halved, so it is possible to maintain a triangular wave that does not become zero. When the output current is reduced in this way, means for reducing the clock cycle becomes effective.

Regarding a high-speed switching power supply characterized by having a means for equalizing the current values flowing through a plurality of coils, the operation of equalizing the current values will be described below using FIG. 17.

In FIG. 17, the configuration and operation from 110 to 145 are the same as those in the second embodiment described using FIG. 4, and the description thereof will be omitted. Reference numerals 146, 147, and 148 are minute resistances, which are inserted in series with the switching elements 129, 130, and 131 on the freewheeling side and the diodes 132, 133, and 134 on the freewheeling side. Numeral 149 is a multiplexer which selects the voltages generated at minute resistors 146, 147, and 148 and sends them to A/D converter 141.

Due to variations in component constants, the currents of each phase in a multiphase switching power supply tend to vary. If there is a coil with a large current and a coil with a small current, there will be a case where the current becomes 0 in the coil with a small current, resulting in a ripple in the laser output current. Therefore, the return current generated in the micro resistances 146, 147, and 148 is measured by switching the multiplexer 149. Secondly, the current flowing through each coil can be made uniform by means of supplying a longer current to the driving switching element of the coil whose current is smaller than the average value of the current flowing through each coil, that is, by turning it ON. Can be done.

Further, the current flowing through each coil can be made uniform by means for changing the power supply voltage and means for changing the resistance value inserted in the freewheeling circuit. The minute resistors 146, 147, 148 can be the same as the attenuation resistors 332, 333, 334 in FIG. 337 is turned off and the return current generated in the damping resistors 332, 333, and 334 is measured through the multiplexer, it is possible to similarly make the current flowing through each of the coils 135, 136, and 137 a uniform value.

The operation of phase compensation will be described with reference to FIG. 18 regarding a switching power supply characterized in that phase compensation is realized by digital control.

The switching element 510 in FIG. 18 is the drive-side switching element 126, 246, 323 in FIGS. 4, 7, and 11, and the diode 511 is the reflux-side diode 132 or the reflux-side switching element 129 in FIGS. 4, 7, and 11. , the diode 252 on the freewheeling side or the switching element 249 on the freewheeling side, the diode 329 on the freewheeling side or the switching element 326 on the freewheeling side, and the coil 512 are the coil 135, the coil 255, the coil 341, and the capacitor 513 in FIGS. The capacitor 138, the capacitor 258, the capacitor 344, the semiconductor laser diode 514 in FIGS. 4, 7, and 11 are the semiconductor laser diode 139, the semiconductor laser diode 259, the semiconductor laser diode 345, and the current detection The resistor 515 is the current detection resistor 140, the current detection resistor 260, and the current detection resistor 346 in FIGS. 4, 7, and 11. The switching element 510, diode 511, and coil 512 can be added in parallel according to the number of phases configuring the circuit, as in FIGS. 4, 7, and 11. Reference numeral 516 is a delay element, which is the delay of the A/D converter 141, A/D converter 261, and A/D converter 347 that receive the output current in FIGS. 4, 7, and 11, and the delay of the FPGA 118, FPGA 232, and FPGA 318. Includes calculation delays. Resistor 517 and resistor 518 are resistors for phase compensation, capacitor 519 is combined with resistor 517 and resistor 518 to determine the characteristic of stepped phase compensation, 520 is an amplifier with a gain of 1, and resistor 521 is for phase compensation. The capacitor 522 is used in combination with the resistor 521 to determine the characteristics of phase compensation that decreases in the high range. , configures the phase compensation 142, phase compensation 262, and phase compensation 348 in FIG. 11, and provides the output of the phase compensation. 523 is an amplifier whose gain has a value of GAIN, which corresponds to the subtraction circuit 145, the subtraction circuit 265, and the subtraction circuit 351 in FIGS. 4, 7, and 11, and which calculates the difference between the phase compensation output and the target current value Vm get. A comparator 524 corresponds to the subtraction circuits 147, 267, and 353 in FIGS. 5, 8, and 12, and receives the output of the amplifier 523 at its negative terminal, and applies a periodic triangular wave to its positive terminal. Outputs a pulse with a time width proportional to the output of The delay element 516, the amplifier 520, the amplifier 523, and the comparator 524 form a negative feedback loop for keeping the output voltage VR of the current detection resistor 515 constant. The resistor 517, the resistor 518, the capacitor 519, the resistor 521, and the capacitor 522 determine the frequency characteristics of the amplifier 523 to constitute a lag-lead compensation method, and have the role of eliminating oscillation elements that are delayed from the VR signal through the negative feedback signal delay circuit. High-speed switching power supplies are subject to many fluctuations when starting up, being affected by the workpiece during processing of the semiconductor laser diode that is the load, and when shutting down. Digital phase compensation, which can dynamically change the phase compensation characteristics depending on the situation, is an effective means to deal with this variation.

By driving a semiconductor laser diode using the switching power supply according to the present invention, it is possible to provide a method for obtaining a sufficiently clean power output without ripples as well as a fast rise time. If high-speed pulse drive becomes possible, precise laser processing can be achieved, and the use of a switching power supply increases efficiency, making it possible to reduce the size and cost.

By applying the high-speed switching power supply according to the present invention to a power supply for driving a laser, a laser beam processing machine with good processability can be realized using a small and highly efficient power supply. It is easy to expand to other applications requiring even faster rise times.

10 Primary power supplies 24, 25, 26, 27 of the switching power supply of the first embodiment Drive-side switching elements 28, 29, 30, 31 of the switching power supply of the first embodiment Freewheel measurement of the switching power supply of the first embodiment Switching elements 36, 37, 38, 39 Coil 40 of the switching power supply of the first embodiment Capacitor 41 of the switching power supply of the first embodiment Semiconductor laser diode 110 of the laser device of the first embodiment Primary power supplies 126, 127, 128 of the switching power supply 129, 130, 131 of the driving side switching elements of the switching power supply of the second embodiment Freewheel measurement switching elements 135, 136, 137 of the switching power supply of the second embodiment Second implementation Coil 138 of the switching power supply of the second embodiment Capacitor 139 of the switching power supply of the second embodiment Semiconductor laser diode 142 of the laser device of the second embodiment Phase compensation circuit 146, 147, 148 of the switching power supply of the second embodiment 149 Multiplexers 210, 211 Primary power supplies 246, 247, 248 of the switching power supply of the third embodiment Drive-side switching elements 249, 250, 251 of the switching power supply of the third embodiment Return current measurement of the switching power supply of the third embodiment Switching elements 255, 256, 257 Coil 258 of the switching power supply of the third embodiment Capacitor 259 of the switching power supply of the third embodiment Semiconductor laser diode 262 of the laser device of the third embodiment Switching power supply of the third embodiment Phase compensation circuit 310 of the switching power supply of the fourth embodiment Primary power supplies 323, 324, 325 Drive-side switching elements 326, 327, 328 of the switching power supply of the fourth embodiment Freewheeling measurement of the switching power supply of the fourth embodiment Switching elements 332, 333, 334 Attenuation resistors 341, 342, 343 Coil 344 of the switching power supply of the fourth embodiment Capacitor 345 of the switching power supply of the fourth embodiment Semiconductor laser diode 348 of the laser device of the fourth embodiment Phase compensation circuit 510 of the switching power supply according to the fourth embodiment Drive-side switching element 512 Coil 513 Capacitor

Claims (11)

Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In a switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of
A high-speed switching power supply characterized in that the total sum of currents flowing through the plurality of coils is controlled to a target value by means for controlling the plurality of switching elements. Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In a switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of
A high-speed switching power supply, characterized in that the sum of currents flowing through the plurality of coils is controlled to a target value by means of changing the power supply voltage . Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In a switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of
A high-speed switching power supply characterized by faster start-up by changing the power supply voltage. Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In a switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of
A high-speed switching power supply characterized in that the sum of currents flowing through the plurality of coils is controlled to a target value by means for changing the resistance value inserted or connected to the plurality of switching elements . Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In a switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of
A high-speed switching power supply characterized in that the power supply is turned down quickly by means of changing the resistance value inserted or connected to the plurality of switching elements . Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In a switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of
A high-speed switching power supply characterized in that a resonance frequency determined by the coil and the capacitor is 0.1 times or more of a switching frequency . Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In a switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of
A high-speed switching power supply characterized by increasing or decreasing the number of coil phases at startup or according to the output current . Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In a switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of
A high-speed switching power supply characterized by changing the clock cycle depending on the output current. Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. A laser device characterized in that the laser is driven by a high-speed switching power supply that keeps the sum of currents flowing through the plurality of coils constant by means of: Consisting of a primary power source, multiple switching elements connected to the primary power source, multiple coils connected to the multiple switching elements, and one capacitor connected to the multiple coils, the power supply voltage is set or selected according to the number of coils. In the switching power supply, the sum of the currents flowing through the plurality of coils is made constant by means of , and the laser is driven by a high-speed switching power supply that sets or selects the power supply voltage as output voltage x (1 + (number of coils where current decreases)/(number of coils where current increases)) laser equipment. A laser device characterized in that a laser is driven by the high-speed switching power supply according to any one of claims 1 to 8.

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