JP7386192B2 - power converter - Google Patents

Description

The present invention relates to a power conversion device, and particularly to a power conversion device suitable for downsizing a voltage doubler rectifier circuit when aligning three-phase alternating current to direct current.

2. Description of the Related Art Conventionally, inverter devices that convert a commercial power source (three-phase 200 V, etc.) to an arbitrary frequency or voltage have been widely used for purposes such as variable speed control of motors. In these inverter devices, the rectifier circuit (converter) at the front stage converts the power into DC power, and the inverter at the rear stage inputs the DC power and outputs AC power with a desired voltage and frequency. . The rectifier circuit included in such an inverter device often performs full-wave rectification of three-phase alternating current, and the DC voltage is approximately √2 times as large as each line voltage input to the rectifier circuit. Generally, the upper limit of the output voltage of an inverter is determined by the DC voltage, and it is not possible to output a voltage higher than that. For this reason, for example, the inverter may not be able to output the voltage necessary to rotate the load motor at high speed. In a typical motor, the induced voltage increases as the rotational speed increases, so the inverter needs to output a higher voltage in order to flow the current required by the motor.

To address this problem, boosting of the DC voltage using a voltage doubler rectifier circuit has been proposed, for example, as disclosed in Patent Document 1 and Patent Document 2. Patent Document 1 proposes a method using a voltage doubler rectifier circuit to switch between a full-wave rectification state and a voltage doubler rectification state depending on the load condition of the motor while improving the power factor. In the technology described in Patent Document 1, in the full-wave rectification state, the voltage doubler rectifier circuit is stopped to make the DC voltage the full-wave rectifier voltage (1x), and when the motor speed exceeds a predetermined value, the voltage doubler is The state changes to a voltage rectification state, and the voltage doubler rectifier circuit is driven to double the DC voltage. On the other hand, in the voltage doubler rectification state, the switching frequency of the three bidirectional switches used in the voltage doubler rectifier circuit is driven at twice the commercial frequency to alternately charge two series capacitors connected to the DC section. This increases the pressure. Further, in Patent Document 2, in the boost mode, the switching frequency of a voltage doubler rectifier circuit configured with two switches is driven at three times the commercial frequency. Another method is to control the DC voltage to less than twice or more than twice by changing the charging current to two series capacitors connected to the DC part by manipulating the on period (duty ratio) of the switch. Proposed.

Japanese Patent Application Publication No. 2017-184397 International Publication No. 2015/186229

When applying a voltage doubler rectifier circuit to an inverter device including a rectifier circuit, it is necessary to modify the DC link capacitor for voltage smoothing into a two-series configuration. When it is desired to minimize modification by utilizing an existing device, it is desirable to add a voltage doubler rectifier circuit equipped with two series capacitors to the inverter device. Such an add-on type voltage doubler rectifier circuit is often additionally mounted on an existing device, so there is a restriction on the mounting area, and it is required to be configured in a small size. Further, an add-on type voltage doubler rectifier circuit has a circuit configuration in which a DC link capacitor included in an inverter device and two series capacitors connected to a DC output section of the voltage doubler rectifier circuit are connected in parallel. In this case, in the methods described in Patent Document 1 and Patent Document 2, when three-phase AC is input, the charging period of two series capacitors connected to the DC output part of the voltage doubler rectifier circuit is equal to the system period. The capacitance is 3 or 6 times larger, and each of the two series capacitors requires a capacitor with sufficient capacitance in order to boost the DC voltage and prevent inrush current in each capacitor. Therefore, the volume of the capacitor mounted in the voltage doubler rectifier circuit (add-on circuit) becomes dominant, making it difficult to miniaturize the add-on circuit.

An object of the present invention is to miniaturize a voltage doubler rectifier circuit in a power conversion device for aligning three-phase alternating current to direct current.

The configuration of the power conversion device of the present invention preferably includes a voltage doubler rectifier circuit that voltage doubles and rectifies the AC voltage of the three-phase AC power supply, and a control circuit that performs switching control of the voltage doubler rectifier circuit. The circuit consists of a first diode bridge connected to a three-phase AC power supply, a DC link capacitor connected to the DC section between the output side and the load side of the first diode bridge, and a three-phase diode bridge connected in parallel to the first diode bridge. A second diode bridge connected to the phase AC power supply, a first switching element and a second switching element connected in parallel to the second diode bridge, and a second diode bridge connected in parallel between the first diode bridge and the DC link capacitor. a first resonant capacitor and a second resonant capacitor; a resonant inductor connected between the first switching element and the second switching element; and between the first resonant capacitor and the second resonant capacitor; and a first diode. A first diode that connects between the cathode side output of the bridge and the cathode side output of the second diode bridge, and a connection between the anode side output of the first diode bridge and the anode side output of the second diode bridge. A second diode is provided.

According to the present invention, it is possible to downsize the voltage doubler rectifier circuit in a power conversion device when aligning three-phase alternating current to direct current.

1 is a circuit configuration diagram of a power conversion device according to Embodiment 1. FIG. FIG. 3 is a configuration diagram of a control block. It is a graph which shows an example of the relationship between resonance frequency Fr, switching frequency Fsw, and power output Pout by a power conversion device. FIG. 6 is a waveform diagram showing an example of each waveform in circuit operation when Fsw≧Fr of the resonant voltage doubler rectifier circuit. FIG. 7 is a diagram showing circuit operation in modes a1 to a6 of the circuit operation when Fsw≧Fr of the resonant voltage doubler rectifier circuit. FIG. 3 is a waveform diagram showing an example of each waveform in circuit operation when Fsw<Fr of the resonant voltage doubler rectifier circuit. FIG. 7 is a diagram showing circuit operation in modes b1 to b6 of circuit operation when Fsw<Fr of the resonant voltage doubler rectifier circuit. It is a graph showing an example of the relationship between power output Pout and power loss Ploss of an add-on circuit. It is a graph showing an example of the relationship between power output Pout and switching frequency Fsw, and switching operation of switching frequency Fsw based on the relationship. It is a flowchart which shows the switching operation of switching frequency Fsw in a control circuit. FIG. 2 is a circuit configuration diagram of a power conversion device according to a second embodiment. 3 is a circuit configuration diagram of a power conversion device according to a third embodiment. FIG.

EMBODIMENT OF THE INVENTION Hereinafter, each embodiment based on this invention is described using FIG. 1 thru|or 12.

[Embodiment 1]
Embodiment 1 of the present invention will be described below with reference to FIGS. 1 to 10.
(I) Circuit configuration of power conversion device First, the circuit configuration of the power conversion device according to the first embodiment will be described using FIGS. 1 and 2.
The power converter 100 of this embodiment is a device that includes a circuit that receives a three-phase AC power source 150 as an input and outputs DC power to a DC link section 110. The power conversion device 100 of this embodiment includes a main three-phase diode bridge 101, an add-on circuit 107, a DC link capacitor Cdc, a voltage sensor 105, and a control circuit 106.

The main three-phase diode bridge 101 includes diodes D1 to D6 bridge-connected for each phase (U phase, V phase, and W phase of three-phase AC), and has a function of rectifying input three-phase AC.

The DC link capacitor Cdc is connected to the DC link section 110 and smoothes the rectified DC voltage.

The add-on circuit 107 includes an auxiliary three-phase diode bridge 102, a half bridge 103, a resonant inductor L, a resonant capacitor section 104, a diode D13, and a diode D14.

The auxiliary three-phase diode bridge 102 has diodes D7 to D12 connected in parallel to the diodes D1 to D6 of the main three-phase diode bridge 101 when viewed from the three-phase AC power supply 150, and the three-phase AC power supply 150 and the half bridge 103 It has a function of rectifying the three-phase alternating current depending on the operation of the half bridge 103.

The half bridge 103 is a circuit that includes a switching element S1 and a switching element S2, and is provided to control current by switching these elements. A switching element S1 (on the emitter side of an IGBT (Insulated Gate Bipolar Transistor)) and a switching element S2 (on the collector side of the IGBT) are connected in series.

Further, in the half bridge 103, a diode Ds1 is arranged in parallel with the switching element S1, and the switching element S1 (the collector side of the IGBT) and the cathode side of the diode Ds1 are connected, and the switching element S1 (the emitter side of the IGBT) and the diode Ds1 are connected to each other. The anode side of is connected. Similarly, a diode Ds2 is arranged in parallel with the switching element S2, the switching element S2 (collector side of the IGBT) and the cathode side of the diode Ds2 are connected, and the switching element S2 (emitter side of the IGBT) and the anode side of the diode Ds2 are connected. ing.

The anode side of the diode D13 is connected to the switching element S1 (collector side of IGBT) and the positive side P' of the DC output side of the auxiliary three-phase diode bridge 102, and the cathode side of the diode D13 is connected to the main three-phase diode bridge 101. The positive side P of the DC output side of the DC link section 110 is connected to the positive side of the DC link section 110. On the other hand, the anode side of the diode D14 is connected to the negative side N of the DC output side of the main three-phase diode bridge 101 and the negative side of the DC link section 110, and the cathode side of the diode D14 is connected to the switching element S2 (emitter side of the IGBT). ) and the negative side N' of the DC output side of the auxiliary three-phase diode bridge 102.

The resonant capacitor section 104 has a configuration in which a resonant capacitor C1 and a resonant capacitor C2 are connected in series.

The resonant inductor L has one end connected in series at the midpoint of the resonant capacitor C1 and the resonant capacitor C2, and the other end connected to the switching element S1 (emitter side of the IGBT) and the switching element S2 (collector side of the IGBT). are connected in series at the midpoints of , and cause series resonance with the resonant capacitor C1 and resonant capacitor C2 of the resonant capacitor section 104.

The control circuit 106 is a circuit that is connected to the switching element S1 (gate side of the IGBT) and the switching element S2 (gate side of the IGBT) and controls the switching operation of each element.
The control circuit 106 controls switching of the switching elements S1 and S2 based on the input DC voltage detection value and the DC voltage command value from the outside (not shown). As a result, the resonant capacitor C1 and the resonant capacitor C2 are alternately charged with the line voltage of the three-phase AC power supply 150, and the DC voltage, which is the total voltage of the resonant capacitor section 104, is boosted to a maximum of about twice the full-wave rectified voltage. .

The voltage sensor 105 is a sensor circuit for measuring the voltage value of the circuit and transmitting it to the control circuit 106 as a detected DC voltage value.

With the circuit configuration described above, the power converter 100 receives three-phase AC power from the three-phase AC power supply 150, converts it to DC power using the main three-phase diode bridge 101 and the add-on circuit 107, and converts it into DC power using the DC link capacitor Cdc. Smooth the voltage. Then, after the DC link capacitor Cdc, a three-phase inverter (not shown) and an arbitrary load such as a motor are connected, and electric power is sent thereto. With this power conversion device 100, when high-speed operation of a motor is required, it is possible to boost the DC voltage and improve the output voltage of a three-phase inverter or the like.

Note that although an IGBT is used as a switching element in FIG. 1, the present invention is not limited to this, and another element such as a MOSFET (Metal-Oxide-Semiconductor Field-Effect Transistor) may be used. For example, when a MOSFET is used, the body diode built into the MOSFET can be used, so the diode Ds1 and the diode Ds2 can be omitted. Although the control circuit 106 is not included in the add-on circuit 107 in FIG. 1, the present invention is not limited to this, and the control circuit 106 may be included in the add-on circuit 107. Any method may be used, such as sharing the control circuit of another power converter connected to the subsequent stage.

Next, the configuration of the control block will be explained using FIG. 2.
An example of a control block of the power conversion device 100 in Embodiment 1 is shown.
The control circuit 106 of the power converter 100 includes a voltage control circuit 403 and a gate signal generation circuit 406, and is a circuit that receives a DC voltage command value Vref401 and a DC voltage detection value Vdt402 and generates a gate signal Vg406.

The DC voltage command value Vref401 is a value input from the outside, and is a DC voltage value desired as an output specification of the power converter 100. The DC voltage command value Vref401 may be inputted by the user using a command or the like, or may be set as an initial value. The DC voltage detection value Vdt402 is a DC voltage detection value detected by the voltage sensor 105.

The voltage control circuit 403 calculates a switching frequency Fsw404 based on the input DC voltage command value 401 and the DC voltage detection value 402, for example, by PI (Proportional Integral) control, etc., and outputs it to the gate signal generation circuit 406.

The gate signal generation block 405 outputs the gate signal Vg406 to the switching elements S1 and S2 according to a period based on the switching frequency Fsw404 to drive them.

By operating the control circuit 106 as described above, even when the voltage of the three-phase AC power supply 150 of the power converter 100 fluctuates or the load fluctuates, the DC voltage to be output can be changed from the full-wave rectified voltage. It can be controlled to an arbitrary value of about twice or less. Although FIG. 2 has been described assuming that only the voltage control circuit 403 controls the output DC voltage, the present invention is not limited to this, and a so-called current mode in which a current control block is provided in a minor loop within the voltage control circuit is also possible. Voltage control may also be used.

(II) Relationship between resonant frequency Fr, switching frequency Fsw, and power output Next, the relationship between resonant frequency Fr, switching frequency Fsw, and power output Pout by the power conversion device will be described using FIG.
As shown in FIG. 3, the power conversion device 100 maximizes the power output Pout by matching the switching frequency Fsw of the switching elements S1 and S2 to the resonance frequency Fr of the resonant inductor L and the resonant capacitors C1 and C2. , there is a characteristic that the output decreases both above and below the resonance frequency Fr. Here, the power output Pout is the power at the right end of the resonant inductor L in FIG. Therefore, in the power conversion device 100, the switching frequency Fsw can be controlled to the same output regardless of whether the switching frequency Fsw is increased or decreased in the vicinity of the resonant frequency Fr.

Below, the circuit operation of the resonant voltage doubler rectifier circuit when Fsw≧Fr will be described in (III), and the circuit operation of the resonant voltage doubler rectifier circuit when Fsw<Fr will be described in (IV).

(III) (Circuit operation when Fsw≧Fr of resonant voltage doubler rectifier circuit)
Next, the circuit operation of the resonant voltage doubler rectifier circuit when Fsw≧Fr will be described using FIGS. 4 and 5.
The gate signal Vg1 of the switching element S1 and the gate signal Vg2 of the switching element S2 are respectively turned on when the signal is high, and turned off when the signal is low.
In FIG. 1, the switching element S1 current I1 is defined as positive in the direction in which it flows from the positive side P' of the DC output section of the auxiliary three-phase diode bridge 102 to the resonant inductor L. On the other hand, the switching element S2 current I2 has a positive direction in the direction in which it flows from the resonant inductor L to the negative side N' of the DC output section of the auxiliary three-phase diode bridge 102 in FIG.
Further, the direction in which the resonant current (resonant inductor L current) IL flows from the half bridge 103 to the resonant capacitor section 104 in FIG. 1 is positive.

The DC voltage (DC link capacitor Cdc voltage) Vdc, the resonant capacitor C1 voltage VC1, and the resonant capacitor C2 voltage VC2 are the voltages applied to the DC link capacitor Cdc, resonant capacitor C1, and resonant capacitor C2, respectively (based on the lower side of Figure 1). ).
Further, the timing of each operation is set as t1 to t6.

Note that the circuit operations shown in FIGS. 4 and 5 are circuit operations when the switching frequency Fsw of the switching elements S1 and S2 is set slightly higher than the resonant frequency Fr of the resonant inductor L and the resonant capacitors C1 and C2. Further, although FIG. 5 representatively shows a case where current flows from the U phase to the V phase of the three-phase AC power supply 150, the same operation occurs when current flows from other phases.
The operation of each mode a1 to a6 will be explained below.

(Mode a1: t1 to t2)
In mode a1, switching element S1 is turned on at time t1, and switching element S2 is turned off. At this time, due to series resonance between the resonant inductor L and the resonant capacitors C1 and C2 in mode a5 and mode a6, the resonant current IL is flowing to the diode Ds1, and the voltage across the switching element S1 is It is only the forward voltage drop of , which is almost zero.

(Mode a2: t2 to t3)
In mode a2, the switching element S1 is on and the switching element S2 is off. The polarity is opposite to mode a5, and the resonant inductor L and the resonant capacitors C1 and C2 cause series resonance, and at time t2, the resonant current IL flowing through the diode Ds1 becomes zero, and the current flows through the switching element S1. start. At this time, the switching element S1 performs so-called zero-voltage switching in which the switching element S1 performs switching with substantially zero voltage across the switching element, and the loss associated with switching (switching loss) can be reduced. At this time, the resonant current IL flows in the direction of discharging the resonant capacitor C1 and charging the resonant capacitor C2. Therefore, the resonant capacitor C1 voltage VC1 decreases and the resonant capacitor C2 voltage VC2 increases. The DC link capacitor Cdc is charged by discharging the resonant capacitor C1.

Then, at time t3, which is the boundary between mode a2 and mode a3, the switching element S1 is turned off. At this time, the resonant current IL cut off by the switching element S1 is smaller than its peak. Therefore, switching loss caused by turning off the switching element S1 can be suppressed.

(Mode a3: t3 to t4)
In mode a3, both switching elements S1 and S2 are off, and this mode is a so-called dead time period in which all flowing currents are zero. When the switching element S1 is turned off at time t3, the resonant current IL commutates to the switching element S2 and flows in a direction to discharge the output capacitance of the switching element S2. When the output capacitance of the switching element S2 is discharged and the voltage across the switching element S2 becomes equal to or less than the forward voltage drop of the diode Ds2, the resonant current IL is commutated to the diode Ds2. Resonant capacitor C1 is still discharging to DC link capacitor Cdc, and resonant capacitor C2 is charged with resonant current IL. Further, the resonant current IL after commutation decreases because a voltage of opposite polarity is applied to the resonant inductor L up to mode a2. At time t4, which is the boundary between mode a3 and mode a4, switching element S2 is turned on.

(Mode a4: t4 to t5)
In mode a4, the switching element S1 is off and the switching element S2 is on. Even when switching element S2 is turned on at time t4, current still flows through diode Ds2. At this time, the voltage across the switching element S2 is approximately zero, which is equal to the forward voltage drop of the diode Ds2.

(Mode a5: t5 to t6)
In mode a5, the mode is symmetrical to mode a2, and the switching element S1 is off and the switching element S2 is on. The resonant inductor L and the resonant capacitors C1 and C2 cause series resonance with the opposite polarity from mode a2 to mode a4, and at time t5, the resonant current IL flowing through the diode Ds2 becomes zero, and the current flows through the switching element S2. start. At this time, the switching element S2 performs zero voltage switching, and switching loss can be reduced. The resonant current IL flows in the direction of charging the resonant capacitor C1 and discharging the resonant capacitor C2. Therefore, VC1 increases and VC2 decreases. The DC link capacitor Cdc is charged by the discharge of the resonant capacitor C2. Switching element S2 is turned off at time t6, which is the boundary between mode a5 and mode a6. At this time, the resonance current IL cut off by the switching element S2 is smaller than its peak. Therefore, switching loss of the switching element S2 can be suppressed.

(Mode a6: t6 to t1)
In mode a6, both switching elements S1 and S2 are off, and this mode is a dead time period. When the switching element S2 is turned off at time t6, the resonant current IL commutates to the switching element S1 and flows in a direction to discharge the output capacitance of the switching element S1. When the output capacitance of the switching element S1 is discharged and the voltage across the switching element S1 becomes equal to or less than the forward voltage drop of the diode Ds1, the resonant current IL is commutated to the diode Ds1. Resonant capacitor C2 is still discharging to DC link capacitor Cdc, and resonant capacitor C1 is charged with resonant current IL. Further, the resonant current IL after commutation rapidly decreases because a voltage is applied to the resonant inductor L with the opposite polarity up to mode a5. When the switching element S1 is turned on at time t1, the mode returns to mode a1.

The above is an example in which the switching elements S1 and S2 are driven at a switching frequency in a slightly higher range from near the resonant frequency of the resonant inductor L and the resonant capacitors C1 and C2 in the circuit operation when Fsw≧Fr of the resonant voltage doubler rectifier circuit. explained. As shown in FIG. 4, switching elements S1 and S2 switch in a complementary manner. Due to this switching operation, the resonant capacitor C1 voltage VC1 and the resonant capacitor C2 voltage VC2 have waveforms in which an alternating current component is superimposed on a direct current component. Further, it can be seen that the DC voltage Vdc is the total voltage of the resonant capacitor C1 voltage VC1 and the resonant capacitor C2 voltage VC2, so it is boosted and further smoothed. This indicates that voltage smoothing is not performed by the resonant capacitors C1 and C2, but only by the DC link capacitor Cdc. In the operation of this resonant voltage doubler rectifier circuit, by utilizing the series resonance between the resonant inductor L and the resonant capacitors C1 and C2, the peak of the charging current can be suppressed without smoothing the voltage of the resonant capacitors C1 and C2. be able to.

Therefore, the capacitance required for the resonant capacitors C1 and C2 can be reduced compared to the conventional technology, and the mounting area can be reduced by reducing the number of parallel resonant capacitors C1 and C2. Furthermore, series resonance allows the switching elements S1 and S2 to perform zero voltage switching, and the interruption current can be suppressed, making it easy to increase the frequency. Therefore, it is possible to further reduce the capacitance of the resonant capacitors C1 and C2 by setting the switching frequencies of the switching elements S1 and S2 to the order of kHz or more. As described above, it is possible to realize miniaturization of the add-on circuit 107.

(IV) (Circuit operation when Fsw<Fr of resonant voltage doubler rectifier circuit)
Next, the circuit operation of the resonant voltage doubler rectifier circuit when Fsw≧Fr will be described using FIGS. 6 and 7.
The circuit operations shown in FIGS. 6 and 7 are circuit operations when the switching frequency Fsw of the switching elements S1 and S2 is set slightly lower than the resonant frequency Fr of the resonant inductor L and the resonant capacitors C1 and C2.
The definition of each waveform is the same as that shown in (III).

FIG. 7 shows each mode of circuit operation when Fsw<Fr of the resonant voltage doubler rectifier circuit, and the circuit operation in each mode b1 to b6 will be explained below.

(Mode b1: t1 to t2)
In mode b1, the switching element S1 is on and the switching element S2 is off. At time t1, switching element S1 is turned on, resonant inductor L and resonant capacitors C1 and C2 cause series resonance, and resonant current IL begins to flow. At this time, the resonant current IL flows in the direction of discharging the resonant capacitor C1 and charging the resonant capacitor C2. Therefore, the resonant capacitor C1 voltage VC1 decreases and the resonant capacitor C2 voltage VC2 increases. The DC link capacitor Cdc is charged by discharging the resonant capacitor C1. The resonant current IL decreases after reaching its peak, and becomes zero at time t2, which is the boundary between mode b1 and mode b2.

(Mode b2: t2 to t3)
In mode b2, similarly to mode b1, switching element S1 is on and switching element S2 is off. When the resonant current IL becomes zero at time t2, no current flows through the resonant inductor L. This is because the resonant capacitor C2 voltage VC2 is higher than the line voltage of the three-phase AC power supply 150, and the auxiliary three-phase diode bridge 102 blocks it. Therefore, in mode b2, both resonant capacitors C1 and C2 discharge into the DC link capacitor Cdc. Then, the switching element S1 is turned off at time t3, which is the boundary between mode b2 and mode b3. At this time, since no current flows through the switching element S1, switching occurs with zero current, so-called zero current switching, and switching loss can be reduced.

(Mode b3: t3 to t4)
In mode b3, both switching elements S1 and S2 are off, and this mode is a dead time period. Similar to mode b2, no current flows through the resonant inductor L, and both the resonant capacitors C1 and C2 operate to discharge to the DC link capacitor Cdc. At time t4, which is the boundary between mode b3 and mode b4, switching element S2 is turned on.

(Mode b4: t4 to t5)
In mode b4, the switching element S1 is off and the switching element S2 is on. At time t4, switching element S2 is turned on, resonant inductor L and resonant capacitors C1 and C2 cause series resonance with the opposite polarity to mode b1, and resonant current IL begins to flow. At this time, the resonant current IL flows in the direction of charging the resonant capacitor C1 and discharging the resonant capacitor C2. Therefore, the resonant capacitor C1 voltage VC1 increases and the resonant capacitor C2 voltage VC2 decreases. The DC link capacitor Cdc is charged by discharging the resonant capacitor C2. The resonant current IL decreases after reaching its peak and becomes zero at time t5, which is the boundary between mode b4 and mode b5.

(Mode b5: t5 to t6)
In mode b5, similarly to mode b4, switching element S1 is off and switching element S2 is on. When the resonant current IL becomes zero at time t5, no current flows through the resonant inductor L. This is because the voltage VC1 of the resonant capacitor C1 is higher than the line voltage of the three-phase AC power supply 150, and the auxiliary three-phase diode bridge 102 blocks it. Therefore, in mode b5, both resonant capacitors C1 and C2 discharge into the DC link capacitor Cdc. At time t6, which is the boundary between mode b5 and mode b6, switching element S2 is turned off. At this time, since no current flows through the switching element S2, zero current switching occurs, and switching loss can be reduced.

(Mode b6: t6 to t1)
In mode b6, both switching elements S1 and S2 are off, and this mode is a dead time period. As in mode b5, no current flows through the resonant inductor L, and both the resonant capacitors C1 and C2 operate to discharge to the DC link capacitor Cdc. At time t1, when the switching element S1 is turned on, the mode returns to mode b1.

As shown above, in the circuit operation of the resonant voltage doubler rectifier circuit when Fsw<Fr, the switching elements S1 and S2 operate at a switching frequency that is slightly lower than the resonant frequency of the resonant inductor L and the resonant capacitors C1 and C2. An example of driving is explained above. As shown in FIG. 6, switching elements S1 and S2 switch complementary to each other. In this circuit operation, the voltage is not smoothed by the resonant capacitors C1 and C2, but only by the DC link capacitor Cdc. Therefore, even in this circuit operation, the capacitance required for the resonant capacitors C1 and C2 can be reduced compared to the conventional technology, so the number of parallel resonant capacitors C1 and C2 is reduced, and the mounting area can be reduced. Furthermore, since the switching elements S1 and S2 can perform zero current switching and the switching frequency can be lowered compared to the case (III), switching loss can be further suppressed. Therefore, it is easy to increase the switching frequency. As described above, it is possible to realize miniaturization of the add-on circuit 107.

(V) Switching operation of switching frequency Fsw Next, the switching operation of switching frequency Fsw in the control circuit 106 of the power conversion device 100 will be explained using FIGS. 8 to 10.

FIG. 8 shows graph 1 in the case of control region 1 in which the switching frequency Fsw of switching elements S1 and S2 is greater than or equal to the resonance frequency Fr, and graph 2 in the case of control region 2 in which the switching frequency Fsw is less than the resonance frequency Fr. . When the output power Pout is relatively large, the power loss Ploss is smaller when the output power Pout is controlled in the control area 1.However, when the power loss is reversed at the predetermined output 1001, and the output is smaller than the predetermined output 1001, the power loss Ploss is smaller when the output power Pout is controlled in the control area 1. The power loss Ploss is smaller when controlled by This is due to the characteristics of control regions 1 and 2 shown below. In control region 1, when the load is heavy, zero-voltage switching results in low switching loss and high efficiency, but when the load becomes light, it is necessary to significantly increase the switching frequency to throttle the output, and efficiency tends to deteriorate due to switching loss. On the other hand, in control region 2, when the load is heavy, the peak current is large and the efficiency is likely to deteriorate due to conduction loss, but at light loads, the switching frequency Fsw is lowered to throttle the output, so the switching loss is reduced and the efficiency is less likely to deteriorate.

Next, the relationship between the output power Pout and the switching frequency Fsw based on this principle and the switching operation of the switching frequency Fsw based on the relationship will be explained using FIG. 9.
As shown in FIG. 9, a region where switching frequency Fsw≧resonant frequency Fr is a control region 1, and a region where switching frequency Fsw<resonant frequency Fr is a control region 2.

In the control region 1, when the output power Pout becomes equal to or less than the first predetermined value 1101, the control circuit 106 switches the control region of the switching elements S1 and S2 from the control region 1 to the control region 2. Further, when the output increases in the control region 2 and becomes equal to or higher than the second predetermined value 1102, the control circuit 106 switches the control region of the switching elements S1 and S2 from the control region 2 to the control region 1. It is desirable that the first predetermined value 1101 and the second predetermined value 1102 be set around the predetermined output 1001 in FIG. 8 . As described above, the switching operation of the switching frequency Fsw in the control circuit 106 is controlled by hysteresis (hysteresis), in which the state of a system changes depending not only on the current action but also on past actions. By operating the add-on circuit 107 in the control region 1 and 2, chattering can be prevented while driving the add-on circuit 107 in a more efficient control region. Therefore, the add-on circuit 107 can be driven with higher efficiency, the coolers for the switching elements S1 and S2 can be simplified, and the add-on circuit 107 can be made smaller.

Next, the switching operation of the switching frequency Fsw in the control circuit 106 will be explained using FIG. 10.
Voltage control calculation is started based on a predetermined control cycle (S1201).
First, it is determined whether or not the vehicle is currently being driven in control region 1 (S1202).
When driving in control region 1 (S1202: YES), it is then determined whether the output power is greater than or equal to the first predetermined value 1101 (S1203).
If the output power is not greater than or equal to the first predetermined value 1101 (S1202: NO), then it is determined whether the output power is greater than or equal to the second predetermined value 1102 (S1204).
In S1203, if the output power is equal to or higher than the first predetermined value 1101 (S1203: YES), driving is performed in control region 1 (S1205).
In S1203, if the output power is less than the first predetermined value 1101 (S1203: NO), driving is performed in control region 2 (S1206).
In S1204, if the output power is equal to or higher than the second predetermined value 1102 (S1204: YES), the process moves to S1205.
In S1204, if the output power is less than the second predetermined value 1102 (S1204: NO), the process moves to S1206.

Then, the control circuit 106 controls the switching elements S1 and S2 so that they are driven in the control region determined in S1205 or S1206 (S1207).

By operating as described above, the add-on circuit 107 can be driven in a more efficient control region while preventing chattering between the control region 1 and the control region 2.

[Embodiment 2]
Embodiment 2 of the present invention will be described below using FIG. 11.
In the first embodiment, a configuration has been described in which AC power is input and DC power is output using an add-on circuit that utilizes series resonance of a resonant inductor and a resonant capacitor.
This embodiment is based on the power conversion device 100 of Embodiment 1, and relates to a circuit in which elements are added. Hereinafter, characteristic parts will be explained in comparison with Embodiment 1.

As shown in FIG. 11, the power conversion device 100 in this embodiment has snubber capacitors Cs1 and Cs2 connected in parallel to switching elements S1 and S2 included in an add-on circuit 107, respectively. . By connecting the snubber capacitors Cs1 and Cs2, it is possible to delay the voltage rise across the switching elements S1 and S2 when they are turned off, thereby further suppressing loss. Furthermore, before the switching elements S1 and S2 turn on due to the commutation of the resonant current IL, the charges of the snubber capacitors Cs1 and Cs2 are extracted and sent to the resonant capacitors C1 and C2, so the energy accumulated in the snubber capacitors Cs1 and Cs2 is In principle, there is no loss. From the above, according to the power conversion device 100 of the second embodiment, since the snubber capacitors Cs1 and Cs2 are added, the power conversion device 100 of the first embodiment can operate more efficiently than the power conversion device 100 of the first embodiment.

[Embodiment 3]
Embodiment 3 of the present invention will be described below using FIG. 12.
In the second embodiment, a configuration of a circuit in which elements are added based on the first embodiment has been described, but this embodiment also relates to a circuit in which elements are added on the basis of the first embodiment. Hereinafter, characteristic parts will be explained in comparison with Embodiment 1.

As shown in FIG. 12, the power conversion device 100 in this embodiment has diodes Dc1 and Dc2 connected in parallel to resonance capacitors C1 and C2 included in an add-on circuit 107, respectively. By connecting the diodes Dc1 and Dc2, the voltages of the resonant capacitors C1 and C2 can be clamped to zero or more. In FIG. 6, in the mode in which the resonant capacitors C1 and C2 discharge, the resonant capacitor C1 voltage VC1 and the resonant capacitor C2 voltage VC2 have decreased to near zero. In such a case, in the circuit configuration shown in FIG. 1 of the first embodiment, the resonant capacitor C1 voltage VC1 and the resonant capacitor C2 voltage VC2 may become negative depending on the load or the step-up ratio of the DC voltage Vdc. For example, when the resonant capacitor C1 voltage VC1 becomes negative, the voltage VC2 becomes higher than the DC voltage Vdc, so the resonant capacitor C2 is discharged to the DC link capacitor Cdc via the resonant inductor L, switching element S1, and diode D13. do. Therefore, after the resonant current IL becomes zero at time t2 in FIG. 6, the resonant current IL starts flowing in the negative direction. If the switching element S1 is turned off while this current is flowing, zero current switching will not be achieved.

Therefore, by connecting the diodes Dc1 and Dc2, the resonant capacitor C1 voltage VC1 and the resonant capacitor C2 voltage VC2 are prevented from becoming negative, and the resonant current IL is prevented from flowing in the negative direction. Zero current switching operation of S1 and S2 can be ensured. From the above, according to the circuit configuration of the second embodiment, the power conversion device 100 can be operated with high efficiency in a wide DC voltage range and load range, and there is no need to increase the coolers of the switching elements S1 and S2. , thereby making it possible to downsize the add-on circuit 107.

DESCRIPTION OF SYMBOLS 100... Power conversion device, 101... Main three-phase diode bridge, 102... Auxiliary three-phase diode bridge, 103... Half bridge, 104... Resonant capacitor part, 105... Voltage sensor, 106... Control circuit, 150... Three-phase AC power supply, D1 to D14, Ds1, Ds2...Diode, S1, S2...Switching element, L...Resonant inductor, C1, C2...Resonant capacitor, Cdc...DC link capacitor

Claims (8)

a voltage doubler rectifier circuit that doubles and rectifies the AC voltage of a three-phase AC power supply;
and a control circuit that performs switching control of the voltage doubler rectifier circuit,
The voltage doubler rectifier circuit is
a first diode bridge connected to a three-phase AC power supply;
a DC link capacitor connected to a DC section between the output side and the load side of the first diode bridge;
a second diode bridge connected to the three-phase AC power supply in parallel with the first diode bridge;
a first switching element and a second switching element connected in parallel to the second diode bridge;
a first resonant capacitor and a second resonant capacitor connected in parallel between the first diode bridge and the DC link capacitor;
a resonant inductor connected between the first switching element and the second switching element and between the first resonant capacitor and the second resonant capacitor;
a first diode connecting between the cathode side output of the first diode bridge and the cathode side output of the second diode bridge;
A power conversion device comprising: a second diode connecting between an anode side output of the first diode bridge and an anode side output of the second diode bridge. The control circuit is characterized in that it performs switching control on the first switching element and the second switching element at a switching frequency near the resonance frequency of the resonant inductor, the first resonant capacitor, and the second resonant capacitor. The power conversion device according to claim 1. The power conversion device according to claim 2, wherein the control circuit controls the output power to the load side by manipulating the switching frequency. The first resonant capacitor and the second resonant capacitor included in the voltage doubler rectifier circuit resonate with the resonant inductor,
The power conversion device according to claim 1, wherein the power conversion device is used for boosting the voltage of a DC link capacitor in the DC section. 2. The power conversion device according to claim 1, further comprising a capacitor connected in parallel to each of the first switching element and the second switching element. 2. The power conversion device according to claim 1, further comprising a diode connected in parallel to each of the first resonant capacitor and the second resonant capacitor. The control circuit is configured to switch the switching frequency from above the resonant frequency to below the resonant frequency, or from below the resonant frequency to above the resonant frequency, depending on the output power of the voltage doubler rectifier circuit to the load side. The power conversion device according to claim 3, characterized in that: When controlling the switching frequency, the control circuit includes a first threshold value for switching the switching frequency from above the resonance frequency to below the resonance frequency, and a second threshold value for switching the switching frequency from below the resonance frequency to above the resonance frequency. The power conversion device according to claim 7, wherein the switching frequency is switched according to the following.

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