JP7501995B2 - CONVERTER CONTROL METHOD AND CONTROL DEVICE - Google Patents

Description

The present invention relates to a converter control method and a control device.

In general, in a motor control system for an electric vehicle or the like, when a desired AC power is supplied from a battery to a motor, the DC power supplied from the battery is boosted by a converter, then converted from DC to AC by an inverter and supplied to the motor. In such a converter, the feedback control gain of the converter is set taking into account the characteristics of the battery's internal resistance due to changes in battery temperature and the change in coil inductance due to the passing current, thereby stabilizing the responsiveness of the converter (for example, Patent Document 1).

JP 2010-68611 A

When controlling the rotational drive of a motor via an inverter, constant power control of the inverter is known as a method of controlling the power operation of the motor. In a normal resistor, the current that flows increases when the applied voltage is increased, but when constant power control is used, the opposite characteristic occurs, that is, a negative resistance characteristic in which the current decreases when the applied voltage is increased, and this negative resistance characteristic may cause the responsiveness of the converter to deteriorate. Furthermore, it is known that as the step-up ratio of the converter increases, the phase lag of the output voltage relative to the input voltage increases, and there is a risk that the output voltage from the converter may become unstable.

According to the technology disclosed in Patent Document 1, the gain used to control the converter is set according to the frequency determined by the internal resistance of the battery due to changes in battery temperature and the inductance value due to the reactor current. As a result, it is not possible to suppress the deterioration of responsiveness due to negative resistance characteristics that occurs when the motor is under constant voltage control, and the deterioration of output voltage stability due to phase lag when the step-up ratio is large, and the supply voltage to the inverter cannot be appropriately controlled, which may result in a decrease in motor control accuracy.

The converter control method of the present invention is a converter control method that boosts the DC input voltage supplied from a battery and supplies power to a motor via an inverter that is constant voltage controlled. In this control method, when the motor is in powered operation, a gain is determined according to the output power from the converter to the inverter, the output voltage, and the boost ratio of the converter, and the output voltage of the converter is controlled by feedback control using the gain.

The converter control method of the present invention makes it possible to suppress instability caused by negative resistance characteristics according to the converter's output power and output voltage, and phase delay according to the boost ratio, when the inverter is subjected to constant voltage control in feedback of the converter's output voltage.

FIG. 1 is a schematic diagram of a motor system according to the present embodiment. FIG. 2 is a circuit diagram of the motor system. FIG. 3 is a detailed circuit diagram in the vicinity of the converter. FIG. 4 is a block diagram of the controller. FIG. 5A is a block diagram of the voltage control unit. FIG. 5B is a block diagram of the current control unit. FIG. 6 is a flowchart of the calculation control of the gain K. FIG. 7 is a graph showing the relationship between the negative resistance characteristic 1/R ₀ and the first gain K ₁ . FIG. 8 is a graph showing the relationship between the negative resistance characteristic 1/R ₀ and the second gain K ₂ . FIG. 9 is a graph showing the relationship between the negative resistance characteristic 1/R ₀ and the first gain K ₁ , the second gain K ₂ , and the gain K. FIG. 10 is a graph showing the relationship between the inverter side voltage _V2 and the inverter current _I0 . FIG. 11 is a block diagram showing a control system of the converter. FIG. 12 is a diagram showing the relationship between the natural angular frequency ω ₀ and the output power and phase lag of the converter according to the step-up ratio of the converter. FIG. 13 is a timing chart of the output power P and the inverter side voltage _V2 in this embodiment. FIG. 14 is a timing chart of the output power P and the inverter side voltage _V2 in the comparative example. FIG. 15 is a schematic diagram of a modified motor system 100. FIG. 16 is a circuit diagram of the motor system 100.

This section describes a converter control method and control device according to an embodiment of the present invention.

Figure 1 is a schematic diagram of a motor system 100 according to this embodiment. When the motor system 100 is used in an electric vehicle, a motor generator 4, which is an example of a load, serves as a drive source for the vehicle.

In the motor system 100, a battery 1, a converter 2, an inverter 3, and a motor generator 4 are connected in series. The motor system 100 further includes an output shaft 5 connected to the motor generator 4, and a controller 6.

Battery 1 is a secondary battery that can be charged and discharged. Converter 2 is a converter capable of stepping up and down voltage, and is configured to step up the DC voltage supplied from battery 1 and supply the stepped-up power to inverter 3. Inverter 3 converts the DC power supplied from converter 2 to AC power, and supplies the converted AC power to motor generator 4.

An output shaft 5 is connected to the motor generator 4. The motor generator 4 functions as either a motor or a generator. When the motor generator 4 is in powered operation, power is supplied from the converter 2 to the inverter 3 and the motor generator 4. When the motor generator 4 is in regenerative operation, the regenerative power generated by the motor generator 4 is charged to the battery 1 via the inverter 3 and the converter 2.

The motor generator 4 is provided with a resolver 41 that detects the rotation angle and rotation speed of the rotor. When the motor system 100 is used in an electric vehicle, the drive wheels (not shown) connected to the output shaft 5 are driven by the rotation output of the motor generator 4.

The controller 6 controls the converter 2 and the inverter 3, and receives the rotation angle and rotation speed of the motor generator 4 from the resolver 41. The controller 6 stores predetermined processing as programs, and is configured to be able to execute the processing corresponding to the programs by executing the programs.

Figure 2 is a circuit diagram of the motor system 100.

DC power supplied from the battery 1 is boosted in the converter 2 and supplied to the inverter 3. A battery side voltage _V1 , which is the terminal voltage on the battery 1 side of the converter 2, and a reactor current _IL flowing in the converter 2 are acquired by a voltage sensor 22 and a current sensor 26 provided in the converter 2. An inverter side voltage _V2 , which is the terminal voltage on the inverter 3 side of the converter 2, is acquired by a voltage sensor 28 provided in the vicinity of the converter 2.

The power output from the converter 2 to the inverter 3 is referred to as the inverter current _I0 . The inverter current _I0 is positive when it is output from the converter 2 to the inverter 3, and is negative when it is input from the inverter 3 to the converter 2. The detailed configuration of the converter 2 will be described later with reference to FIG.

The inverter 3 is a three-phase inverter and is composed of multiple switching elements. The inverter 3 converts the DC power input from the converter 2 into three-phase AC power and supplies it to the motor generator 4. The inverter 3 also converts the regenerated AC power in the motor generator 4 into DC power that can be charged in the battery 1.

The resolver 41 detects the rotation speed N of the motor generator 4 and transmits the detected rotation speed N to the controller 6. The current sensor 42 is provided between the inverter 3 and the motor generator 4, detects the currents of the UVW phases between the inverter 3 and the motor generator 4, and transmits the drive current _Ip to the controller 6. The drive current _Ip indicates the detected currents of the UVW phases. Note that in this diagram, the output shaft 5 connected to the motor generator 4 is omitted.

The controller 6 generates a switching pattern according to a torque command value T ^* for the motor generator 4 calculated by a higher-level device, the rotation speed N detected by the resolver 41, the drive current _Ip detected by the current sensor 42, and the inverter-side voltage _V2 detected by the voltage sensor 28. The controller 6 outputs the generated switching pattern as a gate signal to the inverter 3. The inverter 3 is driven in response to this gate signal, causing the motor generator 4 to rotate with a desired torque.

The controller 6 calculates the power required by the motor generator 4, and determines a target inverter-side voltage _V2 ^* of the converter 2, which is the voltage applied to the inverter 3, based on the required power. The controller 6 generates a duty ratio D based on the target inverter-side voltage _V2 ^* , and outputs the generated duty ratio D to the converter 2 as a gate signal. The switching elements 24a, 24b are controlled based on the duty ratio D, so that the desired inverter-side voltage _V2 can be obtained.

Figure 3 is a detailed circuit diagram of the vicinity of converter 2 in Figure 2.

3, the capacitor 21 is provided between the positive and negative electrodes of the power supply line from the battery 1. The capacitor 21 suppresses noise contained in the power supply from the battery 1 to the converter 2. A voltage sensor 22 is provided near the capacitor 21. The voltage sensor 22 measures the voltage of the capacitor 21 to detect the battery side voltage _V1 , and transmits the detected battery side voltage _V1 to the controller 6.

One end of the reactor 23 (inductance) is connected to the positive electrode of the battery 1, and the other end is connected to one end of the switching element 24a and one end of the switching element 24b. The other end of the switching element 24a becomes the positive electrode of the output side of the inverter 3, and the other end of the switching element 24b becomes the negative electrode of the output side of the inverter 3. The switching elements 24a and 24b are composed of, for example, an IGBT (Insulated Gate Bipolar Transistor).

Diodes 25a and 25b are connected in parallel with switching elements 24a and 24b, respectively. Diode 25a is provided so that the forward direction is from one end of switching element 24a to the other end. Diode 25b is provided so that the forward direction is from the other end of switching element 24a to one end.

In converter 2, switching elements 24a and 24b are controlled to repeatedly switch between a turn-on state and a turn-off state, thereby boosting the voltage. The details of this mechanism are explained below.

First, consider the turn-on state in which switching element 24a is turned off and switching element 24b is turned on. In the turn-on state, the current output from the positive electrode of battery 1 passes through reactor 23 and switching element 24b and flows to the negative electrode of battery 1 (route (a)). Therefore, electrical energy from battery 1 is stored in reactor 23.

Thereafter, the switching element 24a is turned on and the switching element 24b is turned off, resulting in a turn-off state. In the turn-off state, the power stored in the reactor 23 is discharged, and the inverter current _I0 is supplied to the inverter 3 via the switching element 24a (route (b)). Due to this discharge, a voltage higher than the supply voltage of the battery 1 is applied to the inverter 3.

In the converter 2, the inverter side voltage _V2 can be controlled by changing the duty ratio D, which is the ratio of the time during which the turn-on state is maintained to the period during which the turn-on state and the turn-off state are repeated.

A current sensor 26 is provided between the reactor 23 and the switching elements 24a and 24b. The current sensor 26 detects a reactor current I _L flowing through the reactor 23, and transmits the reactor current I _L to the controller 6.

The capacitor 27 is provided between the positive and negative terminals of the converter 2 on the inverter 3 side. The capacitor 27 suppresses voltage ripples caused by switching of the switching elements 24a, 24b. The voltage sensor 28 is provided near the capacitor 27, and measures the voltage of the capacitor 27 to obtain the inverter-side voltage _V2 . The voltage sensor 28 transmits the inverter-side voltage _V2 to the controller 6.

Figure 4 is a block diagram of the controller 6.

This diagram shows the configuration mainly related to the control of the converter 2, among the various controls performed by the controller 6. The controller 6 has a voltage control unit 61, a current control unit 62, a subtractor 63, a divider 64, a PWM generation unit 65, and a control gain calculation unit 66. In this embodiment, when the inverter 3 is under constant voltage control and the motor generator 4 is in power running, the control gain calculation unit 66 adjusts the gain.

The controller 6 receives feedback inputs of the battery side voltage _V1 , the inverter side voltage _V2 , and the reactor current _IL from the converter 2, in addition to the input of the output voltage command value _{Vout_astr} calculated in another section of the controller 6. The controller 6 then outputs a gate signal to the converter 2 in accordance with these inputs.

The voltage control unit 61 performs PI control according to a gain K calculated by a control gain calculation unit 66 based on the output voltage command value V _{out_astr} and a feedback input of the inverter side voltage _V2 . In this way, the voltage control unit 61 calculates an input current command value _{IL_astr} .

The current control unit 62 calculates a reactor voltage command value V _{L_astr} by PI control using a predetermined gain K′, based on the input of the input current command value I _{L_astr} from the voltage control unit 61 and the feedback input of the reactor current I _L from the converter 2.

Here, the detailed configuration of the voltage control unit 61 and the current control unit 62 will be described with reference to Figures 5A and 5B.

As shown in FIG. 5A , in the voltage control unit 61, the subtractor 611 subtracts the inverter side voltage _V2 from the output voltage command value _{Vout_astr} , and in the PI controller 612, PI control is performed on the subtraction result in the subtractor 611 based on the gain K calculated in the control gain calculation unit 66, to calculate the input current command value _{IL_astr} .

5B , in the current control unit 62, a subtractor 621 subtracts the reactor current I _L from the input current command value I _{L_astr} , and a PI controller 622 performs PI control based on a predetermined gain K' on the subtraction result in the subtractor 621 to calculate a reactor voltage command value V _{L_astr} . Note that the reactor voltage command value V _{L_astr} is a command value for a voltage to be applied to the reactor 23. In addition, the gain K' used in the PI control in the PI controller 622 of the current control unit 62 may be the same value as the gain K calculated by the control gain calculation unit 66.

4 again, the subtractor 63 subtracts the reactor voltage command value V _{L_astr} output from the current control unit 62 from the battery side voltage V ₁ that is feedback-input from the converter 2. Then, in a divider 64, the subtraction result by the subtractor 63 is divided by the inverter side voltage V ₂ that is feedback-input from the converter 2 to calculate a duty ratio D and obtain a duty command value.

The PWM generating unit 65 outputs a gate signal having a pulse width according to the duty ratio D to the converter 2. The method of generating the gate signal can be explained as follows.

The relationship between the battery side voltage _V1 and the inverter side voltage _V2 of the converter 2 can be expressed by the following equation using the duty ratio D of the switching element 4a and the reactor voltage _VL .

In equation (1), when the reactor voltage V _L is replaced with the reactor voltage command value V _{L_astr} , the duty ratio D can be expressed by the following equation.

The PWM generating unit 65 converts the duty ratio D calculated by equation (2) through the subtractor 63 and the divider 64 into a gate signal and outputs it to the converter 2. The converter 2 drives the switching element of the inverter 3 based on the input gate signal, thereby outputting the inverter side voltage _V2 .

The control gain calculation unit 66 calculates a gain K used for the PI control of the voltage control unit 61 using the battery side voltage _V1 , the inverter side voltage _V2 , the output power P, and the duty ratio D according to the flowchart shown in Figure 6 described below.

In this embodiment, the gain K is used as a proportional-integral gain (proportional gain and integral gain) in the PI control in the voltage control unit 61, but this is not limiting. For example, the gain K may be used as the proportional gain in P control, or the proportional gain, integral gain, and differential gain in PID control.

Figure 6 is a flowchart of the calculation control of gain K in the control gain calculation unit 66. Note that in the calculation control shown in this figure, it is assumed that the motor generator 4 is in power running operation.

In step S1, the control gain calculation unit 66 acquires the battery side voltage _V1 and the inverter side voltage V2 which are values detected by sensors in the circuit of the converter ₂ , the calculated duty ratio D, and the output power P of the converter 2. Note that the control gain calculation unit 66 may calculate the output power P provided by the circuit of the converter 2 from the inverter current _I0 and the inverter side voltage _V2 , or may acquire detection values using a power sensor.

In step S2, the control gain calculation unit 66 calculates the negative resistance characteristic 1/R ₀ from the following equation: The negative resistance characteristic 1/R ₀ will be described later with reference to FIG.

In step S3, the control gain calculation unit 66 calculates a first gain _K1 based on the negative resistance characteristic 1/ _R0 using a pre-stored table. The first gain _K1 is determined so as to suppress instability caused by the negative resistance characteristic 1/ _R0 that occurs when the inverter 3 is subjected to constant power control. Fig. 7 is a diagram showing the relationship between the first gain _K1 and the negative resistance characteristic 1/ _R0 that is pre-stored. The details of this relationship will be described later.

In step S4, the control gain calculation unit 66 calculates the second gain _K2 based on the negative resistance characteristic 1/ _R0 using a table that is stored in advance and is different from the one used in step S4. The second gain _K2 is determined so as to suppress instability caused by a phase delay of the output voltage relative to the input voltage. FIG. 8 is a diagram showing the relationship between the second gain _K2 stored in advance and the negative resistance characteristic 1/ _R0 , which will be described in detail later. Note that since the phase delay changes depending on the step-up ratio D', in step S4, the second gain _K2 may be calculated depending on the step-up ratio D' without using the table in FIG. 8.

In step S5, the control gain calculation unit 66 calculates the gain K by using the first gain _K1 calculated in step S3 and the second gain _K2 calculated in step S4. Specifically, a value between the first gain _K1 and the second gain _K2 may be obtained as the gain K. Note that the gain K may be obtained as an average value obtained by weighting the first gain _K1 and the second gain K2. As shown in Fig. 9, if the first gain _K1 and the second gain _K2 are combined into one graph, the region between the first gain _K1 and the _second gain _K2 in this graph indicates the possible values of the gain K.

Here, in relation to the calculation method of the first gain _K1 , the relationship between the negative resistance characteristic 1/ _R0 and the first gain _K1 shown in Fig. 7 will be described. The first gain _K1 is a stabilization gain that can suppress instability of the controlled object caused by the negative resistance characteristic 1/ _R0 that occurs when the inverter 3 is subjected to constant power control.

First, an explanation will be given of the principle by which the inverter side voltage V2 of converter 2 oscillates due to the negative resistance characteristic 1/ _R0 associated with the constant power control of inverter 3 when motor generator 4 is in power running, that is, when the output power P provided by converter ₂ is positive.

The relationship between the inverter side voltage _V2 of the converter 2 and the current I flowing into the inverter 3 by the converter 2 can be shown as in FIG.

FIG. 10 is a graph showing the relationship between the inverter side voltage V ₂ and the inverter current I ₀ in the converter 2 .

The power supplied to the motor generator 4 is approximately equal to the output power P (output power P) of the converter 2. This output power P is calculated as the product of the inverter side voltage _V2 and the inverter current _I0 . When the motor generator 4 is under constant power control, the output power P is constant, so that the inverter side voltage _V2 and the inverter current _I0 are inversely proportional to each other.

Therefore, as shown in this graph, the inverter current _I0 decreases as the inverter side voltage _V2 increases. This characteristic is called a negative resistance characteristic 1/ _R0 , and is the opposite of the general characteristic in which the current increases according to the applied voltage.

Then, when the relationship between the inverter side voltage _V2 and the inverter current _I0 at a certain operating point is linearly approximated, the following equation is obtained: When the output power P of the converter 2 is positive, the resistance component _R0 of the impedance of the motor generator 4 is positive.

Here, _Iofs is the intercept of the linear approximation line on the _I0 axis. For example, at a certain operating point of the motor generator 4, if the inverter side voltage _V2 is _V10 and the inverter current _I0 is _I10 , the resistance component _R0 of the motor generator 4 is expressed by the following equation.

If the transfer characteristic from the battery side voltage _V1 in the converter 2 to the inverter side voltage _V2 on the inverter 3 side is expressed by the following equation using the resistance component _R0 etc.

where L [H] is the inductance of reactor 23, C [F] is the capacitance of capacitor 27, R [Ω] is the resistance component of converter 2 in the turned-on state, and D is the duty ratio.

In order for the inverter-side voltage _V2 on the inverter 3 side to be stable without diverging, the coefficient "R/L-1/ _R0C " of "s", which is the dominant parameter in the denominator of the first half of formula (6), must be positive. Therefore, the resistance component _R0 must be set to satisfy the condition of the following formula.

11 is a block diagram showing the control system of the converter 2 when the motor generator 4 is in powering operation. As shown in this diagram, the battery side voltage _V1 is subjected to feedback control in a subtractor 111, filtering by "1/(Ls+R)" by a block 112, and processing according to the duty ratio D by a block 113. In addition, for I _ofs , the inverter side voltage _V2 is fed back and input to a subtractor 115 via a filter 114 with a characteristic of "-1/R ₀ ", and the inverter current I ₀ is calculated.

Then, in subtractor 116, the output of subtractor 115 is subtracted from the output of block 113, and the subtraction result is processed by block 117 having a characteristic of "1/Cs" to calculate inverter side voltage _V2 . Note that when inverter side voltage _V2 is fed back to subtractor 111, it is processed by block 118 according to duty ratio D.

In this system, the influence of _R0 seen from the inverter side voltage _V2 can be expressed as 1/ _R0 . Therefore, by using the following equation (8), which holds true for a typical converter 2, to convert equation (7) into the relationship between the inverter side voltage _V2 of converter 2 and the output power P handled by converter 2, and then rearranging so that 1/ _R0 becomes the left-hand side, equation (9) can be obtained.

By satisfying formula (9), the operation of the converter 2 can be stabilized. Therefore, by using formula (9), the influence of the negative resistance characteristic 1/R ₀ accompanying the constant power control of the inverter 3 is examined. As the output power P of the converter 2 increases, the negative resistance characteristic 1/R ₀ increases, and as the inverter side voltage V ₂ decreases, the negative resistance characteristic 1/R ₀ increases. Therefore, as the negative resistance characteristic 1/R ₀ increases, the responsiveness deteriorates, and therefore it is necessary to increase the first gain K _1. In this way, a relationship is obtained in which the first gain K ₁ increases as the negative resistance characteristic 1/R ₀ increases, as shown in FIG. 7.

7 using equation (9), when the inverter-side voltage _V2 is constant, as the output power P of the converter 2 increases, the negative resistance characteristic 1/ _R0 increases, and therefore the first gain _K1 is increased. Also, when the output power P of the converter 2 is constant, the smaller the inverter-side voltage _V2 is, the larger the negative resistance characteristic 1/ _R0 is, and therefore the first gain _K1 is increased. Furthermore, when the output power P of the converter 2 is small and the inverter-side voltage _V2 of the converter 2 is large, the negative resistance characteristic 1/ _R0 is small, and therefore the first gain _K1 is decreased. That is, the first gain _K1 is determined according to the output power P of the converter 2 and the inverter-side voltage _V2 .

In this manner, in step S3 of FIG. 3, the control gain calculation unit 66 determines the first gain _K1 in accordance with the negative resistance characteristic 1/ _R0 with reference to the graph of FIG. 7, and sets the first gain _K1 to be smaller as the negative resistance characteristic 1/ _R0 is smaller.

Next, a method for calculating the second gain _K2 will be described.

Fig. 8 is a graph showing the relationship between the negative resistance characteristic 1/ _R0 and the second gain _K2 . This graph is used to calculate the second gain _K2 in step S4 shown in Fig. 6. The second gain _K2 is a gain that can suppress vibrations caused by a phase delay in the converter 2.

In general, for the natural angular frequency ω ₀ of the converter 2, the following relationship is established based on the duty ratio D, the inductance L, and the capacitance C.

According to equation (10), the natural angular frequency _ω0 and the duty ratio D are in a proportional relationship. And because the duty ratio D and the step-up ratio D' are in an inverse relationship, the natural angular frequency _ω0 and the step-up ratio D' are inversely proportional to each other. Also, in the converter 2, as the angular frequency ω increases, a phase delay occurs in the inverter side voltage _V2 output from the converter 2, and the delay increases in a step manner with the natural angular frequency _ω0 as a threshold value.

In this embodiment, the second gain _K2 is determined so that the gain margin and phase margin determined by the characteristics of the converter 2, which is the controlled object, are equal to or less than predetermined values in order to stabilize the control system due to the phase delay occurring in the converter 2. For example, the second gain _K2 is determined so that the gain margin and phase margin are not equal to or less than 12 dB and 60°.

Figure 12 shows the relationship between the angular frequency ω and the gain and phase lag of the output power relative to the input voltage in converter 2, depending on the step-up ratio D' of converter 2. Figure 12(A) shows the relationship between the angular frequency ω and the gain of the output voltage relative to the input voltage, and Figure 12(B) shows the relationship between the angular frequency ω and the phase lag of the output voltage generated by converter 2 relative to the input voltage. Note that in Figure 12(B), the downward direction on the y-axis indicates a phase lag. Also, each figure shows three types of relationships, A to C, depending on the magnitude of the step-up ratio D'.

As shown in Fig. 12A, the output power of the converter 2 is maximum when the angular frequency ω is the natural angular frequency _ω0 . The natural angular frequency _ω0 at which the gain is maximum increases as the step-up ratio D' decreases. That is, the natural angular frequency _ω0 increases in the following order: natural angular frequency _ω0A in state A where the step-up ratio D' is large, natural angular frequency _ω0B in state B where the step-up ratio D' is medium, and natural angular frequency _ω0C in state C where the step-up ratio D' is small. Furthermore, referring to Fig. 12B, the phase lag in the converter 2 increases with the natural angular frequency _ω0 as the threshold value. Also, overall, the phase lag increases as the angular frequency ω increases.

In state A where the step-up ratio D' is large, the natural angular frequency _ω0A at which the phase lag increases stepwise is small, and therefore a phase lag is likely to occur. If the second gain _K2 is increased when a phase lag occurs, the control system may vibrate, and therefore the second gain _K2 cannot be increased. Therefore, the larger the step-up ratio D', the smaller the second gain _K2 must be.

Here, in order to conform to the calculation method of the first gain _K1 , the relationship between the step-up ratio _D ' and the second gain K2 is changed to the relationship between the negative resistance characteristic 1/ _R0 and the second gain _K2 . The negative resistance characteristic 1/ _R0 can be expressed as follows using the step-up ratio D'. That is, the larger the step-up ratio D' is, the smaller the negative resistance characteristic 1/ _R0 is.

Therefore, the smaller the negative resistance characteristic 1/ _R0 , the larger the step-up ratio D', and therefore the second gain _K2 is set to be smaller. Therefore, a relationship is required in which the smaller the negative resistance characteristic 1/ _R0 , the smaller the first gain _K1 , as shown in FIG.

In this way, by calculating the second gain _K2 according to the negative resistance characteristic 1/ _R0 in the same way as the calculation of the first gain _K1 , it becomes easy to calculate the gain K taking into consideration the first gain _K1 and the second gain _K2 in the process of step S5 in Fig. 3. Therefore, the gain K can suppress instability caused by the negative resistance characteristic obtained according to the negative resistance characteristic 1/ _R0 , and can also suppress instability caused by a phase delay in the inverter-side voltage _V2 output from the converter 2 that increases with an increase in the step-up ratio D'.

Here, a comparison will be made between the inverter side voltage _V2 from the converter 2 in this embodiment and a comparative example using Figures 13 and 14. Figure 13 shows this embodiment, and Figure 14 shows a comparative example. Note that in both examples, the inverter 3 is assumed to be under constant power control.

In FIG. 13, a time chart of the inverter side voltage _V2 is shown at the top, and a time chart of the output power P is shown at the bottom.

First, in a low power state where the output power P is small, the motor generator 4 operates at a constant low torque and maintains a constant rotation speed. In this low power state, the inverter side voltage _V2 is constant.

Then, with the inverter 3 under constant power control, the output power P changes significantly from low power to high power at a predetermined time _t1 . The gain K is set based on the first gain _K1 taking into account the negative resistance characteristic and the second gain _K2 taking into account the step-up ratio D', so that both the responsiveness and stability of the control system are ensured and the inverter side voltage _V2 does not oscillate.

In contrast, in the feedback control in the comparative example, the feedback control gain of the converter 2 is set in consideration of the characteristics of the battery internal resistance due to the change in battery temperature and the change in coil inductance due to the passing current. In this case, when the output power P handled by the converter 2 is small, the converter 2 has sufficient responsiveness to the negative resistance characteristic 1/ _R0 associated with the constant power control of the inverter 3, and the inverter side voltage _V2 is stable. However, when the output power P becomes large at time _t1 , the influence of the negative resistance characteristic 1/ _R0 becomes large, and the inverter side voltage _V2 oscillates.

Thus, in this embodiment, the gain K used to control the converter 2 during powering operation of the motor is determined in consideration of the first gain _K1 that suppresses instability caused by the negative resistance characteristic 1/ _R0 determined by the output power P and the inverter side voltage _V2 , and the second gain K2 that suppresses instability caused by a phase delay occurring according to the step-up ratio D' of the converter _2. By performing PI control using such gain K, it is possible to achieve both responsiveness and stability of the converter 2.

This embodiment provides the following advantages:

According to the control method for converter 2 of this embodiment, when motor generator 4 is powered by output power P from constant voltage controlled inverter 3, gain K is determined according to output power P from converter 2 to inverter 3, inverter side voltage _V2 which is the output voltage, and boost ratio D' of converter 2. Then, a gate signal for converter 2 is calculated by feedback control using gain K in voltage control unit 61, and the output of converter 2 is controlled.

When the inverter 3 is subjected to constant power control, a negative resistance characteristic 1/ _R0 occurs, and the larger the negative resistance characteristic 1/ _R0 , the worse the responsiveness becomes. Therefore, as shown in Fig. 7, the larger the negative resistance characteristic 1/ _R0 , the larger the first gain _K1 needs to be. Furthermore, since the calculation of the first gain _K1 is premised on the condition of equation (9), the first gain _K1 is determined according to the output power P from the converter 2 to the inverter 3 and the inverter side voltage _V2 .

12, the larger the step-up ratio D', the smaller the natural angular frequency _ω0 of the converter 2 becomes, and a phase delay becomes more likely to occur, so the second gain _K2 cannot be increased. Therefore, the second gain _K2 can be determined according to the step-up ratio D'.

Finally, the gain K is determined by the first gain _K1 and the second gain _K2 . Therefore, the gain K is determined according to the output power P used in the calculation of the first gain _K1 , the inverter side voltage _V2 serving as the output terminal, and the step-up ratio D' used in the calculation of the second gain _K2 . By going through such a calculation process, the gain K takes into consideration the deterioration of responsiveness due to the negative resistance characteristic 1/ _R0 and the delay due to the phase lag in the converter 2 according to the step-up ratio D', so that the output of the converter 2 can be controlled more stably.

According to the control method of the converter 2 of the present embodiment, since the first gain _K1 is calculated in accordance with the negative resistance characteristic 1/ _R0 , the second gain _K2 is also calculated in accordance with the negative resistance characteristic 1/ _R0. This makes it possible to obtain a gain K that can suppress instability caused by the negative resistance characteristic 1/ _R0 in the control of the converter 2 and also suppress instability caused by a phase delay that becomes larger as the step-up ratio D' increases.

According to the formula (11), the negative resistance characteristic 1/ _R0 is determined according to the output power P, the battery side voltage _V1 , and the step-up ratio D'. Therefore, by using the formula (11), the relationship between the step-up ratio D' and the second gain _K2 can be expressed as the relationship between the negative resistance characteristic 1/ _R0 and the second gain _K2 as shown in FIG.

In this way, by calculating both the first gain _K1 and the second gain _K2 according to the negative resistance characteristic 1/ _R0 , the gain K can be calculated using a graph such as that shown in Fig. 9. The gain K determined in this way can suppress instability caused by the negative resistance characteristic determined according to the negative resistance characteristic 1/ _R0 , and can also suppress instability caused by a phase delay that increases according to the step-up ratio D'.

Furthermore, according to the control method for converter 2 of this embodiment, the larger the output power P, the larger the negative resistance characteristic 1/ _R0 , and therefore the larger the first gain _K1 . Also, the smaller the inverter-side voltage V2 output from converter ₂ , the larger the negative resistance characteristic 1/ _R0 , and therefore the larger the first gain _K1 . Here, the larger the output power P and the lower the inverter-side voltage _V2 , the greater the possibility that equation (9) will not be satisfied and converter 2 will become unstable. Therefore, by increasing the first gain _K1 , the responsiveness and stability of converter 2 can be improved.

Furthermore, according to the control method for converter 2 of this embodiment, as shown in Fig. 12, the larger the step-up ratio D', the more likely a phase delay occurs in converter 2, so the second gain _K2 cannot be made large. Furthermore, according to formula (11), the larger the step-up ratio D', the smaller the negative resistance characteristic 1/ _R0 . Therefore, as shown in Fig. 8, the smaller the negative resistance characteristic 1/ _R0 , the larger the step-up ratio D', so the second gain _K2 is made small. In this way, the responsiveness and stability of converter 2 can be improved.

(Modification)
In the above embodiment, an example in which one motor generator 4 is provided in the motor system 100 has been described, but this is not limiting. In this modified example, an example in which two motor generators, a first motor generator 4A and a second motor generator 4B, are provided in the motor system 100 will be described.

Figure 15 is a diagram showing the configuration of a modified motor system 100. For example, the motor system 100 is used in hybrid vehicles. The motor system 100 of this modified example has a first inverter 3A and a first motor generator 4A used to drive the output shaft 5, similar to the motor system 100 shown in Figure 1.

The motor system 100 further includes a second inverter 3B and a second motor generator 4B. The second inverter 3B boosts the voltage supplied from the converter 2 and supplies it to the second motor generator 4B. The second motor generator 4B functions as a starter for the engine 7. A torque transmission device may be provided between the engine 7 and the output shaft 5, and the output torque of the engine 7 may be transmitted to the output shaft 5. The second motor generator 4B is provided with a resolver 41B. The engine 7 is provided with a resolver 71 that detects the rotation angle or rotation speed of the crankshaft.

The controller 6 is configured to be able to control the converter 2, the first inverter 3A, and the second inverter 3B, and receives the rotation speed of the first motor generator 4A from the resolver 41A, the rotation speed of the second motor generator 4B from the resolver 41B, and the rotation speed of the engine 7 from the resolver 71.

16 is a circuit diagram of the motor system 100. The controller 6 receives inputs of a torque command value T _a ^* for the first motor generator 4A and a torque command value T _b ^* for the second motor generator 4B calculated by a higher-level device, the rotation speeds N _a and N _b detected by the resolvers 41A and 41B, drive currents I _pa and I _pb detected based on current sensors 42A and 42B, and an inverter-side voltage V ₂ detected by a voltage sensor 28. The controller 6 controls the converter 2, the first inverter 3A, and the second inverter 3B based on these inputs.

If the required power for the first motor generator 4A is P _a and the required power for the second motor generator 4B is P _b , the output power P of the converter 2 is expressed as the sum of P _a and P _b .

For example, when the first motor generator 4A and the second motor generator 4B are each in a power running operation, P _a and P _b are both positive (P _a >0, P _b >0). When the first motor generator 4A and the second motor generator 4B are each in a regenerative operation, P _a and P _b are both negative (P _a <0, P _b <0).

The controller 6 performs limit control as shown in Fig. 5 on the output power P of the converter 2. In step S4, the controller 6 controls the first motor-generator 4A and the second motor-generator 4B so that the sum of P _a and P _b satisfies the criterion shown in the limit power P _lim .

For example, when the output power P is negative, either the first motor generator 4A or the second motor generator 4B may be operated in power running mode so that the output power P, which is the sum of _Pa and _Pb, does not fall below the input limit power _Pin .

Although the embodiments of the present invention have been described above, the above embodiments merely show some of the application examples of the present invention, and the technical scope of the present invention is not intended to be limited to the specific configurations of the above embodiments. Furthermore, the above embodiments can be combined as appropriate.

Claims (2)

A method for controlling a converter that boosts a DC input voltage supplied from a battery and supplies power to a motor via an inverter that is constant voltage controlled, comprising the steps of:
When the motor is in a power running operation, a gain is determined according to an output power from the converter to the inverter, an output voltage, and a step-up ratio of the converter;
A method for controlling a converter, comprising: controlling an output voltage of the converter by feedback control using the gain. A control device for a converter that boosts a DC input voltage supplied from a battery and supplies power to a motor via an inverter that is constant voltage controlled, comprising:
The control device includes:
When the motor is in a power running operation, a gain is determined according to an output power and an output voltage from the converter to the inverter, and a step-up ratio of the converter;
A converter control device that controls an output voltage of the converter by feedback control using the gain.

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