JP7535782B2 - Control method for grid-connected inverter system - Google Patents

Description

The present invention relates to a method for controlling a grid-connected inverter system, and in particular to a method for controlling an inverter constituting the system using a control system including a feedback control system, a feedforward control system, and a droop control system.

With the recent increase in demand for sustainable energy sources such as solar and wind power, grid-connected inverter systems (hereinafter simply referred to as "systems") have been attracting attention. A grid-connected inverter system includes a grid, an inverter that operates as a power supply source for the grid, and a control system that controls the inverter. The inventors have previously proposed a control system for a grid-connected inverter system that includes a feedback control system and a feedforward control system (see, for example, Patent Document 1). This control system can ensure the stability of the entire system.

In addition, in order to solve the problem of synchronization between the inverter output voltage and the grid voltage in a grid-connected inverter system, the use of a control system including droop control is also being considered. In this case, the control system generally processes sampled current and/or voltage values by dq conversion (see, for example, Non-Patent Document 1).

JP 2019-193494 A

MICROGRID DYNAMICS and CONTROL, John Wiley & Sons, Inc., 2017, pp. 233-243

However, conventional control that combines dq transformation and droop control is complex and therefore difficult to use.

The present invention was made in consideration of these circumstances, and aims to provide a method for controlling a grid-connected inverter system using a control system that is simpler than conventional methods.

In order to solve the above problems, a control method according to the present invention is a method for controlling an inverter constituting a three-phase grid-connected inverter system in an αβ coordinate system using a control system including a feedback control system, a feedforward control system, and a droop control system, and includes: (1) a control system design step of designing a control system; and (2) an inverter control step of controlling the inverter using the control system designed in the control system design step. The control system design step includes: (1A) a first step of designing a sine wave compensator constituting the feedback control system; (1B) a second step of determining a frequency characteristic of an output impedance of the inverter when controlled by a control system including only the feedback control system designed in the first step; (1C) a third step of designing a transfer function of a feedforward control system such that the inverter is passive based on the frequency characteristic determined in the second step; and (1D) a fourth step of designing a droop control system while taking into consideration the response characteristic of the inverter. The inverter control step includes: (2A) determining a frequency characteristic of three-phase currents _iu , _iv , _iw and voltages _vu , _vv , v (2B) a sixth step of determining _a current iαβ and a voltage vαβ in the αβ coordinate system by αβ transformation based on the instantaneous values of the currents _iu , _iv , _iw and voltages _vu , _vv , _vw obtained in the fifth step; (2C) a seventh step of determining a voltage command _{value vrαβ} in the αβ coordinate system for a feedback control system by a droop control system based on _the current _iαβ and voltage _vαβ determined in the sixth step; (2D) an eighth step of determining a control input uαβ in the αβ coordinate system based on the current _iαβ and voltage _vαβ determined in the sixth step and the voltage command value _vrαβ determined in the seventh step ; and (2E) a ninth step of determining _three -phase control inputs _uu , _uv , _uw by inverse αβ transformation based on the control input _uαβ determined in the eighth step.

The above control method may include, in the first step, designing a sine wave compensator using an optimal control method.

In the third step, the control method may be configured such that the transfer function is constructed as a product of a main characteristic part H^ of second order or less and a filter characteristic part F of second order or less that limits the band of the signal transmitted by the main characteristic part H^.

The above control method may also be configured to design the main characteristic part H^ and the filter characteristic part F by fitting in the third step.

The present invention provides a method for controlling a grid-connected inverter system using a control system that is simpler than conventional methods.

1 is a circuit diagram of a grid-connected inverter system to which a control method according to the present invention is applied; FIG. 2 is an equivalent circuit diagram of the system shown in FIG. 1 . FIG. 1 is a block diagram of a control system designed by a control method according to the present invention. FIG. 2 is an equivalent circuit diagram of the system shown in FIG. 1 . FIG. 4 is a flow diagram of a control method according to the present invention. FIG. 11 is a diagram showing the seventh and eighth steps of the control method according to the present invention. FIG. 11 is a waveform diagram showing the frequency characteristics of the output impedance obtained in the second step. FIG. 11 is a waveform diagram showing the frequency characteristics of the output impedance after the third process is executed. FIG. 11 is a waveform diagram showing the results of a first control example according to the present invention. FIG. 11 is a waveform diagram showing the results of a second control example according to the present invention.

Below, we will explain the underlying technologies and concepts that form the basis of the control method of the present invention, with reference to the attached drawings, and then we will explain the overall picture and details of the control method of the present invention.

(Configuration of grid-connected inverter system)
Fig. 1 shows a grid-connected inverter system 10 to which a control method according to the present invention is applied. As shown in the figure, the system 10 includes a three-phase inverter 20 that converts a DC voltage E (hereinafter referred to as "input voltage") output by a photovoltaic power generation device or the like into an AC voltage, a three-phase grid 30, and a filter 40 provided between them.

The inverter 20 includes a u-phase leg 21, a v-phase leg 22, and a w-phase leg 23. The filter 40 includes a filter inductor _Lfu and a filter capacitor _Cfu constituting a u-phase LC filter, a filter inductor _Lfv and a filter capacitor _Cfv constituting a v-phase LC filter, and a filter inductor _Lfw and a filter capacitor _Cfw constituting a w-phase LC filter. The inductors _Lfu , _Lfv , and _Lfw each have an inductance of _Lf , and the capacitors _Cfu , _Cfv , and _Cfw each have a capacitance of _Cf.

Note that Z _su , Z _sv , and Z _sw present between the filter 40 and the grid 30 represent impedance components resulting from wiring and the like, and include the grid inductance _Ls . Furthermore, in this specification, i _ou , i _ov , and i _ow are referred to as "output currents of the inverter" or simply "output currents."

この三相の変数ベクトルａは、次式によりαβ変換することができる。

また、αβ座標系での複素ベクトルａ_αβは、次式の複素記法により表記することができる。

(Complex notation of three-phase state variables)
Any three-phase variable vector a, such as voltages, currents and control variables, can be defined using the transpose symbol T as follows:

This three-phase variable vector a can be αβ transformed by the following equation.

Moreover, a complex vector _aαβ in the αβ coordinate system can be expressed by the following complex notation.

When the α and β components of impedance Z(s) are Z _αα (s) and Z _αβ (s), respectively, and the α and β components of current i _αβ are i _α and i _β , respectively, when current i _αβ flows through impedance Z(s), the voltage v _αβ across impedance Z(s) is given by the following equation.

ただし、式（５）は、α，β成分が対称であることを前提としている。実際には、後述するドループ制御を制御系に含めるとα，β成分は非対称になるので、式（５）ではなくより一般的な式（６）または式（７）を用いることになる。

ただし、

ここで、Ｚ^ｍ _αβ（ｓ）はインピーダンス行列であり、その固有値は、Ｚ_αα（ｓ）＋ｊＺ_αβ（ｓ），Ｚ_αα（ｓ）－ｊＺ_αβ（ｓ）である。前者は正相インピーダンスに対応し、後者は逆相インピーダンスに対応している。 The α and β components v _α and v _β of the voltage v _αβ across both ends can be expressed in a matrix format as shown in the following equation.

However, formula (5) is based on the premise that the α and β components are symmetrical. In reality, when droop control, which will be described later, is included in the control system, the α and β components become asymmetrical, so that the more general formula (6) or (7) is used instead of formula (5).

however,

Here, Z ^m _αβ (s) is an impedance matrix whose eigenvalues are Z _αα (s) + jZ _αβ (s) and Z _αα (s) - jZ _αβ (s), the former corresponding to the positive-sequence impedance and the latter corresponding to the negative-sequence impedance.

ただし、ｙ_ｊは制御出力である。また、状態変数ｘ_ｊ、連続系システム係数行列Ａ_ｃ、連続系入力係数ベクトルｂ_ｃ、連続系外乱係数ベクトルｈ_ｃおよび連続系出力係数ベクトルｃ_ｃは、次式の通りである。

(Discretized state equation of grid-connected inverter system)
If the current (hereinafter referred to as the "inductor current") flowing through the filter inductor _Lfj of each phase (where j = u, v, w; the same applies below) is _iLj , the voltage across the filter capacitor _Cfj of each phase (hereinafter referred to as the "capacitor voltage") is _vCj , the control input of each phase is _uj , and the output current of each phase is _ioj , the three-phase continuous system state equation for _iLj and _Cfj at time t is as follows:

Here, y _j is the control output. Furthermore, the state variable x _j , the continuous system coefficient matrix A _c , the continuous system input coefficient vector b _c , the continuous system disturbance coefficient vector h _c and the continuous system output coefficient vector c _c are as follows.

ただし、αβ座標系における状態変数ｘ_αβは、次式の通りである。

Moreover, the continuous equation of state in the αβ coordinate system can be expressed as follows:

However, the state variable _xαβ in the αβ coordinate system is expressed by the following equation.

Figure 2 (A) is an equivalent circuit diagram of the grid-connected inverter system 10 corresponding to equation (9), and figure 2 (B) is an equivalent circuit diagram of the grid-connected inverter system 10 corresponding to equation (11).

ただし、離散系システム係数行列Ａ、離散系入力係数ベクトルｂ、離散系外乱係数ベクトルｈおよび離散系出力係数ベクトルｃは、次式の通りである。

When the continuous state equation in the αβ coordinate system expressed by equation (11) is discretized with a sampling period T, equation (13), which is a discrete state equation in the αβ coordinate system, is obtained, where time t=iT.

Here, the discrete system coefficient matrix A, the discrete system input coefficient vector b, the discrete system disturbance coefficient vector h, and the discrete system output coefficient vector c are as follows:

(Overview of the control system)
FIG. 3 shows a control system for controlling the grid-connected inverter system 10. As shown in the figure, the control system is composed of a main circuit section and a digital control section. Feedback control, feedforward control, and droop control are realized by this digital control section. In addition, G _s [z] and k _fαβ in the figure are the transfer function and state feedback gain (also called "control gain") of a sine wave compensator constituting the feedback control system, respectively. Also, H[z] is the transfer function of the feedforward control.

(Feedback control system design)
As described above, in the present invention, the feedback control system is configured with a sine wave compensator. If the control gain, fundamental angular frequency, and damping coefficient of the sine wave compensator are _krαβ , _ω0 (= _2πf0 , _f0 is 50/60 [Hz]), and _k0 , respectively, and the reference sine waveform (also called the "voltage command value") is _vCrαβ [i], the auxiliary variable _wαβ [i] and control input _uαβ [i] of the sine wave compensator are expressed by the following equations.

ただし、拡大状態変数ｘ＾_αβ［ｉ］は、次式の通りである。

また、離散系システム係数行列Ａ＾、離散系入力係数ベクトルｂ＾、離散系外乱係数ベクトルｈ＾および基準値係数ベクトルｇ＾は、次式の通りである。

In this case, the discrete augmented state equation in the αβ coordinate system, including the state variable x _αβ [i] and the auxiliary variable w _αβ [i], is as follows:

Here, the extended state variable x^ _αβ [i] is expressed by the following equation.

Moreover, the discrete system coefficient matrix A^, the discrete system input coefficient vector b^, the discrete system disturbance coefficient vector h^, and the reference value coefficient vector g^ are as follows:

ただし、Ｑは拡大状態変数ｘ＾_αβ［ｉ］の評価荷重であり、ｒは制御入力ｕ_αβ［ｉ］の評価荷重である。また、＊は共役転置記号である。 The two control gains (also called "complex gains") _kfαβ and _krαβ of the sine wave compensator can be determined by an optimal control method that minimizes the evaluation function expressed by equation (20).

where Q is the evaluation weight of the augmented state variable x^ _αβ [i], r is the evaluation weight of the control input _uαβ [i], and * is the conjugate transpose symbol.

By using a control system including the feedback control system designed as above, it is possible to make the capacitor voltages v _Cu , v _Cv , and v _Cw of each phase follow a sine wave.

ただし、１は単位行列である。また、系統アドミタンスＹ^ｍ _ｓαβ（ｓ）は、

(System stability)
4, in the frequency domain, the inverter 20 side of the grid-connected inverter system 10 can be expressed by a Thevenin circuit with _an equivalent voltage source ^VvCαβ (s) and an output impedance ^Zm _oαβ (s), and the grid 30 side can be expressed by a grid _voltage ^Vvsαβ (s) and a grid impedance ^Zm _sαβ (s). At this time, the output current ^Iv _oαβ (s) can be expressed by the following equation.

Here, 1 is a unit matrix. Furthermore, the system admittance Y ^m _sαβ (s) is expressed as follows:

By applying the Nyquist stability criterion to the first term on the right-hand side of equation (21), it is possible to derive a stabilization condition for system 10, which is that "since system admittance Y ^m _sαβ (s) is passive, it is sufficient that the phase of output impedance Z ^m _oαβ (s) falls within the range of -90° to +90°." In other words, if the real part of the eigenvalue eig[Z ^m _oαβ (s)] of output impedance Z ^m _oαβ (s) can be made positive, it is possible to realize inverter 20 having passivity, thereby stabilizing the entire system 10. In this case, the following equation is established.

前述した通り、Ｈはフィードフォワード制御の伝達関数である。
なお、式（２４）は、摂動記号^～を用いた周波数領域における状態方程式（式（２５））から式（２６）を導出し、さらにこの式（２６）において等価電圧源Ｖ^～ｖ _Ｃαβ［ｚ］＝０と近似することにより導出したものである。

Here, the output impedance Z ^m _αβ (s) can be calculated by the following equation.

As mentioned above, H is the transfer function of the feedforward control.
Note that equation (24) is derived by deriving equation (26) from the state equation (equation (25)) in the frequency domain using perturbation symbols ^∼ , and further approximating this equation (26) by setting the equivalent voltage source V ^∼v _Cαβ [z] = 0.

As will be described in detail later, in the present invention, at the stage where the design of a feedback control system with excellent tracking performance by the above-mentioned method is completed, the frequency characteristic of the output _{impedance Z moαβ} ⁽ s) calculated by the formula (24) is calculated by frequency simulation to judge whether the system 10 is stable or not (see the formula (23)). If the system is not stable, a feedforward control system (particularly, the transfer function H[z]) is designed by the method described later.

(Feedforward control system design)
In the present invention, the transfer function H[z] of the feedforward control system is composed of the product of a main characteristic part H^[z] and a filter characteristic part F[z] that limits the band of the signal transmitted by the main characteristic part H^[z], as shown in equation (27).

当然ながら、主特性部Ｈ^［ｚ］は、次式のような１次の形式を有していてもよい。

For ease of calculation, the main characteristic part H^[z] is preferably of second order or less, and has, for example, the following form: where a ₀ , a ₁ , a ₂ , b ₁ , and b ₂ are coefficients determined by fitting.

Of course, the main characteristic Ĥ[z] may have a linear form such as:

当然ながら、フィルタ特性部Ｆ［ｚ］は、１次ローパスフィルタＦ_Ｌ［ｚ］と１次ハイパスフィルタＦ_Ｈ［ｚ］との積で表される２次バンドパスフィルタＦ_Ｌ［ｚ］・Ｆ_Ｈ［ｚ］であってもよい。 For the same reason, the filter characteristic F[z] is preferably of second order or less, for example a first order low pass filter or a first order high pass filter of the following form: where c ₀ , c ₁ , and d ₁ are coefficients determined by fitting.

Naturally, the filter characteristic portion F[z] may be a second-order band-pass filter F _L [z]·F _H [z] represented by the product of a first-order low-pass filter F _L [z] and a first-order high-pass filter F _H [z].

これに、遮断角周波数がω_ｃであるローパスフィルタを適用すると、複素測定電力ｓ_ｍ［ｉ］を求めることができる。すなわち、次式により複素測定電力ｓ_ｍ［ｉ］が求められる。なお、Ｇ_ｌは、ローパスフィルタの伝達特性である。

(Droop control system design)
Complex power s[i] at time t = iT can be expressed by the following equation using active power p[i], reactive power q[i], and complex conjugate symbol *.

By applying a low-pass filter with a cutoff angular frequency of _ωc to this, the complex measured power s _m [i] can be obtained. That is, the complex measured power s _m [i] can be obtained by the following equation, where G _l is the transfer characteristic of the low-pass filter.

ただし、ω_０は基本角周波数であり、これを初期値とする。また、ｍ_ｐはドループ制御の電力制御ゲインである。 In droop control, in order to make this complex measured power s _m [i] coincide with the power command value S _r (=active power command value P _r +j reactive power command value Q _r ), the angular frequency ω _αβ is controlled by the following equation.

where ω ₀ is the fundamental angular frequency and is set as the initial value, and m _p is the power control gain of the droop control.

ただし、ｖ_Ｃ０は初期基準電圧であり、系統電圧の振幅を用いる。 At this time, the complex phase angle θ _αβ [i] and the complex voltage command value v _Crαβ [i] by the droop control are respectively expressed by the following equations.

Here, v _C0 is an initial reference voltage, and the amplitude of the system voltage is used .

また、より正確には次式のように表される電力制御ゲインｍ^ｍ _ｐも、一般的にはｍ_ｐα≠ｍ_ｐβであるため、α，β成分が非対称である。

このため、ドループ制御を制御系に含めると、システム１０の制御は、α，β成分において非対称になる。 Here, in the conjugate calculation accompanying the droop control, the α and β components are asymmetric, as can be seen from the following equation.

More precisely, the power control gain ^m _p expressed by the following equation generally satisfies m _pα ≠ m _pβ , and therefore the α and β components are asymmetric.

Therefore, when droop control is included in the control system, the control of the system 10 becomes asymmetric in the α and β components.

(Control method according to the present invention)
As shown in Fig. 5, the control method according to the present invention includes a control system design step of designing a control system including a feedback control system, a feedforward control system, and a droop control system, and an inverter control step of controlling the inverter 20 using the designed control system. The control system design step includes steps S1, S2, S3, and S4 that are executed in sequence, and the inverter control step includes steps S5, S6, S7, S8, S9, and S10 that are executed in sequence. The inverter control step is repeatedly executed at a predetermined switch period while the inverter 20 is operating.

In the first step S1, a sine wave compensator constituting a feedback control system is designed by the above method. More specifically, in the first step S1, two control gains _kfαβ and _krαβ of the sine wave compensator are determined by the optimal control method (see equation (20)). The damping coefficient _k0 can be arbitrarily determined by the user when starting the design.

In a second step S2, the frequency characteristic of the output impedance Z ^moαβ (s) of the inverter 20 is obtained when the inverter 20 is controlled by ^a control system including only the feedback control system designed in the first step S1. The frequency characteristic of _the output impedance Z _moαβ (s) can be obtained by using equation (24) and frequency simulation.

In the third step S3, a transfer function H[z] of the feedforward control system is designed so that the inverter 20 is passive, based on the frequency characteristics obtained in the second step S2. The user can determine in advance what form the main characteristic part H^[z] and the filter characteristic part F[z] will take (see equations (28), (29), and (30)). In addition, the user can determine whether the filter characteristic part F[z] will be a low-pass filter, a high-pass filter, or a band-pass filter, while looking at the frequency characteristics obtained in the second step S2.

In a fourth step S4, a droop control system is designed while taking into consideration the response characteristics of the inverter 20. For example, in the fourth step S4, a change in the output power of the inverter 20 when the power command values ( _Pr , _Qr ) are changed is obtained by a time simulation, and the user determines power control gains m _pα and m _pβ that make this ideal.

In the fifth step S5, instantaneous values of three-phase currents _iu , _iv , _iw and voltages _vu , _vv , _vw are obtained. More specifically, in the fifth step S5, instantaneous values of inductor currents _iLu , _iLv , _iLw and output currents _iou , _iov , _iow are obtained as the currents _iu , _iv , _iw , and instantaneous values of capacitor voltages _vCu , _vCv , _vCw are obtained as the voltages _vu , _vv , _vw . Note that obtaining instantaneous values of three-phase currents _iu , _iv , _iw includes not only simultaneously sampling all of the currents _iu , _iv , _iw using a current sensor, but also simultaneously sampling two of the currents _iu , _iv , _iw using a current sensor and calculating the remaining one. The same applies to obtaining the voltages _vu , _vv , and _vw .

In a sixth step S6, the current iαβ and voltage vαβ in the αβ coordinate system are obtained by _αβ conversion based on the instantaneous values of the three-phase currents _iu , _iv , and _iw and voltages _vu , _vv , and _vw acquired in the fifth step S5. More specifically, in the sixth step S6, the inductor currents _iLu , _iLv , and _iLw are converted into currents _iLα and _iLβ , the output currents _iou , _iov , and _iow are converted into currents _ioα and _ioβ , and the capacitor voltages _vCu , _vCv , and _vCw are converted into voltages _vCα and _vCβ using the formula ₍ 2). As can be seen from equation (4), if the currents _iLα , _iLβ , _ioα , and _ioβ and the voltages _vCα and _vCβ are determined, the currents _iLαβ , _ioαβ and voltages _vCαβ can also be determined.

In the seventh step S7, a voltage command value _vrαβ in the αβ coordinate system for the feedback control system is calculated by droop control based on the current _iαβ and voltage _vαβ calculated in the sixth step S6. More specifically, in the seventh step S7, a complex phase angle _θαβ is calculated by applying the current _ioαβ and voltage _vCαβ calculated in the sixth step S6 to equations (31) to (34), and then a voltage command value _vCrαβ is calculated by equation (35).

In an eighth step S8, a control input _uαβ in the αβ coordinate system is calculated based on the current _iαβ and voltage _vαβ calculated in the sixth step S6 and the voltage command value _vrαβ calculated in the seventh step S7. More specifically, in the eighth step S8, the control input uαβ in the αβ coordinate system is calculated by applying the current _iLαβ and voltage _vCαβ calculated in the sixth step S6 and the voltage command value _vCrαβ calculated in the seventh step S7 to equation ₍₁₆₎ .

When the seventh step S7 and the eighth step S8 are organized, they look like Figure 6.

In a ninth step S9, three-phase control inputs _uu , _uv , and _uw are calculated by inverse αβ transformation based on the control input _uαβ calculated in the eighth step S8. The inverse αβ transformation can be performed by the following equation, which is a modification of equation (2).

In a tenth step S10, the phase legs 21, 22, and 23 of the inverter 20 are controlled based on the three-phase control inputs _uu , _uv , and _uw obtained in the ninth step S9. More specifically, in this tenth step S10, the ON duty of the upper arm constituting the u-phase leg 21 (= the OFF duty of the lower arm) is controlled based on the u-phase control input _uu . The same is true for the v-phase leg 22 and the w-phase leg 23.

In this way, in the control method according to the present invention, the acquired three-phase current and/or voltage values are converted into complex values in the αβ coordinate system for processing. Therefore, according to the present invention, it is possible to concisely describe the control system.

(Control system design example)
The circuit parameters of the grid-connected inverter system 10 were set as shown in Table 1, and the parameters related to the feedback control system (sine wave compensator) were set as shown in Table 2, and the first step S1 was executed. As a result, the two control gains _kfαβ and _krαβ of the sine wave compensator were determined as shown in Table 3.

When control was performed using only this feedback control system, the frequency characteristic of the output impedance Z ^m _oαβ (hereinafter simply referred to as "Z _o ") of the inverter 20 was as shown in FIG. 7. That is, when the second step S2 was performed, the frequency characteristic shown in FIG. 7 was obtained. As shown in FIG. 7(B), the phase of the output impedance Z _o was below -90° in a band of 2 kHz or more. In other words, the inverter 20 was not passive at this stage.

Therefore, by performing the third step S3, a feedforward control system was designed in which the inverter 20 is passive. The coefficients of the main characteristic part H^ (see equation (29)) and the filter characteristic part F (see equation (30)) of the transfer function H determined by fitting are as shown in Table 4.

After the design of the feedforward control system was completed, the frequency characteristics of the output impedance _Z0 were checked again, and the results shown in Figure 8 were obtained. As shown in Figure 8(B), the phase of the output impedance _Z0 was within the range of -90° to +90° in all bands where the simulation was performed. This indicates that the inverter 20 became passive.

Next, a droop control system is designed (fourth step S4). When the cutoff frequency f _c of the low-pass filter used in the droop control was set to 5 [Hz] and the power control gains m _pα and m _pβ were provisionally set to 0.0006, the system 10 became unstable, but when these were reset to 0.0005, the system 10 became stable. For this reason, in this control example, the power control gains m _pα and m _pβ were set to 0.0001 with a margin. However, this alone did not allow the output power of the inverter 20 to accurately follow the power command values ( _Pr , _Qr ), so only the power control gain m _pβ was adjusted, and finally, the power control gains m _pα and m _pβ were set as shown in Table 5.

(First control example according to the present invention)
When the inverter 20 was controlled by setting the power command values P _r and _Qr to 2000 [W] and 0 [VA] using the control system designed as above, the u-phase output current i _ou and the capacitor voltage v _Cu in a steady state were as shown in Fig. 9. It can be seen from Fig. 9 that the capacitor voltage v _Cu can follow the voltage command value v _Cr , and that the output current i _ou can be controlled so that the output power of the inverter 20 coincides with the power command values P _r and _Qr .

(Second control example according to the present invention)
Using the control system designed as above, when the active power command value P _r was changed from 2000 [W] to 3000 [W] to 2000 [W] to 1000 [W] to 2000 [W] while keeping the reactive power command value Q _r at 0 [VA], the active output power P _m and the reactive output power Q _m of the inverter 20 became as shown in Fig. 10. It can be seen from Fig. 10 that no excessive overshoot or ringing occurs in the active output power P _m , and that the transient change in the reactive output power Q _m accompanying the change in the active power command value P _r settles down quickly (in about 0.5 [s]).

Although one embodiment of the control method according to the present invention has been described above, it goes without saying that the configuration of the present invention is not limited to this and various modifications are possible.

10: Grid-connected inverter system 20: Inverter 21: U-phase leg 22: V-phase leg 23: W-phase leg 30: Grid 40: Filter

Claims (5)

A method for controlling an inverter constituting a three-phase grid-connected inverter system in an αβ coordinate system using a control system including a feedback control system, a feedforward control system, and a droop control system, comprising:
(1) a control system design process for designing the control system;
(2) an inverter control step of controlling the inverter using the control system designed in the control system design step;
The control system design process includes:
(1A) a first step of designing a sine wave compensator that constitutes the feedback control system;
(1B) a second step of determining a frequency characteristic of an output impedance of the inverter when the inverter is controlled by the control system including only the feedback control system designed in the first step;
(1C) a third step of designing a transfer function of the feedforward control system based on the frequency characteristic obtained in the second step so that the inverter is passive;
(1D) a fourth step of designing the droop control system while taking into consideration the response characteristics of the inverter;
The inverter control step includes:
(2A) a fifth step of acquiring instantaneous values of three-phase currents _iu , _iv , and _iw and voltages _vu , _vv , and _vw ;
(2B) a sixth step of determining the current _iαβ and the voltage _vαβ in the αβ coordinate system by αβ transformation based on the instantaneous values of the currents _iu , _iv , and _iw and the voltages _vu , _vv , and _vw acquired in the fifth step;
(2C) a seventh step of calculating a voltage command value _vrαβ in the αβ coordinate system for the feedback control system by the droop control system based on the current _iαβ and the voltage _vαβ calculated in the sixth step;
(2D) an eighth step of calculating a control input _uαβ in the αβ coordinate system based on the current _iαβ and voltage _vαβ calculated in the sixth step and the voltage command value _vrαβ calculated in the seventh step ;
(2E) a ninth step of determining three-phase control inputs u _u , u _v , and u _w by inverse αβ transformation based on the control input u _αβ determined in the eighth step. 2. The method of claim 1, wherein in said first step, said sine wave compensator is designed by an optimal control method. The method according to claim 1 or 2, characterized in that in the third step, the transfer function is constructed as a product of a main characteristic part H^ of second order or less and a filter characteristic part F of second order or less that limits the band of the signal transmitted by the main characteristic part H^. 4. The method according to claim 3, wherein in the third step, the main characteristic part H^ and the filter characteristic part F are designed by fitting. A method for controlling an inverter constituting a three-phase grid-connected inverter system in an αβ coordinate system using a control system including a feedback control system, a feedforward control system, and a droop control system, comprising:
(A) Three-phase current i _u , i _v , i _W and voltage v _u , v _v , v _W A first step of acquiring an instantaneous value of
(B) The current i obtained in the first step _u , i _v , i _W and voltage v _u , v _v , v _W Based on the instantaneous value of _αβ and voltage v _αβ A second step of calculating by αβ conversion;
(C) The current i obtained in the second step _αβ and voltage v _αβ Based on this, the droop control system calculates a voltage command value v in the αβ coordinate system for the feedback control system. _rαβ A third step of determining
(D) The current i obtained in the second step _αβ and voltage v _αβ , and the voltage command value v obtained in the third step _rαβ Based on this, the control input u _αβ A fourth step of determining
(E) The control input u obtained in the fourth step _αβ Based on this, the three-phase control input u _u , u _v , u _W The fifth step is to obtain the inverse αβ transformation.
A method comprising:

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