JP7549213B2 - DC-DC Converter - Google Patents

Description

The present invention relates to a DC-DC converter.

A dual bridge DC-DC converter is known that is composed of a first bridge circuit composed of two legs, a transformer, and a second bridge circuit composed of two legs. Patent Document 1 discloses a DC-DC converter that achieves ZVS (zero voltage switching) in switching elements by performing specific control.

JP 2020-5330 A

However, the conventional technology described above had the problem that ZVS could not be achieved in the low output range of the DC-DC converter.

One aspect of the present invention was made in consideration of the above problems, and aims to realize a DC-DC converter that enables ZVS of switching elements even in the low output range.

In order to solve the above problems, a DC-DC converter according to one aspect of the present invention includes a primary bridge circuit consisting of a primary first leg and a primary second leg, each of which has a pair of primary switching elements and a capacitor element connected in parallel to each of the primary switching elements, a secondary bridge circuit consisting of a secondary first leg and a secondary second leg, each of which has a pair of secondary switching elements and a capacitor element connected in parallel to each of the secondary switching elements, a conversion unit having a transformer and connected between the primary bridge circuit and the secondary bridge circuit, and a conversion circuit having a conversion circuit for converting the primary switching elements and the secondary switching elements into a conversion circuit for converting ... and a control unit that controls the switching of the primary side switching elements, the control unit sets the phase difference between the legs of the primary side bridge circuit to 0, controls each of the primary side switching elements to switch every half cycle, and further controls the primary side switching elements of the primary side second leg to switch an even number of times within each half cycle, sets a phase difference between the legs of the secondary side bridge circuit, controls each of the secondary side switching elements to switch every half cycle, and further controls the secondary side switching elements of the secondary side second leg to switch an even number of times within each half cycle.

According to one aspect of the present invention, it is possible to realize a DC-DC converter that can transmit power with low loss when the DC-DC converter is at low output.

1 is a circuit diagram showing a DC-DC converter according to a first embodiment. 4 is a timing chart showing the operation of the DC-DC converter according to the first embodiment. 4 is an enlarged timing chart of the DC-DC converter according to the first embodiment showing a transformer primary voltage, a transformer secondary voltage, a transformer primary current, a transformer secondary current, and an output power. FIG. FIG. 2 is a diagram showing a current path in an equivalent circuit of a DC-DC converter at a certain time. FIG. 2 is a diagram showing a current path in an equivalent circuit of a DC-DC converter at a certain time. FIG. 2 is a diagram showing a current path in an equivalent circuit of a DC-DC converter at a certain time. FIG. 2 is a block diagram showing a configuration of a control unit. 4 is a timing chart of the DC-DC converter at low output.

[Embodiment 1]
Hereinafter, an embodiment of the present invention will be described in detail with reference to FIGS.

(Configuration of DC-DC Converter)
1 is a circuit diagram showing a DC-DC converter 1 according to a first embodiment. The DC-DC converter 1 includes a primary bridge circuit 10, a secondary bridge circuit 20, a conversion unit 30, and a control unit 40. The primary bridge circuit 10 is connected to a DC power source E1. The secondary bridge circuit 20 is connected to a DC power source E2. The output voltage of the DC power source E1 is a primary voltage V1. A primary current flowing through the DC power source E1 is I1. The output voltage of the DC power source E2 is a secondary voltage V2. A secondary current flowing through the DC power source E2 is I2. Here, the primary voltage V1, the primary current I1, the secondary voltage V2, and the secondary current I2 each represent a time average value.

The primary bridge circuit 10 is a full bridge circuit with four switching elements, and a capacitor element C1 is connected in parallel to it. The primary bridge circuit 10 is composed of a primary first leg 11, a primary second leg 12, and a capacitor element C1. The primary first leg is a series connection of switching element S1 and switching element S2. The primary second leg is a series connection of switching element S3 and switching element S4.

The secondary bridge circuit 20 is a full bridge circuit provided with four switching elements, to which a capacitor element C2 is connected in parallel. The secondary bridge circuit 20 is composed of a first secondary leg 21, a second secondary leg 22, and a capacitor element C2. The first secondary leg is formed by connecting switching elements S5 and S6 in series. The second secondary leg is formed by connecting switching elements S7 and S8 in series.

Freewheeling diodes D1 to D8 are connected in parallel to the switching elements S1 to S8, respectively. In addition, snubber capacitors Csnub1 to Csnub8 are connected in parallel to the switching elements S1 to S8, respectively.

The conversion unit 30 includes a transformer Tr, a reactor L1, and a reactor L2. The reactor L1 is connected to the connection point between the switching element S1 and the switching element S2 and the primary winding of the transformer Tr. The reactor L2 is connected to the connection point between the switching element S3 and the switching element S4 and the primary winding of the transformer Tr. The inductance components represented by the reactor L1 and the reactor L2 include the leakage inductance of the transformer Tr. The secondary winding of the transformer Tr is connected to the connection point between the switching element S5 and the switching element S6 and the connection point between the switching element S7 and the switching element S8. Thus, in the circuit diagram of FIG. 1, the inductance components of the conversion unit 30 are equivalently represented as the reactor L1 and the reactor L2 provided on the primary side.

The control unit 40 controls the switching of the switching elements S1 to S8 by referring to the primary voltage V1, the primary current I1, the secondary voltage V2, and the secondary current I2.

(Basic operation of a DC-DC converter)
2 is a timing chart showing the operation of the DC-DC converter 1 according to the first embodiment. As is generally known, in order to prevent a shoot-through current that flows when two switching elements in each leg are simultaneously conductive, it is necessary to provide a dead time for issuing a gate signal that turns off both of these two switching elements when switching on and off. However, in the following explanation and in FIG. 2, the dead time is ignored for ease of understanding. However, when a switching element is turned on, the gate signal is controlled to be turned on with a timing delayed by the dead time.

The control unit 40 generates the switching timing from the command value vref and the first triangular wave vtr1. The first triangular wave vtr1 is a triangular wave with a period of 2π and an amplitude of 1, and is shown by a dotted line in FIG. 2. The command value vref passes through the median value of the first triangular wave vtr1.

The switching timing of the primary bridge circuit 10 is generated based on the magnitude relationship between the command value vref and the first triangular wave vtr1. That is, when the first triangular wave vtr1 is greater than the command value vref, the gate signal of the switching element S1 is turned off, and when the first triangular wave vtr1 is smaller than the command value vref, the gate signal of the switching element S1 is turned on. As a result, since the command value vref is the median value of the first triangular wave vtr1, switching occurs every half cycle π. The gate signal of the switching element S2 is a signal that reverses the on/off of the gate signal of the switching element S1.

Here, we define a second triangular wave vtr2 that is in opposite phase to the first triangular wave vtr1. The second triangular wave vtr2 is a triangular wave that has a different amplitude from the first triangular wave vtr1, and is shown by a dotted line in Figure 2.

The second triangular wave vtr2 and the gate signal of switching element S1 are compared, and the gate signal of switching element S3 is generated based on the magnitude relationship. In other words, when the second triangular wave vtr2 is greater than the gate signal of switching element S1, the gate signal of switching element S3 is turned on, and when the second triangular wave vtr2 is smaller than the gate signal of switching element S1, the gate signal of switching element S3 is turned off. The gate signal of switching element S4 is a signal that reverses the on/off of the gate signal of switching element S3.

Therefore, in the primary side bridge circuit 10, no phase difference is provided between the primary side first leg 11 and the primary side second leg 12. Furthermore, the primary side first leg 11 switches at time t0 and time t7, and the primary side second leg 12 switches at time t0, time t3, time t5, and time t7. The primary side second leg switches three times as often as the primary side first leg.

Next, the switching timing of the secondary first leg 21 is delayed by the inter-bridge phase difference φB from the switching timing of the primary first leg 11. In other words, the switching timing of the secondary first leg 21 is time t1, which is delayed from time t0 by the inter-bridge phase difference φB.

The switching timing of the secondary side first leg 21 and the secondary side second leg 22 is set with a time difference of the inter-leg phase difference φL, and switching is performed in the secondary side second leg 22. Furthermore, for the secondary side second leg 22, in the same manner as in the primary side second leg 12, the magnitude relationship between the second triangular wave vtr2 and the gate signal of the switching element S1 is compared, and switching is performed at a timing delayed by the sum of the inter-bridge phase difference φB and the inter-leg phase difference φL.

In this way, in the secondary bridge circuit 20, an inter-leg phase difference φL is provided between the secondary first leg 21 and the secondary second leg 22. Furthermore, the secondary first leg 21 switches at time t1, and the secondary second leg 22 switches at times t2, t4, and t6. The secondary second leg switches three times as often as the secondary first leg.

Therefore, the phase difference between the legs of the primary bridge circuit is set to 0, and the primary switching elements S1 to S4 are controlled to switch every half cycle. Furthermore, the primary switching elements S3 and S4 of the second primary leg are controlled to switch twice in each half cycle with a first phase interval δ1. The first phase interval δ1 is a value that is indirectly determined by the amplitude of the second triangular wave vtr2.

A phase difference between the legs, φL, is provided between the legs of the secondary bridge circuit, and the secondary switching elements S5 to S8 are controlled to switch every half cycle. Furthermore, the secondary switching elements S7 and S8 of the second secondary leg are controlled to switch twice at a second phase interval δ2 within each half cycle. The second phase interval δ2 is a value that is indirectly determined by the amplitude of the second triangular wave vtr2.

Times t0 to t7 are one half-cycle, and in the remaining half-cycle, the gate signals of switching elements S1 to S8 at each time are all on/off reversed signals.

Here, the second triangular wave vtr2 is compared with the switching element S1 to generate gate signals for the other switching elements, and each signal will be explained below. The on/off voltage difference of switching element S1 is 1. The gate signal of switching element S1 is 0 when it is off and 1 when it is on. Since the second leg of the primary side is switched every half cycle, the median of the second triangular wave vtr2 is 0.5, the same as the median of the gate signal of switching element S1. Therefore, when the amplitude of the second triangular wave vtr2 is 2α, the second triangular wave vtr2 is a triangular wave whose value changes in the range of 0.5-α to 0.5+α.

In the above description, the first phase interval δ1 and the second phase interval δ2 are both values that can be derived from the second triangular wave vtr2, so they are equal, but they may be different values. In this case, the switching timing of the secondary side switching elements S7 and S8 in the secondary side second leg 22 is derived not from the second triangular wave vtr2 but from the third triangular wave vtr3. As with the primary side second leg 12, the secondary side second leg 22 compares the magnitude relationship between the third triangular wave vtr3 and the gate signal of the switching element S1, and switches at a timing delayed by the sum of the bridge phase difference φB and the leg phase difference φL. In addition, the third triangular wave vtr3 is a triangular wave whose value changes in the range from 0.5-β to 0.5+β when the amplitude of the third triangular wave vtr3 is 2β.

By using different triangular waves for the second triangular wave vtr2 and the third triangular wave vtr3, it becomes possible to respond to an environment in which the primary voltage V1 or the secondary voltage V2 of the DC-DC converter 1 fluctuates. In this case, the amplitudes of the second triangular wave vtr2 and the third triangular wave vtr3 are determined so that the effective values of the transformer primary voltage vt1 and the transformer secondary voltage vt2 are approximately equal and the ZVS conditions are satisfied. In addition, by using different triangular waves for the second triangular wave vtr2 and the third triangular wave vtr3, even if voltage fluctuations occur, the output current is still adjusted by adjusting the first phase interval δ1 and the second phase interval δ2.

(One-sided or two-sided reflux)
FIG. 3 is an enlarged timing chart of the DC-DC converter 1 according to the first embodiment, showing the transformer primary voltage vt1, the transformer secondary voltage vt2, the transformer primary current it1, the transformer secondary current it2, and the output power Pout.

As shown in Figures 2 and 3, at time t3, the transformer primary voltage vt1 becomes 0. Figure 4 is a diagram showing the current path in the equivalent circuit of the DC-DC converter 1 at this time. As shown in Figure 4, in the primary bridge circuit, current flows back through the switching element S1 and the freewheel diode D3, and in the secondary bridge circuit, current flows from the freewheel diode D5 through the DC power supply E2 to the freewheel diode D8. The freewheel current is gradually reduced by the DC power supply E2 and eventually reverses.

In this way, when a return current occurs in the primary bridge circuit 10 or the secondary bridge circuit 20, and the transformer primary voltage vt1 or the transformer secondary voltage vt2 becomes zero, this is called a one-sided return current. Note that in Figure 4, the DC-DC converter 1 is represented by an equivalent circuit in which the conversion unit 30 is represented only by an inductance component.

Next, at time t4, the transformer secondary voltage vt2 becomes 0. Therefore, between time t4 and time t5, the transformer primary voltage vt1 and the transformer secondary voltage vt2 are both 0. Figure 5 is a diagram showing the current path in the equivalent circuit of the DC-DC converter 1 at this time. As shown in Figure 5, in the primary bridge circuit, current flows and circulates through the switching element S1 and the freewheel diode D3, and in the secondary bridge circuit, current flows and circulates through the switching element S5 and the freewheel diode D7.

When a return current occurs in the primary bridge circuit 10 and the secondary bridge circuit 20, and both the transformer primary voltage vt1 and the transformer secondary voltage vt2 become zero, this is called a double-sided return current.

Next, at time t5, the second leg 12 on the primary side switches, causing the transformer primary voltage vt1 to become non-zero. Figure 6 is a diagram showing the current path in the equivalent circuit of the DC-DC converter 1 at this time. Therefore, as shown in Figure 6, in the primary bridge circuit, current flows from the freewheel diode D1 through the DC power supply E1 to the freewheel diode D4, and in the secondary bridge circuit, current flows and circulates through the switching element S5 and the freewheel diode D7. The freewheel current is gradually reduced by the DC power supply E1 and eventually reverses again.

Here, the period from time t3 to time t5 during which reflux occurs in the primary bridge circuit 10 corresponds to the first phase interval δ1. Also, the period from time t4 to time t6 during which reflux occurs in the secondary bridge circuit 20 corresponds to the second phase interval δ2.

As shown in FIG. 3, the output power Pout = V2 x I2 decreases from time t3 to time t6. This is because one-sided or two-sided return is in effect.

As a result, in the DC-DC converter 1, the instantaneous current when the secondary switching element is turned on can be made to a sufficient current value to enable ZVS even at low output. Therefore, in the DC-DC converter 1, ZVS is possible even at low output.

(Theoretical calculation of DC-DC converter)
The theoretical current equation from time t0 to time t7 is as follows.

また、平均化した出力電力Ｐｏｕｔの式は次のようになる。ここで、第１位相間隔δ１および第２位相間隔δ２を無視した場合でのパルス幅をτとする。また、簡略化のために、Ｖ２’＝ｎ１／ｎ２×Ｖ２と表記している。

The formula for the average output power Pout is as follows: Here, the pulse width when the first phase interval δ1 and the second phase interval δ2 are ignored is τ. For simplification, V2′=n1/n2×V2 is used.

（１）～（８）式より、トランス１次側電圧ｖｔ１およびトランス２次側電圧ｖｔ２とトランスＴｒの巻き線比の整合が取れている場合において、δ１＝δ２とすることで、第１位相間隔δ１および第２位相間隔δ２の値によらず、各部の電流値は一定となる。また、トランス１次側電圧ｖｔ１およびトランス２次側電圧ｖｔ２に電圧変動が起こった際も、第１位相間隔δ１および第２位相間隔δ２をそれぞれ違う値で調節すれば、各部の電流値は一定になる。

From equations (1) to (8), when the transformer primary voltage vt1 and the transformer secondary voltage vt2 are matched with the winding ratio of the transformer Tr, by setting δ1 = δ2, the current value of each part becomes constant regardless of the values of the first phase interval δ1 and the second phase interval δ2. Also, even when a voltage fluctuation occurs in the transformer primary voltage vt1 and the transformer secondary voltage vt2, if the first phase interval δ1 and the second phase interval δ2 are adjusted to different values, the current value of each part becomes constant.

In addition, Pall in equation (9) represents the theoretical output formula for one-sided pulse width control, and Pδ represents the power that would have been transmitted if the first phase interval δ1 and the second phase interval δ2, which are the one-sided return and two-sided return sections, did not exist. Therefore, by adjusting the first phase interval δ1 and the second phase interval δ2, the averaged transport power can be made low output without reducing the instantaneous current in each section. Therefore, since the instantaneous current in each section remains the same as at high output, losses can be reduced even at low output.

(DC-DC converter control section)
FIG. 7 is a block diagram showing the configuration of the control unit 40.

The bridge phase difference φB and the leg phase difference φL are determined by block diagram 51. The ZVS possible minimum current imin_ZVS is multiplied by a coefficient a, and the difference between the secondary current peak value it2_peak is amplified by PID (Proportional Integral Differential) control. The amplified value is set as the bridge phase difference φB and the leg phase difference φL. Here, the secondary current peak value it2_peak is the maximum value (peak value) of the transformer secondary current it2, for example, the transformer secondary current it2 at time t2 in FIG. 3. The ZVS possible minimum current imin_ZVS and the coefficient a are as follows.

すなわち、１次側ブリッジ回路１０と２次側ブリッジ回路２０との位相差であるブリッジ間位相差φＢと、２次側ブリッジ回路２０のレグ間の位相差であるレグ間位相差φＬとを、変換部３０から２次側ブリッジ回路に流れる２次側電流ピーク値ｉｔ２＿ｐｅａｋの大きさが、ＺＶＳ可能最小電流ｉｍｉｎ＿ＺＶＳに係数ａをかけた値になるように制御している。

That is, the bridge-to-bridge phase difference φB, which is the phase difference between the primary bridge circuit 10 and the secondary bridge circuit 20, and the leg-to-leg phase difference φL, which is the phase difference between the legs of the secondary bridge circuit 20, are controlled so that the magnitude of the secondary side current peak value it2_peak flowing from the conversion unit 30 to the secondary side bridge circuit becomes a value obtained by multiplying the ZVS possible minimum current imin_ZVS by the coefficient a.

The command value vref and the first triangular wave vtr1 are compared in magnitude using block diagram 52. The comparison result becomes the gate signal of switching element S1, and the inverse becomes the gate signal of switching element S2. In addition, the gate signals of switching elements S1 and S2 are delayed in phase by the bridge-to-bridge phase difference to become the gate signals of switching elements S5 and S6, respectively.

According to block diagram 53, the difference between the target power Pout* and the output power Pout is amplified by PID control. The amplified result is a coefficient called amp.

The gate signal of switching element S1 is compared with the second triangular wave vtr2, whose amplitude is set to amp, using block diagram 54. The comparison result becomes the gate signal of switching element S3, and the inverse becomes the gate signal of switching element S4.

Block diagram 55 compares the gate signal of switching element S1 with the second triangular wave vtr2 with an amplitude of amp. The gate signal of switching element S7 is delayed in phase by the bridge-to-bridge phase difference φB and the leg-to-leg phase difference φL from the comparison result, and the inverse of this delay becomes the gate signal of switching element S8. If the first phase interval δ1 and the second phase interval δ2 are not equal, as described above, in block diagram 55, the third triangular wave vtr3 can be used instead of the second triangular wave vtr2.

As described above, the first phase interval δ1 and the second phase interval δ2 are indirectly determined by amp, which determines the transmission power.

(DC-DC converter start-up procedure)
Next, the procedure for actually starting up the DC-DC converter 1 will be described.

The bridge-to-bridge phase difference φB and leg-to-leg phase difference φL are gradually increased from sufficiently small values so that the secondary current peak value it2_peak reaches the ZVS-possible minimum current imin_ZVS. At this time, the bridge-to-bridge phase difference φB and leg-to-leg phase difference φL are set to equal values. This process enables the DC-DC converter 1 to start up in a situation where ZVS is possible. At this time, the amp value is set so that the first phase interval δ1 and the second phase interval δ2 are large values, resulting in a state where almost no power is output.

Next, the value of amp is changed so that the output power Pout becomes the target power Pout*, and the gate signal of the switching element S1 and the gate signal of the switching element S5 are modulated by the second triangular wave vtr2. At this time, the phase difference between the bridges φB and the phase difference between the legs φL are fixed values.

As described above, the bridge phase difference φB and the leg phase difference φL are determined under the condition that they are equal. In addition, the first phase interval δ1 and the second phase interval δ2 are determined according to the power transported from the primary side to the secondary side or from the secondary side to the primary side.

As a result, even at low output, an instantaneous current exceeding the minimum ZVS-possible current imin_ZVS can be guaranteed in the secondary switching elements S5 to S8, making ZVS possible and enabling power transmission with low loss.

Figure 8 is a timing chart of the DC-DC converter 1 at low output. As shown in Figure 8, at low output, including when the DC-DC converter is starting up, the amplitude of the gate signals of switching element S1 and switching element S5 is small relative to the amplitude of the second triangular wave vtr2. As a result, time t3 and time t4 advance, time t5 and time t6 delay, and the first phase interval δ1 and second phase interval δ2 become longer.

By lengthening the first phase interval δ1 and the second phase interval δ2, the period for transmitting power is shortened, making it possible to transmit power at low output. Even in this case, if the relationships in equations (1) to (8) are satisfied and the bridge-to-bridge phase difference φB and the leg-to-leg phase difference φL are equal, the current values at times t0 to t7 in Figures 2 and 8 are equal. Therefore, even at low output, a current value that enables ZVS when the secondary side switching elements S5 to S8 are turned on can be secured, making ZVS possible in the secondary side switching elements S5 to S8. As a result of ZVS, the reactive current can be reduced.

〔summary〕
A DC-DC converter according to a first aspect of the present invention includes a primary bridge circuit including a primary first leg and a primary second leg, each of which has a pair of primary switching elements and a capacitor element connected in parallel to each of the primary switching elements, a secondary bridge circuit including a secondary first leg and a secondary second leg, each of which has a pair of secondary switching elements and a capacitor element connected in parallel to each of the secondary switching elements, a conversion unit having a transformer and connected between the primary bridge circuit and the secondary bridge circuit, and switches for the primary switching elements and the secondary switching elements. and a control unit that controls switching between the legs of the primary side bridge circuit, wherein the control unit sets a phase difference between the legs of the primary side bridge circuit to zero, controls each of the primary side switching elements to perform switching every half cycle, and further controls the primary side switching elements of the primary side second leg to perform switching an even number of times in each half cycle, provides a phase difference between the legs of the secondary side bridge circuit, controls each of the secondary side switching elements to perform switching every half cycle, and further controls the secondary side switching elements of the secondary side second leg to perform switching an even number of times in each half cycle.

The above configuration allows a single-side return section and a double-side return section to be created in the DC-DC converter.

In the DC-DC converter according to aspect 2 of the present invention, in the above aspect 1, the control unit determines the phase difference between the primary bridge circuit and the secondary bridge circuit and the phase difference between the legs of the secondary bridge circuit so that the peak value of the conversion unit secondary current flowing from the conversion unit to the secondary bridge circuit is equal to or greater than a predetermined value.

With the above configuration, it is possible to determine the phase difference between the primary bridge circuit and the secondary bridge circuit, and the phase difference between the legs of the secondary bridge circuit, and it is possible to transmit power from the primary side to the secondary side.

In the DC-DC converter according to aspect 3 of the present invention, in the above aspect 2, the predetermined value is a current value that enables zero-volt switching when the secondary-side switching element is turned on.

The above configuration enables ZVS in both the one-side return section and the two-side return section, reducing losses at low output.

In the DC-DC converter according to aspect 4 of the present invention, in the above aspect 2 or 3, the control unit determines the phase difference between the primary bridge circuit and the secondary bridge circuit and the phase difference between the legs of the secondary bridge circuit under the condition that the phase difference between the primary bridge circuit and the secondary bridge circuit and the phase difference between the legs of the secondary bridge circuit are equal.

The above configuration allows the DC-DC converter to start up.

In the DC-DC converter according to aspect 5 of the present invention, in the above aspects 1 to 4, the control unit controls the primary side switching element of the primary side second leg to perform switching twice at a first phase interval within each half cycle, and controls the secondary side switching element of the secondary side second leg to perform switching twice at a second phase interval within each half cycle, and determines the first phase interval and the second phase interval according to the power transported from the primary side to the secondary side or from the secondary side to the primary side.

With the above configuration, the output power can be adjusted by adjusting the first phase interval and the second phase interval.

[Additional Notes]
The present invention is not limited to the above-described embodiments, and various modifications are possible within the scope of the claims. Embodiments obtained by appropriately combining the technical means disclosed in different embodiments are also included in the technical scope of the present invention.

REFERENCE SIGNS LIST 1 DC-DC converter 10 Primary bridge circuit 11 Primary first leg 12 Primary second leg 20 Secondary bridge circuit 21 Secondary first leg 22 Secondary second leg 30 Conversion section 40 Control section 51-55 Block diagram C1, C2 Capacitor elements Csnub1-Csnub8 Snubber capacitor elements D1-D8 Freewheeling diodes E1, E2 DC power supply I1 Primary current I2 Secondary current it1 Transformer primary current it2 Transformer secondary current L1, L2 Reactor Pout Output power Pout* Target power S1-S8 Switching elements Tr Transformer t0-t7 Time V1 Primary voltage V2 Secondary voltage vt1 Transformer primary voltage vt2 Transformer secondary voltage vref Command value vtr1 First triangular wave vtr2 Second triangular wave φB Phase difference between bridges φL Phase difference between legs δ1 First phase interval δ2 Second phase interval

Claims (5)

a primary bridge circuit including a primary first leg and a primary second leg, each of which has a pair of primary switching elements and a capacitor element connected in parallel to each of the primary switching elements;
a secondary bridge circuit including a secondary first leg and a secondary second leg, each of which has a pair of secondary switching elements and a capacitor element connected in parallel to each of the secondary switching elements;
a conversion unit having a transformer and connected between the primary bridge circuit and the secondary bridge circuit;
a control unit that controls switching of the primary side switching element and the secondary side switching element,
The control unit is
A phase difference between the legs of the primary bridge circuit is set to 0, and each of the primary switching elements is controlled to perform switching every half cycle;
The primary-side switching element of the primary-side second leg is further controlled to perform switching an even number of times within each half cycle,
A phase difference is provided between the legs of the secondary bridge circuit, and each of the secondary switching elements is controlled to perform switching every half cycle;
The DC-DC converter further comprises controlling the secondary-side switching element of the secondary-side second leg so as to perform switching an even number of times within each half cycle. The control unit is
2. The DC-DC converter according to claim 1, wherein a phase difference between the primary bridge circuit and the secondary bridge circuit and a phase difference between the legs of the secondary bridge circuit are determined so that a peak value of a conversion unit secondary current flowing from the conversion unit to the secondary bridge circuit is equal to or greater than a predetermined value. The DC-DC converter according to claim 2, characterized in that the predetermined value is a current value that enables zero-volt switching when the secondary-side switching element is turned on. The control unit is
4. The DC-DC converter according to claim 2 or 3, wherein the phase difference between the primary bridge circuit and the secondary bridge circuit and the phase difference between the legs of the secondary bridge circuit are determined under a condition in which the phase difference between the primary bridge circuit and the secondary bridge circuit and the phase difference between the legs of the secondary bridge circuit are equal. The control unit is
The primary-side switching element of the primary-side second leg is further controlled to perform switching twice at a first phase interval within each half cycle,
The secondary-side switching element of the secondary-side second leg is further controlled to perform switching twice at a second phase interval within each half cycle,
5. The DC-DC converter according to claim 1, wherein the first phase interval and the second phase interval are determined according to power transported from the primary side to the secondary side or from the secondary side to the primary side.

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