JP7590029B2 - Vector Angle Control Method - Google Patents

Description

The present invention belongs to the technical field of power electronics control, and in particular to a control method based on vector angles.

Three-phase large-capacity converters are increasingly used as energy conversion devices in industrial fields such as electrified transportation and ship power systems. This kind of converter usually works under the condition of low carrier ratio, has large control and modulation delay, and insufficient stability margin of the control loop, which affects its dynamic performance.

Three-phase converters are usually controlled by proportional integral controllers (i.e., PI controllers) or proportional resonant controllers (i.e., PR controllers). PI controllers realize the active power control of three-phase currents in a synchronous rotating coordinate system. However, limited by the switching frequency of high-power devices, which is usually less than 1000 Hz, the system control delay can reach the millisecond level, and its phase margin and corresponding dynamic performance are very poor when using the mathematical tools of complex transfer functions for modeling and analysis.

In a three-phase three-wire high power conversion system, there are three-phase asymmetric situations due to the dq-axis impedance asymmetry of the salient pole synchronous motor, three-phase asymmetric load, asymmetric fault conditions, etc. Three-phase asymmetric converters can use PR controllers to realize the active power control of three-phase currents in the stationary coordinate system. However, limited by the switching frequency of high power devices, which is usually less than 1000 Hz, the system control delay can reach the millisecond level, and its phase margin and corresponding dynamic performance are also very insufficient when using mathematical tools such as transfer function matrix and characteristic trajectory analysis for modeling and analysis.

In low carrier ratio operating conditions where a large-capacity converter is deployed, the present invention introduces a new degree of control freedom to the conventional PI controller or PR controller to increase the phase margin and more effectively improve the stability and dynamic performance of the system. To improve the dynamic performance of the large-capacity converter, the present invention provides a vector angle control method.

To achieve the object of the present invention, the present invention provides a vector angle control method used for vector angle proportional integral control or vector angle proportional resonant control. The vector angle control method includes the following steps.

When sampling the current of each phase of the controlled converter and performing vector angle proportional integral control, the currents i _d and i _q in the synchronous coordinate system are obtained by abc/dq coordinate transformation, and the complex vector expression of the current sampled value is defined as i _dq = i _d + j i _q , where i _d and i _q are the d-axis and q-axis current values in the synchronous coordinate system, respectively, and j is the imaginary unit. When performing vector angle proportional resonant control, the currents i _α and i _β in the stationary coordinate system are obtained by abc/αβ coordinate transformation, and the two-dimensional column vector expression of the current sampled value is defined as I _αβ = [i _α i _β ] ^T , where i _α and i _β are the α-axis and β-axis current values in the stationary coordinate system, respectively, and the superscript T is transposition.

Subtract the current sampling value from the current reference value to obtain the current error value.

When the current error value is used as the input of the vector angle proportional integral part or the vector angle proportional resonance part, and the output of the corresponding vector angle proportional integral part or the vector angle proportional resonance part is obtained by calculation to perform vector angle proportional integral control, the calculation formula for the output m _{dq_R} of the vector angle proportional integral part is m _{dq_R} = i _{dq_E} · (K _p + K _i · e ^jθi /s), where K _p is a proportionality coefficient, K _i is an integral coefficient, θ _i is a vector angle, s is a Laplace operator, j is an imaginary unit, and i _{dq_E} is a current error value. When vector angle proportional resonance control is performed, the calculation formula for the output M _{αβ_R} of the vector angle proportional resonance control is as follows:
where _Kpα and _Kpβ are the proportionality coefficients of the α-axis and β-axis in the stationary coordinate system, _Krα and _Krβ are the resonance coefficients of the α-axis and β-axis in the stationary coordinate system, _ω0 is the fundamental angular frequency, s is the Laplace operator, _θrp is the difference vector angle, whose value is a compromise between −90° and 0°, and _{Iαβ_E} is the current error value.

The current sampling value is used as the input to the decoupling part, and the decoupling output is obtained through calculation.

The output of the vector angle proportional integral part or the vector angle proportional resonance part and the decoupling output are added as the input to the delay compensation part, and the output of the delay compensation part is the total output of the control loop.

The dq/abc coordinate or αβ/abc coordinate transformation corresponding to the total output of the control loop is performed to obtain the three-phase modulated wave, which is compared with the carrier wave in the modulation and driving module, and a driving signal is generated to drive the converter topology to realize the electrical energy conversion.

In one embodiment of the present invention, when performing vector angle proportional integral control, before using the current error value as the input of the vector angle proportional integral part to obtain the output of the vector angle proportional integral part through calculation, the processing of the phase equalization part is also included, which involves introducing a phase equalization angle to equalize the current error i _{dq_E} , and then using the equalized current error as the input of the vector angle proportional integral part to obtain the output m _{dq_R} of the vector angle proportional integral part through calculation, and the calculation formula for the output m _{dq_R} of the vector angle proportional integral part is m _{dq_R} = i _{dq_B} · (K _p + K _i1 · e ^jθi /s), where i _{dq_B} = i _{dq_E} · e ^jθb , θ _b is the phase equalization angle, and i _{dq_B} is the phase equalization result.

In one embodiment of the present invention, when vector angle proportional integral control is performed, the calculation formula for the decoupling output m _{dq_D} is m _{dq_D} =i _dq ·jω ₀ L, where L is the AC side inductance value and ω ₀ is the fundamental angular frequency.

In one embodiment of the present invention, when vector angle proportional integral control is performed, the calculation formula for the output _mdq of the delay compensation part is _mdq = _{mdq_RD} or _mdq = _{mdq_RD} ·e ^jnTs·ω0 , where _{mdq_RD} = _{mdq_R} + _{mdq_D} , _Ts is the control period, n is the compensation coefficient, and _ω0 is the fundamental angular frequency.

In one embodiment of the present invention, when vector angle proportional resonance control is performed, the calculation formula for the decoupling output M _{αβ_D} is as follows:
Here, L _α and L _β are equivalent inductance values of the α-axis and β-axis in the stationary coordinate system, respectively, and ω ₀ is the fundamental angular frequency.

In one embodiment of the present invention, when vector angle proportional resonance control is performed, the calculation formula for the output M _αβ of the delay compensation part is as follows:
Here, M _{αβ_RD} =M _{αβ_R} +M _{αβ_D} , n is a compensation coefficient, T _s is a control period, and ω ₀ is a fundamental angular frequency.

In summary, the present invention aims at the control of three-phase converters in the dq synchronous coordinate system, and the existing phase equalization method can only equalize the positive and negative phase margins, but cannot increase the overall phase margin. The present invention proposes a vector angle PI control, which, based on the traditional PI control, can realize simultaneous improvement of the positive and negative phase margins by introducing a new control degree of freedom vector angle θ _i, thereby improving the stability margin and dynamic performance under low carrier ratio operating conditions, and achieving beneficial technical effects. On the other hand, when the carrier ratio is low, in the traditional PR control, the positive and negative phase margins of the three-phase asymmetric converter system are low and even unstable. The present invention proposes a matrix vector angle PR control, which, based on the traditional PR control, can realize simultaneous improvement of the positive and negative phase margins of the characteristic trajectory by introducing a new control degree of freedom difference vector angle θ _rp , thereby improving the stability margin and dynamic performance under low carrier ratio operating conditions, and achieving beneficial technical effects.

FIG. 1 is a schematic diagram of a power conversion circuit. FIG. 2 is an overall control block diagram according to the first embodiment of the present invention. FIG. 2 is a block diagram of a control loop with differential phase compensation resonance control according to a first embodiment of the present invention; FIG. 2 is a block diagram of an implementation of a complex vector in the real field of a control loop according to a first embodiment of the present invention; FIG. 2 is a two-sided frequency domain Bode plot of a vector angle PI controller in accordance with a first embodiment of the present invention; 1 is a diagram showing transient current waveforms in a synchronous coordinate system according to a conventional method and method 1 of the first embodiment; 10A and 10B are diagrams showing transient current waveforms in the first and second modes of the first embodiment in a synchronous coordinate system. FIG. 11 is an overall control block diagram according to a second embodiment of the present invention. FIG. 4 is a block diagram of a control loop with matrix vector angle proportional resonant control according to a second embodiment of the present invention. FIG. 11 is a two-sided frequency domain Bode plot of the characteristic trajectory of a matrix vector angle PR controller in accordance with a second embodiment of the present invention. FIG. 13 is a diagram showing transient current waveforms in the conventional method and the method of the second embodiment.

Figure 1 is a schematic diagram of a power conversion circuit, which is applicable to the first and second embodiments described below. Figures 2 to 7 are schematic diagrams related to a vector angle proportional integral control method according to a first embodiment of the present invention. Figures 8 to 10 are schematic diagrams related to a vector angle proportional resonant control method according to a second embodiment of the present invention.

The first embodiment of the present invention takes the current loop control of a general three-phase bridge inverter topology as an example. The three-phase current is sampled to obtain the AC side currents i _a , i _b , i _c , and the currents i _d , i _q in the stationary coordinate system are obtained by abc/dq coordinate transformation, which are the input of the control loop. Then the control loop outputs the modulated waves m _d , m _q in the stationary coordinate system, and the three-phase modulated waves m _a , _mb , m _c are obtained by dq/abc coordinate transformation, which are compared with the carrier wave in the modulation and driving module, and the driving signal is generated to drive the converter topology to realize electrical energy conversion.

The vector angle proportional integral control method provided by the first embodiment of the present invention includes a phase equalization part, a vector angle PI part, and a delay compensation part. Here, the expression of complex vector and complex transfer function is used. Take the current sampling value i _dq in the stationary coordinate system as an example, the complex vector is i _dq =i _d +ji _q , where j is the imaginary unit, i _d and i _q respectively represent the current values of the d-axis and q-axis, and the definition of the remaining complex vectors with subscripts dq is the same.

The control loop samples the corresponding i _dq from the controlled object, and outputs a modulated wave m _dq to control the controlled object. In the first embodiment of the present invention, the steps of the proportional-integral control method based on the vector angle corresponding to the control loop are as follows:

1) The current of each phase of the controlled converter is sampled, and the currents i _d and i _q in the synchronous coordinate system are obtained by abc/dq coordinate transformation. The current sampled value is defined as i _dq = i _d + ji _q , where i _d and i _q are the current values of the d-axis and q-axis in the synchronous coordinate system, respectively, i _dq is a complex vector, and j is an imaginary unit.

2) The current sampled value i _dq is subtracted from the current reference value i _{dq_R} to obtain a current error value i _{dq_E} .

3) The current error i _{dq_E} is directly used as i _{dq_B} , or is used as the input of the phase equalization part, and i _{dq_B} is obtained by calculation, and the calculation formula of the phase equalization part is as follows:
i _{dq_B} = i _{dq_E} or i _{dq_B} = i _{dq_E}・e ^jθb
where θ _b is the phase equalization angle, whose value can be zero or, typically, set to half the difference between the positive and negative phase margins to equalize the phase margins on both sides.

4) The equalized current error i _{dq_B} is used as the input of the vector angle PI part, and m _{dq_R} is obtained by calculation. The calculation formula for the vector angle PI part is as follows:
m _{dq_R} = i _{dq_B}・(K _p +K _i・e ^jθi /s)
Here, K _p is a proportionality coefficient, K _i is an integral coefficient, θ _i is a vector angle proposed in this embodiment, and s is a Laplace operator.

5) The current sampling value i _dq is used as the input of the decoupling part, and the decoupling output m _{dq_D} is obtained by calculation. The calculation formula of the decoupling part is as follows:
m _{dq_D} = i _dq・jω ₀ L
Here, L is the AC side inductance value, and ω ₀ is the fundamental angular frequency.

6) As the input of the delay compensation part, the output m _{dq_R} of the vector angle PI and the decoupling output m _{dq_D} are added to obtain m _{dq_RD} , and the total output m _dq of the control loop is obtained by calculation. The calculation formula of the delay compensation part is as follows:
m _dq = m _{dq_RD} or m _dq = m _{dq_RD}・e ^jnTs・ω0
where _Ts is the control period and n is a compensation factor, which may have a typical value of 1.5, 0, or any other value.

7) Perform dq/abc coordinate transformation on the sum output m _dq of the control loop to obtain three-phase modulated waves m _a , m _b , and m _c , which are compared with the carrier wave in the modulation and driving module, and the driving signal is generated to drive the converter topology to realize electrical energy conversion.

Next, we briefly explain the implementation of the above complex vector in the real number field. The control loop expression includes an imaginary unit j, which represents the cross-coupling between the d-axis and the q-axis. The feedback decoupling part includes a term ω ₀ L with j in the numerator, i.e., m _{dq_D} =i _dq ·jω ₀ L, and its implementation in the real number field is shown in FIG. 4(a), i.e.,
m _{d_D} =-i _q・ω ₀ L, m _{q_D} = i _d・ω ₀ L
Here, m _{d_D} and m _{q_D} respectively represent the decoupling outputs of the d-axis and q-axis. In addition, the vector angle PI part, the phase equalization part (i _{dq_B} =i _{dq_E} ·e ^jθb ), and the delay compensation part (m _dq =m _{dq_RD} ·e ^jnTs·ω0 ) include exponential functions. Taking the general form of the above exponential function y _dq =u _dq ·e ^jθ as an example, its implementation in the real number field is shown in FIG. 4(b), that is, as follows:
y _d = u _d・cosθ−u _q・sin θ, y _q = u _d・sin θ+u _q・cos θ
Here, _ydq and _udq also use the definition of complex vectors given above, i.e., _yd + _jyq = _ydq , _ud + _juq = _udq , and θ represents the angle used to advance the phase with the exponential function.

The following is an example of application of the first embodiment of the present invention.

For the three-phase power conversion circuit shown in Fig. 1, the general control method is to sample the three-phase currents to obtain the AC side currents i _a , i _b , and i _c , and then obtain the stationary coordinate system currents i _d and i _q through abc/dq coordinate transformation, which are the inputs of the control loop. Here, the specific implementation process of the control loop is the same as above, including phase equalization, PI, feedback decoupling, and delay compensation. For the PI part, compared with the vector angle PI with an additional vector angle control degree of freedom proposed in this embodiment, the traditional method corresponds to the following equation:
m _{dq_R} = i _{dq_B}・(K _p +K _i /s)

The output of the above control loop is the modulated waves _md , _mq in the synchronous coordinate system, and the three-phase modulated waves m _a , _mb , m _c are obtained through dq/abc coordinate transformation, and compared with the carrier wave in the modulation and driving module to generate the driving signal to drive the converter topology to realize electrical energy conversion.

In the case of the control of the three-phase converter in the dq synchronous coordinate system under the low carrier ratio operating condition corresponding to the large-capacity converter, the existing phase equalization method can only equalize the positive and negative phase margins, and cannot increase the overall phase margin. In consideration of the shortcomings of the conventional phase equalization method, this embodiment proposes the vector angle PI, which, based on the conventional PI control, can realize the simultaneous improvement of the positive phase margin and the negative phase margin by introducing the vector angle θ _i of a new control degree of freedom, thereby improving the stability margin and dynamic performance under the low carrier ratio operating condition, and the specific analysis is as follows:

Using the complex transfer function, the PI controller before and after the improvement was analyzed, and the two-sided frequency domain Bode plots shown in FIG. 5 were obtained. In this figure, θ _i is the vector angle proposed in this embodiment, whose value ranges from 0° to 90°. Considering that the larger the value, the stronger the phase lead ability on the right side of the pole, and the larger the value, the smaller the bandwidth on the left side of the pole, the compromise value of 60° is set here for θ _i . After applying the vector angle proposed in this embodiment, it can be seen that the phase lag of the positive frequency band is greatly reduced, and the phase margin of the corresponding positive terminal is greatly increased. For the negative frequency band, when the crossover frequency is set to the left of −10 Hz, the reduction in the phase margin of the corresponding negative terminal is smaller than the increase in the phase margin of the positive terminal, and when it is further set to the left of −26 Hz, the phase margin of the corresponding negative terminal also increases slightly. Therefore, the vector angle PI controller can realize the function of increasing the sum of the phase margins on both sides.

Next, a time domain comparison analysis is performed between the conventional method and the method of this embodiment. In the parameter settings, the frequency carrier ratio is 5, the bandwidth f _c is 60Hz, the proportional coefficient K _p is 2πf _c L, and the integral coefficient K _i is πf _c K _p /2. In the conventional method, the vector angle θ _i and the phase equalization angle θ _b are both 0°, and in the method 1 and method 2 of this embodiment, the vector angle θ _i are both 60°, and the phase equalization angle θ _b is still 0 in method 1, and 41.7° is selected in method 2 to equalize the two-sided phase margin, that is, method 1 corresponds to i _{dq_B} = i _{dq_E} , and method 2 corresponds to i _{dq_B} = i _{dq_E} ·e ^jθb .

Comparing the conventional method with method 1 of this embodiment, as shown in Figure 6, when the active current command jumps from 0 pu to 1 pu in 0.03 seconds, the current response of the conventional PI controller diverges and becomes unstable, while the current response of the vector angle PI controller can achieve nearly critical stability.

Further comparing method 1 of this embodiment with method 2 of this embodiment, as shown in FIG. 7, the vector angle PI controller corrects the current loop from an unstable state to a state where the phase margin is close to 45° and the transient adjustment time is significantly reduced to about 0.03 s under the condition that the frequency carrier ratio is 5 and the design bandwidth is 60 Hz.

Therefore, the vector angle-based proportional-integral control can improve the stability margin and dynamic performance of the converter under low carrier ratio operating conditions of large-capacity converters, achieving beneficial technical effects.

The second embodiment of the present invention takes the current loop control of a general three-phase bridge inverter topology as an example. The three-phase current is sampled to obtain the AC side currents i _a , i _b , and i _c , and the currents i _α and i _β in the stationary coordinate system are obtained through αβ/abc coordinate transformation, which are input to the control loop. Then the control loop outputs the modulated waves m _α and m _β in the stationary coordinate system, and the three-phase modulated waves m _a , _mb , and m _c are obtained through αβ/abc coordinate transformation, which are compared with the carrier wave in the modulation and driving module, and the driving signal is generated to drive the converter topology to realize electrical energy conversion.

The vector angle proportional resonance control method provided by the second embodiment of the present invention includes a matrix vector angle PR part, a feedback decoupling part, and a delay compensation part. Here, the expression of a matrix and a transfer function matrix is used, and the operation of this expression follows the operation rules of a matrix. The control loop samples the corresponding I _αβ from the controlled object, and outputs a modulated wave M _αβ to control the controlled object.

In a second embodiment of the present invention, the steps of a proportional resonance control method based on a matrix vector angle corresponding to a control loop are as follows:

1) The current of each phase of the controlled converter is sampled, and the currents _iα and _iβ in the stationary coordinate system are obtained by abc/αβ coordinate transformation. The current sampled value is defined as _Iαβ = [ _iαiβ ] ^T , where _iα and _iβ are the current values of _{the α-} axis and β-axis in the stationary coordinate system, respectively, _Iαβ is a two-dimensional column vector, and the superscript T is the transpose.

2) The current sampled value I _{αβ is subtracted from the current reference value I αβ_R to} obtain a current error value I _{αβ_E} .

3) The current error I _{αβ_E} is used as the input of the matrix vector angle PR part, and M _{αβ_R} is obtained by calculation. The calculation formula for the matrix vector angle PR part is as follows:
where _Kpα and _Kpβ are respectively proportional coefficients of the α-axis and the β-axis in the stationary coordinate system, _Krα and _Krβ are respectively resonance coefficients of the α-axis and the β-axis in the stationary coordinate system, _θrp is the differential vector angle proposed in the second embodiment of the present invention, _ω0 is the fundamental angular frequency, and s is the Laplace operator.

4) The current sampling value I _αβ is used as the input of the decoupling part, and the decoupling output M _{αβ_D} is obtained by calculation. The calculation formula of the decoupling part is as follows:
Here, L _α and L _β are the equivalent inductance values of the α-axis and β-axis in the stationary coordinate system, respectively.

5) As the input of the delay compensation part, the output M _{αβ_R} of the matrix vector angle PR and the decoupling output M _{αβ_D} are added to obtain M _{αβ_RD} , and the total output M _αβ of the control loop is obtained by calculation. The calculation formula of the delay compensation part is as follows:
where _Ts is the control period and n is a compensation factor, which may have a typical value of 1.5, 0, or any other value.

6) Perform αβ/abc coordinate transformation on the total output M _αβ of the control loop to obtain three-phase modulated waves m _a , m _b , and m _c , which are compared with the carrier wave in the modulation and driving module, and driving signals are generated to drive the converter topology to realize electrical energy conversion.

The following is an example of application of the second embodiment of the present invention.

For the three-phase power conversion circuit shown in Fig. 1, the general control method is to sample the three-phase currents to obtain the AC side currents i _a , i _b , and i _c , and then obtain the currents i _α and i _β in the stationary coordinate system through abc/αβ coordinate transformation, which are the inputs of the control loop. Here, the specific implementation process of the control loop is the same as above, including PR, feedback decoupling, and delay compensation. For the PR part, compared with the matrix vector angle PR with an additional difference vector angle control degree of freedom proposed in the second embodiment of the present invention, the traditional method corresponds to the following equation:

The output of the above control loop is the modulated waves m _α , m _β in the stationary coordinate system, and the three-phase modulated waves m _a , m _b , m _c are obtained through αβ/abc coordinate transformation, and compared with the carrier wave in the modulation and driving module to generate the driving signal to drive the converter topology to realize electrical energy conversion.

In the case of the control of the three-phase converter in the αβ stationary coordinate system under the low carrier ratio operating condition corresponding to the large-capacity converter, in the traditional PR control, the positive and negative phase margins of the system are low, and even unstable. In the second embodiment of the present invention, in consideration of the shortcomings of the traditional PR control method, the matrix vector angle PR is proposed, that is, based on the traditional PR, the difference vector angle θ _rp of the new control degree of freedom is introduced, which can realize the simultaneous improvement of the positive phase margin and the negative phase margin of the characteristic trajectory, thereby improving the stability margin and dynamic performance under the low carrier ratio operating condition, and the specific analysis is as follows:

Using mathematical tools such as transfer function matrix and characteristic trajectory analysis, the PR controller before and after the improvement was analyzed, and the Bode plots in the two-sided frequency domain shown in FIG. 10 were obtained. In this figure, the dotted line represents the conventional method, and the solid line represents the method proposed in the second embodiment of the present invention, and θ _rp is the difference vector angle proposed in the second embodiment of the present invention, whose value ranges from -90° to 0°. The closer the value is to -90°, the more likely the amplitude-frequency characteristic of the characteristic trajectory is to be distorted, and the more likely the system is to become unstable, and the closer the value is to 0°, the less significant the effect of improving the phase margin of the characteristic trajectory is. Therefore, θ _rp is set to a compromise value of -60°. It can be seen that after applying the matrix vector angle PR proposed in the second embodiment of the present invention, the phase margin of the negative frequency band increases from -4.2° to 30.7°, and the phase margin of the positive frequency band increases from 19.9° to 27.5°. Therefore, the matrix vector angle PR controller can realize the function of increasing the phase margin on the positive and negative sides.

Next, a time domain comparison analysis was performed between the conventional method and the method of the second embodiment of the present invention. The parameters were set as follows: frequency carrier ratio 7, bandwidth f _c 50 Hz, proportional coefficient K _pα 2πf _c L _α , K _pβ 2πf _c L _β , resonance coefficient K _rα K _pα ω _c /4, K _rβ K _pβ ω _c /4. In the conventional method, the differential vector angle θ _rp is 0°, and in the method of the second embodiment of the present invention, the differential vector angle θ _rp is -60°.

Comparing the conventional method with the method of the second embodiment of the present invention, as shown in Figure 11, when the active current command jumps from 0 pu to 1 pu in 0.1 seconds, in the conventional method, the current response oscillates many times and then gradually diverges, whereas in the method of the second embodiment of the present invention, the current response converges quickly.

Therefore, the proportional resonant control based on matrix vector angle can improve the stability margin and dynamic performance of the converter under low carrier ratio operating conditions of large-capacity converters, achieving beneficial technical effects.

The present invention is not limited to the above specific embodiments, and those skilled in the art can adopt various other embodiments, such as replacing the feedback decoupling part with a feedforward decoupling, replacing the two-level converter topology with a three-level topology, etc., according to the contents disclosed in the present invention. Therefore, the claims are intended to cover all modifications within the true spirit and scope of the present invention.

Claims (4)

1. A vector angle control method for performing vector angle proportional integral control or vector angle proportional resonant control on a three-phase bridge inverter of a three-phase converter , comprising:
a step of sampling currents of each phase of the three-phase converter to be controlled, in which, when vector angle proportional integral control is performed, currents i _d and i _q in a synchronous coordinate system are obtained by abc/dq coordinate transformation, and a complex vector expression of the current sampling value is defined as i _dq =i _d +ji _q , where i _d and i _q are the d-axis and q-axis current values in the synchronous coordinate system, respectively, and j is an imaginary unit, and when vector angle proportional resonant control is performed, currents i _α , i _β in a stationary coordinate system are obtained by abc/αβ coordinate transformation, and a two-dimensional column vector expression of the current sampling value is defined as I _αβ =[i _α i _β ] ^T , where i _α and i _β are the α-axis and β-axis current values in the stationary coordinate system, respectively, and the superscript T is a transpose;
subtracting the current sampled value from the current reference value to obtain a current error value;
a step of obtaining an output of the vector angle proportional integral part or the vector angle proportional resonance part by calculation using the current error value as an input of the vector angle proportional integral part or the vector angle proportional resonance part; when performing vector angle proportional integral control, the calculation formula for the output m _{dq_R} of the vector angle proportional integral part is m _{dq_R} = i _{dq_E} · (K _p + K _i · e ^jθi /s), where K _p is a proportionality coefficient, K _i is an integral coefficient, θ _i is a vector angle, s is a Laplace operator, j is an imaginary unit, and i _{dq_E} is a current error value; when performing vector angle proportional resonance control, the calculation formula for the output M _{αβ_R} of the vector angle proportional resonance control is as follows:
where _Kpα and _Kpβ are proportional coefficients of the α-axis and β-axis in the stationary coordinate system, _Krα and _Krβ are resonance coefficients of the α-axis and β-axis in the stationary coordinate system, _ω0 is the fundamental angular frequency, s is the Laplace operator, _θrp is the difference vector angle, the value of which is a predetermined value between −90° and 0°, and _{Iαβ_E} is the current error value; and
The current sampling value is used as an input of the decoupling part to obtain a decoupling output by calculation;
A step of adding the output of the vector angle proportional integral part or the vector angle proportional resonance part and the decoupling output and inputting the sum to the delay compensation part, and making the output of the delay compensation part the total output of the control loop . When vector angle proportional integral control is performed, the calculation formula for the output m _dq of the delay compensation part is m _dq = m _{dq_RD} or m _dq = m _{dq_RD} ·e ^jnTs·ω0 , where m _{dq_RD} = m _{dq_R} + m _{dq_D} , T _s is the control period, n is the compensation coefficient, ω ₀ is the fundamental angular frequency, and m _{dq_D} is the decoupling output. When vector angle proportional resonance control is performed, the calculation formula for the output M _αβ of the delay compensation part is as follows:
Here, Mαβ_RD ₌ Mαβ_R ₊ Mαβ_D _, n is the compensation coefficient, Ts _is the control period, ω0 _is the fundamental angular frequency, and _{Mαβ_D} is the decoupling output. Step and
performing dq/abc coordinate or αβ/abc coordinate transformation corresponding to a total output of the control loop to obtain a three-phase modulated wave, and comparing it with a carrier wave in a modulation and driving module to generate a driving signal to drive a three - phase bridge inverter of the three-phase converter to realize electrical energy conversion. 2. The vector angle control method according to claim 1, wherein when performing vector angle proportional integral control, a phase equalization part is also included before the current error value is used as an input of the vector angle proportional integral part to obtain the output of the vector angle proportional integral part through calculation, in which a phase equalization angle is introduced to equalize the current error i _{dq_E} , and then the equalized current error is used as an input of the vector angle proportional integral part to obtain the output m _{dq_R} of the vector angle proportional integral part through calculation, and the calculation formula for the output m _{dq_R} of the vector angle proportional integral part is m _{dq_R} = i _{dq_B} · (K _p + K _i1 · e ^jθi /s), in which i _{dq_B} = i _{dq_E} · e ^jθb , θ _b is the phase equalization angle, and i _{dq_B} is the phase equalization result. The vector angle control method according to claim 1, characterized in that, when performing vector angle proportional integral control, the calculation formula of the decoupling output m _{dq_D is m dq_D =} i _dq · jω ₀ L, where L is the AC side inductance value, and ω ₀ is the fundamental angular frequency. When vector angle proportional resonance control is performed, the calculation formula for the decoupling output M _{αβ_D} is as follows:
2. The vector angle control method according to claim 1, wherein _Lα and _Lβ are equivalent inductance values of the α-axis and β-axis in the stationary coordinate system, respectively, and _ω0 is a fundamental angular frequency.