JP7620591B2 - Motor Control Device - Google Patents

Description

An embodiment of the present invention relates to a motor control device that estimates the rotor speed when starting an induction motor from an initial speed state.

Variable speed motor drive devices that use power converters to drive electric motors at variable speeds are used in a variety of fields. Among electric motors, induction motors are widely used because they have a relatively simple and robust structure. In addition, they have the advantage that they can be driven by directly connecting to a commercial power source as a backup in the unlikely event that the control device breaks down.

When restarting such an induction motor during a period when the rotor is in a free-running state, such as after a momentary power outage or when switching from commercial operation, it is desirable to acquire the rotor speed and restart the motor by matching the output frequency to the rotor speed in order to reduce mechanical shock at startup.

One way to obtain the rotor speed during free-running is to use the induced voltage generated by the secondary magnetic flux remaining in the rotor. Even if the residual magnetic flux is very small, a relatively large induced voltage can be obtained if the rotation speed is high. However, if a long time has passed since the motor stopped operating and the secondary magnetic flux has decreased to near zero, the induced voltage can no longer be observed.

In addition, depending on the installation environment, due to space limitations, a small voltage may be induced in the wiring of the motor due to the influence of wiring of large-capacity power equipment located close to the motor to be controlled. If such a voltage is observed to be larger than the amplitude of the residual voltage of the motor, it may not be possible to accurately estimate the rotor speed.

For example, Patent Document 1 discloses a method for obtaining the rotor speed in a free-running state using residual magnetic flux. According to this method, when it is determined that the residual magnetic flux is small and speed estimation is difficult, a direct current is applied to the motor, and the input voltage vector of the induction motor is detected or estimated during application of the direct current. The secondary magnetic flux is then obtained by integrating the result of subtracting the voltage drop due to the primary resistance of the induction motor and the voltage drop due to the leakage inductance from the voltage vector, and the center of the vector is found from the changes in the vector at three points, and the rotational angular frequency is estimated from the relationship between the passage of time and the phase.

With this method, when the rotor is rotating at a low speed, it takes a long time for the vector locus of the secondary magnetic flux to be obtained. During this time, time passes with DC current flowing through the induction motor, which is thought to result in the generation of a braking torque proportional to the rotation speed.

In response to this problem, Patent Document 2 proposes a method for shortening the time that DC current is applied by integrating the induced voltage minus the voltage drop due to the primary resistance when DC current is applied to an induction motor in a free-running state to estimate the secondary magnetic flux, and estimating the rotation speed using the second derivative of the estimated q-axis component of the secondary magnetic flux. The second derivative of the estimated secondary magnetic flux immediately after the inverter that drives the induction motor starts operating contains a component that is proportional to the rotation speed. Therefore, the rotation speed can be estimated by dividing the second derivative by the product of the excitation inductance, the reciprocal of the secondary time constant, and the excitation current.

Patent No. 3535735 Patent No. 6447183

The methods disclosed in these patent documents all use the amount of magnetic flux that occurs when a current is passed through the motor, so they are considered to be based on the premise that the initial residual magnetic flux is close to zero. For example, if these methods are used in a motor with a relatively long secondary time constant where a relatively large amount of magnetic flux remains, such as when restarting shortly after coasting, there is a risk of torque disturbance occurring due to the current applied to obtain the magnetic flux information used to estimate the speed and the remaining secondary magnetic flux.

Furthermore, these methods only estimate the rotation speed and direction of rotation, and when restarting the motor while the residual magnetic flux is relatively large, there is a risk of a rush current flowing depending on the difference between the induced voltage due to the residual magnetic flux and the initial voltage at the time of restart.

When controlling the torque of a motor, it is expected that the motor will be switched between the operating state and the free-running state relatively frequently, and that it will be restarted in a short time after being placed in the free-running state. In order to suppress current disturbances during restart, it is necessary to apply an output voltage that takes into account the magnitude and phase of the induced voltage due to the residual secondary magnetic flux as the initial value before restarting the motor.

To obtain the residual voltage accurately without causing torque disturbance, it is desirable to provide a voltage detector on the output side of the power converter. However, as time passes after coasting, the residual voltage attenuates, and when the induced voltage becomes lower than the noise components in the installation environment, it becomes difficult to accurately estimate the speed.

Therefore, we provide a motor control device that has a configuration in which a voltage detector is placed on the output side of the power converter, which can minimize the shock at the time of restart even if the residual voltage has decayed, ensure noise resistance, and shorten the time from input of an operation command to restart.

The motor control device of the embodiment includes: a power converter that converts a DC voltage into a three-phase AC voltage with a variable voltage and variable frequency and supplies the AC voltage to an induction motor;
A phase-to-phase voltage detection unit that detects phase-to-phase voltages between two or more phases of the induction motor;
A current detection unit that detects a phase current of the induction motor;
a current adjusting unit that calculates a voltage command value so that a component obtained by converting the detected value of the phase current onto a synchronous coordinate axis coincides with a current command value;
When starting the induction motor from an initial speed state,
A speed estimation excitation operation is performed to provide a step-like current command value having a predetermined magnitude to the current adjustment unit for a predetermined period of time;
and a restart control unit that, after cutting off the output of the power converter, performs convergence calculation processing to obtain the rotational speed and rotational direction of the induction motor and the phase of residual secondary magnetic flux using detection values of the phase-to-phase voltage between two or more phases, and restarts control of the induction motor.
In addition, the motor control device of the embodiment includes a power converter-current adjustment unit similar to the above,
When a request to stop free-run operation is input while the induction motor is in an operating state, or when a main circuit potential of a power converter that drives the induction motor is reduced and it is determined that operation is difficult, and the induction motor transitions to a free-run state, the convergence calculation process is constantly performed;
and a restart control unit that, when an operation command is input again or the main circuit potential is restored, restarts control of the induction motor using the rotational speed and rotational direction and the phase of the residual secondary magnetic flux obtained from the convergence calculation process at that time.

FIG. 1 is a diagram showing a configuration of an electric motor control device according to an embodiment. Functional block diagram showing the configuration of the convergence calculation processing unit Diagram showing the relationship between rotor magnetic flux and induced voltage FIG. 1 shows a DQ synchronous coordinate axis and an MT synchronous coordinate axis. A diagram explaining a method for observing the phase difference between two interphase voltages. Signal waveform diagram showing the state of transition from stop mode to initial speed estimation mode A diagram showing waveforms of phase-to-phase voltage and torque current when an induction motor is restarted from a coasting state. A waveform diagram showing the change in secondary magnetic flux when a current command value is given for a predetermined time. Flowchart showing the processing contents of the control device A timing chart showing a case where the pre-start speed estimation operation is Type 1, the speed estimation excitation operation is enabled, and the pre-start excitation operation is disabled, and the phase-to-phase voltage level determination is NG. A timing chart showing a case where the interphase voltage level determination is NG in the case of pre-start speed estimation operation: Type 1, speed estimation excitation operation: enabled, and pre-start excitation operation: enabled. A timing chart showing a case where the phase-to-phase voltage phase difference determination is NG in the same case as in FIG. Flowchart showing the processing contents of the control device Timing chart showing the cases of pre-start speed estimation operation: Type 2, speed estimation excitation operation: enabled, and pre-start excitation operation: enabled

One embodiment will be described below. As shown in FIG. 1, each phase terminal of a three-phase AC power source 1 is connected to an AC input terminal of a full-wave rectifier circuit 2 consisting of a diode bridge. A smoothing capacitor 3 and a power converter 4 are connected between the DC output terminals of the full-wave rectifier circuit 2. Each phase output terminal of the power converter 4, which is an inverter, is connected to each phase winding terminal of an induction motor 5.

The control device 6, which is a restart control section, controls the power converter 4 to convert the DC voltage into a three-phase AC voltage with variable voltage and variable frequency. Current detection sections 7U, 7W are arranged at the U-phase and W-phase output terminals of the power converter 4, and the U-phase and W-phase currents _iu , _iw detected by these sections are input to a UVW/dq converter 8. The UVW/dq converter 8 converts the three-phase currents into dq-axis currents in vector control and inputs them to a current adjustment section 9 and a speed estimation processing section 10.

Each phase output terminal of the power converter 4 is connected to an input terminal of a phase-to-phase voltage detection unit 11. The phase-to-phase voltage detection unit 11 detects a UV-to-phase voltage _vuv and a WV-to-phase voltage _vwv , and inputs these to a convergence calculation processing unit 12, a phase-to-phase voltage level determination unit 13, and a phase-to-phase voltage phase difference determination unit 14. A gate drive permission signal is provided to one input terminal of an AND gate 16 via a NOT gate 15, and a speed estimation operation command is provided to the other input terminal of the AND gate 16. An output signal of the AND gate 16 is input to the convergence calculation processing unit 12 as a convergence calculation execution command.

The convergence calculation processing unit 12 performs a convergence calculation process, described later, on the input inter-phase voltages to generate and output a secondary flux command value initial value φ _{rd_ref_Init} , a speed command initial value ω _{re_ref_Init} , and an output voltage phase initial value θ _{re_Init} . The secondary flux command value initial value φ _{rd_set_Init} is input to a flux/current command generation unit 17, and the speed command initial value ω _{re_ref_Init} is input to a speed command generation unit 18. The output voltage phase initial value θ _{re_Init} is input to an integrator 27.

The phase-to-phase voltage level determination unit 13 compares the input phase-to-phase voltage with a threshold value and outputs the determination result OK/NG to the restart processing unit 19. The phase-to-phase voltage phase difference determination unit 14 compares the input phase difference between the phase voltages with a threshold value and outputs the determination result OK/NG to the restart processing unit 19.

Further, a secondary flux target value φ _{rd_set} and a secondary flux command value initial value φ _{rd_ref_Init} are input to the flux/current command generating unit 17. Based on these input values, the flux/current command generating unit 17 generates and outputs a d-axis current target value i _{sd_set} , a secondary flux target value φ _{rd_set_Filt} filtered as described below, and a filtered d-axis current target value i _{sd_set_Filt} . Also, the flux/current command generating unit 17 outputs the secondary flux target value φ _{rd_set} as it is.

Further, a speed target value ω _{re_set} is input to the speed command generating unit 18. The speed command generating unit 18 generates a speed command value ω _{re_ref} based on these input values and outputs it to a speed control unit 20. The speed control unit 20 receives an input of a speed estimated value ω _{re_est} output from the speed estimation processing unit 10.

Further, the restart processing unit 19 receives an operation command "ON/OFF", a pre-start speed estimation operation "Type 1/Type 2/invalid", a pre-start speed estimation operation==Type 2 operation "valid/invalid", and a pre-start excitation operation "valid/invalid". Based on these input signals, the restart processing unit 19 generates and outputs signals such as a speed estimation excitation operation command "ON/OFF", a pre-start excitation operation command "ON/OFF", a speed control operation command "ON/OFF", a speed estimation operation command "ON/OFF", an excitation current setting for speed estimation i _{mag_1} , and a switch setting selection. The speed estimation excitation operation command, the pre-start excitation operation command, and the speed control operation command are provided to the input terminals of a three-input OR gate 21. The output signal of the OR gate 21 is input to the power converter 4 and the NOT gate 15 as a gate drive permission signal.

The two input terminals of the current regulator 9 are connected to the changeover switches 22 and 23, respectively. The movable contact of the changeover switch 22 is switched between the fixed contact AB and the fixed contact CD by the switch setting selection described above. The fixed contact AB is provided with the speed estimation excitation current setting i _{mag_1} , and the fixed contact CD is provided with the d-axis current target value i _{sd_set} as i _{mag_2} . The symbols A, B, C, and D attached to the fixed contacts represent the contact symbols of the switches selected in the processing of the control device shown in Figs. 9 and 13.

The movable contact of the changeover switch 23 is switched between a fixed contact ABC and a fixed contact D by switch setting selection. A zero value is applied to the fixed contact ABC, and an output signal of the speed control unit 20 is applied to the fixed contact D. The signals input to the current adjustment unit 9 via the changeover switches 22 and 23 are a D-axis current command i _{sd_ref} and a Q-axis current command i _{sq_ref} , respectively.

The Q-axis current command i _{sq_ref} is input to a slip calculator 24. In addition, a secondary magnetic flux command φ _{rd_ref} is input to the slip calculator 24. The slip calculator 24 calculates a slip frequency ω _slip of the induction motor 5 based on these input values and outputs the calculated value to an adder 25. A movable contact of a changeover switch 26 is connected to the adder 25, and the movable contact is switched to fixed contacts A, BC, and D by switch setting selection. A zero value is applied to the fixed contact A, a speed command initial value ω _{re_ref_Init} is applied to the fixed contact BC, and a speed estimation value ω _{re_est} is applied to the fixed contact D.

The sum of the adder 25 is input to the integrator 27 as the output frequency ω _stat . The integration result of the integrator 27 is input to the UVW/dq converter 8 and the dq/UVW converter 28 as the output voltage phase θ _{re_c} . The current regulator 9 receives D- and Q-axis currents i _d and i _q from the UVW/dq converter 8. The current regulator 9 generates a compensation voltage by, for example, performing a proportional integral calculation on the difference between the input current command value and the currents i _d and i _q , and outputs the compensation voltage to the voltage calculator 29. The voltage calculator 29 generates voltage commands v _{sd_ref} and v _{sq_ref} based on the compensation voltage and the induction motor equation, and outputs the voltage commands to the dq/UVW converter 28. The dq/UVW converter 28 converts the input voltage command into a three-phase voltage command, and generates a PWM signal to output to the power converter 4.

2 is a functional block diagram showing the internal configuration of the convergence calculation processing unit 12. The interphase voltages v _uv , v _wv are converted into three-phase voltages by a UV,WV/UVW converter 31, and are further converted into a d-axis voltage v _dc and a q-axis voltage v _qc via a UVW/αβ converter 32 and an αβ/dq converter 33. These voltages v _dc , v _qc are input to an error calculator 34 and a square sum calculator 35. The error calculator 34 calculates a phase error θ _err by equation (8). The phase error θ _err is given a sign (-) by an encoder 36, and is multiplied by proportional gains K _{P_PLL} , K _{I_PLL} by gain assigners 37P, 37I.

The output terminal of the gain provider 37P is connected to the START fixed contact of the changeover switch 38P, and the output terminal of the gain provider 37I is connected to the START fixed contact of the changeover switch 38I via the Δt provider 39. A zero value is provided to the STOP side fixed contact of the switch 38. The movable contact of the switch 38P is connected to one input terminal of the adder 41. The movable contact of the switch 38I is connected to the other input terminal of the adder 41 via another adder 40.

The output signal of the adder 40 is fed back to the sum input of the adder 40 via the unit delay element 42. The changeover switch 38 is switched by a convergence calculation execution command signal.

The output signal of the adder 41 becomes the estimated speed ω _{re_est_PLL} , and is input to the adder 44 via the Δt provider 43. The output signal of the adder 44 becomes the estimated phase θ _{re_est_PLL} , which is input to the αβ/dq converter 33, and is also output to the outside as the output voltage phase initial value θ _{re_Init} . The output signal of the adder 44 is fed back to the addition value input of the adder 44 via a unit delay element 45. The estimated speed ω _{re_est_PLL} is input to an input B of a divider 46, and is also output to the outside as a speed command initial value ω _{re_ref_Init} . The output signal of the square sum calculator 35 is input to an input A of the divider 46. The divider 46 outputs the division result A/B to the outside as a secondary magnetic flux command initial value φ _{rd_ref_Init} .

Next, a speed estimation method in this embodiment will be described. When vector control is performed on the rotor magnetic flux of an induction motor, if the direction of the magnetic flux is defined as the M axis and the phase 90 degrees ahead of the M axis is defined as the T axis, as shown in Fig. 3, then an induced voltage is generated in the T axis direction. The integral value of θ _re , which corresponds to the actual rotor speed (electrical angle), is set as the phase representing the M axis direction (position information). Since the M axis is the magnetic flux direction, it is set as θ _flux .

In order to estimate the speed and residual magnetic flux direction of a rotor having an initial speed, first, a DQ synchronous coordinate axis corresponding to the estimated MT synchronous coordinate axis is defined. Fig. 4 shows the relationship between the estimated DQ synchronous coordinate axis calculated in the controller and the MT synchronous coordinate axis in an actual motor. The DQ synchronous coordinate axis defines the currently recognized estimated speed as ω _{re_est_PLL} and the direction of the estimated phase θ _{re_est_PLL} corresponding to its time integral as the D axis.

The induced voltage is observed and converted to the DQ synchronous coordinate axis using the estimated phase θ _{re_est_PLL} . The D and Q axis components v _dc and v _qc are obtained as values of the DQ synchronous coordinate axis. When the MT synchronous axis and the DQ synchronous axis are aligned, only v _qc is obtained, so speed estimation should be performed so that v _dc becomes 0.

Next, a description will be given of the above-mentioned calculation process of the convergence calculation processing unit 12. The detection of the phase-to-phase voltages _vuv and _vwv involves detecting the potentials of the U and W phases with the V phase potential as a reference.

２相線間電圧（ｖ_ｕｖ，ｖ_ｗｖ）から３相静止軸上の電圧（ｖ_ｕ，ｖ_ｖ，ｖ_ｗ）には、以下の式で変換できる。

The two-phase line voltage ( _vuv , _vwv ) can be converted to the three-phase stationary axis voltage ( _vu , _vv , _vw ) using the following formula.

３相静止軸上の電圧から２相静止軸上の電圧（ｖ_α，ｖ_β）への変換は、以下の式による。

The three-phase stationary axis voltage is converted to the two-phase stationary axis voltage (v _α , v _β ) by the following equation.

The voltages (v _α , v _β ) are converted into voltage values (v _dc , v _qc ) on the synchronous coordinate axes by a conversion equation using position information θ _{re_est_PLL} corresponding to the time integral value of the estimated rotor speed ω _{re_est_PLL} .

The purpose of this estimation operation is to make the D-Q synchronous coordinate axis, which is the estimated M-T synchronous coordinate axis calculated in the calculator, coincide with the M-T synchronous coordinate axis in the actual motor. If there is no error between θ _{re_est_PLL} and θ _flux , only the Q-axis component of the induced voltage is observed. If the D-Q synchronous coordinate axis has an error θ _err in the leading direction from the M-T synchronous coordinate axis in the actual motor, a positive value is observed as the D-axis component in the controller. If the error between θ _{re_est_PLL} and θ _flux is small, the induced voltage can be calculated using the following approximate formula.

または、以下のように近似しても良い。

Alternatively, the following approximation may be used:

A PLL (Phase Locked Loop) tracking operation is performed by PI control as shown in Fig. 2. As shown in Fig. 4, since the estimated DQ synchronous coordinate axis is ahead of the M-T synchronous coordinate axis of the actual motor by θ _err , PI compensation is performed by changing the sign of θ _err to bring the estimated DQ synchronous coordinate axis closer to the M-T synchronous coordinate axis.

ＰＩ制御のゲインＫ_{Ｐ＿ＰＬＬ}、Ｋ_{Ｉ＿ＰＬＬ}は、以下のように設定する。
Ｋ_{Ｐ＿ＰＬＬ}＝２ζω_ＰＬＬ，Ｋ_{Ｉ＿ＰＬＬ}＝ω_ＰＬＬ ^２ …（１０）

The gains K _{P_PLL} and K _{I_PLL} of the PI control are set as follows.
K _{P_PLL} =2ζω _PLL , K _{I_PLL} =ω _PLL ² ...(10)

In addition, ζ is a damping ratio. The square sum calculator 35 calculates the magnitude of the detected voltage |v _dq | and transmits it to the phase-to-phase voltage level judgment unit 13. If it is lower than a predetermined voltage judgment value, it is judged that the speed estimation using the residual voltage cannot be stably performed, and it is judged as NG. In addition, the magnitude of the voltage may be calculated from the component on the αβ stationary axis.

When an induced voltage caused by residual magnetic flux generated in the induction motor is observed in the detected phase-to-phase voltages _vuv , _vwv , there is ideally a phase difference of 60 degrees between the two phase-to-phase voltages. When the amplitude of the induced voltage becomes small and there is no difference with the detected offset amount, or when the amplitude of the induced voltage becomes smaller than the magnitude of the noise, the phase difference becomes a value far removed from 60 degrees. When the phase-to-phase voltage phase difference determination unit 14 detects this state, it determines that speed estimation using the two-phase line voltages cannot be stably estimated, and judges it as NG.

5 shows an outline of a method for detecting a phase difference implemented by the interphase voltage phase difference determination unit 14. When the polarity of the WV phase is switched, CounterVal is reset.
CNT_A: Time difference between the rising timing between WV and UV, and CNT_B: Time difference between the rising timing between UV and the falling timing between WV. If the initial value of the sign recognition is made to match the sign of the detected voltage value and the estimation operation is started, CounterVal is reset to 0 at the first polarity change of the WV phase as shown in FIG. 5, and the count value up to the second polarity change is stored in CNT_A.

However, at this time, the phase Phase cannot be calculated correctly because CNT_B has not yet been saved and is set to the initial value 0. At the third polarity switch, "CNT_(A+B)" equivalent to "CNT_A+CNT_B" is saved, and CNT_B can be calculated using the previously obtained CNT_A.
CNT_B=CNT_(A+B)-CNT_A
Therefore, the phase from the second polarity change to the third polarity change (between the rising edge of UV and the falling edge of WV) is calculated by the following formula.
Phase=180×CNT_B/CNT_(A+B)

At the fourth polarity change, if CNT_B obtained at the third polarity change is used, CNT_(A+B) can be obtained by the following formula:
CNT_(A+B)=CNT_A+CNT_B
The phase from the third polarity change to the fourth polarity change, that is, the phase from the falling edge of UV to the rising edge of WV, is calculated by the following formula.
Phase=180×CNT_A/CNT_(A+B)
By repeating the above process, the phase difference can be correctly recognized from the third polarity change in the figure. The operation alternates between two values, such as 120 [deg] at the third change, 60 [deg] at the fourth change, and 120 [deg] at the fifth change.

Next, we will explain the principle by which secondary magnetic flux is generated by DC excitation of a motor in a free-running state. The mathematical model of an induction motor can be expressed as follows.

ｉ_ｓｄ［Ａ］：一次電流のＤ軸成分 Ｒ_ｓ［Ω］：一次抵抗
ｉ_ｓｑ［Ａ］：一次電流のＱ軸成分 σＬ_ｓ［Ｈ］：一次インダクタンス
φ_ｒｄ［Ｗｂ］：二次磁束のＤ軸成分 Ｍ［Ｈ］：相互インダクタンス
φ_ｒｑ［Ｗｂ］：二次磁束のＱ軸成分 Ｌ_ｒ［Ｈ］：二次インダクタンス

i _sd [A]: D-axis component of primary current R _s [Ω]: Primary resistance i _sq [A]: Q-axis component of primary current σL _s [H]: Primary inductance φ _rd [Wb]: D-axis component of secondary magnetic flux M [H]: Mutual inductance φ _rq [Wb]: Q-axis component of secondary magnetic flux L _r [H]: Secondary inductance

When performing DC excitation, current control is performed on the DQ synchronous axis with the control phase θ _re =0, i.e., a fixed phase. When the rotor is stopped, ω _re =0, and the command value of the applied current is a DC amount given in steps, so the primary frequency ω _stat , which is the output frequency, is also considered to be ω _stat =0, and when ω _re =ω _stat =0, the slip frequency is also treated as 0. In addition, since the response of the current controller is set to be sufficiently faster than the secondary time constant, it can be considered that a voltage command is given so that the current value is the same as the current command value. In other words, the command current i _sd is constant.

Considering the above, the equation for an induction motor when DC excitation is performed can be expressed as follows.

直流励磁実施時の二次磁束の挙動を求めるために、３行目と４行目を書き出し整理すると、

In order to determine the behavior of the secondary magnetic flux when DC excitation is performed, the third and fourth lines are written out and rearranged as follows:

The current value is given to only the D-axis as a constant value i _sd in a stepped manner, sufficiently quickly with respect to the change in the secondary magnetic flux, by current control. When a Laplace transform is performed on the D-axis component equation (13) with the secondary magnetic flux initial value φ _rd (0) = 0, the following is obtained.

これを整理すると（１６）式となり、（１６）式を逆ラプラス変換すると（１７）式となる。

Rearranging this gives equation (16), which is then inversely Laplace transformed to give equation (17).

From this, it can be seen that by applying current to the D-axis, the secondary magnetic flux proportional to the current value becomes the final value, and rises as a first-order lag response of the secondary time constant. When a current value equivalent to the no-load current is given as the current command, the secondary magnetic flux becomes roughly equivalent to the rated magnetic flux. As for the Q-axis component, since the Q-axis current is controlled to 0, φ _rq = 0. If the voltage command value at this time is rearranged from the first and second lines, the D-axis voltage command value is as follows. Note that the command value on the Q-axis side is omitted since _isq = 0.

From this, it can be seen that until the secondary magnetic flux φ _rd reaches the value of Mi _sd , the second term is a positive value, and therefore a voltage value greater than the voltage drop R _s i _sd due to the winding resistance is applied as the voltage command.

Next, we will show the case where the rotor has an initial speed. Since the rotor is not stopped, ω _re ≠ 0. Since the command value of the applied current is a DC amount given in steps, the primary frequency, which is the output frequency, is considered to be ω _S = 0, and the slip frequency ω _slip is considered to have an absolute value equal to the rotor speed and an opposite polarity. In other words, ω _slip = -ω _re . Also, just as when the rotor is stopped, the response of the current controller is set to be sufficiently faster than the secondary time constant, so it can be considered that a voltage command is given so that the current value is the same as the current command value. In other words, the command current i _sd is constant.

Considering the above, the induction motor equation can be expressed as follows:

磁束の挙動を求めるために、３行目と４行目を書き出して整理すると、

To determine the behavior of the magnetic flux, we write out and rearrange the third and fourth lines, as follows:

Laplace transform is performed on equation (22). If the current value is given to only the D axis as a constant value i _sd in an approximately stepped manner by current control in response to the change in the secondary magnetic flux, the rotation speed is also constant, and the initial value of the secondary magnetic flux is 0, then equation (22) can be rearranged as follows:

これを具体的に計算すると、以下のようになる。

The specific calculation of this is as follows:

ラプラス逆変換を行うことで、電流印加後の磁束の挙動を得ることができる。電流は電流制御されているので一定値とする。

By performing an inverse Laplace transform, we can obtain the behavior of the magnetic flux after the current is applied. Since the current is controlled, it is assumed to be a constant value.

Ｑ軸電流ｉ_ｓｑは０に制御されているので、生成される二次磁束は以下のように示すことができる。

Since the Q-axis current i _sq is controlled to zero, the secondary flux generated can be shown as follows:

（２５）式は、初速ω_ｒｅで回転子が回転している誘導電動機に対し、直流を印加開始から時刻ｔ［ｓ］経過した時点で生じている磁束を表している。

Equation (25) represents the magnetic flux generated at time t [s] after the start of application of direct current to an induction motor whose rotor is rotating at an initial speed ω _re .

If current application ends at time t = t1, the secondary magnetic flux generated in the rotor at that time is expressed as follows:

磁束量を具体的に算出するため、（２６）式をオイラーの公式を用いて極座標に書き換える。

In order to specifically calculate the amount of magnetic flux, equation (26) is rewritten into polar coordinates using Euler's formula.

（２７）式を計算すると、

Calculating equation (27), we get

（２７）式と（２８）式より、（２６）式は以下のように表現できる。

From equations (27) and (28), equation (26) can be expressed as follows:

時間関数部と、回転子速度依存で変化するゲインと位相に分けて表現すると、以下のようになる。

If it is expressed separately as a time function portion and gain and phase that change depending on the rotor speed, it becomes as follows.

Although the rotor speed is an unknown quantity, the secondary time constant can be obtained by offline tuning, etc., so it is desirable to determine the time for applying the current depending on the secondary time constant. Considering the behavior of the magnetic flux when the rotor is stopped, a guide for the application time can be obtained by setting ω _re =0 in equation (30),

となり、（３１）式より、以下のように求められる。

From equation (31), the following can be obtained:

This coincides with equation (17), which is the behavior of the magnetic flux when the rotor is stopped. It can be seen that the level of magnetic flux generated when the rotor speed is zero can be adjusted by the duration of application and the value of the applied current. These amounts are preset.
Next, a specific method for calculating the initial value of the voltage command value will be described. From equation (11), if only the constant-state term is included and the voltage is written on the left side,

電動機がフリーラン中は電流が流れないことに留意し、成分で表すと、

Keeping in mind that no current flows when the motor is coasting, the components are as follows:

検出電圧のＤＱ軸上の値の２乗和を計算すると、

Calculating the sum of the squares of the detection voltages on the DQ axes gives us the following:

これより、残留二次磁束の大きさは、以下のように求められる。

From this, the magnitude of the residual secondary magnetic flux can be calculated as follows.

電動機に与える理想電圧は、電動機の式から定常状態として算出し、ＤＱ軸の成分で表すと以下のようになる。

The ideal voltage to be applied to the motor is calculated from the motor equation in a steady state and expressed in terms of DQ axis components as follows:

（３５）、（３６）式で表される電圧指令値を、再起動時においては、具体的に以下の条件として算出する。
・条件１：速度制御として再起動する場合を想定し、初期トルク指令を０として、トルク電流であるＱ軸電流ｉ_ｓｑの初期値は０と考える。
・条件２：残留二次磁束は定格磁束より低い状態から再起動させることとする。すなわち、励磁電流ｉ_ｓｄは、残留二次磁束と同等の磁束を生成する励磁電流となるように初期値を設定する。

At the time of restart, the voltage command values expressed by the formulas (35) and (36) are calculated under the following conditions:
Condition 1: Assuming a case where restart is performed under speed control, the initial torque command is set to 0, and the initial value of the Q-axis current i _{sq ,} which is the torque current, is considered to be 0.
Condition 2: The motor is restarted from a state in which the residual secondary magnetic flux is lower than the rated magnetic flux. In other words, the initial value of the excitation current i _sd is set to be an excitation current that generates a magnetic flux equivalent to the residual secondary magnetic flux.

When restarting a motor in a free-running state, as a result of a convergence calculation process using the induced voltage due to the residual secondary magnetic flux, an initial value ω _{re_est_Init} of the rotor speed command, which is the initial value of the rotor angular speed ω _{re_est} , and a secondary magnetic flux phase θ _{re_Init} , which is the initial value of the output voltage phase θ _{re_c,} are obtained.

The magnitude of the induced voltage is found from the detected voltage value, and ω _{re_ref_Init} is used as the initial value of the rotor speed command, so that the initial value φ _{rd_ref_Init} of the secondary magnetic flux command value can be obtained as follows.

回転速度が十分に速い時は、以下のように近似してもよい。

When the rotation speed is sufficiently fast, the following approximation can be used:

As for the current value to be applied at startup, if the secondary magnetic flux can be tracked to the D-axis side by convergence calculation processing, the D-axis current i _{sd_ref_Init} that generates the equivalent of the residual secondary magnetic flux can be calculated as follows.
i _{sd_ref_Init} =φ _{rd_ref_Init} /M…(39)

Using these values, the output voltage command value at the time of restart can be calculated as follows: Since the torque command immediately after start-up is set to 0, the torque current i _sq is 0, and the slip frequency is also 0. Considering that the output frequency ω _stat is the same as the speed command value initial value ω _{re_ref_Init} , the output frequency ω stat can be given as follows.

After the initial value is given, if the D-axis current is controlled by the magnetic flux equivalent to the rated magnetic flux of the motor, unnecessary vibration elements can be eliminated by giving a D-axis current of a level that generates the rated magnetic flux as a current command i _{sd_set_Filt} that has passed through a filter equivalent to a leakage time constant T _leak . The initial value of the filter immediately after restarting can be set to i _{sd_ref_Init} shown above. Note that _Tc is the control period.

Regarding the command value of the secondary magnetic flux, since the magnetic flux is not set as the target magnetic flux at the time of restart, in the case where the behavior is such that the rated magnetic flux is gradually reached at a value equivalent to the secondary time constant T _r , unnecessary vibrations are not induced if the secondary magnetic flux command φ _{rd_set_Filt} passed through a filter equivalent to the secondary time constant T _r is used as the magnetic flux command value. The initial value of the filter immediately after restart can be set to φ _{rd_ref_Init} shown above.

A voltage command is calculated from the secondary magnetic flux command and the D-axis current command thus obtained. The value equivalent to the rotor speed may be ω _{re_est} obtained from the speed estimated value in the sensorless vector control. As its initial value, the speed command initial value ω _{re_ref_Init} , which is the final value of the estimated speed ω _{re_est_PLL} in the convergence calculation processing unit, may be used. Furthermore, when a Q-axis current command is given from a torque command obtained by speed control, the component of the Q-axis current command is also added in the same way as the D-axis current. When the Q-axis current is generated, the primary frequency must be set to ω _stat , which is a value to which the slip frequency has been added.

Figure 7 shows an example of restarting the motor by giving the initial voltage command value and the subsequent voltage command as described above. It can be seen that the torque current is not disturbed at all and the motor is restarted without any shock.

Next, we will explain how to set the current application time. To find the behavior of the secondary magnetic flux when DC excitation is performed, we write out the third line from equation (25):

This formula is the same as a first-order lag filter with a time constant of T _r and with Mi _sd as an input.

The change in magnetic flux when current regulator 9 controls i _sd to a constant value can be obtained by the following equation: When expressed in discrete time with T _c as the control period, this becomes equivalent to equation (43).

As shown in FIG. 8, the initial value of φ _{rd_set_Filt} may be equal to the value shown in equation (38). The secondary magnetic flux φ _{rd_set_Filt} shown in equation (48) is calculated while applying _{i sd_set} , and the time T _{mag_2} until the value reaches the secondary magnetic flux command value φ _{rd_set} is applied, i _{sd_set} . If the applied current i _{sd_set} is set to the equivalent of the no-load current, the increase in the secondary magnetic flux requires a time equivalent to the secondary time constant. However, if it is made larger than the no-load current, the time until the magnetic flux reaches the target value can be shortened, and the delay until start-up can be shortened.

Next, the operation of this embodiment will be described. The flowchart shown in FIG. 9 shows the processing in the restart processing unit when the operation command is switched from OFF to ON and a request for restart is received. The flowchart shown in FIG. 13 shows the processing in the restart processing unit when the operation command is switched from ON to OFF and the system is in a free run state, i.e., the power converter output is OFF.

In these flowcharts, there are two types of excitation operation for speed estimation. "Speed estimation excitation operation Type 2" operates so that speed estimation can be performed again by automatically performing excitation operation for speed estimation when it is detected that the phase-to-phase voltage is smaller than the threshold value or the phase difference of the phase-to-phase voltage is smaller than the threshold value in the free-run state after the operation command is turned OFF, and it is determined that speed estimation cannot be performed. On the other hand, "speed estimation excitation operation Type 1" does not have the function of Type 2, and when it is determined that speed estimation cannot be performed, it does not automatically perform excitation operation, ends speed estimation, and enters a standby state for the operation command to be turned ON. The excitation operation is performed after the operation command is turned ON.

The flow when the operation command is switched from OFF to ON will be described with reference to Fig. 9. In the initial state, all commands output by the restart processing unit 19 are OFF. First, the restart processing unit 19 determines whether or not the following condition 1 is satisfied (S1).
{(Pre-startup speed estimation operation==Type1)or
(pre-start speed estimation operation==Type2) and (speed estimation operation counter==0)}
If condition 1 is not satisfied (no), the process proceeds to step S13, and if condition 1 is satisfied (yes), it is determined whether or not the following condition 2 is satisfied (S2).
{(Speed estimation excitation operation == enabled) and
(Interphase voltage level determination = NG) or (Interphase voltage phase difference determination = NG)
Note that NG indicates that each detection is smaller than the threshold.

If condition 2 is not met (no), the process proceeds to step S8, and if condition 2 is met (yes), the following process A1 is executed (S3).
Switch setting selection: A
Excitation time setting for speed estimation: Timer=T _{mag_1}
Excitation current setting for speed estimation: i _{sd_ref} =I _{mag_1} (i _{sq_ref} =0)
Excitation operation command for speed estimation: ON
Then, current application processing is performed (S5) until the Timer, which has been decremented by Δt in step S6, becomes zero (S4; yes).

When the timer reaches zero (S4; no), the speed estimation excitation operation command is turned OFF as process A2 (S7), and then the following process B1 is executed (S8).
Speed estimation time setting: Timer = T _spdest
Speed estimation operation command: ON
In subsequent steps S9 to S11, the speed estimation processing unit 10 performs speed estimation processing (S10) in the same manner as in steps S4 to S6.

When the timer reaches zero (S9; no), the following process B2 is executed (S12).
Speed command initialization: ω _{re_ref_Init1} = ω _{re_ref_Init}
Control axis phase initialization: θ _{re_Init1} = θ _{re_Init1}
Secondary magnetic flux calculation initialization: φ _{r_ref_Init1} = φ _{r_ref_Init1}
Speed command initialization: ω _{re_ref} = ω _{re_ref_Init1}
Control axis phase initialization: θ _{re_c} = θ _{re_Init1}
Secondary magnetic flux calculation initialization: φ _{r_ref} = φ _{r_ref_Init1}
Speed estimation operation command: OFF

In step S13, the following process B3 is executed. Note that this process is executed when the determination in step S1 is "no," so that the pre-start speed estimation operation corresponds to Type 2.
Speed command initialization: ω _{re_ref} = ω _{re_ref_Init2}
Control axis phase initialization: θ _{re_c} = θ _{re_Init2}
Secondary magnetic flux calculation initialization: φ _{r_ref} = φ _{r_ref_Init2}
After step S12 or S13 is executed, the following condition 3 is judged (S14).
{(initial speed command value < threshold value) or (pre-start excitation operation == enabled)}

If condition 3 is not satisfied (no), the process proceeds to step S20, and if it is satisfied (yes), the following process C1 is performed (S15).
Switch setting selection: C
(At startup) Excitation time setting: Timer = T _{mag_2}
(At startup) Excitation current setting: i _{sd_ref} = I _{mag_2}
Pre-start excitation operation command: ON

The following steps S16 to S18 are the same as steps S4 to S6. Then, when the timer reaches zero (S16; no), the pre-start excitation operation command is turned OFF (S19) as process C2. Then, the switch setting selection is set to D and the speed control operation command is turned ON to start speed control (S20).

Next, a flow when the operation command is switched from ON to OFF will be described with reference to Fig. 13. In the initial state, all commands output by the restart processing unit 19 are OFF. The restart processing unit 19 determines whether or not the following condition 1 is satisfied (S21).
(Pre-startup speed estimation operation ==Type1 or Type2)
If condition 1 is not met (no), the process proceeds to step S36, and if condition 1 is met (yes), the following process A1 is executed (S22).
Switch setting selection: A
Speed estimation excitation time setting: Timer = T _spdest
Speed estimation operation counter: Count = 0
Speed estimation operation command: ON

Then, while the operation command is OFF (S23; yes), the speed estimation processing unit 10 performs the speed estimation process (S25) until the Timer, which has been decremented by Δt in step S26, becomes zero (S24; yes).
When the timer reaches zero (S24; no), the following process A2 is executed (S27).
Speed estimation operation counter: Count+=1
Speed estimation excitation setting: Timer = Δt

Then, it is determined whether or not the following condition 2 is satisfied (S28).
{(Interphase voltage level determination = NG) or (Interphase voltage phase difference determination = NG)}
If condition 2 is not satisfied (no), the process returns to step S23. If condition 2 is satisfied (yes), it is determined whether or not the following condition 3 is satisfied (S29).
{(Speed estimation excitation operation==Enabled) and (Pre-start speed estimation operation==Type2)}

If condition 3 is not satisfied (no), the process proceeds to step S36, and if it is satisfied (yes), the following process B1 is performed (S30).
Switch selection setting: B
Speed estimation operation command: OFF
Excitation time setting for speed estimation: Timer=T _{mag_1}
Excitation current setting for speed estimation: i _{sd_ref} =I _{mag_1}
Excitation operation command for speed estimation: ON

In the following steps S31 to S33, a current application process is performed similarly to steps S4 to S6 (S32). When the timer reaches zero (S31; no), the following process B2 is executed (S34).
Excitation operation command for speed estimation: OFF
Speed estimation excitation time setting: Timer = T _spdest
Speed estimation operation command: ON
Then, return to step S23.

When the operation command is turned ON (S23; no), the following process C1 is executed (S35) and then the process ends.
Speed command initialization: ω _{re_ref_Init2} = ω _{re_ref_Init}
Control axis phase initialization: θ _{re_Init2} = θ _{re_Init1}
Secondary magnetic flux calculation initialization: φ _{r_ref_Init2} = φ _{r_ref_Init1}
Speed estimation operation command: OFF

In step S36, the following process C2 is executed, and then the process ends.
Speed command initialization: ω _{re_ref_Init2} = 0
Control axis phase initialization: θ _{re_Init2} = 0
Secondary magnetic flux calculation initialization: φ _{r_ref_Init2} = 0
Speed estimation operation command: OFF

Figures 10 to 12 show an example of operation when using Type 1 excitation operation for speed estimation. Figure 10 is a timing chart showing a case where pre-start speed estimation operation: Type 1, excitation operation for speed estimation: enabled, and pre-start excitation operation: disabled, and the phase-to-phase voltage level determination is NG, while Figure 11 is a timing chart showing a case where pre-start speed estimation operation: Type 1, excitation operation for speed estimation: enabled, and pre-start excitation operation: enabled. Also, Figure 12 shows a case where the phase-to-phase voltage phase difference determination is NG in the same case as Figure 10.

As described above, according to this embodiment, the power converter 4 converts the DC voltage into a three-phase AC voltage with variable voltage and variable frequency and supplies it to the induction motor 5. The phase-to-phase voltage detection unit 11 detects the phase-to-phase voltages v _uv and v _wv of the induction motor 5, and the current detection unit 7 detects the phase currents i _u and i _w of the induction motor 5. The current adjustment unit 9 calculates the voltage command values v sd_ref and v _{sq_ref} so that the components obtained by converting the detection values of the respective phase currents onto the D-Q _synchronous coordinate axes match the current command values i _{sd_ref} and i _{sq_ref} .

The control device 6 is set to constantly perform convergence calculation processing when a request to stop free-run operation is input while the induction motor 5 is in operation, or when the main circuit potential of the power converter 4 that drives the induction motor 5 drops and it is determined that operation is difficult and the control device transitions to a free-run state. Then, when an operation command is input again or the main circuit potential is restored, the control device 6 restarts control of the induction motor using the rotation speed and direction of rotation and the phase of the residual secondary magnetic flux obtained from the convergence calculation processing at that time.

In other words, if the main circuit potential cannot be maintained due to a free-run stop request from the upper controller or a momentary power outage, the power converter 4 stops and starts the convergence calculation process when it transitions to a free-run state. By continuing to perform the convergence calculation during free-running, no matter when an operation command is input or when the momentary power outage is resolved, the rotation speed estimated at that time is used as the speed command, making it possible to immediately restart the system.

In the speed estimation excitation operation Type 1, when it is determined that speed estimation cannot be performed, the speed estimation is terminated and the operation command ON state is entered into a standby state. When the operation command ON is received, the control device 6 performs a speed estimation excitation operation in which a step-like current command value of a predetermined magnitude is given to the current adjustment unit _{9 for a predetermined time when starting the induction motor 5 from a state in which the initial speed is generated, and after cutting off the output of the power converter 4, performs a convergence calculation process to obtain the rotation speed and rotation direction of the induction motor 5 and the phase of the residual secondary magnetic flux using the interphase voltages vuv and} vwv, and restarts the control of the induction motor 5. That is, after performing excitation for speed estimation to bring the residual voltage to a level at which it can be estimated, speed estimation is performed and the induction motor 5 is restarted.

In this configuration, in a configuration in which the phase-to-phase voltage detection unit 11 is placed on the output side of the power converter 4, speed estimation can be performed even when the residual voltage is attenuated, minimizing shocks that occur during restart, improving noise resistance, and shortening the time from receiving an operation command to actually restarting the system.

The control device 6 also performs an excitation operation for speed estimation on the condition that the peak value or effective value of the phase-to-phase voltage detected by the phase-to-phase voltage detection unit 11 is determined to be smaller than a predetermined judgment value, and performs a convergence calculation process after cutting off the output of the power converter 4, and restarts the control of the induction motor 5. In other words, when the detected phase-to-phase voltage is smaller than the judgment value, excitation for speed estimation is performed, thereby making it possible to appropriately perform speed estimation using voltage detection.

In addition, when the phase difference between the two phase voltages is determined to be smaller than the determination value by the phase difference determination unit 14, the control device 6 also performs the excitation operation for speed estimation and similarly restarts the control of the induction motor 5, so that the speed estimation can be performed appropriately in this case as well.

Furthermore, the control device 6 sets the initial value ω _{re_ref_Init} of the speed command value from the rotation speed ω _{re_est_PLL} and the rotation direction in the convergence calculation processing unit 12, rotates the DQ synchronous coordinate axis, in which the phase of the residual secondary magnetic flux obtained by the convergence calculation processing is set as the D axis and the phase 90 degrees ahead of the D axis is set as the Q axis, at a frequency based on the rotation speed ω _{re_est_PLL} , and determines the initial values of the voltage commands v _sd and v _sq using the initial value φ _{re_ref_Init} of the secondary magnetic flux calculated from the interphase voltage value and the rotation speed just before the convergence calculation processing is completed. That is, the direction of the residual secondary magnetic flux, that is, the phase, is set as the D axis, the voltage command value v _sq of the Q axis is matched to the interphase voltage, and the initial value of the secondary magnetic flux used for the calculation of the torque and the voltage command value is also matched, and restarts. This makes it possible to suppress current disturbance at the time of restart.

The control device 6 determines the application time of the D-axis current command value based on the phase-to-phase voltage value immediately before the end of the convergence calculation process, the deviation between the initial value of the secondary magnetic flux and the command value of the secondary magnetic flux calculated from the rotation speed, and a D-axis current command value of a predetermined magnitude, and the current adjustment unit 9 provides a time for applying a current in the D-axis direction according to the D-axis current command value and its application time. The level of the magnetic flux applied for speed estimation is set to, for example, about 25% of the rated magnetic flux in order to minimize disturbance during application. At the time of restart, the magnetic flux level is further reduced, so if a large torque is required immediately after start-up, the magnetic flux is insufficient, so the torque is earned by the current value, and even if the torque command is small, the current may become excessive. Therefore, by setting the magnetic flux level at the time of restart to a state close to the rated state, the increase in current can be suppressed.

The control device 6 also adjusts the speed estimate so that the result of converting the interphase voltage onto a synchronous axis configured to use the time integral value of the speed estimate as the phase during coordinate transformation is the Q-axis component only.

Figure 14 is a timing chart showing the cases of pre-start speed estimation operation: Type 2, speed estimation excitation operation: enabled, and pre-start excitation operation: enabled. In pre-start speed estimation operation Type 2, when the induction motor 5 is in an operating state and a coast-run operation stop request is input, or when the main circuit potential of the power converter 4 driving the induction motor 5 drops and it is judged that operation is difficult and the motor transitions to a coast-run state, the control device 6 performs a convergence calculation process, but when it is judged that the peak value or effective value of the phase-to-phase voltage detected by the phase-to-phase voltage detection unit 11 is smaller than a predetermined judgment value, the convergence calculation process is stopped, and when the current adjustment unit 8 is caused to give a step-like current command value of a predetermined magnitude for a predetermined time, the convergence calculation process is resumed. As a result, when speed estimation is continued during coast-running, if the level of the phase-to-phase voltage becomes small, excitation for speed estimation is performed, and the residual magnetic flux can be maintained at a level at which speed estimation is possible.

Furthermore, when the control device 6 is constantly performing the convergence calculation process, if it is determined that the phase difference between the two phase-to-phase voltages detected by the phase-to-phase voltage detection unit 11 is smaller than a predetermined judgment value, it stops the convergence calculation process, and when it causes the current adjustment unit 8 to give a step-like current command value of a predetermined magnitude for a predetermined time, it resumes the convergence calculation process. As a result, even if the phase difference between the phase-to-phase voltages becomes small while continuing to perform speed estimation during free running, the residual magnetic flux can be maintained at a level that allows speed estimation.

Although several embodiments of the present invention have been described, these embodiments are presented as examples and are not intended to limit the scope of the invention. These novel embodiments can be embodied in various other forms, and various omissions, substitutions, and modifications can be made without departing from the gist of the invention. These embodiments and their modifications are included within the scope and gist of the invention, and are included in the scope of the invention and its equivalents as set forth in the claims.

In the drawing, 1 indicates a three-phase AC power supply, 2 indicates a full-wave rectifier circuit, 3 indicates a smoothing capacitor, 4 indicates a power converter, 5 indicates an induction motor, 6 indicates a control device, 7 indicates a current detection unit, 9 indicates a current adjustment unit, 10 indicates a speed estimation processing unit, 11 indicates a phase-to-phase voltage detection unit, 12 indicates a convergence calculation processing unit, 13 indicates a phase-to-phase voltage level determination unit, and 14 indicates a phase-to-phase voltage phase difference determination unit.

Claims (10)

a power converter that converts a DC voltage into a three-phase AC voltage with variable voltage and frequency and supplies the AC voltage to an induction motor;
A phase-to-phase voltage detection unit that detects phase-to-phase voltages between two or more phases of the induction motor;
A current detection unit that detects a phase current of the induction motor;
a current adjusting unit that calculates a voltage command value so that a component obtained by converting the detected value of the phase current onto a synchronous coordinate axis coincides with a current command value;
When starting the induction motor from an initial speed state,
A speed estimation excitation operation is performed to provide a step-like current command value having a predetermined magnitude to the current adjustment unit for a predetermined period of time;
and a restart control unit that, after cutting off the output of the power converter, performs convergence calculation processing to obtain the rotational speed and rotational direction of the induction motor and the phase of residual secondary magnetic flux using detection values of the phase-to-phase voltages between two or more phases, and restarts control of the induction motor. The motor control device according to claim 1, wherein the restart control unit performs the speed estimation excitation operation on the condition that the peak value or effective value of the phase-to-phase voltage detected by the phase-to-phase voltage detection unit is determined to be smaller than a predetermined determination value, and performs the convergence calculation process after cutting off the output of the power converter, thereby restarting control of the induction motor. a phase-to-phase voltage phase difference determination unit that calculates a phase difference between detection values of two or more phases of the phase-to-phase voltage detected by the phase-to-phase voltage detection unit and compares the phase difference with a predetermined determination value;
3. The motor control device according to claim 2, wherein the restart control unit performs the speed estimation excitation operation even when the phase-to-phase voltage phase difference determination unit determines that the phase difference is smaller than the determination value, and performs the convergence calculation process after cutting off the output of the power converter, thereby restarting control of the induction motor. a power converter that converts a DC voltage into a three-phase AC voltage with variable voltage and frequency and supplies the AC voltage to an induction motor;
A phase-to-phase voltage detection unit that detects phase-to-phase voltages between two or more phases of the induction motor;
A current detection unit that detects a phase current of the induction motor;
a current adjusting unit that calculates a voltage command value so that a component obtained by converting the detected value of the phase current onto a synchronous coordinate axis coincides with a current command value;
When a request to stop free-run operation is input while the induction motor is in an operating state, or when a main circuit potential of a power converter that drives the induction motor is reduced and it is determined that operation is difficult, and the induction motor transitions to a free-run state, the convergence calculation process is constantly performed;
and a restart control unit that, when an operation command is input again or the main circuit potential is restored, restarts control of the induction motor using the rotational speed and rotational direction and the phase of the residual secondary magnetic flux obtained from the convergence calculation processing at that time. The motor control device according to claim 4, wherein the restart control unit stops the convergence calculation process when it is determined that the peak value or effective value of the phase-to-phase voltage detected by the phase-to-phase voltage detection unit is smaller than a predetermined determination value while the convergence calculation process is being performed continuously, and resumes the convergence calculation process when it causes the current adjustment unit to provide a step-like current command value of a predetermined magnitude for a predetermined time. a phase-to-phase voltage phase difference determination unit that calculates a phase difference between detection values of two or more phases of the phase-to-phase voltage detected by the phase-to-phase voltage detection unit and compares the phase difference with a predetermined determination value;
The motor control device according to claim 5 , wherein the restart control unit also stops the convergence calculation process when the interphase voltage phase difference determination unit determines that the phase difference is smaller than the determination value. The restart control unit sets an initial value of a speed command value based on a rotation speed and a rotation direction,
a DQ synchronous coordinate axis, in which the phase of the residual secondary magnetic flux obtained by the convergence calculation process is defined as a D axis and a phase 90 degrees ahead of the D axis is defined as a Q axis, is rotated at a frequency based on the rotation speed;
7. The induction motor control device according to claim 1, wherein an initial value of a voltage command is determined using an initial value of a secondary magnetic flux calculated from the interphase voltage value and the rotation speed immediately before the convergence calculation process is completed. the restart control unit determines an application time of the D-axis current command value based on a deviation between an initial value of a secondary magnetic flux and a command value of a secondary magnetic flux calculated from the interphase voltage value and the rotation speed immediately before the end of the convergence calculation process, and a D-axis current command value of a predetermined magnitude;
8. The induction motor control device according to claim 7, wherein the current adjusting unit sets a time for applying a current in the D-axis direction in accordance with the D-axis current command value and its application time. The induction motor control device according to claim 7, wherein the restart control unit adjusts the speed estimation value so that the result of converting the interphase voltage onto a synchronous axis configured to use the time integral value of the speed estimation value as the phase during coordinate transformation is only the Q-axis component. The induction motor control device according to claim 8, wherein the restart control unit adjusts the speed estimation value so that the result of converting the interphase voltage onto a synchronous axis configured to use the time integral value of the speed estimation value as the phase during coordinate transformation is only the Q-axis component.

Images