JP7624726B2 - Antenna device and sandwich array device - Google Patents

Description

The present invention relates to an antenna device and a sandwich array device.

In recent years, there are high hopes for the practical application of flying cars as vehicles that can be used for commuting to work or school while avoiding city traffic jams, as a new means of transportation on remote islands or in mountainous areas, or for emergency transport or rapid transportation of goods in the event of a disaster. When flying cars are put to practical use, there is much interest in realizing communication between flying cars (flying vehicles).

For example, Patent Document 1 discloses a receiving antenna device that can detect the direction of arrival of a received radio wave in a horizontal plane. According to this, by using a circularly arranged phased array antenna consisting of antenna elements arranged at equal intervals on a circle and an antenna element arranged in the center, it is possible to realize an antenna device that can be constructed small and lightweight and can detect the direction of arrival of a received radio wave with high accuracy.

Japanese Patent Application Publication No. 59-114907

However, the technology disclosed in Patent Document 1 has the problem that although it can estimate the direction of arrival waves in the horizontal direction, it cannot estimate the direction of arrival waves in the elevation angle direction. In other words, since a flying car can freely change its flight direction in the horizontal and vertical directions, unlike conventional automobiles, the targets with which it communicates are not limited to the horizontal plane, and therefore an antenna with radiation characteristics that cover the entire solid angle is required.

The present invention has been made in consideration of the above-mentioned circumstances, and aims to provide an antenna device and a sandwich array device capable of high-speed communication by estimating the direction of arrival waves not only in the horizontal direction but also in the elevation direction.

In order to achieve the above object, an antenna device according to one embodiment of the present invention comprises a circularly arranged phased array antenna consisting of N first antenna elements (N is a natural number equal to or greater than 3) arranged at equal intervals on a circle and one second antenna element arranged approximately at the center of the circle, and a circular base plate whose radius is variable by switching control, and the first antenna elements and the second antenna element are composed of monopole antennas mounted on the base plate.

This configuration makes it possible to estimate the direction of incoming waves not only in the horizontal direction but also in the elevation direction, realizing an antenna device capable of high-speed communication.

Here, for example, the first antenna element and the second antenna element are configured as 1/4 wavelength monopole antennas.

In addition, for example, the antenna may be provided with a subarray control unit that divides the N first antenna elements into N/2 subarrays perpendicular to the direction of the incoming wave, which is a direction parallel to the ground plane of the incoming wave arriving at the circularly arranged phased array antenna, and controls the directivity of the subarrays to independently form the N/2 beams facing the direction of the incoming wave.

This configuration allows the directivity to be controlled based on the estimated direction of the incoming wave, which in turn allows the optimum received signal to be obtained at all times, enabling high-speed communication.

Here, each of the N/2 subarrays is composed of a combination of two antenna elements whose arrangement directions are parallel, and the combination may be changed depending on the direction of the arriving wave.

In addition, a directivity control unit is provided that controls the directivity of the circularly arranged phased array antenna within a predetermined angle range including the elevation angle direction by providing the two first antenna elements closest to the desired communication direction with a phase that is given to form delta directivity and sigma directivity calculated using the path difference in the elevation angle direction, which is the direction in which the incoming waves arriving at the circularly arranged phased array antenna make with the ground plane, and the delta directivity may be a directivity that reduces the radiation gain by realizing anti-phase excitation in the elevation angle direction, and the sigma directivity may be a directivity that increases the radiation gain by realizing in-phase excitation in the elevation angle direction.

This configuration allows directivity to be formed within an angular range in a specified direction, allowing communication only with the target.

The antenna may further include a calculation unit that performs statistical analysis on a first signal and a second signal obtained by multiplying the received signal of each of the first antenna elements by two weights corresponding to the arrangement of the first antenna elements and adding them up, and a third signal which is the received signal of the second antenna element, and calculates a phase difference between the first signal and the third signal, thereby estimating the direction of arrival of the incoming wave, which is the direction of the incoming wave arriving at the circularly arranged phased array antenna relative to the ground plane, and the calculation unit calculates the phase difference between the first signal and the third signal by averaging the phase angle of the voltage of the third signal and the phase angle of the voltage of the first signal, and the phase difference may be approximately proportional to the direction of arrival of the wave.

In order to achieve the above object, a sandwich array device according to one embodiment of the present invention comprises a first antenna device and a second antenna device which are two antenna devices of any of the above aspects, the first antenna device being arranged at the zenith part of the flying vehicle and the second antenna device being arranged at the nadir part of the flying vehicle, and the erection directions of the circularly arranged phased array antennas of the first antenna device and the second antenna device as viewed from the ground plate are opposite to each other.

These general or specific aspects may be realized by a system, method, integrated circuit, computer program, or computer-readable recording medium such as a CD-ROM, or by any combination of a system, method, integrated circuit, computer program, and recording medium.

The antenna device of the present invention can estimate the direction of arrival of waves not only in the horizontal direction but also in the elevation direction, enabling high-speed communication.

FIG. 1 is a diagram illustrating an example of a schematic configuration of an antenna device according to an embodiment. FIG. 2 is a diagram showing a specific configuration example of the circularly arranged phased array antenna shown in FIG. FIG. 3A is a diagram showing an example of the configuration of a base plate whose size is variable according to the embodiment. FIG. 3B is a diagram showing a specific configuration used to switch the size of the main plate shown in FIG. 3A. FIG. 4 is a diagram showing a specific example of the configuration of a sub-array when radio waves arrive from the +X-axis direction of a circularly arranged phased array antenna according to an embodiment. FIG. 5A is a diagram showing an example of a schematic configuration of a sandwich array device according to an embodiment. FIG. 5B is a diagram showing a use case of the sandwich array device shown in FIG. 5A. FIG. 6 is a diagram showing a concept of communication between flying vehicles using a sandwich array device according to an embodiment. FIG. 7A is a diagram showing the positional relationship in the XY plane of the first antenna element excited to form a monopulse directivity. FIG. 7B illustrates the elevation path difference used to design the delta and sigma beam patterns. FIG. 7C illustrates the angles used to determine the electrical length of the excited first antenna element shown in FIG. 7A. FIG. 8 is a diagram showing the relationship between the phase and elevation angle of the first antenna elements #1 and #8 required to form sigma and delta directivities. FIG. 9A is a diagram showing an example of sigma and delta directivities forming a monopulse directivity that realizes the space division monopulse method. FIG. 9B illustrates the concept of two pulses transmitted to form a monopulse directivity. FIG. 10A is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in directional scanning in the elevation angle direction according to an embodiment. FIG. 10B is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in directional scanning in the elevation angle direction according to an embodiment. FIG. 10C is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in directional scanning in the elevation angle direction according to an embodiment. FIG. 10D is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in directional scanning in the elevation angle direction according to an embodiment. FIG. 10E is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in directional scanning in the elevation angle direction according to an embodiment. FIG. 10F is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in directional scanning in the elevation angle direction according to an embodiment. FIG. 11A is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in horizontal directional scanning according to an embodiment. FIG. 11B is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in horizontal directional scanning in the embodiment. FIG. 11C is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in horizontal directional scanning in the embodiment. FIG. 11D is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in horizontal directional scanning in an embodiment. FIG. 11E is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in horizontal directional scanning in an embodiment. FIG. 11F is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in horizontal directional scanning in an embodiment. FIG. 11G is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in horizontal directional scanning in an embodiment. FIG. 11H is a diagram showing an example of an analysis result of sigma directivity and delta directivity of a circularly arranged phased array antenna formed in horizontal directional scanning in an embodiment. FIG. 12 is a diagram showing the voltage of a signal induced in a circularly arranged phased array antenna in the comparative example. FIG. 13A is a diagram showing directional characteristics in the XY plane of a first signal and a third signal in a comparative example. FIG. 13B is a diagram showing phase characteristics in the XY plane of the first signal and the third signal in the comparative example. FIG. 14 is a diagram showing two weighting functions of the first antenna element of the circularly arranged phased array antenna according to the embodiment. FIG. 15A shows the elevation angle directivity of the first signal obtained by performing electromagnetic field analysis using the method of moments. FIG. 15B shows the elevation angle directivity of the third signal obtained by performing electromagnetic field analysis using the method of moments. FIG. 16 is a schematic diagram for explaining the principle of the phase inversion weighting method. FIG. 17 is a diagram illustrating three-dimensional directivities of a first signal and a third signal in the embodiment. FIG. 18A shows the angular characteristics of the directivity and the estimated direction of arrival wave when the elevation angle is 30 degrees. FIG. 18B shows the angular characteristics of the directivity and the estimated direction of arrival wave when the elevation angle is 80 degrees. FIG. 19 is a diagram showing a change in transmission capacity with respect to the elevation angle of an incoming wave when the radius of the base plate is changed in the embodiment. FIG. 20A is a diagram showing three-dimensional radiation directivity of a dipole array antenna in a comparative example. FIG. 20B is a diagram showing three-dimensional radiation directivity of the monopole array antenna according to the embodiment. FIG. 21A is a diagram showing three-dimensional directivity when directional scanning is performed by the sandwich array device according to the embodiment. FIG. 21B is a diagram showing three-dimensional directivity when directional scanning is performed by the sandwich array device according to the embodiment. FIG. 21C is a diagram showing three-dimensional directivity when directional scanning is performed by the sandwich array device according to the embodiment. FIG. 21D is a diagram showing three-dimensional directivity when directional scanning is performed by the sandwich array device according to the embodiment. FIG. 22A is a diagram showing three-dimensional directivity when directional scanning is performed by the sandwich array device according to the embodiment. FIG. 22B is a diagram showing three-dimensional directivity when directional scanning is performed by the sandwich array device according to the embodiment. FIG. 22C is a diagram showing three-dimensional directivity when directional scanning is performed by the sandwich array device according to the embodiment. FIG. 22D is a diagram showing three-dimensional directivity when directional scanning is performed by the sandwich array device according to the embodiment. FIG. 23 is a diagram showing the elevation angle characteristics of the transmission capacity of a MIMO array antenna using a half-wavelength dipole array antenna in a comparative example. FIG. 24A is a diagram showing the azimuth characteristics of the transmission capacity of a MIMO array antenna using a sandwich array device according to an embodiment. FIG. 24B is a diagram showing the elevation angle characteristics of the transmission capacity of a MIMO array antenna using a sandwich array device according to an embodiment. FIG. 25 is a diagram showing the three-dimensional transmission capacity over the entire solid angle of a MIMO array antenna using half-wavelength dipole array antennas in a comparative example. FIG. 26 is a diagram showing the three-dimensional transmission capacity over the entire solid angle of a MIMO array antenna using a sandwich array device according to an embodiment. FIG. 27 is a diagram showing the communication outage probability over the entire solid angle of a MIMO array antenna using a sandwich array device according to an embodiment. FIG. 28 is a diagram showing an example of a multi-element MIMO array antenna according to a modified example. FIG. 29A is a diagram showing an example of a 4×4 MIMO array antenna in a modified example. FIG. 29B is a diagram showing an example of an 8×8 MIMO array antenna in a modified example. FIG. 29C is a diagram showing an example of a 16×16 MIMO array antenna in a modified example. FIG. 29D is a diagram showing an example of a 32×32 MIMO array antenna in a modified example. FIG. 29E is a diagram showing an example of a 64×64 MIMO array antenna in a modified example. FIG. 29F is a diagram showing an example of a 128×128 MIMO array antenna in a modified example. FIG. 29G is a diagram showing an example of a 256×256 MIMO array antenna in a modified example.

Note that each of the embodiments described below represents a specific example of the present invention. The numerical values, shapes, materials, components, component placement and connection forms, steps, and order of steps shown in the following embodiments are merely examples and are not intended to limit the present invention. Furthermore, among the components in the following embodiments, components that are not described in an independent claim that represents the highest concept are described as optional components.

(Embodiment)
[Configuration of Antenna Device]
FIG. 1 is a diagram showing an example of a schematic configuration of an antenna device 10 according to the present embodiment.

As shown in Fig. 1, the antenna device 10 in this embodiment includes a circularly arranged phased array antenna 1, a ground plate 2, a control unit 3, and a calculation unit 4. The antenna device 10 in this embodiment is mounted on, for example, a flying vehicle (also called a flying car). In the example shown in Fig. 1, the circularly arranged phased array antenna 1 has nine antenna elements. θ shown in Fig. 1 represents the elevation angle.

[Circularly arranged phased array antenna 1]
Fig. 2 is a diagram showing a specific configuration example of the circularly arranged phased array antenna 1 shown in Fig. 1. φ shown in Fig. 2 is a direction that intersects with the X-axis in a horizontal plane (XY plane), and represents the direction of an arriving wave (arrival wave direction) in the XY plane.

1 and 2, the circularly arranged phased array antenna 1 comprises N first antenna elements (N is a natural number equal to or greater than 3) arranged at equal intervals on a circle, and one second antenna element arranged approximately at the center of the circle. The first antenna element and the second antenna element are monopole antennas mounted on a ground plate 2 and configured on the ground plate 2. In this embodiment, the first antenna element and the second antenna element are 1/4 wavelength monopole antennas and configured on the ground plate 2 with a radius _rg .

As a result, the circularly arranged phased array antenna 1 not only has a substantially omnidirectional directional characteristic in the plane (horizontal plane) containing the circular area, but also has a directional characteristic in the zenith direction (the +Z axis direction in Figure 1).

Below, we will explain the case where N = 8, i.e., the circularly arranged phased array antenna 1 has eight first antenna elements.

For example, as shown in Fig. 2, a circularly arranged phased array antenna 1 is composed of eight antenna elements (first antenna elements: #1 to #8) arranged at 45 degree intervals on a circumference of a radius a, and one parasitic antenna element (second antenna element: #9) arranged at the center of the circle. Here, the radius a is, for example, 4.9 cm. Furthermore, these nine antenna elements are 1/4 wavelength monopole antennas and are constructed on a ground plate 2 of a radius _rg .

Then, by calculating the difference between the phase of the sum of the received signals of the eight first antenna elements and the phase of the received signal of the single second antenna element located in the center, the arrival direction of the received signal, which also has an angle in the elevation direction, can be estimated. The details of the estimation method will be described later, but the phase angle of the voltage of the signal induced in the second antenna element (#9) by the incoming received radio waves is synthesized using a phase inversion weighting method, which will be described later.

In addition, in the circularly arranged phased array antenna 1, the eight antenna elements (first antenna elements: #1 to #8) are divided and controlled (phase-differential power supply) into sub-arrays consisting of four pairs of elements so that they are orthogonal to the direction of the incoming wave according to the direction of the incoming wave by the control unit 3. This allows the circularly arranged phased array antenna 1 to operate as a directional scanning array antenna such as a MIMO (Multiple Input Multiple Output) array antenna.

In addition, the radiation directivity in the elevation direction of the circularly arranged phased array antenna 1 is optimized by switching control of the radius size of the ground plate 2. In other words, when the circularly arranged phased array antenna 1 is operated as a MIMO array antenna, the radius size of the ground plate 2 is switched and controlled to be different from that when estimating the direction of arrival waves.

Furthermore, in the circularly arranged phased array antenna 1, the directivity in the elevation angle direction is controlled within a predetermined angle range including the elevation angle direction by the control unit 3 using the space division monopulse method described below.

[Main plate 2]
The ground plane 2 is a circular ground plane whose radius _rg is variable by switching control. In this embodiment, the ground plane 2 is a planar (plate-like) conductor, for example as shown in Fig. 1. The reason why the size of the radius _rg of the ground plane 2 is made variable is because the optimal size of the ground plane 2 differs between the case where the circularly arranged phased array antenna 1 is used to estimate the direction of arrival waves and the case where the antenna is used as a MIMO array antenna.

Figure 3A is a diagram showing an example of the configuration of a base plate 2 whose size is variable in this embodiment. Figure 3B is a diagram showing a specific configuration used to switch the size of the base plate 2 shown in Figure 3A.

The radius _rg of the ground plate 2 is switched by the control unit 3. In this embodiment, as shown in FIG. 3A, the diode 301 is controlled to be turned on and off, and the radius _rg of the ground plate 2 is changed to, for example, 7 cm or 10 cm. More specifically, as shown in FIG. 3B, the diode 301 and the choke coil 302 may be connected to the ground plate 2, and a bias voltage _Vd of 5 V or 0 V may be applied to the diode 301 to control the size of the ground plate 2. In this case, for example, when a bias voltage _Vd of 5 V is applied, the diode 301 is turned on, and a current flows through the outer ground plate 2a, so that the radius _rg of the ground plate 2 is changed to 10 cm. Also, for example, when a bias voltage _Vd of 0 V is applied, the diode 301 is turned off, and no current flows through the outer ground plate 2a, and only the inner ground plate 2b is active, so that the radius _rg of the ground plate 2 is changed to 7 cm.

The radius _rg of the base plate 2 is not limited to being switched to either 7 cm or 10 cm, but may be switched to an appropriate size depending on the system to be realized.

[Control unit 3]
The control unit 3 has a subarray control function for dividing and controlling the circularly arranged phased array antenna 1 into subarrays. More specifically, the control unit 3 divides the N first antenna elements into N/2 subarrays orthogonal to the direction of the incoming wave, which is a direction parallel to the ground plane 2 of the incoming wave arriving at the circularly arranged phased array antenna 1. The control unit 3 then controls the directivity of the subarrays in this manner to independently form N/2 beams facing the direction of the incoming wave.

Here, each of the N/2 subarrays is composed of a combination of two antenna elements whose arrangement direction is parallel, and the combination is changed according to the direction of the incoming wave. There are N/2 combination patterns. The control unit 3 changes the combination according to the direction of the incoming wave by selecting, from the N/2 pattern combinations, a combination pattern of two antenna elements that are parallel to the direction of the incoming wave within a specified angle range.

Figure 4 is a diagram showing a specific example configuration of a sub-array when radio waves arrive from the +X-axis direction of the circularly arranged phased array antenna 1 in this embodiment.

For example, since the circularly arranged phased array antenna 1 includes eight first antenna elements, the control unit 3 selects one of four combination patterns according to the direction of the incoming wave. In the example shown in Figure 4, the combinations shown are the first antenna elements #2 and #3, the first antenna elements #1 and #4, the first antenna elements #5 and #8, and the first antenna elements #6 and #7 as subarrays (subarrays 1 to 4).

In general, in a cluster propagation environment, when the subarray arrangement is parallel to the direction of the incoming wave, the correlation of the received signals is high, whereas when it is orthogonal, the correlation of the received signals is low. The control unit 3 divides the eight first antenna elements of the circularly arranged phased array antenna 1 into four so that the combination of subarrays is arranged perpendicular to the direction of the incoming wave, so that the four received signals have low correlation. Furthermore, since the eight first antenna elements are arranged in a circle at 45 degree intervals in the circularly arranged phased array antenna 1, the control unit 3 can rotate the combination of subarrays every 45 degrees by switching the power supply of the subarrays according to the direction of the incoming wave.

The control unit 3 also has a full solid angle directivity control function that controls the directivity of the circularly arranged phased array antenna 1 within a predetermined angle range including the elevation angle direction by using the space division monopulse method. More specifically, the control unit 3 controls the directivity of the circularly arranged phased array antenna 1 within a predetermined angle range including the elevation angle direction by providing the phases given to form delta directivity and sigma directivity to the two first antenna elements closest to the desired communication direction. Here, the delta directivity is a directivity that reduces the radiation gain by realizing anti-phase excitation in the elevation angle direction, and the sigma directivity is a directivity that increases the radiation gain by realizing in-phase excitation in the elevation angle direction. The delta directivity and sigma directivity are calculated using the path difference in the elevation angle direction, which is the direction of the incoming wave arriving at the circularly arranged phased array antenna 1 and the ground plane 2. The space division monopulse method is a method of realizing the SLS function (Side Lobe Suppression) using sigma directivity and delta directivity. Details will be described later.

The control unit 3 also has a switching control function for controlling the radius of the ground plate 2. For example, as shown in Fig. 3B, the control unit 3 may control the radius of the ground plate 2 by controlling the bias voltage _Vd applied to the diode 301 to 0 V or 5 V.

[Calculation unit 4]
The calculation unit 4 has an arrival wave direction estimation function for estimating the arrival wave direction, which is the direction of the arrival wave arriving at the circular array phased array antenna 1. More specifically, the calculation unit 4 statistically analyzes a first signal and a second signal obtained by multiplying the received signals of the first antenna elements by two weights according to the arrangement of the first antenna elements and summing them, and a third signal which is a received signal of the second antenna element, and calculates a phase difference between the first signal and the third signal. Here, the phase difference is approximately proportional to the arrival wave direction. In this manner, the calculation unit 4 estimates the arrival wave direction, which also has an angle in the elevation angle direction of the arrival wave arriving at the circular array phased array antenna 1, by calculating the phase difference between the first signal and the third signal. In this embodiment, the calculation unit 4 calculates the phase angle of the voltage of the third signal by averaging the phase angle of the voltage of the first signal and the phase angle of the voltage of the second signal, thereby calculating the phase difference between the first signal and the third signal. Furthermore, in this embodiment, a calculation for finding an average value is performed as the statistical analysis, but it is not limited to the calculation for finding an average value, and it may be combined with a calculation performed to adapt to a multipath environment.

In this embodiment, each of the eight first antenna elements has two weights according to the arrangement of the first antenna elements, as shown in FIG. 2. Therefore, the calculation unit 4 obtains a first signal by multiplying each of the received signals of the eight first antenna elements by one weight (counterclockwise weight) of each of the eight first antenna elements and adding them up. The calculation unit 4 also obtains a second signal by multiplying each of the received signals of the eight first antenna elements by the other weight (clockwise weight) of each of the eight first antenna elements and adding them up. The calculation unit 4 also obtains a third signal from the second antenna element.

The calculation unit 4 can estimate the direction of arrival of the received signal, which also has an angle in the elevation direction, by calculating the difference between the phase of the voltage of the first signal and the phase of the voltage of the third signal. The phase of the voltage of the third signal is calculated by applying a phase inversion weighting method to combine, i.e., average, the phase of the voltage of the first signal and the phase of the voltage of the second signal.

[Effects, etc.]
As described above, the antenna device 10 of the present embodiment is configured with a circularly arranged phased array antenna 1 on the ground plate 2, which is composed of a plurality of first antenna elements arranged at equal intervals on a circle and one second antenna element arranged approximately at the center of the circle. The first antenna element and the second antenna element are monopole antennas mounted on the ground plate 2. As a result, the antenna device 10 of the present embodiment can estimate the direction of arrival of a wave not only in the horizontal direction but also in the elevation angle direction. In addition, the antenna device 10 of the present embodiment can control directivity with respect to the estimated direction of arrival of a wave. As a result, an optimal reception signal can be obtained at all times, and high-speed communication can be realized. Furthermore, the antenna device 10 of the present embodiment can control directivity to a predetermined angle range including the elevation angle direction, so that communication can be performed only with a target communication partner.

In addition, in the antenna device 10 of this embodiment, the second antenna element, which is a parasitic element arranged at the center of the circle of the circularly arranged phased array antenna 1, contributes to improving the gain by utilizing electromagnetic coupling. In addition, the radiation directivity in the elevation angle direction of the antenna device 10 of this embodiment is optimized by switching control of the size of the base plate 2.

The antenna device 10 of this embodiment has directional characteristics not only in the horizontal direction but also in the zenith direction (+Z-axis direction in Figure 1), but has no directional characteristics in the downward direction of the base plate 2 (-Z direction in Figure 1). Therefore, by arranging the antenna devices 10 in a sandwich shape above and below, it is possible to provide directivity over the entire solid angle. Below, we will explain the sandwich array device 100 in which the antenna devices 10 are arranged in a sandwich shape above and below.

[Sandwich array device 100]
FIG. 5A is a diagram showing an example of a schematic configuration of a sandwich array device 100 according to this embodiment.

As shown in Fig. 5A, the sandwich array device 100 includes two of the above-mentioned antenna devices 10, which are sandwiched vertically (in the Z-axis direction in Fig. 5A). More specifically, the sandwich array device 100 includes two antenna devices 10, a first antenna device 10A and a second antenna device 10B. The standing direction of the circularly arranged phased array antenna 1A as viewed from the ground plane 2A of the first antenna device 10A is opposite to the standing direction of the circularly arranged phased array antenna 1B as viewed from the ground plane 2B of the second antenna device 10B. This configuration allows the sandwich array device 100 to have radiation characteristics over the entire solid angle.

Figure 5B shows a use case of the sandwich array device 100 shown in Figure 5A.

As shown in FIG. 5B, the sandwich array device 100 is mounted on an air vehicle 300 as a use case. More specifically, the first antenna device 10A is arranged at the zenith of the air vehicle 300, and the second antenna device 10B is arranged at the nadir of the air vehicle 300. The erection direction of the circularly arranged phased array antenna 1A as viewed from the ground plate 2A of the first antenna device 10A is opposite to the erection direction of the circularly arranged phased array antenna 1B as viewed from the ground plate 2B of the second antenna device 10B.

The flying vehicle 300 is also called a flying car, and can freely change its flight direction in the horizontal and vertical directions (up and down). In order to perform highly reliable communication between the flying vehicles 300, it is necessary to achieve MIMO communication between the flying vehicles in any direction. In other words, the flying vehicle 300 needs an antenna with radiation characteristics over the entire solid angle. Therefore, by mounting the sandwich array device 100 on the flying vehicle 300, communication between the flying vehicles can be performed.

In the flying vehicle 300, communication between any upper and lower sides can be realized by switching the antenna devices 10A and 10B of the sandwich array device 100 according to the direction of arrival of the received signal. In addition, the radiation directivity in the elevation direction of the antenna devices 10A and 10B can be controlled by changing the size of the ground plate 2 and the ground plate 2B. Of course, for example, if the communication partner is fixed on the zenith side of the flying vehicle 300, the flying vehicle 300 may use only the upper antenna device 10A. Also, if the communication partner is fixed on the nadir side, the flying vehicle 300 may use only the lower antenna device 10B.

As described above, the flying vehicle 300 can estimate the direction of arrival of the received signal over the entire solid angle by mounting the sandwich array device 100, and can direct the directivity to the estimated direction of arrival, thereby realizing highly reliable communication between the flying vehicles. The antenna device 10 constituting the sandwich array device 100 can estimate the direction of arrival of the wave in a simple manner by processing in the RF section. More specifically, as shown in FIG. 5A, the eight first antenna elements constituting the circularly arranged phased array antennas 1A and 1B are arranged in a circle at 45 degree intervals. Therefore, the flying vehicle 300 can rotate the combination of the subarrays every 45 degrees by switching the power supply for dividing and controlling the eight first antenna elements into subarrays according to the direction of arrival of the wave. As a result, even if the flying vehicle 300 freely changes the flight direction up, down, left, and right in space, it is possible to appropriately select the combination of subarrays arranged perpendicular to the direction of arrival of the wave, thereby improving the transmission capacity.

Furthermore, the flying vehicle 300 can use the sandwich array device 100 to direct a beam in the estimated direction of the incoming wave of the received signal, making it possible to network with other flying vehicles and not only prevent collisions with other flying vehicles but also exchange high-definition video content.

In addition, the flying vehicle 300 can form a directivity that is spatially divided over the entire solid angle by forming a directivity while sequentially shifting (spatially dividing) the angular range in a predetermined direction using the sandwich array device 100. This allows the flying vehicle 300 to eliminate synchronous interference and communicate only with the target flying vehicle, even if there are multiple other flying vehicles with which it can communicate in the space.

Below, we explain the spatial division monopulse method, which is used to control directivity within an angular range in a specified direction.

<Space division monopulse method>
Fig. 6 is a diagram showing the concept of communication between flying vehicles using the sandwich array device 100 according to this embodiment. Fig. 6 shows four flying vehicles 300 that can freely change flight direction, and are referred to as flying vehicles 300a, 300b, 300c, and 300d, respectively. Each of the flying vehicles 300a to 300d is equipped with a sandwich array device 100 that is composed of antenna devices 10A and 10B.

As shown in FIG. 6, in order to communicate between multiple flying vehicles 300 (between flying vehicles), each sandwich array device 100 needs to have radiation characteristics over the entire solid angle. However, if there is synchronous interference, communication cannot be performed correctly, and there is a risk of collision between the multiple flying vehicles 300. In other words, for example, in order for flying vehicle 300a to communicate with flying vehicles 300b, 300c, and 300d, communication must be realized between flying vehicles 300b to 300d in any direction. On the other hand, if flying vehicle 300a communicates simultaneously (synchronously) with flying vehicles 300b, 300c, and 300d, it will not be able to distinguish between flying vehicles 300b, 300c, and 300d, and in the worst case scenario, there is a risk of collision. For this reason, a collision prevention function is required.

This collision prevention function is achieved in the sandwich array device 100, i.e., the antenna device 10, by using the space division monopulse method to sequentially shift (spatially divide) the angular range in a specified direction and control the directivity to that angular range in the specified direction.

Below, we will explain a method for calculating the phase that forms delta directivity and sigma directivity as a method for forming directivity within an angular range in a specified direction (hereinafter also referred to as a method for forming monopulse directivity).

Figure 7A shows the positional relationship in the XY plane of the first antenna element excited to form a monopulse directivity. Figure 7B shows the elevation path difference used to design the delta and sigma directivity. Figure 7C shows the angles used to determine the electrical length of the excited first antenna element shown in Figure 7A.

To form a monopulse directivity, for example, of the eight first antenna elements shown in Fig. 7A, four first antenna elements (#1, #2, #7, #8) are excited, and the remaining first antenna elements (#3, #4, #5, #6) are terminated at 50 Ω. In this case, due to the path difference in the elevation angle direction shown in Fig. 7B, the electrical length between first antenna element #1 and first antenna element #2, or the electrical length between first antenna element #7 and first antenna element #8 can be calculated by (Equation 1).

Here, the distance S shown in Fig. 7A, in other words, the distance S between the first antenna element #7 and the first antenna element #8 in the X direction, can be expressed using the radius a of the circularly arranged phased array antenna 1 shown in Fig. 7A and the angle δ shown in Fig. 7C. In other words, it can be expressed as S = a cos δ - a sin δ = a (cos δ - sin δ).

Furthermore, from the addition theorem of trigonometric functions, sin(α - β) = sin α cos β - cos α sin β. If α = π/4 and β = δ, then sin(π/4 - δ) = sin(π/4) cos δ - cos(π/4) sin δ = 1/√2(cos δ - sin δ).

Therefore, we can derive that cos δ - sin δ = √2 sin(π/4 - δ).

In addition, since the angle δ shown in Figure 7C is 22.5 degrees, the distance S can be transformed as follows: S = a(cos δ - sin δ) = √2a sin(π/4 - δ) = √2a sin(π/8).

On the other hand, as shown in Figure 7B, Δl = S · cos θ, which can be transformed to Δl = S · cos θ = √2a · sin(π/8) · cos θ. In this way, the above (Equation 1) can be obtained.

Here, since k = 2π/λ, the sigma directivity can be expressed by the following (Equation 2), and the delta directivity can be expressed by the following (Equation 3). As mentioned above, the sigma directivity is a directivity that increases the radiation gain by realizing in-phase excitation in the elevation angle direction, and the delta directivity is a directivity that decreases the radiation gain by realizing out-of-phase excitation in the elevation angle direction.

Fig. 8 is a diagram showing the relationship between the phase and elevation angle of the first antenna elements #1 and #8 required to form sigma directivity and delta directivity. In Fig. 8, the phase of the first antenna elements #2 and #7 is set to 0. In Fig. 8, the frequency used for the analysis is 2 GHz, the radius a of the circularly arranged phased array antenna 1 in this embodiment is 4.9 cm, and the radius _rg of the ground plate 2 is 7 cm.

The elevation angle is the angle defined as 0 degrees in the horizontal plane (XY plane in Figure 2), +90 degrees in the zenith direction (+Z axis direction in Figure 1), and -90 degrees in the nadir direction (-Z axis direction in Figure 1).

From FIG. 8, when the elevation angle is 45 degrees, φ ₁ =φ ₈ =-45 degrees for sigma directivity, and φ ₁ =φ ₈ =135 degrees for delta directivity. As shown in FIG. 1, FIG. 7A, and FIG. 7C, the eight first antenna elements constituting the circularly arranged phased array antenna 1 have a rotationally symmetric structure. For this reason, four first antenna elements (#1, #2, #3, #8) rotated by 45 degrees are excited, and the other first antenna elements (#4, #5, #6, #7) are terminated with 50Ω, and the phase that realizes in-phase excitation and anti-phase excitation in this case is calculated. In other words, the combination of the four first antenna elements to be excited is rotated by 45 degrees in the azimuth direction, and the phase that realizes in-phase excitation and anti-phase excitation is calculated each time, so that sigma directivity and delta directivity can be formed in all azimuth directions. The azimuth direction is an angle defined as +X-axis direction being 0 degrees and +Y-axis direction being 90 degrees in FIG. 2 or FIG. 7B, and is represented by φ in FIG.

Fig. 9A is a diagram showing an example of sigma directivity and delta directivity forming a monopulse directivity that realizes the space division monopulse method. Fig. 9B is a diagram showing the concept of two pulses _P1 and _P2 transmitted to form the monopulse directivity. As described above, the space division monopulse method is a method for realizing a side lobe suppression function by using sigma directivity and delta directivity.

As shown in Figure 9A, there is an area in the forward direction of the sigma directivity (upward in Figure 9A) where the directivity is 9 dB higher than the delta directivity over a certain angular range. Furthermore, the side lobes of the sigma directivity are completely encompassed by the delta directivity.

In TCAS II (Traffic Alert and Collision Avoidance System II), the interrogator transmits two pulses _P1 and _P2 to the surrounding target aircraft as shown in FIG. 9B. The pulses _P1 and _P2 are transmitted with the same power, with the pulse _P1 being transmitted with sigma directivity and the pulse _P2 being transmitted with delta directivity. The target aircraft responds by providing an ID code or the like only when the pulse _P2 is 9 dB smaller than the pulse _P1 . In other words, in TCAS II, only the transponders of the target aircraft present in this predetermined angle range respond. This allows the interrogator to communicate only with the target aircraft present in the predetermined angle range shown in FIG. 9A, thereby eliminating synchronous interference. Furthermore, in TCAS II, the interrogator selectively performs space division for communication, thereby enabling the interrogation-response sequence to be carried out over the entire space.

In this way, by using the space division monopulse method, it is possible to achieve side lobe suppression functionality similar to that of a large SSR antenna, even with a small antenna.

Figures 10A to 10F are diagrams showing an example of the analysis results of the sigma directivity and delta directivity of the circularly arranged phased array antenna 1 formed in directional scanning in the elevation angle direction in this embodiment. More specifically, Figures 10A to 10F are calculation results of the sigma directivity and delta directivity formed at an elevation angle represented by θ when the phase calculated using Figure 8 is given to the first antenna elements #1 and #8. Figure 10A is the calculation result of the sigma directivity and delta directivity formed at θ = 0 degrees, Figure 10B is the calculation result of θ = 15 degrees, Figure 10C is the calculation result of θ = 30 degrees, Figure 10D is the calculation result of θ = 45 degrees, Figure 10E is the calculation result of θ = 60 degrees, and Figure 10F is the calculation result of the sigma directivity and delta directivity formed at θ = 75 degrees. Also, Figures 10A to 10F show a predetermined angle range δ in which the delta directivity is 9 dB smaller than the sigma directivity in the directivity of each elevation angle based on the relationship between the two pulses forming the monopulse directivity described in Figure 9B.

As can be seen from Figures 10A to 10F, the predetermined angle range δ is around 70 degrees for all elevation angles. This means that targets in a certain angular range in the azimuth direction can be captured regardless of the elevation angle. Furthermore, this value of around 70 degrees is preferable because it allows for a moderate amount of overlapping of adjacent directivities when performing directional scanning in the azimuth direction.

Figures 11A to 11H are diagrams showing an example of the analysis results of the sigma directivity and delta directivity of the circularly arranged phased array antenna 1 formed in horizontal directional scanning in this embodiment. Each of Figures 11A to 11H shows the sigma directivity and delta directivity formed at θ = 30 degrees representing the elevation angle when φ, which is the desired communication direction, is changed. Figure 11A shows the calculation results for φ = 0 degrees, Figure 11B shows φ = 45 degrees, Figure 11C shows φ = 90 degrees, Figure 11D shows φ = 135 degrees, Figure 11E shows φ = 180 degrees, Figure 11F shows φ = 225 degrees, Figure 11G shows φ = 270 degrees, and Figure 11H shows φ = 315 degrees.

As shown in Figures 11A to 11H, by directional scanning at 45 degree intervals from φ = 0 degrees to φ = 315 degrees, it can be seen that the sandwich array device 100 composed of antenna devices 10A and 10B can perform the space division monopulse method for the entire solid angle.

Therefore, the sandwich array device 100 can form monopulse directivity, which is directivity in a specific direction and angle range, using sigma directivity and delta directivity, and perform three-dimensional scanning. This allows the flying vehicle 300 equipped with the sandwich array device 100 to selectively communicate with other flying vehicles by eliminating synchronous interference, thereby preventing collisions with other flying vehicles.

<Phase inversion weighting method>
Next, a phase inversion weighting method will be described, in which the phase of the voltage of the signal induced in the second antenna element (#9) by the received radio wave having an angle in the elevation angle direction is obtained by synthesis. In the following description, the circularly arranged phased array antenna 1 will be described as having eight first antenna elements.

First, as a comparative example, we will explain a method for estimating the direction of an arriving wave that has no angle in the elevation direction.

Figure 12 is a diagram showing the voltage of a signal induced in the circularly arranged phased array antenna 1 in the comparative example. Figure 12 shows the voltage of a signal induced in the circularly arranged phased array antenna 1 in the comparative example by a received radio wave in an arrival wave direction that has no angle in the elevation angle direction.

In the comparative example, the direction of arrival of the wave can be estimated using the phase difference between the voltages of the antenna elements arranged at the center and on the circle. More specifically, the direction of arrival of the received signal can be estimated by calculating the difference between the phase of the sum of the received signals of the eight first antenna elements and the phase of the received signal of the one second antenna element arranged at the center.

Assume that the received radio waves arrive only from the direction that forms an angle φ with the X-axis of the XY plane, which is the plane of the circularly arranged phased array antenna 1. In this case, the signal, i.e., voltage Ei(φ), induced in the i-th first antenna element by the received radio waves can be calculated from the following (Equation 4).

Since the timing of reception for each of the first antenna elements (#1 to #8) is slightly different, a weighting function that reflects this is applied to the voltage Ei calculated from (Equation 4) and the weighted results are then added together for all of the signals from the first antenna elements on the circle. This makes it possible to obtain the voltage E _Δ (φ) of the first signal, which is the sum of the received signals from each of the eight first antenna elements. This voltage E _Δ (φ) of the first signal can be calculated from (Equation 5).

Here, the number of elements shown in (Equation 5) is 8. However, if the number of first antenna elements becomes infinite in order to understand the nature of the voltage E _Δ (φ) of the first signal, the voltage E _Δ (φ) can be expressed by the following (Equation 6) using a Bessel function.

Also, the voltage of the signal (third signal) induced in the second antenna element (#9) by the received radio wave arriving from the direction forming φ with the X-axis is denoted as _EΩ (φ). Furthermore, if the phase angle of the voltage _EΩ (φ) of the third signal is denoted as _∠EΩ and the phase angle of the voltage _EΔ (φ) of the first signal is denoted as _∠EΔ , then the phase difference _φm can be calculated using the following (Equation 7).

Therefore, it can be seen that the phase difference φ _m between the first and third signals is a value that is approximately proportional to the arrival wave angle φ of the received radio wave that has no angle in the elevation angle direction. In this way, by utilizing the phase difference between the voltages of the antenna elements arranged at the center and on the circumference of the circularly arranged phased array antenna 1, the arrival wave direction that has no angle in the elevation angle direction can be estimated.

Figure 13A is a diagram showing the directional characteristics in the XY plane of the first signal and the third signal in the comparative example. Figure 13B is a diagram showing the phase characteristics in the XY plane of the first signal and the third signal in the comparative example. Figures 13A and 13B show the analysis results of an electromagnetic field analysis performed using the moment method, assuming that the received radio waves arrive only from the direction that forms an angle φ with the X axis in the comparative example. The frequency used in the analysis is 2 GHz, and the radius a of the circularly arranged phased array antenna 1 is 4.9 cm.

As shown in Figure 13A, the directional characteristics of the first signal and the third signal, i.e., the directional characteristics when the received signals of the first antenna element are added together and the directional characteristics of the received signal of the second antenna element, are omnidirectional in the XY plane. Figure 13B shows that the phase characteristics of the directionality of the first signal change depending on the azimuth angle. This shows that the difference between the phase characteristics of the directionality of the first signal and the phase characteristics of the directionality of the third signal is proportional to the direction of the arriving wave.

Next, we will explain the method for estimating the direction of an arriving wave having an angle in the elevation direction in this embodiment.

Figure 14 is a diagram showing two weighting functions of the first antenna element of the circularly arranged phased array antenna 1 in this embodiment. In Figure 14, two weighting functions are shown for each of the first antenna elements (#1 to #8): a counterclockwise weighting function (weight) and a clockwise weighting function (weight).

In the circularly arranged phased array antenna 1 shown in Figure 14, assume that the received radio waves having an angle in the elevation direction arrive from a direction that forms an angle φ with the X-axis. In this case, the signal, i.e., voltage Vi(φ), induced in the i-th first antenna element located on the circumference by the received radio waves can be calculated from the following (Equation 8).

Since the reception timing differs for each first antenna element (#1 to #8), in order to reflect this, the voltage Vi calculated from (Equation 8) is weighted by two weighting functions W _i ^a and W _i ^b defined in (Equation 9) and (Equation 10).

Then, by adding up the signals of all the first antenna elements on the circle, it is possible to obtain the first signal voltage E _Δ ^a and the second signal voltage E _Δ ^b, which are the sum of the received signals of each of the eight first antenna elements. More specifically, the first signal voltage E _Δ ^a can be calculated from (Equation 11), and the second signal voltage E _Δ ^b can be calculated from (Equation 12).

Fig. 15A shows the elevation angle directivity of the first signal obtained by performing electromagnetic field analysis using the moment method. Fig. 15B shows the elevation angle directivity of the third signal obtained by performing electromagnetic field analysis using the moment method. In Fig. 15A and Fig. 15B, the frequency used in the analysis is 2 GHz, the radius a of the circularly arranged phased array antenna 1 is 4.9 cm, and the radius _rg of the ground plate is 7 cm.

As shown in Fig. 15A, the directional characteristic of the first signal, i.e., the directional characteristic when the received signal of the first antenna element weighted by the weighting function W _i ^a is added, has good radiation characteristics at high elevation angles, including the zenith direction (+Z axis direction). In contrast, as shown in Fig. 15B, the directional characteristic of the third signal, i.e., the directional characteristic of the received signal of the second antenna element, has a deep null in the zenith direction. This means that even if the difference between the phase characteristic of the directivity of the first signal and the phase characteristic of the directivity of the third signal is calculated using the same method as in the comparative example, it is not possible to estimate the direction of arrival of the wave including the zenith direction.

Therefore, in this embodiment, a method of synthesizing the voltage E _{Ω of the third signal from the voltage E Δa of the first signal and the voltage E Δb of the second signal using a phase inversion weighting method described below has been devised. That is, in this embodiment, a method of synthesizing the phase angle of the voltage E Ω of} ^{the third} _signal ^induced in the second antenna element from the phase angle of the first signal _{E Δa} and the phase angle of _{the voltage E Δb} ^of the second signal has been devised.

FIG. 16 is a schematic diagram for explaining the principle of the phase inversion weighting method. In FIG. 16, the phase angle ∠E _Δ ^a of the voltage E _Δ ^a of the first signal and the phase angle ∠E _Δ ^b of the voltage E _Δ ^b of the second signal are shown. As shown in FIG. 16, the phase angle ∠E _Δ ^a and the phase angle ∠E _Δ ^b have opposite inclinations with respect to the direction of the incoming wave. In other words, the phase angle ∠E _Δ ^a and the phase angle ∠E _Δ ^b are symmetrical. Therefore, the phase angle ∠E _Ω (also called the reference phase) of the voltage E _Ω of the third signal used to estimate the direction of the incoming wave is defined as shown in the following (Equation 13). As a result, the angle φ _m representing the direction of the incoming wave having an angle in the elevation angle direction can be expressed by (Equation 14).

Here, both the voltage E _Δ ^a of the first signal and the voltage E _Δ ^b of the second signal have the elevation angle directivity shown in Fig. 15A. Meanwhile, according to (Equation 13), the directivity of the voltage E _Ω of the third signal is the average value of the directivities of the voltage E _Δ ^a of the first signal and the voltage E _Δ ^b of the second signal. Therefore, the directivity of the voltage E _Ω of the third signal is the same as that of Fig. 15A and has good elevation angle directivity, so it can be seen that it is possible to estimate the direction of arrival waves at high elevation angles.

Figure 17 is a diagram showing the three-dimensional directivity of the first signal and the third signal in this embodiment. Figure 17 shows the three-dimensional directivity of the first signal and the third signal obtained by performing electromagnetic field analysis using the moment method.

The directivity of the voltage _EΩ of the third signal is the average of the directivity of ^the voltage _EΔa of the first signal and the directivity of ^the voltage _EΔb of the second signal, and therefore has the same amplitude characteristics as the voltage _EΔa of the first signal. From Fig. 17, it can be seen that ^the voltage _EΔa of the first signal and the voltage ^EΩ _of the third signal have high gain in the zenith direction, which is above the ground plane (+Z direction).

Fig. 18A shows the directivity and the estimated angle characteristic of the direction of the arrival wave when the elevation angle is 30 degrees. Fig. 18A (a) shows the directivity of the voltage E _Δ ^a of the first signal and the voltage E _Ω of the third signal when the elevation angle is 30 degrees. Fig. 18A (b) shows the angle φ _m defined by the above (Equation 14), that is, the angle φ _m representing the direction of the arrival wave having an elevation angle of 30 degrees.

Fig. 18B shows the directivity and the estimated angle characteristic of the direction of the arrival wave when the elevation angle is 80 degrees. Fig. 18B (a) shows the directivity of the voltage E _Δ ^a of the first signal and the voltage E _Ω of the third signal when the elevation angle is 80 degrees. Fig. 18B (b) shows the angle φ _m defined by the above (Equation 14), that is, the angle φ _m representing the direction of the arrival wave having an elevation angle of 80 degrees.

18A(a) and 18B(a) show that both ^the voltage E _Δa of the first signal and the voltage E _Ω of the third signal maintain a constant high gain in the azimuth direction at a high elevation angle of 80 degrees. Also, 18A(b) and 18B(b) show that the estimated angle φ _m is proportional to the direction of arrival wave at both elevation angles θ = 30 degrees and 80 degrees.

<Confirmation of the relationship between transmission capacity and base plate size>
In the antenna device 10 of this embodiment, the eight first antenna elements of the circularly arranged phased array antenna 1 are divided into four elements to form a combination of sub-arrays arranged perpendicular to the direction of incoming waves, thereby realizing a MIMO array antenna function. By realizing the MIMO array antenna function, the antenna device 10 of this embodiment can improve the transmission capacity.

It is known that the radiation directivity in the elevation direction of a monopole antenna attached to the center of a circular ground plate can be controlled by changing the size of the ground plate. However, the effect of ground plate size on the transmission capacity of a MIMO array antenna has not been fully studied until now. Therefore, the results of confirming the relationship between transmission capacity and ground plate size are explained below.

FIG. 19 is a diagram showing the change in transmission capacity with respect to the elevation angle direction of the incoming wave when the radius r _g of the base plate 2 in this embodiment is changed. FIG. 19 shows the result of a Monte Carlo simulation using a channel model of a cluster propagation environment in which both the azimuth angle and the elevation angle in a three-dimensional coordinate system have Gaussian distribution incoming waves. The radiation characteristics of the entire MIMO array antenna were calculated by the moment method. The analysis frequency is 2 GHz. All antenna elements constituting the circularly arranged phased array antenna 1 are 1/4 wavelength monopole antennas with a length of 37.5 mm and a radius of 0.5 mm. The radius a of the circularly arranged phased array antenna 1 functioning as a 4×4 MIMO array antenna is 4.9 cm, and the spacing between the first antenna elements #2 and #3 is 1/4 wavelength at 2 GHz to form a cardioid directivity.

It can be seen from Fig. 19 that the transmission capacity at the elevation angle changes depending on the radius _rg of the ground plate 2. More specifically, from Fig. 19, the transmission capacity is maximum when the radius _rg of the ground plate 2 is 13 cm in the elevation angle range of 0 to 15 degrees, when the radius _rg of the ground plate 2 is 10 cm in the range of 15 to 60 degrees, and when the radius _rg of the ground plate is 13 cm in the range of 60 to 90 degrees. Therefore, it can be seen that it is necessary to control the optimum radius _rg of the ground plate according to the elevation angle at which communication is desired.

In particular, when the radius _rg of the ground plate 2 is 10 cm, the transmission capacity increases significantly at high elevation angles of 30 degrees to 60 degrees. From this result, taking into consideration the elevation angle characteristics and a practical antenna shape, one of the variable radii _rg of the ground plate 2 in this embodiment is set to 10 cm.

<Confirmation of 3D radiation directivity in sub-array>
Next, the three-dimensional radiation directivity in the sub-array will be described. The following describes the three-dimensional radiation directivity when the first antenna elements #1 and #4 are the sub-array 2 (hereinafter also referred to as the monopole array antenna) in the circularly arranged phased array antenna 1. As a comparative example, the three-dimensional radiation directivity when the first antenna elements #1 and #4 are the sub-array 2 (hereinafter also referred to as the dipole array antenna) in the circularly arranged phased array antenna 1 that does not have the ground plane 2 and is composed of nine dipole antennas will also be described.

Figure 20A is a diagram showing the three-dimensional radiation directivity of a dipole array antenna in a comparative example. Figure 20B is a diagram showing the three-dimensional radiation directivity of a monopole array antenna in this embodiment.

As can be seen by comparing Figures 20A and 20B, the three-dimensional radiation directivity of the dipole array antenna shown in Figure 20A has a deep null in the zenith direction (+Z direction), and radiation at high elevation angles is significantly reduced. On the other hand, the monopole array antenna shown in Figure 20B does not have a deep null in the zenith direction (+Z direction), and shows a significant improvement in radiation gain at high elevation angles compared to the dipole array antenna shown in Figure 20A.

<Confirmation of Three-Dimensional Radiation Directivity in the Sandwich Array Device 100>
Next, a three-dimensional radiation directivity when the sandwich array device 100 of this embodiment performs directional scanning will be described. The sandwich array device 100 has a function of performing directional scanning at 45-degree intervals, but the following description will be given by taking only the three-dimensional radiation directivity at 90-degree intervals as an example.

21A to 22D are diagrams showing three-dimensional directivity when directional scanning is performed by the sandwich array device 100 of this embodiment. In Figs. 21A to 21D, three-dimensional radiation directivity is shown when the azimuth angle desired for communication is changed in the antenna device 10A at the top of the sandwich array device 100, where the radius _rg of the ground plate 2 is 10 cm and the first antenna elements #1 and #4 are the subarray 2. In Figs. 22A to 22D, three-dimensional radiation directivity is shown when the azimuth angle desired for communication is changed in the antenna device 10B at the bottom of the sandwich array device 100, where the radius _rg of the ground plate 2 is 10 cm and the first antenna elements #1 and #4 are the subarray 2. In Figs. 21A to 21D and Figs. 22A to 22D, three-dimensional radiation directivity at 90 degree intervals is shown.

Figures 21A to 22D show that the gain in the azimuth direction of the desired communication is large. Also, as shown in Figures 21A to 22D, by performing directional scanning, the directivity rotates while irradiating the high elevation angle direction. The same can be said for the directivities of the other three subarrays (subarrays 1, 3, and 4).

It can be seen that by linking the upper antenna device 10A and the lower antenna device 10B in a complementary manner, the directional gain in the depression angle direction can be improved and the entire solid angle can be covered.

<Confirmation of transmission capacity of MIMO array antenna>
Next, the transmission capacity when sandwich array device 100 according to the present embodiment is operated as a MIMO array antenna will be described. As a comparative example, the transmission capacity when a circularly arranged phased array antenna 1 (referred to as a half-wavelength dipole array antenna) that does not have a base plate 2 and is composed of nine half-wavelength dipole antennas is operated as a MIMO array antenna will also be described.

Figure 23 is a diagram showing the elevation angle characteristic of the transmission capacity of a MIMO array antenna using a half-wavelength dipole array antenna in a comparative example. Figure 23 shows the analysis result of the transmission capacity of the MIMO array antenna when the azimuth direction of the incoming wave is set to 0 degrees (φ = 0 degrees) and the incoming wave elevation angle is changed. Here, the analysis frequency is 2 GHz, the SNR is 30 dB, and the XPR (cross polarization power ratio) is 10 dB. σ shown in Figure 23 represents the spread angle (standard deviation) of the Gaussian incoming wave in the azimuth direction (horizontal direction) and the elevation direction (elevation angle direction). θ represents the angle of the elevation direction, i.e., the incoming wave elevation angle.

From Figure 23, we can see that a MIMO array antenna using a half-wavelength dipole array antenna has a characteristic in which the transmission capacity is symmetrical with respect to an elevation angle of 0 degrees. It can also be seen that there is a significant decrease in transmission capacity in the zenith direction (θ = +90 degrees) and nadir direction (θ = -90 degrees). When the spread angle is 10 degrees, we can see that the transmission capacity at an elevation angle of 0 degrees is compared to that at elevation angles of ±90 degrees, resulting in a degradation of 22 bits/s/Hz.

These results are thought to be due to the fact that nulls are formed in the zenith and nadir directions in the dipole array antenna, as shown in Figure 20A.

Figure 24A is a diagram showing the azimuth characteristic of the transmission capacity of a MIMO array antenna using a sandwich array device in an embodiment. Figure 24B is a diagram showing the elevation angle characteristic of the transmission capacity of a MIMO array antenna using a sandwich array device 100 in this embodiment. Figure 24A shows the elevation angle characteristic of the transmission capacity with respect to the azimuth angle, and Figure 24B shows the elevation angle characteristic of the transmission capacity with respect to the elevation angle.

In Figures 24A and 24B, the analysis frequency is 2 GHz, the SNR is 30 dB, and the XPR is 10 dB. Note that a line-of-sight (LOS) propagation environment is expected for flying vehicle-to-vehicle communications. Therefore, a propagation environment in which the vertical polarization component is dominant over the horizontal polarization component is assumed, and the XPR, which represents the cross-polarized power ratio, is set to 10 dB.

In addition, in the sandwich array device 100, as a MIMO array antenna, the upper antenna device 10A is selected for an elevation angle target of 0 degrees to 90 degrees, and the lower antenna device 10B is selected for a depression angle target of -90 degrees to 0 degrees.

As can be seen from Figure 24A, the transmission capacity of the MIMO array antenna is almost constant in all azimuth directions for any spread angle. This shows that a stable transmission capacity can be achieved by controlling the beam (directivity) of the MIMO array antenna using the sandwich array device 100 to the azimuth angle of the arriving wave.

Also, from Figure 24B, it can be seen that the transmission capacity of the MIMO array antenna has symmetric characteristics with respect to an elevation angle of 0 degrees due to the switching operation between the upper antenna device 10A and the lower antenna device 10B. Also, from Figure 24B, it can be seen that the fluctuation in the transmission capacity of the MIMO array antenna with respect to the elevation angle is small. This shows that a stable transmission capacity can be achieved by controlling the beam (directivity) of the MIMO array antenna using the sandwich array device 100 to the elevation angle of the arriving wave.

23, 24A, and 24B, the following can be seen. That is, the transmission capacity of the MIMO array antenna using the sandwich array device 100 is almost the same as the maximum transmission capacity of the MIMO array antenna using the half-wavelength dipole array antenna.

This shows that it is appropriate to set the radius _rg of the base plate 2 to 10 cm from FIG. 19 for the purpose of achieving full solid angle irradiation in the above embodiment.

Figure 25 is a diagram showing the three-dimensional transmission capacity over the entire solid angle of a MIMO array antenna using a half-wavelength dipole array antenna in a comparative example. Figure 26 is a diagram showing the three-dimensional transmission capacity over the entire solid angle of a MIMO array antenna using a sandwich array device 100 in this embodiment.

As can be seen from Figure 25, the transmission capacity of a MIMO array antenna using a half-wavelength dipole array antenna has deep nulls in the zenith and nadir directions. On the other hand, as shown in Figure 26, the transmission capacity of a MIMO array antenna using the sandwich array device 100 of this embodiment has a spherical shape of equal radius. This shows that the MIMO array antenna using the sandwich array device 100 of this embodiment has the ability to connect to a target in any direction.

The above-mentioned characteristics of the transmission capacity of the MIMO array antenna using the sandwich array device 100 in this embodiment are obtained by appropriate power synthesis of the θ and φ components of the radiation directivity. In particular, the φ component arises from the current distribution induced in the ground plate 2. For this reason, in principle, the φ component does not exist in a half-wavelength dipole array antenna made up of dipole antennas.

<Verifying that high-speed communication is possible>
Next, it has been confirmed that high-speed communication is possible when sandwich array device 100 in this embodiment is operated as a MIMO array antenna by using outage probability based on transmission capacity, as will be described below.

First, the probability of communication outage is defined as the probability that the transmission capacity falls below a specified threshold, as shown in (Equation 15) below.

In (Equation 15), p(θ,φ) denotes the probability density function (PDF) of the transmission capacity over the entire solid angle, _C denotes the transmission capacity threshold, _φ denotes the azimuth angle corresponding to _C , and _θ denotes the elevation angle corresponding _to C.

Fig. 27 is a diagram showing the communication outage probability over the entire solid angle of a MIMO array antenna using the sandwich array device 100 according to the present embodiment. The present disclosure shown in Fig. 27 shows a MIMO array antenna using the sandwich array device 100. The comparative example shown in Fig. 27 shows a MIMO array antenna using the above-mentioned half-wavelength dipole array antenna. In Fig. 27, the horizontal axis represents the transmission capacity normalized by the maximum transmission capacity _Cmax defined by the following (Equation 16).

From Figure 27, it can be seen that in the comparative example, when the spread angle is 10 degrees (σ = 10 degrees), for example, the normalized transmission capacity r at which the probability of communication outage becomes zero is 22%. On the other hand, in the present disclosure, when the spread angle is 10 degrees, the normalized transmission capacity r at which the probability of communication outage becomes zero is 87%.

This shows that a MIMO array antenna using the sandwich array device 100 of this disclosure, i.e., this embodiment, enables high-speed communication at 87% of the maximum transmission capacity in all directions in three-dimensional space.

From the above confirmations, it has become clear that the sandwich array device 100 of this embodiment 1) can form monopulse directivity by the space division monopulse method, 2) can estimate the direction of arrival of a wave over the entire solid angle, and 3) has the function of a MIMO array antenna. In addition, in the sandwich array device 100 of this embodiment, the radius _rg of the ground plate 2 is set to 7 cm when forming monopulse directivity and when estimating the direction of arrival of a wave over the entire solid angle, and the radius _rg of the ground plate 2 is set to 10 cm when realizing the function of a MIMO array antenna. In other words, in the sandwich array device 100 of this embodiment, in order to realize all the functions 1) to 3) with a single device, the radius _rg of the ground plate 2 is switched by turning the diode on and off.

(Modification)
In the above embodiment, a case has been described in which one sandwich array device 100 composed of two antenna devices 10 is used as a MIMO array antenna, but this is not limiting. A multi-element MIMO array antenna may be formed by modularizing and combining sandwich array device 100. This case will be described below.

Figure 28 is a diagram showing an example of a multi-element MIMO array antenna in a modified example. In Figure 28, an example is shown in which five sandwich array devices 100a, 100b, 100c, 100d, and 100e are combined to form a multi-element MIMO array antenna.

In the above embodiment, the antenna device 10 is used as a MIMO array antenna, but this is not limiting. The antenna device 10 may be modularized and combined to form a multi-element MIMO array antenna. This case will be described below.

Figure 29A is a diagram showing an example of a 4x4 MIMO array antenna in a modified example. Figure 29B is a diagram showing an example of an 8x8 MIMO array antenna in a modified example. Figure 29C is a diagram showing an example of a 16x16 MIMO array antenna in a modified example. Figure 29D is a diagram showing an example of a 32x32 MIMO array antenna in a modified example. Figure 29E is a diagram showing an example of a 64x64 MIMO array antenna in a modified example. Figure 29F is a diagram showing an example of a 128x128 MIMO array antenna in a modified example. Figure 29G is a diagram showing an example of a 256x256 MIMO array antenna in a modified example. Each of Figures 29A to 29G shows a combination of antenna devices 10 when the multi-element MIMO array antenna is viewed from above.

In this way, by modularizing and combining the sandwich array device 100 or the antenna device 10, it can be easily expanded into a multi-element MIMO array antenna.

Other Embodiments
The present invention is not limited to the above-described embodiment. Various modifications and variations can be made to the above-described embodiment within the same scope as the present invention or within an equivalent scope.

In addition, in the above embodiments, each component may be configured by dedicated hardware, or components that can be realized by software may be realized by executing a program.

For example, the modules constituting the antenna device 10 may be realized in the form of IC (integrated circuit), ASIC (application specific integrated circuit), and LSI (large scale integration), or may be realized by a processor based on a CPU such as ARM and a machine such as a PC (personal computer). Each of these modules may be included in many single-function LSIs or one LSI. Although the name used here is LSI, it may also be called IC, system LSI, super LSI, or ultra LSI depending on the degree of integration. Furthermore, the integration method is not limited to LSI, and integration can also be done by dedicated circuits or general-purpose processors. This includes special microprocessors such as DSP (digital signal processors) that can be instructed by program instructions. FPGA (field programmable gate array) that can be programmed after LSI manufacture or reconfigurable processors that can reconfigure the connection or arrangement of LSI can be used for the same purpose. In the future, as manufacturing and processing technologies develop, completely new technologies may replace LSI. Integration can be realized by such technologies.

The present invention can be used in antenna devices that estimate the direction of arrival of received radio waves and control directivity to the estimated direction of arrival to achieve highly reliable communication, and can be used in particular in antenna devices used in flying vehicles, MIMO antennas, etc.

REFERENCE SIGNS LIST 1, 1A, 1B Circularly arranged phased array antenna 2, 2A, 2B, 2a, 2b Ground plate 3 Control unit 4 Calculation unit 10, 10A, 10B Antenna device 100, 100a, 100b, 100c, 100d, 100e Sandwich array device 301 Diode 302 Choke coil

Claims (6)

a circularly arranged phased array antenna including N first antenna elements (N being a natural number of 3 or more) arranged at equal intervals on a circle and one second antenna element arranged approximately at the center of the circle;
A circular base plate whose radius can be changed by switching control;
a subarray control unit that divides the N first antenna elements into the N/2 subarrays orthogonal to a direction of an incoming wave arriving at the circularly arranged phased array antenna, the direction being parallel to the ground plane, and controls directivities of the subarrays to independently form the N/2 beams directed in the direction of the incoming wave,
The first antenna element and the second antenna element are configured as monopole antennas mounted on the ground plane.
Antenna device. The first antenna element and the second antenna element are configured as 1/4 wavelength monopole antennas.
2. The antenna device according to claim 1. Each of the N/2 subarrays is composed of a combination of two antenna elements whose arrangement directions are parallel to each other,
The combination is changed depending on the direction of arrival of the wave.
3. The antenna device according to claim 1 or 2 . a circularly arranged phased array antenna including N first antenna elements (N being a natural number of 3 or more) arranged at equal intervals on a circle and one second antenna element arranged approximately at the center of the circle;
A circular base plate whose radius can be changed by switching control;
a directivity control unit that controls the directivity of the circularly arranged phased array antenna within a predetermined angle range including the elevation angle direction by providing phases to two of the first antenna elements that are closest to a desired communication direction in order to form delta directivity and sigma directivity, the phases being calculated using a path difference in an elevation angle direction, which is a direction of an incoming wave arriving at the circularly arranged phased array antenna and the ground plane,
the first antenna element and the second antenna element are configured as monopole antennas mounted on the ground plane,
The delta directivity is a directivity that reduces a radiation gain by realizing anti-phase excitation in the elevation angle direction,
The sigma directivity is a directivity that increases radiation gain by realizing in-phase excitation in the elevation angle direction.
Antenna device. a circularly arranged phased array antenna including N first antenna elements (N being a natural number of 3 or more) arranged at equal intervals on a circle and one second antenna element arranged approximately at the center of the circle;
A circular base plate whose radius can be changed by switching control;
a calculation unit that performs statistical analysis on a first signal and a second signal obtained by multiplying a received signal of each of the first antenna elements by two weights corresponding to an arrangement of the first antenna elements and summing the result, and a third signal which is a received signal of the second antenna element, and calculates a phase difference between the first signal and the third signal , thereby estimating an arrival wave direction which is a direction of an arrival wave arriving at the circularly arranged phased array antenna relative to the ground plane,
the first antenna element and the second antenna element are configured as monopole antennas mounted on the ground plane,
the calculation unit calculates a phase angle of the voltage of the third signal by averaging a phase angle of the voltage of the first signal and a phase angle of the voltage of the second signal, thereby calculating a phase difference between the first signal and the third signal;
The phase difference is approximately proportional to the direction of the incoming wave.
Antenna device. A first antenna device and a second antenna device are provided,
Each of the first antenna device and the second antenna device
a circularly arranged phased array antenna including N first antenna elements (N being a natural number of 3 or more) arranged at equal intervals on a circle and one second antenna element arranged approximately at the center of the circle;
A circular base plate whose radius is variable by switching control;
the first antenna element and the second antenna element are configured as monopole antennas mounted on the ground plane,
The first antenna device is disposed at the zenith of the flying vehicle;
The second antenna device is disposed at the nadir of the flying vehicle;
The first antenna device and the second antenna device have opposite directions of the circularly arranged phased array antenna as viewed from the ground plane.
Sandwich array device.

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