JP7629255B2 - Steering wheel grip sensor and grip detection method - Google Patents

Description

This disclosure relates to a steering wheel grip sensor and grip detection method that detects contact of a person's hand with a steering wheel.

Patent document 1 discloses a capacitive seating detection system that detects the proximity capacitance of a human body.

U.S. Pat. No. 7,656,169

However, according to Patent Document 1, there is a problem in that detection accuracy deteriorates due to the switching operation caused by the drive signal, frequency fluctuations in the drive signal, and the effects of noise.

Therefore, the present disclosure provides a steering grip sensor and grip detection method that suppresses deterioration of detection accuracy due to frequency fluctuations, noise, etc., and improves detection accuracy.

The steering grip sensor according to one aspect of the present disclosure includes a driven electrode that covers the rim of the steering wheel, a sensor electrode that covers the driven electrode, a sine wave generator that supplies a sine wave voltage to the driven electrode, a charge amplifier that has a feedback capacitance element and detects the charge generated in the sensor electrode, a multiplication processor that multiplies the sine wave voltage by the output voltage of the charge amplifier, an integrator that smoothes the multiplication result by integrating it, and a grip determination unit that determines whether the steering wheel is being gripped depending on the level of the smoothed multiplication result.

In addition, a grip detection method according to one aspect of the present disclosure is a grip detection method for a steering grip sensor that includes a planar driven electrode that follows the rim of the steering wheel and a planar sensor electrode that faces the driven electrode, in which a sine wave voltage is supplied to the driven electrode, a charge amplifier having a feedback capacitance element detects a change in the amount of charge generated by the capacitance of the sensor electrode, the charge amplifier generates an output voltage that represents the detected change in the amount of charge as a change in voltage, the sine wave voltage is multiplied by the output voltage of the charge amplifier, the multiplication result is smoothed by integrating it, and the presence or absence of gripping the steering wheel is determined according to the level of the smoothed multiplication result.

The steering grip sensor and grip detection method disclosed herein can reduce deterioration in accuracy due to frequency fluctuations and noise, improving detection accuracy.

FIG. 1 is a diagram showing an example of an interior layout of a vehicle in which a steering grip sensor according to a first embodiment is arranged. FIG. 2 is a diagram showing an example in which a steering cover of the steering grip sensor in the first embodiment is attached to a rim. FIG. 3 is a diagram showing an example of a cross-sectional configuration of the steering cover in the first embodiment. FIG. 4 is a block diagram showing an example of a circuit configuration of the steering grip sensor in the first embodiment. FIG. 5 is an explanatory diagram of the operation of the sensor of the comparative example in comparison with the steering grip sensor in the first embodiment. FIG. 6 is a block diagram showing an example of a circuit configuration of the steering grip sensor in the second embodiment. FIG. 7 is an explanatory diagram of fault diagnosis in the second embodiment. FIG. 8 is a diagram showing various waveforms during failure diagnosis in the second embodiment. FIG. 9 is a block diagram showing another example of the circuit configuration of the steering grip sensor in the first embodiment. FIG. 10 is a block diagram showing an example of a circuit configuration of the steering grip sensor in the third embodiment. FIG. 11 is a block diagram showing an example of a circuit configuration of a steering grip sensor in the fourth embodiment. FIG. 12 is a diagram showing an example of the correlation between the resistance Rx and the capacitance Cx of the steering grip sensor in the fourth embodiment.

(Findings that formed the basis of this disclosure)
The present inventors have found that the seating detection system of Patent Document 1 described in the "Background Art" section has the following problems.

First, in Patent Document 1, since a switching operation is required for synchronous detection of a sense signal, if degradation occurs in the slew rate of the edge component of the control signal that controls the switching, distortion or detection phase shift occurs in the signal after synchronous detection, resulting in a problem of degraded detection accuracy.

Secondly, since the sensitivity of the sense signal depends on the frequency of the sine wave signal used as the drive signal, there is a problem that if frequency fluctuations occur in the drive signal, the detection accuracy deteriorates.

Thirdly, the higher the frequency of the drive signal, the greater the sense signal, which creates the problem that the higher the frequency of the external high-frequency noise, the worse the detection accuracy becomes.

Therefore, this disclosure provides a steering grip sensor and grip detection method that suppresses deterioration of accuracy due to frequency fluctuations, noise, etc., and improves detection accuracy.

In order to solve such problems, the steering grip sensor according to one aspect of the present disclosure includes a planar driven electrode that follows the rim of the steering wheel, a planar sensor electrode that faces the driven electrode, a sine wave generating unit that supplies a sine wave voltage to the driven electrode, a charge amplifier that has a feedback capacitance element and detects a change in the amount of charge generated by the capacitance of the sensor electrode and outputs the change in the amount of charge as a change in voltage, a multiplication processing unit that multiplies the sine wave voltage by the output voltage of the charge amplifier, an integrator that smoothes the multiplication result of the multiplication processing unit by integrating it, and a grip determination unit that determines whether or not the steering wheel is being gripped depending on the level of the smoothed multiplication result.

This makes it possible to suppress deterioration of accuracy due to frequency fluctuations and the effects of noise, and improve detection accuracy. Specifically, first, the output voltage of the charge amplifier is in phase with the sine wave voltage as the drive signal, i.e., the phase difference is zero, and the detection of the capacitance due to gripping is performed by multiplication detection in the multiplication processing unit. This eliminates the need for switching operations as in conventional technology, suppresses phase shifts and signal distortion due to switching, suppresses deterioration of detection accuracy, and improves detection accuracy. Second, since the sensitivity of the output voltage of the charge amplifier does not depend on the frequency of the sine wave voltage as the drive signal, deterioration of detection accuracy can be suppressed even if frequency fluctuations occur in the sine wave signal. Third, since the magnitude of the output voltage of the charge amplifier does not depend on the frequency of the sine wave voltage as the drive signal, deterioration of detection accuracy due to external high-frequency noise can be suppressed.

Here, the charge amplifier may include a first input terminal to which the sine wave voltage is input, a second input terminal connected to the sensor electrode, an output terminal, the feedback capacitance element connected between the output terminal and the second input terminal, and an operational amplifier connected to the first input terminal, the second input terminal, and the output terminal, and the sine wave voltage output from the output terminal may be supplied to the sensor electrode via the feedback capacitance element.

With this, the charge amplifier is equipped with a feedback capacitance element, and outputs the change in the amount of charge generated in the sensor electrode due to the user's grip as a change in voltage. This charge amplifier can be configured with a general-purpose operational amplifier with two inputs and one output, which reduces circuit costs.

Here, the sine wave generating unit may input a square wave signal and generate the sine wave voltage synchronized with the square wave signal.

This eliminates the need for a quartz crystal oscillator, reducing the cost of the circuit.

Here, the charge amplifier may include a low-pass filter connected in parallel with the feedback capacitance element, and the low-pass filter may pass frequencies lower than the frequency of the sine wave voltage.

As a result, the output voltage from the output terminal indicating the capacitance due to gripping is in phase with the sine wave voltage, while the signal indicating the resistance component such as leakage from the output terminal is out of phase with the sine wave voltage by 90 degrees. The latter signal is offset by the multiplication in the multiplication processing unit, improving disturbance resistance and noise resistance characteristics.

Here, the steering grip sensor further includes a memory, a first AD converter that performs AD conversion (analog-digital conversion) of the output voltage of the charge amplifier in synchronization with the square wave signal and stores the AD conversion result in the memory as first digital data, and a second AD converter that performs AD conversion of the sine wave voltage from the sine wave generating unit in synchronization with the square wave signal and stores the AD conversion result in the memory as second digital data, and the multiplication processing unit may perform a process of averaging each of the waveforms indicated by the first digital data and the waveforms indicated by the second digital data, and multiply the averaged first digital data by the averaged second digital data.

This allows AD conversion and multiplication to be performed with high precision.

Here, the steering grip sensor further includes a memory, a first AD converter that performs AD conversion on the output voltage of the charge amplifier in synchronization with the square wave signal and stores the AD conversion result in the memory as first digital data, and a second AD converter that performs AD conversion on the sine wave voltage from the sine wave generating unit in synchronization with the square wave signal and stores the AD conversion result in the memory as second digital data, and the multiplication processing unit may multiply the first digital data by the second digital data and perform a process of averaging the waveform indicated by the digital data of the multiplication result.

This reduces the amount of data stored in memory and the amount of computation required compared to when averaging is performed before multiplication.

Here, the square wave signal may be switched between at least three frequencies, and the first AD converter and the second AD converter may store the first digital data and the second digital data in the memory for each of the at least three frequencies.

This makes it possible to use the first and second digital data of other frequencies without using the first and second digital data corresponding to the frequency affected by the disturbance noise. It also makes it possible to perform frequency hopping so as to avoid the disturbance noise.

Here, the first AD converter and the second AD converter may generate the first digital data and the second digital data so as to correspond to at least two periods of the waveform of the sine wave voltage.

This allows for improved detection accuracy compared to AD conversion that uses one cycle as a unit.

Here, the first AD converter and the second AD converter may generate the first digital data and the second digital data to correspond to at least three cycles of the waveform of the sine wave voltage, and the multiplication processing unit may compare the at least three cycles of the waveform in the first digital data on a cycle-by-cycle basis and discard the digital data for one cycle that deviates the most.

This allows, for example, digital data from a period affected by disturbance noise to be discarded, improving detection accuracy.

Here, the first AD converter and the second AD converter generate the first digital data and the second digital data so as to correspond to at least three periods of the waveform of the sine wave voltage, and the multiplication processing unit calculates an average value and a deviation for each period of the at least three periods of the waveform in the first digital data, and if the deviation is equal to or greater than a threshold value, discards the digital data for that period.

This allows data from periods that contain disturbance noise to be discarded, improving noise resistance and increasing accuracy.

Here, the multiplication processing unit may change the frequency of the rectangular wave signal when a predetermined number of consecutive periods occur in which the deviation is equal to or greater than the threshold value.

With this, when disturbance noise continues, the effect of the disturbance noise can be reduced by changing the frequency of the square wave signal and the sine wave voltage.

Here, the steering grip sensor further includes a monitoring unit that monitors for faults and an attenuation switch that attenuates or amplifies the sine wave voltage, and the monitoring unit has a normal monitoring mode that does not involve attenuation or amplification of the sine wave voltage by the attenuation switch, and a temporary monitoring mode that involves attenuation or amplification of the sine wave voltage, and the monitoring unit may monitor a ground fault of the driven electrode and a ground fault of the sensor electrode in the normal monitoring mode, and monitor a break in the driven electrode, a break in the sensor electrode, and a short circuit between the driven electrode and the sensor electrode in the temporary monitoring mode.

Here, the monitoring unit may control the attenuation rate and amplification rate in the attenuation switch.

This makes it possible to test the performance of the steering grip sensor by controlling the attenuation rate or amplification rate. It is also possible to set an appropriate attenuation rate or amplification rate according to the individual variations in the steering grip sensor.

Here, the steering grip sensor may further include a correction unit that performs a correction process to suppress fluctuations in the amount of charge generated in the sensor electrode due to environmental changes, and the multiplication processing unit may further shift the phase of the output voltage of the charge amplifier by 90 degrees, multiply the sine wave voltage by the shifted output voltage, and output the multiplication result to the correction unit as an index of the environmental change.

As a result, for example, if the resistance value of the insulator covering the sensor electrode has characteristics that change depending on humidity, the detection accuracy can be improved by performing a correction process that suppresses the fluctuation.

Here, the steering grip sensor further includes another charge amplifier having a feedback capacitance element that detects a change in the amount of charge generated in the driven electrode due to an environmental change and outputs the change in the amount of charge as a change in voltage, and a correction unit that performs a correction process to suppress fluctuations in the amount of charge generated in the driven electrode due to environmental changes, and the multiplication processing unit may further multiply the sine wave voltage by the output voltage of the other charge amplifier and output the multiplication result to the correction unit as an index of the environmental change.

As a result, for example, if the capacitance value of the insulator in contact with the driven electrode has a characteristic that varies depending on humidity or temperature, the detection accuracy can be improved by performing a correction process that suppresses the variation.

Here, the steering grip sensor further includes a current amplifier having a feedback resistor element and outputting a change in the current generated in the driven electrode as a change in voltage, and a correction unit performing a correction process to suppress fluctuations in the current generated in the driven electrode due to environmental changes, and the multiplication processing unit may further multiply the sine wave voltage by the output voltage of the current amplifier and output the multiplication result to the correction unit as an index of the environmental change.

As a result, for example, if the resistance value of the insulator in contact with the driven electrode has a characteristic that varies depending on humidity or temperature, the detection accuracy can be improved by performing a correction process that suppresses the variation.

Here, the low-pass filter may include a first resistive element, a second resistive element connected in series with the first resistive element, a first capacitive element connected to a connection point between the first resistive element and the second resistive element, and a third resistive element connected in series with the first capacitive element.

This makes it possible to limit the gain of the charge amplifier and suppress saturation. Furthermore, by providing a third resistor, it is possible to suppress the gain peak that occurs at a specific frequency (cutoff frequency).

Here, the steering grip sensor may further include a charge supply circuit that supplies a predetermined amount of charge to the charge amplifier in a fault monitoring operation mode, and the monitoring unit may monitor whether the operation of the charge amplifier is normal or not based on the output of the charge amplifier when no charge is being supplied from the charge supply circuit and the output of the charge amplifier when charge is being supplied from the charge supply circuit.

With this, the monitoring unit can monitor whether the charge amplifier is operating normally or not. Furthermore, it can monitor whether the differential amplifier is operating normally or not.

Here, a variable resistor may be provided between the connection point between the sensor electrode and the charge amplifier and ground.

Accordingly, the variable resistor enables correction to suppress fluctuations in the operation of the steering grip sensor due to environmental changes such as temperature and humidity.

In addition, a grip detection method according to one aspect of the present disclosure is a grip detection method for a steering grip sensor that includes a planar driven electrode that follows the rim of the steering wheel and a planar sensor electrode that faces the driven electrode, in which a sine wave voltage is supplied to the driven electrode, a charge amplifier having a feedback capacitance element detects a change in the amount of charge generated by the capacitance of the sensor electrode, the charge amplifier generates an output voltage that represents the detected change in the amount of charge as a change in voltage, the sine wave voltage is multiplied by the output voltage of the charge amplifier, the multiplication result is smoothed by integrating it, and the presence or absence of gripping the steering wheel is determined according to the level of the smoothed multiplication result.

This makes it possible to suppress deterioration in accuracy due to frequency fluctuations and noise, and improve detection accuracy.

These comprehensive or specific aspects may be realized as a system, method, integrated circuit, computer program, or computer-readable recording medium such as a CD-ROM, or may be realized as any combination of a system, method, integrated circuit, computer program, or recording medium.

The following describes the embodiment in detail with reference to the drawings.

The embodiments described below are all comprehensive or specific examples. The numerical values, shapes, materials, components, the arrangement and connection of the components, steps, and the order of steps shown in the following embodiments are merely examples and are not intended to limit the present disclosure.

(Embodiment 1)
[Appearance of the steering grip sensor 100]
FIG. 1 is a diagram showing an example of an interior layout of a vehicle 1 in which a steering grip sensor 100 according to an embodiment is arranged.

As shown in FIG. 1, the vehicle 1 is equipped with a steering wheel 3, a speaker, and a display device such as a liquid crystal display. The speaker and the display device are configured as, for example, an attention warning device.

The steering wheel 3 applies a steering angle to the steered wheels of the vehicle 1. The steering wheel 3 has a rim 31, roughly T-shaped spokes 32 formed integrally with the inner peripheral surface of the rim 31, and a horn switch cover that covers a horn switch located in the center of the spokes 32. The steering grip sensor 100 includes a steering cover 110 and a control device 120.

Figure 2 shows an example of the steering cover 110 of the steering grip sensor 100 in embodiment 1 attached to the rim 31.

As shown in FIG. 2, the steering grip sensor 100 is a device that detects the grip and contact of the steering cover 110 by the hand of the user (human body), and is provided on the rim 31 of the vehicle 1. Contact does not only mean that the user's hand is in direct contact with the steering cover 110, but also includes a state in which the steering grip sensor 100 is separated from the steering cover 110 as long as the steering grip sensor 100 can detect the human hand.

The steering grip sensor 100 is a capacitance type sensor that detects the grip of a user in a vehicle 1 having a steering wheel 3. Specifically, the steering grip sensor 100 detects whether the user's hand is touching the steering wheel 3 by detecting a change in capacitance between the user's hand and an electrode in the steering grip sensor 100.

[Cross-sectional configuration of steering cover 110]
3 is a diagram showing an example of a cross-sectional configuration of the steering cover 110 in the first embodiment. The diagram is a schematic enlarged view of a portion of a cross section of the steering cover 110 wrapped around the rim 31, cut in the lamination direction. As shown in FIG. 3, the steering cover 110 has a skin 11, a sensor electrode 112, a dielectric layer 12, a driven electrode 113, and a urethane layer 13. The rim 31 is grounded to the metal body of the vehicle 1.

The skin 11 is an insulating material such as leather, synthetic leather, or synthetic resin.

The steering cover 110 is not limited to a configuration in which the skin 11, sensor electrode 112, dielectric layer 12, driven electrode 113, and urethane layer 13 are wrapped around the rim 31 in an integrated state. For example, the steering cover 110 may be configured such that a detection unit in which the sensor electrode 112, dielectric layer 12, and driven electrode 113 are integrated is wrapped around the surface of the rim 31 on which the urethane layer 13 is formed, and the skin 11 is then wrapped around that surface.

The sensor electrode 112 is a planar electrode formed of, for example, a mesh-like cloth woven with conductive fibers. The shape of the sensor electrode 112 may be a curved surface that forms part of a ring, a curved surface that forms part of the side surface of a cylinder, or a collection of small flat surfaces. The sensor electrode 112 in Fig. 3 is a curved electrode that corresponds to all or part of a ring.

The driven electrode 113 may have a similar configuration to the sensor electrode 112, or may have a different configuration, for example, a metal wire sewn and fixed onto one side of the dielectric layer 12.

The sensor electrode 112 and the driven electrode 113 face each other with the dielectric layer 12 in between, forming a capacitive element. When the user touches the sensor electrode 112 with his/her hand, a capacitance ΔC is added to the sensor electrode 112, as shown by the dashed line in the figure. The steering grip sensor 100 detects the change in this capacitance ΔC to determine whether or not the hand is touching the sensor electrode.

[Circuit configuration of steering grip sensor 100]
FIG. 4 is a block diagram showing an example of a circuit configuration of the steering grip sensor 100 in the first embodiment.

[Configuration of steering grip sensor 100]
4 is a block diagram showing an example of a circuit configuration of the control device 120 in the steering grip sensor 100 according to the embodiment 1. However, in this figure, the sensor electrode 112 and the driven electrode 113 of the steering cover 110 are added in a schematic manner.

The control device 120 in FIG. 4 includes a sensor circuit 40 and a control processor 50.

The sensor circuit 40 has a sensor terminal T4, a drive terminal T5, a sine wave generating unit 41, a charge amplifier 44, and a differential amplifier 46.

The sensor terminal T4 is connected to the sensor electrode 112 and the second input terminal I2 of the charge amplifier 44, and is a terminal that simultaneously performs input and output. Specifically, the sensor terminal T4 transmits the sinusoidal voltage fed back by the charge amplifier 44 to the sensor electrode 112. At the same time, the sensor terminal T4 transmits to the charge amplifier 44 the change in the amount of charge that occurs due to the change in the capacitance of the sensor electrode 112.

The drive terminal T5 is a terminal connected to the driven electrode 113 and the sine wave generating unit 41. The drive terminal T5 transmits a sine wave voltage as a drive signal from the sine wave generating unit 41 to the driven electrode 113.

The sine wave generating unit 41 inputs a rectangular wave signal from the control unit 51, generates a sine wave voltage synchronized with the rectangular wave signal, and supplies the sine wave voltage as a drive signal to the driven electrode 113 via the drive terminal T5. The drive signal input from the control unit 51 is, for example, a PWM (Pulse Width Modulation) signal. Note that in this embodiment, the sine wave voltage supplied from the sine wave generating unit 41 to the driven electrode 113 may be a DC voltage (for example, 2 V) with a sine wave superimposed thereon.

The charge amplifier 44 has a feedback capacitance element C11, detects a change in the amount of charge generated by the capacitance of the sensor electrode 112, and outputs the change in the amount of charge as a change in voltage. Therefore, the charge amplifier 44 includes a first input terminal I1, a second input terminal I2, an output terminal o1, a feedback capacitance element C11, an operational amplifier A11, resistance elements R11 and R12, and a capacitance element C12. The first input terminal I1 inputs a sine wave voltage from the sine wave generating unit 41 and is connected to the non-inverting input of the operational amplifier A11.

The second input terminal I2 is connected to the sensor terminal T4 and transmits the change in the amount of charge generated by the change in capacitance of the sensor electrode 112 to the inverting input of the operational amplifier A11.

The output terminal o1 outputs a voltage corresponding to the change in the amount of charge generated by the capacitance of the sensor electrode 112. For example, when there is no change in the capacitance of the sensor electrode 112, it outputs a sine wave voltage that is in phase with and has the same amplitude as the sine wave voltage of the first input terminal I1. On the other hand, when there is a change in the capacitance of the sensor electrode 112, it outputs a sine wave voltage that is in phase with the sine wave voltage of the first input terminal I1 and has an amplitude that corresponds to the change in capacitance.

The feedback capacitance element C11 is connected between the output terminal o1 and the second input terminal I2, and feeds back the output voltage from the output terminal o1 to the second input terminal I2. The feedback capacitance element C11 also has the function of detecting a change in the amount of charge generated by a change in the capacitance of the sensor electrode 112.

The operational amplifier A11 outputs an output voltage to the second input terminal I2 via the feedback capacitance element C11, so that the difference between the non-inverting input and the inverting input becomes zero.

The differential amplifier 46 amplifies the difference between the sine wave voltage from the sine wave generator 41 and the output voltage of the charge amplifier 44, and outputs the amplified result as a sense signal to the ADC 62 via the AD terminal Ta.

The circuit consisting of the resistor elements R11, R12, and the capacitor element C12 constitutes a low-pass filter with a cutoff frequency set lower than the frequency of the sine wave voltage. This low-pass filter is connected in parallel with the feedback capacitor element C11, and at frequencies lower than the frequency of the sine wave voltage, it acts more dominantly than the capacitor element C11 to pass and feed back the output voltage o1. As a result, the low-pass filter limits the gain of the operational amplifier A11 to prevent saturation and suppress noise components. On the other hand, at frequencies higher than the cutoff frequency, the feedback is blocked by the action of the capacitor element C12 and the resistor elements R11 and R12. As a result, the capacitor element C11 acts dominantly to pass the output voltage o1. In other words, at frequency components lower than the frequency of the sine wave, the low-pass filter limits the gain to prevent saturation and suppress noise components. Then, at frequency components higher than the frequency of the sine wave, the operational amplifier A11 switches to charge amplification by only the capacitor element C11.

More specifically, as shown in FIG. 4, the sensor electrode 112 is connected to the GND via a resistor Rx. This resistor Rx is a parasitic resistance based on the resistance component of the dielectric layer 12 that is present between the sensor electrode 112 and the GND. As described above, when the sine wave voltage supplied to the driven electrode 113 is a DC voltage with a sine wave superimposed thereon, a DC current flows to the GND via the resistor Rx. Here, when the resistivity of the dielectric layer 12 changes due to a change in temperature or humidity, the value of the DC current flowing to the GND via the resistor Rx changes. In order to reduce the effect of such changes in the DC component, a low-pass filter with a cutoff frequency set lower than the frequency of the sine wave voltage is provided to limit the gain of the operational amplifier A11 and suppress saturation.

In FIG. 4, the low-pass filter is composed of resistor elements R11 and R12 connected in series, and a capacitor element C12 provided between the connection point between resistor elements R11 and R12 and the non-inverting input of operational amplifier A11. However, as shown in FIG. 9, in addition to these, a resistor R21 may be provided in series with capacitor element C12. By providing resistor R21, it is also possible to suppress the gain peak that occurs at a specific frequency (cutoff frequency).

In FIG. 4, the control processor 50 has an AD terminal Ta, an AD terminal Tb, a PWM terminal Tf, a control unit 51, a multiplication processing unit 52, an integrator 54, a grip determination unit 56, and a memory 57. The multiplication processing unit 52 has ADCs 62 and 63, and a multiplier 65. The control processor 50 may be configured by a microcomputer having a CPU, memory 57, ADCs 62 and 63, a multiplier 65, and an input/output port. The microcomputer is an abbreviation for microcomputer or microcontroller. The memory 57 is a general term for ROM, RAM, electrically erasable flash memory, etc. The functional blocks in FIG. 4 can be realized by the CPU executing a program in the memory 57. The multiplier 65 may be a hardware circuit, or may be realized in part or in whole by software.

In the multiplication processing unit 52, the first digital data 72, the second digital data 73, and the first multiplication data 67 are illustrated as data to be processed. These data are temporarily stored in the memory 57, for example.

The AD terminal Ta is connected to the output terminal of the differential amplifier 46 and is a terminal for inputting an analog signal to the ADC 62.

The AD terminal Tb is connected to a terminal that outputs the sine wave voltage of the sine wave generating unit 41, and is a terminal for inputting an analog signal to the ADC 63.

The PWM terminal Tf is connected to the control unit 51 and the sine wave generating unit 41, and is a terminal for transmitting a rectangular wave signal from the control unit 51 to the sine wave generating unit 41. The rectangular wave signal may be a PWM signal.

The control unit 51 generates a variable frequency square wave signal and outputs it to the sine wave generating unit 41 via the PWM terminal Tf. The control unit 51 also controls the ADCs 62 and 63 to sample the analog signal in synchronization with the square wave signal. The number of samples may be, for example, 16 times per period of the sine wave voltage or sense signal, or may be about 8 to 24 times.

The multiplication processing unit 52 multiplies the sine wave voltage from the sine wave generating unit 41 by the output voltage of the charge amplifier 44. The sine wave voltage from the sine wave generating unit 41 is input to the AD terminal Tb. The output voltage of the charge amplifier 44 is input to the AD terminal Ta as a sense signal via the differential amplifier 46. The result of this multiplication corresponds to the magnitude of the capacitance due to gripping of the sensor electrode 112.

The ADC 62 converts the sense signal from the differential amplifier 46 into a digital signal in synchronization with the rectangular wave signal from the control unit 51 as the output voltage of the charge amplifier 44, and stores the AD conversion result in the memory 57 as first digital data 72. The first digital data 72 corresponds to the N-period waveform of the sense signal. Here, N may be 1, 2, 3 or more.

The ADC 63 performs AD conversion on the sine wave voltage from the sine wave generating unit 41 in synchronization with the rectangular wave signal from the control unit 51, and stores the AD conversion result in the memory 57 as second digital data 73. The second digital data 73 corresponds to at least N periods of the sine wave voltage. N is the same as the first digital data 72.

The multiplier 65 multiplies the first digital data 72 and the second digital data 73 together, and stores the multiplication result in the memory 57 as the first multiplied data 67.

The multiplication processing unit 52 may perform a process of averaging the waveform indicated by the first digital data 72 and the waveform indicated by the second digital data 73, and may multiply the averaged first digital data by the averaged second digital data, and use the multiplication result as the first multiplied data 67. In this case, the averaging process may be performed by the integrator 54, instead of the multiplication processing unit 52.

Alternatively, the multiplication processing unit 52 may multiply the first digital data 72 and the second digital data 73 together, and perform a process of averaging the waveform indicated by the first multiplication data 67 resulting from the multiplication. In this case, the averaging process may be performed by the integrator 54, rather than by the multiplication processing unit 52.

The integrator 54 averages, or smoothes, the multiplication result of the multiplication processing unit 52 by integrating it.

The grip determination unit 56 determines whether the steering wheel is being gripped according to the level of the smoothed multiplication result. The level of the smoothed multiplication result corresponds to the capacitance of the sensor electrode 112, and is, for example, larger when the steering wheel is being gripped than when the steering wheel is not being gripped. The grip determination unit 56 may have hysteresis in determining whether the steering wheel is being gripped. Specifically, when the level of the smoothed multiplication result becomes equal to or greater than the first threshold value th1, the grip determination unit 56 determines that the steering wheel has changed from a no-grip state to a gripped state. On the other hand, when the level of the smoothed multiplication result becomes equal to or less than the second threshold value th2, the grip determination unit 56 determines that the steering wheel has changed from a gripped state to a no-grip state. In this case, if the first threshold value th1 is greater than the second threshold value th2, hysteresis can be provided.

[Comparison of steering grip sensor 100 and comparative example]
Next, the steering grip sensor 100 will be described in comparison with a comparative example.

Figure 5 is an explanatory diagram of the operation of a comparative sensor compared to the steering grip sensor 100 in the first embodiment. The comparative sensor in the figure shows a similar configuration based on the sensor in Patent Document 1.

In the circuit configuration of the comparative example, the sensor electrode SE is connected to the inverting input of the operational amplifier A41. The driven electrode DE is connected to the non-inverting input, and a sinusoidal voltage is input.

The current amplifier 47 has a feedback resistor element R, detects the current Δi that flows due to the change ΔC in the capacitance of the sensor electrode SE, and outputs a voltage corresponding to the detected current. In other words, the current amplifier is a current-voltage converter that outputs the detected current as a voltage.

As shown in the "Formula" column of the comparative example, the output voltage V is expressed as V = R x Δi0. Δi0 is the feedback current flowing through the feedback resistor R, and Δi0 is equal to Δi in terms of a virtual short between the non-inverting input and the inverting input. The current Δi flowing due to the change ΔC in the capacitance of the sensor electrode SE is expressed as Δi = ω x ΔC x v. Here, v is the sinusoidal voltage supplied to the non-inverting input and the driven electrode DE. The output voltage V of the current amplifier 47 is expressed as V = ω x (ΔC x v) x R. Here, ω = 2πf. f is the frequency of the sinusoidal voltage v. The output voltage V of the comparative example is proportional not only to the capacitance ΔC but also to the frequency f.

As shown in the "Signal Waveform" column of the comparative example, the output voltage V has a phase shift of 90° with the sinusoidal voltage v. In the "Signal Waveform" column of the comparative example, two types of waveform examples are shown as detection examples of the output voltage V: synchronous switch detection for detecting changes in the resistance component, and phase-shift switch detection for detecting the capacitance component. Both types of detection involve switching operations.

As a result, the comparative example has the following problems. First, because a switching operation is required to detect the output voltage V, if degradation occurs in the slew rate of the edge component of the control signal that controls the switching, distortion or detection phase shift occurs in the signal after synchronous detection, resulting in a problem of degradation in detection accuracy.

Secondly, because the sensitivity of the output voltage V depends on the frequency f, there is a problem that if frequency fluctuations occur in the drive signal, the detection accuracy deteriorates.

Thirdly, the higher the frequency f, the greater the output voltage V, which creates a problem in that the higher the frequency of the external high-frequency noise, the worse the detection accuracy becomes.

In contrast, the charge amplifier 44 in the "Circuit Configuration" column of the "Embodiment" in FIG. 5 detects a change ΔQ in the amount of charge generated by the capacitance ΔC of the sensor electrode 112, as already explained in FIG. 4, and outputs the change ΔQ in the amount of charge as a change in the output voltage V. ΔQ0 is a feedback charge that generates the feedback capacitance element C, and ΔQ0 is equal to ΔQ from the viewpoint of a virtual short between the non-inverting input and the inverting input. The output voltage V of the charge amplifier 44 is expressed as V=(ΔC/C)×v. C is the capacitance value of the feedback capacitance element. This output voltage V is proportional to the capacitance ΔC and does not depend on the frequency f. Also, as shown in the "Signal Waveform" column, the output voltage V and the sine wave voltage v have a phase difference of 0, that is, they are in phase. Therefore, instead of synchronous detection by switching operation, multiplication detection in which the output voltage V and the sine wave voltage v are multiplied is possible. In the "Signal Waveform" column, first multiplication data 67 and second multiplication data 68 are shown as examples of multiplication detection. The first multiplication data 67 indicates a result of multiplying the output voltage V by the sine wave voltage v, and represents the capacitance ΔC of the sensor electrode 112. The second multiplication data 68 indicates a result of multiplying a waveform obtained by shifting the phase of the output voltage V by the sine wave voltage v, and represents the resistance component of the sensor electrode 112.

In the steering grip sensor 100 of the first embodiment, the first problem is solved by multiplication detection. The output voltage V of the charge amplifier 44 does not have the frequency dependency of the sine wave voltage v, so the second and third problems are solved.

As described above, the steering grip sensor 100 according to the first embodiment includes a planar driven electrode 113 that follows the rim of the steering wheel, a planar sensor electrode 112 that faces the driven electrode 113, a sine wave generator 41 that supplies a sine wave voltage v to the driven electrode 113, a charge amplifier 44 that has a feedback capacitance element C11, detects a change in the amount of charge generated by the capacitance of the sensor electrode 112, and outputs the change in the amount of charge as a change in voltage, a multiplication processing unit 52 that multiplies the sine wave voltage v by the output voltage V of the charge amplifier 44, an integrator 54 that smoothes the multiplication result of the multiplication processing unit 52 by integrating it, and a grip determination unit 56 that determines whether or not the steering wheel is being gripped depending on the level of the smoothed multiplication result.

This configuration can reduce deterioration in accuracy due to frequency fluctuations and noise, and improve detection accuracy.

Specifically, firstly, the output voltage of the charge amplifier is in phase with the sinusoidal voltage as the drive signal, i.e. the phase difference is zero, and the detection of the capacitance due to grasping is performed by multiplication detection in the multiplication processing unit. As a result, since there is no need for switching operations as in conventional technology, phase shifts and signal distortion due to switching can be suppressed, deterioration of detection accuracy can be suppressed, and detection accuracy can be improved.

Secondly, the sensitivity of the output voltage of the charge amplifier does not depend on the frequency of the sine wave voltage used as the drive signal, so even if frequency fluctuations occur in the sine wave signal, deterioration of detection accuracy can be suppressed.

Thirdly, the magnitude of the output voltage of the charge amplifier does not depend on the frequency of the sinusoidal voltage used as the drive signal, so degradation of detection accuracy caused by external high-frequency noise can be suppressed.

In addition, in the steering grip sensor 100 shown in FIG. 4, the control processor 50 may switch the square wave signal to at least three frequencies, and the ADC 62 and the ADC 63 may store the first digital data and the second digital data in the memory 57 for each of the at least three frequencies.

This makes it possible to use the first and second digital data of other frequencies without using the first and second digital data corresponding to the frequency affected by the disturbance noise. It also makes it possible to perform frequency hopping so as to avoid the disturbance noise.

In addition, ADC62 and ADC63 may generate the first digital data and the second digital data to correspond to at least two periods of the waveform of the sinusoidal voltage.

This allows for improved detection accuracy compared to AD conversion that uses one cycle as a unit.

The ADC 62 and ADC 63 may generate the first digital data and the second digital data to correspond to at least three cycles of the waveform of the sine wave voltage, and the multiplication processing unit 52 may compare at least three cycles of the waveform in the first digital data on a cycle-by-cycle basis, and discard the digital data for the cycle with the most deviating value.

By using data from periods other than the period with the most deviant values, for example, digital data from periods affected by external noise can be discarded, improving detection accuracy.

In addition, ADC62 and ADC63 generate first digital data and second digital data to correspond to at least three cycles of the waveform of the sine wave voltage, and multiplication processing unit 52 calculates the average value and deviation for each cycle of the waveform of at least three cycles in the first digital data, and if the deviation is equal to or greater than a threshold value, the digital data for that cycle may be discarded.

This allows data from periods that contain disturbance noise to be discarded, improving noise resistance and increasing accuracy.

Furthermore, the multiplication processing unit 52 may change the frequency of the square wave signal if a predetermined number of consecutive periods occur in which the deviation is greater than or equal to the threshold value.

With this, when disturbance noise continues, the effect of the disturbance noise can be reduced by changing the frequency of the square wave signal and the sine wave voltage.

(Embodiment 2)
In the second embodiment, in addition to the first embodiment, a configuration is described that performs a correction process for suppressing fluctuations in the operation of the steering grip sensor 100 due to environmental changes, and fault monitoring for monitoring whether the operation is normal or not.

The above-mentioned environmental changes specifically include changes in the resistance value of the skin 11 in FIG. 3, changes in the capacitance of the driven electrode 113, changes in the resistance value of the urethane layer 13, etc.

For example, the resistance value of the skin 11 has a temperature/humidity dependent characteristic that depends on the temperature and humidity of the skin 11. By utilizing this temperature/humidity dependent characteristic of the skin 11, the steering grip sensor 100 detects resistance changes in the sensor electrode 112 and the skin 11, thereby essentially detecting temperature and humidity without a temperature/humidity sensor. The detected resistance change can be used to make corrections to the temperature/humidity characteristics. Here, temperature/humidity is defined to mean both temperature and humidity, or either temperature or humidity. This definition depends on the material of the skin 11.

The driven electrode 113 faces the rim 31 with the urethane layer 13 in between, or is wrapped around the rim 31 with the urethane layer 13 in between. Therefore, the driven electrode 113 and the rim 31 form a capacitor. Since the rim 31 is grounded, the capacitor formed by the driven electrode 113 and the rim 31 has a constant capacitance value. However, this capacitance value has a characteristic that changes depending on the temperature and humidity of the urethane layer 13. Using this temperature and humidity dependency characteristic, the steering grip sensor 100 detects the temperature and humidity of the driven electrode 113 and the urethane layer 13 by detecting the change in capacitance of the driven electrode 113. This makes it possible to make corrections for temperature and humidity characteristics even without a temperature and humidity sensor.

Figure 6 is a block diagram showing an example of the circuit configuration of the steering grip sensor 100 in the second embodiment. The main differences compared to Figure 4 are that sensor terminals T1 to T3, attenuation switch 42, charge amplifier 43, multiplexer 45, selection control terminal Tc, AD terminal Td, AD terminal Te, CHK terminal Tg, output monitoring terminal Th, ADC 60, timing generation unit 61, ADC 64, third digital data 74, multiplier 66, second multiplication data 68, third multiplication data 69, fault monitoring unit 53, and correction unit 55 have been added, and that a bandpass filter has been added to the differential amplifier 46.

In the following, we will avoid repeating the same points and focus on the differences.

Sensor terminals T1 to T4 are terminals that connect the four sensor electrodes 112 when four pairs of sensor electrodes 112 and driven electrodes 113 are provided inside the steering cover 110.

The attenuation switch 42 switches between attenuating or amplifying the sine wave voltage supplied from the sine wave generator 41 to the driven electrode 113 via the charge amplifier 43, or passing it through without attenuation or amplification.

The charge amplifier 43 has a feedback capacitance element C21, detects a change in the amount of charge generated in the driven electrode 113 due to an environmental change, and outputs the change in the amount of charge as a change in voltage. Therefore, the charge amplifier 43 includes a feedback capacitance element C21, an operational amplifier A21, resistance elements R21 and R22, and a capacitance element C12. This charge amplifier 43 is similar to the charge amplifier 44 already described. Note that the steering grip sensor 100 may be equipped with a current amplifier 47 shown in FIG. 5 instead of the charge amplifier 43. In this case, the current amplifier 47 detects a change in the current generated in the driven electrode 113 due to an environmental change, and outputs the change in the current as a change in voltage.

The multiplexer 45 selects one of the sensor terminals T1 to T4 and connects it to the second input terminal I2 of the charge amplifier 44. This selection is made according to a selection control signal input from the control unit 51 via the selection control terminal Tc. Note that the multiplexer 45 does not have to be configured as a 4:1 connection, but may be configured to switch between M:1 connections. M may be an integer of 2, 3, or 5 or more, and is an integer greater than the number of pairs of sensor electrodes 112 and driven electrodes 113.

In the configuration of FIG. 6, the driven electrodes 113 are arranged in pairs and are all electrically connected. Therefore, the driven electrodes 113 are connected to a single drive terminal T5. However, this configuration is not limited to this, and for example, the driven electrodes 113 of each pair may be selected by another multiplexer and connected to the drive terminal T5. In this case, the driven electrode 113 that pairs with the sensor electrode 112 selected by the multiplexer 45 may be selected by the other multiplexer.

The selection control terminal Tc is connected to the control terminal of the multiplexer 45 and receives the selection control signal output from the control unit 51.

The AD terminal Td is connected to the output terminal of the charge amplifier 43 and is a terminal for inputting an analog signal to the ADC 64.

The AD terminal Te is connected to the sine wave generating unit 41 and is a terminal for inputting the sine wave voltage of an analog signal to the ADC 60.

The CHK terminal Tg is a terminal for transmitting a check signal from the fault monitoring unit 53 to the attenuation switch 42. The check signal is at a low level in the normal monitoring mode in which the attenuation switch 42 does not attenuate or amplify the sine wave voltage, and is at a high level in the temporary monitoring mode in which the sine wave voltage is attenuated or amplified.

The output monitoring terminal Th is a terminal for transmitting the output signal from the output terminal o1 to the fault monitoring unit 53.

The ADC 60 performs AD conversion on the sine wave voltage from the sine wave generating unit 41 and outputs the AD conversion result to the timing generating unit 61. This AD conversion result has a different purpose from the AD conversion result of the ADC 63. The AD conversion result of the ADC 60 is used to generate a timing signal synchronized with the waveform of the sine wave voltage. Therefore, the ADC 60 samples at a higher rate than the ADC 63.

The timing generation unit 61 generates a timing signal synchronized with the waveform of the sine wave voltage based on the AD conversion result of the ADC 60. This timing signal includes the sampling clock signals of the ADCs 62-63. The timing generation unit 61 may generate the timing signal so as to synchronize with the square wave signal (i.e., the PWM signal) from the control unit 51. In this case, the ADC 60 does not need to be provided.

The ADC 64 converts the output voltage of the charge amplifier 43 into a digital signal in synchronization with the timing signal from the timing generation unit 61, and stores the result of the digital conversion in the memory 57 as third digital data 74.

The third digital data 74 corresponds to the N-period waveform of the output voltage of the charge amplifier 43, where N is the same as the number of periods contained in the first digital data 72.

As in FIG. 4, the multiplier 65 multiplies the first digital data 72 and the second digital data 73 together, and stores the multiplication result in the memory 57 as the first multiplication data 67. The first multiplication data 67 corresponds to the capacitance ΔC of the sensor electrode 112.

In addition, the multiplier 65 shifts the phase of the waveform of the first digital data 72 by 90°, multiplies the shifted first digital data 72 by the second digital data 73, and stores the multiplication result in the memory 57 as the second multiplied data 68.

The second multiplication data 68 corresponds to a resistance component that depends on environmental changes of the sensor electrode 112, for example, the temperature and humidity of the epidermis 11.

The multiplier 66 multiplies the second digital data 73 and the third digital data 74 together, and stores the multiplication result in the memory 57 as the third multiplied data 69.

The third multiplication data 69 corresponds to the capacitance value of the urethane layer 13, which varies depending on the environmental changes of the driven electrode 113, such as temperature and humidity.

The correction unit 55 performs a correction process to suppress fluctuations due to environmental changes, using the second multiplication data 68 and the third multiplication data 69 as indexes of environmental changes. As the correction process, the correction unit 55 corrects, for example, the threshold values th1 and th2 used for judgment in the grip judgment unit 56. As the correction process, the correction unit 55 may also correct the level smoothed by the integrator 54. Note that the correction unit 55 may smooth the second multiplication data 68 and the third multiplication data 69, and use the smoothed second multiplication data 68 and the smoothed third multiplication data 69 as indexes.

By the correction process, for example, when the resistance value of the surface layer 11, which is an insulator covering the sensor electrode 112, has a characteristic that changes depending on temperature and humidity, the change can be suppressed to improve the detection accuracy. In this example, specifically, a change in the insulation resistance of the surface layer 11 caused by sweat from the palm gripping the steering wheel penetrating into the surface layer 11 depending on the air temperature can be exemplified. Also, for example, when the capacitance value of the urethane layer 13, which is an insulator in contact with the driven electrode 113, has a characteristic that changes depending on humidity or temperature, the detection accuracy can be improved by the correction process that suppresses the change. In this example, specifically, a change in the dielectric constant of the urethane layer 13 caused by the urethane layer 13 absorbing moisture based on the air temperature and humidity around the steering wheel can be exemplified.

The fault monitoring unit 53 has a normal monitoring mode that does not involve attenuation or amplification of the sine wave voltage by the attenuation switch 42, and a temporary monitoring mode that involves attenuation or amplification of the sine wave voltage, and diagnoses the presence or absence of a fault. In the normal monitoring mode, the fault monitoring unit 53 diagnoses the presence or absence of a ground fault in the driven electrode 113 and the sensor electrode 112. In the temporary monitoring mode, the fault monitoring unit 53 diagnoses the presence or absence of a break in the driven electrode 113, the break in the sensor electrode 112, and a short circuit between the driven electrode 113 and the sensor electrode 112. There are two types of breaks in the driven electrode 113: a break in the cable connected to the driven electrode 113, and an electrode break caused by a break in the planar driven electrode 113. There are two types of breaks in the sensor electrode 112: a break in the cable connected to the sensor electrode 112, and an electrode break caused by a break in the planar sensor electrode 112. In the following explanation, when the word "disconnection" is used, it refers to both types of disconnection.

Next, we will explain fault diagnosis by the fault monitoring unit 53.

Figure 7 is an explanatory diagram of fault diagnosis in embodiment 2. Also, Figure 8 is a diagram showing various waveforms during fault diagnosis in embodiment 2.

The horizontal columns in Figure 7 indicate the types of faults detected by fault diagnosis. The vertical columns indicate the signal changes during a fault and whether fault diagnosis is possible for each fault monitoring mode (normal monitoring mode and temporary monitoring mode). Note that the normal monitoring mode is a mode in which the steering grip sensor 100 monitors for faults while normally operating to detect gripping. The temporary monitoring mode is a mode in which the operation to detect gripping is temporarily stopped and fault monitoring is enabled.

The fault monitoring unit 53 diagnoses the driven electrode 113 for the presence or absence of any of the following: a break, a ground fault (i.e., a short circuit to GND), and a short circuit to the sensor electrode 112. However, it cannot diagnose the type of fault, whether it is a break, a ground fault (i.e., a short circuit to GND), or a short circuit to the sensor electrode 112.

Fault monitoring unit 53 also diagnoses whether the sensor electrode 112 has a break, a ground fault (i.e., a short circuit to GND), or a short circuit to the driven electrode 113. However, it cannot diagnose the type of fault, whether it is a break, a ground fault (i.e., a short circuit to GND), or a short circuit to the driven electrode 113.

Figure 8 shows the CHK signal, the sine wave voltage v, the outputs of the charge amplifiers 43 and 44, the output of the differential amplifier 46, and the output of the multiplication processing unit 52 (multiplication result). Note that Figure 8 shows an example in which the temporary monitoring mode is activated when the CHK signal is at a high level, and the sine wave voltage v is attenuated.

Below, specific examples of each fault diagnosis by the fault monitoring unit 53 will be described with reference to Figures 7 and 8.

First, as shown in FIG. 7, cable breaks cannot be detected in the normal monitoring mode for both the sensor electrode 112 and the driven electrode 113, so as shown in FIG. 8, the fault monitoring unit 53 switches to the temporary monitoring mode as indicated by the CHK signal at a predetermined timing (e.g., once every 0.3 seconds) during grip detection. If the output of the differential amplifier 46 in the temporary monitoring mode shows a smaller change in amplitude than the output in the temporary monitoring mode under normal conditions, the fault monitoring unit 53 determines that the cable is broken.

Incidentally, instead of the output of the differential amplifier 46, the output (multiplication result) of the multiplication processing unit 52 in the temporary monitoring mode may be taken in, and if the multiplication result approaches A compared with the output (multiplication result) in the temporary monitoring mode under normal conditions, the fault monitoring unit 53 may determine that the cable is broken. Here, A is defined as the multiplication result in the normal non-gripping state.

Next, as shown in FIG. 7, electrode breakage cannot be detected in the normal monitoring mode for both the sensor electrode 112 and the driven electrode 113. Therefore, as shown in FIG. 8, the fault monitoring unit 53 switches to the temporary monitoring mode as indicated by the CHK signal at a predetermined timing (e.g., once every 0.3 seconds) during grip detection. If the output of the differential amplifier 46 in the temporary monitoring mode has a reduced amplitude compared to the output in the temporary monitoring mode under normal conditions, the fault monitoring unit 53 determines that the electrode is broken. Here, the reduction in amplitude means that the amplitude is smaller than the normal amplitude in the temporary monitoring mode in the output of the differential amplifier 46 in FIG. 8. The degree to which the amplitude is reduced varies depending on the position of the breakage due to the breakage of the sensor electrode 112 and the driven electrode 113, and the closer the electrode breakage is to the cable, the smaller the amplitude becomes.

In addition, instead of the output of the differential amplifier 46 in the temporary monitoring mode, the output (multiplication result) of the multiplication processing unit 52 may be taken in, and if the multiplication result is smaller than the output (multiplication result) in the temporary monitoring mode under normal conditions, the fault monitoring unit 53 may determine that the electrode is broken. Here, the degree to which the multiplication result is smaller varies depending on the position of the break caused by the breakage of the sensor electrode 112 and the driven electrode 113, and the closer the electrode break is to the cable, the smaller the multiplication result.

Next, as shown in FIG. 7, when determining whether the sensor electrode 112 and the driven electrode 113 are shorted to GND, detection is possible in normal monitoring mode, so as shown in FIG. 8, the fault monitoring unit 53 determines whether the output of the charge amplifiers 43 and 44 is lower than normal during normal grip detection. At this time, in order to improve the accuracy of the determination, the fault monitoring unit 53 may average the outputs of the charge amplifiers 43 and 44, and determine that the sensor electrode 112 and the driven electrode 113 are shorted to GND when the average value is offset to a lower value compared to normal.

Note that since a GND short can be detected even in the temporary monitoring mode, the fault monitoring unit 53 may determine whether a GND short has occurred even in the temporary monitoring mode.

Next, as shown in FIG. 7, when determining whether the sensor electrode 112 and the driven electrode 113 are shorted, detection is possible in normal monitoring mode, as in the case of determining a GND short, so as shown in FIG. 8, the fault monitoring unit 53 determines whether the output of the charge amplifier 43 or the charge amplifier 44 is lower than normal during normal grip detection. At this time, in order to improve the accuracy of the determination, the fault monitoring unit 53 may average the output of the charge amplifier 43 or the charge amplifier 44, and determine that the sensor electrode 112 and the driven electrode 113 are shorted when the average value is offset to a lower value than normal.

In addition, since a short circuit between the sensor electrode 112 and the driven electrode 113 can be detected even in the temporary monitoring mode, the fault monitoring unit 53 may determine a short circuit between the sensor electrode 112 and the driven electrode 113 even in the temporary monitoring mode.

Here, (*1) in FIG. 7 and FIG. 8 means that in the temporary monitoring mode, diagnosis is made based on the change in amplitude of the output (sense signal) from the differential amplifier 46. (*2) means that in either the normal monitoring mode or the temporary monitoring mode, diagnosis is made by offsetting the output (sense signal) of the charge amplifier 44 or the output (driven signal) of the charge amplifier 43, or the average value thereof, at the time of failure (abnormality). The significance of making the diagnosis of (*1) is as follows. In the normal monitoring mode, the level of the output (sense signal) of the differential amplifier 46 at the time of failure hardly changes, making it difficult to diagnose the failure. In contrast, in the temporary monitoring mode, the failure monitoring unit 53 attenuates or amplifies the sine wave voltage, so that the amplitude (level change) that should be generated is reduced at the time of failure (abnormality), making it easier to diagnose the failure. Note that in the temporary monitoring mode, the output (multiplication result) of the multiplication processing unit 52 is also smaller at the time of failure (abnormality) than at normal time, so that the failure monitoring unit 53 may perform the failure diagnosis.

As described above, the steering grip sensor 100 according to the second embodiment further includes a fault monitoring unit 53 that monitors faults and an attenuation switch 42 that attenuates or amplifies the sine wave voltage. The fault monitoring unit 53 has a normal monitoring mode that does not involve attenuation or amplification of the sine wave voltage by the attenuation switch 42, and a temporary monitoring mode that involves attenuation or amplification of the sine wave voltage. In the normal monitoring mode, the fault monitoring unit 53 monitors for ground faults in the driven electrode 113 and for ground faults in the sensor electrode 112, and in the temporary monitoring mode, it monitors for breaks in the driven electrode 113, breaks in the sensor electrode 112, and short circuits between the driven electrode 113 and the sensor electrode 112.

Here, the fault monitoring unit 53 may control the attenuation rate and amplification rate in the attenuation switch 42.

This makes it possible to test the performance of the steering grip sensor by controlling the attenuation rate or amplification rate. It is also possible to set an appropriate attenuation rate or amplification rate according to the individual variations in the steering grip sensor.

Here, the steering grip sensor 100 is equipped with a correction unit 55 that performs a correction process to suppress fluctuations in the amount of charge generated in the sensor electrode 112 due to environmental changes, and the multiplication processing unit 52 may further shift the phase of the output voltage of the charge amplifier 44 by 90 degrees, multiply the sinusoidal voltage by the shifted output voltage, and output the multiplication result to the correction unit 55 as an index of environmental changes.

As a result, for example, if the resistance value of the skin 11, which is an insulator covering the sensor electrode 112, has the characteristic of changing depending on temperature and humidity, the detection accuracy can be improved by performing a correction process that suppresses the fluctuation.

Here, the steering grip sensor 100 further includes another charge amplifier 43 having a feedback capacitance element C21, which detects a change in the amount of charge generated in the driven electrode 113 due to an environmental change and outputs the change in the amount of charge as a change in voltage, and a correction unit 55 that performs a correction process to suppress fluctuations in the amount of charge generated in the driven electrode 113 due to environmental changes, and the multiplication processing unit 52 may further multiply the sine wave voltage by the output voltage of the other charge amplifier 43 and output the multiplication result to the correction unit 55 as an index of the environmental change.

As a result, for example, if the capacitance value of the urethane layer 13, which is an insulator in contact with the driven electrode 113, has the characteristic of fluctuating depending on humidity or temperature, the detection accuracy can be improved by performing a correction process that suppresses the fluctuation.

Here, the steering grip sensor 100 further includes a current amplifier 47 having a feedback resistor element R that outputs a change in the current generated in the driven electrode 113 as a change in voltage, and a correction unit that performs a correction process to suppress fluctuations in the current generated in the driven electrode 113 due to environmental changes. The multiplication processing unit 52 may further multiply the sine wave voltage by the output voltage of the current amplifier 47 and output the multiplication result to the correction unit 55 as an index of the environmental change.

According to this, for example, when the resistance value of the insulator in contact with the driven electrode 113 has a characteristic that varies depending on the humidity or temperature, the detection accuracy can be improved by a correction process that suppresses the variation.

(Embodiment 3)
In the third embodiment, in addition to the second embodiment, a configuration will be described in which failure monitoring is performed to monitor whether or not charge amplifier 44 and differential amplifier 46 are operating normally.

Figure 10 is a block diagram showing an example of the circuit configuration of the steering grip sensor 100 in embodiment 3. The main difference between this figure and Figure 6 is that a charge supply circuit 48 and a CHK terminal Ti have been added.

In the following, we will avoid repeating the same points and focus on the differences.

The charge supply circuit 48 consists of a resistor R43, a capacitor C41, and a switch S41 connected in parallel.

One end of the parallel circuit consisting of resistor R43 and capacitor C41 is connected to one end of switch S41. The other end of this parallel circuit is connected to the ground line.

One end of the switch S41 is connected to one end of a parallel circuit of the resistor R43 and the capacitor C41. The other end of the switch S41 is connected to a wire connecting the output terminal of the multiplexer 45 and the second input terminal I2 of the charge amplifier 44. The switch S41 is turned ON or OFF in response to a second check signal from the fault monitoring unit 53. When the switch S41 is turned ON, the input value to the second input terminal I2 of the charge amplifier 44 changes. The resistance value of the resistor R43 and the capacitance value of the capacitor C41 are predetermined. With this circuit configuration, the charge supply circuit 48 supplies a test charge to the charge amplifier 44 in the fault monitoring operation mode. That is, in the fault monitoring operation mode, the charge supply circuit 48 supplies a predetermined amount of charge to the second input terminal I2 of the charge amplifier 44 by turning on the switch S41. The predetermined amount of charge is set in advance to determine whether the operation is normal or not, and is determined by the resistance value of the resistor R43 and the capacitance value of the capacitor C41.

The CHK terminal Ti is a terminal for transmitting the second check signal from the fault monitoring unit 53 to the charge supply circuit 48. The second check signal is normally at a low level, and is at a high level when monitoring whether the operation of the charge amplifier 44 is normal or not. When the second check signal is at a low level, the switch S41 is OFF, and when it is at a high level, the switch S41 is ON.

When monitoring whether the charge amplifier 44 and the differential amplifier 46 are operating normally, the multiplexer 45 may be connected to any one of the sensor terminals T1 to T4, or may be in an off state where it is not connected to any of the sensor terminals T1 to T4.

The fault monitoring unit 53 monitors whether the charge amplifier 44 and the differential amplifier 46 are operating normally in the fault monitoring operation mode.

First, we will explain how the fault monitoring unit 53 monitors the operation of the charge amplifier 44.

The fault monitoring unit 53 monitors whether the charge amplifier 44 and the differential amplifier 46 are operating normally when the user is not gripping the steering wheel 3 (for example, when the ignition of the vehicle 1 is turned on, or immediately after it is detected that the user is not gripping the steering wheel 3).

First, when the second check signal is at a low level, in other words when the switch S41 is in an OFF state, the fault monitoring unit 53 receives an output signal from the output terminal o1 via the output monitoring terminal Th and stores it in the memory 57.

Next, the fault monitoring unit 53 sets the second check signal to a high level to turn on switch S41, and receives an output signal from output terminal o1 while switch S41 is in the ON state.

Then, the output signal from output terminal o1 when switch S41 is OFF, which is stored in memory 57, is compared with the output signal from output terminal o1 when switch S41 is ON, and the amount of change is calculated.

As described above, the resistance value of resistor R43 and the capacitance value of capacitor C41 are predetermined, so the amount of change in the output signal from output terminal o1 when switch S41 changes from OFF to ON is predictable.

Therefore, the fault monitoring unit 53 compares the calculated amount of change with the amount of change predicted in advance, and if the calculated amount of change is within a predetermined amount of the predicted amount of change, it determines that the operation of the charge amplifier 44 is normal, and if the calculated amount of change deviates from the predicted amount of change by more than a predetermined amount, it determines that the operation of the charge amplifier 44 is abnormal.

Next, we will explain how the fault monitoring unit 53 monitors the operation of the differential amplifier 46.

When the fault monitoring unit 53 determines that the charge amplifier 44 is operating normally, it monitors whether the differential amplifier 46 is operating normally.

First, when the second check signal is at a low level, in other words, when the switch S41 is OFF, the fault monitoring unit 53 receives an output signal from the output terminal of the differential amplifier 46 via the AD terminal Ta and stores it in the memory 57.

Next, the fault monitoring unit 53 turns on the switch S41 by setting the second check signal to a high level, and receives an output signal from the output terminal of the differential amplifier 46 while the switch S41 is in the ON state.

Then, the output signal from the differential amplifier 46 when switch S41 is OFF, which is stored in memory 57, is compared with the output signal from the differential amplifier 46 when switch S41 is ON, and the amount of change is calculated.

As described above, the resistance value of resistor R43 and the capacitance value of capacitor C41 are predetermined, and the operation of charge amplifier 44 is normal, so the amount of change in the output signal from differential amplifier 46 when switch S41 changes from OFF to ON is predictable.

Therefore, the fault monitoring unit 53 compares the calculated amount of change with the amount of change predicted in advance, and if the calculated amount of change is within a predetermined amount of the predicted amount of change, it determines that the operation of the differential amplifier 46 is normal, and if the calculated amount of change deviates from the predicted amount of change by more than a predetermined amount, it determines that the operation of the differential amplifier 46 is abnormal.

As described above, the steering grip sensor 100 according to the third embodiment further includes a charge supply circuit 48, which makes it possible to monitor whether the charge amplifier 44 and the differential amplifier 46 are operating normally.

In addition, without being limited to the above-mentioned embodiment 3, first, switch S41 may be turned ON to store the output signal from output terminal o1 or differential amplifier 46 in memory 57, and then switch S41 may be turned OFF to receive the output signal from output terminal o1 or differential amplifier 46 and compare it with the output signal data stored in memory 57.

(Embodiment 4)
In the fourth embodiment, in addition to the second or third embodiment, a configuration for performing a correction process for suppressing fluctuations in the operation of the steering grip sensor 100 due to environmental changes will be described.

In the fourth embodiment, the environmental changes specifically include changes in both temperature and humidity, or changes in either temperature or humidity, such as changes in the resistance value of the dielectric layer 12, changes in the capacitance of the sensor electrode 112, and changes in the resistance value of the urethane layer 13. In the following, temperature and humidity refers to both temperature and humidity, or temperature or humidity.

The sensor electrode 112 is wrapped around the rim 31 with the dielectric layer 12 and other materials sandwiched between them. The rim 31 is grounded, so in other words, the sensor electrode 112 is connected to GND via a resistor Rx. This resistor Rx is a parasitic resistor based on the resistance component of the dielectric layer 12 that is present between the sensor electrode 112 and GND. The resistor Rx has temperature and humidity dependent characteristics that depend on the temperature and humidity.

The sensor electrode 112 faces the rim 31 with the urethane layer 13 or the like sandwiched between them, or is wrapped around the rim 31 with the urethane layer 13 or the like sandwiched between them. Therefore, the sensor electrode 112 and the rim 31 form a capacitor Cx. Since the rim 31 is grounded, the capacitor Cx formed by the sensor electrode 112 and the rim 31 has a constant capacitance value. However, this capacitance value has the property of changing depending on the temperature and humidity of the dielectric layer 12 and the urethane layer 13, etc.

Here, resistor Rx and capacitor Cx have a correlation. It is preferable that the dielectric layer 12 and urethane layer 13 are provided so that this correlation (hereinafter, the design target relationship) is such that it is unlikely to affect the determination of gripping by the steering grip sensor 100 even if the resistance value of resistor Rx and the capacitance value of capacitor Cx change depending on, for example, temperature and humidity. However, the correlation between resistor Rx and capacitor Cx may deviate from the design target relationship due to individual product variations or deterioration over time.

The steering grip sensor 100 of the fourth embodiment performs a correction process to prevent the correlation between the resistor Rx and the capacitor Cx from deviating from the design target relationship due to individual product variations or deterioration over time. This ultimately prevents fluctuations in the operation of the steering grip sensor 100 due to changes in the environment.

Figure 11 is a block diagram showing an example of the circuit configuration of the steering grip sensor 100 in embodiment 4. This figure differs from Figure 10 mainly in that a variable resistor Rv and a correction control terminal Tk are added.

In the following, we will avoid repeating the same points and focus on the differences.

The variable resistor Rv is connected between the wiring that connects the output terminal of the multiplexer 45 and the second input terminal I2 of the charge amplifier 44, and the GND (signal ground) wiring. It is a variable resistor whose resistance value changes according to the correction control signal output from the correction unit 55 via the correction control terminal Tk.

The correction control terminal Tk is a terminal for transmitting a correction control signal from the correction unit 55 to the variable resistor Rv.

The correction unit 55 determines, from the second multiplication data 68 and the third multiplication data 69, whether the correlation between the resistor Rx and the capacitor Cx deviates from the design target relationship by a predetermined amount or more.

Specifically, the determination is made as follows. There is a relationship between the second multiplication data 68 and the third multiplication data 69, and between the resistance Rx and the capacitor Cx. Therefore, for example, for each temperature and humidity, the second multiplication data 68 and the third multiplication data 69 corresponding to the design target relationship when the user does not grip the steering wheel 3 are stored. Then, if (1) and (2) of (1) the second multiplication data 68 and the third multiplication data 69 when the current user does not grip the steering wheel 3, and (2) the second multiplication data 68 and the third multiplication data 69 when the user does not grip the steering wheel 3 under the same conditions as the current temperature and humidity, are different by a predetermined value or more, it is determined that the correlation between the resistance Rx and the capacitor Cx is different from the design target relationship by a predetermined value or more.

When the correlation between resistor Rx and capacitor Cx deviates from the design target relationship by a predetermined amount or more, the correction unit 55 performs correction by changing the resistance value of the variable resistor Rv so that the correlation between resistor Rx and capacitor Cx approaches the design target relationship.

Specifically, for example, if the resistance value of resistor Rx is small relative to the capacitance value of capacitor Cx and deviates from the design target relationship, the correctable unit 55 transmits a correction control signal to variable resistor Rv and increases the resistance value of variable resistor Rv.

Here, the combined resistance Rs of the resistor Rx and the variable resistor Rv is expressed by Equation 1.

By increasing the resistance value of the variable resistor Rv so that the resistance value of the combined resistor Rs approaches the resistance value of the design target relationship relative to the capacitance value of the capacitor Cx, it is possible to prevent deviation from the design target relationship.

Similarly, for example, if the resistance value of resistor Rx is large relative to the capacitance value of capacitor Cx and deviates from the design target relationship, the correctable unit 55 transmits a correction control signal to variable resistor Rv and reduces the resistance value of variable resistor Rv, thereby bringing the resistance value of combined resistor Rs closer to the resistance value in the design target relationship.

Figure 12 is a diagram showing an example of the correlation between resistor Rx and capacitor Cx. The vertical axis shows the capacitance value of capacitor Cx, and the horizontal axis shows the resistance value of resistor Rx or combined resistor Rs. Solid line L0 shows the design target relationship, for example, the correlation in the initial state at the time of shipment from the factory, or the correlation at room temperature. Dashed lines L1 and L2 show cases in which the correlation between resistor Rx and capacitor Cx deviates from the design target relationship due to individual product variations or deterioration over time. The correction unit 55 corrects the resistance value of combined resistor Rs by increasing or decreasing the resistance value of variable resistor Rv. As shown by the arrows in the figure, the state of dashed line L1 or L2 that deviates from the design target relationship can be brought closer to the design target relationship of solid line L0 by correction.

This configuration allows for correction processing to be performed that prevents the correlation between resistor Rx and capacitor Cx from deviating from the design target relationship due to individual product variations or deterioration over time, and ultimately makes it possible to suppress fluctuations in operation of the steering grip sensor 100 caused by environmental changes.

In each of the above embodiments, each component may be configured with dedicated hardware, or may be realized by executing a software program suitable for each component. Each component may be realized by a program execution unit such as a CPU or processor reading and executing a software program recorded on a recording medium such as a hard disk or semiconductor memory. Here, the software that realizes the steering grip sensor and the like in each of the above embodiments is a program such as the following.

That is, this program causes a computer to execute a grip detection method for a steering grip sensor that includes a planar driven electrode that follows the rim of the steering wheel and a planar sensor electrode that faces the driven electrode. The grip detection method includes supplying a sine wave voltage to the driven electrode, detecting a change in the amount of charge generated by the capacitance of the sensor electrode using a charge amplifier having a feedback capacitance element, generating an output voltage in which the detected change in the amount of charge is a change in voltage using the charge amplifier, multiplying the sine wave voltage by the output voltage of the charge amplifier, smoothing the multiplication result by integrating it, and determining whether or not the steering wheel is being gripped depending on the level of the smoothed multiplication result.

The above describes the steering grip sensor according to one or more aspects based on the embodiment, but the present disclosure is not limited to this embodiment. As long as it does not deviate from the spirit of the present disclosure, various modifications conceivable by a person skilled in the art to this embodiment and forms constructed by combining components in different embodiments may also be included within the scope of one or more aspects.

This disclosure can be used in a steering wheel grip sensor that detects when a person is gripping the steering wheel.

41 Sine wave generating unit 42 Attenuation switch 43, 44 Charge amplifier 48 Charge supply circuit 52 Multiplication processing unit 53 Fault monitoring unit (monitoring unit)
54 Integrator 56 Grip determination unit 57 Memory 62 First AD converter 63 Second AD converter 72 First digital data 73 Second digital data 100 Steering grip sensor 112 Sensor electrode 113 Driven electrode o1 Output terminal A11 Operational amplifier C11 Feedback capacitance element I1 First input terminal I2 Second input terminal R11 Feedback resistance element Rv Variable resistance Rx Resistor

Claims (19)

a planar driven electrode that is aligned along the rim of the steering wheel;
a planar sensor electrode facing the driven electrode;
A sine wave generating unit that supplies a sine wave voltage to the driven electrode;
a charge amplifier having a feedback capacitance element, detecting a change in the amount of charge generated by the capacitance of the sensor electrode, and outputting the change in the amount of charge as a change in voltage;
a multiplication processing unit that multiplies the sine wave voltage by an output voltage of the charge amplifier;
an integrator that smoothes the multiplication result by integrating it;
a grip determination unit that determines whether or not the steering wheel is being gripped according to a level of the smoothed multiplication result ,
moreover,
A monitoring unit that monitors for failures;
an attenuation switch that attenuates or amplifies the sine wave voltage;
the monitoring unit has a normal monitoring mode in which the attenuation switch does not attenuate or amplify the sine wave voltage, and a temporary monitoring mode in which the attenuation switch attenuates or amplifies the sine wave voltage,
the monitoring unit monitors a ground fault of the driven electrode and a ground fault of the sensor electrode in the normal monitoring mode;
In the temporary monitoring mode, a disconnection of the driven electrode, a disconnection of the sensor electrode, and a short circuit between the driven electrode and the sensor electrode are monitored.
Steering grip sensor. The charge amplifier is
a first input terminal to which the sine wave voltage is input;
A second input terminal connected to the sensor electrode;
An output terminal;
the feedback capacitance element connected between the output terminal and the second input terminal;
an operational amplifier connected to the first input terminal, the second input terminal, and the output terminal;
2. The steering grip sensor according to claim 1, wherein the sine wave voltage output from the output terminal is supplied to the sensor electrode via the feedback capacitance element. The steering grip sensor according to claim 1 or 2, wherein the sine wave generating unit receives a square wave signal and generates the sine wave voltage synchronized with the square wave signal. the charge amplifier includes a low-pass filter connected in parallel with the feedback capacitance element;
The steering grip sensor according to any one of claims 1 to 3, wherein the low-pass filter passes frequencies lower than a frequency of the sine wave voltage. moreover,
Memory,
a first AD converter that performs AD conversion on the output voltage of the charge amplifier in synchronization with the rectangular wave signal and stores a result of the AD conversion in the memory as first digital data;
a second AD converter that performs AD conversion on the sine wave voltage from the sine wave generating unit in synchronization with the rectangular wave signal and stores a result of the AD conversion in the memory as second digital data;
4. The steering grip sensor according to claim 3, wherein the multiplication processing unit averages the waveform represented by the first digital data and the waveform represented by the second digital data, and multiplies the averaged first digital data by the averaged second digital data. moreover,
Memory,
a first AD converter that performs AD conversion on the output voltage of the charge amplifier in synchronization with the rectangular wave signal and stores a result of the AD conversion in the memory as first digital data;
a second AD converter that performs AD conversion on the sine wave voltage from the sine wave generating unit in synchronization with the rectangular wave signal and stores the AD conversion result in the memory as second digital data;
The multiplication processing unit multiplies the first digital data and the second digital data,
4. The steering grip sensor according to claim 3, further comprising a process for averaging a waveform indicated by the digital data resulting from the multiplication. The square wave signal is switched between at least three frequencies;
7. The steering grip sensor according to claim 5, wherein the first AD converter and the second AD converter store the first digital data and the second digital data in the memory for each of the at least three frequencies. The first AD converter and the second AD converter generate the first digital data and the second digital data so as to correspond to at least two periods of the waveform of the sine wave voltage. The steering grip sensor according to any one of claims 5 to 7. the first AD converter and the second AD converter generate the first digital data and the second digital data so as to correspond to a waveform of at least three periods of the sine wave voltage;
The steering grip sensor according to any one of claims 5 to 7, wherein the multiplication processing unit compares the at least three cycles of the waveform in the first digital data on a cycle-by-cycle basis and discards the digital data of the cycle with the most deviating value. the first AD converter and the second AD converter generate the first digital data and the second digital data so as to correspond to a waveform of at least three periods of the sine wave voltage;
The multiplication processing unit calculates an average value and a deviation for each of the at least three waveform periods in the first digital data, and if the deviation is equal to or greater than a threshold value, discards the digital data for that period. The steering grip sensor according to any one of claims 5 to 7. The steering grip sensor according to claim 10 , wherein the multiplication processor changes the frequency of the rectangular wave signal when a predetermined number of periods in which the deviation is equal to or greater than the threshold value continue. The steering grip sensor according to claim 1 , wherein the monitoring unit controls the attenuation rate and the amplification rate in the attenuation switch. a correction unit that performs a correction process to suppress fluctuations in the amount of charge generated in the sensor electrode due to environmental changes,
The steering grip sensor according to any one of claims 1 to 12, wherein the multiplication processing unit further shifts the phase of the output voltage of the charge amplifier by 90 degrees, multiplies the sine wave voltage by the shifted output voltage, and outputs a multiplication result to the correction unit as an index of the environmental change. a planar driven electrode that is aligned along the rim of the steering wheel;
a planar sensor electrode facing the driven electrode;
A sine wave generating unit that supplies a sine wave voltage to the driven electrode;
a charge amplifier having a feedback capacitance element, detecting a change in the amount of charge generated by the capacitance of the sensor electrode, and outputting the change in the amount of charge as a change in voltage;
a multiplication processing unit that multiplies the sine wave voltage by an output voltage of the charge amplifier;
an integrator that smoothes the multiplication result by integrating it;
a grip determination unit that determines whether or not the steering wheel is being gripped according to a level of the smoothed multiplication result,
Further, another charge amplifier having a feedback capacitance element detects a change in the amount of charge generated in the driven electrode due to an environmental change, and outputs the change in the amount of charge as a change in voltage;
a correction unit that performs a correction process to suppress a fluctuation in the amount of charge generated in the driven electrode due to an environmental change,
The multiplication processing unit further multiplies the sine wave voltage by an output voltage of the other charge amplifier, and outputs a multiplication result to the correction unit as an index of the environmental change.
Steering grip sensor. a planar driven electrode that is aligned along the rim of the steering wheel;
a planar sensor electrode facing the driven electrode;
A sine wave generating unit that supplies a sine wave voltage to the driven electrode;
a charge amplifier having a feedback capacitance element, detecting a change in the amount of charge generated by the capacitance of the sensor electrode, and outputting the change in the amount of charge as a change in voltage;
a multiplication processing unit that multiplies the sine wave voltage by an output voltage of the charge amplifier;
an integrator that smoothes the multiplication result by integrating it;
a grip determination unit that determines whether or not the steering wheel is being gripped according to a level of the smoothed multiplication result,
a current amplifier having a feedback resistor element and outputting a change in current generated in the driven electrode as a change in voltage;
a correction unit that performs a correction process to suppress fluctuations in current generated in the driven electrode due to environmental changes,
The multiplication processing unit further multiplies the sine wave voltage by the output voltage of the current amplifier, and outputs a multiplication result to the correction unit as an index of the environmental change.
Steering grip sensor. the low-pass filter includes a first resistive element, a second resistive element connected in series with the first resistive element, a first capacitive element connected to a connection point between the first resistive element and the second resistive element, and a third resistive element connected in series with the first capacitive element.
5. The steering grip sensor according to claim 4. moreover,
a charge supply circuit for supplying a predetermined amount of charge to the charge amplifier in a failure monitoring operation mode;
the monitoring unit monitors whether the operation of the charge amplifier is normal based on an output of the charge amplifier when no charge is supplied from the charge supply circuit and an output of the charge amplifier when charge is supplied from the charge supply circuit.
2. The steering grip sensor according to claim 1 . a variable resistor provided between a connection point between the sensor electrode and the charge amplifier and a ground;
The steering grip sensor according to any one of claims 1 to 17 . A grip detection method for a steering grip sensor including a planar driven electrode along a rim of a steering wheel and a planar sensor electrode facing the driven electrode, comprising:
Supplying a sinusoidal voltage to the driven electrode;
A charge amplifier having a feedback capacitance element detects a change in the amount of charge generated by the capacitance of the sensor electrode;
generating an output voltage corresponding to a change in the detected amount of charge by the charge amplifier;
multiplying the sinusoidal voltage by the output voltage of the charge amplifier;
smoothing the multiplication result of the sine wave voltage and the output voltage of the charge amplifier by integrating the multiplication result;
determining whether the steering wheel is being gripped or not according to a level of the smoothed multiplication result ;
moreover,
In a normal monitoring mode without attenuation or amplification of the sine wave voltage, a ground fault of the driven electrode and a ground fault of the sensor electrode are monitored;
In a temporary monitoring mode involving attenuation or amplification of the sine wave voltage, a disconnection of the driven electrode, a disconnection of the sensor electrode, and a short circuit between the driven electrode and the sensor electrode are monitored.
Grasp detection method.

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