JP7629818B2 - AC motor drive control device and drive control method - Google Patents

Description

The present invention relates to a drive control device and drive control method for an AC motor, and is applicable to electric railway vehicles, electric vehicles, industrial inverters, wind power generation systems, diesel generator systems, etc.

Due to the increasing demand for energy saving and reducing the environmental load in recent years, drive control devices for AC motors are becoming increasingly popular in a wide range of applications, such as home appliances, industrial equipment, and infrastructure.
In general, an inverter device that converts DC power into an arbitrary frequency and voltage is used to drive an AC motor at a variable speed. The inverter device is composed of a main circuit using semiconductor switching elements and a controller that controls the switching elements, and controls the voltage and frequency applied to the AC motor by, for example, pulse width modulation control (hereinafter referred to as "PWM control") of the switching elements at an arbitrary carrier frequency, thereby driving the AC motor at a variable speed.

A known method for driving and controlling an AC motor is to control the torque of the AC motor by proportional-integral feedback control so that the current flowing in the direction of the main magnetic flux of the AC motor and the current flowing in a direction perpendicular to the main magnetic flux direction each match a command value.

However, the maximum voltage that can be output by an inverter device is limited by the magnitude of the DC power supply voltage connected to the inverter device. Therefore, if the output voltage of the inverter device becomes saturated when the AC motor is rotating at high speed or the DC power supply voltage drops, and proportional/integral feedback control is performed to match the current flowing through the AC motor to the command value, the deviation between the feedback current and the command value will continue to be integrated, and the current and torque of the AC motor will oscillate and diverge, making the control system prone to instability.

Therefore, when the output voltage of the inverter device becomes saturated, it is necessary to switch the control method of the AC motor to voltage control (synchronous one-pulse control) that outputs a square wave voltage that maximizes the inverter output voltage. A method for switching from current control to voltage control as described above is disclosed, for example, in Patent Document 1.

Patent document 1 discloses a technique in which, when the output voltage of a power conversion device becomes saturated, the integral term of current control is gradually converged to zero, the integral control is stopped, and the control method is switched from current control to voltage control.

JP 2011-72190 A

The technology disclosed in Patent Document 1 is to gradually converge the integral terms of the d-axis and q-axis in the rotating coordinate system to zero and smoothly switch control from current control to voltage control when the voltage applied to the motor becomes saturated due to a drop in the DC power supply voltage of the inverter device, etc. For this reason, it is necessary to determine whether the output voltage of the inverter device is saturated and to switch between current control and voltage control depending on the determined value.

However, when controlling an AC motor at high speeds where the inverter output voltage is saturated, if the DC power supply voltage connected to the inverter fluctuates, the control method will have to be switched frequently, which may cause delays in control response and shocks when switching torque or current.

Therefore, in view of the above problems, the object of the present invention is to provide a drive control device and a drive control method for an AC motor that does not require switching of the control method depending on the saturation state of the inverter output voltage and improves the control stability of vector control.

In order to solve the above problems, one representative drive control device for an AC motor of the present invention is a drive control device for an AC motor driven by a power converter using vector control based on a voltage command, and includes a current command calculation unit that generates current command values for each of a d-axis and a q-axis, which are two orthogonal rotation coordinate axes of the AC motor, current detection means that detects currents flowing through the AC motor and converts them into current detection values for each of the d-axis and q-axis, and a voltage command calculation unit that calculates voltage commands for each of the d-axis and q-axis based on the current command values for each of the d-axis and q-axis, the current detection values for each of the d-axis and q-axis, and the rotation speed of the AC motor, and the voltage command calculation unit: The d-axis voltage command is calculated by adding the result of proportional calculation of the d-axis current command value to the result of proportional calculation of the d-axis current deviation between the d-axis current command value and the d-axis current detection value using d-axis current feedback control based on proportional calculation of the d-axis current deviation between the d-axis current command value and the d-axis current detection value, and subtracting from the sum a value obtained by multiplying the result of proportional and integral calculation of the integral output of the q-axis current deviation between the q-axis current command value and the q-axis current detection value by the rotational speed. The q-axis voltage command is calculated by adding the results of both calculations using q-axis current feedback control based on proportional calculation of the q-axis current deviation and proportional and integral calculation of the q-axis current deviation, and adding to the sum a value obtained by multiplying the result of proportional calculation of the d-axis current command value by the rotational speed and the value of the result of the proportional calculation of the rotational speed, respectively .

According to the present invention, it is possible to avoid delays in control response and the occurrence of current and torque vibrations due to the process of switching control methods when the DC power supply voltage of the inverter device fluctuates, and it is possible to realize stable and highly responsive motor drive control that can be applied from multi-pulse drive using PWM control to synchronous one-pulse drive.
Problems, configurations and effects other than those described above will become apparent from the description of the following embodiments.

1 is a block diagram illustrating an example of the configuration of a drive control device for an AC motor according to a first embodiment of the present invention. FIG. 2 is a diagram showing a coordinate system and definitions of symbols used in drive control of an AC motor according to the present invention. FIG. 2 is a block diagram illustrating a configuration example of a vector control unit according to the first embodiment. FIG. 2 is a block diagram illustrating an example of a configuration of a voltage vector calculation unit according to the first embodiment. FIG. 2 is a block diagram illustrating a configuration example of a dc axis voltage command calculation unit according to the first embodiment. FIG. 2 is a block diagram illustrating a configuration example of a qc-axis voltage command calculation unit according to the first embodiment. FIG. 6 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a second embodiment of the present invention. FIG. 11 is a block diagram illustrating a configuration example of a vector control unit according to a second embodiment. FIG. 11 is a block diagram illustrating a configuration example of a first voltage vector calculation unit according to a second embodiment. FIG. 11 is a block diagram illustrating a configuration example of a second voltage vector calculation unit according to the second embodiment. FIG. 11 is a block diagram illustrating a configuration example of a second dc axis voltage command calculation unit according to the second embodiment. FIG. 11 is a block diagram illustrating a configuration example of a second qc-axis voltage command calculation unit according to the second embodiment. FIG. 11 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a third embodiment of the present invention. FIG. 11 is a block diagram illustrating a configuration example of a vector control unit according to a third embodiment. FIG. 11 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a fourth embodiment of the present invention. FIG. 13 is a block diagram illustrating a configuration example of a vector control unit according to a fourth embodiment. FIG. 13 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a fifth embodiment of the present invention. FIG. 13 is a block diagram illustrating a configuration example of a vector control unit according to a fifth embodiment. FIG. 13 is a block diagram illustrating a configuration example of a drive control device for an AC motor according to a sixth embodiment of the present invention. FIG. 13 is a block diagram illustrating a configuration example of a vector control unit according to a sixth embodiment. FIG. 13 is a block diagram illustrating a configuration example of a voltage vector calculation unit according to a sixth embodiment. FIG. 13 is a block diagram illustrating a configuration example of a dc axis voltage command calculation unit according to a sixth embodiment. FIG. 13 is a block diagram illustrating a configuration example of a qc-axis voltage command calculation unit according to a sixth embodiment. FIG. 13 is a block diagram showing a schematic configuration of a part of a railway vehicle equipped with a drive control device for an AC motor using any one of the first to sixth embodiments, as a seventh embodiment of the present invention.

Below, with reference to the drawings, examples 1 to 7 will be described in detail as modes for carrying out the present invention. In principle, the same reference numerals are used in all drawings to refer to the same elements. Furthermore, duplicated descriptions of parts having the same functions will be omitted. The examples and variations described below may be combined in part or in whole as long as they are not inconsistent.

Fig. 1 is a block diagram showing an example of the configuration of a drive control device for an AC motor according to a first embodiment of the present invention. In Fig. 1, the device includes an AC motor 103 to be controlled, a power converter 102 that drives the AC motor 103, a controller 101 that controls the power converter 102, a command generator 105 that generates a torque command Tm* for the AC motor 103, a phase current detection unit 121 that detects the current flowing through the AC motor 103, and a rotational speed detection unit 124 that detects the rotational speed of the AC motor 103.

The AC motor 103 is a motor that is controlled by the AC power output from the power converter 102. Note that in each embodiment of the present invention, the invention is described using an induction motor as an example of a type of AC motor, but the invention is not limited to this motor and can be applied to all other AC motors. Also, although each embodiment of the present invention uses an AC motor as an example, the same effects as in the case of an AC motor can be obtained even if an AC generator is controlled.

The power converter 102 includes input terminals 123a and 123b that supply DC power to the power converter 102, a main circuit section 132 consisting of six switching elements Sup to Swn, a gate driver 133 that directly drives the main circuit section 132, a DC resistor 134 and a smoothing capacitor 131 that are installed for overcurrent protection of the power converter 102. The power converter 102 also converts the DC power supplied from the input terminals 123a and 123b into AC power based on a gate command signal generated by the controller 101, and supplies the converted AC power to the AC motor 103.

The phase current detection unit 121 detects the AC currents iu and iw that flow from the power converter 102 to the AC motor 103. This phase current detection unit 121 is realized by a current sensor using a Hall element, for example. Note that the phase current detection unit 121 shown in FIG. 1 is configured to detect AC current by two-phase detection, but it may also be configured to detect three phases. Also, instead of using a phase current sensor, it may be possible to use an AC current value estimated from the current value flowing through a DC resistor 134 installed for overcurrent protection of the power converter 102.

The rotation speed detection unit 124 detects the rotation speed ωrm of the AC motor 103. This rotation speed detection unit 124 is realized by, for example, an encoder. Furthermore, the speed calculation unit 115 calculates the electrical angular frequency ωre of the rotation speed of the AC motor 103 from the detected rotation speed ωrm of the AC motor 103, and further the vector control unit 112 calculates the inverter frequency ω1 for vector control.

When it is necessary to detect the rotor position, such as in a permanent magnet synchronous motor, the rotor position can be detected using, for example, a resolver, an encoder, or a magnetic sensor instead of the rotational speed detection unit 124, and the rotational speed of the AC motor can be calculated based on the rotor position.

Furthermore, instead of using a rotational position sensor or a speed sensor, the rotational speed ωr and rotor position θd of the AC motor 103 may be estimated based on a voltage command value and a current detection value, thereby making the configuration speed sensorless and position sensorless.

The command generator 105 is a controller located above the controller 101, and generates a torque command Tm* for the AC motor.

The controller 101 controls the torque generated by the AC motor 103 based on the torque command Tm* from the command generator 105. For example, a current controller is used to control the current flowing through the AC motor 103, or a speed controller or a position controller is used to control the rotational speed or position. In the first embodiment, the controller 101 operates as a torque controller for the purpose of controlling torque, but a speed controller or a position controller may be used as a higher-level controller.

The controller 101 is configured to include a current command calculation unit 111, a vector control unit 112, a current detection unit 113, a dq coordinate conversion unit 114, a speed calculation unit 115, a phase calculation unit 116, a polar coordinate conversion unit 117, a UVW coordinate conversion unit 118, and a PWM signal controller 119.

The controller 101 also generates a gate command signal for driving the switching elements of the power converter 102 from the calculation results of the current control system and the phase control system based on the AC current detection values Iu and Iw, which are the detection values of the AC currents iu and iw flowing through the AC motor 103, the rotation speed ωrm of the AC motor 103 from the rotation speed detection unit 124, and the torque command Tm* from the command generator 105, and supplies the signal to the gate driver 133 of the power converter 102.

2 is a diagram showing a coordinate system and definitions of symbols used in the drive control of the AC motor 103 according to the present invention. Here, as described above, the AC motor 103 will be described by taking an induction motor as an example.
In Fig. 2, the ab-axis coordinate system defined by the a-axis and the b-axis is a stator coordinate system that represents the phase of the stator winding of the induction motor. The a-axis is generally taken as the reference for the u-phase winding phase of the induction motor.

The dq-axis coordinate system defined by the d-axis and q-axis is a rotor coordinate system that represents the direction of magnetic flux excited in the rotor of an induction motor.
The dc-qc axis coordinate system defined by the dc axis and the qc axis is a coordinate system assumed by the controller 101 to be the d axis and q axis directions, and is also called the control axes.
The coordinate axes combined in each coordinate system are all orthogonal to each other.
In each of the above coordinate systems, the phases of the d-axis and dc-axis with respect to the a-axis are represented as θd and θdc, respectively, as shown in Fig. 2. The deviation of the dc-axis from the d-axis is represented as Δθc.

The configuration of the controller 101 shown in FIG. 1 will be described in detail below.
Current command calculation unit 111 outputs current command values Idc* and Iqc* on a dc-qc axis coordinate system calculated based on inverter frequency ω1 output from vector control unit 112, voltage detection value Ecf of smoothing capacitor 131, and torque command Tm* output from command generator 105. Here, the current command values Idc* and Iqc* may be calculated, for example, by using values obtained in advance by testing or analysis that are optimal d-axis and q-axis current command values for the torque command Tm*, voltage detection value Ecf, and inverter frequency ω1 as a reference table, a functional formula, an approximate formula, a design formula, or a theoretical formula.

The vector control unit 112 separates the torque current component (q-axis current component) and the excitation current component (d-axis current component) and controls the currents separately to match the current detection values Idc and Iqc on the dc-qc axis coordinate system output by the dq coordinate conversion unit 114 with the current command values Idc* and Iqc* on the dc-qc axis coordinate system output by the current command calculation unit 111. Based on the result of this current control and the rotation speed ωre of the induction motor 103a, the voltage commands Vdc* and Vqc* on the dc-qc axis, which is the rotating coordinate system, are calculated and output.

The current detection unit 113 calculates three-phase current detection values Iu, Iv, and Iw from the AC currents iu and iw flowing through the induction motor 103a detected by the phase current detection unit 121, and outputs them to the dq coordinate transformation unit 114.

The dq coordinate conversion unit 114 converts the three-phase current detection values Iu, Iv, and Iw output by the current detection unit 113 into current detection values Idc and Iqc on the dc-qc axis coordinate system using the control phase θdc output by the phase calculation unit 116 based on the vector control inverter frequency ω1 output by the vector control unit 112, and outputs the converted values.

The speed calculation unit 115 calculates and outputs the electrical angular frequency ωre of the rotational speed of the induction motor 103a based on the rotational speed ωrm of the induction motor 103a output by the rotational speed detection unit 124.

The phase calculation unit 116 calculates and outputs the control phase θdc based on the vector control inverter frequency ω1 output by the vector control unit 112.

The polar coordinate conversion unit 117 converts the voltage commands Vdc* and Vqc* on the dc-qc axes output by the vector control unit 112 into a voltage amplitude command V1* and a voltage phase command δ and outputs them.

The UVW coordinate conversion unit 118 converts the voltage amplitude command V1* and voltage phase command δ output by the polar coordinate conversion unit 117 and the control phase θdc output by the phase calculation unit 116 into three-phase AC voltage commands Vu*, Vv*, and Vw* and outputs them to the PWM signal controller 119.

The PWM signal controller 119 generates a triangular wave carrier based on an arbitrary carrier frequency fc and the voltage detection value Ecf of the smoothing capacitor 131, compares the magnitude of the triangular wave carrier with that of a modulated wave based on the three-phase AC voltage commands Vu*, Vv*, and Vw*, and performs pulse width modulation. The gate command signal generated as a result of this pulse width modulation controls the on/off of the switching elements of the power converter 102.

Next, the vector control unit 112, which is a characteristic part of the first embodiment, will be described in detail.
FIG. 3 is a block diagram illustrating an example of the configuration of the vector control unit 112 according to the first embodiment.
3 includes a voltage vector calculation unit 201 and an inverter frequency calculation unit 202. Moreover, vector control unit 112 receives as input current command values Idc* and Iqc* on the dc-qc axis coordinate system, current detection values Idc and Iqc on the dc-qc axis coordinate system, and an electrical angular frequency ωre of the rotation speed of induction motor 103a, and outputs voltage commands Vdc* and Vqc* on the dc-qc axes and an inverter frequency ω1 for vector control.

The inverter frequency calculation unit 202 calculates and outputs the vector control inverter frequency ω1 based on the current command values Idc* and Iqc* on the dc-qc axis coordinate system and the electrical angular frequency ωre of the rotational speed of the induction motor 103a.

In the inverter frequency calculation unit 202, when an induction motor is the control target, the inverter frequency ω1 for vector control is calculated by adding the slip frequency ωs calculated based on the current command values Idc* and Iqc* on the dc-qc axis coordinate system to the electrical angular frequency ωre of the rotational speed of the induction motor.

In the first embodiment, the AC motor 103 is described as an induction motor, so the inverter frequency ω1 is determined by adding the electrical angular frequency ωre of the rotational speed of the induction motor to the slip frequency ωs. Alternatively, if the AC motor 103 is a permanent magnet synchronous motor, the slip frequency ωs need not be calculated, and the electrical angular frequency ωre can be set as the inverter frequency ω1 for vector control.

FIG. 4 is a block diagram illustrating an example of the configuration of the voltage vector calculation unit 201 according to the first embodiment.
The voltage vector calculation unit 201 shown in FIG. 4 includes a d-axis current deviation calculation unit 301 , a q-axis current deviation calculation unit 302 , a q-axis integrator 303 , a dc-axis voltage command calculation unit 401 , and a qc-axis voltage command calculation unit 402 .

The voltage vector calculation unit 201 also calculates and outputs voltage commands Vdc* and Vqc* on the dc-qc axes based on the current command values Idc* and Iqc* on the dc-qc axis coordinate system, the current detection values Idc and Iqc on the dc-qc axis coordinate system, and the vector control inverter frequency ω1.

The d-axis current deviation calculation unit 301 subtracts the current detection value Idc on the dc-qc coordinate system from the current command value Idc* on the dc-qc coordinate system, and outputs the result as the dc-axis current deviation amount ΔIdc.

The q-axis current deviation calculation unit 302 subtracts the current detection value Iqc on the dc-qc coordinate system from the current command value Iqc* on the dc-qc coordinate system, and outputs the result as the qc-axis current deviation amount ΔIqc.

The q-axis integrator 303 integrates the qc-axis current deviation amount ΔIqc output by the q-axis current deviation calculation unit 302, and outputs it as the qc-axis current deviation integral value ΔIqc-i.

The dc-axis voltage command calculation unit 401 calculates and outputs the voltage command Vdc* on the dc-qc axes based on the current command value Idc* on the dc-qc axis coordinate system output by the current command calculation unit 111, the dc-axis current deviation amount ΔIdc output by the d-axis current deviation calculation unit 301, the qc-axis current deviation integral value ΔIqc-i output by the q-axis integrator 303, and the inverter frequency ω1 output by the inverter frequency calculation unit 202.

The qc-axis voltage command calculation unit 402 calculates and outputs a voltage command Vqc* on the dc-qc axes based on the current command value Idc* on the dc-qc axis coordinate system output by the current command calculation unit 111, the qc-axis current deviation amount ΔIqc output by the q-axis current deviation calculation unit 302, the qc-axis current deviation integral value ΔIqc-i output by the q-axis integrator 303, and the inverter frequency ω1 output by the inverter frequency calculation unit 202.

FIG. 5 is a block diagram showing an example of the configuration of the dc axis voltage command calculation unit 401 according to the first embodiment.
5 includes proportional units 501, 502, and 503, an adder 801, a subtractor 802, and a multiplier 901. Furthermore, dc axis voltage command calculation unit 401 calculates and outputs a voltage command Vdc* on the dc-qc axes based on the current command value Idc* on the dc-qc axis coordinate system, the dc axis current deviation amount ΔIdc, the qc axis current deviation integral value ΔIqc-i, and the inverter frequency ω1.

In dc-axis voltage command calculation unit 401, proportional unit 501 multiplies the current command value Idc* on the dc-qc axis coordinate system by proportional gain Kr, proportional unit 502 multiplies the dc-axis current deviation amount ΔIdc by proportional gain Kpd, and adder 801 adds the two together. Furthermore, proportional unit 503 and multiplier 901 multiply the qc-axis current deviation integral value ΔIqc-i by proportional gain Kqd and inverter frequency ω1. Then, subtraction of these values is performed by subtractor 802 to obtain voltage command Vdc* on the dc-qc axis. For example, it is calculated using the following formula (1).

ここで、式（２）において、Ｒ１＊は、誘導電動機の一次抵抗の制御設定値、Ｌσ＊は、誘導電動機の一次換算漏れインダクタンスの制御設定値、ωａｃｒ＊は、電流制御の制御応答角周波数である。なお、ｄｃ軸電圧指令演算部４０１の比例ゲインＫｒ、ＫｐｄおよびＫｑｄは、定数でもよいし、電流指令値、電流検出値、速度およびベクトル演算周期などの関数式として演算したものを用いてもよい。 Here, when an induction motor is the control target, the proportional gains Kr, Kpd and Kqd of dc axis voltage command calculation section 401 are calculated, for example, by the following equation (2).

In the formula (2), R1* is a control set value of the primary resistance of the induction motor, Lσ* is a control set value of the primary-equivalent leakage inductance of the induction motor, and ωacr* is a control response angular frequency of the current control. The proportional gains Kr, Kpd, and Kqd of the dc-axis voltage command calculation unit 401 may be constants, or may be calculated as functions of the current command value, the current detection value, the speed, the vector calculation period, etc.

FIG. 6 is a block diagram illustrating an example of the configuration of the qc-axis voltage command calculation unit 402 according to the first embodiment.
6 includes proportional units 504, 505, 506, and 507, an adder 801, and a multiplier 901. The qc-axis voltage command calculation unit 402 calculates and outputs a voltage command Vqc* on the dc-qc axes based on the current command value Idc* on the dc-qc axis coordinate system, the qc-axis current deviation amount ΔIqc, the qc-axis current deviation integral value ΔIqc-i, and the inverter frequency ω1.

In the qc-axis voltage command calculation unit 402, a proportional unit 504 multiplies a current command value Idc* on the dc-qc axis coordinate system by a proportional gain Kdq, and then a multiplier 901 multiplies the inverter frequency ω1. A proportional unit 506 multiplies a qc-axis current deviation amount ΔIqc by a proportional gain Kpq, a proportional unit 505 multiplies a qc-axis current deviation integral value ΔIqc-i by a proportional gain Kiq, and an adder 801 adds both of them. Furthermore, these are added by an adder 801, and the product of the inverter frequency ω1 multiplied by the proportional gain Kf by a proportional unit 507 and the product added by an adder 801 are set as a voltage command Vqc* on the dc-qc axis. For example, it is calculated by the following formula (3).

ここで、式（４）において、Ｒ１＊は、誘導電動機の一次抵抗の制御設定値、Ｌσ＊は、誘導電動機の一次換算漏れインダクタンスの制御設定値、ωａｃｒ＊は、電流制御の制御応答角周波数、Ｍ＊は、誘導電動機の相互インダクタンスの制御設定値、Ｌ２＊は、誘導電動機の二次自己インダクタンスの制御設定値、Φ２ｄ＊は、誘導電動機の二次磁束指令値である。 Here, the proportional gains Kpq, Kiq, Kdq and Kf of the qc-axis voltage command calculation unit 402 are calculated, for example, by the following equation (4) when an induction motor is the control target.

Here, in equation (4), R1* is the control setting value of the primary resistance of the induction motor, Lσ* is the control setting value of the primary-equivalent leakage inductance of the induction motor, ωacr* is the control response angular frequency of current control, M* is the control setting value of the mutual inductance of the induction motor, L2* is the control setting value of the secondary self-inductance of the induction motor, and Φ2d* is the secondary magnetic flux command value of the induction motor.

Note that Φ2d* is calculated by multiplying the current command value Idc* on the dc-qc axis coordinate system by the mutual inductance control setting value M*, and passing it through a first-order lag filter whose filter time constant is equivalent to the secondary time constant T2 of the induction motor.

The proportional gains Kpq, Kiq, Kdq, and Kf of the qc-axis voltage command calculation unit 402 may be constants, or may be calculated as functions of the current command value, current detection value, speed, and vector calculation period.

Next, the principle of stabilization when the AC motor is driven and controlled using the vector control unit 112, which is a feature of the present invention, will be described in detail.
First, by configuring the vector control unit 112 to calculate the voltage commands Vdc* and Vqc* on the dc-qc axes without using an integrator for current control on the dc axis, the same control method can be used regardless of whether the inverter output voltage is saturated. In other words, this is a vector control method that can be used in all speed ranges, including multi-pulse drive of the inverter device and synchronous one-pulse drive in which the voltage utilization rate is maximized, and it becomes unnecessary to stop integral calculation of current control and switch to voltage control. Therefore, it is possible to prevent delays in control response and fluctuations in voltage, current, and torque due to control method switching processing when the DC power supply voltage fluctuates.

In addition, in the calculation of the voltage commands Vdc* and Vqc* on the dc-qc axes by the vector control unit 112, the feedback control of the dc-axis current is proportional control, and the feedback control of the qc-axis current is proportional and integral control (hereinafter abbreviated as "proportional and integral control"), resulting in an asymmetrical control configuration between the dc and qc axes. Here, the integrator output of the qc-axis current control is used to compensate for the voltage of the dq-axis interference term of the dc-axis voltage command Vdc*. As a result, even if the inverter output voltage is saturated, the dc-axis voltage command Vdc* is controlled via the qc-axis current control, so that the torque of the induction motor can be sufficiently controlled even with a configuration that does not use an integrator for the dc-axis current control.

Furthermore, by configuring the vector control unit 112 to calculate the voltage commands Vdc* and Vqc* on the dc-qc axes without using an integrator for current control on the dc axis, stability during high speed regeneration (when the motor torque is negative) is improved compared to using proportional-integral feedback control for current control on the dc axis. This will be explained in more detail below.

ここで、式（５）において、Ｒσは、誘導電動機の二次抵抗を一次側に換算した一次換算抵抗値、Ｌσは、誘導電動機の一次換算漏れインダクタンス値、Ｔ２は、誘導電動機の二次時定数である。 Equation (5) shows the open-loop transfer function Gvd of the transfer loop related to the calculation of the voltage command Vdc* of the dc axis including the induction motor.

Here, in equation (5), Rσ is a primary-converted resistance value obtained by converting the secondary resistance of the induction motor into the primary side, Lσ is a primary-converted leakage inductance value of the induction motor, and T2 is the secondary time constant of the induction motor.

The transfer function related to the calculation of this dc-axis voltage command Vdc* functions as a control loop that stabilizes the secondary magnetic flux of the induction motor during regeneration when the slip frequency is negative. By making the feedback control of the dc-axis current only proportional control, even when the control response angular frequency ωacr* of the current control is low, the DC gain of this transfer loop is multiplied by 1/(Rσ+Kpd) and increases, improving stability during high speed regeneration compared to when the feedback control of the dc-axis current is configured as proportional and integral control. Therefore, this is an effective method for stabilizing the inverter control system in applications where the control response of current control is not high, such as electric railway vehicles and wind power generation systems.

As described above, according to the first embodiment, by using a dc-axis voltage command calculation means that calculates the voltage command Vdc* on the dc-qc axis based on the current command value Idc*, the dc-axis current deviation amount ΔIdc, the qc-axis current deviation integral value ΔIqc-i, and the inverter frequency ω1, and a qc-axis voltage command calculation means that calculates the voltage command Vqc* on the dc-qc axis based on the current command value Idc*, the dc-axis current deviation amount ΔIqc, the qc-axis current deviation integral value ΔIqc-i, and the inverter frequency ω1, it is possible to prevent delays in control response due to the control method switching process when the DC power supply voltage of the inverter device fluctuates, and fluctuations in voltage, current, and torque, and improve the control stability of the drive control device for the AC motor.

In the second embodiment of the present invention, in the calculation of the voltage commands Vdc* and Vqc* on the dc and qc axes by the vector control unit 112 in the first embodiment,
When the AC motor is rotating at medium to high speeds, the feedback control of the dc-axis current is proportional control, and the integrator output of the qc-axis current control is used to compensate for the voltage of the dq-axis interference term of the dc-axis voltage command Vdc*.
When the AC motor is stopped or rotating at a low speed, the feedback control of the current on both the dc axis and the qc axis is configured as proportional-integral control.

This makes it possible to follow the current command with high accuracy, especially in the low-speed range, and also makes it possible to stop or start the integral calculation of the feedback control of the dc-axis current according to the rotation speed of any AC motor, regardless of whether the output voltage of the inverter device is saturated. Therefore, it is possible to realize a drive control device for an AC motor that can improve the accuracy and response of current control in the low-speed range and improve control stability in the high-speed range compared to Example 1.

7 is a block diagram showing a configuration example of a drive control device for an AC motor according to a second embodiment of the present invention. Hereinafter, only the differences in configuration will be described in comparison with the first embodiment shown in FIG.
The drive control device for an AC motor according to the second embodiment shown in FIG. 7 can be realized by using a vector control unit 112b instead of the vector control unit 112 of the first embodiment.

FIG. 8 is a block diagram illustrating an example of the configuration of a vector control unit 112b according to the second embodiment.
A vector control unit 112b shown in FIG. 8 includes a first voltage vector calculation unit 201b, a second voltage vector calculation unit 203b, a dc-axis voltage command switch 601b, a qc-axis voltage command switch 602b, and an inverter frequency calculation unit 202.

The vector control unit 112b receives as input the current command values Idc* and Iqc* on the dc-qc coordinate system, the current detection values Idc and Iqc on the dc-qc coordinate system, and the electrical angular frequency ωre of the rotational speed of the induction motor 103a.

The inverter frequency calculation unit 202 calculates the vector control inverter frequency ω1 based on the current command values Idc* and Iqc* and the electrical angular frequency ωre of the rotational speed of the induction motor 103a, as in the first embodiment.

The first voltage vector calculation unit 201b calculates the first voltage commands Vdc1* and Vqc1* on the dc-qc axes based on the current command values Idc* and Iqc*, the current detection values Idc and Iqc, and the inverter frequency ω1.

The second voltage vector calculation unit 203b calculates second voltage commands Vdc2* and Vqc2* on the dc-qc axes based on the same input as the first voltage vector calculation unit 201b.

The dc-axis voltage command switch 601b switches between the first voltage command Vdc1* output by the first voltage vector calculation unit 201b and the second voltage command Vdc2* output by the second voltage vector calculation unit 203b based on the inverter frequency ω1, and outputs it as the voltage command Vdc* on the dc-qc axes.

Here, the dc-axis voltage command switch 601b outputs the second voltage command Vdc2* output by the second voltage vector calculation unit 203b as the voltage command Vdc* on the dc-qc axes if the inverter frequency ω1 is, for example, 1/3 (threshold) or less of the maximum rotation speed of the AC motor, and outputs the first voltage command Vdc1* output by the first voltage vector calculation unit 201b as the voltage command Vdc* on the dc-qc axes if the inverter frequency ω1 is greater than 1/3 (threshold) of the maximum rotation speed of the AC motor.

The qc-axis voltage command switch 602b switches between the first voltage command Vqc1* output by the first voltage vector calculation unit 201b and the second voltage command Vqc2* output by the second voltage vector calculation unit 203b based on the inverter frequency ω1, and outputs it as the voltage command Vqc* on the dc-qc axes.

Here, the qc-axis voltage command switch 602b outputs the second voltage command Vqc2* output by the second voltage vector calculation unit 203b as the voltage command Vqc* on the dc-qc axes if the inverter frequency ω1 is, for example, 1/3 (threshold) or less of the maximum rotation speed of the AC motor, and outputs the first voltage command Vqc1* output by the first voltage vector calculation unit 201b as the voltage command Vqc* on the dc-qc axes if the inverter frequency ω1 is greater than 1/3 (threshold) of the maximum rotation speed of the AC motor.

The threshold value of the inverter frequency ω1 at which the voltage commands on the dc and qc axes are switched may be set to any value.
In addition, the voltage vector calculation unit that does not output the voltage calculation results as the voltage commands Vdc* and Vqc* on the dc-qc axes should stop the integral calculation of the current deviation amount, and when the voltage calculation results are output as the voltage commands Vdc* and Vqc* on the dc-qc axes and calculation is started, the output of the other integrator on the same axis should be set as the initial value of the integral calculation.

Next, the vector control unit 112b, which is a characteristic feature of the second embodiment, will be described in detail.
FIG. 9 is a block diagram illustrating an example of the configuration of a first voltage vector calculation unit 201b according to the second embodiment.
The first voltage vector calculation unit 201b shown in FIG. 9 includes a d-axis current deviation calculation unit 301, a q-axis current deviation calculation unit 302, a q-axis integrator 303, a first dc-axis voltage command calculation unit 401b, and a first qc-axis voltage command calculation unit 402b.

The first voltage vector calculation unit 201b calculates and outputs first voltage commands Vdc1* and Vqc1* on the dc-qc axes based on the current command values Idc* and Iqc* on the dc-qc axis coordinate system, the current detection values Idc and Iqc on the dc-qc axis coordinate system, and the vector control inverter frequency ω1.
Here, the first voltage vector calculation unit 201b of the second embodiment has the same configuration as the voltage vector calculation unit 201 of the first embodiment shown in FIG.

FIG. 10 is a block diagram illustrating an example of the configuration of the second voltage vector calculation unit 203b according to the second embodiment.
The second voltage vector calculation unit 203b shown in FIG. 10 includes a d-axis current deviation calculation unit 304b, a q-axis current deviation calculation unit 305b, a d-axis integrator 306b, a q-axis integrator 307b, a second dc-axis voltage command calculation unit 403b, and a second qc-axis voltage command calculation unit 404b.

The second voltage vector calculation unit 203b calculates and outputs second voltage commands Vdc2* and Vqc2* on the dc-qc axes based on the current command values Idc* and Iqc* on the dc-qc axis coordinate system, the current detection values Idc and Iqc on the dc-qc axis coordinate system, and the vector control inverter frequency ω1.

The d-axis current deviation calculation unit 304b subtracts the current detection value Idc on the dc-qc coordinate system from the current command value Idc* on the dc-qc coordinate system, and outputs the result as the dc-axis current deviation amount ΔIdc.

The q-axis current deviation calculation unit 305b subtracts the current detection value Iqc on the dc-qc coordinate system from the current command value Iqc* on the dc-qc coordinate system, and outputs the result as the qc-axis current deviation amount ΔIqc.

The d-axis integrator 306b integrates the dc-axis current deviation amount ΔIdc output by the d-axis current deviation calculation unit 304b, and outputs it as the dc-axis current deviation integral value ΔIdc-i.

The q-axis integrator 307b integrates the qc-axis current deviation amount ΔIqc output by the q-axis current deviation calculation unit 305b, and outputs it as the qc-axis current deviation integral value ΔIqc-i.

The second dc-axis voltage command calculation unit 403b calculates and outputs a second voltage command Vdc2* on the dc-qc axis based on the dc-axis current deviation amount ΔIdc output by the d-axis current deviation calculation unit 304b, the dc-axis current deviation integral value ΔIdc-i output by the d-axis integrator 306b, the qc-axis current deviation integral value ΔIqc-i output by the q-axis integrator 307b, and the vector control inverter frequency ω1.

The second qc-axis voltage command calculation unit 404b calculates and outputs a second voltage command Vqc2* on the dc-qc axis based on the qc-axis current deviation amount ΔIqc output by the q-axis current deviation calculation unit 305b, the qc-axis current deviation integral value ΔIqc-i output by the q-axis integrator 307b, the dc-axis current deviation integral value ΔIdc-i output by the d-axis integrator 306b, and the vector control inverter frequency ω1.

FIG. 11 is a block diagram showing an example of the configuration of the second dc axis voltage command calculation unit 403b according to the second embodiment.
The second dc axis voltage command calculation section 403b shown in FIG. 11 includes proportional units 701b, 702b and 703b, an adder 801, a subtractor 802 and a multiplier 901.

In second dc-axis voltage command calculation section 403b, proportional unit 701b multiplies the dc-axis current deviation amount ΔIdc by proportional gain Kpd2, proportional unit 702b multiplies the dc-axis current deviation integral value ΔIdc-i by proportional gain Kid2, and these are added together by adder 801. Furthermore, proportional unit 703b multiplies the qc-axis current deviation integral value ΔIqc-i by proportional gain Kqd2, and further multiplies it by inverter frequency ω1 by multiplier 901. The result obtained by adding and subtracting these values by subtractor 802 is set as second voltage command Vdc2* on the dc-qc axes. For example, it is calculated by the following equation (6).

ここで、式（７）において、Ｒ１＊は、誘導電動機の一次抵抗の制御設定値、Ｌσ＊は、誘導電動機の一次換算漏れインダクタンスの制御設定値、ωａｃｒ＊は、電流制御の制御応答角周波数である。 Here, when an induction motor is the control target, the proportional gains Kpd2, Kid2 and Kqd2 of the second dc axis voltage command calculation section 403b are calculated, for example, by the following equation (7).

Here, in equation (7), R1* is the control setting value of the primary resistance of the induction motor, Lσ* is the control setting value of the primary-equivalent leakage inductance of the induction motor, and ωacr* is the control response angular frequency of the current control.

The proportional gains Kpd2, Kid2, and Kqd2 of the second dc-axis voltage command calculation unit 403b may be constants, or may be calculated as functions of the current command value, current detection value, speed, and vector calculation period.

In addition, in the calculation of the proportional gains Kpd2, Kid2, and Kqd2 of the second dc-axis voltage command calculation unit 403b, the control response angular frequency ωacr* of the current control may be set to a value different from the control response angular frequency ωacr* of the current control in the calculation of the proportional gains Kpd and Kqd of the first voltage vector calculation unit 201b.

FIG. 12 is a block diagram showing an example of the configuration of the second qc-axis voltage command calculation unit 404b according to the second embodiment.
The second qc-axis voltage command calculation unit 404b shown in FIG. 12 includes proportional units 704b, 705b, 706b, and 507b, an adder 801, and a multiplier 901.

In the second qc-axis voltage command calculation unit 404b, a proportional unit 704b multiplies the qc-axis current deviation amount ΔIqc by a proportional gain Kpq2, a proportional unit 705b multiplies the qc-axis current deviation integral value ΔIqc-i by a proportional gain Kiq2, and an adder 801 adds the two together. A proportional unit 706b multiplies the dc-axis current deviation integral value ΔIdc-i by a proportional gain Kdq2, and a multiplier 901 multiplies the same by the inverter frequency ω1. These are added together by an adder 801, and the product of the inverter frequency ω1 multiplied by the proportional gain Kf2 by a proportional unit 507b and the product of the product added together by the adder 801 are used as a second voltage command Vqc2* on the dc-qc axes. For example, this is calculated using the following formula (8).

ここで、式（９）において、Ｒ１＊は、誘導電動機の一次抵抗の制御設定値、Ｌσ＊は、誘導電動機の一次換算漏れインダクタンスの制御設定値、ωａｃｒ＊は、電流制御の制御応答角周波数、Ｍ＊は、誘導電動機の相互インダクタンスの制御設定値、Ｌ２＊は、誘導電動機の二次自己インダクタンスの制御設定値、Φ２ｄ＊は、誘導電動機の二次磁束指令値である。 Here, the proportional gains Kpq2, Kiq2, Kdq2 and Kf2 of the second qc-axis voltage command calculation unit 404b are calculated, for example, by the following equation (9) when an induction motor is the controlled object.

Here, in equation (9), R1* is the control setting value of the primary resistance of the induction motor, Lσ* is the control setting value of the primary-equivalent leakage inductance of the induction motor, ωacr* is the control response angular frequency of current control, M* is the control setting value of the mutual inductance of the induction motor, L2* is the control setting value of the secondary self-inductance of the induction motor, and Φ2d* is the secondary magnetic flux command value of the induction motor.

Note that Φ2d* is calculated by multiplying the current command value Idc* on the dc-qc axis coordinate system by the mutual inductance control setting value M*, and passing it through a first-order lag filter whose filter time constant is equivalent to the secondary time constant T2 of the induction motor.

The proportional gains Kpq2, Kiq2, Kdq2, and Kf2 of the qc-axis voltage command calculation unit 402 may be constants, or may be calculated as functions of the current command value, current detection value, speed, and vector calculation period.

Furthermore, in the calculation of the proportional gains Kpq2, Kiq2, and Kdq2 of the second qc-axis voltage command calculation unit 404b, the control response angular frequency ωacr* of the current control may be set to a value different from the control response angular frequency ωacr* of the current control in the calculation of the proportional gains Kpq and Kiq of the first voltage vector calculation unit 201b.

Therefore, in the second embodiment, based on the inverter frequency ω1 of the vector control,
When the AC motor is rotating at medium to high speeds, the feedback control of the dc-axis current is proportional control, and the integrator output of the qc-axis current control is used to compensate for the voltage of the dq-axis interference term of the dc-axis voltage command Vdc*.
This prevents delays in control response due to the control method switching process when the DC power supply voltage fluctuates, as well as fluctuations in voltage, current, and torque, and improves control stability during high speed regeneration (when the motor torque is negative).

When the AC motor is stopped or rotating at a low speed, the feedback control of current for both the dc axis and the qc axis is configured as proportional/integral control, and the voltage of the dq axis interference term is guaranteed using the integrator output of the current control for each axis.
This makes it possible to compensate for the error between the motor constant and the control set value, thereby achieving high accuracy and high response of current control.

In the third embodiment of the present invention, in the calculation of the voltage commands Vdc* and Vqc* on the dc-qc axes by the vector control unit 112b of the second embodiment, the first voltage commands Vdc1* and Vqc1* output by the first voltage vector calculation unit 201b and the second voltage commands Vdc2* and Vqc2* output by the second voltage vector calculation unit 203b are switched based on the modulation rate of the inverter device, and are output as the voltage commands Vdc* and Vqc* on the dc-qc axes.

This allows the optimum voltage vector calculation method to be selected according to the tolerance up to the maximum voltage that the inverter device can output, making it possible to realize an AC motor drive control device that can improve the accuracy and response of current control at low modulation rates and improve control stability at high modulation rates and when the inverter output voltage is saturated compared to Example 1.

13 is a block diagram showing a configuration example of a drive control device for an AC motor according to a third embodiment of the present invention. Hereinafter, only differences in configuration will be described in comparison with the first embodiment shown in FIG.
The drive control device for an AC motor according to the third embodiment shown in FIG. 13 can be realized by using a vector control unit 112c instead of the vector control unit 112 of the first embodiment.

FIG. 14 is a block diagram illustrating an example of the configuration of a vector control unit 112c according to the third embodiment.
A vector control unit 112c shown in FIG. 14 includes a first voltage vector calculation unit 201b, a second voltage vector calculation unit 203b, a dc axis voltage command switch 601c, a qc axis voltage command switch 602c, an inverter frequency calculation unit 202, and a modulation factor calculation unit 603c.

The vector control unit 112c receives as input the current command values Idc* and Iqc* on the dc-qc coordinate system, the current detection values Idc and Iqc on the dc-qc coordinate system, the electrical angular frequency ωre of the rotational speed of the induction motor 103a, the voltage detection value Ecf of the smoothing capacitor 131, and the voltage amplitude command V1* output from the polar coordinate conversion unit 117.

Similar to the first embodiment (FIG. 3) and the second embodiment (FIG. 8), the inverter frequency calculation unit 202 calculates the inverter frequency ω1 for vector control based on the current command values Idc* and Iqc* and the electrical angular frequency ωre of the rotational speed of the induction motor 103a.
Similar to the second embodiment (FIG. 8), the first voltage vector calculation unit 201b calculates first voltage commands Vdc1* and Vqc1* on the dc-qc axes based on the current command values Idc* and Iqc*, the current detection values Idc and Iqc, and the inverter frequency ω1.

The second voltage vector calculation unit 203b calculates second voltage commands Vdc2* and Vqc2* on the dc-qc axes based on the same input as the first voltage vector calculation unit 201b, as in the second embodiment (Figure 8).

That is, the vector control unit 112c of the third embodiment has the same configuration as the vector control unit 112b of the second embodiment shown in FIG. 8, except for the dc-axis voltage command switch 601c, the qc-axis voltage command switch 602c, and the modulation factor calculation unit 603c.

Next, the dc-axis voltage command switch 601c, the qc-axis voltage command switch 602c, and the modulation factor calculator 603c, which are characteristic parts of the third embodiment, will be described in detail.
The modulation factor calculation unit 603c calculates and outputs a modulation factor command value Vc* for the inverter device based on the voltage detection value Ecf of the smoothing capacitor 131 and the voltage amplitude command V1* output from the polar coordinate conversion unit 117.

The dc-axis voltage command switch 601c switches between the first voltage command Vdc1* output by the first voltage vector calculation unit 201b and the second voltage command Vdc2* output by the second voltage vector calculation unit 203b based on the modulation rate command value Vc* output by the modulation rate calculation unit 603c, and outputs it as the voltage command Vdc* on the dc-qc axes.

Here, the dc-axis voltage command switch 601c outputs the second voltage command Vdc2* output by the second voltage vector calculation unit 203b as the voltage command Vdc* on the dc-qc axes if the modulation factor command value Vc* is equal to or less than 0.7 (threshold value), and outputs the first voltage command Vdc1* output by the first voltage vector calculation unit 201b as the voltage command Vdc* on the dc-qc axes if the modulation factor command value Vc* is greater than 0.7 (threshold value). Note that the threshold value of the modulation factor command value Vc* for switching the voltage command on the dc-qc axes may be set to any value.

The qc-axis voltage command switch 602c switches between the first voltage command Vqc1* output by the first voltage vector calculation unit 201b and the second voltage command Vqc2* output by the second voltage vector calculation unit 203b based on the modulation rate command value Vc* output by the modulation rate calculation unit 603c, and outputs it as the voltage command Vqc* on the dc-qc axes.

Here, for example, if the modulation factor command value Vc* is 0.7 (threshold value) or less, the qc-axis voltage command switch 602c outputs the second voltage command Vqc2* output by the second voltage vector calculation unit 203b as the voltage command Vqc* on the dc-qc axes, and if the modulation factor command value Vc* is greater than 0.7 (threshold value), the qc-axis voltage command switch 602c outputs the first voltage command Vqc1* output by the first voltage vector calculation unit 201b as the voltage command Vqc* on the dc-qc axes.
The threshold value of the modulation factor command value Vc* for switching the voltage commands on the dc and qc axes may be set to any value.

The voltage vector calculation unit that does not output the voltage calculation results as the voltage commands Vdc* and Vqc* on the dc-qc axes should stop the integral calculation of the current deviation amount, and when the voltage calculation results are output as the voltage commands Vdc* and Vqc* on the dc-qc axes and calculation is started, the output of the other integrator on the same axis should be set as the initial value of the integral calculation.

Therefore, in the third embodiment, based on the modulation rate command value Vc*,
When the modulation factor command value Vc* is large, the feedback control of the dc-axis current is proportional control, and the integrator output of the qc-axis current control is used to compensate for the voltage of the dq-axis interference term of the dc-axis voltage command Vdc*.
This prevents delays in control response due to the control method switching process when the DC power supply voltage fluctuates, as well as fluctuations in voltage, current, and torque, and improves control stability during high speed regeneration (negative motor torque).

When the modulation factor command value Vc* is small, the feedback control of the current for both the dc axis and the qc axis is configured as proportional-integral control, and the voltage of the dq axis interference term is guaranteed by using the integrator output of the current control for both.
This makes it possible to compensate for errors between the motor constants and the control set values, thereby achieving high precision and high responsiveness in current control.

In the fourth embodiment of the present invention, in the calculation of the voltage commands Vdc* and Vqc* on the dc-qc axes by the vector control unit 112b of the second embodiment, the first voltage commands Vdc1* and Vqc1* output by the first voltage vector calculation unit 201b and the second voltage commands Vdc2* and Vqc2* output by the second voltage vector calculation unit 203b are switched based on the number of PWM controlled output voltage pulses Npwm output by the inverter device during one period of the electrical angular frequency of the AC motor, and are output as the voltage commands Vdc* and Vqc* on the dc-qc axes.

This allows the optimum voltage vector calculation method to be selected according to the number of PWM control pulses output by the inverter device, realizing an AC motor drive control device that can improve current control accuracy, response, and control stability compared to Example 1.

15 is a block diagram showing a configuration example of a drive control device for an AC motor according to a fourth embodiment of the present invention. In comparison with the first embodiment shown in FIG. 1, only differences in the configuration will be described.
The drive control device for an AC motor according to the fourth embodiment shown in FIG. 15 can be realized by using a vector control unit 112d instead of the vector control unit 112 of the first embodiment shown in FIG.

FIG. 16 is a block diagram illustrating an example of the configuration of a vector control unit 112d according to the fourth embodiment.
A vector control unit 112d shown in FIG. 16 includes a first voltage vector calculation unit 201b, a second voltage vector calculation unit 203b, a dc axis voltage command switch 601d, a qc axis voltage command switch 602d, an inverter frequency calculation unit 202, and a pulse number calculation unit 604d.
Like the vector control unit 112c of the third embodiment shown in FIG. 14, the vector control unit 112d receives as input the current command values Idc* and Iqc* on the dc-qc axis coordinate system, the current detection values Idc and Iqc on the dc-qc axis coordinate system, the electrical angular frequency ωre of the rotational speed of the induction motor 103a, the voltage detection value Ecf of the smoothing capacitor 131, and the voltage amplitude command V1* output from the polar coordinate conversion unit 117.

The inverter frequency calculation unit 202 calculates the vector control inverter frequency ω1 based on the current command values Idc* and Iqc* and the electrical angular frequency ωre of the rotational speed of the induction motor 103a, as in the first embodiment (FIG. 3), the second embodiment (FIG. 8), and the third embodiment (FIG. 14).

The first voltage vector calculation unit 201b calculates the first voltage commands Vdc1* and Vqc1* on the dc-qc axes based on the current command values Idc* and Iqc*, the current detection values Idc and Iqc, and the inverter frequency ω1, similar to the second embodiment (FIG. 8) and the third embodiment (FIG. 14).

The second voltage vector calculation unit 203b calculates second voltage commands Vdc2* and Vqc2* on the dc-qc axes based on the same input as the first voltage vector calculation unit 201b, similar to the second embodiment (FIG. 8) and the third embodiment (FIG. 14).

That is, the vector control unit 112d of the fourth embodiment has the same configuration as the vector control unit 112b of the second embodiment shown in FIG. 8, except for the dc-axis voltage command switch 601d, the qc-axis voltage command switch 602d, and the pulse number calculation unit 604d, and uses the pulse number calculation unit 604d instead of the modulation rate calculation unit 603c of the vector control unit 112c of the third embodiment shown in FIG. 14.

Next, the dc-axis voltage command switch 601d, the qc-axis voltage command switch 602d, and the pulse number calculation unit 604d, which are characteristic parts of the fourth embodiment of the present invention, will be described in detail.
The pulse number calculation unit 604d calculates and outputs the number of PWM controlled output voltage pulses Npwm output by the inverter device during one period of the electrical angular frequency of the AC motor, based on the voltage detection value Ecf of the smoothing capacitor 131, the voltage amplitude command V1* output from the polar coordinate conversion unit 117, and the electrical angular frequency ωre of the rotation speed of the induction motor 103a. Note that the inverter frequency ω1 of the vector control output from the inverter frequency calculation unit 202 may be used instead of the electrical angular frequency ωre of the rotation speed of the induction motor 103a.

Here, the number of pulses Npwm can be calculated, for example, by using a reference table of the number of pulses corresponding to a pre-set PWM control mode based on the voltage detection value Ecf, the modulation factor command value calculated from the voltage amplitude command V1*, and the electrical angular frequency ωre of the rotational speed of the induction motor 103a (or the inverter frequency ω1 for vector control).

The dc-axis voltage command switch 601d switches between the first voltage command Vdc1* output by the first voltage vector calculation unit 201b and the second voltage command Vdc2* output by the second voltage vector calculation unit 203b based on the pulse number Npwm output by the pulse number calculation unit 604d, and outputs it as the voltage command Vdc* on the dc-qc axes.

Here, if the pulse number Npwm is greater than 9 (threshold), the dc-axis voltage command switch 601d outputs the second voltage command Vdc2* output by the second voltage vector calculation unit 203b as the voltage command Vdc* on the dc-qc axes, and if the pulse number Npwm is equal to or less than 9 (threshold), the dc-axis voltage command switch 601d outputs the first voltage command Vdc1* output by the first voltage vector calculation unit 201b as the voltage command Vdc* on the dc-qc axes.

The qc-axis voltage command switch 602d switches between the first voltage command Vqc1* output by the first voltage vector calculation unit 201b and the second voltage command Vqc2* output by the second voltage vector calculation unit 203b based on the pulse number Npwm output by the pulse number calculation unit 604d, and outputs it as the voltage command Vqc* on the dc-qc axes.

Here, for example, if the pulse number Npwm is greater than 9 (threshold value), the qc-axis voltage command switch 602d outputs the second voltage command Vqc2* output by the second voltage vector calculation unit 203b as the voltage command Vqc* on the dc-qc axes, and if the pulse number Npwm is equal to or less than 9 (threshold value), the qc-axis voltage command switch 602d outputs the first voltage command Vqc1* output by the first voltage vector calculation unit 201b as the voltage command Vqc* on the dc-qc axes.
The threshold value of the number of pulses Npwm at which the voltage commands on the dc and qc axes are switched may be set to any value.

The voltage vector calculation unit that does not output the voltage calculation results as the voltage commands Vdc* and Vqc* on the dc-qc axes should stop the integral calculation of the current deviation amount, and when the voltage calculation results are output as the voltage commands Vdc* and Vqc* on the dc-qc axes and calculation is started, the output of the other integrator on the same axis should be set as the initial value of the integral calculation.

Therefore, in the fourth embodiment, based on the number of pulses Npwm,
When the pulse number Npwm is small, the current feedback control system is prone to becoming unstable, so the feedback control of the dc-axis current is made proportional control, and the integrator output of the qc-axis current control is used to compensate for the voltage of the dq-axis interference term of the dc-axis voltage command Vdc*.
This prevents delays in control response due to the control method switching process when the DC power supply voltage fluctuates, as well as fluctuations in voltage, current, and torque, and improves control stability during high speed regeneration (negative motor torque).

When the pulse number Npwm is large, the feedback control of the current for both the dc axis and the qc axis is configured as proportional-integral control, and the voltage of the dq axis interference term is guaranteed by using the integrator output of the current control for both.
This makes it possible to compensate for errors between the motor constants and the control set values, and to achieve a more stable drive control device for the AC motor and higher accuracy and response of current control than in the first embodiment.

In the fifth embodiment of the present invention, in the calculation of the voltage commands Vdc* and Vqc* on the dc-qc axes by the vector control unit 112b of the second embodiment, the first voltage commands Vdc1* and Vqc1* output by the first voltage vector calculation unit 201b and the second voltage commands Vdc2* and Vqc2* output by the second voltage vector calculation unit 203b are switched based on the output voltage update period Tp of the gate command signal by PWM control (hereinafter referred to as the "PWM voltage update period Tp"), and are output as the voltage commands Vdc* and Vqc* on the dc-qc axes.

This allows the optimum voltage vector calculation method to be selected according to the output voltage update period Tp (PWM voltage update period Tp) of the PWM control gate command signal output by the inverter device, thereby realizing an AC motor control device that can improve current control accuracy, improve responsiveness, and improve control stability compared to Example 1.

Fifth embodiment of the present invention will be described below with reference to FIG 17, which is a block diagram showing an example of the configuration of a drive control device for an AC motor according to a fifth embodiment of the present invention.
The drive control device for an AC motor according to the fifth embodiment shown in FIG. 17 can be realized by using a vector control unit 112e and a PWM signal controller 119e instead of the vector control unit 112 and the PWM signal controller 119 of the first embodiment.

The PWM signal controller 119e of the fifth embodiment outputs a gate command signal generated from the calculation result of pulse width modulation based on the voltage detection value Ecf of the smoothing capacitor 131 and the three-phase AC voltage commands Vu*, Vv*, and Vw*, and also outputs the PWM voltage update period Tp of the gate command signal.

FIG. 18 is a block diagram illustrating a configuration example of a vector control unit 112e according to the fifth embodiment.
A vector control unit 112e shown in FIG. 18 includes a first voltage vector calculation unit 201b, a second voltage vector calculation unit 203b, a dc-axis voltage command switch 601e, a qc-axis voltage command switch 602e, and an inverter frequency calculation unit 202.

The vector control unit 112e receives as input the current command values Idc* and Iqc* on the dc-qc coordinate system, the current detection values Idc and Iqc on the dc-qc coordinate system, the electrical angular frequency ωre of the rotational speed of the induction motor 103a, and the PWM voltage update period Tp output from the PWM signal controller 119e.

The inverter frequency calculation unit 202 calculates the vector control inverter frequency ω1 based on the current command values Idc* and Iqc* and the electrical angular frequency ωre of the rotational speed of the induction motor 103a, as in the first to fourth embodiments.

The first voltage vector calculation unit 201b calculates the first voltage commands Vdc1* and Vqc1* on the dc-qc axes based on the current command values Idc* and Iqc*, the current detection values Idc and Iqc, and the inverter frequency ω1, in the same manner as in the second to fourth embodiments.

The second voltage vector calculation unit 203b calculates second voltage commands Vdc2* and Vqc2* on the dc-qc axes based on the same input as the first voltage vector calculation unit 201b, as in the second to fourth embodiments.

That is, the vector control unit 112e of the fifth embodiment has the same configuration as the vector control unit 112b of the second embodiment shown in FIG. 8, except for the dc-axis voltage command switch 601e and the qc-axis voltage command switch 602e.

Next, the dc-axis voltage command switch 601e and the qc-axis voltage command switch 602e, which are characteristic of the fifth embodiment, will be described in detail.
The dc axis voltage command switch 601e switches between the first voltage command Vdc1* output by the first voltage vector calculation unit 201b and the second voltage command Vdc2* output by the second voltage vector calculation unit 203b based on the PWM voltage update period Tp output by the PWM signal controller 119e, and outputs it as the voltage command Vdc* on the dc-qc axes.

Here, the dc-axis voltage command switch 601e, for example, outputs the second voltage command Vdc2* output by the second voltage vector calculation unit 203b as the voltage command Vdc* on the dc-qc axes if the PWM voltage update period Tp is shorter than an arbitrary threshold value, and outputs the first voltage command Vdc1* output by the first voltage vector calculation unit 201b as the voltage command Vdc* on the dc-qc axes if the PWM voltage update period Tp is longer than an arbitrary threshold value.

The qc-axis voltage command switch 602e switches between the first voltage command Vqc1* output by the first voltage vector calculation unit 201b and the second voltage command Vqc2* output by the second voltage vector calculation unit 203b based on the PWM voltage update period Tp output by the PWM signal controller 119e, and outputs it as the voltage command Vqc* on the dc-qc axes.

Here, the qc-axis voltage command switch 602e outputs, for example, the second voltage command Vqc2* output by the second voltage vector calculation unit 203b as the voltage command Vqc* on the dc-qc axes if the PWM voltage update period Tp is shorter than an arbitrary threshold value, and outputs the first voltage command Vqc1* output by the first voltage vector calculation unit 201b as the voltage command Vqc* on the dc-qc axes if the PWM voltage update period Tp is longer than an arbitrary threshold value.

The voltage vector calculation unit that does not output the voltage calculation results as the voltage commands Vdc* and Vqc* on the dc-qc axes should stop the integral calculation of the current deviation amount, and when the voltage calculation results are output as the voltage commands Vdc* and Vqc* on the dc-qc axes and calculation is started, the output of the other integrator on the same axis should be set as the initial value of the integral calculation.

Therefore, in the fifth embodiment, the method of calculating the vector voltage is switched based on the PWM voltage update period Tp of the gate command signal by PWM control. This makes it possible to avoid control instability in the high-speed range when the PWM voltage update period Tp is not sufficiently short compared to one period of the electrical angular frequency of the AC motor, and realizes a drive control device for an AC motor that can improve control stability compared to the first embodiment.

In the sixth embodiment, in the calculation of the voltage commands Vdc* and Vqc* on the dc-qc axes by the vector control unit 112b in the second embodiment, the q-axis integrator 303 of the first voltage vector calculation unit 201b and the q-axis integrator 307b of the second voltage vector calculation unit 203b are shared. This makes it possible to realize a drive control device for an AC motor that has a simpler configuration than the second embodiment and is capable of improving the accuracy and responsiveness of current control in the low speed range and improving control stability in the high speed range.

19 is a block diagram showing an example of the configuration of a drive control device for a current motor according to a sixth embodiment of the present invention. Hereinafter, only the differences in the configuration will be described in comparison with the second embodiment shown in FIG.
The drive control device for an AC motor according to the sixth embodiment shown in FIG. 19 can be realized by using a vector control unit 112f instead of the vector control unit 112b of the second embodiment.

FIG. 20 is a block diagram illustrating a configuration example of a vector control unit 112f according to the sixth embodiment.
A vector control unit 112 f shown in FIG. 20 includes a voltage vector calculation unit 201 f and an inverter frequency calculation unit 202 .

The vector control unit 112f receives as input the current command values Idc* and Iqc* on the dc-qc axis coordinate system, the current detection values Idc and Iqc on the dc-qc axis coordinate system, and the electrical angular frequency ωre of the rotational speed of the induction motor 103a, and calculates and outputs the voltage commands Vdc* and Vqc* on the dc-qc axes.

The inverter frequency calculation unit 202 calculates the vector control inverter frequency ω1 based on the current command values Idc* and Iqc* and the electrical angular frequency ωre of the rotational speed of the induction motor 103a, as in the first to fifth embodiments.

Next, the vector control unit 112f, which is a characteristic feature of the sixth embodiment, will be described in detail.
FIG. 21 is a block diagram illustrating a configuration example of a voltage vector calculation unit 201f according to the sixth embodiment.
A voltage vector calculation unit 201f shown in FIG. 20 includes a d-axis current deviation calculation unit 304f, a q-axis current deviation calculation unit 305f, a d-axis integrator 306f, a q-axis integrator 307f, a dc-axis voltage command calculation unit 405f, and a qc-axis voltage command calculation unit 406f.

The d-axis current deviation calculation unit 304f subtracts the current detection value Idc on the dc-qc coordinate system from the current command value Idc* on the dc-qc coordinate system, and outputs the result as the dc-axis current deviation amount ΔIdc.

The q-axis current deviation calculation unit 305f subtracts the current detection value Iqc on the dc-qc coordinate system from the current command value Iqc* on the dc-qc coordinate system, and outputs the result as the qc-axis current deviation amount ΔIqc.

The d-axis integrator 306f integrates the dc-axis current deviation amount ΔIdc output by the d-axis current deviation calculation unit 304f, and outputs it as the dc-axis current deviation integral value ΔIdc-i.

The q-axis integrator 307f integrates the qc-axis current deviation amount ΔIqc output by the q-axis current deviation calculation unit 305f, and outputs it as the qc-axis current deviation integral value ΔIqc-i.

The dc-axis voltage command calculation unit 405f calculates and outputs a voltage command Vdc* on the dc-qc axes based on the current command value Idc* on the dc-qc axis coordinate system output by the current command calculation unit 111, the dc-axis current deviation amount ΔIdc output by the d-axis current deviation calculation unit 304f, the dc-axis current deviation integral value ΔIdc-i output by the d-axis integrator 306f, the qc-axis current deviation integral value ΔIqc-i output by the q-axis integrator 307f, and the inverter frequency ω1 output by the inverter frequency calculation unit 202.

The qc-axis voltage command calculation unit 406f calculates and outputs a voltage command Vqc* on the dc-qc axis based on the current command value Idc* on the dc-qc axis coordinate system output by the current command calculation unit 111, the qc-axis current deviation amount ΔIqc output by the q-axis current deviation calculation unit 305f, the qc-axis current deviation integral value ΔIqc-i output by the q-axis integrator 307f, the dc-axis current deviation integral value ΔIdc-i output by the d-axis integrator 306f, and the inverter frequency ω1 output by the inverter frequency calculation unit 202.

FIG. 22 is a block diagram showing an example of the configuration of a dc axis voltage command calculation unit 405f according to the sixth embodiment.
22 includes proportional units 501f, 502f, 503f and 702f, a dc axis voltage command switch 601f, an adder 801, a subtractor 802 and a multiplier 901. The dc axis voltage command calculation unit 405f shown in FIG.

In dc axis voltage command calculation section 405f, ΔVdc-r* output from dc axis voltage command switch 601f is multiplied by proportional gain Kpd by proportional unit 502f, and the result is added by adder 801, from which qc axis current deviation integral value ΔIqc-i is multiplied by proportional gain Kqd by proportional unit 503f, which is further multiplied by inverter frequency ω1 by multiplier 901, and the result is subtracted by subtractor 802 to output as voltage command Vdc* on the dc-qc axes. For example, it is calculated using the following equation (10).

ここで、式（１１）において、Ｒ１＊は、誘導電動機の一次抵抗の制御設定値、Ｌσ＊は、誘導電動機の一次換算漏れインダクタンスの制御設定値、ωａｃｒ＊は、電流制御の制御応答角周波数である。 Here, the proportional gains Kr, Kpd, Kid and Kqd of dc axis voltage command calculation section 405f are calculated, for example, by the following equation (11) when an induction motor is the control target.

Here, in equation (11), R1* is the control setting value of the primary resistance of the induction motor, Lσ* is the control setting value of the primary-equivalent leakage inductance of the induction motor, and ωacr* is the control response angular frequency of the current control.

The proportional gains Kr, Kpd, Kid, and Kqd of the dc-axis voltage command calculation unit 405f may be constants, or may be calculated as functions of the current command value, current detection value, speed, and vector calculation period.

In addition, in the calculation of the proportional gains Kpd, Kid, and Kqd of the dc-axis voltage command calculation unit 405f, the control response angular frequency ωacr* of the current control may use different control response angular frequency setting values.

The dc-axis voltage command switch 601f switches between the current command value Idc* on the dc-qc axis coordinate system multiplied by the proportional gain Kr in the proportional converter 501f and the dc-axis current deviation integral value ΔIdc-i multiplied by the proportional gain Kid in the proportional converter 702f based on the inverter frequency ω1, and outputs it as a voltage command ΔVdc-r* that compensates for the voltage drop in the primary resistance due to the dc-axis current.

Here, if the inverter frequency ω1 is, for example, equal to or less than 1/3 (threshold) of the maximum rotation speed of the AC motor, the dc-axis voltage command switch 601f outputs the value obtained by multiplying the dc-axis current deviation integral value ΔIdc-i by the proportional gain Kid as the compensation voltage command ΔVdc-r*. If the inverter frequency ω1 is greater than 1/3 (threshold) of the maximum rotation speed of the AC motor, the switch 601f outputs the value obtained by multiplying the current command value Idc* on the dc-qc axis coordinate system by the proportional gain Kr as the compensation voltage command ΔVdc-r*. Note that the threshold value of the inverter frequency ω1 for switching the compensation voltage command on the dc-qc axis may be set to any value.

FIG. 23 is a block diagram showing an example of the configuration of a qc-axis voltage command calculation unit 406f according to the sixth embodiment.
The qc-axis voltage command calculation unit 406f shown in FIG. 23 includes proportional units 504f, 505f, 506f, 507, and 706f, a qc-axis voltage command switch 602f, an adder 801, and a multiplier 901.

The qc-axis voltage command calculation unit 406f adds, in an adder 801, the product of multiplying the qc-axis current deviation amount ΔIqc by a proportional gain Kpq in a proportional unit 506f and the product of multiplying the qc-axis current deviation integral value ΔIqc-i by a proportional gain Kiq in a proportional unit 505f, to ΔVqc-d* output from the qc-axis voltage command switch 602f, and further adds, in an adder 801, the product of multiplying the inverter frequency ω1 by a proportional gain Kf in a proportional unit 507, and outputs the result as a voltage command Vqc* on the dc-qc axes. For example, it is calculated by the following equation (12).

ここで、式（１３）において、Ｒ１＊は、誘導電動機の一次抵抗の制御設定値、Ｌσ＊は、誘導電動機の一次換算漏れインダクタンスの制御設定値、ωａｃｒ＊は、電流制御の制御応答角周波数、Ｍ＊は、誘導電動機の相互インダクタンスの制御設定値、Ｌ２＊は、誘導電動機の二次自己インダクタンスの制御設定値、Φ２ｄ＊は、誘導電動機の二次磁束指令値である。 Here, the proportional gains Kpq, Kiq, Kdq, Kdq2 and Kf of the qc-axis voltage command calculation unit 406f are calculated, for example, by the following equation (13) when an induction motor is the control target.

Here, in equation (13), R1* is the control setting value of the primary resistance of the induction motor, Lσ* is the control setting value of the primary-equivalent leakage inductance of the induction motor, ωacr* is the control response angular frequency of current control, M* is the control setting value of the mutual inductance of the induction motor, L2* is the control setting value of the secondary self-inductance of the induction motor, and Φ2d* is the secondary magnetic flux command value of the induction motor.

Note that Φ2d* is calculated by multiplying the current command value Idc* on the dc-qc axis coordinate system by the mutual inductance control setting value M*, and passing it through a first-order lag filter whose filter time constant is equivalent to the secondary time constant T2 of the induction motor.

The proportional gains Kpq, Kiq, Kdq, Kdq2, and Kf of the qc-axis voltage command calculation unit 406f may be constants, or may be calculated as functions of the current command value, current detection value, speed, and vector calculation period.

Furthermore, in the calculation of the proportional gains Kpq, Kiq, and Kdq2 of the qc-axis voltage command calculation unit 406f, the control response angular frequency ωacr* of the current control may use different control response angular frequency setting values.

The qc-axis voltage command switch 602f switches between the current command value Idc* on the dc-qc axis coordinate system multiplied by the proportional gain Kdq in the proportional unit 504f and then multiplied by the inverter frequency ω1 in the multiplier 901, and the dc-axis current deviation integral value ΔIdc-i multiplied by the proportional gain Kdq2 in the proportional unit 706f and then multiplied by the inverter frequency ω1 in the multiplier 901, based on the inverter frequency ω1, and outputs it as a voltage command ΔVqc-d* that compensates for the voltage of the dq interference term.

Here, the qc-axis voltage command switch 602f outputs the value obtained by multiplying the dc-axis current deviation integral value ΔIdc-i by the proportional gain Kdq2 and the inverter frequency ω1 as the compensation voltage command ΔVqc-d* if the inverter frequency ω1 is, for example, equal to or less than 1/3 (threshold) of the maximum rotation speed of the AC motor. Also, if the inverter frequency ω1 is greater than 1/3 (threshold) of the maximum rotation speed of the AC motor, the switch 602f outputs the value obtained by multiplying the current command value Idc* on the dc-qc axis coordinate system by the proportional gain Kdq and the inverter frequency ω1 as the compensation voltage command ΔVqc-d*. Note that the threshold value of the inverter frequency ω1 for switching the compensation voltage command on the dc-qc axis may be set to any value.

However, if the dc-axis current deviation integral value ΔIdc-i output from the d-axis integrator 306f is not reflected in the calculation of the voltage commands Vdc* and Vqc* on the dc-qc axes, the integral calculation of the dc-axis current deviation amount must be stopped.

Therefore, in the sixth embodiment, in the calculation of the voltage commands Vdc* and Vqc* on the dc-qc axes by the vector control unit 112b of the second embodiment, the integrator output of the current control on the qc axis is used to compensate for the voltage of the dq-axis interference term of the voltage command Vdc* on the dc axis, and the compensation voltage command calculation based on the current command value Idc* on the dc-qc axis coordinate system and the compensation voltage command calculation based on the dc-axis current deviation integral value ΔIdc-i are switched according to the inverter frequency ω1. This makes it possible to realize a drive control device for an AC motor that has a simpler configuration than the second embodiment, and that is capable of improving the accuracy and responsiveness of current control in the low speed range and improving control stability in the high speed range.

A seventh embodiment of the present invention is an embodiment in which the drive control device for an AC motor according to any one of the first to sixth embodiments is applied to a railway vehicle.
FIG. 24 is a diagram showing a schematic configuration of a part of a railway vehicle equipped with a drive control device for an AC motor using any one of the first to sixth embodiments, as a seventh embodiment of the present invention.

The railway vehicle shown in FIG. 24 has a bogie on which AC motors 103a and 103b are mounted, and a bogie on which AC motors 103c and 103d are mounted, and is equipped with an AC motor drive control device including a controller 101, a power converter 102, a command generator 105, and a phase current detection unit 121.

The railway vehicle according to the seventh embodiment drives the AC motor 103 by converting power supplied from the overhead line via a current collector into AC power by a power converter 102 and supplying it to the AC motor 103 in accordance with a torque command value Tm* generated by a command generator 105 based on an operation command input by a driver via a master controller. The AC motor 103 is connected to the axle of the railway vehicle, and the running of the railway vehicle is controlled by the AC motor 103.

In the railway vehicle according to the seventh embodiment, by applying any one of the first to sixth embodiments as a drive control device for an AC motor, it is possible to improve control stability and control response.

Although the above describes each embodiment of the present invention, the present invention can be applied to other AC motors as well. For example, the same applies to permanent magnet synchronous motors, wound-rotor synchronous motors, and synchronous machines using reluctance torque. The present invention can also be applied to the control of generators. Furthermore, the present invention is not limited to the above-described embodiments, and various modifications are possible without departing from the spirit of the present invention.

101 Controller, 102 Power converter, 103 AC motor, 105 Command generator,
111 Current command calculation unit, 112 Vector control unit, 113 Current detection unit,
114 dq coordinate conversion unit, 115 velocity calculation unit, 116 phase calculation unit,
117 polar coordinate conversion unit, 118 UVW coordinate conversion unit, 119 PWM signal controller,
121 Phase current detection unit, 123 Input terminal, 124 Rotational speed detection unit,
131 smoothing capacitor, 132 main circuit section, 133 gate driver,
134 DC resistor, 201 voltage vector calculation unit,
201b first voltage vector calculation unit, 202 inverter frequency calculation unit,
203b second voltage vector calculation unit, 301, 304b d-axis current deviation calculation unit,
302, 305b q-axis current difference calculation unit, 303, 307b q-axis integrator,
306b d-axis integrator, 401 dc-axis voltage command calculation unit,
401b first dc-axis voltage command calculation unit, 402 qc-axis voltage command calculation unit,
402b: a first qc-axis voltage command calculation unit; 403b: a second dc-axis voltage command calculation unit;
404b: second qc-axis voltage command calculation unit; 405f: dc-axis voltage command calculation unit;
406f qc-axis voltage command calculation unit,
501, 502, 503, 504, 505, 506, 507 proportional device,
601b dc-axis voltage command switch, 602b qc-axis voltage command switch,
603c modulation rate calculation unit, 604d pulse number calculation unit,
701b, 702b, 703b, 704b, 705b, 706b proportional device,
801 adder, 802 subtractor, 901 multiplier

Claims (7)

A drive control device for an AC motor that is driven by a power converter using vector control based on a voltage command,
a current command calculation unit that generates current command values for each of a d-axis and a q-axis, which are two orthogonal rotation coordinate axes of the AC motor;
current detection means for detecting a current flowing through the AC motor and converting the detected current into a current detection value for each of the d-axis and the q-axis;
a voltage command calculation unit that calculates the voltage commands for the d-axis and the q-axis based on current command values for the d-axis and the q-axis, current detection values for the d-axis and the q-axis, and a rotation speed of the AC motor,
The voltage command calculation unit is
the d-axis voltage command is calculated by adding a result of a proportional calculation of the d-axis current command value to a result of the proportional calculation of the d-axis current deviation between the d-axis current command value and the d-axis current detection value by d-axis current feedback control based only on the proportional calculation of the d-axis current deviation between the d-axis current command value and the d-axis current detection value, and subtracting from the added value a value obtained by multiplying a result of a proportional integral calculation of a q-axis current deviation between the q-axis current command value and the q-axis current detection value by the rotation speed;
a voltage command for the q-axis that is calculated by adding together the results of both calculations using q-axis current feedback control based on a proportional calculation of the q-axis current deviation and a proportional integral calculation of the q-axis current deviation, and then adding to the sum a value obtained by multiplying the proportional calculation result of the d-axis current command value by the rotational speed and a value obtained by the proportional calculation result of the rotational speed, respectively. 2. The drive control device for an AC motor according to claim 1,
The voltage command calculation unit is a first voltage command calculation unit, and a second voltage command calculation unit different from the first voltage command calculation unit is further provided,
The second voltage command calculation unit includes:
calculating the second voltage command of the d-axis by adding up the results of both a proportional calculation of the d-axis current deviation and a proportional integral calculation of the d-axis current deviation through d-axis current feedback control based on the proportional integral calculation of the d-axis current deviation, and subtracting from the sum a value obtained by multiplying the proportional integral calculation result of the q-axis current deviation by the rotation speed;
calculating the second voltage command of the q-axis by adding up the results of both proportional and integral calculations of the q-axis current deviation through q-axis current feedback control based on the proportional and integral calculation of the q-axis current deviation, and adding a value obtained by multiplying the proportional and integral calculation result of the d-axis current deviation by the rotation speed and a value obtained by the proportional calculation result of the rotation speed to the sum;
a first voltage command calculation unit and a second voltage command calculation unit that are switched for use in accordance with at least one of a rotation speed of the AC motor, a modulation factor of the power converter, a number of PWM control pulses output by the power converter, and a gate command signal voltage update period output to the power converter. 3. The drive control device for an AC motor according to claim 1,
13. A drive control device for an AC motor, comprising: a power converter that drives the power converter in a range from multi-pulse drive to one-pulse drive based on the voltage commands for the d-axis and the q-axis, respectively. A railway vehicle equipped with a drive control device for an AC motor according to any one of claims 1 to 3. A drive control method for an AC motor, which performs vector control on an AC motor driven by a power converter based on a voltage command, comprising:
generating current command values for each of a d-axis and a q-axis, which are two orthogonal rotation coordinate axes of the AC motor;
A current flowing through the AC motor is converted into a current detection value of each of the d-axis and the q-axis.
As a method of generating the voltage command separately for the d-axis and the q-axis,
the d-axis voltage command is calculated by adding a result of a proportional calculation of the d-axis current command value to a result of the proportional calculation of the d-axis current deviation between the d-axis current command value and the d-axis current detection value by d-axis current feedback control based only on the proportional calculation of the d-axis current deviation between the d-axis current command value and the d-axis current detection value, and subtracting from the added value a value obtained by multiplying a result of a proportional integral calculation of a q-axis current deviation between the q-axis current command value and the q-axis current detection value by a rotation speed of the AC motor ;
a voltage command for the q-axis being calculated by adding together the results of both proportional and integral calculations of the q-axis current deviation through q-axis current feedback control, and then adding to the sum a value obtained by multiplying the proportional calculation result of the d-axis current command value by the rotational speed and a value obtained by the proportional calculation result of the rotational speed, respectively. 6. A method for controlling an AC motor according to claim 5, comprising the steps of:
A method of generating voltage commands separately for the d-axis and the q-axis is defined as a first method, and a method of generating voltage commands separately for the d-axis and the q-axis is defined as a second method.
calculating the d-axis voltage command by adding together the results of both proportional and integral calculations of the d-axis current deviation through d-axis current feedback control based on the proportional and integral calculation of the d-axis current deviation, and subtracting from the sum a value obtained by multiplying the proportional and integral calculation result of the q-axis current deviation by the rotation speed;
a second method for calculating the voltage command of the q-axis by adding the results of both calculations by q-axis current feedback control based on a proportional calculation of the q-axis current deviation and a proportional integral calculation of the q-axis current deviation, and adding a value obtained by multiplying the proportional integral calculation result of the d-axis current deviation by the rotation speed and a value of the proportional calculation result of the rotation speed to the sum,
A drive control method for an AC motor, characterized in that the first method and the second method are switched and executed depending on at least one of a rotation speed of the AC motor, a modulation factor of the power converter, a number of PWM control pulses output by the power converter, and an update period of a gate command signal voltage output to the power converter. 7. A method for controlling an AC motor according to claim 5, further comprising the steps of:
A drive control method for an AC motor, comprising the steps of: operating the power converter in a range from multi-pulse drive to one-pulse drive.

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