JP7658229B2 - MOTOR CONTROL DEVICE, MOTOR CONTROL METHOD, AND ELECTRIC POWER STEERING DEVICE - Google Patents

Description

The present invention relates to a motor control device, a motor control method, and an electric power steering device.

Conventionally, AC motors such as those provided in electric power steering devices (EPS) are supplied with power from a power source capable of controlling the frequency, voltage, current, phase, etc., and their operation is controlled. For example, in electric power steering devices, feedback control of the motor current is performed to appropriately generate torque for steering assist. In the feedback control, the motor applied voltage is adjusted so that the difference between the current command value and the actual current value is reduced. In general, the motor applied voltage is adjusted by PWM (Pulse Width Modulation) control.

An example of a motor used in an electric power steering device is a synchronous motor. Brushless motors, which have excellent durability and maintainability and are quiet, are commonly used as synchronous motors. A known method for driving brushless motors is vector control, which controls the motor current in a dq rotating coordinate system defined by the d-axis and q-axis.

The motor current flowing through a brushless motor contains harmonic components in addition to fundamental components. Torque ripple caused by these harmonic components is one of the causes of vibration and noise. In order to reduce this torque ripple, for example, Patent Document 1 describes a method of superimposing harmonic components with frequencies that are integer multiples of the fundamental component on the current command value of the fundamental component on a dq rotating coordinate system, and controlling the current based on the current command value on which the harmonic components are superimposed.

Patent Document 2 also describes that in order to reduce torque ripple that occurs in the high rotation range of the motor of an electric power steering device, a torque ripple reducing superimposition component is superimposed on the motor current during a period corresponding to the high rotation range.

Patent No. 4484325 JP 2019-88130 A

In Patent Document 1 and Patent Document 2, many functional units are provided for feedback signals in order to reduce torque ripple. The functional units for feedback signals include a functional unit for extracting harmonic components from the motor current flowing through the motor and converting them into a rotating coordinate system, a functional unit for adjusting the harmonic components obtained from the motor current using a gain so as to perform tracking control, and a functional unit for inversely converting the adjusted harmonic components to reflect them in the fundamental wave component. Furthermore, to improve the accuracy of extracting harmonic components from the motor current, a functional unit equipped with a low-pass filter is also required.

Increasing the number of functional parts reduces the responsiveness of the entire device. For example, in a conventional configuration, if you try to improve the accuracy of extracting harmonic components from the motor current by using a low-pass filter or the like, a trade-off inevitably occurs between the accuracy and responsiveness of the control. From the perspective of responsiveness, there is a need for a configuration that can reduce torque ripple while suppressing the number of functional parts in the device.

In view of the above problems, the present invention aims to appropriately reduce torque ripple generated in an AC motor while minimizing the number of functional parts in the device compared to conventional methods.

In order to solve the above problems, the present invention has the following configuration. That is, a motor control device comprising:
a first calculation means for calculating a first two-phase current command value corresponding to a torque command value of the synchronous motor based on an angular velocity of the synchronous motor;
a second calculation means for calculating a second two-phase current command value corresponding to a 6n-th order component in a dq rotating coordinate system based on the torque command value and a rotation angle of the synchronous motor;
a first conversion means for converting a three-phase current value supplied to the synchronous motor into a two-phase current value based on a rotation angle of the synchronous motor;
a derivation means for deriving a first two-phase voltage command value by using a difference between a two-phase current command value obtained by superimposing the first two-phase current command value and the second two-phase current command value and the two-phase current value converted by the first conversion means;
a second conversion means for converting the first two-phase voltage command value into a three-phase voltage command value;
a supply means for supplying a three-phase voltage to the synchronous motor based on the three-phase voltage command value,
The derivation means is
performing PI control using the difference to calculate a second two-phase voltage command value;
calculating a correction value for the second two-phase voltage command value based on the difference and a rotation angle of the synchronous motor;
The second two-phase voltage command value is corrected by the correction value to derive the first two-phase voltage command value.

Another aspect of the present invention has the following configuration. That is, a motor control method comprising:
a first calculation step of calculating a first two-phase current command value corresponding to a torque command value of the synchronous motor based on an angular velocity of the synchronous motor;
a second calculation step of calculating a second two-phase current command value corresponding to a 6n-th order component in a dq rotating coordinate system based on the torque command value and a rotation angle of the synchronous motor;
a first conversion step of converting a three-phase current value supplied to the synchronous motor into a two-phase current value based on a rotation angle of the synchronous motor;
a derivation step of deriving a first two-phase voltage command value by using a difference between a two-phase current command value obtained by superimposing the first two-phase current command value and the second two-phase current command value and the two-phase current value converted in the first conversion step;
a second conversion step of converting the first two-phase voltage command value into a three-phase voltage command value;
a supply step of supplying a three-phase voltage to the synchronous motor based on the three-phase voltage command value,
In the deriving step,
performing PI control using the difference to calculate a second two-phase voltage command value;
calculating a correction value for the second two-phase voltage command value based on the difference and a rotation angle of the synchronous motor;
The second two-phase voltage command value is corrected by the correction value to derive the first two-phase voltage command value.

Another aspect of the present invention has the following configuration: That is, an electric power steering device including a motor control device and a synchronous motor controlled by the motor control device,
The motor control device includes:
a first calculation means for calculating a first two-phase current command value corresponding to a torque command value of the synchronous motor based on an angular velocity of the synchronous motor;
a second calculation means for calculating a second two-phase current command value corresponding to a 6n-th order component in a dq rotating coordinate system based on the torque command value and a rotation angle of the synchronous motor;
a first conversion means for converting a three-phase current value supplied to the synchronous motor into a two-phase current value based on a rotation angle of the synchronous motor;
a derivation means for deriving a first two-phase voltage command value by using a difference between a two-phase current command value obtained by superimposing the first two-phase current command value and the second two-phase current command value and the two-phase current value converted by the first conversion means;
a second conversion means for converting the first two-phase voltage command value into a three-phase voltage command value;
a supply means for supplying a three-phase voltage to the synchronous motor based on the three-phase voltage command value,
The derivation means is
performing PI control using the difference to calculate a second two-phase voltage command value;
calculating a correction value for the second two-phase voltage command value based on the difference and a rotation angle of the synchronous motor;
The second two-phase voltage command value is corrected by the correction value to derive the first two-phase voltage command value.

The present invention makes it possible to appropriately reduce torque ripple generated in AC motors while reducing the number of functional parts in the device compared to conventional methods.

1 is a configuration diagram showing an example of a schematic configuration of an electric power steering device according to an embodiment of the present invention; FIG. 2 is a diagram showing an example of the functional configuration of a control system according to an embodiment of the present invention. FIG. 4 is a diagram for explaining an example of a transfer function of a feedback equation according to an embodiment of the present invention. FIG. 13 is a diagram for explaining another example of the transfer function of the feedback equation according to the embodiment of the present invention. 4 is a flowchart showing a control process according to an embodiment of the present invention. FIG. 13 is a diagram for explaining another example of the transfer function of the feedback equation according to the embodiment of the present invention.

The following describes the embodiment of the present invention with reference to the drawings. Note that the embodiment described below is one embodiment for explaining the present invention, and is not intended to be interpreted as limiting the present invention, and not all of the configurations described in each embodiment are necessarily essential configurations for solving the problems of the present invention. In addition, in each drawing, the same components are given the same reference numbers to indicate their correspondence.

First Embodiment
A first embodiment of the present invention will be described below. Note that the configuration of the electric power steering device shown below is one example, and the present invention is applicable not only to electric power steering devices but also to electric motors in general, including AC motors.

[Configuration Overview]
An example of the configuration of an electric power steering device according to this embodiment is shown in Fig. 1. A steering wheel 1 is a steerable wheel for a driver to steer. A steering shaft 2 of the steering wheel 1 is connected to steered wheels 8L, 8R via a reduction gear (worm gear) 3 constituting a reduction mechanism, universal joints 4a, 4b, a pinion rack mechanism 5, and tie rods 6a, 6b, and further via hub units 7a, 7b.

The steering shaft 2 is formed by connecting an input shaft on the steering wheel 1 side to an output shaft on the pinion rack mechanism 5 side via a torsion bar 9. The pinion rack mechanism 5 has a pinion 5a connected to a pinion shaft (not shown) to which steering force is transmitted from the universal joint 4b, and a rack 5b that meshes with the pinion 5a. The rotational motion transmitted to the pinion 5a is converted into linear motion in the vehicle width direction by the rack 5b.

The steering shaft 2 is provided with a torque sensor 10 for detecting a steering torque _Tdct applied to the torsion bar 9. The steering shaft 2 is also provided with a steering angle sensor 14 for detecting a steering angle _θh indicating a rotation angle of the steering shaft 2 about the axis on the steering wheel 1 side (input shaft side). The steering shaft 2 is also provided with an output shaft angle sensor 15 for detecting an output shaft angle _θc indicating a rotation angle of the steering shaft 2 about the axis on the pinion rack mechanism 5 side (output shaft side). In other words, the steering angle sensor 14 detects the rotation angle of the input shaft side relative to the torsion bar 9 as the steering angle _θh , and the output shaft angle sensor 15 detects the rotation angle of the output shaft side relative to the torsion bar 9 as the output shaft angle _θc . The torque sensor 10 detects the steering torque _Tdct based on the twist of the torsion bar 9 caused by the difference between the steering angle _θh and the output shaft angle _θc .

The steering angle sensor 14 and the output shaft angle sensor 15 may be integrated into one sensor. In FIG. 1, the steering shaft 2 and the torque sensor 10 are shown separately for ease of explanation, but they may be integrated into one sensor. The configuration of the torque sensor 10 is not particularly limited, and for example, a sleeve type or a ring type that detects torque from the torsion of the torsion bar 9 may be used. In the above configuration, the steering torque T _dct is detected based on the torsion of the torsion bar 9 caused by the difference between the steering angle θ _h and the output shaft angle θ _c , but the present invention is not limited to this. For example, the torque value may be detected using the difference between the angle signal on the steering wheel 1 side of the torsion bar 9 and the angle signal on the pinion rack mechanism 5 side. In the following description, the steering wheel 1 side of the steering shaft 2 is also referred to as the upstream side, and the pinion rack mechanism 5 side is also referred to as the downstream side.

The steering torque _Tdct detected by the torque sensor 10 includes a driver torque based on the operation of the steering wheel 1 by the driver, as well as a torque generated by an input (disturbance, etc.) from the downstream side. A command value based on the steering torque _Tdct is corrected so as to suppress vibration caused by the input from the downstream side. The suppression method here is not particularly limited, and any method may be used.

A steering assist motor 20 that assists the steering force applied to the steering wheel 1 is connected to the steering shaft 2 via a reduction gear 3. An EPS-ECU (Electronic Control Unit) 30, which is a controller that controls the electric power steering (EPS) device, is supplied with power from a battery 13 and receives an ignition key signal via an ignition (IGN) key 11.

The steering assist motor 20 (hereinafter also simply referred to as motor 20) according to this embodiment is, for example, a three-phase AC motor, and has a permanent magnet field or a wound field. The motor 20 rotates when three-phase AC currents, the phases of which differ by 120°, are supplied to coils (not shown) of the u-, v-, and w-phases. A rotation angle sensor 219, which is composed of a resolver, a rotary encoder, or the like, is installed on the shaft of the rotor (not shown) of the motor 20. The rotation angle θ of the motor 20 detected by the rotation angle sensor 219 is output to the EPS-ECU 30. The use of the rotation angle θ detected here will be described later.

The EPS-ECU 30 calculates a voltage command value as an assist command value based on the steering torque T _dct detected by the torque sensor 10 and the vehicle speed _Vh detected by the vehicle speed sensor 12. Furthermore, the EPS-ECU 30 controls the power (voltage value V _ref ) supplied to the steering assist motor 20 in accordance with the voltage command value based on the steering torque T _dct and the voltage command value based on the driving assist function. The steering assist motor 20 operates the reduction gear 3 based on the voltage value V _ref input from the EPS-ECU 30, and performs assist control for the steering wheel 1.

The EPS-ECU 30 may include, for example, a computer including a processor and peripheral components such as a storage device. The processor may be, for example, a CPU (Central Processing Unit) or an MPU (Micro-Processing Unit). The storage device may include any of a semiconductor storage device, a magnetic storage device, and an optical storage device. The storage device may include memories such as a register, a cache memory, and a ROM (Read Only Memory) and a RAM (Random Access Memory) used as a main storage device. The functions of the EPS-ECU 30 described below are realized, for example, by the processor of the EPS-ECU 30 executing a computer program stored in the storage device.

The EPS-ECU 30 may be formed of dedicated hardware for executing each of the information processes described below. For example, the EPS-ECU 30 may include a functional logic circuit configured in a general-purpose semiconductor integrated circuit. For example, the EPS-ECU 30 may include a programmable logic device (PLD) such as a field programmable gate array (FPGA).

[Functional configuration of vector control]
The motor current flowing through an AC motor contains harmonic components in addition to fundamental components. Torque ripple caused by these harmonic components can be a cause of vibration and noise. In the case of a three-phase AC motor, the main component of the torque ripple is the 6nth order component in electrical angle, and it is known that the harmonic component of the winding current that causes this is the (6n±1)th order component. In a dq rotating coordinate system synchronized with the angular velocity ω, which is the fundamental component of the motor, this (6n±1)th order component is observed as a 6nth order component (6nω).

Therefore, appropriately controlling the 6nth order component (6nω) in the dq rotating coordinate system is equivalent to appropriately controlling the (6n±1)th order component of the winding current, which makes it possible to reduce torque ripple. In this embodiment, motor control is performed with the above in mind.

Figure 2 is a diagram showing an example of the functional configuration of a control system for vector control of the motor 20 according to this embodiment. The control system shown in Figure 2 is configured using two-phase feedback vector control. This control system is configured, for example, by the EPS-ECU 30.

In the control system, first, the torque command value T _rq * is input to the current command value calculation unit 201 and the n-th current command value calculation unit 202. The torque command value T _rq * may be input from, for example, another control unit (not shown) included in the EPS-ECU 30. In the following description, the symbol "*" indicates a command value. The current command value calculation unit 201 calculates current command values i d0 * and i _q0 * of the zeroth component on the d-axis and q-axis, respectively, based on the angular velocity ω of the motor 20 input from the angular velocity calculation unit 221 so that the torque generated by the motor 20 coincides with the torque command value _{T rq} *. The current command values i _{d0 and} i _q0 are calculated, for example, by multiplying the torque command value T _rq * by a predetermined proportionality constant.

The d-axis and q-axis are axes in a dq rotating coordinate system. The d-axis indicates the direction of magnetic flux of a rotor (not shown) included in the motor 20. The q-axis indicates a direction perpendicular to the d-axis. The current command value i _d0 * of the d-axis component calculated by the current command value calculation unit 201 is output to the adder 203. The current command value i _q0 * of the q-axis component calculated by the current command value calculation unit 201 is output to the adder 205.

The n-th current command value calculation unit 202 calculates current command values i dn *, i qn * of the 6nth order component (6nω) on the d-axis and q-axis, respectively, based on the rotation angle θ and the torque command value T _rq * input from the rotation angle sensor _{219. The current command values i dn *, i qn * may} be calculated using, for example, a table in which the rotation angle θ and the torque command value T _rq * are input and the current command values i _dn *, i _qn * are output, or a calculation formula that is specified in advance. The d-axis current command value i _dn * calculated by the n-th current command value calculation unit 202 is output to the adder 203. The q-axis current command value i _qn * calculated by the n-th current command value calculation unit 202 is output to the adder 205.

The adder 203 adds the current command value i _d0 * from the current command value calculation unit 201 and the current command value i _dn * from the nth current command value calculation unit 202 to obtain a current command value i _d *, and outputs it to the subtractor 204. The adder 205 adds the current command value i _q0 * from the current command value calculation unit 201 and the current command value i _qn * from the nth current command value calculation unit 202 to obtain a current command value i _q *, and outputs it to the subtractor 204.

The subtractor 204 subtracts the current value i _d output from the three-phase/two-phase conversion unit 220 from the current command value i _d *, and outputs the result as a deviation Δi _d * to the PI control unit 207 and the correction value calculation unit 208. The subtractor 206 subtracts the current value i _q output from the three-phase/two-phase conversion unit 220 from the current command value i _q *, and outputs the result as a deviation Δi _q * to the PI control unit 210 and the correction value calculation unit 211.

The PI control unit 207 calculates a voltage command value by using a transfer function shown in the following equation (1) for the deviation Δi _d * input from the subtractor 204. Then, the PI control unit 207 outputs the calculated voltage command value to the adder 209.

The correction value calculation unit 208 calculates a correction value based on the internal model principle in order to make the actual current value i _d , which is a 6n-th order component, follow the current command value without delay. The correction value calculation unit 208 calculates a correction value for the voltage command value based on the deviation Δi _d * input from the subtractor 204 and the angular velocity ω of the motor 20 input from the angular velocity calculation unit 221, and outputs the correction value to the adder 209. In this embodiment, the correction value calculation unit 208 calculates the correction value using the transfer function shown in the following formula (2). In formula (2), s ² + (6nω) ² is included in the denominator polynomial of the transfer function in order to correspond to the 6n-th order component. In addition, the transfer function shown in formula (2) is derived from a cosine function.

The adder 209 adds the voltage command value from the PI control unit 207 and the correction value from the correction value calculation unit 208 , and outputs the result to the two-phase/three-phase conversion unit 213 as a voltage command value v _d * for the d-axis.

The PI control unit 210 calculates a voltage command value for the deviation Δi _q * input from the subtractor 206, using the transfer function shown in the above equation (1), similarly to the PI control unit 207. Then, the PI control unit 210 outputs the calculated voltage command value to the adder 212.

The correction value calculation unit 211 calculates a correction value based on the internal model principle in order to make the actual current value _iq , which is a 6n-th order component, follow the current command value. The correction value calculation unit 211 calculates a correction value for the voltage command value based on the deviation _Δiq * input from the subtractor 206 and the angular velocity ω of the motor 20 input from the angular velocity calculation unit 221, and outputs the correction value to the adder 212. In this embodiment, like the correction value calculation unit 208, the correction value calculation unit 211 calculates the correction value using the transfer function shown in the above formula (2).

The adder 212 adds the voltage command value from the PI control unit 210 and the correction value from the correction value calculation unit 211 , and outputs the result to the two-phase/three-phase conversion unit 213 as a voltage command value v _q * for the q axis.

The two-phase/three-phase converter 213 receives the two-phase voltage command values (the voltage command value _vd * from the adder 209 and the voltage command value _vq * from the adder 212) and the rotation angle θ of the motor 20 from the rotation angle sensor 219, and performs coordinate conversion of the two-phase voltage command values into three-phase voltage command values _vu *, _vv *, _vw * corresponding to the motor 20 so as to be synchronized with the rotation angle θ. The two-phase/three-phase converter 213 then outputs the three-phase voltage command values _vu *, _vv *, _vw * to the PWM controller 214.

The PWM control unit 214 derives the duty ratio of each phase based on the three-phase voltage command values v _u *, v _v *, and v _w * from the two-phase/three-phase conversion unit 213 , and outputs it to the inverter 215 .

Based on an input from the PWM control unit 214, the inverter 215 converts the power of the AC power supply supplied by the battery 13 into three-phase voltages _vu , _vv , and _vw , and supplies the voltage to the motor 20. Note that in Fig. 1, a voltage value _Vref is shown as the voltage output from the EPS-ECU 30 to the motor 20, which corresponds to the three-phase voltages _vu , _vv , and _vw .

Current detection circuits 216, 217, 218 for detecting current values are provided on the connection line between the inverter 215 and the motor 20, and the values of three-phase currents _iu , _iv , _iw are detected. The current detection circuits 216, 217, 218 output the detected current values to the three-phase/two-phase conversion unit 220. The current detection circuits 216, 217, 218 may be provided inside the inverter 215.

The rotation angle sensor 219 detects the rotation angle θ of the motor 20 and outputs it to the 2-phase/3-phase conversion unit 213, the 3-phase/2-phase conversion unit 220, the angular velocity calculation unit 221, and the n-th current command value calculation unit 202.

The three-phase/two-phase conversion unit 220 receives the three-phase current values (values of three-phase currents _iu , _iv , _iw supplied from the inverter 215 to the motor 20) and the rotation angle θ of the motor 20 from the rotation angle sensor 219, and performs coordinate conversion of the three-phase current values into two-phase actual current values _id , _iq corresponding to the d-axis and q-axis, respectively, of a dq rotating coordinate system. The three-phase/two-phase conversion unit 220 then outputs the actual current value _id to the subtractor 204, and outputs the actual current value _iq to the subtractor 206.

The angular velocity calculation unit 221 differentiates the rotation angle θ from the rotation angle sensor 219 to calculate the angular velocity ω of the motor 20, and outputs it to the current command value calculation unit 201, the correction value calculation unit 208, and the correction value calculation unit 211.

3 and 4 are diagrams for explaining the feedback control system around the PI control unit 207 and the PI control unit 210 shown in FIG. 2, and the motor 20. In FIG. 3 and FIG. 4, the input is U(s) and the output is Y(s). The model of the PI control unit 207 and the correction value calculation unit 208 (and the PI control unit 210 and the correction value calculation unit 211) is _Gc (s). The motor model around the motor 20 is shown as the controlled object _Gp (s). The transfer function of the controlled object _Gp (s) uses the following equation (3).

Figures 3 and 4 show the cases where n = 1 and 2. In this embodiment, the above formula (2) used in the correction value calculation unit 208 and the correction value calculation unit 211 is a transfer function derived from a cos function. This transfer function may be connected in parallel as shown in Figure 3. Alternatively, the transfer function may be connected in series as shown in Figure 4.

[Processing flow]
5 is a flowchart of a control process according to this embodiment, which is executed by the EPS-ECU 30. In order to simplify the description, the EPS-ECU 30 will be collectively referred to as the processing entity.

In S501, the EPS-ECU 30 acquires a torque command value T _rq *. The torque command value T _rq * may be acquired from another portion (not shown) included in the EPS-ECU 30, or may be acquired from an ADAS-ECU (not shown) or the like included in the vehicle (not shown).

In S502, the EPS-ECU 30 calculates zero-order current command values i _d0 *, i _q0 * in the dq rotating coordinate system based on the angular velocity ω of the motor 20 from the torque command value T _rq * acquired in S501.

In S503, the EPS-ECU 30 calculates n-th order current command values i _dn *, i _qn * for reducing the torque ripple based on the rotation angle θ of the motor 20 from the torque command value T _rq * acquired in S501.

In S504, the EPS-ECU 30 superimposes the zeroth-order current command value i _d0 * on the d-axis calculated in S502 and the nth-order current command value i _dn * on the d-axis calculated in S503 to calculate a current command value I _d *. Similarly, the EPS-ECU 30 superimposes the zeroth-order current command value i _q0 * on the q-axis calculated in S502 and the nth-order current command value i _qn * on the q-axis calculated in S503 to calculate a current command value i _q *.

In S505, the EPS-ECU 30 calculates deviations Δi _d *, Δi _q * between the current command values i _d *, i _q * calculated in S505 and the actual current values i _d , i _q of the motor 20.

In S506, the EPS-ECU 30 calculates a voltage command value by PI control using the deviations Δi _d * and Δi _q * of the current values of the d-axis component and the q-axis component, respectively, calculated in S505. In the PI control here, the transfer function shown in the above-mentioned formula (1) may be used.

In S507, the EPS-ECU 30 calculates a correction value for the voltage command value using the deviations Δi _d * and Δi _q * of the current values of the d-axis component and q-axis component, respectively, calculated in S505, and the angular velocity ω of the motor 20. Here, the transfer function shown in the above-mentioned equation (2) may be used.

In S508, the EPS-ECU 30 calculates a voltage command value v _d * for the d-axis component by correcting the voltage command value for the d-axis component calculated in S506 with the correction value for the d-axis component calculated in S507. Similarly, the EPS-ECU 30 calculates a voltage command value v q * for the q-axis component by correcting the voltage command value for the q-axis component calculated in S506 with the correction value for the _q -axis component calculated in S507.

In S509, the EPS-ECU 30 converts the voltage command values v _d *, v _q * calculated in S508 into three-phase voltage command values v _u *, v _v *, v _w * for controlling the motor 20 based on the rotation angle θ of the motor 20.

In S510, the EPS-ECU 30 uses the voltage command values _vu *, _vv *, _vw * obtained in S509 to derive the duty of each phase by the PWM control unit 214, and controls the three-phase voltages _vu , _vv , _vw based on this to control the motor 20. Then, this process flow ends. This process flow is repeatedly executed while the motor 20 is being controlled.

As described above, this embodiment makes it possible to appropriately reduce the torque ripple generated in the electric motor while reducing the number of functional parts in the device. In particular, since the number of functional parts related to feedback can be reduced compared to conventional configurations, it is possible to improve the responsiveness of the entire device.

[Modification]
In the above example, the formula (2) is used as the transfer function used in the correction value calculation unit 208 and the correction value calculation unit 211. However, this is not limited to this, and other transfer functions may be used.

The transfer function shown in the above formula (2) is a transfer function derived from a cosine function. Alternatively, a transfer function shown in the following formula (4) derived from a sine function may be used. As in formula (2), formula (4) also includes ^s2 + (6nω) ² in the denominator polynomial of the transfer function to accommodate the 6nth order component.

Figure 6 is a diagram for explaining the control system around the PI control unit 207, the PI control unit 210, and the motor 20 shown in Figure 2. The difference from Figure 3 is that the correction value calculation unit 208 and the correction value calculation unit 211 use the above formula (4). Here, an example configuration in which transfer functions are connected in parallel is shown.

<Other embodiments>
Although an electric power steering device has been described as an example of an application of a synchronous motor using the control according to the present invention, the configuration of the electric power steering device is not limited to the configuration shown in Fig. 1. For example, the electric power steering device may be configured with a Steer-By-Wire (SBW) mechanism in which the steering wheel 1 side and the pinion rack mechanism 5 side are mechanically separated.

The present invention can also be realized by providing a program or application for implementing the functions of one or more of the above-described embodiments to a system or device via a network or storage medium, and having one or more processors in the computer of the system or device read and execute the program.

As such, the present invention is not limited to the above-described embodiment, and the invention also contemplates the mutual combination of the various components of the embodiment, as well as modifications and applications by those skilled in the art based on the description in the specification and well-known technology, and these are included in the scope of the protection sought.

As described above, the present specification discloses the following:
(1) A first calculation means (e.g., 201) for calculating first two-phase current command values (e.g., i _d0 *, i _q0 *) corresponding to a torque command value (e.g., T _rq *) of a synchronous motor (e.g., 20) based on an angular velocity (e.g., ω) of the synchronous motor;
a second calculation means (e.g., 202) for calculating second two-phase current command values (e.g., i _dn *, i _qn *) corresponding to 6n-th order components in a dq rotating coordinate system based on the torque command value and a rotation angle of the synchronous motor;
a first conversion means (e.g., 220) for converting three-phase current values (e.g., _iu , _iv , _iw ) supplied to the synchronous motor into two-phase current values (e.g., _id , _iq ) based on a rotation angle of the synchronous motor;
a derivation means (e.g., 207 to 211) for deriving a first two-phase voltage command value (e.g., v _d *, v _q *) using a difference (e.g., Δi d *, Δi q * ₎ between a two-phase current command value (e.g., i _d *, i _q *) obtained by superimposing the first two-phase current command value and the second two-phase current command value and the two-phase current value converted by the first conversion means _;
a second conversion means (e.g., 213) for converting the first two-phase voltage command values into three-phase voltage command values (e.g., _vu *, _vv *, _vw *);
a supply means (e.g., 214, 215) for supplying a three-phase voltage to the synchronous motor based on the three-phase voltage command value;
The derivation means is
PI control is performed using the difference to calculate a second two-phase voltage command value (e.g., 207, 210);
Calculating a correction value for the second two-phase voltage command value based on the difference and the rotation angle of the synchronous motor (e.g., 208, 211);
The motor control device according to claim 1, wherein the first two-phase voltage command value is derived by correcting the second two-phase voltage command value with the correction value (e.g., 209, 212).
According to this configuration, it is possible to appropriately reduce the torque ripple generated in the AC motor while suppressing the number of functional parts in the device.

(2) The motor control device according to (1), wherein the derivation means calculates the correction value using a transfer function based on a sine function.
According to this configuration, the sixth-order component of the actual two-phase current of the synchronous motor is reflected in the current command value without delay using a sine function, thereby making it possible to appropriately reduce the torque ripple.

(3) The motor control device according to (1), wherein the derivation means calculates the correction value using a transfer function based on a cosine function.
According to this configuration, the sixth-order component of the actual two-phase current of the synchronous motor is reflected in the current command value without delay using a cosine function, thereby making it possible to appropriately reduce the torque ripple.

(4) The motor control device according to (2) or (3), wherein the transfer function includes {s ² +(6nω) ² } in a denominator polynomial.
According to this configuration, the sixth-order component of the actual two-phase current of the synchronous motor can be focused on and reflected in the current command value.

(5) The motor control device according to any one of (1) to (4), wherein the correction value corresponds to a 6n-th order component of the two-phase current value converted by the first conversion means.
According to this configuration, it is possible to derive a current command value by focusing on the sixth-order component of the actual two-phase current of the synchronous motor, which is a cause of torque ripple.

(6) a first calculation step of calculating first two-phase current command values corresponding to a torque command value of the synchronous motor based on an angular velocity of the synchronous motor;
a second calculation step of calculating a second two-phase current command value corresponding to a 6n-th order component in a dq rotating coordinate system based on the torque command value and a rotation angle of the synchronous motor;
a first conversion step of converting a three-phase current value supplied to the synchronous motor into a two-phase current value based on a rotation angle of the synchronous motor;
a derivation step of deriving a first two-phase voltage command value by using a difference between a two-phase current command value obtained by superimposing the first two-phase current command value and the second two-phase current command value and the two-phase current value converted in the first conversion step;
a second conversion step of converting the two-phase voltage command values derived in the derivation step into three-phase voltage command values;
a supply step of supplying a three-phase voltage to the synchronous motor based on the three-phase voltage command value,
In the deriving step,
performing PI control using the difference to calculate a second two-phase voltage command value;
calculating a correction value for the second two-phase voltage command value based on the difference and a rotation angle of the synchronous motor;
a correction value for correcting the second two-phase voltage command value, thereby deriving the first two-phase voltage command value.
According to this configuration, it is possible to appropriately reduce the torque ripple generated in the AC motor while suppressing the number of functional parts in the device.

(7) A motor control device according to any one of (1) to (5),
and a synchronous motor controlled by the motor control device.
According to this configuration, it is possible to appropriately reduce torque ripple generated in an AC motor mounted in an electric power steering device while suppressing the number of functional units in the motor control device.

REFERENCE SIGNS LIST 1 Steering wheel 2 Steering shaft 3 Reduction gears 4a, 4b Universal joint 5 Pinion rack mechanism 6a, 6b Tie rods 7a, 7b Hub units 8L, 8R Steering wheels 9 Torsion bar 10 Torque sensor 11 Ignition (ING) key 12 Vehicle speed sensor 13 Battery 14 Steering angle sensor 20 Steering assist motor (motor)
30 EPS (Electric Power Steering) - ECU (Electronic Control Unit))
201 Current command value calculation unit 202 n-th order current command value calculation unit 203, 205, 209, 212 Adder 204, 206 Subtractor 207, 210 PI control unit 208, 211 Correction value calculation unit 213 2-phase/3-phase conversion unit 214 PWM control unit 215 Inverter 216, 217, 218 Current detection circuit 219 Rotation angle sensor 220 3-phase/2-phase conversion unit 221 Angular velocity calculation unit

Claims (7)

a first calculation means for calculating a first two-phase current command value corresponding to a torque command value of the synchronous motor based on an angular velocity of the synchronous motor;
a second calculation means for calculating a second two-phase current command value corresponding to a 6n-th order component in a dq rotating coordinate system based on the torque command value and a rotation angle of the synchronous motor;
a first conversion means for converting a three-phase current value supplied to the synchronous motor into a two-phase current value based on a rotation angle of the synchronous motor;
a derivation means for deriving a first two-phase voltage command value by using a difference between a two-phase current command value obtained by superimposing the first two-phase current command value and the second two-phase current command value and the two-phase current value converted by the first conversion means;
a second conversion means for converting the first two-phase voltage command value into a three-phase voltage command value;
a supply means for supplying a three-phase voltage to the synchronous motor based on the three-phase voltage command value,
The derivation means is
performing PI control using the difference to calculate a second two-phase voltage command value;
calculating a correction value for the second two-phase voltage command value based on the difference and the angular velocity of the synchronous motor;
a correction value for correcting the second two-phase voltage command value, thereby deriving the first two-phase voltage command value. The motor control device according to claim 1, characterized in that the derivation means calculates the correction value using a transfer function based on a sine function. The motor control device according to claim 1, characterized in that the derivation means calculates the correction value using a transfer function based on a cosine function. 4. The motor control device according to claim 2, wherein the transfer function includes {s ² +(6nω) ² } in a denominator polynomial. The motor control device according to any one of claims 1 to 4, characterized in that the correction value corresponds to the 6nth order component of the two-phase current value converted by the first conversion means. a first calculation step of calculating a first two-phase current command value corresponding to a torque command value of the synchronous motor based on an angular velocity of the synchronous motor;
a second calculation step of calculating a second two-phase current command value corresponding to a 6n-th order component in a dq rotating coordinate system based on the torque command value and a rotation angle of the synchronous motor;
a first conversion step of converting a three-phase current value supplied to the synchronous motor into a two-phase current value based on a rotation angle of the synchronous motor;
a derivation step of deriving a first two-phase voltage command value by using a difference between a two-phase current command value obtained by superimposing the first two-phase current command value and the second two-phase current command value and the two-phase current value converted in the first conversion step;
a second conversion step of converting the first two-phase voltage command value into a three-phase voltage command value;
a supply step of supplying a three-phase voltage to the synchronous motor based on the three-phase voltage command value,
In the deriving step,
performing PI control using the difference to calculate a second two-phase voltage command value;
calculating a correction value for the second two-phase voltage command value based on the difference and the angular velocity of the synchronous motor;
a correction value for correcting the second two-phase voltage command value, thereby deriving the first two-phase voltage command value. A motor control device according to any one of claims 1 to 5,
and a synchronous motor controlled by the motor control device.

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