JP7697767B2 - Bidirectional current resonant DC/DC converter and bidirectional power supply device - Google Patents

Description

The present invention relates to a bidirectional current resonant DC/DC converter and a bidirectional power supply device equipped with the bidirectional current resonant DC/DC converter.

In recent years, CLLC-type (also called CLLLC-type) bidirectional current-resonant DC/DC converters, which are an LLC-type current-resonant DC/DC converter made bidirectional, have been attracting attention as an isolated bidirectional DC/DC converter.

LLC type current resonant DC/DC converters have the advantages of being small, highly efficient, with a small number of parts and low cost, but they also have the disadvantages that the output power varies depending on the input voltage and output voltage, making them difficult to control, the controllable voltage range is narrow, and the output power cannot be reduced to zero by frequency control alone.

CLLC-type bidirectional current resonant DC/DC converters have the same drawbacks as LLC-type converters, in that frequency control alone cannot reduce the output power to zero, causing a power jump (instantaneous power fluctuation) when switching the direction of power transmission. For this reason, CLLC-type bidirectional current resonant DC/DC converters have the problem of being unable to seamlessly switch the direction of power transmission.

In the case of conventional voltage-current type DC/DC converters, when the voltage applied to the output side becomes higher than the output voltage value being controlled, the current flow naturally reverses and the direction of power transmission is reversed, so there is no need to change the control method and the direction of power transmission can be switched seamlessly. However, in the case of the LLC or CLLC method, the direction of control and the direction of current flow (direction of power transmission) are the same, so in order to seamlessly switch the direction of power transmission, in addition to the problem of power jumps when switching, the control method must also be changed, making it difficult to switch the direction of power transmission seamlessly.

Figure 10 is a diagram showing the change in drive frequency and output power when the power transmission direction of the conventional CLLC bidirectional current resonant DC/DC converter is switched. The third quadrant of Figure 10 is the amount of power transmission from the primary side to the secondary side, with the horizontal axis Fd being the drive frequency and the vertical axis Pd being the forward output power, while the first quadrant is the amount of power transmission from the secondary side to the primary side, with the horizontal axis Fc being the drive frequency and the vertical axis Pc being the reverse output power. In the case of the third quadrant, the forward output power Pd decreases as the drive frequency Fd increases, but the rate of decrease in the output power Pd decreases as the drive frequency Fd increases. Similarly, in the case of the first quadrant, the reverse output power Pc decreases as the drive frequency Fc increases, but the rate of decrease in the output power Pc decreases as the drive frequency Fc increases.

For example, when switching direction from drive point Ac to drive point Ad, the drive frequency of one drive circuit is increased while decreasing the output to move to point Bc, which is the maximum frequency, and then the drive circuit on one side is stopped and at the same time the drive circuit on the other side starts driving at point Bd, which is the maximum frequency, and the drive frequency is decreased to move to point Ad. In this case, as the drive circuit is switched, there is an instantaneous transition from point Bc to point Bd, which causes a power jump. The same is true for switching from drive point Ad to drive point Ac.

In response to this problem, Patent Document 1 proposes a control method that switches between pulse width modulation control or phase modulation control with a fixed frequency at the resonant frequency, and frequency modulation control below the resonant frequency. However, with this control method, performing pulse width modulation control at the resonant frequency results in hard switching, which creates the problem of switching noise.

Patent Document 2 proposes a method for seamlessly switching the direction of power transmission in a CLLC-type bidirectional current resonant DC/DC converter, in which the driving frequency on the primary (power supply) side is maximized (step 3), the driving of the secondary side is started from a delay of the maximum phase difference with respect to the primary side (step 4), the phase difference is reduced by phase shift control (step 5), and when the phase difference becomes a minimum, the direction of power transmission is switched (step 6).

However, the general method of driving the CLLC system involves bridge driving the primary (power supply) side with a duty of 50% and diode or synchronous rectification of the secondary side, and is not a method of driving with a phase difference between the primary and secondary sides. For this reason, the method described in Patent Document 2 cannot directly apply, develop, or reuse commonly known design methods for the LLC system or the CLLC system, or existing hardware and software assets and knowledge. Furthermore, Patent Document 2 does not disclose a specific method of driving the secondary bridge circuit, or the maximum and minimum phase differences between the drive signals on the primary and secondary sides, which are necessary to realize seamless switching of the power transmission direction.

Non-Patent Document 1 proposes a method for seamlessly switching the power transmission direction in a CLLC-type bidirectional current resonant DC/DC converter by maximizing the drive frequency of the primary (power supply side) bridge circuit and decreasing the phase delay of the secondary side relative to the primary side from +0.05, passing through 0, and then increasing it to -0.05, thereby switching to the opposite direction.

The method described in Non-Patent Document 1, like the method described in Patent Document 2, cannot utilize the design methods of the general LLC or CLLC methods, and in addition, the phase difference between the primary and secondary sides of ±0.05 is very narrow, being only half the range of the dead time (less than about 0.5 μsec if the drive frequency is about 100 kHz), making it difficult to control. Specifically, a highly accurate control instruction means is required, and since the primary and secondary sides are insulated, it is necessary to detect the phase taking into account the detection delay, and it is also difficult to detect the phase difference.

In addition, the methods of Patent Document 2 and Non-Patent Document 1 provide a phase difference between the primary bridge circuit and the secondary bridge circuit, but since the phase of the secondary side also changes depending on the input voltage and the load to which it is connected, it cannot be easily controlled even if the phase difference with respect to the primary drive signal is assumed. In addition, since the primary and secondary sides are insulated, insulation detection is required, which leads to an increase in the detection circuit, high costs, and detection delays. In addition, although it depends on the input voltage, when the drive frequency of the primary side is maximized, the current value of the secondary side is minimized, so there is a problem that it is difficult to accurately detect the phase difference even if the current on the secondary side is detected.

JP 2014-79145 A U.S. Pat. No. 1,058,7201

Soichiro Kawano and five others, "CLLC-type resonant converter capable of bidirectional power transmission", [online], IEICE Technical Report EE2021-5, [Retrieved February 22, 2022], Internet <URL: https://www.ieice.org/ken/paper/20210527MCEF/>

The present invention has been made in consideration of the above circumstances, and its objective is to provide a bidirectional current resonant DC/DC converter and a bidirectional power supply device that can seamlessly switch the power transmission direction without power jumps when switching, with relatively simple control.

In order to solve the above problems, the bidirectional current resonant DC/DC converter according to the present invention comprises:
A bidirectional current resonant DC/DC converter including a power supply unit and a control unit,
The power supply unit includes:
a transformer including a primary coil and a secondary coil;
a first bridge circuit including a first leg and a second leg connected in parallel, each leg including an upper arm and a lower arm connected in series, each arm including a switching element, and connected to the primary coil;
a second bridge circuit including a third leg and a fourth leg connected in parallel, each leg including an upper arm and a lower arm connected in series, each arm including a switching element, and connected to the secondary coil;
a primary-side resonant circuit provided between the first bridge circuit and the primary-side coil, the primary-side resonant circuit including a first resonant coil and a first resonant capacitor;
a secondary resonant circuit provided between the second bridge circuit and the secondary coil, the secondary resonant circuit including a second resonant coil and a second resonant capacitor;
Equipped with
The control unit is
a forward control in which the first bridge circuit is used as a drive circuit, and synchronous rectification control or diode rectification control is performed on the second bridge circuit to transmit power from the first bridge circuit to the second bridge circuit;
a reverse control in which the second bridge circuit is used as a drive circuit, synchronous rectification control or diode rectification control is performed on the first bridge circuit, and power is transmitted from the second bridge circuit to the first bridge circuit;
and increasing the drive frequency of the drive circuit and then switching the direction of power transmission.
The switching control is characterized in that the phase shift control between each leg of the drive circuit is performed with the phase shift amount ranging from 0 to a maximum value, or PWM control is performed on each switching element of the drive circuit to change the on-ratio of the switching element in the range from 50% to a minimum value.

According to this configuration, by performing phase shift control between each leg of the drive circuit or PWM control that changes the on-ratio of each switching element, seamless switching of the power transmission direction at zero power, which cannot be achieved by frequency control alone, is possible. In other words, in this configuration, by performing phase shift control of the drive signal between each leg in the range from 0 to the maximum value, or by performing PWM control that changes the on-ratio of the switching element in the range from 50% to the minimum value, it is not necessary to use a signal between the primary side and secondary side for phase shift control, and it is also not necessary to control the phase difference between the first bridge circuit and the second bridge circuit. Therefore, according to this configuration, it is possible to use the design method of the conventional LLC method or CLLC method, and relatively simple control is possible (for example, control is possible over a range about 10 times wider than the dead time), and costs can be reduced.

In the bidirectional current resonant DC/DC converter,
The control unit is
In the forward control, the synchronous rectification control of the second bridge circuit is performed;
In the switching control during the forward control,
a first process of increasing a drive frequency of the first bridge circuit to a predetermined maximum frequency;
a second process of performing a phase shift control between the legs of the first bridge circuit and increasing a first phase shift amount of the phase shift control from 0, or performing a PWM control of each switching element of the first bridge circuit and decreasing a first on ratio of each switching element of the first bridge circuit from 50%;
a third process of stopping the synchronous rectification control of the second bridge circuit when an output current value of the second bridge circuit becomes equal to or lower than a predetermined threshold value;
a fourth process of setting the first phase shift amount to a maximum value or the first on ratio to a minimum value, and stopping the driving of the first bridge circuit when an output voltage value of the second bridge circuit reaches a predetermined upper limit value;
a fifth process of setting a drive frequency of the second bridge circuit to a predetermined maximum frequency and a second phase shift amount between each leg of the second bridge circuit to a maximum value to start phase shift control between each leg of the second bridge circuit, or setting a second on-ratio of each switching element of the second bridge circuit to a minimum value to start PWM control of each switching element of the second bridge circuit;
a sixth process of starting the synchronous rectification control of the first bridge circuit when an output current value of the first bridge circuit becomes equal to or greater than a predetermined threshold value;
and a seventh process of lowering the drive frequency of the second bridge circuit from the maximum frequency by setting the second phase shift amount to 0 or setting the second on ratio to 50%.

With this configuration, the power transmission direction is switched based on the output current value and output voltage value of the second bridge circuit, making it possible to seamlessly switch the power transmission direction without an external control signal.

In the bidirectional current resonant DC/DC converter,
The control unit is
At least the fifth process may be executed while performing constant voltage control on the second bridge circuit.

With this configuration, the power transmission direction is switched while controlling the second bridge circuit at a constant voltage, making it possible to switch only the current direction.

In the bidirectional current resonant DC/DC converter,
The control unit is
execute the third process before executing the second process;
The seventh process may be executed before the sixth process is executed.

In the bidirectional current resonant DC/DC converter,
The control unit is
In the forward control, the diode rectification control of the second bridge circuit is performed;
In the switching control during the forward control,
a first process of increasing a drive frequency of the first bridge circuit to a predetermined maximum frequency;
a second process of performing a phase shift control between the legs of the first bridge circuit and increasing a first phase shift amount of the phase shift control from 0, or performing a PWM control of each switching element of the first bridge circuit and decreasing a first on ratio of each switching element of the first bridge circuit from 50%;
a fourth process of setting the first phase shift amount to a maximum value or the first on ratio to a minimum value, and stopping the driving of the first bridge circuit when an output voltage value of the second bridge circuit reaches a predetermined upper limit value;
a fifth process of setting a drive frequency of the second bridge circuit to a predetermined maximum frequency and a second phase shift amount between each leg of the second bridge circuit to a maximum value to start phase shift control between each leg of the second bridge circuit, or setting a second on-ratio of each switching element of the second bridge circuit to a minimum value to start PWM control of each switching element of the second bridge circuit;
and a seventh process of reducing the drive frequency of the second bridge circuit from the maximum frequency by setting the second phase shift amount to 0 or setting the second on ratio to 50%.

In order to solve the above problems, a bidirectional power supply device according to the present invention comprises:
The bidirectional current resonant DC/DC converter;
a bidirectional DC/AC inverter that performs a DC/AC conversion operation and an AC/DC conversion operation;
A bidirectional power supply device comprising:
When the direction of power transmission is switched, the voltage at the connection point between the bidirectional current resonant DC/DC converter and the bidirectional DC/AC inverter becomes a constant voltage or a voltage determined by the input voltage of the bidirectional current resonant DC/DC converter.

With this configuration, the connection point voltage (link voltage) between the bidirectional current resonant DC/DC converter and the bidirectional DC/AC inverter is a constant voltage or a voltage determined by the power supply voltage on the bidirectional current resonant DC/DC converter side, so there is no need to charge or discharge a capacitor (e.g., an electrolytic capacitor) provided to stabilize the link voltage when switching the power transmission direction, making it possible to instantly switch the power transmission direction without unnecessary energy transmission.

In the bidirectional power supply device,
The bidirectional current resonant DC/DC converter is configured to be connectable to an electric vehicle,
The electric vehicle may be configured to perform charging and discharging operations.

The present invention provides a bidirectional current resonant DC/DC converter and a bidirectional power supply device that can seamlessly switch the power transmission direction without a power jump when switching, with relatively simple control.

1 is a diagram showing an example of a circuit configuration of a bidirectional current resonant DC/DC converter according to the present invention; 1 is a flow diagram showing an example of a first flow of control when switching the power transmission direction of the bidirectional current resonant DC/DC converter according to the present invention; FIG. 11 is a flow diagram showing an example of a second flow of control when switching the power transmission direction of the bidirectional current resonant DC/DC converter according to the present invention. FIG. 11 is an example of a flow diagram showing a third flow of control when switching the power transmission direction of the bidirectional current resonant DC/DC converter according to the present invention. 1 is a timing diagram showing an example of a change in phase difference when switching the power transmission direction of the bidirectional current resonant DC/DC converter according to the present invention; FIG. 1 is a system configuration diagram showing an example of an application of a bidirectional power supply device according to the present invention. FIG. 4 is a diagram showing a schematic diagram of changes in drive frequency and output power when the power transmission direction of the bidirectional power supply device according to the present invention is switched. 1 is a timing diagram showing an example of a change in duty of PWM control when switching the power transmission direction of the bidirectional current resonant DC/DC converter according to the present invention; FIG. 4 is a diagram showing a schematic diagram of a change in link voltage due to switching of the power transmission direction of the bidirectional power supply device according to the present invention. 1 is a diagram showing a schematic diagram of changes in drive frequency and output power when switching the power transmission direction of a conventional bidirectional current resonant DC/DC converter.

Below, an embodiment of a bidirectional current resonant DC/DC converter and a bidirectional power supply device according to the present invention will be described with reference to the attached drawings.

[Bidirectional current resonant DC/DC converter]
1 shows a bidirectional current resonant DC/DC converter 1 according to an embodiment of the present invention. The bidirectional current resonant DC/DC converter 1 includes a DC/DC power supply unit 10 (corresponding to the "power supply unit" of the present invention), a control unit 20, and terminals T1 to T4.

The DC/DC power supply unit 10 includes a transformer Tr, a first bridge circuit 11, a second bridge circuit 12, a primary resonant circuit 13, a secondary resonant circuit 14, and various detection circuits (not shown). The DC/DC power supply unit 10 has a symmetrical circuit configuration, so that power can be transmitted in both directions using LLC resonance. Therefore, the circuit configuration of the DC/DC power supply unit 10 is called the CLLC (or CLLLC including the excitation inductance Lm) system.

The transformer Tr is a high-frequency insulating transformer, and includes a primary coil and a secondary coil. The primary coil is connected to the first bridge circuit 11 via the primary resonant circuit 13. The secondary coil is connected to the second bridge circuit 12 via the secondary resonant circuit 14. The transformer Tr may be configured with one or more transformers depending on the capacity. In addition, the excitation inductance Lm of the transformer Tr is not shown in the figure as it is included in the transformer Tr.

The first bridge circuit 11 includes a first leg and a second leg connected in parallel, and each leg includes an upper arm and a lower arm connected in series. The upper arm of the first leg includes a switching element Q1, a diode D1, and a capacitor C1, the lower arm of the first leg includes a switching element Q2, a diode D2, and a capacitor C2, the upper arm of the second leg includes a switching element Q3, a diode D3, and a capacitor C3, and the lower arm of the second leg includes a switching element Q4, a diode D4, and a capacitor C4.

The second bridge circuit 12 includes a third leg and a fourth leg connected in parallel, and each leg includes an upper arm and a lower arm connected in series. The upper arm of the third leg includes a switching element Q5, a diode D5, and a capacitor C5, the lower arm of the third leg includes a switching element Q6, a diode D6, and a capacitor C6, the upper arm of the fourth leg includes a switching element Q7, a diode D7, and a capacitor C7, and the lower arm of the fourth leg includes a switching element Q8, a diode D8, and a capacitor C8.

The switching elements Q1 to Q8 are indicated by the circuit symbols for transistors, but any power semiconductor switching element capable of switching at high frequencies, such as an IGBT (insulated gate bipolar transistor), MOSFET (metal oxide semiconductor field effect transistor), SiC (silicon carbide)-MOSFET, or GaN (gallium nitride)-MOSFET, can be used.

Diodes D1 to D8 are connected in parallel in the reverse direction to the current paths of switching elements Q1 to Q8. Diodes D1 to D8 are described as external diodes independent of switching elements Q1 to Q8, but they may be parasitic diodes of switching elements Q1 to Q8, or both.

Capacitors C1 to C8 are capacitors for partial resonance and are connected in parallel to the current paths of switching elements Q1 to Q8 and diodes D1 to D8. Capacitors C1 to C8 may be parasitic capacitors of switching elements Q1 to Q8, or may be external capacitors independent of switching elements Q1 to Q8, or may be both.

The primary resonant circuit 13 includes a resonant coil Lr1, which corresponds to the "first resonant coil" of the present invention, and a resonant capacitor Cr1, which corresponds to the "first resonant capacitor" of the present invention, connected in series to the primary coil of the transformer Tr.

The secondary resonant circuit 14 includes a resonant coil Lr2, which corresponds to the "second resonant coil" of the present invention, and a resonant capacitor Cr2, which corresponds to the "second resonant capacitor" of the present invention, connected in series to the secondary coil of the transformer Tr.

The resonant coil Lr1 and the resonant capacitor Cr1 are placed on both ends of the primary coil, but they may be placed on one side of the primary coil. Alternatively, the constants of the resonant capacitor Cr1 and the resonant coil Lr1 may be divided and both may be placed on both ends of the primary coil. The same applies to the resonant coil Lr2 and the resonant capacitor Cr2. Furthermore, the resonant coils Lr1 and Lr2 may be partially or entirely composed of leakage flux from the transformer Tr, or may be composed of individual coils, or both.

The control unit 20 includes drive circuits for the switching elements Q1 to Q8 for turning the switching elements Q1 to Q8 on and off, and a control circuit for sending control signals to each drive circuit. The control unit 20 may be composed of analog circuits, digital circuits such as a microcontroller or DSP, or a combination of analog and digital circuits.

The control unit 20 monitors the primary side voltage V1, the secondary side voltage V2, and the primary and secondary side currents (not shown) based on the detection values of various detection circuits (not shown), and turns the switching elements Q1 to Q8 on and off.

The control unit 20 executes forward control to transmit power from the first bridge circuit 11 on the primary side to the second bridge circuit 12 on the secondary side, reverse control to transmit power from the second bridge circuit 12 on the secondary side to the first bridge circuit 11 on the primary side, and switching control to switch the direction of power transmission.

Specifically, during forward control, the control unit 20 performs frequency control for the first bridge circuit 11, which serves as the drive circuit, to drive the switching elements Q1 to Q4 within a specified frequency range, while performing synchronous rectification control for the second bridge circuit 12. During forward control and switching control, the control unit 20 performs phase shift control between the first leg (switching elements Q1, Q2) and the second leg (switching elements Q3, Q4) of the first bridge circuit 11.

During reverse control, the control unit 20 performs frequency control for the second bridge circuit 12, which serves as the drive circuit, to drive the switching elements Q5 to Q8 within a specified frequency range, while performing synchronous rectification control for the first bridge circuit 11. During reverse control and switching control, the control unit 20 performs phase shift control between the third leg (switching elements Q5, Q6) and the fourth leg (switching elements Q7, Q8) of the second bridge circuit 12.

Figures 2 to 4 show the flow of control (switching control) when switching the power transmission direction of the bidirectional current resonant DC/DC converter 1. Specifically, they show the flow of control when the output power drops during forward power transmission in the bidirectional current resonant DC/DC converter 1 under constant voltage control, the output voltage rises, and the power transmission direction is switched to the reverse direction, and the flow of control when the reverse output power drops after switching the power transmission direction to the reverse direction, and the power transmission direction is switched to the forward direction.

In FIG. 2, the flow of processing when the output drops during forward power transmission will be described. The bidirectional current resonant DC/DC converter 1 is assumed to transmit power in the direction from the first bridge circuit 11 to the second bridge circuit 12. In this case, the control unit 20 executes forward control, performs frequency control on the first bridge circuit 11 with the input voltage V1 and the phase difference between the first leg (switching elements Q1, Q2) and the second leg (switching elements Q3, Q4) at 0° (no phase difference), and performs synchronous rectification control and constant voltage control of voltage V2 on the second bridge circuit 12 (S101).

The control unit 20 increases the drive frequency F1 of the first bridge circuit 11 to a predetermined maximum frequency F1max in response to the decrease in output power due to the increase in voltage V2 (Yes in S102). Step S101 is executed until the drive frequency F1 reaches the maximum frequency F1max (No in S102). If the output power further decreases, the control unit 20 starts phase shift control and increases the phase difference between the first leg (switching elements Q1, Q2) and the second leg (switching elements Q3, Q4) of the first bridge circuit 11, i.e., the first phase shift amount θ1, from 0° (S103). Note that if the output power decreases due to the increase in voltage V2 in step S103, the first phase shift amount θ1 is increased, and if the output power increases due to the decrease in voltage V2, the first phase shift amount θ1 is decreased. That is, the control unit 20 increases the first phase shift amount θ1 when the output power is greater than the target power, and decreases the first phase shift amount θ1 when the output power is smaller than the target power. By controlling the phase shift in this way, it is possible to control without being affected by resonance due to the parasitic capacitance of the elements that make up the circuit.

If the output current (secondary side current) of the second bridge circuit 12 is greater than a predefined threshold (No in S104), the control unit 20 continues the synchronous rectification control of the second bridge circuit 12 (S105). If the first phase shift amount θ1 is not 0° (No in S106), the control unit 20 executes steps S103 and onward. If the first phase shift amount θ1 is 0° (Yes in S106), the control unit 20 executes steps S101 and onward. On the other hand, if the output current of the second bridge circuit 12 is equal to or less than the threshold (Yes in S104), the control unit 20 stops the synchronous rectification control of the second bridge circuit 12 (S107).

The output power further decreases, and steps S103 and onwards are executed until the first phase shift amount θ1 reaches a maximum value (preset maximum value or 180°) (No in S108). On the other hand, if the first phase shift amount θ1 reaches a maximum value (Yes in S108) and the output from the first bridge circuit 11 to the second bridge circuit 12 is controlled to a minimum value (0 or a preset minimum value), but the output voltage V2 rises and reaches a predetermined upper limit value (Yes in S109), the control unit 20 stops driving the first bridge circuit 11 (S110). As a result, the control unit 20 determines that the power transmission direction has started to be reversed (reverse power transmission from the second bridge circuit 12 to the first bridge circuit 11), and proceeds to FIG. 3 "A" (S201 and onwards). Note that if the output voltage V2 does not rise after the first phase shift amount θ1 reaches a maximum value (No in S109), steps S103 and onwards are executed.

Next, in FIG. 3, the flow of the startup process during reverse power transmission after FIG. 2 "A" will be described. While stopping the drive of the first bridge circuit 11, the control unit 20 performs constant voltage control of the input voltage V2 for the second bridge circuit 12, sets the phase difference between the third leg (switching elements Q5, Q6) and the fourth leg (switching elements Q7, Q8) to the maximum value (preset maximum value or 180°) and sets the drive frequency F2 to the maximum frequency F2max to start phase shift control (S201).

The control unit 20 controls the phase difference between the third leg (switching elements Q5, Q6) and the fourth leg (switching elements Q7, Q8), i.e., the second phase shift amount θ2, according to the state of the voltage V2, and reduces the second phase shift amount θ2 from the maximum value when the voltage V2 rises (S202). When the output current (primary side current) of the first bridge circuit 11 becomes equal to or greater than a predetermined threshold value due to the phase shift control (Yes in S203), the control unit 20 starts synchronous rectification control of the first bridge circuit 11 (S204). Note that in step S202, when the voltage V2 rises, the second phase shift amount θ2 is reduced, and when the voltage V2 drops, the second phase shift amount θ2 is increased.

When the output power further increases and the second phase shift amount θ2 reaches 0° (Yes in S205), the control unit 20 performs frequency control to reduce the drive frequency F2 of the second bridge circuit 12 from the maximum frequency F2max (S206) to bring the output power closer to the target value. When the output power reaches the target value, if the control unit 20 continues the control (Yes in S207), it executes steps S202 and onward, and if the control does not continue (No in S207), it ends the process.

On the other hand, if the output current (primary side current) of the first bridge circuit 11 is less than the predetermined threshold (No in S203), the synchronous rectification control of the first bridge circuit 11 is stopped (S208), and if the second phase shift amount θ2 is less than the maximum value (No in S209), steps S202 and after are executed. If the second phase shift amount θ2 becomes the maximum value (Yes in S209) and the voltage V2 becomes equal to or less than a predetermined lower limit value (Yes in S210), the control unit 20 stops driving the second bridge circuit 12 since the reverse current has decreased and reverse operation is no longer necessary (S211). As a result, the control unit 20 determines that the power transmission direction has started to be reversed (forward power transmission from the first bridge circuit 11 to the second bridge circuit 12), and the process continues to "B" (S301 and after) in FIG. 4.

Next, in FIG. 4, the flow of the startup process during forward power transmission after FIG. 3 "B" will be described. While stopping the drive of the second bridge circuit 12, the control unit 20 performs constant voltage control of the first bridge circuit 11 with the input voltage V1 and the voltage V2, while setting the phase difference between the first leg (switching elements Q1, Q2) and the second leg (switching elements Q3, Q4) to the maximum value (preset maximum value or 180°) and setting the drive frequency F1 to the maximum frequency F1max to start phase shift control (S301).

The control unit 20 performs the same control as steps S202 to S207. That is, the control unit 20 controls the phase difference between the first leg (switching elements Q1, Q2) and the second leg (switching elements Q3, Q4), i.e., the first phase shift amount θ1, according to the state of the voltage V2. When the voltage V2 decreases, the control unit 20 decreases the first phase shift amount θ1 from the maximum value (S302). When the output current of the second bridge circuit 12 becomes equal to or greater than a predetermined threshold value (Yes in S303), the control unit 20 starts synchronous rectification control of the second bridge circuit 12 (S304). Until the first phase shift amount θ1 reaches 0° (No in S305), the control unit 20 executes steps S302 and after. Note that in step S302, when the voltage V2 decreases, the control unit 20 decreases the first phase shift amount θ1, and when the voltage V2 increases, the control unit 20 increases the first phase shift amount θ1.

When the output power further increases and the first phase shift amount θ1 of the first bridge circuit 11 reaches 0° (Yes in S305), frequency control is performed to reduce the drive frequency F1 of the first bridge circuit 11 from the maximum frequency F1max to match the output power (S306). If the control is to be continued (Yes in S307), the state transitions from "C" in FIG. 2 to step S101. If the control is not to be continued (No in S307), the process ends.

On the other hand, if the output current of the second bridge circuit 12 is less than the predetermined threshold value (No in S303), the synchronous rectification control of the second bridge circuit 12 is stopped (S308), and if the first phase shift amount θ1 is less than the maximum value (No in S309), steps S302 and after are executed. When the first phase shift amount θ1 becomes the maximum value (Yes in S309), steps S302 and after are executed until a voltage is applied to the second bridge circuit 12 from the outside and the voltage V2 becomes a predetermined upper limit value (No in S310), and when it becomes equal to or greater than the upper limit value (Yes in S310), the state transitions from "D" in FIG. 2 to step S110.

In the above description, the process in which the answer is Yes in step S102 and the drive frequency F1 of the first bridge circuit 11 is maximized corresponds to the "first process" of the present invention. The process in which the first phase shift amount θ1 of the first bridge circuit 11 is increased in step S103 corresponds to the "second process" of the present invention. The process in which the output current of the second bridge circuit 12 is below the threshold value in steps S104 and S107 and the synchronous rectification control of the second bridge circuit 12 is stopped corresponds to the "third process" of the present invention. The process in which the first phase shift amount θ1 of the first bridge circuit 11 is maximized in steps S108 to S110 and the voltage V2 is above the upper limit value and the drive of the first bridge circuit 11 is stopped corresponds to the "fourth process" of the present invention.

The process of starting phase shift control for the second bridge circuit 12 with the maximum phase difference and maximum frequency F2max while stopping the drive of the first bridge circuit 11 in step S201 corresponds to the "fifth process" of the present invention. The process of decreasing the second phase shift amount θ2 of the second bridge circuit 12 in steps S202 to S204 and starting synchronous rectification control of the first bridge circuit 11 when the output current of the first bridge circuit 11 becomes equal to or greater than the threshold corresponds to the "sixth process" of the present invention. The process of performing frequency control in which the second phase shift amount θ2 of the second bridge circuit 12 becomes 0° and the drive frequency F2 is reduced from the maximum frequency F2max in steps S205 and S206 corresponds to the "seventh process" of the present invention.

In the above explanation, the "third process" is performed after the "second process" and the "seventh process" is performed after the "sixth process". However, whether the synchronous rectification control is stopped or started depends on the output current threshold and is unrelated to the operation of the phase shift control of the drive circuit, so the synchronous rectification control may be stopped when the phase shift control is performed. In other words, the "second process" may be performed after the "third process", and the processes of steps S203 and S204 of the "sixth process" may be performed after the "seventh process".

Figure 5 is a timing diagram showing the change in the phase difference θ (first phase shift amount θ1) between the first leg and the second leg of the first bridge circuit 11.

Figure 5 (A) shows a state in which the phase difference θ between the first leg (switching elements Q1, Q2) and the second leg (switching elements Q3, Q4) is 0° (no phase difference), which corresponds to the state of step S101 in Figure 2.

In FIG. 5(B), there is a slight phase difference θ between the first leg and the second leg, and in FIG. 5(C), the phase difference θ is large. These correspond to the state of step S103 in FIG. 2. In FIG. 5(D), the phase difference θ between the first leg and the second leg is θ = 180° (maximum value), which corresponds to the Yes state in step S108 in FIG. 2.

[Bidirectional power supply]
6 shows a configuration diagram of a power supply system using a bidirectional power supply device 2 according to an embodiment of the present invention. In FIG. 6, the bidirectional power supply device 2 is applied to a V2H (Vehicle to Home) system.

The bidirectional power supply 2 comprises a DC/DC power supply unit 10, a DC/AC power supply unit 30, a control unit 21 that controls these, and terminals T1 to T6. The DC/DC power supply unit 10 and the part of the control unit 21 that controls the DC/DC power supply unit 10 (i.e., the part that has the functions of the control unit 20) correspond to the bidirectional current resonant DC/DC converter 1, and the DC/AC power supply unit 30 and the part of the control unit 21 that controls the DC/AC power supply unit 30 correspond to the bidirectional DC/AC inverter of the present invention.

The DC/DC power supply unit 10 has the same configuration as the DC/DC power supply unit 10 of the bidirectional current resonant DC/DC converter 1. An electrolytic capacitor (not shown) for voltage stabilization is connected to the connection point (between terminals T3 and T4) between the DC/DC power supply unit 10 and the DC/AC power supply unit 30.

The DC/AC power supply unit 30 includes multiple switching elements and inductors, and is configured to perform DC/AC conversion operations to convert DC power to AC power and AC/DC conversion operations to convert AC power to DC power under the control of the control unit 21 (e.g., under PWM control).

The control unit 21 has the functions of the control unit 20 described above, and is further configured to perform switching control (e.g., PWM control) for turning on/off multiple switching elements of the DC/AC power supply unit 30. Like the control unit 20, the control unit 21 may be configured with an analog circuit, or may be configured with a digital circuit such as a microcontroller or DSP, or may be configured with a circuit that combines an analog circuit and a digital circuit.

In the bidirectional power supply 2, the DC/DC power supply unit 10 and the DC/AC power supply unit 30 are connected to each other via terminals T3 and T4. The DC side terminals T1 and T2 of the bidirectional power supply 2 are connected to an electric vehicle (EV) 40 via a charge/discharge connector (not shown). The AC side terminals T5 and T6 of the bidirectional power supply 2 are connected to a load 50 such as a household electrical appliance or electrical equipment, and are also connected to a power conditioner device (PCS) 61 of a solar panel (PV) 60, and are further connected to a power system 71 (system power source V4) via a system disconnector (SW) 70.

The bidirectional power supply device 2 has an operation unit (e.g., an operation panel) not shown, and when a charge command is received by the operation unit, the device performs a charging operation on the electric vehicle 40, whereas when a discharge command is received by the operation unit, the device performs a discharging operation on the electric vehicle 40.

Specifically, during charging operation, the bidirectional power supply device 2 operates as a charging power supply, converting the system power supplied to terminals T5 and T6 and/or the power generated by the solar panel 60 that has been converted to AC by the power conditioner device 61 into DC power, and supplies the DC power to the electric vehicle 40 under constant current control.

When discharging, the bidirectional power supply device 2 operates as a regenerative power supply when the power grid 71 is energized, converting the DC power stored in the battery of the electric vehicle 40 into AC power of commercial frequency, and outputting the AC power from terminals T5 and T6 via grid connection.

On the other hand, in the event of a power outage in which the power grid 71 is out, the bidirectional power supply device 2 during discharging operation operates the system disconnector 70 to disconnect from the power grid 71, performs independent operation, and supplies AC power to the load 50. The power conditioner device 61 can supply AC power to the load 50 by connecting (pseudo-connecting) to the AC power supplied by the bidirectional power supply device 2.

When there is surplus power generated by the solar panel 60, the bidirectional power supply 2 can use the surplus power to charge the electric vehicle 40, and can store power in the battery of the electric vehicle 40 even during a power outage in the power grid 71. This allows the bidirectional power supply 2 to operate autonomously for long periods of time.

Therefore, when surplus power is generated and the current flow is reversed during discharging operation from terminals T1, T2 to terminals T5, T6 under constant voltage control, the bidirectional power supply 2 must smoothly and seamlessly switch the power transmission direction without a power jump during switching. In addition, when the surplus power decreases after switching the power transmission direction and the current flow returns to normal, the power transmission direction must be smoothly and seamlessly returned from charging to discharging.

During discharge operation, the DC/DC power supply unit 10 receives voltage V1, which is the battery voltage of the electric vehicle 40, and performs constant voltage control of link voltage V2 (corresponding to the "connection point voltage" of the present invention), which is the voltage at the connection point (between terminals T3 and T4) with the DC/AC power supply unit 30. The DC/AC power supply unit 30 converts the direct current link voltage V2 into a commercial AC voltage (voltage V3) through a DC/AC conversion operation, and performs constant voltage control of voltage V3.

When the power generated by the solar panel 60 becomes greater than the power consumption of the load 50 and surplus power is generated, the voltage V3 rises, and the DC/AC power supply unit 30 starts a power conversion operation (AC/DC conversion operation) in the reverse direction (charging direction). As a result, in the bidirectional power supply 2, the direction of the current is reversed and the link voltage V2 rises. The DC/DC power supply unit 10, under the control (switching control) of the control unit 21, can smoothly and seamlessly reverse the current direction and reverse the power transmission direction without a power jump at the time of switching, even if the link voltage V2 rises. In addition, after the power transmission direction is switched, when the difference between the power consumption of the load 50 and the power generated by the solar panel 60 decreases and the surplus power disappears, the voltage V3 drops, and the DC/AC power supply unit 30 starts a power conversion operation (DC/AC conversion operation) in the original discharge direction. As a result, in the bidirectional power supply 2, the direction of the current is reversed and the link voltage V2 decreases. The DC/DC power supply unit 10, under the control of the control unit 21 (switching control), can smoothly and seamlessly reverse the current direction and return the power transmission direction to the original direction without a power jump when switching, even if the link voltage V2 decreases.

Figure 7 shows a schematic diagram of the changes in drive frequency and output power when the power transmission direction of the DC/DC power supply unit 10 in the bidirectional power supply device 2 is switched.

The third quadrant of FIG. 7 shows the amount of power transmission on the discharge side (from the V1 side to the V3 side), with the horizontal axis Fd representing the drive frequency of the DC/DC power supply unit 10 and the vertical axis Pd representing the discharge power. The first quadrant of FIG. 7 shows the amount of power transmission on the charge side (from the V3 side to the V1 side), with the horizontal axis Fc representing the drive frequency of the DC/DC power supply unit 10 and the vertical axis Pc representing the charge power.

During the discharging operation, in the bidirectional power supply 2, the control unit 21 controls the DC/DC power supply unit 10 and the DC/AC power supply unit 30 so that the discharge power (output power of the bidirectional power supply 2) is at point Ad. When the power difference between the power consumed by the load 50 and the power generated by the solar panel 60 subsequently decreases, the drive frequency of the first bridge circuit 11 of the DC/DC power supply unit 10 increases, the discharge power decreases to point Bd, and the drive frequency reaches the maximum frequency on the discharge side.

When the drive frequency reaches the maximum frequency on the discharge side, the control unit 21 starts phase shift control of the first bridge circuit 11 and increases the first phase shift amount θ1 between the first leg (switching elements Q1, Q2) and the second leg (switching elements Q3, Q4) from 0°. When the phase shift control causes the discharge power to drop to point Cd, the control unit 21 stops the synchronous rectification control of the second bridge circuit 12 of the DC/DC power supply unit 10. When the output power further drops and the first phase shift amount θ1 reaches its maximum value, the discharge power reaches point Dd and becomes 0. When the discharge power becomes 0, the control unit 21 stops driving the first bridge circuit 11.

When surplus power (the power difference between the power generated by the solar panel 60 and the power consumed by the load 50) occurs in the power generated by the solar panel 60, the DC/AC power supply unit 30 starts reverse power conversion operation (AC/DC conversion operation) under the control of the control unit 21, and the link voltage V2 rises. The control unit 21 starts phase shift control with the drive frequency at the maximum frequency and with the second phase shift amount θ2 between the third leg (switching elements Q5, Q6) and the fourth leg (switching elements Q7, Q8) at the maximum value for the second bridge circuit 12 so that the charging power (the output power of the bidirectional power supply device 2) is at point Dc.

When the surplus power increases, the control unit 21 reduces the second phase shift amount θ2. When the charging power increases to point Cc due to phase shift control, the control unit 21 starts synchronous rectification control of the first bridge circuit 11. When the second phase shift amount θ2 becomes 0°, the charging power increases to point Bc. When the charging power increases to point Bc, the control unit 20 performs frequency control to reduce the driving frequency of the second bridge circuit 12 from the maximum frequency, and brings the charging power closer to the target value of point Ac.

The relationship lines of the charge power and discharge power versus drive frequency in FIG. 7 vary depending on the input voltage, and the higher the input voltage, the higher the output value even at the same drive frequency. The maximum frequency during discharge and the maximum frequency during charging may be the same or different. Although no hysteresis is provided for the power values at the start/stop of synchronous rectification control on the discharge side and charge side, it is preferable to provide hysteresis to prevent hunting. The maximum frequencies that differ for each of these operating conditions may be stored in a memory unit (not shown) in the control unit 20 or control unit 21 for the input voltage range during charging/discharging, and the optimal maximum frequency may be read out and controlled depending on the operating conditions.

The bidirectional current resonant DC/DC converter 1 and the bidirectional power supply 2 perform phase shift control of the drive circuit of the DC/DC power supply unit 10, thereby enabling seamless switching of the power transmission direction at zero power (no power jump at switching), which cannot be achieved by frequency control alone. In other words, the bidirectional current resonant DC/DC converter 1 and the bidirectional power supply 2 perform phase shift control of the drive signal between each leg of the drive circuit of the DC/DC power supply unit 10 in the range from 0 to the maximum value, making it unnecessary to use a signal between the primary side and the secondary side for phase shift control, and also unnecessary to control the phase difference between the first bridge circuit 11 and the second bridge circuit 12. This allows the use of design methods of the conventional LLC method or CLLC method, making relatively simple control possible (for example, control over a range about 10 times wider than the dead time) and reducing costs.

In the bidirectional current resonant DC/DC converter 1 and the bidirectional power supply 2, the power transmission direction is switched based on the output current value and output voltage value of the DC/DC power supply unit 10, so that the power transmission direction can be switched seamlessly without an external control signal. In addition, in the bidirectional current resonant DC/DC converter 1 and the bidirectional power supply 2, the power transmission direction is switched while the second bridge circuit 12 of the DC/DC power supply unit 10 is subjected to constant voltage control, so that only the current direction can be switched.

In this embodiment, the phase shift between the drive legs is changed from 0° to the maximum value or 180° to reduce the output from the maximum drive frequency (maximum frequencies F1max and F2max). However, this is not limiting. The on ratio of the switching element may be reduced from 50% to the minimum value (preset minimum value or 0%) using PWM control to limit the load current and reduce the output.

Figures 8 (A) to (D) are timing diagrams showing the change in duty D of the PWM control of the first bridge circuit 11. Figures 8 (A) to (D) correspond to Figures 5 (A) to (D). Figure 8 (A) shows a state where duty D = 50%, in Figure 8 (B) duty D is smaller than 50%, and in Figure 8 (C) duty D is even smaller. And in Figure 8 (D) duty D = 0%. Note that in Figure 8, duty D is changed by controlling the timing of turn-off, but duty D may be changed by controlling the timing of turn-on, or by other control.

The above describes the embodiments of the bidirectional current resonant DC/DC converter and bidirectional power supply device according to the present invention, but the present invention is not limited to the above embodiments.

[Diode rectification control]
In the above embodiment, synchronous rectification control is performed on the second bridge circuit 12 during forward control, and synchronous rectification control is performed on the first bridge circuit 11 during reverse control, but diode rectification control may be performed instead of synchronous rectification control.

In the case of diode rectification control, during forward control, the switching elements Q5 to Q8 of the second bridge circuit 12 are kept in a continuous off state, causing the second bridge circuit 12 to perform diode rectification, and during reverse control, the switching elements Q1 to Q4 of the first bridge circuit 11 are kept in a continuous off state, causing the first bridge circuit 11 to perform diode rectification.

In the case of diode rectification control, for example, in switching control during power transmission from the first bridge circuit 11 to the second bridge circuit 12 (forward control), the third and sixth processes can be omitted, and the first, second, fourth, fifth, and seventh processes can be executed.

That is, in the case of diode rectification control,
(1) A “first process” in which the drive frequency F1 of the first bridge circuit 11 is maximized;
(2) A “second process” in which the first phase shift amount θ1 between the legs of the first bridge circuit 11 is increased from 0 by phase shift control (or the on-ratio (first on-ratio) of the switching elements Q1 to Q4 is decreased from 50% by PWM control);
(3) A fourth process in which the first phase shift amount θ1 becomes a maximum value (or the first on-ratio becomes a minimum value), the voltage V2 becomes equal to or greater than the upper limit value, and the driving of the first bridge circuit 11 is stopped;
(4) A "fifth process" in which, while the driving of the first bridge circuit 11 is stopped, the second bridge circuit 12 is caused to start phase shift control with a maximum phase difference and a maximum frequency (or the second bridge circuit 12 is caused to start PWM control with a minimum on-ratio (second on-ratio) of the switching elements Q5 to Q8 and a maximum frequency);
(5) A "seventh process" in which the second phase shift amount θ2 between the legs of the second bridge circuit 12 becomes 0° (or the second on ratio becomes 50%) and frequency control is performed to reduce the drive frequency F2 from the maximum frequency F2max; and
Just execute the following.

[Link voltage control]
FIG. 9 shows a schematic diagram of a change in the link voltage V2 when the bidirectional power supply 2 according to the embodiment switches between a discharging operation and a charging operation.

In the case of FIG. 9(A), the link voltage V2 of the DC/DC power supply unit 10 during discharging is voltage a, but the link voltage V2 of the DC/AC power supply unit 30 during charging is voltage b (where voltage b is less than voltage a). Therefore, when switching between charging and discharging, the electrolytic capacitor for voltage stabilization connected to the connection point (between terminals T3 and T4) where the link voltage V2 is generated is charged and discharged, resulting in unnecessary power consumption.

On the other hand, in the case of FIG. 9(B), the link voltage V2 of the DC/DC power supply unit 10 during discharging operation and the link voltage V2 of the DC/AC power supply unit 30 during charging operation are both c voltages. Therefore, even when switching between charging and discharging operation, no charging or discharging occurs in the electrolytic capacitor for voltage stabilization connected to the connection point (between terminals T3 and T4), so no unnecessary power is consumed. Therefore, in the case of FIG. 9(B), when switching the power transmission direction, it is possible to switch the power transmission direction immediately without waiting for a change in the link voltage V2. Note that the link voltage V2 may be variable or fixed depending on, for example, the voltage V1.

In other words, the bidirectional power supply device 2 can instantly switch the power transmission direction without waiting for a change in the link voltage V2 by setting the link voltage V2 between the DC/DC power supply unit 10 and the DC/AC power supply unit 30 at a constant voltage or a voltage determined by the input voltage (voltage V1) of the DC/DC power supply unit 10 when switching the power transmission direction.

[Other Modifications]
The bidirectional current resonant type DC/DC converter according to the present invention includes a transformer, a first bridge circuit, a second bridge circuit, a primary side resonant circuit, a secondary side resonant circuit, and a control unit, and the control unit executes forward control, in which the first bridge circuit is used as a drive circuit, and performs synchronous rectification control or diode rectification control on the second bridge circuit to transmit power from the first bridge circuit to the second bridge circuit, reverse control, in which the second bridge circuit is used as a drive circuit, and performs synchronous rectification control or diode rectification control on the first bridge circuit to transmit power from the second bridge circuit to the first bridge circuit, and switching control, in which the drive frequency of the drive circuit is increased and then the direction of power transmission is switched, and the configuration can be changed as appropriate in the switching control, as long as the phase shift amount between the legs of the drive circuit is controlled within a range from 0 to a maximum value (a preset maximum value or 180°) or the on ratio of the switching element is changed within a range from 50% to a minimum value (a preset minimum value or 0%) by PWM control.

In other words, in the embodiment, one drive frequency is set to the maximum value, the phase shift amount is increased from 0 to the maximum value, and then the drive frequency is switched to the other drive frequency, the phase shift amount is reduced from the maximum value to 0, and then the drive frequency is reduced from the maximum value. However, it is also possible to increase the drive frequency while increasing the phase shift amount from 0 with one drive frequency set to a sufficiently large value (for example, about 1.5 to 2 times the resonant frequency determined by the resonant coil and resonant capacitor), switch to the other drive frequency with a sufficiently large value and a maximum phase shift amount, reduce the phase shift amount from the maximum value, and reduce the drive frequency from a sufficiently large value, and continue to reduce the drive frequency after the phase shift amount becomes 0.

When the drive frequency is higher than the resonant frequency, the rate of gain reduction due to an increase in drive frequency decreases, while the heat generated by the transformer and resonant coil increases as the drive frequency increases. Therefore, it is preferable to determine the maximum drive frequency (maximum frequencies F1max and F2max) taking both into consideration.

REFERENCE SIGNS LIST 1 Bidirectional current resonant DC/DC converter 2 Bidirectional power supply device 10 DC/DC power supply unit 11 First bridge circuit 12 Second bridge circuit 13 Primary side resonant circuit 14 Secondary side resonant circuit 2 Bidirectional power supply devices 20, 21 Control unit 30 DC/AC power supply unit 40 Electric vehicle 50 Load 60 Solar panel 61 Power conditioner device 70 System disconnector 71 Power system

Claims (7)

A bidirectional current resonant DC/DC converter including a power supply unit and a control unit,
The power supply unit includes:
a transformer including a primary coil and a secondary coil;
a first bridge circuit including a first leg and a second leg connected in parallel, each leg including an upper arm and a lower arm connected in series, each arm including a switching element, and connected to the primary coil;
a second bridge circuit including a third leg and a fourth leg connected in parallel, each leg including an upper arm and a lower arm connected in series, each arm including a switching element, and connected to the secondary coil;
a primary-side resonant circuit provided between the first bridge circuit and the primary-side coil, the primary-side resonant circuit including a first resonant coil and a first resonant capacitor;
a secondary resonant circuit provided between the second bridge circuit and the secondary coil, the secondary resonant circuit including a second resonant coil and a second resonant capacitor;
Equipped with
The control unit is
a forward control in which the first bridge circuit is used as a drive circuit, and synchronous rectification control or diode rectification control is performed on the second bridge circuit to transmit power from the first bridge circuit to the second bridge circuit;
a reverse control in which the second bridge circuit is used as a drive circuit, synchronous rectification control or diode rectification control is performed on the first bridge circuit, and power is transmitted from the second bridge circuit to the first bridge circuit;
and increasing the drive frequency of the drive circuit and then switching the direction of power transmission.
a bidirectional current resonant DC/DC converter characterized in that in the switching control, a phase shift control between each leg of the drive circuit is performed with a phase shift amount in a range from 0 to a maximum value, or PWM control is performed on each switching element of the drive circuit to change an on-ratio of the switching element in a range from 50% to a minimum value. The control unit is
In the forward control, the synchronous rectification control of the second bridge circuit is performed;
In the switching control during the forward control,
a first process of increasing a drive frequency of the first bridge circuit to a predetermined maximum frequency;
a second process of performing a phase shift control between the legs of the first bridge circuit and increasing a first phase shift amount of the phase shift control from 0, or performing a PWM control of each switching element of the first bridge circuit and decreasing a first on ratio of each switching element of the first bridge circuit from 50%;
a third process of stopping the synchronous rectification control of the second bridge circuit when an output current value of the second bridge circuit becomes equal to or lower than a predetermined threshold value;
a fourth process of setting the first phase shift amount to a maximum value or the first on ratio to a minimum value, and stopping the driving of the first bridge circuit when an output voltage value of the second bridge circuit reaches a predetermined upper limit value;
a fifth process of setting a drive frequency of the second bridge circuit to a predetermined maximum frequency and a second phase shift amount between each leg of the second bridge circuit to a maximum value to start phase shift control between each leg of the second bridge circuit, or setting a second on-ratio of each switching element of the second bridge circuit to a minimum value to start PWM control of each switching element of the second bridge circuit;
a sixth process of starting the synchronous rectification control of the first bridge circuit when an output current value of the first bridge circuit becomes equal to or greater than a predetermined threshold value;
and a seventh process of reducing a drive frequency of the second bridge circuit from the maximum frequency by setting the second phase shift amount to 0 or setting the second on ratio to 50%. The control unit is
3. The bidirectional current resonant DC/DC converter according to claim 2, wherein at least the fifth process is executed while performing a constant voltage control on the second bridge circuit. The control unit is
execute the third process before executing the second process;
3. The bidirectional current resonant DC/DC converter according to claim 2, wherein the seventh process is executed before the sixth process is executed. The control unit is
In the forward control, the diode rectification control of the second bridge circuit is performed;
In the switching control during the forward control,
a first process of increasing a drive frequency of the first bridge circuit to a predetermined maximum frequency;
a second process of performing a phase shift control between the legs of the first bridge circuit and increasing a first phase shift amount of the phase shift control from 0, or performing a PWM control of each switching element of the first bridge circuit and decreasing a first on ratio of each switching element of the first bridge circuit from 50%;
a fourth process of setting the first phase shift amount to a maximum value or the first on ratio to a minimum value, and stopping the driving of the first bridge circuit when an output voltage value of the second bridge circuit reaches a predetermined upper limit value;
a fifth process of setting a drive frequency of the second bridge circuit to a predetermined maximum frequency and a second phase shift amount between each leg of the second bridge circuit to a maximum value to start phase shift control between each leg of the second bridge circuit, or setting a second on-ratio of each switching element of the second bridge circuit to a minimum value to start PWM control of each switching element of the second bridge circuit;
and a seventh process of reducing a drive frequency of the second bridge circuit from the maximum frequency by setting the second phase shift amount to 0 or setting the second on ratio to 50%. A bidirectional current resonant DC/DC converter according to any one of claims 1 to 5,
a bidirectional DC/AC inverter that performs a DC/AC conversion operation and an AC/DC conversion operation;
A bidirectional power supply device comprising:
a bidirectional power supply device characterized in that, when switching the direction of power transmission, a connection point voltage between the bidirectional current resonant DC/DC converter and the bidirectional DC/AC inverter becomes a constant voltage or a voltage determined by the input voltage of the bidirectional current resonant DC/DC converter. The bidirectional current resonant DC/DC converter is configured to be connectable to an electric vehicle,
7. The bidirectional power supply device according to claim 6, wherein the bidirectional power supply device performs charging and discharging operations for the electric vehicle.

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