JP7743262B2 - Motor control device - Google Patents

Description

The present invention relates to a motor control device that detects phase currents using current detection elements connected to the DC side of an inverter circuit and drives a motor.

Conventional motor control devices for driving motors use PWM (Pulse Width Modulation) control of the switching elements of each of the UVW phases of a three-phase inverter circuit, but to detect the phase currents used for control, they are equipped with current detection elements (shunt resistors or Hall elements) that detect the current on the DC side of the inverter circuit.

In particular, when using a single shunt resistor, known as a single-shunt current detection method, the current detection timing is set according to the current pattern generated by the PWM signal, allowing the maximum and minimum phase currents of the UVW phases to be detected, ultimately determining the phase current for each phase. Furthermore, when sufficient current detection time cannot be secured, current detection is made possible by shifting the phase of the PWM signal (pulse shift) (see, for example, Patent Documents 1 and 2).

Patent No. 5951208 JP 2010-88260 A

However, due to factors such as the current detection timing and pulse shift, the detected current has an error compared to the sinusoidal phase current (ideal current) that flows when an ideal sinusoidal voltage is applied. This error is called a ripple current, and because the magnitude of this ripple current varies depending on the detection timing, a different error (ripple current) is included in the detected current each time the current is detected.

This problem affects, for example, when estimating rotor position using sensorless control, resulting in poor controllability. Therefore, in the aforementioned Patent Document 1, the same current pattern (PWM pattern) is output twice, the same phase current is detected twice at different times, and the degree of change is calculated from the difference between these to correct the error. However, this has the disadvantage of complicating control, such as requiring the use of carriers with different waveforms for each phase and the need to make the current detection cycle faster than the current control cycle.

The present invention was made to solve these conventional technical problems, and provides a motor control device that can accurately correct the detected current, which includes ripple current, which is an error from the ideal current, thereby improving controllability.

A motor control device of the present invention includes an inverter circuit made up of a plurality of switching elements, and a control unit that generates a current conduction pattern for switching the switching elements. By switching the switching elements according to the current conduction pattern, DC voltage is converted into AC voltage to drive a motor. The motor control device includes a current detection element connected to the DC side of the inverter circuit and generating a signal corresponding to a current value, a phase current detection unit that detects the phase current of the motor based on the signal generated by the current detection element, a predicted ripple current calculation unit that predicts the ripple current contained in the phase current detected by the phase current detection unit, and a phase current correction unit that corrects the phase current detected by the phase current detection unit based on the ripple current predicted by the predicted ripple current calculation unit. The predicted ripple current calculation unit calculates the amount of change in ripple current per unit time from the DC voltage, the phase voltage, and the inductance of the circuit including the motor, and predicts the ripple current based on the current conduction pattern, a predetermined reference point which becomes an ideal current, the detection timing of the phase current detection unit, and the calculated amount of change in ripple current per unit time . The control unit generates the current conduction pattern based on the phase current corrected by the phase current correction unit.

A motor control device according to a second aspect of the present invention is characterized in that, in the above-mentioned invention , the predicted ripple current calculation section calculates the amount of change in ripple current per unit time in accordance with a pattern of the line voltage within the carrier cycle.

A motor control device of a third invention is characterized in that, in each of the above inventions, the current detection element is connected to a DC bus, and the phase current detection unit detects the phase current of the motor based on the signal generated by the current detection element and the current flow pattern.

According to the present invention, a motor control device includes an inverter circuit composed of multiple switching elements and a control unit that generates a conduction pattern for switching the switching elements. The motor drives a motor by converting DC voltage to AC voltage by switching the switching elements according to the conduction pattern. A current detection element connected to the DC side of the inverter circuit generates a signal corresponding to the current value, and a phase current detection unit detects the motor's phase current based on the signal generated by the current detection element. The predicted ripple current calculation unit predicts the ripple current contained in the phase current detected by the phase current detection unit, and the phase current correction unit corrects the phase current detected by the phase current detection unit based on the ripple current predicted by the predicted ripple current calculation unit. The control unit generates a conduction pattern based on the phase current corrected by the phase current correction unit. Therefore, the phase current (detected current) including the ripple current detected by the phase current detection unit is corrected to an ideal current or a value close to it by the phase current correction unit, and the control unit can generate a conduction pattern for switching the inverter circuit's switching elements based on the corrected phase current.

This eliminates or suppresses the effects of the ripple current, improving motor controllability. In this case, the predicted ripple current calculation unit predicts the ripple current contained in the phase current, but by predicting the ripple current based on the current conduction pattern, a predetermined reference point that represents the ideal current, the detection timing of the phase current detection unit, and the amount of change in ripple current per unit time, it becomes possible to accurately predict the ripple current and accurately correct the detected current.

Furthermore, since the amount of change in ripple current varies depending on the DC voltage, phase voltage, and inductance of the circuit including the motor, the predicted ripple current calculation unit calculates the amount of change in ripple current per unit time from these values, which makes it possible to predict and calculate the ripple current with high accuracy.

Furthermore, since the amount of change in ripple current per unit time varies depending on the pattern of the line-to-line voltage within the carrier cycle, the predicted ripple current calculation unit may calculate the amount of change in ripple current per unit time in accordance with the pattern of the line-to-line voltage within the carrier cycle, as in the second aspect of the present invention.

Furthermore, the above inventions are extremely effective in the case of so-called one-shunt current detection, in which a current detection element is connected to a DC bus as in the third invention , and the phase current detection unit detects the phase current of the motor based on the signal generated by the current detection element and the current flow pattern.

1 is an electrical circuit diagram of a motor control device according to an embodiment of the present invention; FIG. 2 is a functional block diagram of the motor control device of FIG. 1 . 3 is a diagram showing a current conduction pattern (PWM signal) for one carrier period output by the control unit in FIG. 2 and ripple currents of maximum and minimum phases. FIG. 1 is a diagram illustrating the relationship between an ideal current that flows when an ideal sine wave voltage is applied to a motor and a current that flows due to a voltage that is actually applied according to a current conduction pattern (PWM signal). FIG. 5 is a diagram illustrating a method for determining the gradient of the ripple current (amount of change in the ripple current per unit time) in FIG. 4 . FIG. 6 is a circuit diagram for explaining a method for determining the slope of the ripple current in FIG. 5 . FIG. 4 is a diagram showing seven phase regions within one carrier period of FIG. 3.

Embodiments of the present invention will now be described in detail with reference to the drawings. Figure 1 is an electrical circuit diagram of a motor control device 1 according to one embodiment of the present invention, and Figure 2 is a functional block diagram of the motor control device 1.

(1) Motor control device 1
The motor control device 1 of the embodiment includes an inverter circuit 4 and a control board 6, and is configured to convert a DC voltage (hereinafter referred to as an HV voltage _VHV ) supplied from a vehicle HV battery 7, which is a DC power source, into a three-phase AC voltage of a predetermined frequency and apply the AC voltage to a motor 8 to drive the motor 8.

Motor 8, driven and controlled by motor control device 1 in the embodiment, drives the compression mechanism of an on-board electric compressor (not shown) and is composed of a permanent magnet synchronous motor (PMSM), typified by an interior permanent magnet synchronous motor (IPMSM). This motor 8 (IPMSM in the embodiment) is composed of a stator with concentrated windings 2U, 2V, and 2W (Figure 2) on the UVW axes wound around the teeth, and a rotor with embedded permanent magnets that form magnetic poles, positioned inside the stator with a specified air gap.

(2) Inverter circuit 4
The inverter circuit 4 converts the HV voltage V _HV (e.g., 350 V) of the HV battery 7 into a three-phase AC voltage and applies it to the motor 8. The inverter circuit 4 has a U-phase half-bridge circuit 11U, a V-phase half-bridge circuit 11V, and a W-phase half-bridge circuit 11W, and each of the half-bridge circuits 11U to 11W for each phase has upper arm switching elements 12A, 12B, and 12C and lower arm switching elements 12D, 12E, and 12F. Furthermore, a flywheel diode 13 is connected in anti-parallel to each of the switching elements 12A to 12F.

The upper ends of the upper arm switching elements 12A to 12C of the inverter circuit 4 are connected to the DC bus 10 on the positive side of the HV battery 7. On the other hand, the lower ends of the lower arm switching elements 12D to 12F of the inverter circuit 4 are connected to the DC bus 15 on the negative side of the HV battery 7.

In this embodiment, these multiple switching elements 12A-12F are composed of insulated gate bipolar transistors (IGBTs) with a MOS structure incorporated in the gate portion. Furthermore, the upper arm switching element 12A and the lower arm switching element 12D of the U-phase half-bridge circuit 11U are connected in series, the upper arm switching element 12B and the lower arm switching element 12E of the V-phase half-bridge circuit 11V are connected in series, and the upper arm switching element 12C and the lower arm switching element 12F of the W-phase half-bridge circuit 11W are connected in series.

A connection point P1 (U-phase voltage V _u ) between the upper arm switching element 12A and the lower arm switching element 12D of the U-phase half-bridge circuit 11U is connected to the U-phase winding 2U of the motor 8, a connection point P2 (V-phase voltage V _v ) between the upper arm switching element 12B and the lower arm switching element 12E of the V-phase half-bridge circuit 11V is connected to the V-phase winding 2V of the motor 8, and a connection point P3 (W-phase voltage V _w ) between the upper arm switching element 12C and the lower arm switching element 12F of the W-phase half-bridge circuit 11W is connected to the W-phase winding 2W of the motor 8. The inverter circuit 4 switches the HV voltage V _HV (DC voltage) input from the HV battery 7 by the ON-OFF operation of these switching elements 12A to 12F, converts it into a three-phase AC voltage, and applies it to the motor 8.

(3) Current detection element 16
A current detection element 16 is connected to the negative DC bus 15 (the DC side of the inverter circuit 4) between the HV battery 7 and the inverter circuit 4. The current detection element 16 generates a signal corresponding to the value of the DC current (phase current: shunt current) flowing through the inverter circuit 4. In this embodiment, the current detection element 16 is connected to the DC bus 15, but may also be connected to the positive DC bus 10. The current detection element 16 can be configured using a shunt resistor or a Hall element, but in this embodiment, it is configured using a shunt resistor.

(4) Control board 6
Meanwhile, a microcomputer having a processor and a control circuit are mounted on the control board 6 of the motor control device 1 of this embodiment, and an LV battery 14 that serves as the power source for these elements is also connected to the control board 6 (FIG. 1). FIG. 2 shows the functional blocks (mainly the functions of the microcomputer) of the control board 6 of this motor control device 1. In FIG. 2, the motor control device 1 includes, as functions of the microcomputer mounted on the control board 6, a control unit 21, a phase current correction unit 22, a DC voltage detection unit 23, a phase current detection unit 24, and a predicted ripple current calculation unit 26.

(4-1) Phase current detection unit 24
The phase current detection unit 24 detects and outputs the phase currents (U-phase current iU, V-phase current iV, W-phase current _iW ) of the motor 8 based on the signal ( _signal corresponding to the value of the DC current) generated by the current detection element 16 and the current conduction pattern for switching each of the switching elements _12A to 12F.

For example, in a current conduction pattern in which upper arm switching element 12A of U-phase half-bridge circuit 11U is ON, and lower arm switching element 12E of V-phase half-bridge circuit 11V and lower arm switching element 12F of W-phase half-bridge circuit 11W are ON within one carrier period, U-phase current _iU flows from switching element 12A to U-phase winding 2U of motor 8, V-phase current _iV flows from flywheel diode 13 connected in parallel to switching element 12E to V-phase winding 2V of motor 8, and W-phase current _iW flows from W-phase winding 2W of motor 8 to switching element 12F. In this case, U-phase current _iU flows through current detection element 16 and can be detected.

On the other hand, in a current conduction pattern in which upper arm switching element 12A of U-phase half-bridge circuit 11U and upper arm switching element 12B of V-phase half-bridge circuit 11V are ON and lower arm switching element 12F of W-phase half-bridge circuit 11W is ON within one carrier period, U-phase current _iU and V-phase current _iV flow from switching elements 12A and 12B to U-phase and V-phase windings 2U and 2V of motor 8, respectively, and W-phase current _iW flows from W-phase winding 2W of motor 8 to switching element 12F. In this case, W-phase current _iW flows through current detection element 16 and can be detected.

From the U-phase current _iU and W-phase current _iW thus determined, the remaining V-phase current _iV can be determined by applying Kirchhoff's current law at the neutral point P _C of the windings 2U, 2V, and 2W. Note that the phase currents (detected currents) _iU , _iV , and _iW detected by the phase current detection unit 24 contain ripple currents (ripple currents Δi _rpli and Δi _rplk described below) that are errors from the ideal currents, which will be described in detail later.

(4-2) DC voltage detection unit 23
The DC voltage detection unit 23 detects and outputs the HV voltage _VHV of the HV battery 7. The HV voltage _VHV detected and output by the DC voltage detection unit 23 and the current conduction pattern (PWM signal) calculated and output by the control unit 21 as described below are input to a predicted ripple current calculation unit 26.

(4-3) Predicted ripple current calculation unit 26
The predicted ripple current calculation unit 26 calculates the maximum phase ripple current Δi _rpli and the minimum phase ripple current Δi _rplk based on the HV voltage V _HV , the current conduction pattern, the circuit inductance, etc., and predicts and outputs these ripple currents Δi _rpli and Δi _rplk . These ripple currents Δi _rpli and Δi _rplk are included in the phase currents output by the phase current detection unit 24, and the predictive control of the maximum phase ripple current Δi _rpli and the minimum phase ripple current Δi _rplk in the predicted ripple current calculation unit 26 will be described in detail later.

The maximum phase ripple current Δi _rpli and minimum phase ripple current Δi _rplk calculated (predicted) and output by this predicted ripple current calculation unit 26, and the U-phase current i _U , V-phase current i _V , and W-phase current i _W (phase currents) detected and output by the phase current detection unit 24 are input to the phase current correction unit 22.

(4-4) Phase current correction section 22
The phase current correction unit 22 corrects and outputs the U-phase current iU, V-phase current iV, and W-phase current _iW (phase currents) detected by the phase current detection unit 24 based on the maximum phase ripple _{current Δirpli} and minimum phase ripple current _Δirplk calculated (predicted) by the predicted ripple current calculation unit _26. For example, if the U phase is the maximum phase, the phase current correction unit 22 subtracts the ripple current _Δirpli from the U-phase current _iU and outputs the corrected U-phase current _iU '. Similarly, if the W phase is the minimum phase, the phase current correction unit 22 subtracts the ripple current _Δirplk from the _W -phase current iW and outputs the corrected U-phase current _iW '. Note that for the V phase, which is an intermediate phase, the corrected V-phase current _iV ' = _-iU ' - _iW ' is calculated and output according to Kirchhoff's current law. The calculated corrected phase currents i _U ′, i _V ′, and i _W ′ are then input to the control unit 21 .

(4-5) Control Unit 21
The control unit 21 of this embodiment estimates the electrical angular velocity of the motor 8 from the phase currents _iU ', _iV ', and _iW ' corrected by the phase current correction unit 22, and generates a current conduction pattern (PWM signal) for switching each of the switching elements 12A to 12F of the inverter circuit 4 based on the estimated electrical angular velocity and the d-axis current and q-axis current obtained from the electrical angular velocity command value, and outputs the current conduction pattern to the inverter circuit 4. In this way, the motor 8 is driven by position sensorless vector control.

(5) Predictive Control of Ripple Currents Δi _rpli and Δi _rplk Next, the predictive control of ripple currents Δi _rpli and Δi _rplk by the predicted ripple current calculation unit 26 of the motor control device 1 of this embodiment will be described with reference to Figures 3 to 7. Figure 3 shows the energization pattern (PWM signal) in one carrier cycle. In the figure, the top row represents the energization pattern for maximum phase i, below that for intermediate phase j, and below that for minimum phase k. During the rising period, upper arm switching elements 12A to 12C are ON and lower arm switching elements 12D to 12F are OFF, and during the falling period, lower arm switching elements 12D to 12F are ON and upper arm switching elements 12A to 12C are OFF.

In other words, the phase for which the upper arm switching elements 12A to 12C are ON for the longest period is the maximum phase i, the phase for the shortest period is the minimum phase k, and the intermediate phase is the intermediate phase j. For example, if the upper arm switching element 12A of the U-phase half-bridge circuit 11U is ON for the longest period, the U-phase is the maximum phase i, and if the upper arm switching element 12C of the W-phase half-bridge circuit 11W is ON for the shortest period, the W-phase is the minimum phase k, and the intermediate V-phase is the intermediate phase. Figure 3 shows, for example, the above current conduction pattern.

In each half-bridge circuit 11U-11W, when the upper arm switching elements 12A-12C are ON, the lower arm switching elements 12D-12F are OFF, and when the lower arm switching elements 12D-12F are ON, the upper arm switching elements 12A-12C are OFF. Also, to prevent short circuits in each half-bridge circuit 11U-11W, a dead time is actually provided in the ON-OFF timing of the upper and lower arm switching elements 12A-12F.

Furthermore, in Figure 3, if the time from the beginning (0) of the carrier cycle until the maximum phase i (U phase) rises is _tion and the time until the middle phase j (V phase) rises is _tjon , then at current detection timing _tad1 in the period between _tion and _tjon when only the maximum phase i rises, the current ii of the maximum phase _i (U phase current _iU in Figure 3) can be detected as described above, and at current detection timing _tad2 in the period from _tjon to the rise of the minimum phase k, the current _ik of the minimum phase k (W phase current _iW in Figure 3) can be detected as described above.

(5-1) Ripple current i _rpl
The lowest line in Figure 3 shows the ripple current Δi _rplk of the smallest phase, and the highest line shows the ripple current Δi _rpli of the largest phase. This ripple current (collectively referred to as i _rpl ) will be explained with reference to Figure 4 . The ripple current i _rpl is the difference (error: i real - i ideal ) between the sinusoidal phase current that flows when an ideal sinusoidal voltage is applied (hereinafter referred to as the ideal current i _ideal ) and the current i _real that flows due to the voltage actually applied by the PWM output ( _pulse -shaped applied voltage). The aforementioned phase currents (detected currents) i _U , i _V , and i _W are this current _{i real .} In other words, the phase currents (detected currents) i _U , i _V , and i _W include the ripple current i _rpl as an error from the ideal current i _ideal ( Figure 4 ).

(5-2) Principle of Ripple Current (i _rpl ) Prediction Next, the principle of predicting this ripple current i _rpl will be explained with reference to Figures 5 and 6. First, consider the case of a single-phase circuit as shown in Figure 6. When the change in effective voltage (sinusoidal AC) during one carrier period is sufficiently small, the ideal voltage v _ideal during that period (dashed line in Figure 5) can be set to the command voltage v _com (solid horizontal line in Figure 5), and the difference Δv between the voltage v _real actually applied by the PWM output (the height of the pulse-shaped applied voltage in Figure 5) and the ideal voltage v _ideal is given by the following formula (I):
Δv=v _real -v _com ...(I)

Therefore, the difference between the voltage v _real actually applied by the PWM output and the ideal voltage v _ideal can be expressed by the following equation (II) using the ripple current i _rpl . Note that L in equation (II) is the inductance of the single-phase circuit shown in Figure 6. Also, di _rpl /dt is the amount of change per unit time of the ripple current i _rpl (hereinafter referred to as the slope), and if the current conduction pattern is constant and v _real can be considered to be constant, the slope of the ripple current i _rpl can also be considered to be constant.

(5-3) Predictive Calculation of Ripple Currents Δi _rpli and Δi _rplk Considering the case of a three-phase motor 8 as shown in Figure 3 of the embodiment, the ripple current is zero at the beginning of the carrier cycle, so this beginning can be considered as a reference point for the ideal current. If the maximum phase current (estimated value) at the beginning of the carrier cycle is i _i ', the minimum phase current (estimated value) is i _k ', the maximum phase current is i _i , and the minimum phase current is i _k , the maximum phase ripple Δi _rpli is the difference between the maximum phase current i _i ' and the maximum phase detected current i _i at the beginning of the carrier cycle (error: i _i - _{i i} '), and the minimum phase ripple Δi _rplk is the difference between the minimum phase current i _k ' at the beginning of the carrier cycle and the minimum phase detected current i _k (error: i _k - _{i k} ').

3 is divided into seven phase regions F1 to F7 shown in Fig. 7 based on the combination of ON-OFF states of each phase. Furthermore, the voltage between each line in one carrier cycle is divided into three types of line voltage patterns Pt1 to Pt3 for the current conduction pattern (PWM signal), and each phase region F1 to F7 corresponds to each of the line voltage patterns Pt1 to Pt3 as follows:
Line voltage pattern Pt1: phase regions F1, F4, F7
Line voltage pattern Pt2: phase regions F2 and F6
Line voltage pattern Pt3: phase regions F3 and F5

Since the amount of change in the ripple currents Δi _rpli and Δi _rplk per unit time varies for each line voltage pattern within one carrier period, the amount of change (slope) in the ripple currents Δi _rpli and Δi _rplk per unit time is calculated for each of the line voltage patterns Pt1 to Pt3.

Now, ignoring the effect of mutual inductance, and assuming that the total current flowing into the neutral point P _C of the motor 8 is 0 (ignoring the effect of parasitic capacitance, no charge accumulates at the neutral point, and leakage current is 0), the slope di _rpli /dt of the maximum phase ripple current Δi _rpli and the slope di _rplk /dt of the minimum phase ripple current Δi _rplk in the line voltage pattern Pt1 can be calculated using the following equations (III) and (IV), respectively.

Note that L _i , L _j , and L _k are the inductances of the maximum phase, the intermediate phase, and the minimum phase, respectively. These inductances L _i , L _j , and L _k are the inductances of the paths from the respective connection points P1, P2, and P3 to the neutral point _PC (the inductances of the circuit including the motor 8). Also, V _HV is the HV voltage (DC voltage) mentioned above. Furthermore, V _i , V _j , and V _k are the effective voltages (phase voltages) of the maximum phase, the intermediate phase, and the minimum phase, respectively. These effective voltages are found from command values, but the command values themselves may also be used (the same applies below).

Similarly, the gradient di _rpli /dt of the maximum phase ripple current Δi _rpli and the gradient di _rplk /dt of the minimum phase ripple current Δi _rplk in the line voltage pattern Pt2 can be calculated using the following formulas (V) and (VI), respectively.

Similarly, the gradient di _rplk /dt of the ripple current Δi _rplk of the minimum phase in the line voltage pattern Pt3 can be calculated as shown in the following formula (VII).

The maximum phase ripple current Δi _rpli and the minimum phase ripple current Δi _rplk in Figure 3 are the change (slope) of the ripple current from the beginning (0) of the carrier cycle in Figure 3 to the current detection timings t _ad1 and t _ad2 , so they can be calculated (predicted) as shown in the following formulas (VIII) and (IX) using the current pattern (PWM signal: t _ion , t _jon ), the beginning (reference point) of the carrier, each current detection timing t _ad1 and t _ad2 , and the slope di _rpli /dt of the maximum phase ripple current Δi _rpli and the slope di _rplk /dt of the minimum phase ripple current Δi _rplk obtained using formulas (III) to (VII).

The slopes used in equations (VIII) and (IX) can be expressed as follows in relation to the line voltage patterns Pt1 to Pt3 described above:

The first term on the right side of equation (VIII) is the slope (equation (III)) of the ripple current of the maximum phase in the line voltage pattern Pt1. Multiplying this slope by the time t _ion from the beginning (0) of the carrier cycle until the maximum phase i (U phase) rises gives the amount of change in the ripple current Δi _rpli from the beginning (0) until the maximum phase i (U phase) rises (t _ion ). The second term on the right side is the slope (equation (V)) of the ripple current in the line voltage pattern Pt2. Multiplying this slope by the time (t _{ad1 -} t _ion ) from t _ion to the current detection timing t _ad1 gives the amount of change in the ripple current _{Δi rpli} from t ion to the current detection timing t ad1. The sum of these first and second terms gives the final value of the maximum phase ripple current Δi _rpli .

The first term on the right side of equation (IX) is the slope (equation (IV)) of the ripple current of the smallest phase in the line voltage pattern Pt1, and multiplying this slope by the time t _ion from the beginning (0) of the carrier cycle until the rise of the largest phase i (U phase) is the amount of change in the ripple current Δi _rplk from the beginning (0) to the rise of the largest phase i (U phase) (t _ion ). The second term on the right side is the slope (equation (VI)) of the ripple current of the smallest phase in the line voltage pattern Pt2, and multiplying this slope by the time (t _jon - _{t ion} ) from t _ion until the rise of the middle phase j (V _phase ) is the amount of change in the ripple current Δi _rplk from t _ion to t _jon . The third term on the right side is the gradient of the ripple current of the minimum phase in the line voltage pattern Pt3 (formula (VII)), and multiplying this gradient by the time from t _jon to current detection timing t _ad2 (t _ad2 - t _jon ) is the amount of change in ripple current Δi _rplk from t _jon to current detection timing t _ad2 . The final value of the ripple current Δi _rplk of the minimum phase is calculated by adding together the first, second, and third terms.

(6) Control of Motor 8 Taking Ripple Currents Δi _rpli and Δi _rplk into Account The predicted maximum-phase ripple current Δi _rpli and minimum-phase ripple current Δi _rplk calculated in this manner by the predicted ripple current calculation unit 26 are input to the phase current correction unit 22, as described above. The phase current correction unit 22 corrects the U-phase current i _U , V-phase current i _V , and W-phase current i _W by subtracting the maximum-phase ripple current Δi _rpli and minimum-phase ripple current Δi _rplk from the U-phase current i _U , V-phase current i _V , and W-phase current i _W (phase currents) detected by the phase current detection unit 24, as described above. As a result, the U-phase current i _U , V-phase current i _V , and W-phase current i _W are corrected to ideal currents or values close to them, and these corrected phase currents i _U ', i _V ', and i _W ' are input to the control unit 21.

Then, the control unit 21 estimates the electrical angular velocity of the motor 8 from the phase currents _iU ', _iV ', _iW ' corrected by the phase current correction unit 22 as described above, and generates a current conduction pattern (PWM signal) for switching each of the switching elements 12A to 12F of the inverter circuit 4 based on the estimated electrical angular velocity and the d-axis current and q-axis current obtained from the electrical angular velocity command value, and outputs the current conduction pattern to the inverter circuit 4.

As described above in detail, according to the present invention, the phase currents (detected currents) _iU , _iV , _iW , including ripple currents, detected by the phase current detection unit 24 are corrected by the phase current correction unit 22 to ideal currents or values close to them, and the control unit ₂₁ can generate a current conduction pattern ( _PWM signal) for switching the switching elements 12A to 12F of the inverter circuit 4 based on the corrected phase currents _iU ', iV', iW'.

This makes it possible to eliminate or suppress the influence of the ripple current and improve the controllability of the motor 8. In this case, the predicted ripple current calculation unit 26 predicts the ripple current contained in the phase current, and therefore, as in the above embodiment, by predicting the ripple current based on the current conduction pattern (the time t _ion from the start (0) of the carrier cycle until the maximum phase i (U phase) rises, and the time t _jon until the middle phase j (V phase) rises), a predetermined reference point (the start of the carrier cycle in this embodiment) which becomes the ideal current, the detection timings t _ad1 and t _ad2 of the phase current detection unit 24, and the amount of change (slope) of the ripple current per unit time, it becomes possible to accurately predict the ripple current and correct the detected current with high precision.

In addition, since the amount of change in ripple current per unit time varies for each line voltage pattern within one carrier cycle, in this embodiment, the predicted ripple current calculation unit 26 calculates the amount of change (slope) in ripple current per unit time in accordance with the line voltage patterns Pt1 to Pt3 within one carrier cycle.

Furthermore, since the amount of change in the ripple current varies depending on the HV voltage V _HV (DC voltage), the phase voltages (V _i , V _j , V _k ), and the inductances (L _i , L _j , L _k ) of the circuit including the motor 8, in this embodiment the predicted ripple current calculation unit 26 calculates the amount of change (slope) in the ripple current per unit time from these values. This makes it possible to predict and calculate the ripple current with high accuracy.

The present invention is extremely effective in the case of so-called one-shunt current detection, in which the current detection element 16 is connected to the DC bus 15 as in the embodiment, and the phase current detection unit 24 detects the phase currents _iU , _iV , _iW of the motor 8 based on the signal generated by the current detection element 16 and the current conduction pattern (PWM signal).

In the above-described embodiment, the beginning of the carrier cycle is used as the reference point for the ideal current, and the ripple current is predicted based on the time from the beginning to the detection timing of the phase current detection unit 24. However, this is not limiting. For example, in the case of a control method in which the center of the carrier cycle can be considered the ideal current, the center may be used as the reference point.

Furthermore, although the ripple current is predicted using a formula in the embodiment, it may also be predicted using a data table. In this case, a position sensor such as a Hall IC or rotary encoder is disposed on the rotor of the motor 8 to detect the rotor position. A data table of ripple currents set corresponding to the HV voltage _VHV (DC voltage), phase voltages ( _VU , _VV , VW or _Vi , _Vj , _{Vk )} , and rotor position of the motor 8 is written in advance to the predicted ripple current calculation unit 26, and the ripple current is predicted by extracting the ripple current from the data table based on the HV voltage _VHV (DC voltage), phase voltages ( _VU , _VV , _VW or _Vi , _Vj , _Vk ), and the rotor position detected by the position sensor.

In addition, in the embodiment, the saliency of the interior permanent magnet rotor is taken into account for the inductance of each phase ( _Li , _Lj , _Lk ), but this is not limiting, and the inductance may be set to a fixed value L and used in the ripple prediction calculation.

Furthermore, although the embodiments have been described in terms of so-called one-shunt current detection, the present invention is not limited to this, and is also effective in the case of so-called three-shunt current detection in which current detection elements are inserted into each of the U, V, and W phases for detection, for example, when the current detection timing is shifted.

Furthermore, in the embodiment, the present invention is applied to a motor control device 1 that controls the drive of the motor 8 of an electric compressor, but is not limited to this and is effective for controlling the drive of motors in various types of equipment.

1 Motor control device 4 Inverter circuit 7 HV battery (DC power supply)
8 Motor 12A to 12F Switching elements 15 DC bus 16 Current detection element 21 Control unit 22 Phase current correction unit 24 Phase current detection unit 26 Predicted ripple current calculation unit

Claims (3)

A motor control device includes an inverter circuit configured with a plurality of switching elements, and a control unit that generates a current pattern for switching the switching elements, and converts a DC voltage into an AC voltage by switching the switching elements according to the current pattern,
a current detection element connected to a DC side of the inverter circuit and generating a signal corresponding to a current value;
a phase current detection unit that detects a phase current of the motor based on a signal generated by the current detection element;
a predicted ripple current calculation unit that predicts a ripple current included in the phase current detected by the phase current detection unit;
a phase current correction unit that corrects the phase current detected by the phase current detection unit based on the ripple current predicted by the predicted ripple current calculation unit;
The predicted ripple current calculation unit calculates a change in ripple current per unit time from the DC voltage, the phase voltage, and the inductance of a circuit including the motor, and predicts the ripple current based on the current conduction pattern, a predetermined reference point that becomes an ideal current, the detection timing of the phase current detection unit, and the calculated change in ripple current per unit time, and
The motor control device is characterized in that the control unit generates the current conduction pattern based on the phase current corrected by the phase current correction unit. 2. The motor control device according to claim 1, wherein the predicted ripple current calculation unit calculates the amount of change in the ripple current per unit time in accordance with a pattern of a line voltage within a carrier cycle. the current detection element is connected to a DC bus;
3. The motor control device according to claim 1, wherein the phase current detection unit detects the phase current of the motor based on the signal generated by the current detection element and the current conduction pattern.