JP7743337B2 - Power Conversion Device - Google Patents

Description

The present invention relates to a power conversion device.

A known prior art technology is a non-isolated uninterruptible power supply, a power conversion device that switches between two-phase modulation and three-phase modulation to perform PWM control.

For example, Patent Document 1 discloses a power conversion device that performs PWM control using a two-phase modulation method that turns off the switching elements of the inverter circuit, and then performs PWM control using a three-phase modulation method while the two-phase modulation phases are being switched. This reduces distortion in the output AC waveform when the two-phase modulation phases are being switched.

Japanese Patent Application Publication No. 09-149660

In a power conversion device configured with a built-in DC power supply (storage battery) to continue power supply in the event of a power outage or momentary interruption, the input filter, output filter, and DC filter of the DC circuit may vibrate, causing a large amount of ripple current to flow through the DC filter. As in the configuration of Patent Document 1, performing PWM control using three-phase modulation while switching the phases of two-phase modulation can suppress the current flowing through the DC filter. However, because the value of the ripple current flowing through the DC filter varies depending on the duration of three-phase modulation (hereinafter also referred to as the three-phase modulation control period), it is necessary to optimize the duration of three-phase modulation.

The present invention therefore provides a power conversion device that optimizes the three-phase modulation control period during phase switching of two-phase modulation.

A power conversion device according to one aspect of this embodiment includes:
a converter circuit connected to a three-phase AC power source and converting AC from the three-phase AC power source into DC;
a smoothing circuit connected to a DC output terminal of the converter circuit and including a capacitor;
a DC circuit connected to the DC output terminal of the converter circuit and connected in parallel with the smoothing circuit, the DC circuit comprising a DC power supply;
an inverter circuit connected to a DC output terminal of the smoothing circuit and a DC output terminal of the DC circuit, for converting the DC from the smoothing circuit or the DC from the DC circuit into AC and outputting the AC to an AC load;
a converter control unit that PWM-controls the converter circuit using a two-phase modulation method or a three-phase modulation method;
an inverter control unit that PWM controls the inverter circuit using a two-phase modulation method or a three-phase modulation method;
Equipped with
the converter circuit and the inverter circuit are configured as three-phase full-bridge circuits made up of switching elements,
the converter circuit has an input filter;
the DC circuit includes a DC filter;
the inverter circuit has an output filter;
a 180° cycle of a control period of the converter circuit and the inverter circuit includes a two-phase modulation control period in which PWM control is performed by a two-phase modulation method and a three-phase modulation control period in which PWM control is performed by a three-phase modulation method;
A first resonance period is defined as a resonance period of the input filter, the DC filter, and the output filter;
When the second resonance period is defined as the resonance period of the DC filter and the output filter,
The three-phase modulation control period is set to be 3/8 or more and 5/8 or less of the first resonance period, or 3/8 or more and 5/8 or less of the second resonance period.

According to the present invention, the three-phase modulation control period during the phase switching of two-phase modulation can be optimized, thereby suppressing fluctuations in the value of the ripple current flowing through the DC filter.

1 is a circuit diagram showing a circuit configuration of a power conversion device according to an embodiment of the present invention. FIG. 2 is a block diagram showing a circuit configuration of a converter control unit according to the present embodiment. FIG. 2 is a block diagram showing a circuit configuration of an inverter control unit according to the present embodiment. 5A and 5B are diagrams illustrating internal signal waveforms of an inverter control unit according to the present embodiment. FIG. 2 is a diagram showing a waveform obtained by adding up the output voltages of the phases of the inverter circuit according to the present embodiment. 1 is an equivalent circuit for harmonic frequencies during normal operation according to the present embodiment. This is an equivalent circuit obtained by rearranging the equivalent circuit in FIG. 1 is an equivalent circuit for harmonic frequencies during battery operation according to this embodiment. This is an equivalent circuit obtained by rearranging the equivalent circuit in FIG. 8. When the three-phase modulation control period is set to 0, this is the waveform of the ripple current flowing through the DC filter during normal operation. This is a waveform of a ripple current flowing through a DC filter in normal operation when the three-phase modulation control period is set to 1/2 of the first resonance period of the filter circuit. FIG. 10 is a diagram showing the magnitude of a ripple current during normal operation with respect to a three-phase modulation control period.

This embodiment will now be described with reference to the drawings. For the sake of convenience, explanations of components with the same reference numbers as components already explained in the description of the embodiment will be omitted. Furthermore, for the sake of convenience, the dimensions of each component shown in the drawings may differ from the actual dimensions of each component.

Figure 1 is a diagram showing the circuit configuration of a power conversion device 1 according to this embodiment. As shown in Figure 1, a three-phase AC power supply 91 is connected to device input terminals 31a-31c of the power conversion device 1, and an AC load 92, to which three-phase AC output from an inverter circuit I is supplied, is connected to device output terminals 71a-71c of the power conversion device 1. The power conversion device 1 includes an input filter F1, a converter circuit C, a smoothing circuit S, an inverter circuit I, an output filter F2, a DC filter F3, a DC circuit D, a converter control unit 12, and an inverter control unit 13.

The input filter F1 has input filter capacitors 2a-2c and input filter reactors 3a-3c. One terminal of each of the input filter reactors 3a-3c is connected to the device input terminals 31a-31c of the power conversion device 1. The other terminals of each of the input filter reactors 3a-3c are connected to the three-phase (r-phase, s-phase, t-phase) AC input terminals 41a-41c of the converter circuit C, respectively.

One terminal of input filter capacitors 2a to 2c is connected to the terminal of input filter reactors 3a to 3c on the device input terminals 31a to 31c side, respectively. The other terminal of input filter capacitors 2a to 2c is connected to neutral conductor n1, respectively. Input filter F1 passes three-phase AC from three-phase AC power supply 91 to converter circuit C, while preventing signals at the carrier frequency (the frequency that determines the pulse width modulation period in the PWM control system) generated by converter circuit C from flowing into three-phase AC power supply 91.

Converter circuit C has six semiconductor switching elements 4a-4f connected in a three-phase bridge. The semiconductor switching elements 4a-4f are composed of, for example, IGBTs (insulated gate bipolar transistors) and anti-parallel freewheeling diodes. Converter circuit C is composed of a three-phase (r-phase, s-phase, t-phase) full-bridge circuit. Converter circuit C converts the three-phase AC input from a three-phase AC power source 91 into DC.

The smoothing circuit S includes a smoothing capacitor 5 of the converter circuit C. The smoothing capacitor 5 is, for example, an electrolytic capacitor, and is connected to the DC output terminals 42a and 42b of the converter circuit C. The smoothing capacitor 5 smoothes the output of the converter circuit C.

Inverter circuit I has six semiconductor switching elements 6a-6f connected in a three-phase bridge. Semiconductor switching elements 6a-6f are composed, for example, of IGBTs and anti-parallel freewheeling diodes. Inverter circuit I is composed of a three-phase (u-phase, v-phase, w-phase) full-bridge circuit. DC input terminals 61a and 61b of inverter circuit I are connected to both pole terminals of smoothing capacitor 5 and one terminal of each DC filter reactor 9a and 9b, respectively. Through the switching operation of semiconductor switching elements 6a-6f, inverter circuit I converts DC from smoothing circuit S (DC output by converter circuit C) to AC, or converts DC from DC circuit D to AC.

The output filter F2 has output filter reactors 7a-7c and output filter capacitors 8a-8c. One terminal of each of the output filter reactors 7a-7c is connected to the AC output terminals 62a-62c of the three-phase (u-phase, v-phase, w-phase) inverter circuit I. The other terminal of each of the output filter reactors 7a-7c is connected to the device output terminals 71a-71c of the power conversion device 1, respectively.

One terminal of output filter capacitors 8a to 8c is connected to the terminal of output filter reactors 7a to 7c on the device output terminals 71a to 71c side, respectively. The other terminal of output filter capacitors 8a to 8c is connected to neutral conductor n1, respectively. Output filter F2 passes the AC output from inverter circuit I to AC load 92, while preventing carrier frequency signals generated by inverter circuit I from flowing into AC load 92.

The DC filter F3 has DC filter reactors 9a and 9b and DC filter capacitors 10a and 10b. The DC circuit D has a DC power supply (storage battery) 11. One terminal of each of the DC filter reactors 9a and 9b is connected to the DC input terminals 61a and 61b of the inverter circuit I, respectively. The other terminals of the DC filter reactors 9a and 9b are connected to the DC filter capacitors 10a and 10b and the DC power supply 11, respectively.

DC circuit D is connected in parallel with smoothing circuit S and is configured to continue supplying power to AC load 92 in the event of a power outage or momentary interruption. Specifically, during normal operation of power conversion device 1, DC from smoothing circuit S (DC output by converter circuit C) is input to inverter circuit I. During battery operation of power conversion device 1, DC from DC circuit D is input to inverter circuit I.

Figure 2 shows the circuit configuration of the converter control unit 12 according to this embodiment. The converter control unit 12 includes a converter modulation command generation unit 121, a first two-phase modulation control unit 122, a first PWM signal generation unit 123, and a two-phase modulation signal generation unit 124, and controls the switching of the semiconductor switching elements 4a to 4f of the converter circuit C to the desired state.

The converter modulation command generation unit 121 receives three-phase AC from the three-phase AC power supply 91, generates converter modulation command signals Vr, Vs, and Vt consisting of three-phase AC sine waves, and outputs them to the first two-phase modulation control unit 122. The two-phase modulation signal generation unit 124 receives three-phase AC from the three-phase AC power supply 91, generates converter two-phase modulation signals AC1 to AC6, and outputs them to the first two-phase modulation control unit 122. The first two-phase modulation control unit 122 receives the converter modulation command signals Vr, Vs, and Vt from the converter modulation command generation unit 121 and the converter two-phase modulation signals AC1 to AC6 from the two-phase modulation signal generation unit 124. The first two-phase modulation control unit 122 further generates converter two-phase modulation command signals Vr', Vs', and Vt' and outputs them to the first PWM signal generation unit 123.

The first PWM signal generator 123 has a comparator 123a and a carrier signal generator 123b. The comparator 123a receives the converter two-phase modulation command signals Vr', Vs', and Vt' and the carrier signal Vc, such as a triangular wave, generated by the carrier signal generator 123b, compares them, and outputs PWM drive signals Vgr, Vgs, and Vgt. The PWM drive signal Vgr and its inverted signal /Vgr, the PWM drive signal Vgs and its inverted signal /Vgs, and the PWM drive signal Vgt and its inverted signal /Vgt are each input to the gates of the semiconductor switching elements 4a to 4f of the converter circuit C. In other words, the PWM drive signals Vgr, Vgs, and Vgt have waveforms that turn on the upper semiconductor switching elements 4a, 4c, and 4e and turn off the lower semiconductor switching elements 4b, 4d, and 4f when the converter two-phase modulation command signals Vr', Vs', and Vt' are greater than the carrier signal Vc. Furthermore, when the converter two-phase modulation command signals Vr', Vs', and Vt' are smaller than the carrier signal Vc, the PWM drive signals Vgr, Vgs, and Vgt have waveforms that turn on the lower semiconductor switching elements 4b, 4d, and 4f and turn off the upper semiconductor switching elements 4a, 4c, and 4e.

Figure 3 shows the circuit configuration of the inverter control unit 13 according to this embodiment. The inverter control unit 13 includes an inverter modulation command generation unit 131, a second two-phase modulation control unit 132, a second PWM signal generation unit 133, a two-phase modulation signal generation unit 134, and an inverter sine wave reference generation unit 135, and controls the switching of the semiconductor switching elements 6a-6f of the inverter circuit I to the desired state.

The inverter modulation command generation unit 131 receives the ideal sine wave generated by the inverter sine wave reference generation unit 135, generates inverter modulation command signals Vu, Vv, and Vw consisting of three-phase AC sine waves, and outputs them to the second two-phase modulation control unit 132. The two-phase modulation signal generation unit 134 receives the ideal sine wave from the inverter sine wave reference generation unit 135, generates inverter two-phase modulation signals AI1 to AI6, and outputs them to the second two-phase modulation control unit 132. The second two-phase modulation control unit 132 receives the inverter modulation command signals Vu, Vv, and Vw from the inverter modulation command generation unit 131 and the inverter two-phase modulation signals AI1 to AI6 from the two-phase modulation signal generation unit 134. The second two-phase modulation control unit 132 generates two-phase modulation command signals Vu', Vv', and Vw' and outputs them to the second PWM signal generation unit 133.

The configuration of the second PWM signal generation unit 133 is similar to that of the first PWM signal generation unit 123 of the converter control unit 12 shown in Figure 2, so a detailed description will be omitted.

Figure 4 shows internal signal waveforms of the inverter control unit 13 according to this embodiment. In Figure 4, the upper row shows inverter modulation command signals Vu, Vv, and Vw, the middle row shows inverter two-phase modulation signals AI1 to AI6, and the lower row shows inverter two-phase modulation command signals Vu', Vv', and Vw'.

The inverter modulation command signals Vu, Vv, and Vw are three-phase AC sine waves that are 120° out of phase with one another. The inverter two-phase modulation signals AI1 to AI6 are signals for two-phase modulating the inverter modulation command signals Vu, Vv, and Vw. The inverter two-phase modulation command signals Vu', Vv', and Vw' are signals that are partially two-phase modulated from the inverter modulation command signals Vu, Vv, and Vw based on the inverter two-phase modulation signals AI1 to AI6.
Specifically, the inverter two-phase modulation command signal Vu' is a signal in which the inverter modulation command signal Vu is fixed to High during the two-phase modulation control period t1 in which the inverter two-phase modulation signal AI1 is High. Also, the inverter two-phase modulation command signal Vu' is a signal in which the inverter modulation command signal Vu is fixed to Low during the two-phase modulation control period t1 in which the inverter two-phase modulation signal AI4 is High.
Similarly, the inverter two-phase modulation command signal Vv' is a signal in which the inverter modulation command signal Vv is fixed to High during the two-phase modulation control period t2 in which the inverter two-phase modulation signal AI2 is High. Also, the inverter two-phase modulation command signal Vv' is a signal in which the inverter modulation command signal Vv is fixed to Low during the two-phase modulation control period t2 in which the inverter two-phase modulation signal AI5 is High.
Similarly, the inverter two-phase modulation command signal Vw' is a signal in which the inverter modulation command signal Vw is fixed to High during the two-phase modulation control period t2 in which the inverter two-phase modulation signal AI3 is High. Also, the inverter two-phase modulation command signal Vw' is a signal in which the inverter modulation command signal Vw is fixed to Low during the two-phase modulation control period t2 in which the inverter two-phase modulation signal AI5 is High.

As shown in FIG. 4, the inverter two-phase modulation command signals Vu', Vv', and Vw' are two-phase modulation signals in which one of the three phases is fixed to either High or Low during two-phase modulation control periods t1, t2, and t3, and the remaining two phases are modulated. The inverter two-phase modulation command signals Vu', Vv', and Vw' are three-phase modulation signals in which all three phases are modulated during three-phase modulation control periods tr1, tr2, and tr3. The three-phase modulation control period tr1 is between the two-phase modulation control period t1 and the two-phase modulation control period t3, the three-phase modulation control period tr2 is between the two-phase modulation control period t2 and the two-phase modulation control period t1, and the three-phase modulation control period tr3 is between the two-phase modulation control period t3 and the two-phase modulation control period t2.

Furthermore, the two-phase modulation control periods t1, t2, and t3 are times corresponding to a phase of 60°-Δθ, and the three-phase modulation control periods tr1, tr2, and tr3 are times corresponding to a phase of Δθ. Therefore, during the control periods with a phase cycle of 180°, the inverter control unit 13 performs PWM control using the two-phase modulation method during the two-phase modulation control periods t1, t2, and t3, and performs PWM control using the three-phase modulation method during the three-phase modulation control periods tr1, tr2, and tr3.

In this embodiment, the three-phase modulation control periods tr1, tr2, and tr3 are each set based on the first resonance period T1 or the second resonance period T2 described below. The two-phase modulation control periods t1, t2, and t3 are each set to a value obtained by subtracting the above-set three-phase modulation control periods tr1, tr2, and tr3 from the time corresponding to a phase of 60°.

The configuration in which the inverter control unit 13 according to this embodiment PWM controls the inverter circuit I using two-phase modulation or three-phase modulation has been described using Figure 4. The converter control unit 12 according to this embodiment may also be configured to similarly PWM control the converter circuit C using two-phase modulation or three-phase modulation.

Figure 5 shows the waveform obtained by adding up the output voltages of each phase of inverter circuit I according to this embodiment. As shown in Figure 5, the fundamental waves (waveforms of commercial components) of each phase of inverter circuit I are summed to 0V. As a result, the voltage during the three-phase modulation control periods tr1, tr2, and tr3 is 0V, and the voltage during the two-phase modulation control periods t1, t2, and t3 remains as a high or low waveform. Furthermore, the waveform obtained by adding up the output voltages of each phase of inverter circuit I fluctuates significantly when transitioning from the two-phase modulation control periods t1, t2, and t3 to the three-phase modulation control periods tr1, tr2, and tr3, or when transitioning from the three-phase modulation control periods tr1, tr2, and tr3 to the two-phase modulation control periods t1, t2, and t3.

Fig. 6 shows an equivalent circuit for harmonic frequencies during normal operation when power is supplied from a three-phase AC power supply 91 to an AC load 92. As shown in Fig. 6, the equivalent circuit is made up of input filter capacitors 2a to 2c and input filter reactors 3a to 3c of an input filter F1, output filter capacitors 8a to 8c and output filter reactors 7a to 7c of an output filter F2, and DC filter capacitors 10a and 10b and DC filter reactors 9a and 9b of a DC filter F3.
In Fig. 6, symbol C' represents the waveform obtained by adding up the output voltages of the phases of converter circuit C, and symbol I' represents the waveform obtained by adding up the output voltages of the phases of inverter circuit I (the waveform shown in Fig. 5). When the fluctuating voltage waveform shown in Fig. 5 is applied to the equivalent circuit shown in Fig. 6, a harmonic current flows.

Figure 7 shows an equivalent circuit diagram that combines waveform C', which is the sum of the output voltages of the phases of converter circuit C, and waveform I', which is the sum of the output voltages of the phases of inverter circuit I, in the equivalent circuit diagram of Figure 6. Note that waveform C', which is the sum of the output voltages of the phases of converter circuit C, and waveform I', which is the sum of the output voltages of the phases of converter circuit C, are nearly identical voltage waveforms. Because the inductances L3a-L3c of input filter reactors 3a-3c of input filter F1 and the inductances L7a-L7c of output filter reactors 7a-7c of output filter F2 are typically the same value, Lac1 can be combined as L3a/6. Similarly, because the capacitances C2a-C2c of input filter capacitors 2a-2c of F1 and the capacitances C8a-C8c of output filter capacitors 8a-8c of output filter F2 are the same value, Cac1 can be combined as C2a x 6. For the DC filter F3, the filter values on the P-side and N-side are the same, so the inductances L9a and L9b of the DC filter reactors 9a and 9b can be combined as Ldc1 = L9a/2, and the capacitances C10a and C10b of the DC filter capacitors 10a and 10b of the DC filter F3 can be combined as Cdc1 = C10a × 2.
A harmonic current flows when a voltage waveform C' obtained by adding up the output voltages of the phases of the converter circuit C or a voltage waveform I' obtained by adding up the output voltages of the phases of the inverter circuit I is applied to the equivalent circuit shown in Fig. 7. The magnitude of the harmonic current differs depending on the three-phase modulation control periods tr1, tr2, and tr3.

7 is defined as the resonance period of the input filter F1, the DC filter F3, and the output filter F2. Specifically, the first resonance period T1 is defined by the following equation:

Fig. 8 shows an equivalent circuit for harmonic frequencies during battery operation when power is supplied from DC circuit D to AC load 92. As shown in Fig. 8, the equivalent circuit is made up of output filter capacitors 8a to 8c and output filter reactors 7a to 7c of output filter F2, and DC filter reactors 9a and 9b and DC filter capacitors 10a and 10b of DC filter F3.
When a waveform obtained by adding up the output voltages of the phases of the inverter circuit I shown in FIG. 5 is applied, a harmonic current flows in the equivalent circuit.

Figure 9 shows an equivalent circuit that is a rearrangement of the equivalent circuit in Figure 8. Because the inductances L7a to L7c of output filter reactors 7a to 7c of output filter F2 are typically the same value, they can be combined as Lac2 = L7a/3. Similarly, because the capacitances C8a to C8c of output filter capacitors 8a to 8c of F2 are the same value, they can be combined as Cac2 = C8a x 3. For DC filter F3, the filter values on the positive and negative sides are the same, so the inductances L9a and L9b of DC filter reactors 9a and 9b are Ldc2 = L9a/2, and the capacitances C10a and C10b of DC filter capacitors 10a and 10b of DC filter F3 can be combined as Cdc2 = C10a x 2.
When a voltage waveform I' obtained by adding up the output voltages of the phases of the inverter circuit I is applied to the equivalent circuit shown in Fig. 9, a harmonic current flows. The magnitude of the harmonic current differs depending on the three-phase modulation control periods tr1, tr2, and tr3.

9 is defined as the resonance period of the DC filter F3 and the output filter F2. Specifically, the second resonance period T2 is defined by the following equation.

Figure 10 shows, for comparison with this embodiment, the ripple current waveform flowing through the DC filter F3 during normal operation when the three-phase modulation control periods tr1, tr2, and tr3 are set to 0. Because the three-phase modulation control periods tr1, tr2, and tr3 are set to 0, the power conversion device 1 does not perform PWM control using the three-phase modulation method, but only performs PWM control using the two-phase modulation method.

Figure 11 shows the ripple current waveform flowing through the DC filter F3 during normal operation when the three-phase modulation control periods tr1, tr2, and tr3 are set to 1/2 the first resonant period T1 of the filter circuit in this embodiment.

Comparing Figures 10 and 11, it can be seen that when PWM control using three-phase modulation is performed while switching the phases of two-phase modulation, the ripple current flowing through DC filter F3 during phase switching of two-phase modulation can be reduced to less than half compared to when only PWM control using two-phase modulation is performed.

Figure 12 shows the magnitude of ripple current during normal operation versus the three-phase modulation control periods tr1, tr2, and tr3. As shown in Figure 12, by setting the three-phase modulation control periods tr1, tr2, and tr3 to be greater than or equal to 3/8 and less than or equal to 5/8 of the first resonant period T1 of the filter circuit, respectively, it is possible to suppress the ripple current flowing through the DC filter F3 while the phases of the two-phase modulation are switched. Furthermore, by setting the three-phase modulation control periods tr1, tr2, and tr3 to be 1/2 of the first resonant period T1 of the filter circuit, it is possible to most effectively suppress the ripple current flowing through the DC filter F3 while the phases of the two-phase modulation are switched.

Figures 10 to 12 illustrate the ripple current characteristics flowing through the DC filter F3 during phase switching of two-phase modulation during normal operation. Similar ripple current characteristics are observed during battery operation. In other words, performing PWM control using three-phase modulation during phase switching of two-phase modulation is more effective at suppressing the ripple current flowing through the DC filter F3 than performing PWM control using only two-phase modulation. Furthermore, by setting the three-phase modulation control periods tr1, tr2, and tr3 to be greater than or equal to 3/8 and less than or equal to 5/8 of the second resonant period T2 of the filter circuit, the ripple current flowing through the DC filter F3 during phase switching of two-phase modulation can be suppressed. Furthermore, by setting the three-phase modulation control periods tr1, tr2, and tr3 to be 1/2 of the second resonant period T2 of the filter circuit, the ripple current flowing through the DC filter F3 during phase switching of two-phase modulation can be most effectively suppressed.

The present embodiment has been described above, but it goes without saying that the technical scope of this embodiment should not be interpreted as being limited by the description of this embodiment. This embodiment is merely an example, and it will be understood by those skilled in the art that various modifications of the embodiment are possible within the scope of the invention described in the claims. The technical scope of this embodiment should be determined based on the scope of the invention described in the claims and its equivalents.

1: Power conversion devices 2a to 2c: Input filter capacitors 3a to 3c: Input filter reactors 4a to 4f: Semiconductor switching elements 5: Smoothing capacitors 6a to 6f: Semiconductor switching elements 7a to 7c: Output filter reactors 8a to 8c: Output filter capacitors 9a, 9b: DC filter reactors 10a, 10b: DC filter capacitor 11: DC power supply (storage battery)
12: Converter control unit 13: Inverter control unit 121: Converter modulation command generation unit 131: Inverter modulation command generation unit 122: First two-phase modulation control unit 132: Second two-phase modulation control unit 123: First PWM signal generation unit 133: Second PWM signal generation unit 123a, 133a: Comparators 123b, 133b: Carrier signal generation units 124, 134: Two-phase modulation signal generation unit 135: Inverter sine wave reference generation units 31a to 31c: Device input terminals 41a to 41c 1c: AC input terminals 42a to 42b: DC output terminals 61a to 61b: DC input terminals 62a to 62c: AC output terminals 71a to 71c: device output terminal 91: three-phase AC power supply 92: AC load C: converter circuit I: inverter circuit D: DC circuit F1: input filter F2: output filter F3: DC filter Lac1, Ldc1, Lac2, Ldc2: combined inductance Cac1, Cdc1, Cac2, Cdc2: combined capacitance n1: neutral wire S: smoothing circuit

Claims (3)

a converter circuit connected to a three-phase AC power source and converting AC from the three-phase AC power source into DC;
a smoothing circuit connected to a DC output terminal of the converter circuit and including a capacitor;
a DC circuit connected to the DC output terminal of the converter circuit and connected in parallel with the smoothing circuit, the DC circuit comprising a DC power supply;
an inverter circuit connected to a DC output terminal of the smoothing circuit and a DC output terminal of the DC circuit, for converting the DC from the smoothing circuit or the DC from the DC circuit into AC and outputting the AC to an AC load;
a converter control unit that performs PWM control of the converter circuit using two-phase modulation or three-phase modulation; and an inverter control unit that performs PWM control of the inverter circuit using two-phase modulation or three-phase modulation.
Equipped with
the converter circuit and the inverter circuit are configured as three-phase full-bridge circuits made up of switching elements,
the converter circuit has an input filter;
the DC circuit includes a DC filter;
the inverter circuit has an output filter;
a 180° cycle of a control period of the converter circuit and the inverter circuit includes a two-phase modulation control period in which PWM control is performed by a two-phase modulation method and a three-phase modulation control period in which PWM control is performed by a three-phase modulation method;
A first resonance period is defined as a resonance period of the input filter, the DC filter, and the output filter;
When the second resonance period is defined as the resonance period of the DC filter and the output filter,
the three-phase modulation control period is set to 3/8 or more and 5/8 or less of a first resonance period, or 3/8 or more and 5/8 or less of a second resonance period;
Power conversion device. The power conversion device of claim 1, wherein the input filter, the DC filter, and the output filter each consist of a filter capacitor and a filter reactor. The power conversion device of claim 1 or 2, wherein the three-phase modulation control period is greater than or equal to 3/8 and less than or equal to 5/8 of the first resonant period when the three-phase AC power supply is input to the converter circuit, and is greater than or equal to 3/8 and less than or equal to 5/8 of the second resonant period when the three-phase AC power supply is not input to the converter circuit.