JP7757738B2 - Control method for electric vehicle, control device for electric vehicle - Google Patents

Description

The present invention relates to a control method for an electric vehicle and a control device for an electric vehicle.

Patent Document 1 discloses the application of vibration suppression control to a field-winding synchronous motor in order to reduce torsional vibration in the drive shaft connecting the motor and the drive wheels. In Patent Document 1, vibration is suppressed by controlling the q-axis current of the motor.

Japanese Patent Application Laid-Open No. 2000-270589

However, when the motor is stopped, for example, and the f-axis current applied to the rotor winding (field winding) is set to a small value, it is difficult to sufficiently reduce vibration simply by controlling the q-axis current.

The present invention aims to provide a control method for an electric vehicle and a control device for an electric vehicle that can reduce vibration in the driving force transmission system even when the f-axis current applied to the rotor winding is set to a small value.

The control method for an electric vehicle according to the present invention uses a field-winding synchronous motor having a rotor with a rotor winding and a stator with a stator winding as the drive motor, sets a motor torque command value based on vehicle information, calculates a first vertical axis current command value which is a vertical axis command value of the orthogonal component of the stator current, a first quadrature axis current command value which is a horizontal axis command value of the stator current, and a first f-axis current command value which is a rotor current command value based on the motor torque command value and the vehicle information, controls the stator current using the first vertical axis current command value and the first quadrature axis current command value, and controls the rotor current using the first f-axis current command value. This control method calculates a second f-axis current command value that is the same as the first f-axis current command value when there is no disturbance to the driving force transmission system of the electric vehicle, but that is higher than the first f-axis current command value when the absolute value of the motor torque command value is equal to or less than a predetermined value and a disturbance is detected. It also estimates a magnetic flux estimate based on the first vertical axis current command value and the second f-axis current command value. It also calculates a first torque command value based on the first quadrature axis current command value and the magnetic flux estimate. It also calculates a vibration-damping torque command value by performing a calculation process on the first torque command value to suppress vibrations in the driving force transmission system. It also calculates a second quadrature axis current command value based on the magnetic flux estimate and the vibration-damping torque command value. The stator current is then controlled based on the second quadrature axis current command value and the first vertical axis current command value, and the rotor current is controlled based on the second f-axis current command value.

According to the present invention, even when the f-axis current applied to the rotor winding is set to a small value, the f-axis current is increased in response to a disturbance, thereby making it possible to sufficiently suppress vibrations in the driving force transmission system that may occur due to the disturbance.

FIG. 1 is a diagram showing an example of the basic configuration of an electric vehicle equipped with a control device for an electric vehicle according to this embodiment. FIG. 2 is a flowchart of the motor current control performed by the control device of this embodiment. FIG. 3 is a diagram showing an example of an accelerator opening-torque table. FIG. 4 is a block diagram showing a part of the control device for the electric vehicle according to this embodiment. FIG. 5 is a block diagram of the q-axis current control unit. FIG. 6 is a block diagram of the d-axis current control unit. FIG. 7 is a block diagram of the f-axis current control unit. FIG. 8 is a block diagram of the vibration suppression control unit of this embodiment. FIG. 9 shows a control flow of the vibration damping control unit of this embodiment. FIG. 10 is a block diagram of a flux estimator. FIG. 11 is a block diagram of the vibration damping torque command value calculation unit. FIG. 12 is a diagram showing a model of the drive train system of an electric vehicle. FIG. 13 is a diagram showing the frequency characteristics of the band-pass filter H(s). FIG. 14 is a time chart showing the response of the drive shaft torque, drive motor rotation speed, q-axis current, d-axis current, and f-axis current when the parking lock is released, and compares the cases where control by the control device for an electric vehicle of this embodiment is not executed with the case where it is executed.

[System configuration of electric vehicle]
1 is a block diagram showing the system configuration of an electric vehicle 1 equipped with a control device (control method) for an electric vehicle 1 according to this embodiment. The electric vehicle 1 is a vehicle that is equipped with a motor 6 (drive motor) as part or all of the vehicle's drive source and can run using the driving force of the motor 6, and includes electric vehicles and hybrid vehicles.

Signals indicating vehicle conditions such as vehicle speed V, accelerator opening, electrical angle θ _re of motor 6, three-phase AC current (iu, iv, iw) of the stator of motor 6, and f-axis current if of the _rotor of motor 6 are input as digital signals to motor controller 2 (control device). Based on the input signals (vehicle information), motor controller 2 generates a PWM signal for controlling motor 6. Based on the generated PWM signal, it also generates a drive signal for inverter 3 via a drive circuit.

The inverter 3 has two pairs of switching elements (e.g., power semiconductor elements such as IGBTs and MOS-FETs) for each phase to control the current in the stator of the motor 6. By turning the switching elements on and off, it converts the direct current supplied from the battery 4 into alternating current and passes the desired current through the stator windings of the motor 6.

The inverter 3 also has two pairs of switching elements on both ends of the rotor winding for controlling the rotor current of the motor 6, and by turning the switching elements on and off in response to a drive signal, the desired current flows from the battery 4 to the rotor winding. However, if the current flowing to the rotor is in only one direction, the two diagonally positioned switching elements of the two pairs of switching elements may be replaced with diodes.

The motor 6 (three-phase AC motor) generates driving force using the AC current supplied by the inverter 3, and transmits the driving force to the left and right drive wheels 81, 82 via the reducer 7 and drive shaft 8. Furthermore, when the motor 6 rotates in conjunction with the drive wheels 81, 82 while the vehicle is running, it generates regenerative driving force, thereby recovering the vehicle's kinetic energy as electrical energy. In this case, the inverter 3 converts the AC current generated during regenerative operation of the motor 6 into DC current and supplies it to the battery 4.

The current sensor 5 detects three-phase AC currents (iu, iv, iw) flowing through the stator of the motor 6 and an f-axis current i/ _f flowing through the rotor of the motor 6. However, since the sum of the three-phase AC currents (iu, iv, iw) is zero, the currents of any two phases may be detected and the current of the remaining phase may be calculated.

The rotation sensor 9 is, for example, a resolver or an encoder, and detects the electrical angle θ _re of the rotor of the motor 6 .

[System-wide control flow]
2 is a flowchart of the motor current control performed by the control device (motor controller 2) of this embodiment. The processes from step S201 to step S204 are executed at regular intervals while the vehicle system is running.

In step S201, a signal indicating the vehicle state (vehicle information) is input to the motor controller 2. Here, the vehicle speed V (km/h), accelerator opening (%), and electrical angle _θre of the motor 6 are input. Also input are the rotation speed Nm (rpm) of the motor 6, the three-phase AC current (iu, iv, iw) flowing through the stator of the motor 6, the f-axis current if flowing through the rotor of the motor 6, the DC voltage value Vdc (V) of the battery 4, a brake signal, and a park lock release signal (shift information, a signal indicating that the shift lever of the electric vehicle 1 is in a position other than park).

Vehicle speed V (km/h) is obtained via communication from a vehicle speed sensor (not shown) or another controller.

The accelerator opening (%) is obtained from an accelerator opening sensor (not shown) or via communication from another controller, such as a vehicle controller (not shown).

The electrical angle θ _re (rad) of the motor 6 is obtained from the rotation sensor 9. The electrical angular velocity ω _re is determined by differentiating the electrical angle θ _re .

The rotation speed Nm [rpm] of the motor 6 is calculated by dividing the electrical angular velocity _ωre by the number of poles of the electric motor to obtain the mechanical angular velocity _ωrm [rad/s], which is the mechanical angular velocity of the electric motor, and then multiplying this by a unit conversion coefficient (60/2π) from [rad/s] to [rpm].

The three-phase AC current (iu, iv, iw) [A] flowing through the stator of the motor 6 and the f-axis _current if [A] flowing through the rotor of the motor 6 are acquired from the current sensor 5 .

The DC voltage value V _dc [V] is detected by a voltage sensor (not shown) provided on the DC power supply line between the battery 4 and the inverter 3. Note that the DC voltage value V _dc [V] may also be detected from a signal related to the power supply voltage value transmitted from a battery controller (not shown).

The brake signal is detected as a hardware signal or obtained via communication from another controller such as a shift controller.

The park lock release signal is obtained via communication from the shift position signal, the park lock release switch signal on the shift knob, and other controllers such as the shift controller.

In step S202, the motor controller 2 (motor torque command value setting means) sets a motor torque command value Tm ^* as a basic target torque. Specifically, the motor controller 2 sets the motor torque command value Tm ^* by referring to the accelerator opening-torque table shown in Figure 3 based on the accelerator opening and vehicle speed V input in step S201. However, the accelerator opening-torque table is an example and is not limited to the one shown in Figure 3.

In step S203, the motor controller 2 (more specifically, the vibration suppression control unit 211 (FIGS. 4 and 8)) performs vibration suppression control calculation processing. Specifically, the motor torque command value Tm ^* set in step S202, the mechanical angular velocity _ωrm , etc. are input, and a second q-axis current command value i q2 * (second quadrature-axis current command value), a first d-axis current command value i d1 * (first vertical-axis current command value), and a second f-axis current command value i _f2 ^* (described later) are calculated to suppress vibrations (such as torsional vibrations of the drive shaft 8) in the driving force transmission system (a mechanism that transmits driving force from the motor 6 to the drive _shaft ⁸ (drive wheels 81, ₈₂ )) without sacrificing the torque response of the drive shaft ^8. Details of the vibration suppression control calculation processing will be described later (FIG. 9).

In step S204, current control calculation processing is performed by the motor controller 2 (FIG. 4). Specifically, PWM_Duty drive signals (Duu ^{*, Dul*} , Dvu*, _Dvl *, Dwu ^* , Dwl* ⁾ for the stator switching elements of the _inverter 3 and PWM_Duty drive signals ( _Dfu ^* , _Dfl ^* ) for the rotor switching elements of the inverter 3 are generated based on the second q-axis current command value _iq2 ^* , the first d _{- axis} current command ^value id1 ^* _{, and} the ^second _f - ^axis _current command value if2 ^* .

[Motor Controller 2]
4 is a block diagram showing a part of the control device (motor controller 2) of the electric vehicle 1 of this embodiment. The motor controller 2 of this embodiment executes the vibration damping control calculation process and the current control calculation process.

The motor controller 2 of this embodiment includes a stator PWM converter 201, a rotor PWM converter 202, a look-ahead compensation unit 203, a first coordinate converter 204, a decoupling control unit 205, a q-axis current control unit 206 (current control means), a d-axis current control unit 207 (current control means), an f-axis current control unit 208 (current control means), a voltage command value calculation unit 209, a second coordinate converter 210, and a vibration suppression control unit 211.

The stator PWM converter 201 generates PWM_Duty drive signals ( _Duu ^* , _Dul ^* , Dvu ^*, _Dvl *, _Dwu *, _Dwl ^{* )} for the switching elements for the stator of the inverter 3 based on the three ^- phase voltage command values ( _vu ^* , _vv ^{* ,} _{vw *} ).

The rotor PWM converter 202 generates PWM_Duty drive signals (D _fu ^* , D _fl ^* ) for the rotor switching elements of the inverter 3 based on the f-axis voltage command value (v _f ^* ).

The look-ahead compensation unit 203 receives the electrical angle θ _re and the electrical angular velocity ω _re , and calculates a look-ahead compensated electrical angle θ _{re ′} by adding the product of the electrical angular velocity ω _re and the dead time of the control system to the electrical angle θ re.

The first coordinate converter 204 (three-phase/dq-axis coordinate converter) converts a three-phase AC coordinate system (u-axis, v-axis, w-axis) into a two-axis DC coordinate system (d-axis (e.g., vertical axis) and q-axis (e.g., horizontal axis)) that are mutually orthogonal and rotate _at an electrical angular velocity ωre. Specifically, the u-phase current iu, the v-phase current iv, and the electrical angle _θre are input, and the d-axis current _id and the q-axis current _iq are calculated using the following equation (1):

The decoupling control unit 205 receives the electrical angular velocity _ωre and the outputs of the current control unit, namely, the d-axis current reference response i _{d — ref} , the q-axis current reference response i _{q — ref} , the f-axis current reference response i _{f — ref} , the differential value s·i d _— ref of the d-axis current reference response, and the differential value s·i _{f — ref} of the f-axis current reference response, and calculates decoupling voltages (v _{d — dcpl} , v _{q — dcpl , v f — dcpl ) required to cancel out the interference voltages between the d-axis, q-axis} , and _f- axis. The voltage equation of the wound-field synchronous motor is given by the following mathematical expression (2).

however,
: f-axis current vd: d-axis voltage vq: q-axis voltage vf: f-axis voltage Ld: d-axis inductance Lq: q-axis inductance Lf: f-axis inductance Mf: mutual inductance between stator and rotor Ld': d-axis dynamic inductance Lq': q-axis dynamic inductance Lf': f-axis dynamic inductance Mf': dynamic mutual inductance between stator and rotor Ra: stator winding resistance Rf: rotor winding resistance _ωre : electrical angular velocity Furthermore, s is the Laplace operator. Here, if the decoupling control unit 205 functions ideally, the voltage equation of Equation (2) can be diagonalized as shown in the following Equation (3).

That is, the characteristics from the voltage to the current of the d-axis, q-axis, and f-axis have a time lag as shown in the following equations (4), (5), and (6), respectively.

The q-axis current control unit 206, the d-axis current control unit 207, and the f-axis current control unit 208 respectively cause the measured (actual) q-axis current ^iq , d-axis current id, and f-axis _current if to follow the second q-axis current command value _iq2 ^* , the first d-axis current command value _{id1 *} , and the second f-axis current command ^value _{if2 *} with the desired responsiveness without any steady-state deviation.

<q-axis current control unit 206>
5 is a block diagram of the q-axis current control unit 206. The q-axis current control unit 206 includes a first-order lag responder 2061, a subtractor 2062, a proportional gain 2063, an integral gain 2064, an integrator 2065, and an adder 2066.

The second q-axis current command value _iq2 ^* is input to the first-order lag responder 2061. The first-order lag responder 2061 has a first-order lag transfer characteristic that simulates the response delay of the actual current with respect to the second q-axis current command value _iq2 ^* . Therefore, the first-order lag responder 2061 superimposes this transfer characteristic on the input second q-axis current command value _iq2 ^* and outputs the q-axis current reference response _{iq_ref} . Note that _τq is the q-axis current reference response time constant.

The subtractor 2062 calculates the difference between the second q-axis current command value i _q2 ^* and the q-axis current i _q (detected value), and outputs it to the proportional gain 2063 and the integral gain 2064 .

The proportional gain 2063 multiplies the difference by a gain K _pq and outputs the result to an adder 2066. The integral gain 2064 multiplies the difference by a gain K _iq and outputs the result to an integrator 2065. The integrator 2065 integrates the output of the integral gain 2064 and outputs the result to the adder 2066.

An adder 2066 adds the output of the proportional gain 2063 and the output of the integrator 2065 and outputs a first q-axis voltage command value _{vq_dsh} .

Here, the gains K _pq and K _iq are set as shown in the following formulas (7) and (8). This allows the transfer characteristics from the q-axis current command values (first q-axis current command value i _q1 ^* , second q-axis current command value i _q2 ^* ) to the q-axis current i _q to match the reference response of the following formula (9).

<d-axis current control unit 207>
6 is a block diagram of the d-axis current control unit 207. The d-axis current control unit 207 includes a first-order lag responder 2071, a subtractor 2072, a proportional gain 2073, an integral gain 2074, an integrator 2075, an adder 2076, and a differentiator 2077.

The first d-axis current command value i _d1 ^* is input to the first-order lag responder 2071. The first-order lag responder 2071 has a first-order lag transfer characteristic that simulates the response delay of the actual current with respect to the first d-axis current command value i _d1 ^* . Therefore, the first-order lag responder 2071 superimposes this transfer characteristic on the input first d-axis current command value i _d1 ^* and outputs the d-axis current reference response i _{d_ref} . Note that τ _d is the d-axis current reference response time constant.

The first d-axis current command value i _d1 ^* is input to the differentiator 2077. The differentiator 2077 has a transfer characteristic of a first-order delay time derivative that simulates the response delay of the actual current with respect to the first d-axis current command value i _d1 ^* . Therefore, the differentiator 2077 superimposes this transfer characteristic on the input first d-axis current command value i _d1 ^* and outputs a differential value s·i _{d_ref} of the d-axis current reference response i _{d_ref} .

The subtractor 2072 calculates the difference between the first d-axis current command value i _d1 ^* and the d-axis current i _d (detected value), and outputs it to the proportional gain 2073 and the integral gain 2074 .

The proportional gain 2073 multiplies the difference by a gain K _pd and outputs the result to an adder 2076. The integral gain 2074 multiplies the difference by a gain K _id and outputs the result to an integrator 2075. The integrator 2075 integrates the output of the integral gain 2074 and outputs the result to the adder 2076.

An adder 2076 adds the output of the proportional gain 2073 and the output of the integrator 2075 and outputs a first d-axis voltage command value v _{d — dsh} .

Here, the gains K _pd and K _id are set as shown in the following formulas (10) and (11). This allows the transfer characteristics from the d-axis current command values (first d-axis current command value id1*, second d-axis current command value _id2 ^* ) to the d-axis current _id to match the reference response of the following formula (12).

<f-axis current control unit 208>
7 is a block diagram of the f-axis current control unit 208. The f-axis current control unit 208 includes a first-order lag responder 2081, a subtractor 2082, a proportional gain 2083, an integral gain 2084, an integrator 2085, an adder 2086, and a differentiator 2087.

The second f-axis current command value i _f2 ^* is input to the first-order lag responder 2081. The first-order lag responder 2081 has a first-order lag transfer characteristic that simulates the response delay of the actual current with respect to the second f-axis current command value i _f2 ^* . Therefore, the first-order lag responder 2081 superimposes this transfer characteristic on the input second f-axis current command value i _f2 ^* and outputs the f-axis current reference response i _{f_ref} . Note that τ _f is the f-axis current reference response time constant.

The second f-axis current command value i _f2 ^* is input to the differentiator 2087. The differentiator 2087 has a transfer characteristic of a first-order time differential that simulates the response delay of the actual current with respect to the second f-axis current command value i _f2 ^* . Therefore, the differentiator 2087 superimposes this transfer characteristic on the input second f-axis current command value i _f2 ^* and outputs a differential value s·i _{f_ref} of the f-axis current reference response.

The subtractor 2082 calculates the difference between the second f-axis current command value i _f2 ^* and the f-axis current i _f (detected value), and outputs it to the proportional gain 2083 and the integral gain 2084 .

The proportional gain 2083 multiplies the difference by a gain K _pf and outputs the result to an adder 2086. The integral gain 2084 multiplies the difference by a gain K _if and outputs the result to an integrator 2085. The integrator 2085 integrates the output of the integral gain 2084 and outputs the result to the adder 2086.

An adder 2086 adds the output of the proportional gain 2083 and the output of the integrator 2085 and outputs a first f-axis voltage command value _{vf_dsh} .

Here, the gains K _pf and K _if are set as shown in the following formulas (13) and (14). This allows the transfer characteristics from the f-axis current command values (first f-axis current command value i _f1 ^* , second f-axis current command value i _f2 ^* ) to the f-axis current i _f to match the reference response of the following formula (15).

Let's return to Figure 4 and continue the explanation.

The voltage command value calculation unit 209 adds the first q-axis voltage command value _{vq_dsh,} which is the output of the q-axis current control unit 206, and the q-axis decoupling voltage _{vq_dpcl,} which is the output of the decoupling control unit 205, to generate a second q-axis voltage command value _vq ^* , and outputs this to the second coordinate converter 210.

In addition, the voltage command value calculation unit 209 adds the first d-axis voltage command value v _{d _dsh,} which is the output of the d-axis current control unit 207, and the d-axis decoupling voltage v _{d _dpcl,} which is the output of the decoupling control unit 205, to generate a second d-axis voltage command value v _d ^* , and outputs this to the second coordinate converter 210.

Furthermore, the voltage command value calculation unit 209 adds the first f-axis voltage command value v _{f — dsh,} which is the output of the f-axis current control unit 208, and the f-axis decoupling voltage v _{f — dpcl,} which is the output of the decoupling control unit 205, to generate a second f-axis voltage command value v _f ^* , and outputs this to the rotor PWM converter 202.

The second coordinate converter 210 (dq-axis/three-phase coordinate converter) converts a two-axis DC coordinate system (d-axis, q-axis) into a three-phase AC coordinate system (u-axis, v-axis, w-axis). Specifically, the second d-axis voltage command value v _d ^* , the second q-axis voltage command value v _q ^* , and the post-lookahead compensation electrical angle θ _re ′ are input, and the second d-axis voltage command value v _d ^* and the second q-axis voltage command value v _q ^* are subjected to coordinate conversion processing using the following equation (16), thereby calculating and outputting a u-phase voltage command value v _u ^* , a v-phase voltage command value v _v ^* , and a w-phase voltage command value v _w ^* .

The vibration suppression control unit 211 receives the motor torque command value _Tm ^* , mechanical angular velocity _ωrm , _DC voltage value Vdc, q-axis current _iq detected by current sensor 5, the parking lock release signal, the brake signal, and vehicle speed V, and calculates a second q-axis current command value _iq2 ^* , a first d-axis current command value _id1 ^* , and a second f-axis current command value _if2 ^* based on these values. Details of the vibration suppression control calculation process (Fig. 8) will be described later.

<Vibration control unit 211>
Fig. 8 is a block diagram of vibration suppression control unit 211. As shown in Fig. 8, vibration suppression control unit 211 includes a first current command value calculator 212, a second current command value calculator 213, a magnetic flux estimator 214, a first torque command value calculator 215, a vibration suppression torque command value calculator 216, and a second q-axis current command value calculator 217.

When the first current command value calculator 212 (first current command value calculation means) receives the motor torque command value T _m ^* , the motor rotation speed (mechanical angular velocity ω _rm ), and the DC voltage value V _dc as input, it calculates the first q-axis current command value i _q1 ^* , the first d-axis current command value i _d1 ^* , and the first f-axis current command value i _f1 ^* . The first q-axis current command value _iq1 ^* , the first d-axis current command value _id1 ^* , and the first f-axis current command value _if1 ^* can be determined by _pre -storing map data in memory that defines the relationship between the motor torque command value _Tm ^* , motor rotation speed (mechanical angular velocity _ωrm ), and _DC voltage value Vdc ^and the first q-axis current command value _iq1 ^* , the first d-axis current command value _{id1*, and the first f-axis current command value if1} ^* , and by referencing this map data.

The second current command value calculator 213 (second current command value calculation means) receives as input the motor torque command value T _m ^* , motor rotation speed (mechanical angular velocity ω _rm ), first f-axis current command value i _f1 ^* , q-axis current i _q (detected value), park lock release signal, brake signal, and vehicle speed V. Then, the second current value calculator 2112 generates a second f-axis current command value i _f2 ^* based on a predetermined control flow and outputs this to the magnetic flux estimator 214 and the f-axis current control unit 208 ( FIGS. 4 and 7 ). Details of the control flow will be described later ( FIG. 9 ).

The first d-axis current command value i _d1 ^* and the second f-axis current command value i _f2 ^* are input to a magnetic flux estimator 214 (magnetic flux estimation means). The magnetic flux estimator 214 then calculates a magnetic flux estimation value φ^ based on the input command values and outputs it to a first torque command value calculator 215 and a second q-axis current command value calculator 217. Details of the magnetic flux estimator 214 will be described later ( FIG. 10 ).

The first q-axis current command value _iq1 ^* and the magnetic flux estimation value φ^ are input to the first torque command value calculator 215 (torque command value calculation means). The first torque command value calculator 215 calculates the first torque command value _Tm1 ^* by multiplying the product of the first q-axis current command value _iq1 ^* and the magnetic flux estimation value φ^ by the number of pole pairs _pn , and outputs this to the vibration damping torque command value calculator 216.

The motor rotation speed (mechanical angular velocity ω _rm ) and first torque command value T _m1 ^* are input to vibration damping torque command value calculator 216 (vibration damping torque command value calculation means). Vibration damping torque command value calculator 216 generates final torque command value T _mfin ^* (vibration damping torque command value) based on the motor rotation speed (mechanical angular velocity ω _rm ) and first torque command value T _m1 ^* , and outputs this to second q-axis current command value calculator 217. Details of vibration damping torque command value calculator 216 will be described later ( FIG. 11 ).

The final torque command value T _mfin ^* and the magnetic flux estimation value φ^ are input to the second q-axis current command value calculator 217 (third current command value calculation means). The second q-axis current command value calculator 217 calculates a second q-axis current command value i _q2 ^* based on the following equation (17), and outputs it to the q-axis current control unit 206 (FIG. 4).

[Vibration suppression control calculation processing]
9 shows a control flow of the vibration suppression control unit 211 of this embodiment. The vibration suppression control unit 211 of this embodiment has a program installed therein for executing the control method of the present invention, and executes the control flow in accordance with this program.

Prior art (e.g., Patent Document 1) suppresses vibration by controlling the q-axis current _iq of a wound-field synchronous motor. Therefore, when the f-axis current i/ _f is zero or a very small value, the desired motor torque cannot be generated. Therefore, when the parking lock is released on a slope, as described below, if the f-axis current i/ _f is zero or a very small value, vibration cannot be suppressed.

Specifically, the vehicle is parked on a slope with the gear shifted to P and the foot brake or parking brake not applied. In this case, the park lock mechanism is activated by shifting to P, allowing the vehicle to remain parked even when the brake is released. However, because the drive wheels 81, 82 are not fixed, the drive shaft 8 twists a certain amount depending on the slope of the gradient to stop the vehicle. To start the vehicle from this stopped state, the parking brake must be applied to release the park lock. At this time, releasing the P range releases the park lock mechanism, releasing the torsion that had accumulated in the drive shaft 8, causing drive shaft torsional vibration (torsional vibration caused by a disturbance to the drive force transmission system), causing the occupants to feel an unexpected shock (hereinafter, this shock will be referred to as the park lock release shock).

Therefore, in this embodiment, as will be described below, control is executed that can sufficiently suppress torsional vibration of the drive shaft 8 due to disturbances even when the f-axis current i _f is set to a small value.

In the initial state (when there is no disturbance), the second f-axis current command value i _f2 ^* output by the second current command value calculator 213 is equal to the first f-axis current command value i _f1 ^* . As a result, in the initial state (when there is no disturbance), the second q-axis current command value i _q2 ^* output by the second q-axis current command value calculator 217 is equal to the first q-axis current value output by the first current command value calculator 212. Here, the first f-axis current command value i _f1 ^* is a command value set in response to a request from the electric vehicle side, and is set regardless of disturbance.

In step S301, the second current command value calculator 213 determines whether a disturbance has been detected from the range of fluctuations in the parking lock or the motor rotation speed; if YES, the process proceeds to step S302; if NO, the process proceeds to RETURN.

In step S302, the second current command value calculator 213 determines whether the motor torque command value T _m ^* is equal to or less than a predetermined value (for example, an absolute value of 1 [N·m]), and if YES, proceeds to step S303, and if NO, proceeds to RETURN.

In step S303, the second current command value calculator 213 determines whether or not the motor torque command value T _m ^* is zero. If YES, the process proceeds to step S304, and if NO, the process proceeds to step S305.

In step S304, when the second current command value calculator 213 detects a disturbance, it sets the second f-axis current command value _if2 ^* to a value obtained by offsetting the first f-axis current command value _if1 ^* by a predetermined value (a predetermined value that increases the absolute value). That is, when the second current command value calculator 213 detects a disturbance, it adds a predetermined value to the second f-axis current command value _if2 ^* to change it like a step function (steeply). In this way, when the motor torque command value _Tm ^* is zero, the second f-axis current command value _if2 ^* is immediately increased after the disturbance is detected, thereby quickly converging the shock caused by the disturbance.

In step S305, when the second current command value calculator 213 detects a disturbance, it offsets the first f-axis current command value i _f1 ^* by a predetermined value through low-pass filter (LPF) processing (a predetermined value that increases the absolute value) as the second f-axis current command value i _f2 ^* . That is, when the second current command value calculator 213 detects a disturbance, it increases the first f-axis current command value i f2 * based on the characteristics (first-order response) of the low-pass filter, as shown in Equation (18) below, and converges it to a value that is higher by a predetermined value (A) than the first f-axis current command value i _f1 ^* .

Here, the time constant (τ _fa ) is set so that the rise of the second f-axis current command value i _f2 ^* after detecting a disturbance is approximately the same as the rise of the second q-axis current command value i _q2 ^* .

If the motor torque command value T _m ^* is not zero, increasing the second f-axis current command value i _f2 ^* stepwise as in step S304 may cause fluctuations in motor torque, resulting in unexpected acceleration/deceleration or shocks. However, by increasing the second f-axis current command value i _f2 ^* according to the above equation (18), motor torque fluctuations can be suppressed.

Even if the motor torque command value T _m ^* is zero, when a disturbance is detected, it is possible to increase the second f-axis current command value i _f2 ^* according to formula (18).Furthermore, formula (18) may be approximated to a linear function, and the second f-axis current command value i _f2 ^* may be increased at a predetermined increase rate (rate limit).

In step S306, the second current command value calculator 213 determines whether a predetermined time (e.g., 1 second) has elapsed since the disturbance was detected. If the result is YES, the process proceeds to step S307. If the result is NO, the process proceeds to step S310. The predetermined time is a time set for determining that the shock (torsional vibration) caused by the disturbance has converged. At this time, the second current command value calculator 213 determines that the fluctuations in the rotation speed Nm of the motor 6 or the q-axis current _iq, which are affected by the disturbance, have converged within a predetermined threshold value after the predetermined time has elapsed. The predetermined time may be constant, or the second current command value calculator 213 may set the predetermined time based on the fluctuation range of the rotation speed Nm of the motor 6 caused by the disturbance (proportional to the time until the torsional vibration converges).

In step S307, the second current command value calculator 213 determines whether the motor torque command value T _m ^* exceeds a predetermined value (for example, an absolute value of 1 [N·m]) within the specified time. If the determination is YES, the process proceeds to step S310. If the determination is NO, the process proceeds to step S308. If the determination is YES in step S307, that is, if the motor torque command value T _m ^* is equal to or greater than the predetermined value even before the specified time has elapsed (the driver is operating the accelerator), the first f-axis current command value i _f1 ^* also has a predetermined magnitude, and the motor 6 is in a state where it can generate a predetermined torque. Therefore, it is possible to suppress shock to the driving force transmission system due to a disturbance without making the second f-axis current command value i _f2 ^* higher than the first f-axis current command value i _f1 ^* , and the process can proceed to the next flow before the specified time has elapsed.

In step S308, the second current command value calculator 213 determines whether the absolute value of the vehicle speed V of the electric vehicle 1 exceeds a predetermined value (e.g., 5 km/s). If the answer is YES, the process proceeds to step S310; if the answer is NO, the process proceeds to step S309. If the absolute value of the vehicle speed V exceeds the predetermined value even before the specified time has elapsed, it can be determined that the shock has subsided, and the process can proceed to the next flow before the specified time has elapsed.

In step S309, the second current command value calculator 213 determines whether the brake has transitioned from the ON state to the OFF state within the specified time. If the answer is YES, the process proceeds to step S310; if the answer is NO, the process proceeds to step S306. If the brake has transitioned from the ON state to the OFF state even before the specified time has elapsed, it can be determined that the shock caused by the disturbance has subsided, and the process can proceed to the next flow before the specified time has elapsed.

By executing the flow from step S307 to step S309, unnecessary power consumption can be reduced.

In step S310, the second current command value calculator 213 determines whether or not the motor torque command value T _m ^* is zero. If YES, the process proceeds to step S311, and if NO, the process proceeds to step S312.

In step S311, the second current command value calculator 213 makes the second f-axis current command value i _f2 ^* coincide with the first f-axis current command value i _f1 ^* in a step function manner, and then proceeds to RETURN.

In step S312, the second current command value calculator 213 reduces the second f-axis current command value i _f2 ^* based on the characteristics (first-order response) of the low-pass filter, as shown in equation (19) below, and transitions to RETURN, converging it to the first f-axis current command value i _f1 ^* .

Here, the time constant (τ _fb ) is an arbitrary time constant greater than zero.

When the motor torque command value T _m ^* is not zero, immediately matching the second f-axis current command value i _f2 ^* to the first f-axis current command value i _f1 ^* may cause fluctuations in motor torque, which may result in unexpected acceleration/deceleration or shock. Therefore, by providing a predetermined response characteristic, it is possible to prevent abrupt changes in motor torque and to converge shocks due to disturbances.

<Magnetic Flux Estimator 214>
10 is a block diagram of the magnetic flux estimator 214. The magnetic flux estimator 214 includes a transfer characteristic section 2141, a gain 2142, a transfer characteristic section 2143, a gain 2144, and an adder 2145. The transfer characteristic section 2141 and the gain 2142 configure a reluctance torque equivalent magnetic flux estimator, and the transfer characteristic section 2143 and the gain 2144 configure a field magnetic flux estimator.

The first d-axis current command value i _d1 ^* is input to the transfer characteristic unit 2141. The transfer characteristic unit 2141 has a transfer characteristic in which the response of the q-axis current i _q is advanced and compensated for with respect to a first-order lag transfer characteristic that simulates the response delay of the d-axis current i _d . The transfer characteristic unit 2141 superimposes the transfer characteristic on the input first d-axis current command value i _d1 ^* and outputs the result to the gain 2142.

A gain 2142 multiplies the output of the transfer characteristic section 2141 by the difference L _d −L _q between the d-axis inductance L _d and the q-axis inductance L _q to calculate a reluctance torque equivalent magnetic flux estimate value φ _r ^ and outputs it to an adder 2145 .

As described above, _Ld is the d-axis inductance and _Lq is the q-axis inductance, but values at a certain representative operating point may be used, or they may be found by referring to map data.

The reluctance torque generated by the d-axis current i _d and the q-axis current i _q is expressed as p _n (L _d - L _q ) i _d · i _q , so the term (L _d - L _q ) i _d is defined as the equivalent magnetic flux of the reluctance torque, where p _n is the number of pole pairs.

As described above, by using the reluctance torque equivalent magnetic flux estimate _φr ^ that has been compensated for the response lead of the q-axis current _iq , it is possible to calculate the second q-axis current command value _iq2 ^* that takes into account the response delay of the q-axis current _iq .

The second f-axis current command value i _f2 ^* is input to the transfer characteristic unit 2143. The transfer characteristic unit 2143 has a transfer characteristic in which the response of the q-axis current i _q is advanced and compensated for with respect to a first-order lag transfer characteristic that simulates the response delay of the f-axis current i _f . The transfer characteristic unit 2143 superimposes the transfer characteristic on the input second f-axis current command value i _f2 ^* and outputs the result to the gain 2144.

A gain 2144 calculates a field flux estimation value φ _f ^ by multiplying the output of the transfer characteristic section 2143 by the mutual inductance M _f between the stator and rotor, and outputs this to an adder 2145. Here, the mutual inductance M _f may be a value at a certain representative operating point, or may be found by referring to map data.

As described above, by using the field flux estimated value φ _f ^ that has been compensated for the response lead of the q-axis current _iq , it is possible to calculate the second q-axis current command value i _q2 ^* that takes into account the response delay of the q-axis current _iq .

The adder 2145 adds the reluctance torque equivalent magnetic flux estimate value _φr ^ and the field magnetic flux estimate value φf^ to generate a magnetic flux estimate value φ^, and outputs this to the first torque command value calculator 215 (Figure 8) and the second q-axis current command value calculator 217 (Figure 8).

<Vibration damping torque command value calculator 216>
11 is a block diagram of vibration suppression torque command value calculator 216. Vibration suppression torque command value calculator 216 is made up of a feedforward compensator 2161 and a feedback compensator (adder 2162, angular velocity estimator 2163, subtractor 2164, and transfer characteristic section 2165).

The first torque command value T _m1 ^* is input to the feedforward compensator 2161. The feedforward compensator 2161 has a transfer function ( _Gr (s)/Gp(s ₎ ) for suppressing torsional vibration of the drive shaft 8. The feedforward compensator 2161 superimposes the transfer function on the input first torque command value T _m1 ^* to generate a second torque command value T _m2 ^* , and outputs this to the adder 2162 and the angular velocity estimator 2163.

The adder 2162 adds the second torque command value T _m2 ^* and the third torque command value T _m3 ^* output from the transfer characteristic unit 2165 to generate a final torque command value T _mfin ^* , and outputs this to the second q-axis current command value calculator 217 (Figure 8).

The second torque command value T _m2 ^* is input to the angular velocity estimator 2163. The angular velocity estimator 2163 has a transfer function (G _p (s)) for estimating the angular velocity of the motor 6. The angular velocity estimator 2163 superimposes the transfer function on the input second torque command value T _m2 ^* to generate an angular velocity estimate value ω _rm ^ of the motor 6 and outputs this to the subtractor 2164.

The subtractor 2164 calculates the difference (ω _rm ^−ω _rm ) by subtracting the mechanical angular velocity ω _rm (measured value) of the motor 6 from the angular velocity estimate value ω _rm ^ of the motor 6 , and outputs this to the transfer characteristic section 2165 .

The transfer characteristic unit 2165 has an inverse characteristic filter (H(s)/G _p (s)) which is the product of the inverse characteristic of the transfer function (G _p (s)) of the angular velocity estimator 2163 and a band-pass filter H(s). Here, the transfer function (H(s)) is set so that the difference between the denominator order and the numerator order is equal to or greater than the difference between the denominator order and the numerator order of the transfer function (G _p (s)).

The transfer characteristic unit 2165 generates a third torque command value T _m3 ^* by superimposing an inverse characteristic filter on the difference (ω _rm ^-ω _rm ) input from the subtractor 2164 , and outputs this to the adder 2162 .

<Model of the driving force transmission system of an electric vehicle>
Fig. 12 is a diagram showing a model of the drive transmission system of the electric vehicle 1. Fig. 13 is a diagram showing the frequency characteristics of the bandpass filter H(s). A model of the electric vehicle 1 to which the present invention is applied will be described.

Based on FIG. 12, the equations of motion of the electric vehicle 1 are expressed by the following equations (20)-(24).

Here, the parameters are as follows:

J _m : Motor inertia J _w : Drive shaft inertia (for one shaft)
M: Mass of vehicle _Kd : Torsional rigidity of drive shaft _Kt : Coefficient of friction between tire and road surface _Nal : Overall gear ratio r: Tire load radius _ωm : Motor angular velocity _ωw : Drive wheel angular velocity _Tm : Motor torque _Td : Drive shaft torque F: Driving force (for two shafts)
V: Vehicle speed

(20)-(24) When the transfer characteristics from the torque command value T _m to the motor angular velocity ω _m are obtained by Laplace transform, the following formulas (25) and (26) are obtained.

However, each parameter is as shown in the following formula (27).

Equation (26) can be rearranged and expressed as the following equation (28): where ζ _p and ω _p are the damping coefficient and natural vibration frequency of the driven torsional vibration system.

If the reference model G _m (s) is expressed by the following formula (29), the linear filter G inv (s) (transfer function of the feedforward compensator 2161) having the inverse characteristic of the linearly approximated vehicle transfer characteristic G _p (s) and the characteristic G _m (s)/G _p (s) consisting of the reference model G _m (s) is expressed by the following formula (30).

where ζm and ωm are the damping coefficient and natural vibration frequency of the reference model.

Next, the transfer function H(s) will be described. When H(s) is configured as a band-pass filter, it becomes a feedback element that reduces only vibration. In this case, the greatest effect can be obtained by setting the filter characteristics as shown in FIG. 13. That is, the transfer function H(s) is set so that the attenuation characteristics on the low-pass side and the high-pass side are approximately the same and the torsional resonance frequency of the drivetrain is near the center of the passband on a logarithmic axis (log scale). For example, when H(s) is configured with a first-order high-pass filter and a first-order low-pass filter, it is configured as shown in Equation (31) below, where frequency _fp is the torsional resonance frequency of the drivetrain and k is an arbitrary value.

However, τ _L =1/(2πf _HC ), f _HC =k·f _p , τ _H =1/(2πf _LC ), and f _LC = f _p /k.

[Time chart]
Figure 14 is a time chart showing the response of the drive shaft torque, drive motor rotation speed, q-axis current _iq , d-axis current _id , and f-axis _current if when the parking lock is released, and compares the case where control is performed by the control device of the electric vehicle 1 of this embodiment with the conventional technology where this control is not performed.

Figure 14 compares the results when the control of this embodiment is executed and when it is not executed when the electric vehicle 1 is stopped on a slope using the parking lock, causing the drive shaft 8 to twist, and then the brake is applied and the parking lock is released, resulting in a parking lock release shock (disturbance).

Before time t1, the electric vehicle 1 is stopped, the shift is set to parking, and the parking lock signal is OFF. At this time, a torsional torque is applied to the drive shaft 8. Meanwhile, the d-axis current i _d (first d-axis current command value i _d1 ^* ), the q-axis current i _q (first q-axis current command value i _q1 ^* , second q-axis current command value i _q2 ^* ), and the f-axis current i _f (first f-axis current command value i _f1 ^* , second f-axis current command value i _f2 ^* ) are all zero.

A conventional technology that does not implement this embodiment corresponds to a case where the second current command value calculator 213 shown in Figure 8 is omitted, and the q-axis current _iq1 ^* is input to the q-axis current control unit ₂₀₆ shown in Figure 4, and the first f-axis current command value _if1 ^* is input to the f-axis current control unit 208, thereby controlling the q-axis current _iq , the d-axis current _id , and the f-axis current if.

In the case of conventional technology, at time t1, the brake is applied and the shift lever is moved from Park to a position other than Park. This causes the park lock signal to switch from OFF to ON. The torsion of the drive shaft 8 is released, causing a park lock release shock (torsional vibration) in the drive shaft 8. The drive shaft torque [N·m] and rotation speed [rpm] then each produce waveforms that oscillate around zero in accordance with the natural frequency of the torsional vibration of the drive shaft 8.

On the other hand, the q-axis current _iq (first q-axis current command value _iq1 ^* ), d-axis current _id (first d-axis current command value _id1 ^* ), and f-axis current _if (first f-axis current command value _ifl ^* ) remain zero even after time t1, so it is not possible to apply drive shaft torque (set to a value other than zero) to the drive shaft 8 via the motor 6 in order to reduce the park lock release shock.

Furthermore, even if the q-axis current _iq (first q-axis current command value _iq1 ^* ) is set to a value higher than zero at time t1, as long as the f-axis current _if (first f-axis current command value _if1 ^* ) is zero, it is not possible to apply drive shaft torque to the drive shaft 8 via the motor 6 to suppress the park lock release shock.

On the other hand, in this embodiment, a drive shaft torque is applied to the drive shaft 8 via the motor 6 by the q-axis current _iq (second q-axis current command value _iq2 ^* ), the d-axis current _id (first d-axis current command value _id1 ^* ), and the _f -axis current if (second f-axis current command value _if2 ^* ).

As shown in FIG. 14, the q-axis current i _q (second q-axis current command value i _q2 ^* ) and the f-axis current i _f (second f-axis current command value i _f2 ^* ) rise when the park lock signal is turned on.

As shown in Figures 8 and 11, the q-axis current _iq (second q-axis current command value _iq2 ^* ) is generated via vibration damping torque command value calculator 216 (feedback control using mechanical angular velocity _ωrm ) and has a waveform (e.g., an impulse response waveform with boosted high frequency) that offsets the park lock release shock. Furthermore, the f-axis current _if (second f-axis current command value _if2 ^* ) increases to rise with predetermined response characteristics at time t1 and converges to a value higher than the first f-axis current command value _ifl ^* by a predetermined value. The drive shaft torque applied to motor 6 and drive shaft 8 is approximately proportional to the product of q-axis current _iq and f-axis current _if . However, since q-axis current _iq and f-axis current _if are not zero immediately after time t1, it is possible to generate a drive shaft torque (to be set to a value other than zero). Therefore, as shown in FIG. 14, for example, the park lock release shock generated by the drive shaft torque and the rotational speed [rpm] of the motor 6 and the subsequent torsional vibration can be suppressed.

In addition, after a predetermined time has elapsed since the drive shaft torque, motor 6 rotation speed, and q-axis current _iq have all converged within predetermined threshold values, the f-axis current i _f (second f-axis current command value i _f2 ^* ) converges to the same value as the first f-axis current command value i _f1 ^* at time t2, thereby suppressing unnecessary power consumption.

14 illustrates a case where the parking lock of the electric vehicle 1 stopped on a slope is released, but the same applies to a case where, for example, the electric vehicle 1 is traveling at a low speed (when the f-axis current i _f is extremely small) and suddenly enters a steep slope, causing fluctuations in the rotation speed to exceed a predetermined value (step S301 in FIG. 9). In this case, the d-axis current i _d may also be a constant value other than zero.

In addition, in this embodiment, for example, the first d-axis current command value i _d1 ^* output from the first current command value calculator 212 may be input to the second current command value calculator 213, the second current command value calculator 213 may generate a second d-axis current command value i _d2 ^* and output it to the magnetic flux estimator 214, and the magnetic flux estimator 214 may generate a first torque command value T _m1 ^* based on the second d-axis current command value i _d2 ^* and the second f-axis current command value i _f2 *. In this case, when there is no disturbance, the second current command value calculator 213 sets the second d-axis current command value i _d2 ^* to the same value as the first d-axis current command value i _d1 ^* , but when a disturbance is detected, it may control the second d-axis current command value i _d2 ^* so that its absolute value is higher by a predetermined value than the first d-axis current command value i _d1 ^* within a specified time, as with the second f-axis current command value i _f2 ^* .

[Effects of this embodiment]
According to the control method for the electric vehicle 1 of this embodiment, a field-winding synchronous motor including a rotor having a rotor winding and a stator having a stator winding is used as the drive motor (motor 6), a motor _torque command value _Tm ^{* is} set based on vehicle information, a first vertical axis current command value (first d-axis current command value _id1 ^* ) which is a command value for the vertical axis (e.g., d-axis) of the orthogonal component of the stator current, a first quadrature axis current command value (first q-axis current command value _iq1 ^* ) which is a command value for the quadrature axis (e.g., q-axis) of the stator current, and a first f-axis current command value ifl ^* which is a command value for the rotor current are calculated based on the motor torque command value _Tm * and the vehicle information, and the stator current is controlled using the first vertical axis current command value (first d-axis current command value _id1 ^* ) and the first quadrature axis current command value (first q-axis current command value _iq1 ^* _). a first torque command value T m1 ^* based on the first q-axis current command value (first q-axis current command value i _q1 ^{* )} and _the magnetic flux estimated value φ^; and a damping torque command value (final torque command value _{T mfin} ^{* )} obtained _by performing a calculation process on the first torque ^command value _{T m1} ^* to suppress vibrations ₍ torsional ^{vibrations )} in ^the driving ^force _transmission system _. ) is calculated, a second quadrature axis current command value (second q-axis current command value i _q2 ^* ) is calculated based on the magnetic flux estimation value φ^ and the vibration damping torque command value (final torque command value T _mfin ^* ), and the stator current is controlled based on the second quadrature axis current command value (second q-axis current command value i _q2 ^* ) and the first vertical axis current command value (first d-axis current command value i _d1 ^* ), and the rotor current is controlled based on the second f-axis current command value i _f2 ^* .

With the above method, even when the f-axis current _if (first f-axis current command value _if1 ^* ) applied to the rotor winding is set small, the f-axis current _if (second f-axis current command value _if2 ^* ) is increased in response to a disturbance, thereby making it possible to sufficiently suppress vibrations (e.g., shocks and torsional vibrations when the parking lock is released) in the driving force transmission system (mechanism including the drive shaft 8) that may occur due to disturbances.

In this embodiment, when the second f-axis current command value i _f2 ^* before detecting a disturbance matches the first f-axis current command value i _f1 ^* and the absolute value of the motor torque command value T _m ^* when detecting a disturbance is zero, the second f-axis current command value i _f2 ^* is increased in a step function manner, thereby enabling vibrations caused by disturbances in the driving force transmission system (e.g., shocks and torsional vibrations when the parking lock is released) to be suppressed with high precision.

In this embodiment, when the second f-axis current command value i _f2 ^* before the disturbance is detected matches the first f-axis current command value i _f1 ^* , and the absolute value of the motor torque command value T _m ^* when the disturbance is detected is greater than zero and less than a predetermined value (e.g., 1 [N·m]), the second f-axis current command value i _f2 ^* is increased based on a predetermined response characteristic (e.g., equation (18)).

If the motor torque command value T _m ^* is not zero, increasing the second f-axis current command value i _f2 ^* in a step function manner may cause motor torque fluctuations. However, as in the above method, by increasing the second f-axis current command value i _f2 ^* in accordance with a predetermined response characteristic (e.g., equation (18)), motor torque fluctuations can be suppressed.

In this embodiment, disturbances are detected based on shift information that indicates the position of the shift lever of the electric vehicle 1. By detecting the movement of the shift lever operated by the driver from park to a position other than park as a disturbance, delays in control can be avoided.

In this embodiment, disturbances are detected based on fluctuations in the rotation speed of the drive motor (motor 6). This makes it possible to detect disturbances other than park lock release disturbances, such as disturbances caused by the electric vehicle 1 suddenly entering a steep slope.

In this embodiment, the vertical axis is the rotor's d-axis, and the horizontal axis is the rotor's q-axis. This allows the drive motor (motor 6) to more reliably generate torque to suppress vibration.

In this embodiment, the specified time during which the second f-axis current command value i _f2 ^* is higher than the first f-axis current command value i _f1 ^* is set based on the fluctuation range of the q-axis current i _q . This makes it possible to set the time (specified time) during which the second f-axis current command value i _f2 ^* is higher than the first f-axis current command value i _f1 ^* until vibrations occurring in the driving force transmission system are suppressed, thereby enabling more reliable vibration suppression.

In this embodiment, the specified time during which the second f-axis current command value i _f2 ^* becomes higher than the first f-axis current command value i _f1 ^* is set to a predetermined time, thereby making it possible to set the time during which the second f-axis current command value i _f2 ^* becomes higher than the first f-axis current command value i _f1 ^* with a simple configuration.

In this embodiment, the specified time during which the second f-axis current command value i _f2 ^* becomes higher than the first f-axis current command value i _f1 ^* is set based on the fluctuation range of the rotation speed of the drive motor (motor 6). This makes it possible to set the time (specified time) during which the second f-axis current command value i _f2 ^* becomes higher than the first f-axis current command value i _f1 ^* until vibrations occurring in the driving force transmission system are suppressed, thereby enabling more reliable vibration suppression.

In this embodiment, if the motor torque command value T _m ^* exceeds a predetermined value (e.g., 1 [N·m]) within a specified time, the second f-axis current command value i _f2 ^* is made to coincide with the first f-axis current command value i _f1 ^* in a step function manner regardless of the specified time, or the second f-axis current command value i _f2 ^* is made to converge to the first f-axis current command value i _f1 ^* via a predetermined response characteristic (e.g., Equation (19)). At this time, the first f-axis current command value i _f1 ^* also has a predetermined magnitude, and the electric motor (motor 6) is in a state where it can generate a predetermined torque. Therefore, shock to the driving force transmission system due to a disturbance can be suppressed without making the second f-axis current command value i _f2 ^* higher than the first f-axis current command value i _f1 ^* , and power consumption can be reduced accordingly.

In this embodiment, if the vehicle speed V of the electric vehicle 1 exceeds a predetermined value (for example, 5 m/s) within a specified time, the second f-axis current command value i _f2 ^* is made to coincide with the first f-axis current command value i _f1 ^* in a step function manner regardless of the specified time, or the second f-axis current command value i _f2 ^* is made to converge to the first f-axis current command value i _f1 ^* via predetermined response characteristics (for example, Equation (19)). As a result, if the absolute value of the vehicle speed V exceeds the predetermined value (for example, 5 m/s) even before the specified time has elapsed, it can be determined that the shock has subsided, and the second f-axis current command value i _f2 ^* can be made to coincide with the first f-axis current command value i _f1 ^* quickly before the specified time has elapsed, thereby reducing power consumption.

If the brake of the electric vehicle 1 changes from an on state to an off state within a specified time, the second f-axis current command value i _f2 ^* is made to coincide with the first f-axis current command value i _f1 ^* in a step function manner regardless of the specified time, or the second f-axis current command value i _f2 ^* is made to converge to the first f-axis current command value i _f1 ^* via a predetermined response characteristic (for example, formula (19)). As a result, if the brake transitions from an on state to an off state even before the specified time has elapsed, it can be determined that the shock due to the disturbance has subsided, and power consumption can be reduced by quickly making the second f-axis current command value i _f2 ^* coincide with the first f-axis current command value i _f1 ^* before the specified time has elapsed.

Furthermore, the control device (motor controller 2) for the electric vehicle 1 of this embodiment controls the electric vehicle 1, whose drive motor (motor 6) is a field-winding type synchronous motor including a rotor having a rotor winding and a stator having a stator winding, and includes motor torque command value setting means (motor controller 2 ⁾ that sets a motor _torque command value _Tm ^* based on vehicle information, first current command value calculation means (first current command value calculator 212) that calculates a first vertical axis current command value (first d-axis current command value _id1 ^* ) that is a command value for the vertical axis (e.g., d-axis) of the orthogonal component of the stator current based on the motor torque command value _Tm * and the vehicle information, a first quadrature axis current command value (first q-axis current command value _iq1 ^* ) that is a command value for the quadrature axis of the stator current, and a first f-axis current command value _ifl ^* that is a command value for the rotor current ^, and a current control unit (206, 207, 208) for controlling a stator current by a first q-axis current command value i _q1 ^* and a first quadrature-axis current command value i _{q1 *, and for controlling a rotor current by a first f-axis current command value i f1} ^* , the control unit (206, 207, 208) further comprising: a second current command value calculation unit ( ₂₁₃ ) for calculating a second f-axis current command value i _f2 * that is the same as the first f-axis current command value i _f1 ^* when there is no disturbance to the driving force transmission system of the electric vehicle (1 ⁾ , but that is higher than the first f-axis current command value i _f1 ^* when the absolute value of the motor torque command value _{T m} ^* is equal to or less than a predetermined ^value and when a disturbance is detected _; a torque command value calculator (first torque command value calculator 215) that calculates a first torque command value T m1 ^* based on the first quadrature axis current command value (first q-axis current command value i _q1 ^* ) and the magnetic flux estimated value φ^; an oscillation-damping torque command value calculator (oscillation-damping torque command value calculator 216 ⁾ that calculates a damping torque command value (final torque _{command value T mfin *} ⁾ by performing an arithmetic process on the first torque command value T _m1 ^* to suppress vibrations in the driving force transmission system; and a second quadrature axis current command value (second q-axis current command value i q2 *) that calculates a second quadrature axis current command value (second q-axis current command value i _q2 ^*) based on the magnetic flux estimated value φ^ and the damping torque command value (final torque command value T _mfin ^* ). and a third current command value calculation means (second q-axis current command value calculator 217) that calculates a second q-axis current command value i q2 * and a first y-axis current command value i d1 *, and current control means (q-axis current control unit 206, d-axis current control unit 207, f-axis current control unit 208) control the stator current based on the second q-axis current command value i _q2 ^* and the first y-axis current command value (first d-axis current command value i _d1 ^* ), and control the rotor current based on the second f-axis current command value i _f2 ^* .

With the above configuration, even when the f-axis current _if (first f-axis current command value _if1 ^* ) applied to the rotor winding is set small, the f-axis current _if (second f-axis current command value _if2 ^* ) is increased in response to a disturbance, thereby making it possible to sufficiently suppress vibrations (e.g., shocks and torsional vibrations when the parking lock is released) in the driving force transmission system (mechanism including the drive shaft 8) that may occur due to disturbances.

The above describes embodiments of the present invention, but these embodiments merely illustrate some of the application examples of the present invention, and are not intended to limit the technical scope of the present invention to the specific configurations of the above embodiments. Furthermore, the above embodiments can be combined as appropriate.

1. Electric vehicle, 2. Motor controller, 6. Motor, 206. q-axis current control unit, 207. d-axis current control unit, 208. f-axis current control unit, 212. First current command value calculator, 213. Second current command value calculator, 214. Magnetic flux estimator, 215. First torque command value calculator, 216. Damping torque command value calculator, 217. Second q-axis current command value calculator

Claims (13)

a first vertical-axis current command value that is a vertical-axis command value of an orthogonal component of a stator current, a first quadrature-axis current command value that is a quadrature-axis command value of the stator current, and a first f-axis current command value that is a rotor current command value, based on the motor torque command value and the vehicle information; and a first vertical-axis current command value that is a vertical-axis command value of an orthogonal component of a stator current, and a first f-axis current command value that is a rotor current command value, the stator current being controlled by the first vertical-axis current command value and the first quadrature-axis current command value, and the rotor current being controlled by the first f-axis current command value,
calculating a second f-axis current command value that is the same as the first f-axis current command value when there is no disturbance to a driving force transmission system of an electric vehicle, and that is higher than the first f-axis current command value when the absolute value of the motor torque command value is equal to or less than a predetermined value and the disturbance is detected;
estimating a magnetic flux estimation value based on the first vertical axis current command value and the second f-axis current command value;
calculating a first torque command value based on the first quadrature axis current command value and the magnetic flux estimation value;
calculating a vibration-damping torque command value by performing a calculation process on the first torque command value to suppress vibrations in the driving force transmission system;
calculating a second quadrature axis current command value based on the magnetic flux estimation value and the vibration suppression torque command value;
A control method for an electric vehicle, comprising: controlling the stator current based on the second quadrature-axis current command value and the first vertical-axis current command value; and controlling the rotor current based on the second f-axis current command value. A control method for an electric vehicle according to claim 1, wherein the second f-axis current command value is increased in a step function manner when the second f-axis current command value before the disturbance is detected matches the first f-axis current command value and the absolute value of the motor torque command value when the disturbance is detected is zero. The control method for an electric vehicle described in claim 1, wherein when the second f-axis current command value before the disturbance is detected matches the first f-axis current command value, and the absolute value of the motor torque command value when the disturbance is detected is greater than zero and less than or equal to the predetermined value, the second f-axis current command value is increased based on predetermined response characteristics. The control method for an electric vehicle according to any one of claims 1 to 3, wherein the disturbance is detected based on shift information representing the position of a shift lever of the electric vehicle. The control method for an electric vehicle according to any one of claims 1 to 3, wherein the disturbance is detected based on fluctuations in the rotation speed of the drive motor. The control method for an electric vehicle described in any one of claims 1 to 5, wherein the vertical axis is the d-axis of the rotor and the horizontal axis is the q-axis of the rotor. The control method for an electric vehicle described in claim 6, wherein the specified time during which the second f-axis current command value is higher than the first f-axis current command value is set based on the fluctuation range of the q-axis current. A control method for an electric vehicle according to any one of claims 1 to 5, wherein the specified time during which the second f-axis current command value becomes higher than the first f-axis current command value is set to a predetermined time. 5. The method for controlling an electric vehicle according to claim 1, wherein the specified time during which the second f-axis current command value becomes higher than the first f-axis current command value is set based on a fluctuation range of the rotation speed of the drive motor. A control method for an electric vehicle according to any one of claims 7 to 9, wherein, if the motor torque command value exceeds a predetermined value within the specified time, the second f-axis current command value is made to match the first f-axis current command value in a step function manner regardless of the specified time, or the second f-axis current command value is made to converge to the first f-axis current command value via a predetermined response characteristic. A control method for an electric vehicle according to any one of claims 7 to 9, wherein, if the vehicle speed of the electric vehicle exceeds a predetermined value within the specified time, the second f-axis current command value is made to coincide with the first f-axis current command value in a step function manner regardless of the specified time, or the second f-axis current command value is made to converge to the first f-axis current command value via a predetermined response characteristic. A control method for an electric vehicle according to any one of claims 7 to 9, wherein, if the brake of the electric vehicle changes from an on state to an off state within the specified time, the second f-axis current command value is made to coincide with the first f-axis current command value in a step function manner, regardless of the specified time, or the second f-axis current command value is made to converge to the first f-axis current command value via a predetermined response characteristic. The control target is an electric vehicle using a field-winding type synchronous motor as a drive motor, the field-winding type synchronous motor including a rotor having a rotor winding and a stator having a stator winding,
a motor torque command value setting means for setting a motor torque command value based on vehicle information;
a first current command value calculation means for calculating a first vertical axis current command value which is a vertical axis command value of an orthogonal component of a stator current, a first quadrature axis current command value which is a quadrature axis command value of the stator current, and a first f-axis current command value which is a rotor current command value, based on the motor torque command value and the vehicle information;
a current control unit that controls the stator current based on the first vertical axis current command value and the first quadrature axis current command value, and controls the rotor current based on the first f-axis current command value,
a second current command value calculation means for calculating a second f-axis current command value that is the same as the first f-axis current command value when there is no disturbance to a driving force transmission system of an electric vehicle, and that is higher than the first f-axis current command value when the absolute value of the motor torque command value is equal to or less than a predetermined value and the disturbance is detected;
a magnetic flux estimation means for estimating a magnetic flux estimated value based on the first vertical axis current command value and the second f-axis current command value;
a torque command value calculation means for calculating a first torque command value based on the first quadrature axis current command value and the magnetic flux estimation value;
a vibration-damping torque command value calculation means for calculating a vibration-damping torque command value by performing a calculation process on the first torque command value to suppress vibrations in the driving force transmission system;
a third current command value calculation means for calculating a second quadrature axis current command value based on the magnetic flux estimation value and the vibration suppression torque command value,
the current control means controls the stator current based on the second quadrature-axis current command value and the first vertical-axis current command value, and controls the rotor current based on the second f-axis current command value.