JP7789587B2 - radar equipment - Google Patents

Description

This disclosure relates to a radar device.

In recent years, research has been progressing on radar devices that use short-wavelength radar transmission signals, including microwaves and millimeter waves, which provide high resolution. Furthermore, to improve outdoor safety, there is a demand for the development of radar devices (wide-angle radar devices) that can detect small objects, such as pedestrians and fallen objects, in addition to vehicles, over a wide angle range.

One type of radar device with a wide detection range uses an array antenna consisting of multiple antennas (antenna elements) to receive reflected waves, and estimates the angle of arrival (direction of arrival) of the reflected waves using a signal processing algorithm based on the received phase difference relative to the element spacing (antenna spacing) (Direction of Arrival (DOA) estimation). Examples of angle of arrival estimation methods include the Fourier method, or methods that provide high resolution include the Capon method, MUSIC (Multiple Signal Classification), and ESPRIT (Estimation of Signal Parameters via Rotational Invariance Techniques).

Furthermore, radar devices have been proposed that have multiple antennas (array antennas) in the transmitting section as well as the receiving section, and perform beam scanning through signal processing using the transmitting and receiving array antennas (sometimes called MIMO (Multiple Input Multiple Output) radar) (see, for example, Non-Patent Document 1).

Japanese Patent Application Laid-Open No. 2008-304417 Special Publication No. 2011-526371 JP 2014-119344 A International Publication No. 2019/054504

J. Li, and P. Stoica, "MIMO Radar with Colocated Antennas", Signal Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007 M. Kronauge, H. Rohling, "Fast two-dimensional CFAR procedure", IEEE Trans. Aerosp. Electron. Syst., 2013, 49, (3), pp. 1817-1823 Direction-of-arrival estimation using signal subspace modeling Cadzow, J.A.; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28 , Issue: 1 Publication Year: 1992 , Page(s): 64 - 79

However, methods for detecting targets in radar devices (e.g., MIMO radar) have not been fully studied.

Non-limiting examples of the present disclosure contribute to providing a radar device that can detect targets with high accuracy.

A radar device according to one embodiment of the present disclosure includes a transmission circuit that outputs, for each transmission period, a first transmission signal having a first center frequency and a second transmission signal having a second center frequency that is a center frequency higher than the first center frequency, and a transmission antenna that transmits the first transmission signal and the second transmission signal, wherein the second center frequency is a frequency that is higher than (1+1/ _Nc ) times the first center frequency ( _Nc is an integer indicating the number of times that each of the first transmission signal and the second transmission signal is transmitted for each transmission period within a predetermined period).

Note that these comprehensive or specific embodiments may be realized as a system, device, method, integrated circuit, computer program, or recording medium, or as any combination of a system, device, method, integrated circuit, computer program, and recording medium.

According to one embodiment of the present disclosure, a radar device can detect targets with high accuracy.

Further advantages and benefits of one embodiment of the present disclosure will become apparent from the specification and drawings. While such advantages and/or benefits may be provided by some of the embodiments and features described in the specification and drawings, not all of them necessarily need to be provided to obtain one or more identical features.

FIG. 1 shows an example of non-uniform Doppler multiplex transmission. Block diagram showing an example of the configuration of a radar device A diagram showing an example of a chirp signal FIG. 1 is a diagram showing an example of a radar transmission signal. A diagram showing an example of a Doppler peak A diagram showing an example of a Doppler peak A diagram showing an example of a Doppler peak A diagram showing an example of a Doppler peak FIG. 10 is a diagram showing an example of Doppler determination processing; Another example of a chirp signal FIG. 1 is a diagram showing an example of the configuration of a radar device. FIG. 1 is a diagram showing an example of a radar transmission signal. FIG. 1 shows an example of Doppler multiplexing. FIG. 1 shows an example of Doppler multiplexing. A diagram showing an example of a chirp signal FIG. 1 shows an example of Doppler multiplexing. A diagram showing an example of the configuration of a radar receiver. FIG. 1 is a diagram showing an example of the configuration of a radar device. FIG. 1 is a diagram showing an example of the configuration of a radar device. FIG. 1 is a diagram showing an example of the configuration of a radar device. A block diagram showing another example of the configuration of a radar device. FIG. 10 is a diagram showing another example of a radar transmission signal.

MIMO radar transmits multiplexed signals (radar transmission waves) using, for example, time division, frequency division, or code division from multiple transmitting antennas (also called transmitting array antennas), receives signals (radar reflection waves) reflected by surrounding objects using multiple receiving antennas (also called receiving array antennas), and separates and receives the multiplexed transmission signals from each received signal. Through this processing, MIMO radar can extract the propagation path response expressed as the product of the number of transmitting antennas and the number of receiving antennas, and performs array signal processing on these received signals as a virtual receiving array.

In addition, in MIMO radar, by appropriately spacing the elements in the transmitting and receiving array antennas, it is possible to virtually expand the antenna aperture and improve angular resolution.

[Time division multiplexing transmission]
For example, Patent Document 1 discloses a MIMO radar (hereinafter referred to as "time-division multiplexing MIMO radar") that uses time-division multiplexing, which transmits signals by shifting the transmission time for each transmitting antenna, as a multiplexing method for MIMO radar. Time-division multiplexing can be implemented with a simpler configuration than frequency-multiplexing or code-multiplexing. Furthermore, time-division multiplexing can maintain good orthogonality between transmitted signals by sufficiently widening the transmission time interval. Time-division multiplexing MIMO radar outputs transmission pulses, which are an example of transmission signals, while sequentially switching between transmitting antennas at a predetermined cycle. Time-division multiplexing MIMO radar receives signals that are transmitted as reflected pulses from an object using multiple receiving antennas, and performs correlation processing between the received signals and the transmitted pulses, followed by, for example, spatial Fast Fourier Transform (FFT) processing (processing to estimate the direction of arrival of the reflected waves).

Time-division multiplexed MIMO radar sequentially switches the transmitting antennas that transmit transmit signals (e.g., transmit pulses or radar transmit waves) at a predetermined cycle. Therefore, compared to frequency-division or code-division transmission, time-division multiplexed transmission can require a longer time to complete transmission of transmit signals from all transmit antennas. For this reason, when transmitting transmit signals from each transmit antenna and detecting the Doppler frequency (e.g., the relative velocity of a target) from the received phase change, as in Patent Document 2, for example, the time interval (e.g., sampling interval) for observing the received phase change becomes longer when applying Fourier frequency analysis to detect the Doppler frequency. This reduces the maximum Doppler frequency range based on the sampling theorem (e.g., the Doppler frequency range that can be detected without aliasing, or the range of detectable relative velocity of a target).

Furthermore, when a reflected wave signal from a target with a Doppler frequency exceeding the maximum Doppler frequency based on the sampling theorem is expected to be received, the radar device may observe a Doppler frequency that is an aliased component that differs from the true frequency. In this case, it is difficult for the radar device to determine whether the reflected wave signal is an aliased component, resulting in ambiguity in the Doppler frequency (e.g., the relative velocity of the target).

For example, if a radar device transmits a transmission signal (transmission pulse) by sequentially switching Nt transmitting antennas at a predetermined period _Tr , the transmission time until all transmitting antennas have transmitted the transmission signal is _Tr × Nt. If such time-division multiplexed transmission is repeated _Nc times and Fourier frequency analysis is applied to detect the Doppler frequency (detection of relative velocity), the Doppler frequency range in which the Doppler frequency can be detected without aliasing is ±1/( _2Tr × Nt) according to the sampling theorem. Therefore, the Doppler frequency range in which the Doppler frequency can be detected without aliasing decreases as the number of transmitting antennas Nt increases, and Doppler frequency ambiguity is more likely to occur even at slower relative velocities.

[Doppler multiplexing]
Since time division multiplexing MIMO radar may have the above-mentioned Doppler frequency ambiguity, the following will focus on, as an example, a method of simultaneously multiplexing and transmitting transmission signals from multiple transmission antennas.

One method for simultaneously multiplexing and transmitting transmission signals from multiple transmitting antennas is to transmit signals in such a way that the receiving unit can separate the multiple transmission signals on the Doppler frequency axis (hereinafter referred to as Doppler multiplexing) (see, for example, Non-Patent Document 3).

In Doppler multiplexing, the transmitter applies a Doppler shift greater than the Doppler frequency bandwidth of the received signal to the transmit signal transmitted from a transmit antenna other than the reference transmit antenna, for example, relative to the transmit signal transmitted from the reference transmit antenna, and the transmit signals are transmitted from multiple transmit antennas in the same transmission cycle (same transmit slot). In Doppler multiplexing, the receiver separates and receives the transmit signals transmitted from each transmit antenna by filtering on the Doppler frequency axis.

In Doppler multiplexing, signals are transmitted from multiple transmitting antennas at the same transmission period, which shortens the time interval for observing received phase changes when applying Fourier frequency analysis to detect Doppler frequency (or relative velocity) compared to time division multiplexing. However, in Doppler multiplexing, the transmitted signals from each transmitting antenna are separated by filtering on the Doppler frequency axis, which limits the effective Doppler frequency bandwidth per transmitted signal.

For example, in Doppler multiplexing, a radar device will be described that transmits transmission signals from Nt transmitting antennas at a period _Tr . If such Doppler multiplexing is repeated _Nc times within a predetermined period and Fourier frequency analysis is applied to detect the Doppler frequency (or relative velocity), the Doppler frequency range in which the Doppler frequency can be detected without aliasing is ±1/(2× _Tr ) according to the sampling theorem. For example, the Doppler frequency range in which the Doppler frequency can be detected without aliasing in Doppler multiplexing is expanded by Nt times compared to the case of time division multiplexing (e.g., ±1/( _2Tr × Nt)). Note that the predetermined period is composed of a Doppler multiplexing period (period _Tr × Nc) + a non-transmission period.

However, in Doppler multiplexing, as described above, transmitted signals are separated by filtering on the Doppler frequency axis. Therefore, the effective Doppler frequency bandwidth per transmitted signal is narrower than the Doppler frequency range in which Doppler frequencies can be detected without aliasing. For example, if the Doppler frequency range ±1/(2× _Tr ) in which Doppler frequencies can be detected without aliasing is equally divided into Nt transmitted signals, the effective Doppler frequency range of each signal is limited to 1/( _Tr × Nt), resulting in a Doppler frequency range similar to that in the case of time-division multiplexing. Furthermore, in Doppler multiplexing, in a Doppler frequency band exceeding the effective Doppler frequency range per transmitted signal, signals in the Doppler frequency bands of other transmitted signals different from the transmitted signal are mixed, which may make it difficult to correctly separate the transmitted signals.

[Unequal Interval Doppler Multiplexing]
One method for expanding the maximum detectable Doppler frequency range in such Doppler multiplex transmission is to equally divide the Doppler frequency range ±1/( _2Tr ) in which Doppler frequencies can be detected without aliasing into Nt+1 parts, assign Nt Doppler shift amounts among the Nt+1 divided Doppler shift amounts to Nt transmission signals, and transmit the transmission signals simultaneously from Nt transmission antennas (see, for example, Patent Document 4).

In this Doppler multiplexing transmission, for example, since no transmission signal is assigned to some of the Nt+1 equally divided Doppler shift amounts, the Doppler shift intervals assigned to the Doppler-multiplexed transmission signals (hereinafter referred to as "Doppler multiplexing intervals") are unequal. Hereinafter, this type of Doppler multiplexing transmission will be referred to as "unequal interval Doppler multiplexing transmission."

Next, we will explain an example of how radar reflected waves are received and processed when using non-uniform Doppler multiplexing.

In the output obtained by applying Fourier frequency analysis for Doppler frequency detection (relative velocity detection), for example, among the Nt+1 equally divided Doppler shift amounts, the received power level of the Doppler corresponding to the Doppler shift amount to which the transmitted signal is not assigned is lower than the received power level of the Doppler corresponding to the Doppler shift amount to which the transmitted signal is assigned. The radar device may, for example, utilize this difference in received power level to estimate the Doppler frequency. This estimation process enables the radar device to estimate the Doppler frequency of the radar reflection wave within the Doppler frequency range ±1/( _2Tr ).

In this way, uneven Doppler multiplex transmission, which imparts Doppler shifts at uneven intervals in the Doppler frequency domain, allows the radar device to suppress ambiguity in Doppler frequencies and expand the maximum detectable Doppler frequency to 1/ _2Tr by detecting the unevenly spaced Doppler domains, even when targets with Doppler frequencies outside the divided Doppler frequency domain ±1/( _2Tr × (Nt+1)) are included. As a result, with uneven Doppler multiplex transmission, the detectable Doppler frequency range is expanded by Nt times compared to, for example, the method described in Patent Document 3.

For example, in Patent Document 4, due to the constraints of the sampling theorem of Fourier frequency analysis, Doppler frequencies (or relative velocities) exceeding the maximum detectable Doppler frequency _½Tr cannot be detected. For example, shortening the transmission period Tr can expand the Doppler detection range, but shortening the transmission period _Tr while maintaining the detectable distance range or distance _resolution requires using an A/D converter with a faster sampling rate, which complicates the hardware configuration. Furthermore, increasing the sampling rate of the A/D converter can increase the power consumption or heat generation of the radar device. On the other hand, if the transmission period _Tr is shortened under the constraints of the sampling rate of the A/D converter, the detectable distance range may be reduced or distance resolution may be degraded, which may degrade the distance detection range or distance separation performance of the radar device.

Furthermore, in unevenly spaced Doppler multiplex transmission, for example, if there are multiple reflected waves from approximately the same distance from the radar device, and the Doppler intervals of these reflected waves match the Doppler multiplex interval (e.g., represented as "Δf _DDM ") or a multiple of the Doppler multiplex interval, the radar device is more likely to make errors in detecting the unevenly spaced Doppler regions, which is more likely to result in errors in separating the multiple waves or errors in measuring the angles of the multiple reflected waves.

For example, as shown in FIG. 1, a radar device uses Nt=2 transmitting antennas to perform unequal interval Doppler multiplexing transmission using two of the Doppler shift amounts equally divided into 3 (=Nt+1) amounts, and receives reflected waves #1 and #2 from targets at the same distance from the radar device, with the difference in Doppler frequency between reflected waves #1 and #2 being Δf _DDM .

1(a), (b), and (c) show outputs (e.g., the output of a frequency analysis unit) obtained by applying Fourier frequency analysis for Doppler frequency detection (or relative velocity detection). 1(a) shows the received power of reflected wave #1, 1(b) shows the received power of reflected wave #2, and 1(c) shows the combined received signals of reflected waves #1 and #2. Since the difference in Doppler frequency between reflected waves #1 and #2 is Δf _DDM , reflected wave #2 in 1(b) is shifted by +Δf _DDM on the Doppler frequency axis from reflected wave #1 in 1(a).

As shown in Figures 1(a) and (b), of the Nt+1 equally divided Doppler shift amounts, the received power level of the Doppler frequency corresponding to the Doppler shift amount to which the transmitted signal is not assigned is lower (approximately the noise level) than the received power level of the Doppler corresponding to the Doppler shift amount to which the transmitted signal is assigned. In contrast, in Figure 1(c), the received power level tends to be higher because it includes the received power of the other reflected wave.

1C, for example, the Doppler frequency -½T _r +2Δf _DDM in reflected wave #1, which corresponds to a Doppler shift amount to which no transmission signal is assigned, tends to have a high received power level because it matches the Doppler frequency corresponding to the Doppler shift amount assigned to the transmission signal in the other reflected wave #2. Similarly, the Doppler frequency -½T _r in reflected wave #2, which corresponds to a Doppler shift amount to which no transmission signal is assigned, tends to have a high received power level because it matches the Doppler frequency corresponding to the Doppler shift amount assigned to the transmission signal in the other reflected wave #1.

Furthermore, since a received signal is composed of phase and amplitude, the amplitude of the received power obtained by combining multiple transmitted signals varies depending on the phase value. For example, in the case of (c) in Figure 1, the Doppler frequency components of -1/ _2Tr + Δf _DDM correspond to the Doppler shift amounts assigned to the transmitted signals in reflected wave #1 and reflected wave #2, so the received power is a combination of these components, and the amplitude of the combination varies depending on the phase value.

In a radar device using non-uniform Doppler multiplex transmission, a radar receiver separates non-uniform Doppler multiplex signals by, for example, using a Doppler multiplex separation unit (described later) to detect Doppler peak positions that coincide with the Doppler multiplex intervals to which the transmission signals are assigned, and separates the Doppler multiplex transmission signals. At this time, the Doppler multiplex separation unit utilizes the fact that the received power of the Doppler frequency components of the Doppler multiplex intervals to which the Doppler multiplex transmission signals are not assigned to be sufficiently low,
Separating the Doppler multiplexed signals.

By utilizing this difference in received power level, the Doppler frequency can be uniquely estimated within the Doppler frequency range of ±1/( _2Tr ), and Doppler multiplexed transmission signals can be separated. For example, in Figure 1(a), the received power of reflected wave #1 detects the Doppler peak positions of the Doppler frequency components of -1/ _2Tr , which matches the Doppler multiplexing interval _ΔfDDM , and -1/ _2Tr + _ΔfDDM . Furthermore, the received power of the Doppler frequency component (-1/ _2Tr + _2ΔfDDM ), which is shifted by the Doppler multiplexing interval _ΔfDDM from these Doppler peak positions, is sufficiently low, allowing the Doppler frequency to be estimated and the Doppler multiplexed transmission signals to be separated.

However, for example, the received power at a Doppler frequency of -1/ _2Tr + _2ΔfDDM in Figure 1(c) is at a higher Doppler position than the received power at a Doppler frequency of -1/ _2Tr + _2ΔfDDM in Figure 1(a), making it easier to erroneously estimate the Doppler frequency of reflected wave #1 and degrading the separation performance of the transmitting antenna.Similarly, for example, the received power at a Doppler frequency of -1/ _2Tr in Figure 1(c) is at a higher Doppler position than the received power at a Doppler frequency of -1/ _2Tr in Figure 1(b), making it easier to erroneously estimate the Doppler frequency of reflected wave #2 and degrading the separation performance of the transmitting antenna.

In addition, in (c) of Figure 1, at the Doppler frequency (-1/2T _r +1Δf _DDM ), the Doppler of Tx #2 of reflected wave #1 and the Doppler of Tx #1 of reflection #2, which have different phases and amplitudes, are combined, resulting in changes in phase and amplitude from the states in (a) and (b) of Figure 1, and deterioration of angle measurement accuracy.

For example, in (c) of Figure 1, the difference in Doppler frequency between reflected wave #1 and reflected wave #2 is +Δf _DDM , so the Doppler frequency component of reflected wave #2 to which the transmission signal from Tx #2 is assigned is received overlapping with the Doppler frequency component (solid line x mark, -1/2T _r +2Δf _DDM ) corresponding to the Doppler shift amount to which the transmission signal in reflected wave #1 is not assigned.

Furthermore, since the difference in Doppler frequency between reflected wave #2 and reflected wave #1 is -Δf _DDM , the Doppler frequency component of reflected wave #1 to which the transmission signal by Tx #1 is assigned is received overlapping with the Doppler frequency component (dotted circle, -1/2T _r ) corresponding to the Doppler shift amount to which no transmission signal is assigned in reflected wave #2.

Therefore, when reflected waves #1 and #2 are received from targets at the same distance and the difference in Doppler frequency between reflected waves #1 and #2 is Δf _DDM , the Doppler frequency corresponding to the amount of Doppler shift to which the transmitted signal is not assigned (solid line x or dotted line circle in Figure 1(c)) will be lower than the Doppler reception power level corresponding to the amount of Doppler shift to which the transmitted signal is assigned (Tx #1 and Tx #2 of reflected wave #1 in Figure 1(a) and Tx #1 and Tx #2 of reflected wave #2 in Figure 1(b)).

However, the Doppler frequency corresponding to the Doppler shift amount to which no transmission signal is assigned tends to be high because it includes the received power of the other reflected wave. For example, for the solid line x in Fig. 1(a), the combined received power at Doppler frequency = -1/ _2Tr + 2Δf _DDM in Fig. 1(c) tends to be high because it includes Tx#2 of reflected wave #2 in Fig. 1(b).

For this reason, when the radar device is in the state shown in Figure 1(c), the probability of an error in estimating the Doppler frequency increases. If the radar device makes an error in estimating the Doppler frequency, it is more likely to fail to properly separate the transmitting antennas, and angle measurement errors are also more likely to increase.

Furthermore, even if the Doppler frequency is estimated correctly, if the Doppler multiplexed signal of reflected wave #1 and the Doppler multiplexed signal of reflected wave #2 contain the same Doppler component (Doppler frequency is −½T _r +1Δf _DDM ), as shown in FIG. 1C, they are added (combined) as complex signals, which changes the amplitude component or phase component from the state shown in FIG. 1A or 1B, and the angle measurement accuracy of the radar device is likely to deteriorate.

Therefore, in a non-limiting embodiment of the present disclosure, a method is described for expanding the range of Doppler frequencies in which aliasing does not occur (e.g., no ambiguity occurs) in Doppler multiplex transmission. This enables a radar device according to one embodiment of the present disclosure to accurately detect targets over a wider Doppler frequency range.

Furthermore, in a non-limiting embodiment of the present disclosure, a method is described that enables the separate detection of each reflected wave even when the Doppler interval of each reflected wave from multiple targets at similar distances from the radar device matches the Doppler multiplex interval (or a multiple of the Doppler multiplex interval).

Furthermore, non-limiting examples of the present disclosure describe a method for expanding the range of Doppler frequencies (relative velocities) in which aliasing does not occur in Doppler multiplex transmission, and for enabling the separate detection of each reflected wave even when the Doppler intervals of reflected waves from multiple targets at similar distances from the radar device match the Doppler multiplex interval (or a multiple of the Doppler multiplex interval).

Note that a radar device according to an embodiment of the present disclosure may be mounted on a moving body such as a vehicle. The positioning output (information related to the estimation results) of the radar device mounted on the moving body may be output to a control ECU (Electronic Control Unit) (not shown) such as an Advanced Driver Assistance System (ADAS) that improves collision safety or an autonomous driving system, and may be used for vehicle drive control or alarm generation control.

A radar device according to an embodiment of the present disclosure may be attached to a relatively high structure (not shown), such as a roadside utility pole or traffic light. Such a radar device may be used, for example, as a sensor in an assistance system that improves the safety of passing vehicles or pedestrians, or in a system that prevents the intrusion of suspicious individuals. The positioning output of the radar device may be output to a control device (not shown) in the safety-improving assistance system or the system that prevents the intrusion of suspicious individuals, and used for alarm generation control or abnormality detection control.

Note that the uses of the radar device are not limited to these and it may also be used for other purposes.

An embodiment of the present disclosure will be described in detail below with reference to the drawings. Note that in the embodiments, identical components are designated by the same reference numerals, and redundant descriptions will be omitted.

The following describes a radar device configuration (e.g., a MIMO radar configuration) in which different transmit signals are simultaneously multiplexed from multiple transmit antennas in the transmit branch, and each transmit signal is separated and received in the receive branch.

Furthermore, the following describes, as an example, the configuration of a radar system using frequency-modulated pulse waves such as chirp pulses (also referred to as fast chirp modulation). However, the modulation system is not limited to frequency modulation. For example, an embodiment of the present disclosure can also be applied to a radar system using pulse compression radar that transmits a pulse train after phase modulation or amplitude modulation.

(Embodiment 1)
[Radar Device Configuration]
The radar device 10 in FIG. 2 includes a radar transmitter (transmitting branch) 100 and a radar receiver (receiving branch) 200.

The radar transmitter 100 generates a radar signal (radar transmission signal) and transmits the radar transmission signal at a predetermined transmission period using a transmission array antenna composed of multiple transmission antennas 106-1 to 106-Nt.

The radar receiver 200 receives reflected wave signals, which are radar transmission signals reflected by targets (not shown), using a receiving array antenna including multiple receiving antennas 202-1 to 202-Na. The radar receiver 200 processes the reflected wave signals received by each receiving antenna 202, for example, to detect the presence or absence of a target or estimate the direction of arrival of the reflected wave signals.

Note that a target object is an object that the radar device 10 detects, and includes, for example, a vehicle (including two-wheeled and four-wheeled vehicles), a person, a block, or a curb.

[Configuration of radar transmitter 100]
The radar transmitter 100 includes a radar transmission signal generator 101, a signal generation controller 104, Doppler shifters 105-1 to 105-Nt, and transmitting antennas 106-1 to 106-Nt. For example, the radar transmitter 100 includes Nt transmitting antennas 106, each connected to a separate Doppler shifter 105.

The radar transmission signal generation unit 101 generates a radar transmission signal, for example, under control of the signal generation control unit 104. The radar transmission signal generation unit 101 includes, for example, a modulation signal generation unit 102 and a VCO (Voltage Controlled Oscillator) 103. Each component of the radar transmission signal generation unit 101 is described below.

The modulation signal generator 102 periodically generates a modulation signal, for example, a sawtooth-shaped modulation signal, where the radar transmission period is _Tr .

VCO 103 generates a frequency modulation signal (hereinafter referred to as a frequency chirp signal or chirp signal, for example) based on the modulation signal output from modulation signal generator 102, and outputs it to Doppler shifters 105-1 to 105-Nt and radar receiver 200 (mixer unit 204, described below).

The signal generation control unit 104 controls the radar transmission signal generation unit 101 (e.g., the modulation signal generation unit 102 and VCO 103) to generate the radar transmission signal. For example, the signal generation control unit 104 may set parameters (e.g., modulation parameters) related to the chirp signal so that chirp signals with different center frequencies are transmitted alternately.

In the following, the two chirp signals with different center frequencies will be referred to as the "first chirp signal" and the "second chirp signal," respectively.

Figure 3 shows examples of chirp signals (e.g., a first chirp signal and a second chirp signal).

3, modulation parameters for the chirp signal may include, for example, center frequency f _c (q), frequency sweep bandwidth B _w (q), sweep start frequency f _cstart (q), sweep end frequency f _cend (q), frequency sweep time T _sw (q), and frequency sweep change rate D _m (q). Note that D _m (q) = B _w (q) / T _sw (q). Also, B _w (q) = f _cend (q) - f _cstart (q) and f _c (q) = (f _cstart (q) + f _cend (q)) / 2. Also, for example, q = 1 or 2, where q = 1 represents the modulation parameters for the first chirp signal and q = 2 represents the modulation parameters for the second chirp signal.

The frequency sweep time _Tsw (q) corresponds to, for example, a time range (also called a range gate) for capturing A/D sample data in an A/D conversion unit 207 of the radar receiver 200 (described later). The frequency sweep time _Tsw (q) may be set to the entire interval of the chirp signal, as shown in Fig. 3(a), or may be set to a partial interval of the chirp signal, as shown in Fig. 3(b).

Note that Figure 3 shows an example of an up-chirp waveform in which the modulation frequency gradually increases over time, but this is not limited to this; a down-chirp waveform in which the modulation frequency gradually decreases over time may also be applied. Similar effects can be obtained regardless of whether the modulation frequency is an up-chirp or a down-chirp.

The signal generation control unit 104 may, for example, set (or select) a center frequency f _c (q) that satisfies a predetermined condition (an example will be described later).

In the following, as an example, a case will be described in which the center frequencies f _c (q) of the modulation parameters set for the first chirp signal and the second chirp signal are different from each other, but the other modulation parameters are the same (or common). However, this is not limiting, and application of an embodiment of the present disclosure is not limited to this. For example, it is sufficient that the resolution of the distance axis of the first chirp signal and the second chirp signal is the same, and therefore it is sufficient to set chirp signals with the same frequency sweep bandwidth B _w (q) (an example will be described later).

In the following description, the signal generation control section 104 may control the modulation signal generating section 102 and the VCO 103 so that two chirp signals with different center frequencies f _c (q) are alternately transmitted Nc times.

Figure 4 shows an example of a chirp signal output by the radar transmission signal generation unit 101 based on the control of the signal generation control unit 104.

In Fig. 4, the transmission period T _r1 of the first chirp signal and the transmission period T _r2 of the second chirp signal may be different (T _r1 ≠ T _r2 ) or may be the same (T _r1 = T _r2 ). In the following, the combined period of the transmission periods T _r1 and T _r2 will be referred to as "T _rs ". For example, the transmission period T _rs indicates the transmission period in which a set of the first chirp signal and the second chirp signal is transmitted, and is T _rs = T _r1 + T _r2 . In the following description, unless otherwise specified, the transmission periods T _r1 and T _r2 represent parameters with the same value (e.g., T _r1 = T _r2 ), and for convenience, they may be referred to as the transmission period T _r .

Similarly, unless otherwise specified, the frequency sweep bandwidth, frequency sweep time (also called range gate), and frequency sweep change rate represent parameters with the same values for the first chirp signal and the second chirp signal, and may be expressed as _Bw (1)= _Bw (2)= _Bw , _Tsw (1)= _Tsw (2)= _Tsw , and _Dm (1)= _Dm (2)= _Dm .

Furthermore, the frequency sweep bandwidths of each chirp signal with a different center frequency may not include overlapping bands, as shown in Figure 4(a), or may include overlapping bands, as shown in Figure 4(b). In one embodiment of the present disclosure, similar effects can be achieved regardless of whether the frequency sweep bandwidths include overlapping bands, as long as the relationship in center frequency between the first chirp signal and the second chirp signal satisfies a predetermined condition.

In one embodiment of the present disclosure, the transmission period T _rs may be set to, for example, several hundred μs or less, and the transmission time interval of the radar transmission signal may be set to a relatively short time. As a result, even if the center frequencies of the first chirp signal and the second chirp signal are different, the frequency of the beat signal of the received reflected wave (e.g., the beat frequency index) does not change, and the radar device 10 can detect this as a change in Doppler frequency.

Each chirp signal output from the radar transmission signal generation unit 101 (e.g., VCO 103) is input, for example, to each mixer unit 204 of the radar reception unit 200 and to Nt Doppler shift units 105.

The Doppler shifter 105 applies a phase rotation φ _n to the chirp signal input from the VCO 103 in order to impart a Doppler shift amount DOP _n to the chirp signal for each transmission period (e.g., _Tr1 or _Tr2 ) of the chirp signal, and outputs the Doppler-shifted signal to the transmitting antenna 106, where n = 1, ..., Nt. An example of a method for imparting a Doppler shift amount DOP _n (e.g., phase rotation φ _n ) in the Doppler shifter 105 will be described later.

The output signals from Doppler shift units 105-1 to 105-Nt are amplified to a predetermined transmission power and radiated into space from each transmitting antenna 106 (e.g., Tx#1 to Tx#Nt).

[Configuration of radar receiver 200]
2 , the radar receiver 200 includes Na receiving antennas 202 (e.g., Rx#1 to Rx#Na) forming an array antenna. The radar receiver 200 also includes Na antenna system processors 201-1 to 201-Na, a CFAR (Constant False Alarm Rate) unit 211, a Doppler demultiplexing unit 212, a Doppler determination unit 213, and a direction estimation unit 214.

The CFAR unit 211 may include, for example, a CFAR unit 211-1 and a CFAR unit 211-2 corresponding to the first chirp signal and the second chirp signal, respectively, which have different center frequencies. Similarly, the Doppler demultiplexing unit 212 may include, for example, a Doppler demultiplexing unit 212-1 and a Doppler demultiplexing unit 212-2 corresponding to the first chirp signal and the second chirp signal, respectively, which have different center frequencies. While FIG. 2 shows a configuration in which CFAR units 211 are provided in parallel (CFAR units 211-1 and 211-2), it is also possible to provide a single CFAR unit 211 and sequentially switch its input for processing. While FIG. 2 shows a configuration in which Doppler demultiplexing units 212 are provided in parallel (Doppler demultiplexing units 212-1 and 212-2), it is also possible to provide a single Doppler demultiplexing unit 212 and sequentially switch its input for processing.

Each receiving antenna 202 receives a reflected wave signal, which is a radar transmission signal reflected by a target, and outputs the received reflected wave signal to the corresponding antenna system processing unit 201 as a received signal.

Each antenna system processing unit 201 has a receiving radio unit 203 and a signal processing unit 206.

The receiving radio unit 203 includes a mixer unit 204 and an LPF (low pass filter) 205. In the receiving radio unit 203, the mixer unit 204 mixes the received reflected wave signal (received signal) with a chirp signal, which is a transmitted signal. Furthermore, by passing the output of the mixer unit 204 through the LPF 205, a beat signal is extracted, the frequency of which corresponds to the delay time of the reflected wave signal. For example, the difference frequency between the frequency of the transmitted signal (transmitted frequency modulated wave) and the frequency of the received signal (received frequency modulated wave) is obtained as the beat frequency (or beat signal).

The signal processing unit 206 of each antenna system processing unit 201-z (where z = 1 to Na) includes an A/D conversion unit 207, a beat frequency analysis unit 208, and a Doppler analysis unit 210. The Doppler analysis unit 210 may include, for example, a Doppler analysis unit 210-1 and a Doppler analysis unit 210-2 corresponding to a first chirp signal and a second chirp signal, respectively, which have different center frequencies.

The signal (e.g., beat signal) output from the LPF 205 is converted into discrete sample data by the A/D conversion unit 207 in the signal processing unit 206.

The beat frequency analyzer 208 performs FFT processing on N _data pieces of discrete sample data obtained within a predetermined time range (range gate) for each transmission period T _r. Here, the range gate sets a frequency sweep time T _sw (q). Here, for example, q = 1 or 2, where q = 1 represents the frequency sweep time of the first chirp signal, and q = 2 represents the frequency sweep time of the second chirp signal. As a result, the signal processor 206 outputs a frequency spectrum in which a peak appears at a beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave). During FFT processing, the beat frequency analyzer 208 may multiply the data by a window function coefficient, such as a Han window or a Hamming window. Using the window function coefficient can suppress side lobes that occur around the beat frequency peak.

Here, the beat frequency response output from the beat frequency analysis unit 208 in the zth signal processing unit 206, obtained by transmitting the mth chirp pulse of the qth chirp signal, is represented by RFT _z,q (f _b , m). Here, f _b represents the beat frequency index, which corresponds to the FFT index (bin number). For example, f _b = 0, to, N _data /2-1, z = 1, to, Na, m = 1, to, N _C , and q = 1 or 2. The smaller the beat frequency index f _b , the shorter the delay time of the reflected wave signal (e.g., the closer the distance to the target) is, indicating a beat frequency.

The beat frequency index f _b can be converted into distance information R(f _b ) using the following equation (1): Therefore, hereinafter, the beat frequency index f _b will be referred to as the "distance index f _b ".

Here, _Bw represents the frequency sweep bandwidth of the chirp signal, and _C0 represents the speed of light. Also, in equation (1), _C0 / _2Bw represents the range resolution. Hereinafter, the range resolution ΔR is expressed as ΔR= _C0 / _2Bw .

The output switching unit 209 selectively switches the output of the beat frequency analysis unit 208 to one of the two Doppler analysis units 210 depending on the transmission period of the first chirp signal or the transmission period of the second chirp signal, based on a control signal output from the signal generation control unit 104. For example, the output switching unit 209 outputs the output of the beat frequency analysis unit 208 for the transmission period T _r1 of the first chirp signal (e.g., q=1) to the Doppler analysis unit 210-1. Also, for example, the output switching unit 209 outputs the output of the beat frequency analysis unit 208 for the transmission period T _r2 of the second chirp signal (e.g., q=2) to the Doppler analysis unit 210-2.

The qth Doppler analysis unit 210 (also referred to as Doppler analysis unit 210-q) performs Doppler analysis for each distance index f b using beat frequency responses RFT _z,q (f _b , 1), RFT _z,q (f _b , 2), ..., RFT _z,q (f _b , N _C ) obtained by transmitting chirp pulses of the _qth chirp signal N _C times, which are output from the output switching unit 209. For example, the qth Doppler analysis unit 210 may estimate the Doppler frequency from a reflected wave signal of the qth chirp signal reflected by a target.

For example, when _Nc is a power of 2, FFT processing can be applied in Doppler analysis. In this case, the FFT size is _Nc , and the maximum Doppler frequency at which aliasing does not occur, as derived from the sampling theorem, is ±1/( _2Trs ). Furthermore, the Doppler frequency interval of the Doppler frequency index _fs is 1/( _Nc × _Trs ), and the range of the Doppler frequency index _fs is _fs = _-Nc /2, ∼, 0, ∼, _Nc /2-1.

In the following, an example will be described in which _Nc is a power of 2. If _Nc is not a power of 2, FFT processing can be performed with a data size of a power of 2 by, for example, including zero-padded data. Furthermore, the Doppler analyzer 210 may multiply the data by a window function coefficient such as a Han window or a Hamming window during FFT processing. Applying a window function can suppress side lobes that occur around the beat frequency peak.

For example, the output VFT _z,q (f _b , f _s ) of the q-th Doppler analyzer 210 of the z-th signal processor 206 is shown in the following equation (2), where j is the imaginary unit, z=1 to Na, and q=1, 2.

The above explains the processing performed by each component of the signal processing unit 206.

2, the CFAR unit 211 performs CFAR processing (e.g., adaptive threshold determination) using outputs from the Doppler analysis units 210 of the first to Na-th signal processing units 206, and extracts a distance index f _{b — cfar} and a Doppler frequency index f _{s — cfar} that provide a local peak signal. As shown in FIG. 2, the CFAR unit 211 may include a first CFAR unit 211 (also referred to as CFAR unit 211-1) that performs CFAR processing using outputs from the first Doppler analysis units 210, and a second CFAR unit 211 (also referred to as CFAR unit 211-2) that performs CFAR processing using outputs from the second Doppler analysis units 210.

The qth CFAR unit 211 (q=1,2) adds the power of the outputs VFT1 _{,q(fb,fs), VFT2,q} ( _fb , _fs ), to VFTNa _,q ( _fb , _{fs )} of the qth Doppler analyzers 210 of the first to Nath signal processors 206, as shown in the following equation (3 ₎ , for example, and performs two _- dimensional CFAR processing consisting of a distance axis and a Doppler frequency axis (corresponding to relative velocity), or CFAR processing that combines one-dimensional CFAR processing. For the two-dimensional CFAR processing or the CFAR processing that combines one-dimensional CFAR processing, the processing disclosed in Non-Patent Document 2, for example, may be applied.

The qth CFAR unit 211 adaptively sets a threshold value and outputs to the qth Doppler demultiplexing unit 212 a distance index f _{b_cfar} (q), a Doppler frequency index f _{s_cfar} (q), and received power information PowerFT(f _{b_cfar} (q), f _{s_cfar} (q)) that result in received power greater than the threshold value.

The Doppler demultiplexing unit 212 may include a first Doppler demultiplexing unit 212 (or referred to as Doppler demultiplexing unit 212-1) that performs Doppler demultiplexing processing using the outputs of the first Doppler analysis unit 210 and the first CFAR unit 211, and a second Doppler demultiplexing unit 212 (or referred to as Doppler demultiplexing unit 212-2) that performs Doppler demultiplexing processing using the outputs of the second Doppler analysis unit 210-2 and the second CFAR unit 211.

The q-th Doppler demultiplexing unit 212 (q=1, 2) separates the transmission signals (e.g., reflected wave signals corresponding to the transmission signals) transmitted from each transmitting _antenna 106 from the Doppler- _multiplexed signals (hereinafter referred to as "Doppler multiplexed signals") using the output from the q-th Doppler analysis unit 210, based on information input from the q-th CFAR unit 211 (e.g., distance index _{fb_cfar} (q), Doppler frequency index _{fs_cfar} (q), and received power information PowerFT(fb_cfar(q), fs_cfar(q))). The q-th Doppler demultiplexing unit 212 outputs information related to the separated signals to, for example, the Doppler determination unit 213 and the direction estimation unit 214. The information about the separated signals may include, for example, a distance index f _{b_cfar} (q) and Doppler frequency indices (hereinafter sometimes referred to as separation index information) (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q), ..., f _{demul_Tx#Nt} (q)) corresponding to the separated signals. In addition, the qth Doppler demultiplexing unit 212 outputs the output from the qth Doppler analysis unit 210 to the direction estimation unit 214.

Below, an example of the operation of the qth Doppler demultiplexing unit 212 will be explained, along with the operation of the Doppler shift unit 105.

[How to set the Doppler shift amount]
First, an example of a method for setting the amount of Doppler shift applied in the Doppler shifter 105 will be described.

The Doppler shift units 105-1 to 105-Nt apply different Doppler shift amounts DOP _n to the chirp signals input thereto. In one embodiment of the present disclosure, the intervals (Doppler shift intervals) of the Doppler shift amounts DOP _n between the Doppler shift units 105-1 to 105-Nt (for example, between the transmitting antennas 106-1 to 106-Nt) are not equal, but are set so that at least one Doppler interval is different.

For example, the nth Doppler shifter 105 applies a phase rotation φ _n (m) to the input mth qth chirp signal, resulting in a different Doppler shift amount DOP _n , and outputs the result. As a result, different Doppler shift amounts are applied to the transmission signals transmitted from the multiple transmitting antennas 106. For example, in one embodiment, the Doppler multiplexing number N _DM =Nt, where m is an integer from 1 to _NC , n is an integer from 1 to Nt, and q is 1 or 2.

Furthermore, in the q-th Doppler analyzer 210, the range of Doppler frequency _fd in which aliasing does not occur, which is derived from the sampling theorem, is −1/(2T _rs )≦f _d <1/(2T _rs ).

From this, for the transmission signals transmitted from Nt transmitting antennas 106, the phase rotation φ _n (m) that makes the Doppler shift intervals equal 1/(Nt×T _rs ) is expressed by the following equation (4).

Here, φ ₀ is the initial phase, and Δφ ₀ is the reference Doppler shift phase. Furthermore, round(x) is a round function that outputs an integer value obtained by rounding off a real value x. Note that the term round(N _C /N _t ) is introduced for the purpose of making the amount of phase rotation an integer multiple of the Doppler frequency interval in the Doppler analysis unit 210.

For example, if the phase rotation φ _n (m) shown in equation (4) is used, the intervals of the phase rotation between the transmission signals imparted to the mth qth chirp signal will all be equal, 2πround(N _C /N _t )/N _C.

As an example, in equation (4), when phase rotation φ _n (m) is applied with Nt=2, Δφ ₀ =0, and φ ₀ =0, the Doppler shift amounts are DOP ₁ =0 and DOP ₂ =−1/(2T _rs ).

For example, the intervals between the amounts of Doppler shift imparted to the transmission signals transmitted from the multiple transmitting antennas 106 are set to equal intervals within the Doppler frequency range (e.g., a Doppler frequency range in which aliasing does not occur) in the radar device 10 (radar receiving unit 200). For example, the intervals between the amounts of Doppler shift imparted to the transmission signals transmitted from Nt=2 transmitting antennas 106 are set to intervals (1/( _2Trs ) in the above example) obtained by dividing the Doppler frequency range in which aliasing does not occur (e.g., -1/(2Trs) ≦ _fd < 1/( _2Trs )) by the number of transmitting antennas 106 (e.g., _Nt =2). Therefore, the Doppler interval between Doppler peak P1 and Doppler peak P2 is 1/( _2Trs ).

Figure 5 shows an example of a Doppler peak obtained by Doppler analysis (FFT) in the Doppler analysis unit 210 when Doppler shift amounts of DOP ₁ = 0 and DOP ₂ = -1/(2T _rs ) are used for transmission signals transmitted from Nt=2 transmitting antennas 106 (hereinafter referred to as Tx#1 and Tx#2).

As shown in FIG. 5, Nt (Nt=2 in FIG. 5) Doppler peaks occur for the Doppler frequency (target doppler) _{fd_TargetDoppler} of one target to be measured.

As an example, in Figure 5, _we compare the positional relationship between _{the Doppler} peak that occurs when a reflected wave signal is received in response to a transmission signal transmitted from transmitting antenna Tx#1 and the Doppler peak that occurs when a reflected wave signal is received in response to a transmission signal transmitted from transmitting antenna Tx#2 when _{the Doppler frequency} of the target to be measured is fd_TargetDoppler = -1/(4Trs) and fd_TargetDoppler = 1/(4Trs).

<When the target Doppler frequency f _{d_TargetDoppler} =-1/(4T _rs )>
When _{fd_TargetDoppler} = -1/( _4Trs ), as shown in Figure 5, the Doppler peak that occurs when a reflected wave signal is received from the transmission signal from transmitting antenna Tx#2 is output in the FFT as a peak (P2A) of the aliased signal. Therefore, when _{fd_TargetDoppler} = 1/( _4Trs ), as shown in Figure 5, the positional relationship between the Doppler peak (P1) that occurs when a reflected wave signal is received from the transmission signal from transmitting antenna Tx#1 and the Doppler peak (P2A) of the aliased signal is as follows. The Doppler interval between the Doppler peak (P1) and the Doppler peak (P2A) is 1/( _2Trs ). Note that although the Doppler peak (P2') is shown as the signal before aliasing for reference, it does not actually exist in the FFT output.

<When the target Doppler frequency is f _{d_TargetDoppler} =1/(4T _rs )>
When _{fd_TargetDoppler} = 1/( _4Trs ), the positional relationship between the Doppler peak (P1) generated when a reflected wave signal is received in response to a transmission signal from transmitting antenna Tx#1 and the Doppler peak (P2) generated when a reflected wave signal is received in response to a transmission signal from transmitting antenna Tx#2 is as shown in Figure 5. The Doppler interval between the Doppler peak (P1) and the Doppler peak (P2) is 1/( _2Trs ).

Thus, when _{fd_TargetDoppler} = -1/( _4Trs ) and when _{fd_TargetDoppler} = 1/( _4Trs ), the Doppler interval between the Doppler peak (P1) corresponding to transmitting antenna Tx#1 and the Doppler peak (P2 or P2A) corresponding to transmitting antenna Tx#2 is 1/( _2Trs ). Therefore, when _{fd_TargetDoppler} = -1/( _4Trs ) and 1/( _4Trs ), the positional relationship between the Doppler peaks corresponding to Tx#1 and Tx#2 becomes indistinguishable, resulting in ambiguity. Therefore, in the example shown in Fig. 5, the Doppler frequency range of the target where ambiguity does not occur is, for example, -1/( _4Trs ) ≦ _{fd_TargetDoppler} < 1/( _4Trs ).

6, the Doppler shift unit 105 according to an embodiment of the present disclosure sets Doppler shift amounts that result in unequal Doppler multiplexing intervals. For example, the Doppler shift unit 105 may impart Doppler shift amounts at intervals obtained by dividing the Doppler frequency range that is the target for determining the number of Doppler frequency aliasings into unequal intervals. For example, the Doppler shift unit 105 imparts Doppler shift amounts DOP _n (or phase rotations φ _n (m)) to the transmission signal transmitted from the transmitting antenna 106 at intervals that differ by at least one.

Furthermore, for example, the Doppler shift unit 105 applies the Doppler shift DOP n to the transmission signals transmitted from the Nt transmission antennas ₁₀₆ so that the intervals between the Doppler shifts are as large as possible and the intervals between at least one of the phase rotations φ _n (m) are different, thereby improving the Doppler multiplexing separation performance.

For example, the nth Doppler shifter 105 applies a phase rotation φ n (m) to the mth input first chirp signal or second chirp signal, as shown in the following equation (5), which results in a Doppler shift amount _{DOP n} that differs between each Doppler shifter.

Here, A is a coefficient that gives a positive or negative polarity of 1 or −1. δ is an integer equal to or greater than 1. The term round(N _C /(Nt+δ)) is introduced for the purpose of making the amount of phase rotation an integer multiple of the Doppler frequency interval in Doppler analysis unit 210, but this is not limitative, and 2π/(Nt+δ) may be used instead of the term (2π/N _C )×round(N _C /(Nt+δ)) in equation (5).

For example, the radar device 10 performs unequal Doppler multiplexing on the first chirp signal and the second chirp signal at the same Doppler multiplexing interval.

As an example, in equation (5), when Nt=2, Δφ ₀ =0, φ ₀ =0, A=1, δ=1, and N _C is a multiple of 3 and a phase rotation φ _n (m) is applied, the Doppler shift amounts are DOP ₁ =0 and DOP ₂ =1/(3T _rs ).

FIG. 6 shows an example of a Doppler peak obtained by Doppler analysis in the Doppler analysis unit 210 when Doppler shift amounts of DOP ₁ =0 and DOP ₂ =1/(3T _rs ) are used for transmission signals transmitted from Nt=2 transmitting antennas 106 (hereinafter referred to as Tx#1 and Tx#2).

As shown in FIG. 6, Nt (Nt=2 in FIG. 6) Doppler peaks occur for the Doppler frequency (target doppler) _{fd_TargetDoppler} of one target to be measured.

As an example, Figure 6 compares the positional relationship between the _Doppler peak that occurs when a reflected wave signal is received from a transmission signal transmitted from transmitting antenna _Tx #1 and the Doppler peak that occurs when a reflected wave signal is received from a transmission signal transmitted from transmitting antenna Tx#2 when the Doppler frequency of the _target being measured is _{fd_TargetDoppler} = -1/(4Trs) and fd_TargetDoppler = 1/(4Trs) in the output of the Doppler analysis unit 210.

<When the target Doppler frequency is f _{d_TargetDoppler} =-1/(4T _rs )>
When _{fd_TargetDoppler} = -1/( _4Trs ), the positional relationship between the Doppler peak (P1) generated when a reflected wave signal is received in response to a transmission signal from transmitting antenna Tx#1 and the Doppler peak (P2) generated when a reflected wave signal is received in response to a transmission signal from transmitting antenna Tx#2 is as shown in Figure 6. The Doppler interval between Doppler peak P1 and Doppler peak P2 is 1/( _3Trs ).

<When the target Doppler frequency f _{d_TargetDoppler} = 1/(4T _rs )>
When _{fd_TargetDoppler} = 1/( _4Trs ), as shown in Fig. 6, the Doppler peak that occurs when a reflected wave signal is received from the transmission signal from transmitting antenna Tx#2 is output as a peak (P2A) of the aliased signal by FFT. Therefore, when _{fd_TargetDoppler} = 1/( _4Trs ), the positional relationship between the Doppler peak (P1) that occurs when a reflected wave signal is received from the transmission signal from transmitting antenna Tx#1 and the Doppler peak (P2A) of the aliased signal is as follows: The Doppler interval between the Doppler peak (P1) and the peak (P2A) is 2/( _3Trs ).

As shown in Figure 6, when the target Doppler frequency _{fd_TargetDoppler} = -1/( _4Trs ) and when _{fd_TargetDoppler} = 1/( _4Trs ), the positional relationship between the Doppler peak (P1) corresponding to transmitting antenna Tx#1 and the Doppler peak (P2 or P2A) corresponding to transmitting antenna Tx#2 is different from each other.

Therefore, in the example shown in Figure 6, the Doppler demultiplexing unit 212 can distinguish between the case where the target Doppler frequency _{fd_TargetDoppler} = -1/( _4Trs ) (e.g., when there is no aliasing) and the case where _{fd_TargetDoppler} = 1/( _4Trs ) (e.g., when there is aliasing).

For example, when the assumed target Doppler frequency is -1/( _2Trs ) ≦ _{fd_TargetDoppler} < 1/( _2Trs ), the Doppler demultiplexing unit 212 can determine that no aliasing signals are included when the target Doppler frequency _{fd_TargetDoppler} = -1/( _4Trs ). Therefore, for example, when _{fd_TargetDoppler} = -1/( _4Trs ) as shown in Fig. 6, the Doppler demultiplexing unit 212 can determine that no aliasing signals are included and that the signals are reflected wave signals of the transmission signals from the transmitting antennas Tx#1 and Tx#2, based on the Doppler peaks with the smallest frequencies.

Furthermore, for example, if the assumed target Doppler frequency is -1/(2T _rs ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ), the Doppler multiplexing separation unit 212 can determine that a folded Doppler peak (e.g., P2A) is included when the target Doppler frequency f _{d_TargetDoppler} = 1/(4T _rs ), and can determine that the Doppler frequency is f _{d_TargetDoppler} = 1/(4T _rs ). Therefore, for example, when _{fd_TargetDoppler} = 1/( _4Trs ) shown in Figure 6, an aliasing signal (P2A) is included, so the Doppler demultiplexing unit 212 can determine that the higher of the Doppler peaks, whose interval is 2/( _3Trs ), is the reflected wave signal corresponding to transmitting antenna Tx#1, and the lower Doppler peak is the reflected wave signal corresponding to transmitting antenna Tx#2. Note that, although P1' and P2' are shown in Figure 6 for ease of explanation, they do not actually exist in the output of the Doppler analysis unit 210.

Next, as another example, in Figure 6, _we compare the positional relationship between the _Doppler peak that occurs when a reflected wave signal is received for a transmission signal transmitted from transmitting antenna Tx#1 and _{the Doppler peak} that occurs when a reflected wave signal is received for a transmission signal transmitted from transmitting antenna Tx#2 when the Doppler frequency of the target to be measured is fd_TargetDoppler = -1/(2Trs) and fd_TargetDoppler = 1/(2Trs).

<When the target Doppler frequency is f _{d_TargetDoppler} =-1/(2T _rs )>
When _{fd_TargetDoppler} = -1/( _2Trs ), the positional relationship between the Doppler peak (P1) generated when a reflected wave signal is received in response to a transmission signal from transmitting antenna Tx#1 and the Doppler peak (P2) generated when a reflected wave signal is received in response to a transmission signal from transmitting antenna Tx#2 is as shown in Figure 6. The Doppler interval between the Doppler peak (P1) and the Doppler peak (P2) is 1/( _3Tr ).

<When the target Doppler frequency is f _{d_TargetDoppler} =1/(2T _rs )>
When _{fd_TargetDoppler} = 1/( _2Trs ), as shown in Figure 6, the Doppler peak generated when a reflected wave signal from the transmission signal from transmitting antenna Tx#1 is received is output as the Doppler peak (P1A) of the aliased signal by FFT, and the Doppler peak generated when a reflected wave signal from the transmission signal from transmitting antenna Tx#2 is output as the Doppler peak (P2A) of the aliased signal by FFT. Therefore, the positional relationship between the Doppler peak (P1A) generated when a reflected wave signal from the transmission signal from transmitting antenna Tx#1 is obtained and the Doppler peak (P2A) of the aliased signal is as follows: The Doppler interval between the Doppler peak (P1A) and the Doppler peak (P2A) is 1/(3Tr).

Thus, when the target Doppler frequency _{fd_TargetDoppler} = -1/( _2Trs ) and when _{fd_TargetDoppler} = 1/( _2Trs ), the Doppler interval between the Doppler peak (P1) corresponding to transmitting antenna Tx#1 and the Doppler peak (P2 or P2A) corresponding to transmitting antenna Tx#2 is 1/( _3Trs ). Therefore, when _{fd_TargetDoppler} = -1/( _2Trs ) and when _{fd_TargetDoppler} = 1/( _2Trs ), it becomes impossible to distinguish the positional relationship between the Doppler peaks corresponding to Tx#1 and Tx#2, and ambiguity occurs. Therefore, in the example shown in FIG. 6, the target Doppler frequency range in which no ambiguity occurs in the Doppler demultiplexing unit 212 is, for example, −1/(2T _rs )≦f _{d — TargetDoppler} <1/(2T _rs ).

Therefore, with the Doppler shift setting in Figure 6, the Doppler frequency range of targets where ambiguity does not occur can be expanded by Nt times (e.g., twice in Figure 6) compared to time division multiplexing or Doppler multiplexing where the Doppler shift amounts are equally spaced (e.g., see Figure 5).

In this embodiment, we will explain a method for further expanding the Doppler frequency range of targets without ambiguity through processing by the Doppler determination unit 213, which will be described later.

Next, we will explain an example of a method for separating signals corresponding to each transmitting antenna 106 in the Doppler demultiplexing unit 212.

As an example, we will explain the operation of the Doppler demultiplexing unit 212 when Nt=2.

In the following, as an example, a case will be described in which the phase rotation φ _n (m) shown in equation (5) is applied in the Doppler shift unit 105. Note that in the following, as an example, Δφ ₀ = 0, φ ₀ = 0, δ = 1, and N _C is a multiple of 3. When A = 1, the amount of Doppler shift for each transmitting antenna 106 is DOP ₁ = 0, DOP ₂ = 1/( _3Tr ), and when A = -1, the amount of Doppler shift for each transmitting antenna 106 is DOP ₁ = 0, DOP ₂ = -1/( _3Tr ).

In this case, the qth Doppler multiplexing separation unit 212 separates the Doppler multiplexed signal using the peak (distance index f _{b_cfar} (q) and Doppler frequency index f _{s_cfar} (q)) having a received power greater than the threshold input from the qth CFAR unit 211.

For example, the qth Doppler demultiplexing unit 212 determines which of the transmission signals transmitted from the transmitting antennas Tx#1 to Tx#Nt corresponds to a reflected wave signal for a plurality of Doppler frequency indexes f _{s_cfar} (q) having the same distance index f _{b_cfar} (q). The Doppler demultiplexing unit 212 separates and outputs the reflected wave signal for each of the determined transmitting antennas Tx#1 to Tx#Nt.

The following describes the operation when there are Ns Doppler frequency indexes _{fs_cfar} (q) with the same distance index _{fb_cfar} (q). For example, let _{fs_cfar} (q)∈{fd _#1 , fd _#2 ..., fd _#Ns }.

Here, Nt=2 Doppler peaks are generated for one target Doppler frequency _{fd_TargetDoppler} due to the Doppler shift amounts _DOP1 and _DOP2 imparted to the transmission signals transmitted from transmitting antennas Tx#1 and Tx#2, respectively. The Doppler index interval corresponding to the Doppler interval between these Doppler peaks is round(Nc/(Nt+1)) from the difference between the phase rotation _φ1 (m) for transmitting antenna Tx#1 and the phase rotation _φ2 (m) for transmitting antenna _Tx #2, as shown in the following equation (6). Furthermore, when aliasing signals are included, the Doppler index interval corresponding to the Doppler interval between the Doppler peaks is _Nc -round( _Nc /(Nt+1)).

The qth Doppler demultiplexing unit 212, for example, uses the distance index f_{b_cfar}(q) is the same as multiple Doppler frequency indexes f_{s_cfar}(q) ∈{fd_#1,fd_#2…,fd_#Ns}, the q-th Doppler demultiplexing unit 212 calculates the Doppler index interval round(N_c/(Nt+1)), or the Doppler index interval (N_c-round(N_cSearch for a Doppler frequency index that matches (Nt/(Nt+1)).

The qth Doppler demultiplexing unit 212 performs the following processing based on the results of the above search.

(1) If there is a Doppler frequency index that matches the index interval round(N _c /(Nt+1)), which corresponds to the interval of the Doppler shift amount when no aliasing signal is included, the qth Doppler multiplex separation unit 212 outputs a pair of these Doppler frequency indexes (for example, represented as fd _#p , fd _#q ) as separation index information (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q)) of the Doppler multiplexed signal.

Here, when the amount of Doppler shift for transmitting antennas Tx#1 and Tx#2 has a relationship of _DOP1 < _DOP2 , the qth Doppler demultiplexing unit 212 determines the larger of fd _#p and fd _#q to be the Doppler frequency index f _{demul_Tx#2} (q) corresponding to Tx#2, and determines the lower one to be the Doppler frequency index f _{demul_Tx#1} (q) corresponding to Tx#1. On the other hand, when the amount of Doppler shift for transmitting antennas Tx#1 and Tx#2 has a relationship of _DOP1 > _DOP2 , the Doppler demultiplexing unit 212 determines the larger of fd _#p and fd _#q to be the Doppler frequency index f _{demul_Tx#1} (q) corresponding to Tx#1, and determines the lower one to be the Doppler frequency index f _{demul_Tx#2} (q) corresponding to Tx#2.

(2) If there is a Doppler frequency index that matches the index interval _Nc - round( _Nc /(Nt+1)), which corresponds to the interval of the Doppler shift amount when a return signal is included, the qth Doppler multiplex separation unit 212 outputs a pair of these Doppler frequency indexes (e.g., fd _#p , fd _#q ) as separation index information ( _{fdemul_Tx#1} (q), _{fdemul_Tx#2} (q)) of the Doppler multiplexed signal.

Here, when the amount of Doppler shift for transmitting antennas Tx#1 and Tx#2 has a relationship of _DOP1 < _DOP2 , the qth Doppler demultiplexing unit 212 determines the larger of fd _#p and fd _#q to be the Doppler frequency index f _{demul_Tx#1} (q) corresponding to Tx#1, and the smaller of fd #p and fd #q to be the Doppler frequency index f _{demul_Tx#2} (q) corresponding to Tx#2. On the other hand, when the amount of Doppler shift for transmitting antennas Tx#1 and Tx#2 has a relationship of _DOP1 > _DOP2 , the qth Doppler demultiplexing unit 212 determines the larger of fd _#p and fd _#q to be the Doppler frequency index f _{demul_Tx#2} (q) corresponding to Tx#2, and the smaller of fd #p and fd #q to be the Doppler frequency index f _{demul_Tx#1} (q) corresponding to Tx#1.

(3) If there is no Doppler frequency index that matches the index interval round( _Nc /(Nt+1)) corresponding to the interval of the Doppler shift amount when no aliasing signal is included, and if there is no Doppler frequency index that matches the index interval _Nc -round( _Nc /(Nt+1)) corresponding to the interval of the Doppler shift amount when aliasing signals are included, the q-th Doppler demultiplexing unit 212 determines that the generated Doppler peak is a noise component. In this case, the Doppler demultiplexing unit 212 may omit outputting the separation index information ( _{fdemul_Tx#1} (q), _{fdemul_Tx#2} (q)) of the Doppler multiplexed signal.

In this way, the qth Doppler multiplexing separation unit 212 can separate the Doppler multiplexed signal.

Note that while an example of Doppler multiplexing operation when Nt = 2 has been described, the number of transmitting antennas Nt is not limited to two and may be three or more. Below, as another example, the operation of the radar device 10 when Nt = 3 will be described.

In the following, as an example, a case will be described in which the phase rotation φ _n (m) shown in equation (5) is applied in the Doppler shift unit 105. Note that in the following, as an example, it is assumed that Δφ ₀ = 0, φ ₀ = 0, A = 1, and δ = 1. In this case, the Doppler shift amounts for each transmitting antenna 106 are DOP ₁ = 0, DOP ₂ = 1/(4T _rs ), and DOP ₃ = -1/(2T _rs ).

When such a Doppler shift amount is used, for example, Nt Doppler peaks (three in FIG. 7) are generated for one target Doppler frequency _{fd_TargetDoppler} to be measured, as shown in FIG. 7. Note that FIG. 7 is a diagram showing changes in Nt=3 Doppler peaks when the horizontal axis represents the target Doppler frequency and the vertical axis represents the output of the q-th Doppler analyzer 210 (FFT).

<When the target Doppler frequency is 0≦ f _{d_TargetDoppler} < 1/(2T _rs )>
7, the Doppler interval between the Doppler peak (solid line) generated when a reflected wave signal is received from a transmission signal from transmitting antenna Tx#1 and the Doppler peak (dashed line) generated when a reflected wave signal is received from a transmission signal from transmitting antenna Tx#3 is 1/( _2Trs ). In this case, when 0≦ _{fd_TargetDoppler} <1/( _4Trs ), no aliasing signals are included for any of the transmitting antennas Tx#1, Tx#2, and Tx#3, and therefore the qth Doppler demultiplexing unit 212 can determine from the low-frequency Doppler peaks that the reflected wave signals are from the transmission signals from transmitting antennas Tx#3, Tx#1, and Tx#2, respectively.

In this case, a return signal is included for Tx#2 when 1/( _4Trs )≦ _{fd_TargetDoppler} <1/( _2Trs ). Therefore, the qth Doppler demultiplexing unit 212 can determine that the higher Doppler peak (solid triangle) among the Doppler peaks spaced 1/( _2Trs ) is the reflected wave signal corresponding to transmitting antenna Tx#1, the lower Doppler peak (solid square) is the reflected wave signal corresponding to transmitting antenna Tx#3, and the remaining Doppler peaks are the reflected wave signals from transmitting antenna Tx#2.

<When the target Doppler frequency is -1/(2T _rs )≦ f _{ｄ_TargetDoppler} <0>
7, since a return signal is included for Tx#1, the Doppler interval between the Doppler peak (solid line) generated when a reflected wave signal for the transmission signal from transmitting antenna Tx#1 is received and the Doppler peak (dotted line) generated when a reflected wave signal for the transmission signal from transmitting antenna Tx#2 is received is 1/(4T _rs ). Also, the Doppler interval between the Doppler peak (dotted line) generated when a reflected wave signal for the transmission signal from transmitting antenna Tx#2 is received and the Doppler peak (constant dashed line) generated when a reflected wave signal for the transmission signal from transmitting antenna Tx#3 is received is 1/(4T _rs ).

In addition, in this case, since the transmitting antenna Tx#1 contains a return signal, the qth Doppler demultiplexing unit 212 can determine from the low-frequency Doppler peaks that they are reflected wave signals of the transmitted signals from the transmitting antennas Tx#1, Tx#2, and Tx#3, respectively.

Therefore, in the example shown in FIG. 7, the Doppler frequency range of the target in which no ambiguity occurs is, for example, −1/(2T _rs )≦f _{d — TargetDoppler} <1/(2T _rs ).

Next, we will explain an example of a method for separating signals corresponding to each transmitting antenna 106 in the qth Doppler demultiplexing unit 212.

In the following, as an example, a case will be described in which the phase rotation φ _n (m) shown in equation (5) is applied in the Doppler shift unit 105. Note that in the following, as an example, Nt=3, Δφ ₀ =0, φ ₀ =0, δ=1, and N _C is a multiple of 3. When A=1, the Doppler shift amounts for each transmitting antenna 106 are DOP ₁ =0, DOP ₂ =1/(4T _rs ), and DOP ₃ =1/(2T _rs )=-1/(2T _rs ), and when A=-1, the Doppler shift amounts for each transmitting antenna 106 are DOP ₁ =0, DOP ₂ =-1/(4T _rs ), and DOP ₃ =-1/(2T _rs ).

The q-th Doppler demultiplexing unit 212 separates the Doppler multiplexed signal using the peaks (distance index f _{b — cfar} (q) and Doppler frequency index f _{s — cfar} (q)) whose received power is greater than the threshold input from the q-th CFAR unit 211 .

For example, the q-th Doppler demultiplexing unit 212 determines which of the transmission signals transmitted from the transmitting antennas Tx#1 to Tx#Nt corresponds to a reflected wave signal for a plurality of Doppler frequency indexes _{fs_cfar} (q) having the same distance index _{fb_cfar} (q). The q-th Doppler demultiplexing unit 212 separates and outputs the reflected wave signal for each of the determined transmitting antennas Tx#1 to Tx#Nt.

The following describes the operation when there are Ns Doppler frequency indexes _{fs_cfar} (q) with the same distance index _{fb_cfar} (q). For example, let _{fs_cfar} (q)∈{fd _#1 , fd _#2 ..., fd _#Ns }.

For example, the qth Doppler demultiplexing unit 212 calculates the Doppler index interval for multiple Doppler frequency indexes _{fs_cfar} (q)∈{fd _#1 , fd _#2 ..., fd _#Ns } having the same distance index _{fb_cfar} (q).Then, the qth Doppler demultiplexing unit 212 searches for a combination of Doppler frequency indexes such that the interval between two Doppler indexes, when viewed in ascending order, matches the index interval corresponding to the interval between Doppler shift amounts when no aliasing signals are included.Alternatively, the qth Doppler demultiplexing unit 212 searches for a combination of Doppler frequency indexes such that the interval between two Doppler indexes, when viewed in ascending order, matches the index interval corresponding to the interval between Doppler shift amounts when aliasing signals are included.

The qth Doppler demultiplexing unit 212 performs the following processing based on the results of the above search.

(1) If there is a combination of Doppler frequency indexes that does not include a return signal and matches the index interval corresponding to the interval of the Doppler shift amount, the qth Doppler multiplex separation unit 212 outputs the set of Doppler frequency indexes (for example, represented as fd _#p1 , fd _#p2 , fd _#p3 ) as separation index information of the Doppler multiplexed signal (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q), f _{demul_Tx#3} (q)).

Here, if the Doppler shift amounts for transmitting antennas Tx#1 to Tx#3 are in the relationship _DOP3 < _DOP1 < _DOP2 , the qth Doppler multiplex separation unit 212 determines the Doppler frequency indexes f demul_Tx _#2 (q), f _{demul_Tx#1 (} q), and f _{demul_Tx#3} ( _q ) corresponding to Tx#2, Tx#1, and Tx#3, respectively, from the larger of fd #p1, fd #p2, and fd _#p3 (0≦f _{d_TargetDoppler} < 1/(4T _rs ) in Figure 7). Furthermore, if the Doppler shift amounts for transmitting antennas Tx#1 to Tx#3 are in the relationship DOP ₁ > DOP ₂ > DOP ₃ , the qth Doppler multiplexing separation unit 212 determines the Doppler frequency indexes f demul_Tx _{#1 ( q)} , f _{demul_Tx#2 (} q), and f _{demul_Tx#3} (q) corresponding to Tx#1, Tx#2, and Tx#3, respectively, from the larger of fd #p1, fd #p2, and fd _# p3.

(2) If there is a combination of Doppler frequency indexes that includes a return signal and matches the index interval corresponding to the interval of the Doppler shift amount, the qth Doppler multiplex separation unit 212 outputs the set of Doppler frequency indexes (for example, represented as fd _#q1 , fd _#q2 , fd _#q3 ) as separation index information of the Doppler multiplexed signal (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q), f _{demul_Tx#3} (q)).

For example, if the Doppler shift amounts for transmitting antennas Tx#1 to Tx#3 are in the relationship DOP ₃ < DOP ₁ < DOP ₂ , and there is a combination of Doppler frequency indices where the Doppler frequency corresponding to DOP ₃ is a aliasing signal, the qth Doppler multiplex separation unit 212 determines the Doppler frequency indexes f demul_Tx# ₂ (q), f _{demul_Tx} # ₁ (q), and f _{demul_Tx#3} (q) corresponding to Tx#3, Tx#2, and Tx#1, respectively, from the largest of fd _# q1, fd #q2, and fd _# q3 (-1/(2T _rs ) ≦ f _{ｄ_TargetDoppler} < 0 in Figure 7). Furthermore, if the Doppler shift amounts for transmitting antennas Tx#1 to Tx#3 are in the relationship DOP ₁ > DOP ₂ > DOP ₃ and there is a combination of Doppler frequency indexes where the Doppler frequency corresponding to DOP ₃ is a aliasing signal, the qth Doppler multiplex separation unit 212 determines the largest of fd _#q1 , fd _#q2 , and fd _#q3 to be f _{demul_Tx#} 2 (q), f demul_Tx#3 (q), and f _{demul_Tx #1 (q) corresponding to Tx#3, Tx#} 1, and Tx#2, respectively.

(3) If there is a combination of Doppler frequency indexes that includes a return signal and matches the index interval corresponding to the interval of the Doppler shift amount, the qth Doppler multiplex separation unit 212 outputs the set of Doppler frequency indexes (for example, represented as fd _#u1 , fd _#u2 , fd _#u3 ) as separation index information of the Doppler multiplexed signal (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q), f _{demul_Tx#3} (q)).

Here, if the Doppler shift amounts for transmitting antennas Tx#1 to Tx#3 are in the relationship _DOP3 < _DOP1 < _DOP2 , and there is a combination of Doppler frequency indexes where the Doppler frequency corresponding to _DOP2 is a aliasing signal, the qth Doppler multiplex separation unit 212 determines f _{demul_Tx#3 (q)} , f _{demul_Tx#2 (q)} , f _{demul_Tx#1} (q) corresponding to Tx#1, Tx#3, and Tx#2, respectively, from the larger of _{fd #u1, fd #u2} , and fd _#u3 (1/(4T _rs ) ≦ f _{ｄ_TargetDoppler} < 1/(2T _rs ) in Figure 7).

Furthermore, if the Doppler shift amounts for transmitting antennas Tx#1 to Tx#3 are in the relationship DOP ₁ > DOP ₂ > DOP ₃ and there is a combination of Doppler frequency indexes in which the Doppler frequency corresponding to DOP ₁ is a folded signal, the qth Doppler multiplex separation unit 212 determines f demul_Tx _{#3 (q)} , f demul_Tx# _{1 (q)} , and f _{demul_Tx#2} (q) corresponding to Tx#2, Tx#3, and Tx#1, respectively, from the larger of fd _#u1 , fd _#u2 , and fd #u3.

(4) The q-th Doppler demultiplexing unit 212 determines that a Doppler peak corresponding to a Doppler frequency index that does not fall under any of the above (1), (2), or (3) is a noise component. In this case, the q-th Doppler demultiplexing unit 212 may omit outputting the separation index information (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q), f _{demul_Tx#3} (q)) of the Doppler multiplexed signal.

In this way, the Doppler multiplexing separation unit 212 can separate the Doppler multiplexed signal.

In addition, the case where the phase rotation φ _n (m) shown in equation (5) is used as an example of the phase rotation corresponding to the Doppler shift amount DOP _n applied to the transmission signal has been described. However, the phase rotation is not limited to the phase rotation φ _n (m) shown in equation (5).

As another example, the nth Doppler shifter 105 may impart a phase rotation φ n (m) of the following equation (7) to the input mth chirp signal (transmission signal), which results in a Doppler shift amount _{DOP n} different from that obtained when equation (5) is used. Note that the term round(N _C /(Nt + δ)) is introduced for the purpose of making the phase rotation amount an integer multiple of the Doppler frequency interval in the Doppler analyzer 210, but this is not limitative, and 2π/Nt may be used instead of the term (2π/N _C ) × round(N _C /Nt) in equation (7).

Here, _dpn is a component that causes the phase rotation to be unevenly spaced within the Doppler frequency range. For example, _dp1 , _dp2 , ..., _dpNt are values within the range of -round( _Nc /Nt)/2 < _dpn < round( _Nc /Nt)/2, and do not all have the same value, but include at least one component with a different value. Note that the term round( _Nc /Nt) is introduced for the purpose of making the amount of phase rotation an integer multiple of the Doppler frequency interval in the Doppler analysis unit 210.

As an example, in equation (7), when a phase rotation _φn (m) is applied with Nt=2, _Δφ0 =0, _φ0 =0, A= ₁ , dp1=0, and _dp2 =π/5, the Doppler shift amounts are _DOP1 =0, _DOP2 =1/( _2Trs )+1/( _10Trs )=6/( _10Trs )=-4/( _10Trs ).

8 is a diagram showing the change in Doppler peak when Nt = 2, _DOP1 = 0, and _DOP2 = -4/( _10Trs ), with the horizontal axis representing the target Doppler frequency and the vertical axis representing the output of the Doppler analysis unit 210 (FFT). In this case, _DOP1 > _DOP2 .

<When the target Doppler frequency is -1/(10T _rs )≦ f _{d_TargetDoppler} < 1/(2T _rs )>
As shown in Figure 8, the Doppler interval between the Doppler peak (solid line) that occurs when a reflected wave signal is received for a transmission signal from transmitting antenna Tx#1 and the Doppler peak (dotted line) that occurs when a reflected wave signal is received for a transmission signal from transmitting antenna Tx#2 is 4/(10T _rs ).

In this case, neither of the transmitting antennas Tx#1 and Tx#2 includes a return signal.
Therefore, the qth Doppler multiplex separation unit 212 can determine that the higher Doppler peak (solid triangle) among the Doppler peaks whose spacing is 4/(10T _rs ) is the reflected wave signal corresponding to transmitting antenna Tx#1, and the lower Doppler peak (solid square) is the reflected wave signal corresponding to transmitting antenna Tx#2.

<When the target Doppler frequency is -1/(2T _rs )≦ f _{ｄ_TargetDoppler} ＜ -1/(10T _rs )>
As shown in Figure 8, the Doppler interval between the Doppler peak (solid line) that occurs when a reflected wave signal is received for a transmission signal from transmitting antenna Tx#1 and the Doppler peak (dotted line) that occurs when a reflected wave signal is received for a transmission signal from transmitting antenna Tx#2 is 6/(10T _rs ).

In this case, a return signal is included for Tx#2 (solid circle). Therefore, the qth Doppler demultiplexing unit 212 can determine, for example, from the low-frequency Doppler peaks (dotted triangles) that they are reflected wave signals from the transmission signals from the transmitting antennas Tx#1 and Tx#2, respectively.

Therefore, in the example shown in FIG. 8, the target Doppler frequency range in which no ambiguity occurs is, for example, −1/(2T _rs )≦f d _{— TargetDoppler} <1/(2T _rs ).

The above describes an example of the operation of the Doppler demultiplexing unit 212. Note that in the above description of the example of the operation of the Doppler demultiplexing unit 212, the radar device 10 performs unequal Doppler multiplexing on the first chirp signal and the second chirp signal at the same (unequally divided) Doppler multiplexing interval. However, this is not limited to this, and the radar device 10 may perform unequal Doppler multiplexing on the first chirp signal and the second chirp signal using different parameters, such as different (unequally divided) Doppler multiplexing intervals.

For example, when Nt=2, the Doppler shift amount for each transmitting antenna 106 in the Doppler shift unit 105 for the first chirp signal may be set to DOP ₁ =0 and DOP ₂ =1/( _3Tr ), and the Doppler shift amount for each transmitting antenna 106 in the Doppler shift unit 105 for the second chirp signal may be set to DOP ₁ =0 and DOP ₂ =1/( _4Tr ), and unequal interval Doppler multiplexing may be performed using parameters that result in different Doppler multiplexing intervals between the first chirp signal and the second chirp signal.

Alternatively, for example, when Nt=2, the Doppler shift amount for each transmitting antenna 106 in the Doppler shift unit 105 may be set to DOP ₁ =0 and DOP ₂ =1/( _3Tr ) for the first chirp signal, and the Doppler shift amount for each transmitting antenna 106 in the Doppler shift unit 105 may be set to DOP ₁ =1/( _4Tr ) and DOP ₂ =-1/( _2Tr ) for the second chirp signal, and unequal interval Doppler multiplexing may be performed using parameters that result in different Doppler multiplexing intervals between the first chirp signal and the second chirp signal.

Even in such a case, the radar receiving unit 200 processes the first chirp signal in the first Doppler analysis unit 210 and the first Doppler demultiplexing unit 212, and processes the second chirp signal independently in the second Doppler analysis unit 210 and the second Doppler demultiplexing unit 212, so the operations described above can be applied. In this way, since the radar device 10 processes each chirp signal individually, it is not necessary to share the parameters for performing uneven Doppler multiplexing between chirp signals.

2 , the Doppler determination unit 213 determines the Doppler frequency corresponding to the Doppler peak based on the outputs of the first Doppler demultiplexing unit 212 and the second Doppler demultiplexing unit 212. For example, even when a target having a Doppler frequency _{fd_TargetDoppler} outside the Doppler frequency range −1/( _2Trs )≦ _{fd_TargetDoppler} <1/( _2Trs ) is included, the Doppler determination unit 213 can further expand the Doppler detection range by determining the Doppler frequency of the target.

For example, the Doppler determination unit 213 determines the Doppler frequency of a target including a Doppler frequency exceeding the Doppler frequency range -1/( _{2T rs ) ≦ f d_TargetDoppler <} 1/(2T rs ) using the separation index information (f _{demul_Tx#1} (1), f _{demul_Tx#2} (1), ..., f _{demul_Tx#Nt} (1)) of the Doppler multiplexed signal output from the first Doppler multiplex separation unit 212 and the separation index information (f _{demul_Tx#1} (2), f _{demul_Tx#2} (2), ..., f _{demul_Tx#Nt} (2)) of the Doppler multiplexed signal output from the second _Doppler multiplex separation unit ₂₁₂ , which have the same distance indexes f b_cfar (1) and _{f b_cfar} (2).

The Doppler frequency determination unit 213 determines the Doppler frequency by utilizing the fact that the center frequencies of the first chirp signal and the second chirp signal, which are radar transmission signals generated by the signal generation control unit 104 and the radar transmission signal generation unit 101, are different from each other.

The operating principles of the Doppler frequency determination process and an example of the operation of the Doppler determination unit 213 are explained below.

In the following, an example will be described in which the Doppler determination unit 213 performs processing using separation _index information of the Doppler _multiplexed signals output from the first Doppler multiplex separation unit 212 and the second Doppler multiplex separation unit 212, in which the distance indexes f b_cfar (1) and f b_cfar (2) are common. For this reason, in the following, the distance index will be abbreviated as "f _{b_cfar} " (= f _{b_cfar} (1) = f _{b_cfar} (2)).

For example, if the center frequency of the first chirp signal and the center frequency of the second chirp signal differ, the Doppler frequency of the reflected wave also changes. For example, if the radar device 10 is stationary and a target is moving toward the radar device 10 at a speed v, the Doppler frequency _fd (1) observed using the first chirp signal is _fd (1)=2v× _fc (1)/ _C0 , and the Doppler frequency _fd (2) observed using the second chirp signal is _fd (2)=2v× _fc (2)/ _C0 . Therefore, the relationship between the two Doppler frequencies is expressed as _fd (2)/ _fd (1)= _fc (2)/ _fc (1). For example, f _d (2) can be calculated by multiplying f _d (1) by the center frequency ratio f _c (2)/f _c (1) (f _d (2)=(f _c (2)/f _c (1))×f _d (1)), where C ₀ represents the speed of light.

Furthermore, for example, if the Doppler frequency _{fd_TargetDoppler} of the target is assumed to exceed the Doppler frequency range -1/( _2Trs ) ≦ _{fd_TargetDoppler} < 1/( _2Trs ), and the Doppler frequency estimate value detected in the first Doppler analyzer 210 and the first Doppler demultiplexer 212 is _{fd_VFT} (1), the Doppler frequency _{fd_TargetDoppler} of the target is expressed by the following equation (8), taking Doppler aliasing into consideration: Here, n _al represents the number of Doppler aliasing and takes an integer value.
f _{d_TargetDoppler} = f _{d_VFT} (1)+ n _al /T _rs (8)

Since it is difficult to determine the number of Doppler aliasings n _al from the outputs of the first Doppler analysis unit 210 and the first Doppler demultiplexing unit 212, the conditions under which it can be determined using the outputs of the second Doppler analysis unit 210 and the second Doppler demultiplexing unit 212 are derived below. Here, the Doppler frequency observed using the second chirp signal is obtained by multiplying equation (8) by the center frequency ratio f _c (2)/f _c (1), as shown in the following equation (9).
f _c (2)/f _c (1)×f _{d_TargetDoppler} = f _c (2)/f _c (1)×(f _{d_VFT} (1)+ n _al /T _rs ) (9)

For example, the Doppler frequency observed using the second chirp signal is f _c (2)/f _c (1)×f _{d_VFT} (1) when the Doppler aliasing number n _al =0, f _c (2)/f _c (1)×(f _{d_VFT} (1)+1/T _rs ) when the Doppler aliasing number n _al =1, and f _c (2)/f _c (1)×(f _{d_VFT} (1)-1/T _rs ) when the Doppler aliasing number n _al =-1. The same applies to other Doppler aliasing numbers.

In this way, the Doppler frequency difference due to the difference in the number of Doppler aliasing times n _al has a relationship of being an integer multiple of f _c (2)/f _c (1)/T _rs .

Here, if the difference between f _c (2)/f _c (1)/T _rs and 1/T _rs , which is the aliasing frequency interval of the second Doppler analysis unit 210, is larger than the Doppler frequency resolution Δf _d in the second Doppler analysis unit 210, the Doppler determination unit 213 becomes able to detect (e.g., determine) the Doppler frequency difference due to differences in the number of Doppler aliasing times n _al .

Therefore, for example, the radar device 10 (e.g., the signal generation control unit 104) may determine the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal so as to satisfy the condition (also called the determinable condition) shown in the following equation (10).

Here, for example, when f _c (2)>f _c (1), the determination condition for f _c (1) and f _c (2) shown in equation (10) is expressed by the following equation (11).

Furthermore, for example, when f _c (2)<f _c (1), the determination condition for f _c (1) and f _c (2) shown in equation (10) is expressed by the following equation (12).

Furthermore, for example, the determination condition shown in the following equation (13) may be used, where α≧1.

Equation (13) specifies the conditions for the center frequencies f _c (1) and f c (2) that satisfy the condition that the difference between f _c (2 ₎ /f _c (1)/T _rs and 1/T _rs , which is the aliasing frequency interval of the second Doppler analyzer 210, is greater than an integer multiple α of the Doppler frequency resolution Δf _d in the second Doppler analyzer 210. The determination conditions shown in equation (13) enable the Doppler determination unit 213 to more easily determine the Doppler frequency difference due to differences in the number of Doppler aliasing n _al , compared to the determination conditions shown in equation (10). For example, the determination conditions shown in equation (13) enable improved determination accuracy even when there is noise influence, such as when the received signal level is low, compared to the determination conditions shown in equation (10).

Here, for example, when f _c (2)>f _c (1), the determination condition for f _c (1) and f _c (2) shown in equation (13) is expressed by the following equation (14).

Furthermore, for example, when f _c (2)<f _c (1), the determination condition for f _c (1) and f _c (2) shown in equation (13) is expressed by the following equation (15).

As an example, when f _c (1)=78 GHz, assuming that f _c (2)>f _c (1), based on equation (14), f _c (2) is set to be greater than 78.61 GHz when α=1 and Nc=128, and f _c (2) is set to be greater than 79.22 GHz when α=2 and Nc=128.

In this way, by setting the center frequencies f _c (1) and f _c (2) that satisfy any of the determination conditions of equations (10) to (15), the Doppler determination unit 213 can determine the Doppler frequency of the target even when the target includes a Doppler frequency that exceeds the Doppler frequency range -1/(2T _rs ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ) (for example, when Doppler aliasing occurs).

For example, the Doppler determination unit 213 may perform the following Doppler determination processing using separation index information (f _{demul_Tx #1} (1), f _{demul_Tx#2} (1), to f _{demul_Tx#Nt} (1)) of the Doppler multiplexed signal in the distance index f _{b_cfar} (1) output from the first Doppler multiplex separation unit 212, and separation index information (f _{demul_Tx#1} (2), f _{demul_Tx#2} (2) to f _{demul_Tx#Nt} (2)) of the Doppler multiplexed signal in the distance index f b_cfar (2) output from the second Doppler multiplex separation unit 212.

For example, when there is one separation index information (f _{demul_Tx#1} (1), f demul_Tx _{#2 (1) to f demul_Tx#Nt} (1)) of the Doppler multiplexed signal in the distance index f _{b_cfar} (1) output from the first Doppler multiplexed separation unit 212, and there is one separation index information (f _{demul_Tx# 1 (2), f demul_Tx #2} (2) to f _{demul_Tx#Nt} (2)) of the Doppler multiplexed signal in the distance index f _{b_cfar} (2) output from the second Doppler multiplexed separation unit 212, the Doppler determination unit 213 may perform the following Doppler determination operation.

Furthermore, for example, when there is a plurality of pieces of separation index information for Doppler multiplexed signals in the distance index f _{b_cfar} (1) output from the first Doppler multiplexed separation unit 212, and there is a plurality of pieces of separation index information for Doppler multiplexed signals in the distance index f _{b_cfar} (2) output from the second Doppler multiplexed separation unit 212, the Doppler determination unit 213 may compare the received power indicated by these indexes and associate the separated index information with similar received power levels as respective pairs.

The Doppler determination unit 213 may then perform the following Doppler determination operation using the separation index information of the Doppler multiplexed signal output from the first Doppler multiplex separation unit 212 and the separation index information of the Doppler multiplexed signal output from the second Doppler multiplex separation unit 212, which are associated as a pair. For example, the Doppler determination unit 213 may sequentially perform the following Doppler determination operation for each associated pair of separation index information of the Doppler multiplexed signal output from the first Doppler multiplex separation unit 212 and separation index information of the Doppler multiplexed signal output from the second Doppler multiplex separation unit 212, and repeat the following operation for all pairs until it has been completed.

An example of the Doppler determination operation performed by the Doppler determination unit 213 is described below.

First, the Doppler determination unit 213 calculates a Doppler frequency estimate value f _{d_VFT} (1) based on the separation index information (f _{demul_Tx#1} (1), f _{demul_Tx#2} (1), ..., f _{demul_Tx#Nt} (1)) of the Doppler multiplexed signal in the distance index f b_cfar (1) output from the first Doppler multiplex separation unit ₂₁₂ , assuming that the Doppler frequency of the target is within the Doppler frequency range -1/(2T _rs ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ).

Here, the separation index information (f _{demul_Tx#1} (1), f _{demul_Tx#2} (1) to f _{demul_Tx#Nt} (1)) of the Doppler multiplexed signal in the distance index f _{b_cfar} (1) output from the first Doppler multiplex separation unit 212 includes a predetermined Doppler shift component assigned to each transmitting antenna 106 by the radar transmitter 100. For example, the Doppler determination unit 213 may calculate a Doppler frequency estimation value f _{d_VFT} (1) from which the Doppler shift component has been removed.

Similarly, the Doppler determination unit 213 calculates a Doppler frequency estimate f _{d_VFT} (2) based on the separation index information (f _{demul_Tx#1} (2), f _{demul_Tx#2} (2) to f _{demul_Tx#Nt} (2)) of the Doppler multiplexed signal in the distance index f b_cfar (2) output from the second Doppler multiplex separation unit 212, assuming that the Doppler frequency of the target is within the _Doppler frequency range -1/(2T _rs ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ).

Here, the separation index information (f _{demul_Tx#1} (2), f _{demul_Tx#2} (2) to f _{demul_Tx#Nt} (2)) of the Doppler multiplexed signal in the distance index f _{b_cfar} (2) output from the second Doppler multiplex separation unit 212 includes a component of a predetermined Doppler shift amount assigned to each transmitting antenna 106 by the radar transmitter 100. For example, the Doppler determination unit 213 may calculate a Doppler frequency estimation value f _{d_VFT} (2) from which the component of the Doppler shift amount has been removed.

Next, the Doppler determination unit 213 calculates the Doppler aliasing number n _al that minimizes the following equation (16).

Here, the Doppler aliasing number n _al is an integer value, and is calculated within a range of integer values that covers the Doppler frequency range of the expected target.

Furthermore, f _est (f _{d_VFT(1)} , n _al ) represents a function that calculates the Doppler frequency (f _{d_VFT (1)} +n _al /T _rs ) when the Doppler frequency estimate f d_VFT (1) has a Doppler aliasing count of n _al , as shown in the following equation (17), and outputs a value obtained by multiplying this by f _c (2)/f _c (1) to convert it into a Doppler frequency observed using a second chirp signal with a center frequency of _f c (2). For example, f _est (f _{d_VFT(1)} , n _al ) represents the Doppler frequency estimate corresponding to the second chirp signal estimated based on the Doppler frequency estimate f _{d_VFT} (1).

In equation (17), Fmod[x] is a function that calculates the Doppler frequency taking into account aliasing of ±½T _rs in the Doppler analysis unit 210, and when x≧1/(2T _rs ), it calculates nmod=floor((x−1/(2T _rs ))/T _rs )+1 and outputs x−nmod/T _rs . When x<−1/(2T _rs ), Fmod[x] calculates nmod=ceil((|x|−1/(2T _rs ))/T _rs ) and outputs x+nmod/T _rs . Here, floor(x) is a floor function, which outputs the largest integer value that does not exceed x. Also, ceil(x) is a ceiling function, which outputs the smallest integer value that exceeds x.

Hereinafter, the Doppler aliasing number n _al when the Doppler frequency f _est (f _{d_VFT(1)} , n _al ) is varied and the Doppler frequency estimated value f _{d_VFT} (2) matches (or matches, or is close to) the highest (for example, n _al when equation (16) is minimized ₎ will be referred to as the "Doppler aliasing number estimated value n _alist ."

Next, an example of the above-mentioned Doppler determination operation will be described using Figure 9. In Figure 9, the horizontal axis represents the Doppler frequency of the target, and the vertical axis represents the Doppler frequency estimate based on the outputs of the first Doppler demultiplexing unit 212 and the second Doppler demultiplexing unit 212.

9, the solid line represents the Doppler frequency estimate (e.g., f _{d_VFT} (1)) when a first chirp signal with a center frequency of f _c (1) is used, and the dotted line represents the Doppler frequency estimate (f _{d_VFT} (2)) when a second chirp signal with a center frequency of f _c (2) is used. However, FIG. 9 shows an example where f _c (2)>f _c (1).

9, the circles represent Doppler frequency estimates for the Doppler frequency f _{d_VFT} (1)+n _al /T _rs based on the output of the first Doppler demultiplexing unit 212. As shown in FIG. 9, the Doppler frequency estimates (values on the vertical axis) indicated by the circles are values (e.g., f _{d_VFT} (1)) that do not depend on the number of Doppler aliasing n _al .

9, square marks represent Doppler frequency estimates based on the output of the second Doppler demultiplexing unit 212 for the Doppler frequency _{fd_VFT} (1)+n _al /T _rs . As shown in FIG. 9, the Doppler frequency estimates (values on the vertical axis) indicated by the square marks are values that depend on the Doppler aliasing count n _al (e.g., f _est ( _{fd_VFT(1)} , n _al )). For example, the intervals between the square marks along the vertical axis in FIG. 9 are greater than the Doppler frequency resolution _Δfd in the second Doppler analysis unit 210, and different Doppler frequency estimates can be detected depending on the Doppler aliasing count n _al .

The Doppler determination unit 213 may, for example, calculate the square plot shown in Fig. 9 based on the Doppler frequency estimate _{fd_VFT} (1) using _fest ( _{fd_VFT(1)} , _nal ) shown in equation (17). Then, the Doppler determination unit 213 may set the Doppler aliasing number _nal (e.g., nal at which equation (16) is minimized) that most closely matches the calculated value (the value on the vertical axis of the square plot shown in Fig. 9) with the Doppler frequency estimate _{fd_VFT} (2) (the dotted _line shown in Fig. 9) when the second chirp signal with center frequency _fc (2) is used, as the Doppler aliasing number estimate _nalest .

In this way, the Doppler determination unit 213 may estimate the Doppler peak of the reflected wave signal corresponding to the second chirp signal having the center frequency f _c (2) (e.g., the square mark in FIG _{. 9 , the second peak position) based on the Doppler peak observed in the reflected wave signal corresponding to the first chirp signal having the center frequency f c ( 1} ) (e.g., the circle mark in FIG. 9 , the first peak position) and the ratio f _c (2)/f c (1) between the center frequencies f _c (1) and f c (2), and determine the number of aliasing n aless based on the degree of match (closeness) between the estimated Doppler peak and the Doppler peak observed in the reflected wave _signal corresponding to the second chirp signal (e.g., the dotted line in FIG. 9 , the third peak position). Note that the degree of match is a closeness that takes into account the fact that the value of the Doppler peak changes due to aliasing.

Note that the larger f _c (2)/f _c (1) when f _c (2)>f _c (1), or the smaller f _c (2)/f _c (1) when f _c (2)<f _c (1), the larger the difference in f _est (f _{d_VFT(1)} , n _al ) according to the number of aliases n _al (e.g., the difference from f _{d_VFT(1)} ), and therefore the Doppler determination unit 213 can more easily distinguish Doppler frequencies according to the number of aliases n _al (e.g., determine the number of aliases n _al ).

On the other hand, when the difference in Doppler frequency due to the number of aliases n _al becomes large and exceeds ±1/(2T _rs ), ambiguity in the Doppler frequency occurs, which may increase the probability of an error in the estimated number of Doppler aliases n _alist .

Here, when the number of Doppler aliasings is n _al , the condition under which the difference between the aliasing component n _al × _{f c (} 2)/f _c (1) /T _rs of the Doppler frequency observed using the second chirp signal with center frequency f c (2) and n _al /T rs, which is the frequency interval of the number of aliasings n _al in the second Doppler analysis unit ₂₁₀ , does not exceed ±1/(2T _rs ), is expressed by the following equation (18).

For example, when n _al is positive, Δn _al is in the range of ±1/(2T _rs ) up to the maximum n _al that satisfies the following equation (19), and the Doppler determination unit 213 can estimate aliasing without ambiguity. The maximum n _al that satisfies the following equation (19) is denoted as "n _almax ". Also, for example, when n _al is negative, n _al = -n _almax similarly satisfies the following equation (19). From these facts, the detection range of the Doppler frequency is expanded by n _almax times, for example, compared to the Doppler frequency range when one transmitting antenna is used.

For example, when n _almax = 2, if f _c (2) > f _c (1), f _c (2) may be set to a frequency lower than 1.25 times f _c (1), and if f _c (2) < f _c (1), f _c (1) may be set to a frequency lower than 1.25 times f _c (2). Furthermore, for example, when n _almax = 3, if f _c (2) > f _c (1), f _c (2) may be set to a frequency lower than (7/6) times f _c (1), and if f _c (2) < f _c (1), f _c (1) may be set to a frequency lower than (7/6) times f _c (2). Note that the value of n _almax is not limited to 2 or 3 and may be other values.

For example, if f _c (1) and f _c (2) satisfy the condition for n _almax = 2, the detection range of the Doppler frequency f _d becomes ±2/(T _rs ), which is twice the Doppler frequency range when one transmitting antenna is used.

Note that f _c (1) and f _c (2) may be set within the pass frequency range of the radar transmitter 100 or the radar receiver 200, and the expansion of the Doppler frequency detection range may also be restricted by the pass frequency characteristics of the radar transmitter 100 or the radar receiver 200.

For example, in setting f _c (1) and f _c (2), the maximum number of Doppler aliasing n _almax may be determined based on the maximum Doppler frequency of an assumed target, and f _c (1) and f _c (2) that satisfy the determination condition and equation (17) may be determined. Furthermore, f _c (1) and f _c (2) may be determined so that they fall within the passing frequency range of the _radar transmitter 100 or the _radar receiver 200.

The above explains an example of the operation of the Doppler determination unit 213.

In Figure 2, the direction estimation unit 214 performs target direction estimation processing based on information input from the qth Doppler multiplex separation unit 212 (e.g., distance index f _{b_cfar} (q) and separation index information of the Doppler multiplex signal (q) (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q) to f _{demul_Tx#Nt} (q))).

For example, the direction estimation unit 214 extracts the output of the qth Doppler analysis unit 210 from the output of the qth Doppler multiplex separation unit 212 based on the distance index f _{b_cfar} (q) and separation index information (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q) to f _{demul_Tx#Nt} (q)) of the Doppler multiplexed signal (q), generates the qth virtual receiving array correlation vector h _q (f _{b_cfar} (q), f _{demul_Tx#1} (q), f _{demul_Tx#2} (q) to f _{demul_Tx#Nt} (q)) as shown in the following equation (20), and performs direction estimation processing, where q=1, 2.

As shown in equation (20), the qth virtual receiving array correlation vector h _q (f _{b_cfar} (q), f _{demul_Tx#1} (q), f demul_Tx _#2 (q) to f _{demul_Tx#Nt} (q)) includes Nt×Na elements, which is the product of the number of transmitting antennas Nt and the number of receiving antennas Na. The virtual receiving array correlation vector h _q (f _{b_cfar} (q), f _{demul_Tx#1} (q), f _{demul_Tx#2} (q) to f _{demul_Tx#Nt} (q)) is used in the process of performing direction estimation based on the phase difference between each receiving antenna 202 with respect to the wave signal reflected from the target. Here, the integer z is 1 to Na. Note that the direction estimation unit 214 performs direction estimation processing using the outputs of the first and second Doppler demultiplexing units 212 with the same distance index, so f _{b_cfar} (1) = f _{b_cfar} (2) = f _{b_cfar} .

In equation (20), h _cal[b] is an array correction value that corrects the phase deviation and amplitude deviation between the transmitting array antennas and the receiving array antennas, where b is an integer between 1 and (Nt×Na).

The direction estimation unit 214 calculates a spatial profile by varying the azimuth direction θu in the direction estimation evaluation function value P _H ( _θu , _{fb_cfar} , _{fdemul_Tx#1} (1) to _{fdemul_Tx#Nt} (1), fdemul_Tx _#1 (2) to _{fdemul_Tx#Nt (} 2)) within a predetermined angle range. The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial profile in descending order, and outputs the azimuth directions of the maximum peaks as direction-of-arrival estimates (e.g., positioning outputs).

Note that there are various methods for calculating the direction estimation evaluation function value P _H (θ _u , f _{b_cfar} , f _{demul_Tx#1} (1) to f demul_Tx# _Nt (1), f _{demul_Tx#1} (2) to f _{demul_Tx#Nt} (2)) depending on the direction of arrival estimation algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

For example, when Nt × Na virtual receiving arrays are arranged linearly at equal intervals _dH , the beamformer method can be expressed as in the following equation (21). In addition to the beamformer method, methods such as Capon and MUSIC can also be applied. In equation (21), the superscript H is the Hermitian transpose operator.

In addition, in equation (21), a _q (θ _u ) indicates the direction vector of the virtual receiving array for the arriving wave in the azimuth direction θ _u , and is expressed by equation (22). In equation (22), λ _q is the wavelength of the radar transmission signal (for example, the qth chirp signal) when the center frequency is f _c (q), and λ _q =C ₀ /f _c (q).

The azimuth direction θ _u is a vector obtained by varying the azimuth range in which the direction of arrival estimation is performed at a predetermined azimuth interval β _1. For example, θ _u is set as follows.
θ _u ＝θmin + uβ ₁ , integer u=0～NU
NU=floor[(θmax-θmin)/β ₁ ]+1
Here, floor(x) is a function that returns the maximum integer value that does not exceed the real number x.

For example, instead of the direction vector _aq ( _θu ) shown in equation (22), the direction estimator 214 may commonly use the direction vector a( _θu ) of the virtual receiving array for the arriving wave in the azimuth direction θ at the average center frequency of the center frequencies _fc (1) and _fc (2), as shown in equation (23). Here, _λa is the wavelength of the radar transmission signal when the center frequency is ( _fc (1)+ _fc (2))/2, and _λa = _2C0 /( _fc (1)+ _fc (2)). In this case, the direction estimator 214 can commonly use the direction vector a( _θu ) of the virtual receiving array for processing each chirp signal, which also has the effect of reducing the memory capacity required to store the direction vectors of the virtual receiving array.

In the above example, the direction estimation unit 214 calculates the azimuth direction as the direction of arrival estimate, but this is not limited to this. Direction of arrival estimation can also be performed in the elevation direction, or by using MIMO antennas arranged in a rectangular grid, direction of arrival estimation can also be performed in the azimuth direction and elevation direction. For example, the direction estimation unit 214 may calculate the azimuth direction and elevation direction as the direction of arrival estimate, and use these as the positioning output.

Through the above operations, the direction estimation unit 214 may output, as positioning outputs, distance index f _{b_cfar} (q) and arrival direction estimation values in separation index information of Doppler multiplexed signals (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q), ..., f _{demul_Tx#Nt} (q)). The direction estimation unit 214 may further output, as positioning outputs, distance index f _{b_cfar} (q) and separation index information of Doppler multiplexed signals (f _{demul_Tx#1} (q), f _{demul_Tx#2} (q), ..., f _{demul_Tx#Nt} (q)). The direction estimation unit 214 may output the positioning output (or the positioning result) to, for example, a vehicle control device in the case of an on-board radar, or an infrastructure control device in the case of an infrastructure radar, both not shown.

In addition, the direction estimation unit 214 may output, for example, either one or both of the Doppler frequency information f _{d_VFT(1)} +n _alest /T _rs determined by the Doppler determination unit 213 and f _c (2)/f _c (1)(f _{d_VFT(1)} +n _alest /T _rs ).

Furthermore, the distance index f _{b_cfar} may be converted into distance information using equation (1) and output.

Furthermore, the Doppler frequency information determined by the Doppler determination unit 213 may be converted into relative velocity information and output. The Doppler frequency information f _{d — VFT(1)} + n _alest /T _rs based on the center frequency f _c (1) determined by the Doppler determination unit 213 can be converted into the relative velocity v _d using the following equation (24).

Similarly, when the Doppler frequency information f _c (2)/f _c (1) (f _{d_VFT(1)} +n _alest /T _rs ) determined by the Doppler determination unit 213 based on the center frequency f _c (2) is converted into relative velocity v _d , the value obtained is the same as equation (24), as shown in the following equation (25), and therefore the relative velocity component information may be output as a common value (or a unified value) for different center frequencies.

As described above, in this embodiment, the radar device 10 alternates between, for example, a first center frequency and a second center frequency that satisfy one of equations (10) to (15) for each transmission period in which a transmission signal is transmitted from the transmitting antenna 106. This allows the radar device 10 to determine the number of aliasings based on the deviation in Doppler frequency in Doppler analysis that corresponds to differences in center frequency. Therefore, the radar device 10 can expand the Doppler frequency range (or the maximum value of relative velocity) in which Doppler multiplexed signals can be separated, for example, depending on the number of aliasings that can be determined.

As described above, according to this embodiment, the Doppler frequency range (or maximum value of relative velocity) in which no ambiguity occurs can be expanded. This allows the radar device 10 to accurately detect targets (e.g., direction of arrival) over a wider Doppler frequency range.

Furthermore, in this embodiment, since the Doppler frequency range in which a Doppler multiplexed signal can be separated is expanded by setting the center frequency of the chirp signal, it is possible to omit the application of a method such as increasing the sampling rate of the A/D converter. Therefore, this embodiment can prevent the hardware configuration of the radar device 10 from becoming complicated and also prevent an increase in power consumption or heat generation in the radar device 10. Furthermore, in this embodiment, since the Doppler frequency range in which a Doppler multiplexed signal can be separated is expanded by setting the center frequency of the chirp signal, it is possible to omit the application of a method such as shortening the transmission period T _r . Therefore, this embodiment can prevent a reduction in the detectable distance range of the radar device 10 or a deterioration in distance resolution.

In the present embodiment, the first chirp signal and the second chirp signal have the same modulation parameters other than the center frequency, but the present disclosure is not limited to this. For example, the present disclosure can be applied to chirp signals that have the same distance resolution and the same frequency sweep bandwidth _Bw (q).

For example, as shown in Fig. 10, the first chirp signal and the second chirp signal may be set by modulation parameters that satisfy _Bw (1) = _Bw (2), _Tsw (1) ≠ _Tsw (2), and _Dm (1) ≠ _Dm (2). In this case, although the frequency sweep times _Tsw of the first chirp signal and the second chirp signal are different, the frequency sweep bandwidths _Bw are the same and the range resolutions ΔR (= _C0 / _2Bw ) match, so the radar device 10 can obtain similar effects by performing the operation according to the embodiment of the present disclosure described above.

10 , when the beat signals output from each radio reception unit 203 are discretely sampled in the A/D conversion unit 207 of each signal processing unit 206 by setting T _sw (1)≠T _sw (2), the number of discrete sampling data obtained in a predetermined time range (range gate) T _sw (1)≠T _sw (2) differs. Therefore, the beat frequency analysis unit 208 may perform the following operation instead of performing FFT processing on N _data pieces of discrete sampling data obtained in a predetermined time range (range gate) T _sw for each transmission period _Tr .

For example, the beat frequency analysis unit 208 may perform FFT processing on N _data (1) discrete sampled data obtained in a predetermined time range (range gate) T _sw (1) during the period in which the first chirp signal is transmitted, and may also perform FFT processing on N _data (2) discrete sampled data obtained in a predetermined time range (range gate) T _sw (2) during the period in which the second chirp signal is transmitted.The beat frequency analysis unit 208 may then perform subsequent processing using the smaller of N _data (1) and N _data (2), as the N _data .

(Embodiment 2)
The radar device according to this embodiment may be similar to the radar device 10 shown in FIG.

In the first embodiment, the conditions under which the number of Doppler aliasing can be determined for the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal have been described. For example, in the first embodiment, the operation of determining the number of Doppler aliasing in the Doppler determining unit 213 when the determination conditions are satisfied has been described, and it has been explained that the Doppler frequency range can be expanded to more than twice the Doppler frequency range when one transmitting antenna is used.

Here, for example, in a MIMO radar using unevenly spaced Doppler multiplexing, if there are multiple reflected waves from similar distances, and if the Doppler spacing of these reflected waves matches the Doppler multiplexing spacing (or a multiple of the Doppler multiplexing spacing), the radar device 10 is more likely to erroneously detect the unevenly spaced Doppler frequency range, which could result in incorrect separation of the Doppler multiplexed signal or large angle measurement errors for the multiple reflected waves.

In this embodiment, a radar device that can improve detection performance even in such a situation, in addition to the effects of embodiment 1, will be described. For example, in this embodiment, conditions under which a plurality of reflected waves can be determined regarding the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal will be described. For example, the setting conditions (determinable conditions) for the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal set in the signal generation control unit 104 of the radar transmitter 100, and the operation of the Doppler determination unit 213 are different from those in embodiment 1. Below, the operation of this embodiment that differs from embodiment 1 will be mainly described.

[Judgment possible conditions]
First, the setting conditions (determinable conditions) for the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal set in the signal generation control unit 104 of the radar transmitter 100 will be described.

The Doppler shifter 105 of the radar transmitter 100 may use, for example, the phase rotation φ _n (m) shown in equation (4). In this case, the Doppler multiplexing interval Δf _DDM is expressed by the following equation (26).

A portion of the Doppler multiplexing interval Δf _DDM shown in equation (26) is not used for Doppler multiplexing and no transmit signals are assigned to it, resulting in uneven intervals that are multiples of the Doppler multiplexing interval. Here, δ is an integer greater than or equal to 1. When δ is 1, the Doppler multiplexing interval Δf _DDM is widest, and as δ increases, the Doppler multiplexing interval Δf _DDM narrows. For example, the narrower the Doppler multiplexing interval, the greater the mutual interference between Doppler multiplexed signals, so it is more preferable to set δ to a smaller integer, such as δ=1.

For example, when the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} of reflected waves from two targets (hereinafter referred to as "Target #1" and "Target #2") detected at the same distance index arrive at an interval equal to the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval N _mul × Δf _DDM ), the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} are expressed by the following equations (27) and (28), and the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} have the relationship shown in equation (29). Note that this equation shows the relationship when a first chirp signal with center frequency f _c (1) is used.
f _{d1_T#1} = f _{d_T#1_VFT} (1)+ n _{al_T#1} /T _rs (27)
f _{d1_T#2} = f _{d1_T#2_VFT} (1)+ n _{al_T#2} /T _rs (28)
|f _{d1_T#1} - f _{d1_T#2} |= N _mul ×Δf _DDM (29)

Here, f _{d_T#1_VFT} (1) is the estimated Doppler frequency of Target #1 detected by the first Doppler analysis unit 210 and the first Doppler demultiplexing unit 212, and f _{d1_T#2_VFT} (1) is the estimated Doppler frequency of Target #2 detected by the first Doppler analysis unit 210 and the first Doppler demultiplexing unit 212. Furthermore, n _{al_T#1} represents the number of Doppler aliasing events of Target #1, and n _{al_T#2} represents the number of Doppler aliasing events of Target #2. Furthermore, n _{al_T#1} and n _{al_T#2} take integer values.

Furthermore, _Nmul is a natural number within the assumed Doppler frequency range. For example, _Nmul may be set to _Nmul ∈ {1, ..., (Nt+δ) × _nalmax } using the maximum number of aliasing _nalmax that satisfies Equation (19), or may be set to a value within a range narrower than _Nmul ∈ {1, ..., (Nt+δ) × _nalmax }.

For the relational expressions for two targets (Target #1, Target #2) expressed as above, when the radar device 10 observes the Doppler frequency using the second chirp signal with center frequency f _c (2), the following relational expressions (30), (31), and (32) are obtained. Note that the relational expressions (30), (31), and (32) utilize the fact that the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} when the second chirp signal with center frequency f _c (2) is used can be calculated by multiplying the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} by the center frequency ratio f _c (2)/f _c (1).
f _{d2_T#s1} = f _c (2)/f _c (1)×(f _{d_T#1_VFT} (1)+ n _{al_T#1} /T _rs ) (30)
f _{d2_T#2} = f _c (2)/f _c (1)×(f _{d1_T#2_VFT} (1)+ n _{al_T#2} /T _rs ) (31)
|f _{d2_T#1} - f _{d2_T#2} |= f _c (2)/f _c (1)×(N _mul ×Δf _DDM ) (32)

Here, the center frequency ratio f _{c (2)/f c (1) may be set so that the difference between f c (2)/f c} (1)×(N _mul ×Δf _DDM ) and the Doppler multiplex interval N _mul ×Δf _DDM (for example, the difference between the value of equation (32) and the value of _equation ( ₂₉ )) is greater than the Doppler frequency resolution Δf _d in the second Doppler analysis unit 210.

For example, the radar device 10 may determine the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal so as to satisfy the "second determination condition (1)" of the following equation (33).

As a result, for example, when using the outputs of the second Doppler analysis unit 210 and the second Doppler demultiplexing unit 212 using the second chirp signal with center frequency f _c (2), the Doppler frequencies of Target #1 and Target #2 are observed as Doppler frequencies that deviate from the Doppler frequency estimates of Target #1 and Target #2 detected in the first Doppler analysis unit 210 and the first Doppler demultiplexing unit 212 (for example, an interval of the Doppler multiplexing interval Δf _DDM (or a multiple of Δf _DDM )) by at least more than the Doppler frequency resolution Δf _d in the second Doppler analysis unit 210.

Therefore, the radar device 10 can suppress overlap of Doppler multiplexed signal components between two reflected waves observed at the same distance, for example, and improve the separation performance of the Doppler multiplexed signals of each of the two reflected waves. For example, the second determination condition (1) is a condition under which, even if the Doppler frequency estimates of Target #1 and Target #2 detected by the first Doppler analysis unit 210 and the first Doppler demultiplexing unit 212 are spaced apart by the Doppler multiplexing interval Δf _DDM (or a multiple of Δf _DDM ), and Doppler demultiplexing of Target #1 and Target #2 is difficult, Doppler demultiplexing of Target #1 and Target #2 can be performed based on the outputs of the second Doppler analysis unit 210 and the second Doppler demultiplexing unit 212 using the second chirp signal, and the Doppler frequency estimates of Target #1 and Target #2 can be obtained.

Here, when f _c (2)>f _c (1), the condition for f _c (1) and f _c (2) in the second determinable condition (1) is expressed by the following equation (34).

For example, if N _mul =1 is satisfied, the case of N _mul >1 is also satisfied, and therefore the second determinable condition (1) may be expressed as in the following equation (35).

Similarly, when f _c (2)<f _c (1), the condition for f _c (1) and f _c (2) in the second determinable condition (1) is expressed by the following equation (36).

For example, if N _mul =1 is satisfied, the case of N _mul >1 is also satisfied, and therefore the second determinable condition (1) may be expressed as in the following equation (37).

Similarly, when the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} of reflected waves from two targets (e.g., Target #1 and Target #2) detected at the same distance index arrive at an interval equal to the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval N _mul × Δf _DDM ), the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} are expressed by the following equations (38) and (39), and the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} have the relationship shown in equation (40). Note that this equation shows the relationship when a second chirp signal with center frequency f _c (2) is used.
f _{d2_T#1} = f _{d_T#1_VFT} (2)+ n _{al_T#1} /T _rs (38)
f _{d2_T#2} = f _{d1_T#2_VFT} (2)+ n _{al_T#2} /T _rs (39)
|f _{d2_T#1} - f _{d2_T#2} | = N _mul ×Δf _DDM (40)

Here, f _{d_T#1_VFT} (2) is the estimated Doppler frequency of Target #1 detected by the second Doppler analysis unit 210 and the second Doppler demultiplexing unit 212, and f _{d1_T#2_VFT} (2) is the estimated Doppler frequency of Target #2 detected by the second Doppler analysis unit 210 and the second Doppler demultiplexing unit 212. Furthermore, n _{al_T#1} represents the number of Doppler aliasing events of Target #1, and n _{al_T#2} represents the number of Doppler aliasing events of Target #2. Furthermore, n _{al_T#1} and n _{al_T#2} take integer values.

For the relational expressions for two targets (Target #1, Target #2) expressed as above, when the radar device 10 observes the Doppler frequency using a first chirp signal with a center frequency of f _c (1), the following relational expressions (41), (42), and (43) are obtained. Note that the relational expressions (41), (42), and (43) utilize the fact that the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} when the first chirp signal with a center frequency of f _c (1) is used can be calculated by multiplying the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} by the center frequency ratio f _c (1)/f _c (2).
f _{d1_T#1} = fc(1)/fc(2)×(f _{d_T#1_VFT} (2)+ n _{al_T#1} /T _rs ) (41)
f _{d1_T#2} = fc(1)/fc(2)×( f _{d1_T#2_VFT} (2)+ n _{al_T#2} /T _rs ) (42)
|f _{d1_T#1} - f _{d1_T#2} |= fc(1)/fc(2)×(N _mul ×Δf _DDM ) (43)

Here, the center frequency ratio f _c (1)/f _c (2) may be set so that the difference between f c (1)/f c (2)×(N _mul ×Δf _DDM ) and the Doppler multiplex interval N _mul ×Δf _DDM (for example, the difference between the value of equation (43) and the _value of _equation (40)) is greater than the Doppler frequency resolution Δf _d in the first Doppler analysis unit 210.

For example, the radar device 10 may determine the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal so as to satisfy the "second determination condition (2)" of the following equation (44).

As a result, for example, when the outputs of the first Doppler analysis unit 210 and the first Doppler demultiplexing unit 212 using the first chirp signal with center frequency f _c (1) are used, the Doppler frequencies of Target #1 and Target #2 are observed as Doppler frequencies that deviate from the Doppler frequency estimates of Target #1 and Target #2 (for example, the Doppler multiplexing interval Δf _DDM (or an interval that is a multiple of Δf _DDM )) detected in the second Doppler analysis unit 210 and the second Doppler demultiplexing unit 212 by at least the Doppler frequency resolution Δf _d in the first Doppler analysis unit 210.

Therefore, the radar device 10 can suppress overlap of Doppler multiplexed signal components between two reflected waves observed at the same distance, for example, and improve the separation performance of the Doppler multiplexed signals of each of the two reflected waves. For example, the second determination condition (2) is a condition under which, even if the Doppler frequency estimates of Target #1 and Target #2 detected by the second Doppler analysis unit 210 and the second Doppler demultiplexing unit 212 are the Doppler multiplexing interval Δf _DDM (or an interval that is a multiple of Δf _DDM ) and Doppler demultiplexing of Target #1 and Target #2 is difficult, Doppler demultiplexing of Target #1 and Target #2 is possible based on the outputs of the first Doppler analysis unit 210 and the first Doppler demultiplexing unit 212 using the first chirp signal, and the Doppler frequency estimates of Target #1 and Target #2 can be obtained.

Here, when f _c (2)>f _c (1), the condition for f _c (1) and f _c (2) in the second determinable condition (2) is expressed by the following equation (45).

For example, when N _mul =1 is satisfied (meaning that N _mul >1 also holds), the second determinable condition (2) may be expressed as the following equation (46).

Furthermore, if the second determinable condition (2) is satisfied as in the following equation (47), the second determinable condition (1) is also satisfied.

Therefore, for example, the following equation (48) is called the "second determinable condition" when f _c (2)>f _c (1).

Similarly, when f _c (2)<f _c (1), the condition for f _c (1) and f _c (2) in the second determinable condition (2) is expressed by the following equation (49).

For example, when N _mul =1 is satisfied (meaning that N _mul >1 also holds), the second determinable condition (2) may be expressed as the following equation (50).

Furthermore, for example, from equations (37) and (50), if the second determinable condition (2) is satisfied as in the following equation (51), the second determinable condition (1) is also satisfied.

Therefore, for example, the following equation (52) is called the "second determinable condition" when f _c (2)<f _c (1).

Furthermore, for example, the second determinable condition shown in equation (33) or equation (44) may be such that the center frequencies f _c (1) and f _c (2) satisfy a condition that they are greater than an integer multiple (e.g., α times) of the Doppler frequency resolution Δf _d (where α≧1). In this case, the larger α is, the easier it is for the radar device 10 to perform Doppler separation even when there is noise influence, such as when the received signal level is low. Furthermore, for example, the second determinable condition shown in equation (48) when f _c (2)>f _c (1) and the second determinable condition shown in equation (52) when f _c (2)<f _c (1) may be expressed as the following equations (53) and (54), respectively.

As an example, when f _c (1)=78 GHz and f _c (2)>f _c (1), f _c (2) is set to be greater than 80.51 GHz when α=1, Nc=128, Nt=3, and δ=1, and f _c (2) is set to be 83.2 GHz when α=2, Nc=128. Furthermore, for example, f _c (2) is set to be 79.23 GHz when α=1, Nc=256, Nt=3, and δ=1, and f _c (2) is set to be greater than 80.51 GHz when α=2, Nc=256.

The second determinable condition is a condition in which the difference |f _c (2)−f _c (1)| between the center frequencies of the chirp signals is wider than that in the first determinable condition. For example, if the second determinable condition is satisfied, the first determinable condition is also satisfied.

Furthermore, f _c (1) and f _c (2) may be set within the pass frequency range of the radar transmitter 100 or the radar receiver 200, and the expansion of the Doppler frequency detection range may also be restricted by the pass frequency characteristics of the radar transmitter 100 or the radar receiver 200.

Furthermore, the second determination condition may also be set taking into consideration the frequency condition constraints shown in equation (18) or equation (19). This allows the same effects as in embodiment 1 to be obtained.

Furthermore, the second determination condition may include a frequency condition that the difference in Doppler frequency observed between f _c (1) and f _c (2) is less than the Doppler multiplex interval Δf _DDM . For example, f _c (1) and f _c (2) may be set so as to satisfy the conditions shown in the following equations (55) and (56).

Here, _Nmul is a natural number within the range of the assumed Doppler frequency. For example, _Nmul may be set to _Nmul ∈ {1, ..., (Nt+δ) × _nalmax } using the maximum number of aliasing _nalmax that satisfies Equation (19), or may be set to a value within a narrower range than _Nmul ∈ {1, ..., (Nt+δ) × _nalmax }.

The above explains the conditions for determination.

[Doppler determination method]
Next, the Doppler determination unit 213 may perform the following operation using the center frequencies f _c (1) and f _c (2) that satisfy the second determination possible condition described above.

For example, when the distance index f _{b_cfar} includes the same number of pieces of separation index information for the Doppler multiplexed signals output from the first Doppler multiplex separation unit 212 and the same number of pieces of separation index information for the Doppler multiplexed signals output from the second Doppler multiplex separation unit 212, the Doppler determination unit 213 may perform the same operation as in embodiment 1. This provides the same effect as in embodiment 1.

On the other hand, the Doppler determination unit 213 may perform the following process, for example, when the distance index f _{b_cfar} does not include the same number of pieces of separation index information for the Doppler multiplexed signals output from the first Doppler multiplex separation unit 212 and the second Doppler multiplex separation unit 212. For example, the Doppler determination unit 213 may perform the following process when, for example, there are multiple reflected waves from approximately the same distance to the radar device 10, and there is a possibility that multiple reflected waves will be separated in one of the Doppler analysis of the first chirp signal and the Doppler analysis of the second chirp signal, but will not be separated in the other.

<Case 1>
Case 1 will be described as a case where, for example, the distance index f _{b_cfar} includes multiple pieces of separation index information for Doppler multiplexed signals output from the first Doppler multiplexed separation unit 212, and does not include separation index information for Doppler multiplexed signals output from the second Doppler multiplexed separation unit 212. For example, case 1 is a case where a Doppler multiplexed signal is separated in the first Doppler multiplexed separation unit 212, but is not separated in the second Doppler multiplexed separation unit 212, such as the following case.

When two reflected waves are present at approximately the same distance from the radar device 10, it is expected that the separation index information for the Doppler multiplexed signal output from the first Doppler multiplexed separation unit 212 and the separation index information for the Doppler multiplexed signal output from the second Doppler multiplexed separation unit 212 will each contain the separation index information for the Doppler multiplexed signal for the two reflected waves.

However, by transmitting using the second chirp signal, for example, as shown in (c) of Figure 1, if the reflected waves #1 and #2 match the Doppler multiplexing interval Δf _DDM or a multiple of the Doppler multiplexing interval, some Doppler multiplexed signals may be received as overlapping Doppler frequency components between the reflected waves #1 and #2. This makes it difficult to separate the Doppler multiplexed signals, and the Doppler multiplexed signals output from the second Doppler multiplexing separation unit 212 may not contain separation index information.

On the other hand, even if such a reflected wave is present, the radar device 10 satisfies the second determination condition by transmitting using the first chirp signal, and therefore some Doppler multiplexed signals will not be received as overlapping Doppler frequency components in reflected wave #1 and reflected wave #2. This means that the Doppler multiplexed signal output from the first Doppler multiplex separation unit 212 will contain separation index information. In such a case, the following operation is performed.

The following describes, as an example, a case in which the separation index information of the Doppler multiplexed signal output from the first Doppler multiplexing separation unit 212 includes separation index information of the Doppler multiplexed signals of reflected waves from two targets (e.g., Target #1 and Target #2).

For example, when using a chirp signal with a center frequency of f _c (2), the Doppler determination unit 213 may determine (or may assume) that the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} of the reflected waves from two targets (Target #1 and Target #2) arrive at a Doppler frequency interval equal to the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval N _mul ×Δf _DDM ). In this case, the Doppler determination unit 213 may perform the following Doppler determination process when the arriving waves overlap.

For example, the Doppler determination unit 213 may calculate the number of Doppler aliasing n _{al_ #T1} of Target #1 and the number of Doppler aliasing n _{al_ #T2} of Target #2 that minimize the following equation (57).

Here, the Doppler aliasing numbers n _{al_#T1} and n _{al_#T2} are integer values and may be calculated within a range of integer values that cover the Doppler frequency range of the expected target. For example, n _{al_#T1} and n _{al_#T2} may be calculated within the range of ±n _{almax using the maximum aliasing number n almax that} satisfies equation (19). Furthermore, when the Doppler aliasing numbers n _{al_#T1} and n _{al_#T2} are varied, the Doppler aliasing numbers that minimize equation (57) are denoted as n _{alist_#T1} and n _{alist_#T2} , respectively. Furthermore, mod[x,y] is a function that represents the remainder when x is divided by y.

For example, when a chirp signal with a center frequency f _c (2) is used, equation (57) is minimized when the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} of the reflected waves from two targets (Target #1 and Target #2) satisfy the condition that they arrive at a Doppler frequency interval equal to the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval N _mul ×Δf _DDM ). The Doppler determination unit 213 utilizes this fact to estimate the Doppler aliasing numbers n _{alest_#T1} and n _{alest_#T2} of Target #1 and Target #2.

In equation (57), for example, the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} for Doppler aliasing numbers n _{al_#T1} and n _{al_#T2} are estimated using f _est (f _{d_T#1_VFT(1)} , n _{al_T#1} ) and f _est (f _{d_T#2_VFT(1)} , n _{al_T#2} ), respectively. Here, f _est (f _{d_VFT(1)} , n _al ) may be the function shown in equation (17). Furthermore, each of the Doppler frequency estimates _{fd_T#1_VFT(1)} and _{fd_T#2_VFT(1)} is a Doppler frequency estimate based on, for example, the separation index information of the Doppler multiplexed signal in the distance index _{fb_cfar} output from the first Doppler multiplex separation unit 212, assuming that the Doppler frequency of the target is within the Doppler frequency range -1/( _2Trs ) ≦ _{fd_TargetDoppler} < 1/( _2Trs ).

<Case 2>
Case 2 will be described as a case where, for example, the distance index f _{b_cfar} includes multiple pieces of separation index information for Doppler multiplexed signals output from the second Doppler multiplexed separation unit 212, and does not include separation index information for Doppler multiplexed signals output from the first Doppler multiplexed separation unit 212. For example, case 2 is a case where a Doppler multiplexed signal is not separated in the first Doppler multiplexed separation unit 212, but is separated in the second Doppler multiplexed separation unit 212, such as the following case.

When two reflected waves are present at approximately the same distance from the radar device 10, it is expected that the separation index information for the Doppler multiplexed signal output from the first Doppler multiplexed separation unit 212 and the separation index information for the Doppler multiplexed signal output from the second Doppler multiplexed separation unit 212 will each contain the separation index information for the Doppler multiplexed signal for the two reflected waves.

However, by transmitting using the first chirp signal, for example, as shown in FIG. 1(c), if the reflected waves #1 and #2 match the Doppler multiplexing interval Δf _DDM or a multiple of the Doppler multiplexing interval, some of the Doppler multiplexed signals between the reflected waves #1 and #2 may be received as overlapping Doppler frequency components.

This makes it difficult to separate the Doppler multiplexed signals, and the Doppler multiplexed signals output from the first Doppler multiplexing separation unit 212 may not contain separation index information.

On the other hand, even if such a reflected wave is present, the radar device 10 satisfies the second determination condition by transmitting using the second chirp signal, and therefore some Doppler multiplexed signals will not be received as overlapping Doppler frequency components in reflected wave #1 and reflected wave #2. This means that the Doppler multiplexed signal output from the second Doppler multiplex separation unit 212 will contain separation index information. In such a case, the following operation is performed.

The following describes, as an example, a case in which the separation index information of the Doppler multiplexed signal output from the second Doppler multiplexing separation unit 212 includes separation index information of the Doppler multiplexed signals of reflected waves from two targets (e.g., Target #1 and Target #2).

For example, when using a chirp signal with a center frequency of f _c (1), the Doppler determination unit 213 may determine (or may assume) that the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} of the reflected waves from two targets (Target #1 and Target #2) arrive at a Doppler frequency interval equal to the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval N _mul ×Δf _DDM ). In this case, the Doppler determination unit 213 may perform the following Doppler determination process when the arriving waves overlap.

For example, the Doppler determination unit 213 may calculate the Doppler aliasing numbers n _{al_ #T1} and n _{al_ #T2} of Target #1 and Target #2 that minimize the following equation (58).

Here, the Doppler aliasing numbers n _{al_#T1} and n _{al_#T2} are integer values and may be calculated within a range of integer values that cover the Doppler frequency range of the expected target. For example, n _{al_#T1} and n _{al_#T2} may be calculated within a range of ±n _{almax using the maximum aliasing number n almax that} satisfies equation (19). Furthermore, when the Doppler aliasing numbers n _{al_#T1} and n _{al_#T2} are varied, the Doppler aliasing numbers that minimize equation (58) are denoted as n _{alist_#T1} and n _{alist_#T2} , respectively.

For example, when a chirp signal with a center frequency f _c (1) is used, equation (58) is minimized when the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} of reflected waves from two targets (Target #1 and Target #2) satisfy the condition that they arrive at a Doppler frequency interval equal to the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval N _mul ×Δf _DDM ). The Doppler determination unit 213 utilizes this fact to estimate the Doppler aliasing numbers n _{alest_#T1} and n _{alest_#T2} of Target #1 and Target #2.

In equation (58), for example, the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} for Doppler aliasing numbers n _{al_#T1} and n _{al_#T2} are estimated using f _est2 (f _{d_T#1_VFT(2)} , n _{al_T#1} ) and f _est2 (f _{d_T#2_VFT(2)} , n _{al_T#2} ), respectively. Furthermore, each of the Doppler frequency estimates _{fd_T#1_VFT(2)} and _{fd_T#2_VFT(2)} is a Doppler frequency estimate based on the separation index information of the Doppler multiplexed signal in the distance index _{fb_cfar} output from the second Doppler multiplex separation unit 212, assuming that the Doppler frequency of the target is within the Doppler frequency range -1/( _2Trs ) ≦ _{fd_TargetDoppler} < 1/( _2Trs ).

Furthermore, f _est2 (f _{d_VFT(2)} , n _al ) is a function shown in the following equation (59), which calculates the Doppler frequency (f d_T# _{1_VFT(2)} +n _{al /} T rs) assuming that the Doppler frequency estimate value f d_VFT(2) calculated based on the separation index information of the Doppler multiplexed signal output from the second Doppler multiplex _{separation unit 212 has a Doppler aliasing count of n al, and outputs a value multiplied by f c ( 1 ) /} f _c (2) to convert it to a Doppler frequency observed using a first chirp signal with a center frequency of f c (1).

By the operation of the Doppler determination unit 213 as described above, even if the distance index f _{b_cfar} does not include the same number of separation index information for the Doppler multiplexed signals output from the first Doppler multiplex separation unit 212 and the second Doppler multiplex separation unit 212, the Doppler determination unit 213 can estimate the number of Doppler aliasing occurrences based on the separation index information of the Doppler multiplexed signals obtained in one of the Doppler multiplex separation units 212, as in case 1 or case 2, and assume that the Doppler frequency of the reflected wave when using a chirp signal of the other center frequency arrives at a Doppler frequency interval that is the Doppler multiplexing interval Δf _DDM (or a multiple of the Doppler multiplexing interval N _mul × Δf _DDM ).

For example, the Doppler determination unit 213 estimates the Doppler peaks of each of the multiple targets in the Doppler analysis corresponding to the first chirp signal or the Doppler analysis corresponding to the second chirp signal, based on the results of one Doppler analysis in which multiple reflected wave signals are separated, and determines the number of times the Doppler frequency of each of the multiple targets folds based on the estimated intervals between the Doppler peaks of the multiple targets and the Doppler multiplex interval.

In this embodiment, the subsequent processing of the radar device 10 may be the same as in the first embodiment. Furthermore, the radar device 10 may output a Doppler frequency estimate as a positioning output, for example, using the estimated Doppler aliasing numbers n _{alast_#T1} and n _{alast_#T2} . As described above, in this embodiment, the radar device 10 can obtain the effects of the first embodiment by using the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal that satisfy the second determination condition.

Furthermore, in the radar device 10, for example, the Doppler frequencies of reflected waves from two targets may arrive at a Doppler frequency interval that is the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval, N _mul × Δf _DDM ), making it difficult to perform Doppler demultiplexing of those targets using one of the chirp signals. Even in this case, the radar device 10 can perform Doppler demultiplexing of the two targets by making the Doppler frequency of the other chirp signal not coincident with the Doppler frequency interval that is the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval, N _mul × Δf _DDM ), but differing by at least more than the Doppler resolution of the Doppler analysis unit 210.

As a result, in this embodiment, the radar device 10 can improve its detection performance for multiple waves arriving from the same distance, improving the probability of target detection in the radar device 10 and reducing the probability of non-detection, thereby improving radar detection performance.

(Modification of the second embodiment)
As a modification of the second embodiment, when the Doppler frequency _{fd_TargetDoppler} of an assumed target is within the Doppler frequency range of -1/( _2Trs ) ≦ _{fd_TargetDoppler} < 1/( _2Trs ), the radar device 10 may use the center frequency _fc (1) of the first chirp signal and the center frequency _fc (2) of the second chirp signal that satisfy the second determination condition, and perform processing without estimating the number of aliasings in the Doppler determination unit 213. For example, the radar device 10 may perform processing to separate and detect multiple reflected waves without estimating the number of aliasings.

In this case, the Doppler determination unit 213 may, for example, perform the following processing of Case 1a and Case 2a instead of the processing of Case 1 and Case 2 described above.

<Case 1a>
In case 1a, for example, the distance index f _{b_cfar} includes multiple pieces of separation index information for the Doppler multiplexed signal output from the first Doppler multiplexed separation unit 212, and does not include separation index information for the Doppler multiplexed signal output from the second Doppler multiplexed separation unit 212, such as the following case.

When two reflected waves are present at approximately the same distance from the radar device 10, it is expected that the separation index information for the Doppler multiplexed signal output from the first Doppler multiplexed separation unit 212 and the separation index information for the Doppler multiplexed signal output from the second Doppler multiplexed separation unit 212 will each contain the separation index information for the Doppler multiplexed signal for the two reflected waves.

However, by transmitting using the second chirp signal, for example, as shown in Figure 1(c), if the reflected waves #1 and #2 match the Doppler multiplexing interval Δf _DDM or a multiple of the Doppler multiplexing interval, some Doppler multiplexed signals may be received as overlapping Doppler frequency components between the reflected waves #1 and #2.

This makes it difficult to separate the Doppler multiplexed signals, and the Doppler multiplexed signals output from the second Doppler multiplexing separation unit 212 will not contain separation index information. However, even if such reflected waves exist, the radar device 10 satisfies the second determination condition by transmitting using the first chirp signal, so that some Doppler multiplexed signals will not be received as overlapping Doppler frequency components in reflected wave #1 and reflected wave #2.

As a result, the Doppler multiplexed signal output from the first Doppler multiplexing separation unit 212 will contain separation index information (assuming a case in which reflected waves #1 and #2 arrive, similar to case 1). In such a case, the following describes the operations that differ from case 1.

The following describes, as an example, a case in which the separation index information of the Doppler multiplexed signal output from the first Doppler multiplexing separation unit 212 includes separation index information of the Doppler multiplexed signals of reflected waves from two targets (e.g., Target #1 and Target #2).

For example, when a chirp signal with a center frequency f _c (2) is used, the Doppler determination unit 213 may determine (or may assume) that the Doppler frequencies f _{d2_T#1} and f _{d2_T#2} of the reflected waves from two targets (Target #1 and Target #2) arrive at a Doppler frequency interval that is the Doppler multiplexing interval Δf _DDM (or a multiple of the Doppler multiplexing interval N _mul ×Δf _DDM ). In this case, the Doppler determination unit 213 may output separation index information of the Doppler multiplexed signal output from the first Doppler multiplex separation unit 212 to the direction estimation unit 214.

The direction estimation unit 214 performs direction estimation processing using the distance index f _{b_cfar} and separation index information of the Doppler multiplexed signal (q=1) that are Doppler demultiplexed and output from the first Doppler demultiplexing unit 212. The direction estimation unit 214 may perform direction estimation processing using q=1 in equation (21), for example.

<Case 2a>
In case 2a, for example, the distance index f _{b_cfar} includes multiple pieces of separation index information for the Doppler multiplexed signal output from the second Doppler multiplexed separation unit 212, and does not include separation index information for the Doppler multiplexed signal output from the first Doppler multiplexed separation unit 212, such as the following case.

When two reflected waves are present at approximately the same distance from the radar device 10, it is expected that the separation index information for the Doppler multiplexed signal output from the first Doppler multiplexed separation unit 212 and the separation index information for the Doppler multiplexed signal output from the second Doppler multiplexed separation unit 212 will each contain the separation index information for the Doppler multiplexed signal for the two reflected waves.

However, when the first chirp signal is used for transmission and the reflected waves #1 and #2 coincide with the Doppler multiplexing interval Δf _DDM or a multiple of the Doppler multiplexing interval, as shown in FIG. 1(c), for example, some of the Doppler multiplexed signals may be received as overlapping Doppler frequency components between the reflected waves #1 and #2.

This makes it difficult to separate the Doppler multiplexed signals, and the Doppler multiplexed signals output from the first Doppler multiplexing separation unit 212 may not contain separation index information.

On the other hand, even if such a reflected wave is present, the radar device 10 satisfies the second determination condition by transmitting using the second chirp signal, and therefore some Doppler multiplexed signals will not be received as overlapping Doppler frequency components in reflected wave #1 and reflected wave #2. This results in a case where the Doppler multiplexed signal output from the second Doppler multiplex separation unit 212 contains separation index information (assuming a case where reflected waves #1 and #2 arrive similar to case 2). In such a case, the following describes the operation that differs from case 2.

The following describes, as an example, a case in which the separation index information of the Doppler multiplexed signal output from the second Doppler multiplexing separation unit 212 includes separation index information of the Doppler multiplexed signals of reflected waves from two targets (e.g., Target #1 and Target #2).

For example, when a chirp signal with a center frequency f _c (1) is used, the Doppler determination unit 213 may determine (or may assume) that the Doppler frequencies f _{d1_T#1} and f _{d1_T#2} of the reflected waves from two targets (Target #1 and Target #2) arrive at a Doppler frequency interval that is the Doppler multiplexing interval Δf _DDM (or a multiple of the Doppler multiplexing interval N _mul ×Δf _DDM ). In this case, the Doppler determination unit 213 may output separation index information of the Doppler multiplexed signal output from the second Doppler multiplex separation unit 212 to the direction estimation unit 214.

The direction estimation unit 214 performs direction estimation processing using the distance index f _{b_cfar} and separation index information of the Doppler multiplexed signal (q=2) that are Doppler demultiplexed and output from the second Doppler demultiplexing unit 212. The direction estimation unit 214 may perform direction estimation processing using, for example, q=2 in equation (21).

Case 1a and Case 2a have been explained above.

In this way, by using the center frequency f _c (1) of the first chirp signal and the center frequency f _c (2) of the second chirp signal that satisfy the second determination condition in the Doppler frequency range where the _{target Doppler} frequency f d_TargetDoppler is -1 _/ (2T _rs ) ≦ f d_TargetDoppler < 1/(2T rs ), the following effects can be obtained.

For example, there may be cases where the Doppler frequencies of reflected waves from two targets arrive at a Doppler frequency interval equal to the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval, N _mul × Δf _DDM ), making it difficult to perform Doppler demultiplexing of the targets based on one of the chirp signals. Even in this case, by using the other chirp signal (a chirp signal with a different center frequency), it is possible to make the Doppler frequency interval of the reflected waves from the two targets different from the Doppler frequency interval equal to the Doppler multiplex interval Δf _DDM (or a multiple of the Doppler multiplex interval, N _mul × Δf _DDM ), and to make the interval at least greater than the Doppler resolution of the Doppler analysis unit 210. Thus, the radar device 10 enables Doppler demultiplexing of two targets.

In this way, in a variation of embodiment 2, the radar device 10 may perform direction estimation processing based on the results of either the Doppler analysis of the first chirp signal or the Doppler analysis of the second chirp signal, whichever results in separating multiple reflected wave signals.

As a result, even if, for example, in a Doppler analysis of a reflected wave signal corresponding to one chirp signal, the spacing between the Doppler frequencies (Doppler peaks) corresponding to multiple reflected waves at similar distances from the radar device 10 matches the Doppler multiplexing interval (or a multiple of the Doppler multiplexing interval), the radar device 10 can separate and detect the Doppler multiplexing signals corresponding to each of the multiple reflected waves based on the results of the Doppler analysis of the reflected wave signal corresponding to the other chirp signal.

As a result, in the modified version of embodiment 2, the radar device 10 can improve its detection performance for multiple reflected waves arriving from the same distance, thereby improving the probability of target detection by the radar device 10 and reducing the probability of non-detection. Therefore, according to the modified version of embodiment 2, the radar detection performance of the radar device 10 can be improved.

(Embodiment 3)
In the first embodiment, an example of the configuration of a radar device including one radar transmission signal generator has been described. However, the configuration of the radar device is not limited to this, and a radar device including a plurality of radar transmission signal generators may also be used.

For example, Fig. 11 shows a configuration example in which the radar transmitter 100a of the radar device 10a includes two radar transmission signal generators 101. In the first embodiment, the radar device 10 includes a single radar transmission signal generator 101, and the radar transmission waves (e.g., chirp signals) with different center frequencies are alternately transmitted by switching over time every transmission period T _r . In contrast, in the present embodiment, as shown in Fig. 11, the radar device 10a includes multiple radar transmission signal generators 101, and simultaneously transmits radar transmission waves (e.g., chirp signals) with different center frequencies from multiple transmitting antennas 106 every transmission period T _r . With this configuration, as in the first embodiment, the effect of expanding the detectable Doppler frequency range can be obtained.

The following describes the operation of this embodiment, focusing mainly on examples of operation that differ from embodiment 1.

[Configuration example of radar transmitter 100a]
11 shows, as an example, a configuration in which the radar transmitter 100a of the radar device 10a includes two radar transmission signal generators 101. Hereinafter, the two radar transmission signal generators 101 will be referred to as the "first radar transmission signal generator 101 (or radar transmission signal generator 101-1)" and the "second radar transmission signal generator 101-2 (or radar transmission signal generator 101-2)."

In FIG. 11, the configuration of each radar transmission signal generation unit 101 may be the same as in embodiment 1. Each radar transmission signal generation unit 101 generates a radar transmission signal, for example, under control from the signal generation control unit 104.

The signal generation control unit 104 controls the generation of radar transmission signals by the first and second radar transmission signal generation units 101 (e.g., the modulation signal generation unit 102 and the VCO 103). For example, the signal generation control unit 104 may set parameters (e.g., modulation parameters) related to the chirp signals so that the first and second radar transmission signal generation units 101 each transmit chirp signals with different center frequencies. Hereinafter, the chirp signal generated by the first radar transmission signal generation unit 101 will be referred to as the "first chirp signal," and the chirp signal generated by the second radar transmission signal generation unit 101 will be referred to as the "second chirp signal."

As in the first embodiment, the modulation parameters for the chirp signal may include, for example, the center frequency f _c (q), frequency sweep bandwidth B _w (q), sweep start frequency f _cstart (q), sweep end frequency f _cend (q), frequency sweep time T _sw (q), and frequency sweep change rate D _m (q). Note that D _m (q) = B _w (q) / T _sw (q). Also, B _w (q) = f _cend (q) - f _cstart (q) and f _c (q) = (f _cstart (q) + f _cend (q)) / 2. Also, for example, q = 1 or 2, where q = 1 represents the modulation parameters of the first chirp signal and q = 2 represents the modulation parameters of the second chirp signal.

As in the first embodiment, the signal generation control unit 104 may set (or select) a center frequency f _c (q) that satisfies a predetermined condition. In the following, as an example, a case will be described in which, among the modulation parameters set for the first chirp signal and the second chirp signal, the center frequencies f _c (q) are different from each other, but the other modulation parameters are the same (or common). However, this is not limiting, and application of an embodiment of the present disclosure is not limited to this. For example, it is sufficient that the resolution of the distance axis is the same for the first chirp signal and the second chirp signal, and therefore it is sufficient to set chirp signals with the same frequency sweep bandwidth B _w (q).

Furthermore, the signal generation control unit 104 may control the modulation signal generating unit 102 and the VCO 103 so that two chirp signals with different center frequencies f _c (q) are simultaneously transmitted (or output) Nc times each.

Figure 12 shows an example of chirp signals (e.g., first chirp signal and second chirp signal) output by the first and second radar transmission signal generation units 101 based on the control of the signal generation control unit 104. Note that Figure 12 shows an example of an up-chirp waveform in which the modulation frequency gradually increases over time, but this is not limited to this, and a down-chirp waveform in which the modulation frequency gradually decreases over time may also be applied. Similar effects can be obtained regardless of whether the modulation frequency is an up-chirp or down-chirp.

12, the first chirp signal and the second chirp signal are simultaneously transmitted with a transmission period T _r . In the following, unless otherwise specified, the frequency sweep bandwidth, frequency sweep time (also called range gate), and frequency sweep change rate represent parameters with the same values for the first chirp signal and the second chirp signal, and may be expressed as B _w (1) = B _w (2) = B _w , T _sw (1) = T _sw (2) = T _sw , and D _m (1) = D _m (2) = D _m , respectively.

Furthermore, the frequency sweep bandwidths of each chirp signal with a different center frequency may not include overlapping bands, as shown in Figure 12(a), or may include overlapping bands, as shown in Figure 12(b). In one embodiment of the present disclosure, similar effects can be achieved regardless of whether the frequency sweep bandwidths include overlapping bands, as long as the relationship in center frequency between the first chirp signal and the second chirp signal satisfies a predetermined condition.

In one embodiment of the present disclosure, the transmission period T _r may be set to, for example, several hundred μs or less, and the transmission time interval of the radar transmission signal may be set to a relatively short time. As a result, even if the center frequencies of the first chirp signal and the second chirp signal are different, the frequency of the beat signal of the received reflected wave (e.g., the beat frequency index) does not change, and the radar device 10 a can detect this as a change in Doppler frequency.

The first chirp signal output from the first radar transmission signal generation unit 101 (e.g., VCO 103) is input to, for example, N1 of the Nt Doppler shift units 105 (e.g., represented as Doppler shift units 105-1 to 105-N1). Furthermore, the first chirp signal output from the first radar transmission signal generation unit 101 is input to, for example, N3 of the Na mixer units 204 of the radar reception unit 200a (e.g., mixer units 204 of antenna system processing units 201-1 to 201-N3).

On the other hand, the second chirp signal output from the second radar transmission signal generation unit 101 (e.g., VCO 103) is input to, for example, N2 Doppler shift units 105 (e.g., Doppler shift units 105-N1+1 to 105-Nt) out of the Nt Doppler shift units 105. Furthermore, the second chirp signal output from the second radar transmission signal generation unit 101 is input to, for example, N4 mixer units 204 (e.g., antenna system processing units 201-N3+1 to 201-Na) out of the Na mixer units 204 of the radar reception unit 200a.

Here, N1 + N2 = Nt, and N3 + N4 = Na. Note that N1 and N2 can each be 2 or greater, and Nt can be 4 or greater. Also, N3 and N4 can each be 1 or greater, and Na can be 2 or greater.

The output signals of the N1 Doppler shifters 105 to which the first chirp signal is input are amplified to a predetermined transmission power and radiated into space from each transmitting antenna 106 (e.g., Tx#1 to Tx#N1). The output signals of the N2 Doppler shifters 105 to which the second chirp signal is input are amplified to a predetermined transmission power and radiated into space from each transmitting antenna 106 (e.g., Tx#N1+1 to Tx#Nt). As a result, the first chirp signal and the second chirp signal are simultaneously transmitted every transmission period T _r .

Hereinafter, the N1 Doppler shift units 105 (e.g., Doppler shift units 105-1 to 105-N1) to which the first chirp signal is input and the respective transmit antennas 106 (e.g., Tx#1 to Tx#N1) that transmit the output signals of these N1 Doppler shift units 105 are referred to as the "first transmit sub-block." Furthermore, the N2 Doppler shift units 105 (e.g., Doppler shift units 105-N1+1 to 105-Nt) to which the second chirp signal is input and the respective transmit antennas 106 (e.g., Tx#N1+1 to Tx#Nt) that transmit the output signals of these N2 Doppler shift units 105 are referred to as the "second transmit sub-block."

For example, the Doppler shift unit 105 included in the first transmission sub-block applies a phase rotation φ _nsub1 to the first chirp signal to impart a Doppler shift amount DOP _nsub1 to the first chirp signal for each transmission period T _r of the chirp signal, and outputs the Doppler-shifted signal to a transmission antenna 106 (e.g., Tx#1 to Tx#N1). Here, nsub1 is an integer between 1 and N1. The Doppler shift unit 105 included in the second transmission sub-block applies a phase rotation φ _nsub2 to the second chirp signal to impart a Doppler shift amount DOP _nsub2 to the second chirp signal for each transmission period T _r of the chirp signal, and outputs the Doppler-shifted signal to a transmission antenna 106 (e.g., Tx#N1+1 to Tx#Nt). Here, nsub2 is an integer between 1 and N2.

An example of a method for imparting the Doppler shift amount DOP _nsub1 (or phase rotation φ _nsub1 ) and the Doppler shift amount DOP _nsub2 (or phase rotation φ _nsub2 ) in the Doppler shift unit 105 included in the first and second transmission subblocks will be described later.

Also, if Nt is an even number, the number of transmit antennas 106 included in the first and second transmission subblocks may be made the same, for example, by setting N1 = N2. Also, if Nt is an odd number, the number of transmit antennas 106 included in the first and second transmission subblocks may be made approximately the same, with a difference of one transmit antenna, by setting N1 = (Nt + 1)/2 or N1 = (Nt - 1)/2. In this way, the number of transmit antennas 106 included in the first subblock (e.g., transmit antennas 106 transmitting the first chirp signal) and the number of transmit antennas 106 included in the second transmission subblock (e.g., transmit antennas 106 transmitting the second chirp signal) may be set to be the same or different by one. By setting the number of transmitting antennas 106 included in the first and second sub-blocks to be equal or approximately equal, the radar device 10a can achieve the effect of further increasing (for example, approximately doubling) the Doppler multiplexing interval when performing Doppler multiplexing using the first chirp signal and the second chirp signal compared to embodiment 1.

Furthermore, in the first embodiment, the first chirp signal and the second chirp signal are switched in a time-division manner (for example, the first chirp signal and the second chirp signal are alternately transmitted every transmission period T _r ). Therefore, Doppler-multiplexed signals are multiplexed and transmitted in a Doppler frequency range of ±½T _rs (where T _rs > T _r ). On the other hand, in this embodiment, as shown in FIG. 12 , the first chirp signal and the second chirp signal are simultaneously transmitted every transmission period T _r , so that Doppler-multiplexed signals can be multiplexed in a Doppler frequency range of ±½T _r . For example, compared to the case of Doppler-multiplexed transmission at T _rs =2T _r in the first embodiment, this embodiment enables multiplexed transmission of Doppler-multiplexed signals in a Doppler frequency range twice as wide. Therefore, by making the number of transmitting antennas 106 included in the first and second transmitting sub-blocks the same or approximately the same, the Doppler multiplexing interval when Doppler multiplexing using the first chirp signal and the second chirp signal can be expanded by approximately four times compared to embodiment 1.

For example, if the Doppler multiplexing interval during Doppler multiplexing transmission is close, interference between Doppler multiplexed signals is more likely to occur in targets with spread Doppler components, resulting in poor direction estimation accuracy and poor target detection accuracy. In this embodiment, the Doppler multiplexing interval during Doppler multiplexing can be extended, reducing the occurrence of such interference between Doppler multiplexed signals and suppressing deterioration in direction estimation accuracy and target detection accuracy.

In addition, in this embodiment, the Doppler multiplexing interval can be extended when performing Doppler multiplexing, so even if more transmitting antennas 106 are used for Doppler multiplexing, interference between Doppler multiplexed signals can be reduced, and degradation in direction estimation accuracy and target detection accuracy can be suppressed. Therefore, in this embodiment, more transmitting antennas 106 can be used for Doppler multiplexing compared to embodiment 1. In this way, when performing Doppler multiplexing using more transmitting antennas 106, this embodiment is more preferable than embodiment 1.

[Configuration example of radar receiver 200a]
11 , the radar receiver 200a includes, for example, Na receiving antennas 202 (e.g., Rx#1 to Rx#Na) forming an array antenna. The radar receiver 200a also includes, for example, Na antenna system processors 201-1 to 201-Na, a CFAR unit 211, a Doppler demultiplexing unit 212, a Doppler determination unit 213, and a direction estimation unit 214.

Each receiving antenna 202 receives a reflected wave signal, which is a radar transmission signal (e.g., a first chirp signal and a second chirp signal) reflected from a target, and outputs the received reflected wave signal to the corresponding antenna system processing unit 201 as a received signal.

Each antenna system processing unit 201 has a receiving radio unit 203 and a signal processing unit 206.

The receiving radio unit 203 includes a mixer unit 204 and an LPF 205. In the receiving radio unit 203, the mixer unit 204 mixes the received reflected wave signal (received signal) with a chirp signal, which is a transmission signal.

Here, the first chirp signal output from the first radar transmission signal generation unit 101-1 (e.g., VCO 103) is input to the mixer units 204 of, for example, N3 antenna system processing units 201 (e.g., antenna system processing units 201-1 to 201-N3) out of the Na antenna system processing units 201. In the antenna system processing units 201-1 to 201-N3, the output of the mixer units 204 is passed through the LPFs 205, so that the output of the mixer units 204 corresponding to the reflected wave of the second chirp signal has a high frequency outside the passband of the LPFs 205. This makes it easier for the LPFs 205 to output a beat signal with a frequency corresponding to the delay time of the reflected wave signal of the first chirp signal.

Similarly, the second chirp signal output from the second radar transmission signal generator 101-2 (e.g., VCO 103) is input to the mixer units 204 of, for example, N4 antenna system processors 201 (e.g., antenna system processors 201-N3+1 to 201-Na) among the Na antenna system processors 201. In the antenna system processors 201-N3+1 to 201-Na, the output of the mixer units 204 is passed through the LPFs 205, so that the output of the mixer units 204 corresponding to the reflected wave of the first chirp signal has a high frequency outside the passband of the LPF. This makes it easier for the LPFs 205 to output a beat signal with a frequency corresponding to the delay time of the reflected wave signal of the second chirp signal.

Therefore, antenna system processing units 201-1 to 201-N3 process the reflected wave signals of the first chirp signal received by receiving antennas 202-1 to 202-N3. Hereinafter, the antenna system processing unit 201 (e.g., receiving radio unit 203 and signal processing unit 206) that processes the reflected wave of the first chirp signal, and the receiving antenna 202 connected to the antenna system processing unit 201 that processes the reflected wave of the first chirp signal, will be referred to as the "first receiving sub-block."

Furthermore, antenna system processing units 201-N3+1 to 201-Na process the reflected wave signals of the second chirp signal received by receiving antennas 202-N3+1 to 202-Na. Hereinafter, the antenna system processing unit 201 (e.g., receiving radio unit 203 and signal processing unit 206) that processes the reflected wave of the second chirp signal, and the receiving antenna 202 connected to the antenna system processing unit 201 that processes the reflected wave of the second chirp signal, will be referred to as the "second receiving sub-block."

Here, N3 + N4 = Na. Note that N3 and N4 can each be 1 or greater, and Na can be 2 or greater.

The signal processing unit 206 of each antenna system processing unit 201-z _q included in the q-th receiving sub-block has an A/D conversion unit 207, a beat frequency analysis unit 208, and a Doppler analysis unit 210. Here, when q=1, z ₁ = any one of 1 to N3, and when q=2, z ₂ = any one of N3+1 to Na.

The signal (e.g., beat signal) output from the LPF 205 is converted into discrete sample data by the A/D conversion unit 207 in the signal processing unit 206.

The beat frequency analyzer 208 included in the qth receiving subblock performs FFT processing on N _data pieces of discrete sample data obtained within a predetermined time range (range gate) for each transmission period Tr. Here, the range gate sets the frequency sweep time _Tsw (q). For example, q=1 or 2. When q=1, _Tsw (1) represents the frequency sweep time of the first chirp signal, and when q=2, _Tsw (2) represents the frequency sweep time of the second chirp signal. As a result, the signal processor 206 outputs a frequency spectrum in which a peak appears at the beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave). During FFT processing, the beat frequency analyzer 208 may multiply the data by a window function coefficient, such as a Han window or a Hamming window. Using the window function coefficient can suppress side lobes that occur around the beat frequency peak.

Here, the beat frequency response output from the beat frequency analysis unit 208 in the zqth signal processing unit 206 of the qth receiving sub-block, obtained by transmitting the mth chirp pulse of the qth _chirp signal, is represented by "RFT _zq (f _b , m)." Here, f _b represents the beat frequency index, which corresponds to the FFT index (bin number). For example, f _b = 0, to, N _data /2-1, z ₁ = 1 to N3, z ₂ = N3+1 to Na, m = 1, to, N _C , and q = 1 or 2. The smaller the beat frequency index f _b , the smaller the delay time of the reflected wave signal (e.g., the closer the distance to the target).

The Doppler analyzer 210 in the _zqth signal processor 206 of the qth receiving sub-block performs Doppler analysis for each range index f b using beat frequency responses RFT _zq (f _b , 1), RFT _zq (f _b , 2), ..., RFT _zq (f _b , _NC ) obtained by transmitting chirp pulses of the qth chirp signal N _{C times} . For example, the Doppler analyzer 210 of the qth receiving sub-block may estimate the Doppler frequency from the reflected wave signal of the qth chirp signal reflected by the target.

For example, if _Nc is a power of 2, FFT processing can be applied in Doppler analysis. In this case, the FFT size is _Nc , and the maximum Doppler frequency at which aliasing does not occur, as derived from the sampling theorem, is ±1/( _2Tr ). Furthermore, the Doppler frequency interval of the Doppler frequency index _fs is 1/( _Nc × _Tr ), and the range of the Doppler frequency index _fs is _fs = _-Nc /2, ∼, 0, ∼, _Nc /2-1.

In the following, an example will be described in which _Nc is a power of 2. If _Nc is not a power of 2, FFT processing can be performed with a data size of a power of 2 by, for example, including zero-padded data. Furthermore, the Doppler analyzer 210 may multiply the data by a window function coefficient such as a Han window or a Hamming window during FFT processing. Applying a window function can suppress side lobes that occur around the beat frequency peak.

For example, the output VFT _zq ( _fb , _fs ) of the Doppler analyzer 210 in the _zqth signal processor 206 of the qth receiving subblock is shown in the following equation (60), where j is the imaginary unit, _z1 = 1 to N3, _z2 = N3 + 1 to Na, and q = 1, 2.

The above explains the processing performed by each component of the signal processing unit 206.

The CFAR unit 211 may, for example, include a first CFAR unit 211 (or CFAR unit 211-1) and a second CFAR unit 211 (or CFAR unit 211-2) corresponding to the first chirp signal and the second chirp signal, respectively, which have different center frequencies. Similarly, the Doppler demultiplexing unit 212 may, for example, include a first Doppler demultiplexing unit 212 (or Doppler demultiplexing unit 212-1) and a second Doppler demultiplexing unit 212 (or Doppler demultiplexing unit 212-2) corresponding to the first chirp signal and the second chirp signal, respectively, which have different center frequencies.

Note that while Figure 11 shows a configuration in which CFAR units 211 are provided in parallel (CFAR units 211-1 and 211-2), it is also possible to provide a single CFAR unit 211 and sequentially switch and process its inputs. Note that while Figure 11 shows a configuration in which Doppler demultiplexing units 212 are provided in parallel (Doppler demultiplexing units 212-1 and 212-2), it is also possible to provide a single Doppler demultiplexing unit 212 and sequentially switch and process its inputs.

In FIG. 11 , a CFAR unit 211 performs CFAR processing (e.g., adaptive threshold determination) using the output from the Doppler analysis unit 210 of the signal processing unit 206 included in the q-th receiving sub-block, and extracts a distance index f _{b_cfar} and a Doppler frequency index f _{s_cfar} that give a local peak signal.

As shown in FIG. 11, the CFAR unit 211 may include a first CFAR unit 211 (or CFAR unit 211-1) that performs CFAR processing using the output of the Doppler analysis unit 210 of the signal processing unit 206 included in the first reception sub-block, and a second CFAR unit 211 (or CFAR unit 211-2) that performs CFAR processing using the output of the Doppler analysis unit 210 of the signal processing unit 206 included in the second reception sub-block.

The qth CFAR unit 211 (q=1, 2) performs power addition on the outputs of the Doppler analysis units 210 of the signal processing units 206 included in the qth receiving sub-block, and performs two-dimensional CFAR processing consisting of a distance axis and a Doppler frequency axis (corresponding to relative velocity), or CFAR processing that combines one-dimensional CFAR processing, as shown in the following equation (61). As for the two-dimensional CFAR processing or the CFAR processing that combines one-dimensional CFAR processing, for example, the processing disclosed in Non-Patent Document 2 may be applied.
Here, z ₁ = 1 to N3, and z ₂ = N3 + 1 to Na.

The qth CFAR unit 211 adaptively sets a threshold value and outputs to the qth Doppler demultiplexing unit 212 a distance index f _{b_cfar} (q), a Doppler frequency index f _{s_cfar} (q), and received power information PowerFT(f _{b_cfar} (q), f _{s_cfar} (q)) that result in received power greater than the threshold value.

The Doppler demultiplexing unit 212 may include a first Doppler demultiplexing unit 212 (or referred to as Doppler demultiplexing unit 212-1) that performs Doppler demultiplexing processing using the outputs of the Doppler analysis unit 210 and first CFAR unit 211 of the signal processing unit 206 included in the first receiving sub-block, and a second Doppler demultiplexing unit 212 (or referred to as Doppler demultiplexing unit 212-2) that performs Doppler demultiplexing processing using the outputs of the Doppler analysis unit 210-2 and second CFAR unit 211 of the signal processing unit 206 included in the second receiving sub-block.

The q-th Doppler multiplexing separation unit 212 (q=1, 2) separates the transmission signals transmitted from each transmitting antenna ₁₀₆ (e.g., reflected wave signals corresponding to the transmission signals) from the Doppler _multiplexed signals (hereinafter referred to as "Doppler multiplexed signals") using the output from the Doppler analysis unit 210 included in the q-th receiving sub-block, based on information input from the q-th CFAR unit 211 (e.g., distance index f _{b_cfar} (q), _Doppler frequency index f s_cfar (q), and reception power information PowerFT(f b_cfar (q), f s_cfar (q))).

The q-th Doppler demultiplexing unit 212 outputs, for example, information about the separated signals to the Doppler determination unit 213 and the direction estimation unit 214. The information about the separated signals may include, for example, a distance index f _{b_cfar} (q) corresponding to the separated signals and a Doppler frequency index (hereinafter sometimes referred to as separation index information). Here, the separation index information of the first Doppler demultiplexing unit 212 is the Doppler frequency index obtained by separating the signals transmitted from the transmitting antennas Tx#1, Tx#2, to, Tx#N1 included in the first transmission sub-block, and is represented as (f _{demul_Tx#1} , f _{demul_Tx#2} , to, f _{demul_Tx#N1} ) corresponding to each. Similarly, the separation index information of the second Doppler multiplex separation unit 212 is the Doppler frequency index separated from the signals transmitted from the transmitting antennas Tx#N1+1, Tx#N1+2, ..., Tx#Nt included in the second transmission sub-block, and is represented as (f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , ..., f _{demul_Tx#Nt} ) corresponding to each.

Furthermore, the qth Doppler demultiplexing unit 212 outputs the output from the qth Doppler analysis unit 210 to the direction estimation unit 214. Note that the qth Doppler demultiplexing unit 212 may output the output from the Doppler analysis unit 210 included in the qth reception sub-block to the direction estimation unit 214 based on information input from the qth CFAR unit _{211 (e.g., distance index f b_cfar (} q), Doppler frequency index f _{s_cfar} (q), and reception power information PowerFT(f _{b_cfar} (q), f s_cfar (q))).

The operation of the qth Doppler demultiplexing unit 212 will be explained below, along with the operation of the Doppler shift unit 105 in the radar transmitter 100a.

[How to set the Doppler shift amount]
The Doppler shift units 105-1 to 105-N1 included in the first transmission sub-block impart a phase rotation φ _nsub1 to the first chirp signal in order to impart a Doppler shift amount DOP _nsub1 to the first chirp signal for each transmission period T _{r of} the chirp signal, and output the Doppler-shifted signal to the transmitting antennas 106 (for example, Tx#1 to Tx#N1). Here, nsub1 = 1 to N1. In this embodiment, the intervals between the Doppler shift amounts DOP _nsub1 (Doppler shift intervals) between the Doppler shift units 105-1 to 105-N1 (or between the transmitting antennas 106-1 to 106-N1) may not be equal, but may be set to unequal intervals so that at least one Doppler interval is different.

For example, the nsub1th Doppler shifter 105 applies a phase rotation φ _nsub1 (m) to the mth input first chirp signal, resulting in a different Doppler shift amount DOP _nsub1 , and outputs the applied signal. As a result, different Doppler shift amounts are applied to the transmission signals transmitted from the multiple transmitting antennas 106. For example, in this embodiment, the Doppler multiplexing number N _DM =N1 may be used. Here, m is an integer from 1 to _NC , and nsub1 is an integer from 1 to N1.

Similarly, the Doppler shift units 105-N1+1 to 105-Nt included in the second transmission sub-block impart a phase rotation φ _nsub2 to the second chirp signal to impart a Doppler shift amount DOP _nsub2 to the second chirp signal for each transmission period T _r of the chirp signal, and output the Doppler-shifted signal to the transmitting antennas 106 (e.g., Tx#N1+1 to Tx#Nt). Here, nsub2 = 1 to N2. In this embodiment, the intervals between the Doppler shift amounts DOP _nsub2 (Doppler shift intervals) between the Doppler shift units 105-N1+1 to 105-Nt (or between the transmitting antennas 106-N1+1 to 106-Nt) may not be equal, but may be set to unequal intervals so that at least one Doppler interval is different.

In this way, by setting the Doppler shift amounts in the Doppler shift units 105 of the first and second transmission sub-blocks, a first Doppler demultiplexing unit 212 (described later) can separate the Doppler indices Tx#1 to Tx#N1 and calculate Doppler frequencies within the range of ±1/ _Tr by operating in the same manner as the Doppler demultiplexing unit 212 in Embodiment 1. Furthermore, a second Doppler demultiplexing unit 212 (described later) can separate the Doppler indices Tx#N1+1 to Tx#Nt and calculate Doppler frequencies within the range of ±1/ _Tr by operating in the same manner as the Doppler demultiplexing unit 212 in Embodiment 1. Based on the outputs of these Doppler demultiplexing units 212, a Doppler determination unit 213 (described later) can determine whether or not the Doppler frequency has aliased based on the difference in Doppler frequency and can calculate a Doppler frequency that exceeds the range of ±1/ _Tr by operating in the same manner as the Doppler determination unit 213 in Embodiment 1.

The operation of the Doppler shift units 105-1 to 105-N1 included in the first transmission sub-block is the same as the operation described in the description of the operation between the Doppler shift units 105-1 to 105-Nt (or between the transmitting antennas 106-1 to 106-Nt) in embodiment 1, except that "Nt" is replaced with "N1" and "T _rs " is replaced with "T _r ", and therefore a detailed description of the operation will be omitted.

Furthermore, the operation of Doppler shift units 105-N1+1 to 105-Nt included in the second transmission sub-block is the same as the operation described in the description of the operation between Doppler shift units 105-N1+1 to 105-Nt (or between transmitting antennas 106-N1+1 to 106-Nt) in embodiment 1, except that "Nt" is replaced with "N2" and "T _rs " is replaced with "T _r ", and therefore a detailed description of the operation will be omitted.

For example, in relation to equation (5) described in embodiment 1, in this embodiment, the Doppler shift units 105-1 to 105-N1 of the first transmission sub-block impart a phase rotation φ nsub1 (m) as shown in the following equation (62) to the input mth first chirp signal, which results in a Doppler shift amount DOP _nsub1 that differs between each Doppler shift unit ₁₀₅ .

Similarly, the Doppler shift units 105-N1+1 to 105-Nt of the second transmission sub-block apply a phase rotation φ _nsub2 (m) to the input mth second chirp signal, as shown in the following equation (63), which results in a Doppler shift amount DOP _nsub2 that differs between each Doppler shift unit 105.

Here, A is a coefficient that assigns a positive or negative polarity of 1 or -1. δ1 and δ2 are integers equal to or greater than 1. The terms round(N _C /(N1 + δ1)) and round(N _C /(N2 + δ2)) are introduced for the purpose of making the amount of phase rotation an integer multiple of the Doppler frequency interval in Doppler analysis unit 210, but this is not limitative. 2π/(N1 + δ1) may be used instead of the term (2π/N _C ) × round(N _C /(N1 + δ1)) in equation (62). Similarly, 2π/(N2 + δ2) may be used instead of the term (2π/N _C ) × round(N _C /(N2 + δ2)) in equation (63).

Furthermore, φ ₀₁ and φ ₀₂ are initial phases, and may be equal to or different from each other. For example, whether φ ₀₁ and φ ₀₂ are equal to or different from each other, the Doppler frequencies will match. Furthermore, Δφ ₀₁ and Δφ ₀₂ are reference Doppler shift phases, and may be equal to or different from each other. For example, whether Δφ ₀₁ and Δφ ₀₂ are equal to or different from each other, the Doppler frequencies will match.

For example, the radar device 10a performs unequal Doppler multiplexing on the first chirp signal and the second chirp signal with Doppler multiplexing numbers N1 and N2, respectively.

In the following, (2π/N _C )×round(N _C /(N1 + δ1)) or 2π/(N1 + δ1) in equation (62) will be referred to as the Doppler multiplexing interval "ΔDOP _min1 " of the Doppler multiplexed signal assigned to the first chirp signal. Similarly, (2π/N _C )×round(N _C /(N2 + δ2)) or 2π/(N2 + δ2) in equation (63) will be referred to as the Doppler multiplexing interval "ΔDOP _min2 " of the Doppler multiplexed signal assigned to the second chirp signal.

As an example when the number of transmitting antennas Nt=4, in equation (62), N1=2, Δφ ₀₁ =0, φ ₀₁ =0, A=1, δ1=1, and N _C is a multiple of 3, and a phase rotation φ _nsub1 (m)=2π(nsub1-1)×(m-1)/3 is applied to the first chirp signal for each transmission period T _r , so the Doppler shift amounts are DOP ₁ =φ ₁ (m)/{2π(m-1)T _r }=0, DOP ₂ =φ ₂ (m)/{2π(m-1)T _r }=1/(3T _r ). The upper rows of each of (a) to (d) in Figure 13 show examples of arrangement of Doppler multiplexed signals when transmitting the first chirp signal.

As an example when the number of transmitting antennas is Nt=4, in equation (63), Nt=4, N2=2, Δφ ₀₂ =0, φ ₀₂ =0, A=1, δ2=1, and N _C is a multiple of 3, and a phase rotation φ _nsub2 (m)=2π(nsub2-1)×(m-1)/3 is applied to the second chirp signal for each transmission period T _r , so the Doppler shift amounts are DOP ₁ =0, DOP ₂ =1/(3T _r ). The lower rows of each of (a) to (d) in Figure 13 show examples of arrangement of Doppler-multiplexed signals when transmitting the second chirp signal.

The arrangement of the Doppler multiplexed signals for the first chirp signal and the second chirp signal shown in Figure 13(a) is the same. This arrangement makes it easy for the Doppler determination unit 213 (described later) to calculate the Doppler frequency shift in the Doppler multiplexed signal multiplexed onto the first chirp signal and the second chirp signal when the first chirp signal and the second chirp signal are received.

Note that the arrangement of the Doppler multiplexed signals is not limited to the example shown in Figure 13(a). For example, the arrangement of the Doppler multiplexed signals for the first chirp signal and the second chirp signal may be different, as shown in Figure 13(b), Figure 13(c), and Figure 13(d).

For example, in FIG. 13(b), by setting Δφ ₀₂ ≠ 0 in equation (63), the Doppler multiplexed signal for the second chirp signal may be arranged with an offset by a predetermined Doppler frequency from the arrangement of the Doppler multiplexed signal for the first chirp signal.

Also, for example, in (c) of Figure 13, two Doppler-multiplexed signals are assigned to each of the first chirp signal and the second chirp signal, out of the Doppler range of ±1/ _Tr divided into three (=(N2+δ2)). In (c) of Figure 13, nsub1 = 1, 2 in equation (62) may be assigned to the Doppler-multiplexed signal for the first chirp signal, and nsub2 = 2, 3 in equation (63) may be assigned to the Doppler-multiplexed signal for the second chirp signal. For example, the assignment of Doppler-multiplexed signals within the (N1+δ1) (=(N2+δ2)) division may be made different for the Doppler-multiplexed signals between the first chirp signal and the second chirp signal.

Also, for example, in (d) of Figure 13, by setting δ2 = 2 in equation (63), a different Doppler multiplexing interval (e.g., ΔDOPmin1 ≠ ΔDOPmin2) may be applied to the Doppler multiplexed signal for the second chirp signal than to the Doppler multiplexing interval of the Doppler multiplexed signal for the first chirp signal.

Also, as an example when the number of transmitting antennas Nt=5, in equation (62), N1=2, Δφ ₀₁ =0, φ ₀₁ =0, A=1, δ1=2, and N _C is a multiple of 4, a phase rotation φ _nsub1 (m)=π(nsub1-1)×(m-1)/2 is applied to the first chirp signal for each transmission period T _r , so the Doppler shift amounts are DOP ₁ =φ ₁ (m)/{2π(m-1)T _r }=0, DOP ₂ =φ ₂ (m)/{2π(m-1)T _r }=1/(4T _r ). The upper rows of each of (a) to (d) in Figure 14 show examples of arrangement of Doppler multiplexed signals when transmitting the first chirp signal.

As an example of the case where the number of transmitting antennas is Nt=5, in equation (63), Nt=5, N2=3, Δφ ₀₂ =0, φ ₀₂ =0, A=1, δ2=1, and N _C is a multiple of 4, and a phase rotation φ _nsub2 (m)=π×(m-1)/2 is applied to the second chirp signal for each transmission period T _r , so the Doppler shift amounts are DOP ₁ =0, DOP ₂ =1/(4T _r ). The lower rows of each of (a) to (d) in Figure 14 show examples of arrangement of Doppler-multiplexed signals when transmitting the second chirp signal.

The arrangement of the Doppler multiplexed signals for the first chirp signal and the second chirp signal shown in (a) of Figure 14 is the same. This arrangement makes it easy for the Doppler determination unit 213 (described later) to calculate the Doppler frequency shift in the Doppler multiplexed signal multiplexed onto the first chirp signal and the second chirp signal when the first chirp signal and the second chirp signal are received.

Note that the arrangement of the Doppler multiplexed signals is not limited to the example shown in Figure 14(a). For example, the arrangement of the Doppler multiplexed signals for the first chirp signal and the second chirp signal may be different, as shown in Figure 14(b), Figure 14(c), and Figure 14(d).

For example, in FIG. 14(b), by setting Δφ ₀₂ ≠ 0 in equation (63), the Doppler multiplexed signal for the second chirp signal may be arranged with an offset by a predetermined Doppler frequency from the arrangement of the Doppler multiplexed signal for the first chirp signal.

Also, for example, in (c) of Figure 14, the Doppler range of ±1/ _Tr is divided into four (=(N2+δ2)), and two or three Doppler-multiplexed signals are assigned to each of the first chirp signal and the second chirp signal. In (c) of Figure 14, nsub1 = 1, 2 in equation (62) may be assigned to the Doppler-multiplexed signal for the first chirp signal, and nsub2 = 3, 4, 5 in equation (63) may be assigned to the Doppler-multiplexed signal for the second chirp signal. For example, the assignment of Doppler-multiplexed signals within the (N1+δ1) (=(N2+δ2)) division may be made different for the Doppler-multiplexed signals between the first chirp signal and the second chirp signal.

Also, for example, in (d) of Figure 14, by setting δ1 = 1 in equation (62), a different Doppler multiplexing interval (e.g., ΔDOPmin1 ≠ ΔDOPmin2) may be applied to the Doppler multiplexed signal for the second chirp signal than to the Doppler multiplexing interval of the Doppler multiplexed signal for the first chirp signal.

[Example of operation of Doppler demultiplexing unit 212]
The first Doppler multiplex separation unit 212 separates and receives signals that have been transmitted such that the intervals between the Doppler shift amounts DOP _nsub1 (Doppler shift intervals) between the Doppler shift units 105-1 to 105-N1 included in the first transmission sub-block (for example, between the transmission antennas 106-1 to 106-N1) are not equal, but at least one Doppler interval is different.

The first Doppler demultiplexing unit 212 demultiplexes the signals that have been Doppler multiplexed at uneven intervals based on the outputs of the Doppler analysis unit 210 and the CFAR unit 211 in the first reception sub-block using the same operation as in Embodiment 1. The first Doppler demultiplexing unit 212 then outputs to the Doppler determination unit 213 the distance index f _{b_cfar} (1), separation index information (f _{demul_Tx #1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) of the N1 Doppler-multiplexed signals multiplexed and transmitted in the first transmission sub-block at the distance index f b_cfar (1), and the output of the Doppler analysis unit 210.

Note that the operation of the Doppler shift units 105-1 to 105-N1 included in the first transmission sub-block is the same as the operation described in the description of the operation between the Doppler shift units 105-1 to 105-Nt (for example, between the transmitting antennas 106-1 to 106-Nt) in embodiment 1, but with N1 replacing Nt and T _rs replacing T _r . Similarly, the operation of the first Doppler demultiplexing unit 212 is the same as the operation described in the description of the operation of the Doppler shift unit 105 in embodiment 1, but with N1 replacing Nt and T _rs replacing T _r . This enables the separation of signals Doppler-multiplexed between the transmitting antennas 106-1 to 106-N1. Therefore, a detailed description of the operation of the first Doppler demultiplexing unit 212 will be omitted.

Furthermore, the second Doppler demultiplexing unit 212 separates and receives signals that have been transmitted with the Doppler shift amounts DOP _nsub2 not spaced at equal intervals (Doppler shift intervals) between the Doppler shift units 105-N1+1 to 105-Nt included in the second transmission sub-block (for example, between the transmission antennas 106-N1+1 to 106-Nt) but with at least one Doppler interval set to be different.

The second Doppler demultiplexing unit 212 demultiplexes the Doppler-multiplexed signals at uneven intervals based on the outputs of the Doppler analysis unit 210 and the CFAR unit 211 in the second reception sub-block using the same operation as in Embodiment 1. The second Doppler demultiplexing unit 212 then outputs to the Doppler determination unit 213 the distance index f _{b_cfar} (2), separation index information (f _{demul_Tx #N1+1} , f _{demul_Tx#N1+2} , ..., f _{demul_Tx#Nt(=N1+N2)) of the N2 Doppler-multiplexed signals multiplexed and transmitted in the second transmission sub-block at the distance index f b_cfar (2} ), and the output of the Doppler analysis unit 210.

Incidentally, the operation of Doppler shift units 105-N1+1 to 105-Nt included in the second transmission sub-block is the same as the operation described in the description of the operation between Doppler shift units 105-1 to 105-Nt (for example, between transmitting antennas 106-1 to 106-Nt) in embodiment 1, but with N2 replacing Nt and _Trs replacing _Tr . Similarly, the operation of second Doppler demultiplexing unit 212 is the same as the operation described in the description of the operation of Doppler shift unit 105 in embodiment 1, but with N2 replacing Nt and _Trs replacing _Tr . This enables signals Doppler-multiplexed between transmitting antennas 106-N1+1 to 106-Nt to be separated. Therefore, a detailed description of the operation of second Doppler demultiplexing unit 212 will be omitted.

[Example of operation of Doppler determination unit 213]
11 , the Doppler determination unit 213 determines the Doppler frequency corresponding to the Doppler peak based on the outputs of the first Doppler demultiplexing unit 212 and the second Doppler demultiplexing unit 212. For example, even when a target having a Doppler frequency _{fd_TargetDoppler} outside the Doppler frequency range −1/( _2Tr )≦ _{fd_TargetDoppler} <1/(2Tr ₎ is included, the Doppler determination unit 213 can further expand the Doppler detection range by determining the Doppler frequency of the target.

For example, the Doppler determination unit 213 determines the Doppler frequency of a target including a Doppler frequency exceeding the Doppler frequency range -1/( _{2T r ) ≦ f d_TargetDoppler <} 1/(2T r ) using the separation index information (f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) of the Doppler multiplexed signal output from the first Doppler multiplex separation unit 212 and the separation index information (f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , ..., f _{demul_Tx#Nt} ) of the Doppler multiplexed signal output from the second _Doppler multiplex separation unit ₂₁₂ , which have the same distance indexes f b_cfar ( ₁ ) and f b_cfar (2).

The principle of determining the Doppler frequency of a target including a Doppler frequency that exceeds the Doppler frequency range −1/(2T _r )≦ f _{d_TargetDoppler} < 1/(2T _r ) in the Doppler determination unit 213 is, as in the first embodiment, that the center frequencies of the first chirp signal and the second chirp signal, which are radar transmission signals generated by the signal generation control unit 104 and the radar transmission signal generation unit 101, are different from each other.

In the first embodiment, the Doppler frequency is determined based on changes in the Doppler frequency in the Doppler multiplexed signal for the same transmitting antenna 106. In contrast, in the present embodiment, the Doppler determination unit 213 determines the Doppler frequency based on changes in the Doppler frequency in the Doppler multiplexed signal for a different transmitting antenna 106. When a different transmitting antenna 106 is used, the reception phase changes, but the received Doppler frequency does not change. Therefore, similar to the first embodiment, the Doppler determination unit 213 can determine the Doppler frequency of a target including a Doppler frequency that exceeds the Doppler frequency range -1/( _2Tr ) ≦ _{fd_TargetDoppler} < 1/( _2Tr ).

The operating principle of the Doppler frequency determination process and an example of the operation of the Doppler determination unit 213 differ from the operating principle of the Doppler frequency determination process and the example of the operation of the Doppler determination unit 213 described in embodiment 1 in the following points (1), (2), and (3).
(1) Replacing "T _rs " with "T _r ",
(2) In embodiment 1, the Doppler determination unit 213 calculates the Doppler frequency estimate value f _{d_VFT} (1) based on the separation index information (f _{demul_Tx#1} (1), f _{demul_Tx#2} (1), ..., f _{demul_Tx#Nt} (1)) of the Doppler multiplexed signal in the distance index f b_cfar (1) output from the first Doppler _multiplex separation unit 212, assuming that the Doppler frequency of the target is within the Doppler frequency range -1/(2T _rs ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ). In contrast to this, in this embodiment, the Doppler determination unit 213 calculates a Doppler frequency estimate f _{d_VFT} (1) assuming that the Doppler frequency of the target is within the Doppler frequency range −1/(2T r )≦f _{d_TargetDoppler} <1/(2T _r ) based on the separation index information (f _{demul_Tx#1} , f _{demul_Tx# 2} , . . . , f demul_Tx# _N1 ) of the Doppler multiplexed signal in the distance index f _{b_cfar} (1) output from the first Doppler multiplex separation unit 212; and
(3) In embodiment 1, the Doppler determination unit 213 calculates the Doppler frequency estimate value f _{d_VFT} (2) based on the separation index information (f _{demul_Tx#1} (2), f _{demul_Tx#2} (2), ..., f _{demul_Tx#Nt} (2)) of the Doppler multiplexed signal in the distance index f b_cfar (2) output from the second Doppler _multiplex separation unit 212, assuming that the Doppler frequency of the target is within the Doppler frequency range -1/(2T _rs ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ). In contrast to this, in this embodiment, the Doppler determination unit 213 calculates the Doppler frequency estimate value f _{d_VFT} (2) based on the separation index information (f _{demul_Tx#N1+1} , f _{demul_Tx# N1+2} , ..., f _{demul_Tx#Nt} ) of the Doppler multiplexed signal in the distance index f b_cfar (2) output from the second Doppler multiplex separation unit 212, assuming that the Doppler frequency of the target is within the _Doppler frequency range -1/(2T _r ) ≦ f _{d_TargetDoppler} < 1/(2T _r ).

In this embodiment, the operation of the Doppler determination unit 213, except for the three differences mentioned above, is the same as in embodiment 1, so a description of that operation will be omitted.

Furthermore, similarly to the Doppler determination unit 213 in the first embodiment, by setting center frequencies f c (1) and f _c (2) that satisfy the determination condition of any one of equations (10) to (15) in which _{" T} rs " is replaced with " _T r ", the Doppler determination unit 213 can determine the Doppler frequency of a target even when a target with a Doppler frequency outside the Doppler frequency range -1/(2T _r ) ≦ f _{d_TargetDoppler} < 1/(2T _r ) is included (for example, when Doppler aliasing occurs). Note that although equations (10) and (13) include T _rs , by rearranging them, equations that do not include T _rs , such as equations (11), (12), (14), and (15), can be obtained. Therefore, the center frequencies f _c (1) and f _c (2) that satisfy the determination condition have the same conditions as those in the first embodiment.

Incidentally, equation (18) used in the description of the first embodiment can be expressed as the following equation (64) by replacing T _rs with T _r .

Equation (64) expresses the condition that, when the Doppler aliasing number is n _al , the difference between the aliasing component n _al ×{ _{f c (} 2)/f _c (1)}/ _Tr of the Doppler frequency observed using the second chirp signal with a center frequency f c (2) and n _al / _Tr , the frequency interval of the aliasing number n _al in the first Doppler analysis unit 210, does not exceed ±1/( _2Tr ). For example, when n _al is positive, Δn _al is in the range of ±1/( _2Tr ) up to the maximum n _al that satisfies Equation (19), and the Doppler determination unit 213 can estimate the aliasing without ambiguity. The maximum n _al that satisfies Equation (19) is denoted as "n _almax ." For example, when n _al is negative, setting n _al = -n _almax similarly satisfies Equation (19).

In the first embodiment, the first chirp signal or the second chirp signal is transmitted at a period of T _rs , and the Doppler frequency detection range is expanded, for example, by n _almax times compared to the Doppler frequency range when one transmitting antenna is used. On the other hand, in the present embodiment, the first chirp signal and the second chirp signal are transmitted at a period of T _r , and the Doppler frequency detection range is expanded, for example, by 2×n _almax times compared to the Doppler frequency range when one transmitting antenna is used. Therefore, in this embodiment, the Doppler frequency detection range can be expanded further for the same chirp signal center frequencies f _c (1) and f _c (2) compared to the first embodiment.

For example, if f _c (1) and f _c (2) satisfy the condition when n _almax =1, the detection range of the Doppler frequency f _d becomes ±1/(T _r ), which is doubled compared to the Doppler frequency range of ±1/(2T _r ) when there is one transmitting antenna. Also, for example, if f _c (1) and f _c (2) satisfy the condition when n _almax =2, the detection range of the Doppler frequency f _d becomes ±2/(T _r ), which is quadrupled compared to the Doppler frequency range of ±1/(2T _r ) when there is one transmitting antenna.

The above explains an example of the operation of the Doppler determination unit 213.

[Example of operation of direction estimation unit 214]
In Figure 11, the direction estimation unit 214 extracts the output of the first Doppler analysis unit 210 and the output of the second Doppler analysis unit 210 based on information input from the first Doppler multiplex separation unit _{212 (e.g., distance index fb_cfar (} 1) and separation index information of the Doppler multiplexed signal at distance index _{fb_cfar} (1) (fdemul_Tx# ₁ , _{fdemul_Tx#2} , ..., _{fdemul_Tx#N1} )) and information input from the second Doppler multiplex separation unit 212 (e.g., distance index _{fb_cfar (2) and separation index information of the Doppler multiplexed signal at distance index fb_cfar} (2) ( _{fdemul_Tx#N1+1} , fdemul_Tx# _N1+2 , ..., _{fdemul_Tx#Nt} )), and performs target direction estimation processing.

In this embodiment, N1 x N3 MIMO virtual receive antennas are formed between the N1 transmit antennas 106 included in the first transmit subblock and the N3 receive antennas 202 included in the first receive subblock. Similarly, N2 x N4 MIMO virtual receive antennas are formed between the N2 transmit antennas 106 included in the second transmit subblock and the N4 receive antennas 202 included in the second receive subblock. The direction estimation unit 214 may perform direction estimation processing using these two sets of virtual receive antennas.

For example, the direction estimation unit 214 extracts the output of the first Doppler analysis unit 210 from the output of the first Doppler multiplex separation unit 212 based on the distance index f _{b_cfar} (1) and the separation index information of the Doppler multiplexed signal (f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ), and generates a first virtual receiving array correlation vector h ₁ (f _{b_cfar} (1), f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) consisting of N1 × N3 elements as shown in the following equation (65).

Furthermore, the direction estimation unit 214 extracts the output of the second Doppler analysis unit 210 from the output of the second Doppler multiplex separation unit 212, for example, based on the distance index f _{b_cfar} (2) and the separation index information of the Doppler multiplexed signal (f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , . . . , f _{demul_Tx#Nt} ), and generates a second virtual receiving array correlation vector h ₂ (f _{b_cfar} (2), f _{demul_Tx#N1+1 , f demul_Tx#N1+ 2} , . . . , f _{demul_Tx#Nt} ) consisting of N2 × N4 elements as shown in the following equation (66).

In addition, since the direction estimation unit 214 performs direction estimation processing using the outputs of the first and second Doppler demultiplexing units 212 with the same distance index, in equations (65) and (66), f _{b_cfar} (1) = f _{b_cfar} (2) = f _{b_cfar} .

In equation (65), h _1cal[b] is an array correction value that corrects the phase deviation and amplitude deviation between the transmitting antennas Tx#1 to Tx#N1 and between the receiving antennas Rx#1 to #N3, where b is an integer between 1 and (N3×N1).

In addition, in equation (66), _h2cal[b] is an array correction value that corrects the phase deviation and amplitude deviation between the transmitting antennas Tx#N1+1 to Tx#Nt and between the receiving antennas Rx#N3+1 to #Na, where b is an integer between 1 and (N4 × N2).

The direction estimation unit 214 calculates a spatial profile by varying the azimuth direction _θu in the direction estimation evaluation function value P _H ( _θu , f _{b_cfar} , f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#Nt} ) within a predetermined angle range. The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial profile in descending order, and outputs the azimuth directions of the maximum peaks as direction-of-arrival estimates (e.g., positioning outputs). Here, f _{b_cfar} represents a distance index such that f _{b_cfar} (1) = f _{b_cfar} (2).

Note that there are various methods for calculating the direction estimation evaluation function value P _H (θ _u , f _{b_cfar} , f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#Nt} ) depending on the direction of arrival estimation algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

For example, the beamformer method can be expressed as in the following equation (67): In addition to the beamformer method, methods such as Capon and MUSIC can also be applied. In equation (67), the superscript H is the Hermitian transpose operator.

In equation (67), a ₁ (θ _u ) represents the direction vector (column vector with N1×N3 elements) of the N1×N3 MIMO virtual receiving antennas formed between the N1 transmitting antennas 106 included in the first transmitting sub-block and the N3 receiving antennas 202 included in the first receiving sub-block, and represents the phase response or complex amplitude response at each virtual receiving antenna forming the N1×N3 MIMO virtual receiving antennas when a reflected wave arrives from the θ _u direction.

Similarly, in equation (67), a ₂ (θ _u ) represents the direction vector (column vector with N2 × N4 elements) of the N2 × N4 MIMO virtual receiving antennas formed between the N2 transmitting antennas 106 included in the second transmitting sub-block and the N4 receiving antennas 202 included in the second receiving sub-block, and represents the phase response or complex amplitude response at each virtual receiving antenna forming the N2 × N4 MIMO virtual receiving antennas when a reflected wave arrives from the θ _u direction.

The direction vector _aq ( _θu ) may be the phase response or complex amplitude response at each virtual receiving antenna when the wavelength of the radar transmission signal (e.g., the qth chirp signal) at the center frequency _fc (q) is used. Alternatively, the direction vector a( _θu ) of the virtual receiving array for the arriving wave in the azimuth direction θ at the average center frequency of the center frequencies _fc (1) and _fc (2) may be used in common.

The azimuth direction θ _u is a vector obtained by varying the azimuth range in which the direction of arrival estimation is performed at a predetermined azimuth interval β _1. For example, θ _u is set as follows.
θ _u ＝θmin + uβ ₁ , integer u=0～NU
NU=floor[(θmax-θmin)/β ₁ ]+1
Here, floor(x) is a function that returns the maximum integer value that does not exceed the real number x.

In the above example, the direction estimation unit 214 calculates the azimuth direction as the direction of arrival estimate, but this is not limited to this. Direction of arrival estimation can also be performed in the elevation direction, or by using MIMO antennas arranged in a rectangular grid, direction of arrival estimation can also be performed in the azimuth direction and elevation direction. For example, the direction estimation unit 214 may calculate the azimuth direction and elevation direction as the direction of arrival estimate, and use these as the positioning output.

Through the above operations, the direction estimation unit 214 may output, as positioning outputs, distance index f _{b_cfar} and arrival direction estimates for separation index information of Doppler multiplexed signals (f _{demul_Tx#1} , f _{demul_Tx#2} , . . . , f _{demul_Tx#Nt} ). The direction estimation unit 214 may further output, as positioning outputs, distance index f _{b_cfar} and separation index information of Doppler multiplexed signals (f _{demul_Tx#1} , f _{demul_Tx#2} , . . . , f _{demul_Tx#Nt} ). The direction estimation unit 214 may output the positioning output (or the positioning result) to, for example, a vehicle control device in the case of an on-board radar, or an infrastructure control device in the case of an infrastructure radar, both not shown.

In addition, the direction estimation unit 214 may output, for example, either one or both of the Doppler frequency information f _{d_VFT(1)} +n _alest /T _s determined by the Doppler determination unit 213 and f _c (2)/f _c (1)(f _{d_VFT(1)} +n _alest /T _s ).

Furthermore, the distance index f _{b_cfar} may be converted into distance information using, for example, equation (1) and then output.

Furthermore, the Doppler frequency information determined by the Doppler determination unit 213 may be converted into relative velocity information and output. The Doppler frequency information f _{d — VFT(1)} + n _alest /T _s determined by the Doppler determination unit 213 at the center frequency f _c (1) can be converted into the relative velocity v _d using the following equation (68).

Similarly, when the Doppler frequency information f _c (2)/f _c (1) (f _{d_VFT(1)} + n _alest /T _r ) determined by the Doppler determination unit 213 based on the center frequency f _c (2) is converted into relative velocity v _d , the value obtained is the same as equation (68), as shown in the following equation (69). Therefore, the relative velocity information may be output as a common value (or a unified value) for different center frequencies.

As described above, in this embodiment, the radar device 10a includes multiple radar transmission signal generators 101 and transmits transmission signals from the transmission antenna 106 at predetermined transmission intervals using, for example, a first center frequency and a second center frequency that satisfy any one of equations (10) to (15). This allows the radar device 10a to determine the number of aliases in the Doppler determination unit 213 based on the Doppler frequency shift detected in the Doppler analysis unit 210 and the Doppler demultiplexing unit 212, which corresponds to differences in center frequency. Therefore, the radar device 10a can expand the Doppler frequency range (or the maximum value of relative velocity) in which a Doppler multiplexed signal can be separated, for example, depending on the number of aliases that can be determined.

In addition, in this embodiment, the first chirp signal and the second chirp signal are simultaneously transmitted every transmission period T _r , which enables multiplexing of Doppler multiplexed signals within a Doppler frequency range of ±½T _r . For example, compared to the case of Doppler multiplexed transmission at T _rs =2T _r as in embodiment 1, in this embodiment, Doppler multiplexed signals can be multiplexed within a Doppler frequency range (or maximum value of relative velocity) that is twice as wide.

Furthermore, for example, by making the number of transmitting antennas 106 included in the first and second transmitting sub-blocks the same or approximately the same, the Doppler multiplexing interval when Doppler multiplexing using the first chirp signal and the second chirp signal can be expanded by approximately four times compared to embodiment 1.

Note that if the Doppler multiplexing interval during Doppler multiplexing is close, interference between Doppler multiplexed signals is more likely to occur in the case of targets with spread Doppler components, resulting in deterioration of direction estimation accuracy and target detection accuracy. In this embodiment, the Doppler multiplexing interval can be extended compared to embodiment 1, thereby reducing the occurrence of such interference between Doppler multiplexed signals and suppressing deterioration of direction estimation accuracy and target detection accuracy. Furthermore, in this embodiment, the Doppler multiplexing interval can be extended compared to embodiment 1, thereby reducing the occurrence of interference between Doppler multiplexed signals and suppressing deterioration of direction estimation accuracy and target detection accuracy even when more transmit antennas are used for Doppler multiplexing. Therefore, in this embodiment, Doppler multiplexing can be performed using more transmit antennas 106 than in embodiment 1.

As described above, according to this embodiment, the Doppler frequency range (or maximum value of relative velocity) in which no ambiguity occurs can be expanded. This allows the radar device 10a to accurately detect targets (e.g., direction of arrival) over a wider Doppler frequency range.

Furthermore, in this embodiment, the Doppler frequency range in which a Doppler multiplexed signal can be separated is expanded by setting the center frequency of the chirp signal, so it is possible to omit the application of a method such as increasing the sampling rate of the A/D converter. Therefore, this embodiment can prevent the hardware configuration of the radar device 10a from becoming complicated and also prevent an increase in power consumption or heat generation in the radar device 10a. Furthermore, in this embodiment, the Doppler frequency range in which a Doppler multiplexed signal can be separated is expanded by setting the center frequency of the chirp signal, so it is possible to omit the application of a method such as shortening the transmission period T _r . Therefore, this embodiment can prevent a reduction in the detectable distance range of the radar device 10a or a deterioration in distance resolution.

In this embodiment, the radar device 10a includes two radar transmission signal generators 101, and transmits a transmission signal from the transmitting antenna 106 at predetermined transmission intervals using, for example, a first center frequency and a second center frequency that satisfy any one of equations (10) to (15). However, this is not limiting. The radar device 10a may also be configured to include, for example, more radar transmission signal generators 101 (three or more).

For example, the radar device 10a may include three radar transmission signal generators 101, and transmit a transmission signal from the transmitting antenna 106 at a predetermined transmission period using a first center frequency fc(1) and a second center frequency fc(2) that satisfy any one of equations (10) to (15). The second center frequency fc(2) and a third center frequency fc(3) that also satisfy any one of equations (10) to (15). Note that fc(1) > fc(2) > fc(3) or fc(1) < fc(2) < fc(3) may be satisfied. In this case, the radar receiver 200a may have three reception sub-blocks, and outputs from the three radar transmission signal generators 101 may be input to the mixer unit 204 of each reception sub-block. Furthermore, the radar receiver 200a may have a CFAR unit 211 and a Doppler demultiplexing unit 212 for each reception sub-block, each of which may perform the same operation as in this embodiment. This allows multiplexed received signals for the three chirp signals to be obtained, and the Doppler determination unit 213 can perform Doppler determination based on these output signals, thereby expanding the Doppler frequency detection range.

Furthermore, for example, the radar device 10a may be configured to include three or more radar transmission signal generators 101, and the number of transmission antennas included in each transmission subblock may be the same or approximately the same. This has the effect of further expanding the Doppler multiplexing interval when Doppler multiplexing using multiple chirp signals. Therefore, since the Doppler multiplexing interval can be expanded when Doppler multiplexing, interference between Doppler multiplexed signals can be reduced, and deterioration of direction estimation accuracy and target detection accuracy due to interference can be suppressed. Furthermore, since the Doppler multiplexing interval can be expanded when Doppler multiplexing, even if more transmission antennas 106 are used for Doppler multiplexing transmission compared to embodiment 1, interference between Doppler multiplexed signals can be reduced, and deterioration of direction estimation accuracy and target detection accuracy can be suppressed.

In addition, in the present embodiment, the case where the modulation parameters for the first chirp signal and the second chirp signal have other parameters other than the center frequency in common has been described, but this is not limiting. To apply an embodiment of the present disclosure, it is sufficient that the distance resolutions are the same and that the frequency sweep bandwidths B _w (q) are the same.

For example, as shown in Fig. 15, the first chirp signal and the second chirp signal may be set by modulation parameters that satisfy _Bw (1) = _Bw (2), _Tsw (1) ≠ _Tsw (2), and _Dm (1) ≠ _Dm (2). In this case, although the frequency sweep times _Tsw of the first chirp signal and the second chirp signal are different, the frequency sweep bandwidths _Bw are the same and the range resolutions ΔR (= _C0 / _2Bw ) match, so the radar device 10a can obtain similar effects by performing the operation according to the embodiment of the present disclosure described above.

15 , when the beat signals output from each radio reception unit 203 are discretely sampled in the A/D conversion unit 207 of each signal processing unit 206 by setting T _sw (1)≠T _sw (2), the number of discrete sampling data obtained in a predetermined time range (range gate) T _sw (1)≠T _sw (2) differs. Therefore, the beat frequency analysis unit 208 may perform the following operation instead of performing FFT processing on N _data pieces of discrete sampling data obtained in a predetermined time range (range gate) T _sw for each transmission period _Tr .

For example, the beat frequency analysis unit 208 in the first receiving sub-block may perform FFT processing on N _data (1) discrete sampled data obtained in a predetermined time range (range gate) T _sw (1) during the period in which the first chirp signal is transmitted. The beat frequency analysis unit 208 in the second receiving sub-block may perform FFT processing on N _data (2) discrete sampled data obtained in a predetermined time range (range gate) T _sw (2) during the period in which the second chirp signal is transmitted. The beat frequency analysis unit 208 may then use the smaller of N _data (1) and N _data (2) as N _data for subsequent processing (processing in the CFAR unit 211, Doppler demultiplexing unit 212, Doppler determination unit 213, and direction estimation unit 214).

The radar device 10a of this embodiment also includes multiple radar transmission signal generators 101, and transmits transmission signals from the transmission antenna 106 at predetermined transmission intervals using, for example, a first center frequency and a second center frequency that satisfy any one of equations (10) to (15). This allows the radar device 10a to determine the number of aliasing events based on the deviation in Doppler frequency in Doppler analysis that corresponds to differences in center frequency. Therefore, the radar device 10a can expand the Doppler frequency range (or maximum relative velocity) in which Doppler multiplexed signals can be separated, for example, depending on the number of aliasing events that can be determined. Furthermore, compared to embodiment 1, the Doppler frequency range (or maximum relative velocity) in which Doppler multiplexed signals can be separated can be doubled.

In addition, in this embodiment, the Doppler detection range in the Doppler analysis unit 210 is twice as wide as in embodiment 1. Furthermore, in this embodiment, the number of transmit antennas N1 or N2 used for Doppler multiplexing is smaller than the number of transmit antennas Nt used for Doppler multiplexing in embodiment 1. Therefore, in this embodiment, the Doppler multiplexing interval can be increased at least twice as wide as in embodiment 1. For example, if the Doppler multiplexing interval is close, interference between Doppler multiplexed signals is more likely to occur in the case of targets with spread Doppler components. In contrast, in this embodiment, multiplexing can be performed using a larger number of transmit antennas 106.

(Variation 1 of Embodiment 3)
In this embodiment, the operation of the Doppler shift units 105-1 to 105-N1 and the Doppler shift units 105-N1+1 to 105-Nt included in the first and second transmission sub-blocks has been described in which the intervals (Doppler shift intervals) between the Doppler shift amounts DOP _nsub1 and DOP _nsub2 for the first and second chirp signals are not set to equal intervals, but rather to unequal intervals with at least one Doppler interval being different.

However, this is not limited to this, and at least one of the Doppler intervals set by the Doppler shift unit 105 in the first or second transmission sub-block may be set to different unequal intervals rather than to equal intervals, and the other may be set to equal intervals.

As an example, the Doppler shift units 105-1 to 105-N1 included in the first transmission sub-block impart a phase rotation φ _nsub1 (m) as shown in equation (62) to the input m-th first chirp signal, which results in a Doppler shift amount DOP _nsub1 that differs between the Doppler shift units, and perform uneven Doppler multiplexing with a Doppler multiplexing number of N1. In this case, the Doppler shift units 105-N1+1 to 105-Nt included in the second transmission sub-block may impart a phase rotation φ nsub2 (m) as shown in the following equation (70) to the input m-th second chirp _signal , which results in a Doppler shift amount DOP _nsub2 that differs between the Doppler shift units 105, and perform even Doppler multiplexing with a Doppler multiplexing number of N2.

Alternatively, as another example, the Doppler shift units 105-1 to 105-N1 included in the first transmission sub-block may apply a phase rotation φ _nsub1 (m) as shown in the following equation (71) to the input m-th first chirp signal, resulting in a Doppler shift amount DOP _nsub1 that differs between the Doppler shift units 105, thereby performing equidistant Doppler multiplexing with a Doppler multiplexing number N1. In this case, the Doppler shift units 105-N1+1 to 105-Nt included in the second transmission sub-block may apply a phase rotation φ _nsub2 (m) as shown in equation (63) to the input m-th second chirp signal, resulting in a Doppler shift amount DOP _nsub2 that differs between the Doppler shift units 105, thereby performing unequal Doppler multiplexing with a Doppler multiplexing number N2.

Here, in equations (70) and (71), A is a coefficient that gives a positive or negative polarity of 1 or -1. The terms round(N _C /N1) and round(N _C /N2) are introduced for the purpose of making the amount of phase rotation an integer multiple of the Doppler frequency interval in the Doppler analysis unit 210. However, this is not limiting. 2π/N1 may be used instead of the term (2π/N _C )×round(N _C /N1) in equation (71). Similarly, 2π/N2 may be used instead of the term (2π/N _C )×round(N _C /N2) in equation ₍ 70). φ _{01 and} φ ₀₂ are initial phases, which may be equal or different. Whether φ 01 and φ 02 are equal or different, the Doppler frequencies will match. Δφ ₀₁ and Δφ ₀₂ are reference Doppler shift phases, which may be equal or different. Whether Δφ ₀₁ and Δφ ₀₂ are equal or different, the Doppler frequencies match.

As an example when the number of transmitting antennas Nt=4, in equation (62), N1=2, Δφ ₀₁ =0, φ ₀₁ =0, A=1, δ1=1, and N _C is a multiple of 6, and a phase rotation φ _nsub1 (m)=2π(nsub1-1)×(m-1)/3 is applied to the first chirp signal for each transmission period T _r , so the Doppler shift amounts are DOP ₁ =φ ₁ (m)/{2π(m-1)T _r }=0, DOP ₂ =φ ₂ (m)/{2π(m-1)T _r }=1/(3T _r ). The upper part of (a) of Figure 16 shows an example of the arrangement of Doppler multiplexed signals when transmitting the first chirp signal.

As an example when the number of transmitting antennas is Nt=4, in equation (70), Nt=4, N2=2, Δφ ₀₂ =0, φ ₀₂ =0, A=1, and N _C is a multiple of 6, and a phase rotation φ _nsub2 (m)=2π(nsub2-1)×(m-1)/2 is applied to the second chirp signal for each transmission period T _r , so the Doppler shift amounts are DOP ₁ =0 and DOP ₂ =1/T _r . The lower part of (a) of Figure 16 shows an example of the arrangement of Doppler-multiplexed signals when transmitting the second chirp signal.

Also, as an example when the number of transmitting antennas Nt=5, if in equation (71) N1=3, Δφ ₀₁ =0, φ ₀₁ =0, A=1, and N _C is a multiple of 3, a phase rotation φ _nsub1 (m)=2π(nsub1-1)×(m-1)/3 is applied to the first chirp signal for each transmission period T _r , and the Doppler shift amounts are DOP ₁ =φ ₁ (m)/{2π(m-1)T _r }=0, DOP ₂ =φ ₂ (m)/{2π(m-1)T _r }=1/(3T _r ), and DOP ₃ =φ ₃ (m)/{2π(m-1)T _r }=2/(3T _r )=-1/(3T _r ). The upper part of FIG. 16(b) shows an example of the arrangement of Doppler multiplexed signals when transmitting the first chirp signal.

As an example of the case where the number of transmitting antennas is Nt=5, if in equation (63) Nt=5, N2=2, Δφ ₀₂ =0, φ ₀₂ =0, A=1, δ2=1, and N _C is a multiple of 3, a phase rotation φ _nsub2 (m)=π×(m-1)/2 is imparted to the second chirp signal for each transmission period T _r , and the Doppler shift amounts are DOP ₁ =0, DOP ₂ =1/(3T _r ). The lower part of (b) of Figure 16 shows an example of the arrangement of Doppler-multiplexed signals when transmitting the second chirp signal.

In this way, when the Doppler intervals between Doppler multiplexed signals set in the Doppler shift unit 105 in the first or second transmission sub-block are equal, it becomes difficult for the Doppler demultiplexing unit 212 in the radar receiving unit 200a to estimate the Doppler frequency in the Doppler frequency range of ±1/ _Tr , and therefore it becomes difficult to output separation index information for the Doppler multiplexed signals. The radar receiving unit 200a performs separation processing, for example, using the output of the Doppler demultiplexing unit 212 for received signals corresponding to signals of transmission sub-blocks in which the Doppler intervals between Doppler multiplexed signals set in the Doppler shift unit 105 in the first or second transmission sub-block are unequal.

Figure 17 shows an example configuration of the radar receiver 200a in the radar device 10a when the Doppler shifter 105 in the first transmission sub-block performs unequal-interval Doppler multiplexing and the Doppler shifter 105 in the second transmission sub-block performs equal-interval Doppler multiplexing.

The first receiving sub-block performs reception processing of the reflected waves from the first transmitting sub-block. The radar receiving unit 200a shown in FIG. 17 differs from the radar receiving unit 200a shown in FIG. 11 in that, for example, the output from the first Doppler demultiplexing unit 212 is input to the second Doppler demultiplexing unit 212. Below, we will explain an example of the operation of the Doppler demultiplexing unit 212 in the radar receiving unit 200a shown in FIG. 17, which differs from the radar receiving unit 200a shown in FIG. 11.

The separation operation of the Doppler multiplexed signal in the first Doppler multiplexed separation unit 212 is similar to the operation described above (operation in Figure 11), but the first Doppler multiplexed separation unit 212 outputs distance index f _{b_cfar} (1) information and separation index information (f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) of the N1 Doppler multiplexed signals multiplexed and transmitted in the first transmission sub-block at distance index f _{b_cfar} (1) to the second Doppler multiplexed separation unit 212.

The second Doppler demultiplexing unit 212 calculates a Doppler frequency estimate f _{d_VFT} (1) assuming that the Doppler frequency of the target is within the Doppler frequency range −1/(2T _r ) ≦ f d_TargetDoppler < 1/(2T _r ) based on, for example, separation index information (f _{demul_Tx#1} , f _{demul_Tx#2} , to , f _{demul_Tx #Nt} ) of the Doppler multiplexed signal in the distance index f _{b_cfar} (1) output from the first Doppler demultiplexing unit 212. For example, the second Doppler demultiplexing unit 212 may calculate a Doppler frequency estimate f _{d_VFT} (1) from which a component of a known predetermined Doppler shift amount assigned to each transmitting antenna 106 in the radar transmitter 100 a has been removed.

Then, based on the calculated Doppler frequency estimate value _{fd_VFT} (1), the second Doppler demultiplexing unit 212 demultiplexes the Doppler multiplexed signal using peaks (distance index _{fb_cfar} (2) and Doppler frequency index _{fs_cfar} (2)) having reception power greater than the threshold input from the second CFAR unit 211. For example, the second Doppler demultiplexing unit 212 uses the _Doppler frequency estimate value _{fd_VFT} (1) to determine which of the transmission signals transmitted from transmitting _antennas Tx#N1+1 to Tx#Nt the reflected wave signal corresponds to for multiple Doppler frequency indexes _{fs_cfar} (2) ∈ {fd _#N1+1 , fd _#N1+2 ..., fd _#Nt } for which distance index fb_cfar(1)=fb_cfar(2).

For example, the Doppler shift amount that the Doppler shifter 105 in the second transmission sub-block applies to each of the second chirp signals output from the transmitting antennas Tx#N1+1 to Tx#Nt is known, and therefore the second Doppler demultiplexer 212 can calculate the received Doppler frequencies for the transmitting antennas Tx#N1+1 to Tx#Nt assuming that the Doppler frequency of the target is the Doppler frequency estimated value f _{d_VFT} (1).

In this case, the reception Doppler frequencies for the transmitting antennas Tx#N1+1 to Tx#Nt are represented as {fdRef _#N1+1 , fdRef _#N1+2 , . . . fdRef _#Nt }.

The second Doppler demultiplexing unit 212 can generate such signals for each of the transmitting antennas 106. For example, the second Doppler demultiplexing unit 212 determines, for these Doppler frequencies, the Doppler frequency for which the difference between each Doppler frequency index f _{s_cfar} (2) from the second CFAR unit 211 is the smallest and which is smaller than ±ΔDOP _min2 /2, as the reception Doppler frequency for the transmitting antennas Tx#N1+1 to Tx#Nt.

Then, the second Doppler demultiplexing unit 212 separates and outputs the reflected wave signals for each of the determined transmitting antennas Tx#N+1 to Tx#Nt. For example, the second Doppler demultiplexing unit 212 outputs to the Doppler determination unit 213 distance index f _{b_cfar} (2) information, separation index information (f _{demul_Tx #N1+1} , f demul_Tx _#N1+ 2 , to , f _{demul_Tx#Nt) of the N2 Doppler multiplexed signals multiplexed and transmitted in the first transmission sub-block for distance index f b_cfar (2} ), and the output of the Doppler analysis unit 210.

The premise for enabling such operation of the second Doppler demultiplexing unit 212 is that the difference in the Doppler frequency of the target resulting from the difference between the center frequencies of the first chirp signal and the second chirp signal is within ±ΔDOP _min2 /2, and the relative velocity of the target is assumed to be within this range.

Alternatively, in the radar device 10a, the Doppler shift unit 105 in the second transmission sub-block may perform uneven Doppler multiplexing, and the Doppler shift unit 105 in the first transmission sub-block may perform equal Doppler multiplexing. In this case, as in the example of Figure 17, in the radar receiver 200a, the output from the second Doppler demultiplexing unit 212 is input to the first Doppler demultiplexing unit 212, enabling Doppler demultiplexing in the first Doppler demultiplexing unit 212.

In this way, the Doppler shift unit 105 in at least one of the first or second transmission sub-blocks may be set to perform unequal-interval Doppler multiplexing, and the Doppler shift unit 105 in the remaining transmission sub-blocks may be set to perform equal-interval Doppler multiplexing. This allows the radar device 10a to achieve the effect of further increasing the Doppler multiplexing interval when Doppler multiplexing using the first chirp signal and the second chirp signal. By increasing the Doppler multiplexing interval when Doppler multiplexing, interference between Doppler-multiplexed signals can be reduced, and deterioration in direction estimation accuracy and target detection accuracy can be suppressed. Note that the Doppler shift unit 105 in subsequent embodiments can similarly be set to both unequal-interval Doppler multiplexing and equal-interval Doppler multiplexing, achieving similar effects.

(Modification 2 of Embodiment 3)
11 may be realized by combining a plurality of transmitting and receiving chips, as shown in Fig. 18. In the example of Fig. 18, the radar device 10a is configured from transmitting and receiving chip #1 and transmitting and receiving chip #2.

Transmit/receive chip #q includes a radar transmission signal generation unit 101-q, a qth transmission sub-block, a qth reception sub-block, a qth CFAR unit 211 (or a CFAR unit 211-q), and a qth Doppler demultiplexing unit 212 (or a Doppler demultiplexing unit 212-q). Here, q = 1 or 2.

Note that at least one of the Doppler determination unit 213 and the direction estimation unit 214 may be implemented on a separate signal processing chip or ECU, or may be incorporated into either the transmission/reception chip.

The difference between Figure 18 and Figure 11 is that it includes a synchronization signal generation unit 215, and the synchronization signal (e.g., a reference signal) output from the synchronization signal generation unit 215 is output to the radar transmission signal generation unit 101 of the radar transmitter 100a of each transmit/receive chip. This makes it possible to output chirp signals so that the frequency difference between the first chirp signal output from the transmit sub-block of transmit/receive chip #1 and the second chirp signal output from the transmit sub-block of transmit/receive chip #2 is within a predetermined allowable error. The configuration shown in Figure 18 also achieves the same effects as in embodiment 3, and by combining general-purpose transmit/receive chips, costs can be reduced.

(Fourth embodiment)
In the third embodiment, a MIMO antenna configuration with Nt transmitting antennas and Na receiving antennas has been described, and the direction estimation unit 214 performs direction estimation processing using N1×N3 MIMO virtual receiving antennas and N2×N4 MIMO virtual receiving antennas. Here, N1+N2=Nt and N3+N4=Na. In this case, the number of antennas available in the direction estimation unit 214 is less than Nt×Na.

In this embodiment, when radar transmission signals of different frequencies are transmitted simultaneously every transmission period, as in embodiment 3, a method is described in which the number of MIMO virtual receiving antennas available in the direction estimation unit 214 is set to Nt × Na, as in embodiment 1.

For example, Fig. 19 shows a configuration example in which a radar transmitter 100b of a radar device 10b includes two radar transmission signal generators 101. As shown in Fig. 19, the radar device 10b, like the configuration of the radar device 10a in the third embodiment shown in Fig. 11, simultaneously transmits radar transmission signals (e.g., chirp signals) with different center frequencies generated in the multiple radar transmission signal generators 101 from multiple transmission antennas 106 every transmission period T _r .

On the other hand, the radar receiver 200b shown in Figure 19 differs from Figure 11 in that it includes a switching unit 216 that switches the output destination of the multiple radar transmission signal generators 101 to one of the mixer units 204 in the two receiving sub-blocks. The radar receiver 200b shown in Figure 19 also includes an output switching unit 217 that, in conjunction with the switching operation of the switching unit 216, switches the output destination of the beat frequency analysis unit 208 to one of the two Doppler analysis units 210 for output.

The following describes the operation of this embodiment, focusing mainly on examples of operation that differ from embodiment 3.

As an example, Figure 19 shows a configuration in which the radar transmitter 100b of the radar device 10b includes two radar transmission signal generators 101. Below, the two radar transmission signal generators 101 are referred to as the "first radar transmission signal generator 101 (or radar transmission signal generator 101-1)" and the "second radar transmission signal generator 101 (or radar transmission signal generator 101-2)."

In FIG. 19, the configuration of each radar transmission signal generation unit 101 may be the same as in embodiment 1. Each radar transmission signal generation unit 101 generates a radar transmission signal, for example, under control of the signal generation control unit 104.

The signal generation control unit 104 controls the generation of radar transmission signals by the first and second radar transmission signal generation units 101 (e.g., the modulation signal generation unit 102 and the VCO 103). For example, the signal generation control unit 104 may set parameters (e.g., modulation parameters) related to the chirp signals so that the first and second radar transmission signal generation units 101 each transmit chirp signals with different center frequencies. Hereinafter, the chirp signal generated by the first radar transmission signal generation unit 101 will be referred to as the "first chirp signal," and the chirp signal generated by the second radar transmission signal generation unit 101 will be referred to as the "second chirp signal."

As in the first embodiment, the signal generation control unit 104 may set (or select) a center frequency f _c (q) that satisfies a predetermined condition. In the following, as an example, a case will be described in which, among the modulation parameters set for the first chirp signal and the second chirp signal, the center frequencies f _c (q) are different from each other, but the other modulation parameters are the same (or common). However, this is not limiting, and application of an embodiment of the present disclosure is not limited to this. For example, it is sufficient that the resolution of the distance axis of the first chirp signal and the second chirp signal is the same, and therefore it is sufficient to set chirp signals with the same frequency sweep bandwidth B _w (q). Here, q = 1 or 2.

Furthermore, the signal generation control unit 104 may control the modulation signal generating unit 102 and the VCO 103 to simultaneously transmit (or output) two chirp signals with different center frequencies f _c (q) 2Nc times each, as shown in FIG. 12 , for example, in the same manner as in the third embodiment.

In one embodiment of the present disclosure, the transmission period T _r may be set to, for example, several hundred μs or less, and the transmission time interval of the radar transmission signal may be set to a relatively short time. As a result, even if the center frequencies of the first chirp signal and the second chirp signal are different, the frequency of the beat signal of the received reflected wave (e.g., the beat frequency index) does not change, and the radar device 10b can detect this as a change in Doppler frequency.

The first chirp signal output from the first radar transmission signal generation unit 101 (e.g., VCO 103) is input to, for example, N1 of the Nt Doppler shift units 105 (e.g., represented as Doppler shift units 105-1 to 105-N1). The first chirp signal output from the first radar transmission signal generation unit 101 is input to the first switching unit 216 (also referred to as switching unit 216-1) and the second switching unit 216 (also referred to as switching unit 216-2).

On the other hand, the second chirp signal output from the second radar transmission signal generation unit 101 (e.g., VCO 103) is input to, for example, N2 Doppler shift units 105 (e.g., Doppler shift units 105-N1+1 to 105-Nt) out of the Nt Doppler shift units 105. Furthermore, the second chirp signal output from the second radar transmission signal generation unit 101 is input to the first and second switching units 216.

Here, let N1 + N2 = Nt.

The output signals of the N1 Doppler shifters 105 to which the first chirp signal is input are amplified to a predetermined transmission power and radiated into space from each transmitting antenna 106 (e.g., Tx#1 to Tx#N1). The output signals of the N2 Doppler shifters 105 to which the second chirp signal is input are amplified to a predetermined transmission power and radiated into space from each transmitting antenna 106 (e.g., Tx#N1+1 to Tx#Nt). In this way, the first chirp signal and the second chirp signal are simultaneously transmitted every transmission period T _r .

The first and second switching units 216, for example, switch between the first chirp signal input from the first radar transmission signal generation unit 101 and the second chirp signal input from the second radar transmission signal generation unit 101 for each transmission period of the radar transmission signal, and output the signal to one of the mixer units 204 of the Na antenna system processing units 201 in the radar receiving unit 200b.

For example, in odd-numbered transmission periods, the first and second switching units 216 may input the first chirp signal to the mixer units 204 of N3 antenna system processing units 201 (e.g., antenna system processing units 201-1 to 201-N3) out of Na antenna system processing units 201 in the radar receiving unit 200b. Also, for example, in odd-numbered transmission periods, the first and second switching units 216 may input the second chirp signal to the mixer units 204 of N4 antenna system processing units 201 (e.g., antenna system processing units 201-N3+1 to 201-Na) out of Na antenna system processing units 201 in the radar receiving unit 200b.

Here, N3 + N4 = Na.

Furthermore, for example, in an even-numbered transmission period, the first and second switching units 216 may input the first chirp signal to the mixer units 204 of N4 antenna system processing units 201 (e.g., antenna system processing units 201-N3+1 to 201-Na) out of the Na antenna system processing units 201 in the radar receiving unit 200b. Furthermore, for example, in an even-numbered transmission period, the second chirp signal may be input to the mixer units 204 of N3 antenna system processing units 201 (e.g., antenna system processing units 201-1 to 201-N3) out of the Na antenna system processing units 201 in the radar receiving unit 200b.

Note that the output destinations of the first and second chirp signals (the switching destinations of the first and second switching units 216) in odd-numbered and even-numbered transmission cycles may be reversed. In this way, the output destinations of the first and second chirp signals (the receiving sub-blocks described below) may alternate for each transmission cycle.

Hereinafter, the N1 Doppler shift units 105 to which the first chirp signal is input and the respective transmit antennas 106 (e.g., Tx#1 to Tx#N1) that transmit the output signals of these N1 Doppler shift units 105 will be referred to as the "first transmit sub-block." Furthermore, the N2 Doppler shift units 105 to which the second chirp signal is input and the respective transmit antennas 106 (e.g., Tx#N1+1 to Tx#Nt) that transmit the output signals of these N2 Doppler shift units 105 will be referred to as the "second transmit sub-block." Here, N1 + N2 = Nt. Note that N1 and N2 are each 2 or greater, and Nt is 4 or greater.

For example, the Doppler shift unit 105 included in the first transmission sub-block applies a phase rotation φ _nsub1 to the first chirp signal to impart a Doppler shift amount DOP _nsub1 to the first chirp signal for each transmission period T _r of the chirp signal, and outputs the Doppler-shifted signal to a transmission antenna 106 (e.g., Tx#1 to Tx#N1). Here, nsub1 is an integer between 1 and N1. The Doppler shift unit 105 included in the second transmission sub-block applies a phase rotation φ _nsub2 to the second chirp signal to impart a Doppler shift amount DOP _nsub2 to the second chirp signal for each transmission period T _r of the chirp signal, and outputs the Doppler-shifted signal to a transmission antenna 106 (e.g., Tx#N1+1 to Tx#Nt). Here, nsub2 is an integer between 1 and N2.

Note that an example of a method for applying the Doppler shift amount DOP _nsub1 (or phase rotation φ _nsub1 ) and the Doppler shift amount DOP _nsub2 (or phase rotation φ _nsub2 ) in the Doppler shift unit 105 included in the first and second transmission sub-blocks differs from that of embodiment 3, and an example will be described later.

Also, if Nt is an even number, the number of transmit antennas 106 included in the first and second transmission subblocks may be made the same by setting N1 = N2, for example. Also, if Nt is an odd number, the number of transmit antennas 106 included in the first and second transmission subblocks may be made approximately the same, with a difference of one transmit antenna, by setting N1 = (Nt + 1)/2 or N1 = (Nt - 1)/2, for example. In this way, the number of transmit antennas 106 included in the first subblock (e.g., transmit antennas 106 transmitting the first chirp signal) and the number of transmit antennas 106 included in each second transmission subblock (e.g., transmit antennas 106 transmitting the second chirp signal) may be set to be the same or different by one. By setting the number of transmitting antennas 106 included in the first and second sub-blocks to be equal or approximately equal, the radar device 10b can achieve the effect of further increasing the Doppler multiplexing interval when performing Doppler multiplexing using the first chirp signal and the second chirp signal compared to embodiment 1 (for example, by approximately double).

For example, if the Doppler multiplexing interval during Doppler multiplexing transmission is close, interference between Doppler multiplexed signals is more likely to occur in targets with spread Doppler components, resulting in poor direction estimation accuracy and poor target detection accuracy. In this embodiment, the Doppler multiplexing interval during Doppler multiplexing can be extended, reducing the occurrence of such interference between Doppler multiplexed signals and suppressing deterioration in direction estimation accuracy and target detection accuracy.

In addition, in this embodiment, the Doppler multiplexing interval can be extended when performing Doppler multiplexing, so even if more transmitting antennas 106 are used for Doppler multiplexing, interference between Doppler multiplexed signals can be reduced, and degradation in direction estimation accuracy and target detection accuracy can be suppressed. Therefore, in this embodiment, more transmitting antennas 106 can be used for Doppler multiplexing compared to embodiment 1. In this way, when performing Doppler multiplexing using more transmitting antennas 106, this embodiment is more preferable than embodiment 1.

[Configuration example of radar receiver 200b]
19, the radar receiver 200b includes, for example, Na receiving antennas 202 (e.g., Rx#1 to Rx#Na) forming an array antenna. The radar receiver 200b also includes, for example, Na antenna system processors 201-1 to 201-Na, a CFAR unit 211, a Doppler demultiplexing unit 212, a Doppler determination unit 213, and a direction estimation unit 214.

Each receiving antenna 202 receives a reflected wave signal, which is a radar transmission signal (e.g., a first chirp signal and a second chirp signal) reflected from a target, and outputs the received reflected wave signal to the corresponding antenna system processing unit 201 as a received signal.

Each antenna system processing unit 201 has a receiving radio unit 203 and a signal processing unit 206.

The receiving radio unit 203 includes a mixer unit 204 and an LPF 205. In the receiving radio unit 203, the mixer unit 204 mixes the received reflected wave signal (received signal) with a chirp signal, which is a transmission signal.

Here, for example, in an odd-numbered transmission period, the first chirp signal output from the first or second switching unit 216 is input to the mixer unit 204 in the receiving radio unit 203 of each of the N3 antenna system processing units 201 out of the Na antenna system processing units 201. In the antenna system processing units 201-1 to 201-N3, by passing the output of the mixer unit 204 through the LPF 205, the output of the mixer unit 204 corresponding to the reflected wave of the second chirp signal becomes a high frequency outside the passband of the LPF 205, and the LPF 205 is therefore more likely to output a beat signal with a frequency corresponding to the delay time of the reflected wave signal of the first chirp signal.

Furthermore, for example, in an odd-numbered transmission period, the second chirp signal output from the first or second switching unit 216 is input to the mixer unit 204 in the receiving radio unit 203 of each of N4 of the Na antenna system processing units 201. In the antenna system processing units 201-N3+1 to 201-Na, by passing the output of the mixer unit 204 through the LPF 205, the output of the mixer unit 204 corresponding to the reflected wave of the first chirp signal becomes a high frequency outside the passband of the LPF 205, and the LPF 205 is therefore more likely to output a beat signal whose frequency corresponds to the delay time of the reflected wave signal of the second chirp signal.

On the other hand, for example, in an even-numbered transmission period, the second chirp signal output from the first or second switching unit 216 is input to the mixer unit 204 in the receiving radio unit 203 of each of the N3 antenna system processing units 201 out of the Na antenna system processing units 201. In the antenna system processing units 201-1 to 201-N3, by passing the output of the mixer unit 204 through the LPF 205, the output of the mixer unit 204 corresponding to the reflected wave of the first chirp signal becomes a high frequency outside the passband of the LPF 205, and the LPF 205 is therefore more likely to output a beat signal whose frequency corresponds to the delay time of the reflected wave signal of the second chirp signal.

Furthermore, for example, in an even-numbered transmission period, the first chirp signal output from the first or second switching unit 216 is input to the mixer unit 204 in the receiving radio unit 203 of each of N4 of the Na antenna system processing units 201. In the antenna system processing units 201-N3+1 to 201-Na, by passing the output of the mixer unit 204 through the LPF 205, the output of the mixer unit 204 corresponding to the reflected wave of the second chirp signal becomes a high frequency outside the passband of the LPF 205, and the LPF 205 is therefore more likely to output a beat signal with a frequency corresponding to the delay time of the reflected wave signal of the first chirp signal.

Therefore, antenna system processing units 201-1 to 201-N3 process, for example, the reflected wave signals of the first chirp signal received by receiving antennas 202-1 to 202-N3 in odd-numbered transmission periods, and process the reflected wave signals of the second chirp signal received by receiving antennas 202-1 to 202-N3 in even-numbered transmission periods. Hereinafter, the antenna system processing units 201 (e.g., receiving radio unit 203 and signal processing unit 206) that process the reflected waves of the first chirp signal in odd-numbered transmission periods and the reflected waves of the second chirp signal in even-numbered transmission periods, and the receiving antennas 202 connected to these antenna system processing units 201, will be referred to as the "first receiving sub-block."

Furthermore, antenna system processing units 201-N3+1 to 201-Na process, for example, reflected wave signals of the second chirp signal received by receiving antennas 202-N3+1 to 202-Na in odd-numbered transmission periods, and process reflected wave signals of the first chirp signal received by receiving antennas 202-N3+1 to 202-Na in even-numbered transmission periods. Hereinafter, antenna system processing units 201 (e.g., receiving radio unit 203 and signal processing unit 206) that process reflected waves of the second chirp signal in odd-numbered transmission periods and reflected waves of the first chirp signal in even-numbered transmission periods, and receiving antennas 202 connected to these antenna system processing units 201, are referred to as the "second receiving sub-block."

Here, N3 + N4 = Na. Note that N3 and N4 can each be 1 or greater, and Na can be 2 or greater.

For example, the first receiving sub-block (e.g., corresponding to the first receiving circuit) mixes the signal received by the receiving antenna 202 with the first chirp signal in odd-numbered transmission periods to output a reflected wave signal resulting from the first chirp signal reflected by the target, and the second receiving sub-block (e.g., corresponding to the second receiving circuit) mixes the signal received by the receiving antenna 202 with the second chirp signal in odd-numbered transmission periods to output a reflected wave signal resulting from the second chirp signal reflected by the target, and the second receiving sub-block (e.g., corresponding to the second receiving circuit) mixes the signal received by the receiving antenna 202 with the second chirp signal in odd-numbered transmission periods to output a reflected wave signal resulting from the second chirp signal reflected by the target, and the second receiving sub-block (e.g., corresponding to the second receiving circuit) mixes the signal received by the receiving antenna 202 with the first chirp signal in even-numbered transmission periods to output a reflected wave signal resulting from the first chirp signal reflected by the target.

Note that the chirp signals to be processed in each receiving sub-block in odd-numbered and even-numbered transmission cycles (e.g., the switching destination of the first and second switching units 216) may be reversed. In this way, the chirp signals to be processed in each receiving sub-block may alternate for each transmission cycle.

The signal processing unit 206 of each antenna system processing unit 201-z _q included in the q-th receiving sub-block has an A/D conversion unit 207, a beat frequency analysis unit 208, an output switching unit 217, and a Doppler analysis unit 210. Here, when q=1, z ₁ = any one of 1 to N3, and when q=2, z ₂ = any one of N3+1 to Na.

The signal (e.g., beat signal) output from the LPF 205 is converted into discrete sample data by the A/D conversion unit 207 in the signal processing unit 206.

The beat frequency analyzer 208 included in the first receiver sub-block performs FFT processing on N _data pieces of discrete sample data obtained within a predetermined time range (range gate) for each transmission period T _r. Here, the range gate sets the frequency sweep time T _sw (q). For example, q = 1 or 2. When q = 1, T _sw (1) represents the frequency sweep time of the first chirp signal, and when q = 2, T _sw (2) represents the frequency sweep time of the second chirp signal. The beat frequency analyzer 208 included in the first receiver sub-block may set q = 1 in odd-numbered transmission periods and q = 2 in even-numbered transmission periods. As a result, the signal processor 206 of the first receiver sub-block outputs a frequency spectrum in which peaks appear at beat frequencies corresponding to the delay times of the reflected wave signals (radar reflected waves) for the first chirp signal in odd-numbered transmission periods and the second chirp signal in even-numbered transmission periods.

The beat frequency analyzer 208 included in the second receiver sub-block performs FFT processing on N _data pieces of discrete sample data obtained within a predetermined time range (range gate) for each transmission period T _r. Here, the range gate sets the frequency sweep time T _sw (q). For example, q = 1 or 2. When q = 1, T _sw (1) represents the frequency sweep time of the first chirp signal, and when q = 2, T _sw (2) represents the frequency sweep time of the second chirp signal. The beat frequency analyzer 208 included in the second receiver sub-block may set q = 2 for odd-numbered transmission periods and q = 1 for even-numbered transmission periods. As a result, the signal processor 206 of the second receiver sub-block outputs a frequency spectrum in which peaks appear at beat frequencies corresponding to the delay times of the reflected wave signals (radar reflected waves) for the second chirp signal in odd-numbered transmission periods and the first chirp signal in even-numbered transmission periods.

Here, the beat frequency response output from the beat frequency analysis unit 208 in the _zqth signal processing unit 206 of the qth receiving sub-block, obtained by transmitting the mth chirp pulse, is represented by "RFT _zq (f _b , m)." Here, f _b represents the beat frequency index, which corresponds to the FFT index (bin number). For example, f _b = 0, to, N _data /2-1, z ₁ = 1 to N3, z ₂ = N3+1 to Na, m = 1 to 2N _C , and q = 1 or 2. The smaller the beat frequency index f _b , the smaller the delay time of the reflected wave signal (for example, the closer the distance to the target) indicating the beat frequency.

For example, when m is an odd number, RFT _z1 ( _fb ,m) represents a frequency spectrum in which a peak appears at a beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave) from the first chirp signal, and RFT _z2 ( _fb ,m) represents a frequency spectrum in which a peak appears at a beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave) from the second chirp signal.On the other hand, when m is an even number, RFT _z1 ( _fb ,m) represents a frequency spectrum in which a peak appears at a beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave) from the second chirp signal, and RFT _z2 ( _fb ,m) represents a frequency spectrum in which a peak appears at a beat frequency corresponding to the delay time of the reflected wave signal (radar reflected wave) from the first chirp signal.

The output switching unit 217 in the _z1st signal processing unit 206 of the first receiving sub-block performs a switching operation to output the beat frequency response RFT _z1 (f _b , m) input from the beat frequency analysis unit 208 to the first Doppler analysis unit 210 (or Doppler analysis unit 210-1) if m is odd, and to the second Doppler analysis unit 210 (or Doppler analysis unit 210-2) if m is even.

In addition, the output switching unit 217 in the _z2nd signal processing unit 206 of the second receiving sub-block performs a switching operation to output the beat frequency response RFT _z2 (f _b , m) input from the beat frequency analysis unit 208 to the second Doppler analysis unit 210 if m is an odd number, and to output it to the first Doppler analysis unit 210 if m is an even number.

As a result, one of the first Doppler analysis units 210 in each receiving sub-block processes the reflected wave signals for the first chirp signal received by receiving antennas 202-1 to 202-N3 in odd-numbered transmission periods, and processes the reflected wave signals for the first chirp signal received by receiving antennas 202-N3+1 to 202-Na in even-numbered transmission periods. Furthermore, one of the second Doppler analysis units 210 in each receiving sub-block processes the reflected wave signals for the second chirp signal received by receiving antennas 202-N3+1 to 202-Na in odd-numbered transmission periods, and processes the reflected wave signals for the second chirp signal received by receiving antennas 202-1 to 202-N3 in even-numbered transmission periods.

Furthermore, for example, the first receiving sub-block processes the reflected wave signals for the first chirp signal received by receiving antennas 202-1 to 202-N3 in odd-numbered transmission periods, and processes the reflected wave signals for the second chirp signal received by receiving antennas 202-1 to 202-N3 in even-numbered transmission periods.Furthermore, for example, the second receiving sub-block processes the reflected wave signals for the second chirp signal received by receiving antennas 202-N3+1 to 202-Na in odd-numbered transmission periods, and processes the reflected wave signals for the first chirp signal received by receiving antennas 202-N3+1 to 202-Na in even-numbered transmission periods.

The first Doppler analyzer 210 in the _z1st signal processor 206 of the first receiving subblock performs Doppler analysis for each range index f _b using beat frequency responses RFT _z1 (f _b ,1), RFT _z1 (f _b ,3), ... obtained by transmitting chirp pulses of the first chirp signal N _C times. For example, the first Doppler analyzer 210 may estimate the Doppler frequency from a reflected wave signal of the first chirp signal reflected by a target.

Furthermore, the second Doppler analyzer 210 in the _z1st signal processor 206 of the first receiving subblock performs Doppler analysis for each range index f _b using beat frequency responses RFT _z1 (f _b ,2), RFT _z1 (f _b ,4), ... obtained by transmitting chirp pulses of the second chirp signal N _C times. For example, the second Doppler analyzer 210 may estimate the Doppler frequency from a reflected wave signal of the second chirp signal reflected by a target.

Furthermore, the first Doppler analyzer 210 in the _z2nd signal processor 206 of the second receiving subblock performs Doppler analysis for each range index _fb using beat frequency responses RFT _z2 ( _fb ,2), RFT _z2 ( _fb ,4), ... obtained by transmitting chirp pulses of the first chirp signal N _C times. For example, the first Doppler analyzer 210 may estimate the Doppler frequency from a reflected wave signal of the first chirp signal reflected by a target.

Furthermore, the second Doppler analyzer 210 in the _z2nd signal processor 206 of the second receiving subblock performs Doppler analysis for each range index _fb using beat frequency responses RFT _z2 ( _fb ,1), RFT _z2 ( _fb ,3), ... obtained by transmitting chirp pulses of the second chirp signal N _C times. For example, the second Doppler analyzer 210 may estimate the Doppler frequency from a reflected wave signal of the second chirp signal reflected by a target.

For example, if _Nc is a power of 2, FFT processing can be applied in Doppler analysis. In this case, the FFT size is _Nc , and the maximum Doppler frequency at which aliasing does not occur, as derived from the sampling theorem, is ±1/( _4Tr ). Furthermore, the Doppler frequency interval of the Doppler frequency index _fs is 1/( _Nc × _2Tr ), and the range of the Doppler frequency index _fs is _fs = _-Nc /2, ∼, 0, ∼, _Nc /2-1.

In the following, an example will be described in which _Nc is a power of 2. If _Nc is not a power of 2, FFT processing can be performed with a data size of a power of 2 by, for example, including zero-padded data. Furthermore, the Doppler analyzer 210 may multiply the data by a window function coefficient such as a Han window or a Hamming window during FFT processing. Applying a window function can suppress side lobes that occur around the beat frequency peak.

For example, in the z1 _-th signal processing unit 206 of the first receiving sub-block, the output VFT _z1,1 ( _fb , _fs ) of the first Doppler analyzer 210 and the output VFT _z1,2 ( _fb , _fs ) of the second Doppler analyzer 210 are shown in the following equations (72) and (73), where j is the imaginary unit and _z1 = 1 to N3.

Furthermore, for example, in the _z2nd signal processing unit 206 of the second receiving sub-block, the output VFT _z2,1 ( _fb , _fs ) of the first Doppler analyzer 210 and the output VFT _z2,2 ( _fb , _fs ) of the second Doppler analyzer 210 are shown in the following equations (74) and (75), where j is the imaginary unit and _z2 = N3 + 1 to Na.

The above explains the processing performed by each component of the signal processing unit 206.

The CFAR unit 211 may, for example, include a first CFAR unit 211 (or CFAR unit 211-1) and a second CFAR unit 211 (or CFAR unit 211-2) corresponding to the first chirp signal and the second chirp signal, respectively, which have different center frequencies. Similarly, the Doppler demultiplexing unit 212 may, for example, include a first Doppler demultiplexing unit 212 (or Doppler demultiplexing unit 212-1) and a second Doppler demultiplexing unit 212 (or Doppler demultiplexing unit 212-2) corresponding to the first chirp signal and the second chirp signal, respectively, which have different center frequencies.

Note that while Figure 19 shows a configuration in which CFAR units 211 are provided in parallel (CFAR units 211-1 and 211-2), it is also possible to provide a single CFAR unit 211 and sequentially switch and process its inputs. Note that while Figure 19 shows a configuration in which Doppler demultiplexing units 212 are provided in parallel (Doppler demultiplexing units 212-1 and 212-2), it is also possible to provide a single Doppler demultiplexing unit 212 and sequentially switch and process its inputs.

In FIG. 19 , the CFAR unit 211 performs CFAR processing (e.g., adaptive threshold determination) using the output from the Doppler analysis unit 210 of the signal processing unit 206 included in the first and second reception sub-blocks, and extracts a distance index f _{b_cfar} and a Doppler frequency index f _{s_cfar} that give a local peak signal.

In FIG. 19, the CFAR unit 211 may include a first CFAR unit 211 (or CFAR unit 211-1) that performs CFAR processing using the output of the first Doppler analysis unit 210 of the signal processing unit 206 included in the first and second reception sub-blocks, and a second CFAR unit 211 (or CFAR unit 211-2) that performs CFAR processing using the output of the second Doppler analysis unit 210 of the signal processing unit 206 included in the first and second reception sub-blocks.

The first CFAR unit 211 performs CFAR processing (e.g., adaptive threshold determination) using the output from the first Doppler analysis unit 210 of the signal processing unit 206 included in the first and second reception sub-blocks, which is, for example, the result of estimating the Doppler frequency from the reflected wave signal of the first chirp signal reflected by the target, and extracts a distance index f _{b_cfar} and a Doppler frequency index f _{s_cfar} that give a local peak signal.

Furthermore, the second CFAR unit 211 performs CFAR processing (e.g., adaptive threshold determination) using the output from the second Doppler analysis unit 210 of the signal processing unit 206 included in the first and second reception sub-blocks, which is, for example, the result of estimating the Doppler frequency from the reflected wave signal of the second chirp signal reflected by the target, and extracts a distance index f _{b_cfar} and a Doppler frequency index f _{s_cfar} that give a local peak signal.

The qth CFAR unit 211 (q=1, 2) adds the power of the outputs of the qth Doppler analyzers 210 of the signal processors 206 included in the first and second reception sub-blocks, and performs two-dimensional CFAR processing consisting of a distance axis and a Doppler frequency axis (corresponding to relative velocity), or CFAR processing that combines one-dimensional CFAR processing, as shown in the following equation (76). The two-dimensional CFAR processing or the CFAR processing that combines one-dimensional CFAR processing may be, for example, the processing disclosed in Non-Patent Document 2. Here, _z1 = 1 to N3, and _z2 = N3 + 1 to Na.

The qth CFAR unit 211 adaptively sets a threshold value and outputs to the qth Doppler demultiplexing unit 212 a distance index f _{b_cfar} (q), a Doppler frequency index f _{s_cfar} (q), and received power information PowerFT(f _{b_cfar} (q), f _{s_cfar} (q)) that result in received power greater than the threshold value.

The Doppler demultiplexing unit 212 may include a first Doppler demultiplexing unit 212 (also referred to as Doppler demultiplexing unit 212-1) that performs Doppler demultiplexing processing using the output of the first CFAR unit 211 and the output of the first Doppler analysis unit 210 of the signal processing unit 206 included in each receiving sub-block, and a second Doppler demultiplexing unit 212 (also referred to as Doppler demultiplexing unit 212-2) that performs Doppler demultiplexing processing using the output of the second CFAR unit 211 and the output of the second Doppler analysis unit 210-2 of the signal processing unit 206 included in each receiving sub-block.

The q-th Doppler multiplexing separation unit 212 (q=1, 2) separates the transmission signals transmitted from each transmitting antenna ₁₀₆ (e.g., reflected wave signals corresponding to the transmission signals) from the Doppler _multiplexed signals (hereinafter referred to as "Doppler _{multiplexed signals") using the output from the q-th Doppler analysis unit 210 included in each receiving sub-block, based on information input from the q-th CFAR unit 211 (e.g., distance index f b_cfar (} q), Doppler frequency index f s_cfar (q), and reception power information PowerFT(f b_cfar (q), f s_cfar (q))).

The q-th Doppler demultiplexing unit 212 outputs, for example, information about the separated signals to the Doppler determination unit 213 and the direction estimation unit 214. The information about the separated signals may include, for example, a distance index f _{b_cfar} (q) corresponding to the separated signals and a Doppler frequency index (hereinafter sometimes referred to as separation index information). Here, the separation index information of the first Doppler demultiplexing unit 212 is the Doppler frequency index obtained by separating the signals transmitted from the transmitting antennas Tx#1, Tx#2, to, Tx#N1 included in the first transmission sub-block, and is represented as (f _{demul_Tx#1} , f _{demul_Tx#2} , to, f _{demul_Tx#N1} ) corresponding to each. Similarly, the separation index information of the second Doppler multiplex separation unit 212 is the Doppler frequency index separated from the signals transmitted from the transmitting antennas Tx#N1+1, Tx#N1+2, ..., Tx#Nt included in the second transmission sub-block, and is represented as (f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , ..., f _{demul_Tx#Nt} ) corresponding to each.

Furthermore, the qth Doppler demultiplexing unit 212 outputs the output from the qth Doppler analysis unit 210 to the direction estimation unit 214. Note that the qth Doppler demultiplexing unit 212 may output the output from the Doppler analysis unit 210 included in the qth reception sub-block to the direction estimation unit 214 based on information input from the qth CFAR unit _{211 (e.g., distance index f b_cfar (} q), Doppler frequency index f _{s_cfar} (q), and reception power information PowerFT(f _{b_cfar} (q), f s_cfar (q))).

Below, an example of the operation of the qth Doppler demultiplexing unit 212 will be explained, along with the operation of the Doppler shift unit 105 in the radar transmitter 100b.

[How to set the Doppler shift amount]
An example of a method for setting the amount of Doppler shift applied in the Doppler shifter 105 will be described.

In this embodiment, the input of the first chirp signal and the second chirp signal to the mixer unit 204 is switched between odd-numbered and even-numbered transmission cycles, and the same transmission phase change is imparted to these received signals. Therefore, this embodiment differs from embodiment 3 in that the Doppler shift unit 105 imparts the same phase rotation to the odd-numbered and even-numbered transmission cycles. For example, in odd-numbered transmission cycles, the Doppler shift unit 105 imparts a phase rotation to the chirp signal that sets the same Doppler shift amount as in embodiment 3, and in even-numbered transmission cycles, it imparts a phase rotation to the chirp signal that is the same as the phase rotation imparted to the immediately preceding odd-numbered transmission cycle.

For example, the following points are different in this embodiment from equations (62) and (63) described in embodiment 3.

The Doppler shifters 105-1 to 105-N1 of the first transmission sub-block apply a phase rotation φ nsub1 (2u-1) as shown in the following equation (77) to the input odd-numbered m=2u-1-th first chirp signal, which results in a Doppler shift amount DOP _nsub1 that differs between the Doppler shifters ₁₀₅ , and output the Doppler-shifted signal to the transmitting antennas 106 (for example, Tx#1 to Tx#N1), where nsub1=1 to N1 and u=1 to Nc.

In addition, the Doppler shift units 105-1 to 105-N1 of the first transmission sub-block apply a phase rotation φ _nsub1 (2u) as shown in the following equation (78) to the subsequently input even-numbered m=2u-th first chirp signal, which results in a Doppler shift amount DOP _nsub1 that differs between each Doppler shift unit 105, just like the odd-numbered ones.

As a result, different Doppler shift amounts are imparted to the transmission signals transmitted from the multiple transmission antennas 106 (e.g., Tx#1 to Tx#N1) included in the first transmission sub-block in each transmission period. Furthermore, the transmission signals transmitted from the transmission antennas 106 (e.g., Tx#1 to Tx#N1) included in the first transmission sub-block may be Doppler multiplexed with a Doppler multiplexing number N _DM =N1, for example.

Similarly, the Doppler shifters 105-N1+1 to 105-Nt of the second transmission sub-block apply a phase rotation φ _nsub2 (m) as shown in the following equation (79) to the input odd-numbered m=2u−1-th second chirp signal, which results in a Doppler shift amount DOP _nsub2 that differs between the Doppler shifters 105, and output the Doppler-shifted signal to the transmitting antennas 106 (for example, Tx# N1+1 to Tx#Nt), where nsub2=1 to N2 and u=1 to Nc.

In addition, the Doppler shift units 105-N1+1 to 105-Nt of the second transmission sub-block apply a phase rotation φ nsub2 (2u) as shown in the following equation (80) to the subsequently input even-numbered m=2u-th second chirp signal, which results in a Doppler shift amount DOP _nsub2 that differs between each Doppler shift unit ₁₀₅ , just like the odd-numbered ones.

As a result, different Doppler shift amounts are imparted to the transmission signals transmitted from the multiple transmission antennas 106 (e.g., Tx#N1+1 to Tx#Nt) included in the second transmission sub-block in each transmission period. Furthermore, the transmission signals transmitted from the transmission antennas 106 (e.g., Tx#N1+1 to Tx#Nt) included in the second transmission sub-block may be Doppler multiplexed with a Doppler multiplexing number N _DM =N2, for example.

Note that the present embodiment is not limited to equations (62) and (63), and the setting of the Doppler shift amount in the Doppler shift unit 105 described in embodiment 3 may also be applied.

For example, the radar device 10b may perform uneven Doppler multiplexing on the first chirp signal and the second chirp signal at Doppler multiplexing numbers N1 and N2, respectively. Alternatively, the radar device 10b may perform uneven Doppler multiplexing on at least one of the first chirp signal and the second chirp signal at Doppler multiplexing numbers N1 and N2, respectively.

[Example of operation of Doppler demultiplexing unit 212]
The first Doppler multiplex separation unit 212 separates and receives signals that have been transmitted such that the intervals between the Doppler shift amounts DOP _nsub1 (Doppler shift intervals) between the Doppler shift units 105-1 to 105-N1 included in the first transmission sub-block (for example, between the transmission antennas 106-1 to 106-N1) are not equal, but at least one Doppler interval is different.

The first Doppler demultiplexing unit 212 demultiplexes the Doppler-multiplexed signals at uneven intervals based on the outputs of the first Doppler analysis unit 210 and the first CFAR unit 211 in the first and second reception sub-blocks. The first Doppler demultiplexing unit 212 then outputs to the Doppler determination unit 213 the distance index f _{b_cfar} (1), separation index information (f _{demul_Tx #1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) of the N1 Doppler-multiplexed signals multiplexed and transmitted in the first transmission sub-block at distance index f b_cfar (1), and the output of the Doppler analysis unit 210.

In this embodiment, the input of the first chirp signal and the second chirp signal to the mixer unit 204 is switched between odd-numbered and even-numbered transmission periods, and the same transmission phase change is applied to these received signals. Therefore, the Doppler shift unit 105 applies the same phase rotation to the odd-numbered and even-numbered transmission periods. Accordingly, the operation of the first Doppler demultiplexing unit 212 is the same as that of the Doppler shifting unit 105 of the first embodiment, except that "Nt" is replaced with "N1" and " _Trs " is replaced with " _2Tr ." This allows the signals Doppler-multiplexed between the transmitting antennas 106-1 to 106-N1 to be separated. Therefore, a detailed description of the operation of the first Doppler demultiplexing unit 212 will be omitted.

Similarly, the second Doppler demultiplexing unit 212 separates and receives signals that have been transmitted with the Doppler shift amounts DOP _nsub2 not spaced at equal intervals (Doppler shift intervals) between the Doppler shift units 105-N1+1 to 105-Nt included in the second transmission sub-block (for example, between the transmission antennas 106-N1+1 to 106-Nt) but with at least one Doppler interval set to be different.

The second Doppler demultiplexing unit 212 demultiplexes the Doppler-multiplexed signals at uneven intervals based on the outputs of the second Doppler analysis unit 210 and the second CFAR unit 211 in the first and second reception sub-blocks. The second Doppler demultiplexing unit 212 then outputs to the Doppler determination unit 213 the distance index f _{b_cfar} (2), separation index information (f _{demul_Tx #N1+1} , f _{demul_Tx#N1+2} , to , f _{demul_Tx#Nt} ) of the N2 Doppler-multiplexed signals multiplexed and transmitted in the second transmission sub-block at the distance index f b_cfar (2), and the output of the Doppler analysis unit 210.

In this embodiment, the mixer unit 204 switches between inputting the first chirp signal and the second chirp signal in odd-numbered and even-numbered transmission periods, and the same transmission phase change is applied to these received signals. Therefore, the Doppler shifter 105 applies the same phase rotation in odd-numbered and even-numbered transmission periods. Accordingly, the second Doppler demultiplexer 212 operates in a manner similar to that described in the operation of the Doppler shifter 105 of the first embodiment, but with "Nt" replaced with "N2" and " _Trs " replaced with " _2Tr. " This enables the separation of signals Doppler-multiplexed between the transmitting antennas 106-N1+1 to 106-Nt. Therefore, a detailed description of the operation of the second Doppler demultiplexer 212 will be omitted.

[Example of operation of Doppler determination unit 213]
19 , the Doppler determination unit 213 determines the Doppler frequency corresponding to the Doppler peak based on the outputs of the first Doppler demultiplexing unit 212 and the second Doppler demultiplexing unit 212. For example, even when a target having a Doppler frequency _{fd_TargetDoppler} outside the Doppler frequency range −1/(2T _r )≦ _{fd_TargetDoppler} <1/(2T _r ) is included, the Doppler determination unit 213 can further expand the Doppler detection range by determining the Doppler frequency of the target.

For example, the Doppler determination unit 213 determines the Doppler frequency of a target including a Doppler frequency exceeding the Doppler frequency range -1/( _{2T r ) ≦ f d_TargetDoppler <} 1/(2T r ) using the separation index information (f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) of the Doppler multiplexed signal output from the first Doppler multiplex separation unit 212 and the separation index information (f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , ..., f _{demul_Tx#Nt} ) of the Doppler multiplexed signal output from the second _Doppler multiplex separation unit ₂₁₂ , which have the same distance indexes f b_cfar ( ₁ ) and f b_cfar (2).

The principle of determining the Doppler frequency of a target including a Doppler frequency that exceeds the Doppler frequency range −1/(2T _r )≦ f _{d_TargetDoppler} < 1/(2T _r ) in the Doppler determination unit 213 is, as in the first embodiment, that the center frequencies of the first chirp signal and the second chirp signal, which are radar transmission signals generated by the signal generation control unit 104 and the radar transmission signal generation unit 101, are different from each other.

In the first embodiment, the Doppler frequency is determined based on changes in the Doppler frequency in the Doppler multiplexed signal for the same transmitting antenna 106. In contrast, in the present embodiment, the Doppler determination unit 213 determines the Doppler frequency based on changes in the Doppler frequency in the Doppler multiplexed signal for a different transmitting antenna 106. When a different transmitting antenna 106 is used, the reception phase changes, but the received Doppler frequency does not change. Therefore, similar to the first embodiment, the Doppler determination unit 213 can determine the Doppler frequency of a target including a Doppler frequency that exceeds the Doppler frequency range -1/( _2Tr ) ≦ _{fd_TargetDoppler} < 1/( _2Tr ).

The operating principle of the Doppler frequency determination process and an example of the operation of the Doppler determination unit 213 differ from the operating principle of the Doppler frequency determination process and the example of the operation of the Doppler determination unit 213 described in embodiment 1 in the following points (1), (2), and (3).
(1) Replacing "T _rs " with "2T _r ",
(2) In embodiment 1, the Doppler determination unit 213 calculates the Doppler frequency estimate value f _{d_VFT} (1) based on the separation index information (f _{demul_Tx#1} (1), f _{demul_Tx#2} (1), ..., f _{demul_Tx#Nt} (1)) of the Doppler multiplexed signal in the distance index f b_cfar (1) output from the first Doppler _multiplex separation unit 212, assuming that the Doppler frequency of the target is within the Doppler frequency range -1/(2T _rs ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ). In contrast to this, in this embodiment, the Doppler determination unit 213 calculates a Doppler frequency estimate value f _{d_VFT} (1) assuming that the Doppler frequency of the target is within the Doppler frequency range −1/(4T r )≦f _{d_TargetDoppler} <1/(4T _r ) based on the separation index information (f _{demul_Tx#1} , f demul_Tx _#2 , . . . , f _{demul_Tx#N1 )} of the Doppler multiplexed signal in the distance index f _{b_cfar} (1) output from the first Doppler multiplex separation unit 212; and
(3) In embodiment 1, the Doppler determination unit 213 calculates the Doppler frequency estimate value f _{d_VFT} (2) based on the separation index information (f _{demul_Tx#1} (2), f _{demul_Tx#2} (2), ..., f _{demul_Tx#Nt} (2)) of the Doppler multiplexed signal in the distance index f b_cfar (2) output from the second Doppler _multiplex separation unit 212, assuming that the Doppler frequency of the target is within the Doppler frequency range -1/(2T _rs ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ). In contrast to this, in this embodiment, the Doppler determination unit 213 calculates the Doppler frequency estimate value f _{d_VFT} (2) based on the separation index information (f _{demul_Tx#N1+1} , f _{demul_Tx# N1+2} , ..., f _{demul_Tx#Nt} ) of the Doppler multiplexed signal in the distance index f b_cfar (2) output from the second Doppler multiplex separation unit 212, assuming that the Doppler frequency of the target is within the _Doppler frequency range -1/(4T _r ) ≦ f _{d_TargetDoppler} < 1/(4T _r ).

In this embodiment, the operation of the Doppler determination unit 213, except for the three differences mentioned above, is the same as in embodiment 1, so a description of that operation will be omitted.

Furthermore, similarly to the Doppler determination unit 213 in the first embodiment, by setting center frequencies f _c (1) and f _c (2) that satisfy the determination condition of any one of equations (10) to (15) in which " _{T rs} " is replaced with "2T r ", the Doppler determination unit 213 can determine the Doppler frequency of a target even when a target with a Doppler frequency outside the Doppler frequency range -1/(2T _r ) ≦ f _{d_TargetDoppler} < 1/(2T _r ) is included (for example, when Doppler aliasing occurs). Note that although equations (10) and (13) include T _rs , by rearranging them, equations that do not include T _rs , such as equations (11), (12), (14), and (15), can be obtained. Therefore, the center frequencies f _c (1) and f _c (2) that satisfy the determination condition have the same conditions as those in the first embodiment.

Incidentally, the equation (18) used in the explanation of the first embodiment can be expressed as the following equation (81) by replacing T _rs with 2T _r .

Equation (81) expresses the condition that, when the Doppler aliasing number is n _al , the difference between the aliasing component n _al ×{ _{f c (} 2)/f _c (1)}/ _2Tr of the Doppler frequency observed using the second chirp signal with a center frequency f c (2) and 2n _al / _Tr , the frequency interval of the aliasing number n _al in the first Doppler analysis unit 210, does not exceed ±1/( _4Tr ). For example, when n _al is positive, Δn _al is in the range of ±1/(4Tr) up to the maximum n _al that satisfies equation ( ₁₉ ), and the Doppler determination unit 213 can estimate the aliasing without ambiguity. Note that the maximum n _al that satisfies equation (19) is denoted as "n _almax ." For example, when n _al is negative, n _al = -n _almax similarly satisfies equation (19).

In the first embodiment, the first chirp signal or the second chirp signal is transmitted at a period of _Trs , and the Doppler frequency detection range is expanded, for example, by n _almax times compared to the Doppler frequency range when one transmitting antenna is used. On the other hand, in this embodiment, a signal corresponding to the first chirp signal or the second chirp signal is input to the first or second Doppler analyzer 210 at a period of _2Tr via the output switching unit 217, and frequency analysis processing is performed. Therefore, in this embodiment, the Doppler frequency detection range is the same as in the first embodiment, and is expanded, for example, by n _almax times compared to the Doppler frequency range when one transmitting antenna is used.

The above explains an example of the operation of the Doppler determination unit 213.

[Example of operation of direction estimation unit 214]
19 , the direction estimation unit 214 receives information input from the first Doppler multiplex separation unit 212 (for example, distance index f _{b_cfar} (1) and separation index information of the Doppler multiplexed signal at distance index f _{b_cfar} (1) (f _{demul_Tx#1} , f _{demul_Tx#2} , . . . f _{demul_Tx#N1} )), information input from the second Doppler multiplex separation unit 212 (for example, distance index f _{b_cfar} (2) and separation index information of the Doppler multiplexed signal at distance index f _{b_cfar} (2) (f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , . . . f _{demul_Tx#Nt} )), and Doppler frequency information f _dest =f _{d_VFT(1)} +n _alast /(2T _r ) or f _c (2)/f _c Based on (1)(f _{d_VFT(1)} +n _alast /(2T _r )), the output of the first Doppler analyzer 210 and the output of the second Doppler analyzer 210 are extracted and the direction of the target is estimated.

In this embodiment, N1×(N3+N4)=N1×Na MIMO virtual receive antennas are formed between the N1 transmit antennas 106 included in the first transmit subblock, the N3 receive antennas 202 included in the first receive subblock, and the N4 receive antennas 202 included in the second receive subblock. Similarly, N2×(N3+N4)=N2×Na MIMO virtual receive antennas are formed between the N2 transmit antennas 106 included in the second transmit subblock, and the N3 receive antennas 202 included in the first receive subblock and the N4 receive antennas 202 included in the second receive subblock. The direction estimation unit 214 may perform direction estimation processing using these two sets of virtual receive antennas, for example, Nt×Na MIMO virtual receive antennas.

For example, the direction estimation unit 214 extracts _{the output of the first Doppler analysis unit 210 from the output of the first Doppler multiplex separation unit 212 based on the distance index f b_cfar (} 1) and the separation index information (f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) of the Doppler multiplex signal at the distance index f b_cfar (1), and generates a first virtual receiving array correlation vector h ₁ (f _{b_cfar} (1), f _{demul_Tx#1} , f demul_Tx _#2 , ..., f _{demul_Tx#N1} ) consisting of N1 × (N3 + N4) elements as shown in the following equation (82).

Furthermore, the direction estimation unit 214 extracts _{the output of the second Doppler analysis unit 210 from the output of the second Doppler multiplex separation unit 212, for example, based on the distance index f b_cfar (} 2) and the separation index information (f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , to, f _{demul_Tx#Nt} ) of the Doppler multiplexed signal at the distance index f b_cfar (2), and generates a second virtual receiving array correlation vector h ₂ (f _{b_cfar} (2), f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , to, f _{demul_Tx#N1 ) consisting of N2 × (N3 + N4} ) elements as shown in the following equation (83).

Since the direction estimation unit 214 performs direction estimation processing using the outputs of the first and second Doppler demultiplexing units 212 with the same distance index, in equations (82) and (83), f _{b_cfar} (1) = f _{b_cfar} (2) = f _{b_cfar} .

In equation (82), h ₁₁ (f _{b_cfar} , f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) represents a column vector consisting of N1 × N3 elements, as shown in the following equation (84), and is a vector whose elements are the outputs of the first Doppler analysis unit 210 obtained by receiving signals transmitted from the N1 transmitting antennas 106 included in the first transmitting sub-block by the N3 receiving antennas 202 included in the first receiving sub-block.

Also, in equation (82), h ₂₁ (f _{b_cfar} , f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#N1} ) represents a column vector consisting of N1 × N4 elements, as shown in the following equation (85), and is a vector whose elements are the outputs of the second Doppler analysis unit 210 obtained by receiving signals transmitted from the N1 transmitting antennas 106 included in the first transmitting sub-block by the N4 (= Na - N3) receiving antennas 202 included in the second receiving sub-block.

Furthermore, in equation (83), h ₁₂ (f _{b_cfar} , f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , to, f _{demul_Tx#Nt} ) represents a column vector consisting of N2 × N3 elements, as shown in the following equation (86), and is a vector whose elements are the outputs of the second Doppler analysis unit 210 obtained by receiving signals transmitted from the N2 (= Nt - N1) transmitting antennas 106 included in the second transmitting sub-block by the N3 receiving antennas 202 included in the first receiving sub-block.

Furthermore, in equation (83), h ₂₂ (f _{b_cfar} , f _{demul_Tx#N1+1} , f _{demul_Tx#N1+2} , to, f _{demul_Tx#Nt} ) represents a column vector consisting of N2 × N4 elements, as shown in the following equation (87), and is a vector whose elements are the outputs of the second Doppler analysis unit 210 obtained by receiving signals transmitted from N2 (= Nt - N1) transmitting antennas 106 included in the first transmitting sub-block by N4 (= Na - N3) receiving antennas 202 included in the second receiving sub-block.

Furthermore, the reception timing of _h21 ( _{fb_cfar} , fdemul_Tx _#1 , fdemul_Tx _#2 , to, _{fdemul_Tx#N1} ) is delayed _{by Tr} relative to the reception timing of _h11 ( _{fb_cfar} , _{fdemul_Tx#1} , _{fdemul_Tx#2} , to, _{fdemul_Tx#N1} ). The term exp[ _-j2πfdestTr ] included in equation ( ₈₂ ) is a correction term that corrects the phase fluctuation due to the delay, based on the Doppler frequency _fdest of the target estimated in the Doppler determination unit 213.

Furthermore, the reception timing of _h12 ( _{fb_cfar} , _{fdemul_Tx#1} , fdemul_Tx _#2 , to _{fdemul_Tx#N1} ) is delayed by T _r relative to the reception timing of _h22 ( _{fb_cfar} , _{fdemul_Tx#1} , _{fdemul_Tx#2} , to _{fdemul_Tx#N1} ). The term exp[-j2πf _dest T _r ] included in equation (83) is a correction term that corrects the phase fluctuation due to the delay, based on the Doppler frequency f _dest of the target estimated in the Doppler determination unit 213.

In addition, in equations (84) and (85), h _1cal[b] is an array correction value that corrects the phase deviation and amplitude deviation between the transmitting antennas Tx#1 to Tx#N1 and between the receiving antennas Rx#1 to #Na, where b is an integer between 1 and (Na×N1).

In addition, in equations (86) and (87), h _2cal[bb] is an array correction value that corrects the phase deviation and amplitude deviation between the transmitting antennas Tx#N1+1 to Tx#Nt and between the receiving antennas Rx#1 to #Na, where bb is an integer from 1 to (Na×N2).

The direction estimation unit 214 calculates a spatial profile by varying the azimuth direction _θu in the direction estimation evaluation function value P _H ( _θu , f _{b_cfar} , f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#Nt} ) within a predetermined angle range. The direction estimation unit 214 extracts a predetermined number of maximum peaks from the calculated spatial profile in descending order, and outputs the azimuth directions of the maximum peaks as direction-of-arrival estimates (e.g., positioning outputs). Here, f _{b_cfar} represents a distance index such that f _{b_cfar} (1) = f _{b_cfar} (2).

Note that there are various methods for calculating the direction estimation evaluation function value P _H (θ _u , f _{b_cfar} , f _{demul_Tx#1} , f _{demul_Tx#2} , ..., f _{demul_Tx#Nt} ) depending on the direction of arrival estimation algorithm. For example, the estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

For example, the beamformer method can be expressed as in the following equation (88): In addition to the beamformer method, methods such as Capon and MUSIC can also be applied. In equation (88), the superscript H is the Hermitian transpose operator.

In equation (88), a ₁ (θ _u ) represents the direction vector (column vector with N1×Na elements) of the N1×Na MIMO virtual receiving antennas configured between the N1 transmitting antennas 106 included in the first transmitting subblock and the Na receiving antennas 202 included in the first and second receiving subblocks, and represents the phase response or complex amplitude response at each virtual receiving antenna that configures the N1×Na MIMO virtual receiving antennas when a reflected wave arrives from the _θu direction.

Also, in equation (88), a ₂ (θ _u ) represents the direction vector (column vector with N2 × Na elements) of the N2 × Na MIMO virtual receiving antennas configured between the N2 transmitting antennas 106 included in the second transmitting sub-block and the Na receiving antennas 202 included in the first and second receiving sub-blocks, and represents the phase response or complex amplitude response at each virtual receiving antenna that configures the N2 × Na MIMO virtual receiving antennas when a reflected wave arrives from the θ _u direction.

The direction vector _aq ( _θu ) may be the phase response or complex amplitude response at each virtual receiving antenna when the wavelength of the radar transmission signal (e.g., the qth chirp signal) at the center frequency _fc (q) is used. Alternatively, the direction vector a( _θu ) of the virtual receiving array for the arriving wave in the azimuth direction θ at the average center frequency of the center frequencies _fc (1) and _fc (2) may be used in common.

The azimuth direction θ _u is a vector obtained by varying the azimuth range in which the direction of arrival estimation is performed at a predetermined azimuth interval β _1. For example, θ _u is set as follows.
θ _u ＝θmin + uβ ₁ , integer u=0～NU
NU=floor[(θmax-θmin)/β ₁ ]+1
Here, floor(x) is a function that returns the maximum integer value that does not exceed the real number x.

In the above example, the direction estimation unit 214 calculates the azimuth direction as the direction of arrival estimate, but this is not limited to this. Direction of arrival estimation can also be performed in the elevation direction, or by using MIMO antennas arranged in a rectangular grid, direction of arrival estimation can also be performed in the azimuth direction and elevation direction. For example, the direction estimation unit 214 may calculate the azimuth direction and elevation direction as the direction of arrival estimate, and use these as the positioning output.

Through the above operations, the direction estimation unit 214 may output, as positioning outputs, distance index f _{b_cfar} and arrival direction estimates for separation index information of Doppler multiplexed signals (f _{demul_Tx#1} , f _{demul_Tx#2} , . . . , f _{demul_Tx#Nt} ). The direction estimation unit 214 may further output, as positioning outputs, distance index f _{b_cfar} and separation index information of Doppler multiplexed signals (f _{demul_Tx#1} , f _{demul_Tx#2} , . . . , f _{demul_Tx#Nt} ). The direction estimation unit 214 may output the positioning output (or the positioning result) to, for example, a vehicle control device in the case of an on-board radar, or an infrastructure control device in the case of an infrastructure radar, both not shown.

In addition, the direction estimation unit 214 may output, for example, either one or both of the Doppler frequency information f _{d_VFT(1)} +n _alest /T _rs determined by the Doppler determination unit 213 and f _c (2)/f _c (1)(f _{d_VFT(1)} +n _alest /T _rs ).

Furthermore, the distance index f _{b_cfar} may be converted into distance information using, for example, equation (1) and then output.

Furthermore, the Doppler frequency information determined by the Doppler determination unit 213 may be converted into relative velocity information and output. The Doppler frequency information f _{d — VFT(1)} + n _alast /2T _r based on the center frequency f _c (1) determined by the Doppler determination unit 213 can be converted into the relative velocity v _d using the following equation (89).

Similarly, when the Doppler frequency information f _c (2)/f _c (1) (f _{d_VFT(1)} +n _alest /T _r ) determined by the Doppler determination unit 213 based on the center frequency f _c (2) is converted into relative velocity v _d , the value obtained is the same as that of equation (89), as shown in the following equation (90). Therefore, the relative velocity component information may be output as a common value (or a unified value) for different center frequencies.

As described above, in this embodiment, the radar device 10b includes multiple radar transmission signal generators 101 and transmits transmission signals from the transmission antenna 106 at predetermined transmission intervals using, for example, a first center frequency and a second center frequency that satisfy any one of equations (10) to (15). This allows the radar device 10b to determine the number of aliases in the Doppler determination unit 213 based on the Doppler frequency shift detected in the Doppler analysis unit 210 and the Doppler demultiplexing unit 212, which correspond to differences in center frequency. Therefore, the radar device 10b can expand the Doppler frequency range (or the maximum value of relative velocity) in which a Doppler multiplexed signal can be separated, for example, depending on the number of aliases that can be determined.

As described above, according to this embodiment, the Doppler frequency range (or the maximum value of relative velocity) in which no ambiguity occurs can be expanded. This allows the radar device 10b to accurately detect targets (e.g., direction of arrival) over a wider Doppler frequency range.

Furthermore, in this embodiment, the Doppler frequency range in which a Doppler multiplexed signal can be separated is expanded by setting the center frequency of the chirp signal, so it is possible to omit the application of a method such as increasing the sampling rate of the A/D converter. Therefore, this embodiment can prevent the hardware configuration of the radar device 10b from becoming complicated and also prevent an increase in power consumption or heat generation in the radar device 10b. Furthermore, in this embodiment, the Doppler frequency range in which a Doppler multiplexed signal can be separated is expanded by setting the center frequency of the chirp signal, so it is possible to omit the application of a method such as shortening the transmission period T _r . Therefore, this embodiment can prevent a reduction in the detectable distance range or a deterioration in distance resolution in the radar device 10b.

Furthermore, in this embodiment, a larger number of MIMO virtual receive antennas can be used in the direction estimation unit 214 compared to embodiment 3. This allows the radar device 10b to improve the SNR and direction estimation accuracy. Furthermore, in this embodiment, a larger number of MIMO virtual receive antennas can be used compared to embodiment 3, so the aperture length of the MIMO virtual receive antennas can be increased and the angular resolution can also be improved.

(Variation 1 of the fourth embodiment)
The configuration of the radar device 10b shown in Fig. 19 may be realized by combining multiple transceiver chips, as shown in Fig. 20. In the example of Fig. 20, the radar device 10b is configured from transceiver chip #1 and transceiver chip #2.

Transmit/receive chip #q includes a radar transmission signal generation unit 101-q, a qth transmission sub-block, a qth reception sub-block, a qth CFAR unit 211, a qth Doppler demultiplexing unit 212, and a qth switching unit 216. Here, q = 1 or 2.

Note that at least one of the Doppler determination unit 213 and the direction estimation unit 214 may be implemented on a separate signal processing chip or ECU, or may be incorporated into either the transmission/reception chip.

The difference between Figure 20 and Figure 19 is that it includes a synchronization signal generator 215, and the synchronization signal (e.g., a reference signal) output from the synchronization signal generator 215 is output to the radar transmission signal generator 101 of the radar transmitter 100b of each transmit/receive chip. This makes it possible to output chirp signals so that the frequency difference between the first chirp signal output from the first transmit sub-block of transmit/receive chip #1 and the second chirp signal output from the second transmit sub-block of transmit/receive chip #2 is within a predetermined allowable error. The configuration shown in Figure 20 also achieves the same effects as in embodiment 4, and by combining general-purpose transmit/receive chips, costs can be reduced.

The above describes one embodiment of the present disclosure.

Other Embodiments
(Variation 1)
For example, in the first embodiment, an example has been shown in which the present invention is applied to a Doppler multiplexing transmission MIMO radar in which the Doppler multiplexing intervals are unequal, but the present invention is not limited to this, and can also be applied to a radar in which there is one transmitting antenna (Nt=1) (for example, a SIMO (Single Input Multiple Output) radar).

Furthermore, the operation according to embodiment 1 can also be applied to radars with a single receiving antenna (Na=1) (e.g., MISO (Multiple Input Single Output) radar).

Furthermore, the operation according to embodiment 1 can also be applied to radars (e.g., SISO (Single Input Single Output) radars) that have one transmitting antenna (Nt-1) and one receiving antenna (Na-1).

Note that when there is one receiving antenna (Na=1), this corresponds to setting Na=1 in embodiment 1, for example, and the same effects as those described in embodiment 1 can be obtained.

Furthermore, when there is one transmitting antenna (Nt=1), this corresponds to, for example, setting Nt=1 in embodiment 1, and the same effects as those described in embodiment 1 can be obtained.

Note that when there is one transmitting antenna (Nt=1), Doppler multiplexing may be omitted. Therefore, Doppler shifting in the Doppler shifter 105 may be omitted. Furthermore, the Doppler demultiplexer 212 does not need to perform Doppler demultiplexing, and may simply perform peak extraction processing on the Doppler analyzer 210 for the index indicated by the output from the CFAR unit 211. Therefore, when there is one transmitting antenna (Nt=1), the radar device configuration shown in FIG. 21 may be used, for example.

Figure 21 is a block diagram showing an example configuration of a radar device 10c. Note that in Figure 21, components similar to those in embodiment 1 (e.g., Figure 2) are assigned the same reference numerals, and their description will be omitted. For example, the radar device 10c shown in Figure 21 may include one transmitting antenna 106c (e.g., Tx#1). Furthermore, the radar device 10c may omit the Doppler shift unit 105 in the radar device 10 shown in Figure 1. Furthermore, the radar device 10c may include, for example, a qth peak extraction unit 218 instead of the qth Doppler demultiplexing unit 212.

Below, we will explain the operations of radar device 10c that differ from those of radar device 10.

In Figure 21, the radar transmission signal output from the radar transmission signal generation unit 101 is output (radiated) from the transmitting antenna Tx#1 without passing through the Doppler shift unit 105.

21 , the qth CFAR unit 211 adaptively sets a threshold value, and outputs distance index f _{b_cfar} (q), Doppler frequency index f _{s_cfar} (q) information, and received power information PowerFT _q (f _{b_cfar} (q), f _{s_cfar} (q)) that result in received power greater than the threshold value to the qth peak extraction unit 218, where q=1, 2.

The q-th peak extraction unit 218 sets, for example, information related to the Doppler frequency index f _{s_cfar} (q) as Doppler index information (f _{demul_Tx#1} (q)). The q-th peak extraction unit 218 also outputs, for example, the distance index f _{b_cfar} (q) and the Doppler index information (f _{demul_Tx#1} (q)) to the Doppler determination unit 213c. The q-th peak extraction unit 218 also outputs, for example, the distance index f _{b_cfar} (q) as well as the Doppler index information (f _{demul_Tx#1} (q)) and the output of the q-th Doppler analysis unit 210 corresponding to the Doppler index information (f _{demul_Tx#1} (q)) to the direction estimation unit 214c. Here, q=1, 2.

The Doppler determination unit 213c determines the Doppler frequency assuming a case in which the target _Doppler frequency fd_TargetDoppler includes a target having a Doppler frequency outside the Doppler frequency range -1/( _2Trs ) ≦ _{fd_TargetDoppler < 1/} (2Trs), using the Doppler index information ( _{fdemul_Tx#1} (1)) output from the first peak extraction unit 218 and the _Doppler index information ( _{fdemul_Tx} #1( ₂ )) output from the second peak extraction unit 218, which have the same distance indexes fb_cfar(1) and fb_cfar( ₂ ). Note that the operation of the Doppler determination unit 213c is similar to that of the Doppler determination unit 213 in the first embodiment, and therefore description thereof will be omitted.

The direction estimation unit 214c extracts the output of the q-th Doppler analysis unit 210 based on the distance index f _{b_cfar} (q) and Doppler index information (f _{demul_Tx#1} (q)) from the q-th peak extraction unit 218, generates the q-th virtual receiving array correlation vector h _q (f _{b_cfar} (q), f _{demul_Tx#1} (q), f _{demul_Tx#2} (q), ..., f _{demul_Tx#Nt} (q)), and performs direction estimation processing, where q = 1, 2.

The above operation allows the same effects as in embodiment 1 to be achieved even in a radar device 10 (e.g., a SIMO radar) with one transmitting antenna (Nt-1).

(Variation 2)
In the above embodiments, the radar device 10 has been described as transmitting radar transmission signals simultaneously from multiple transmission antennas 106 (Doppler multiplex transmission). However, the present invention is not limited to this. For example, the radar transmission signals may be transmitted by switching between the multiple transmission antennas 106.

For example, as shown in FIG. 22 , when the radar device 10 switches between two transmitting antennas 106 to transmit radar transmission signals (e.g., chirp signals), the radar device 10 may transmit a first chirp signal having a center frequency f _c (1) with a period T _rs1 from the first transmitting antenna 106 (first transmitting antenna) and a second chirp signal having a center frequency f _c (2) with a period T _rs1 , and perform the same processing as in the first embodiment on the received signals obtained from the reflected waves of these transmitted signals. This enables the radar device 10 to determine the Doppler frequency of the target even when the Doppler frequency f _{d_TargetDoppler} exceeds the Doppler frequency range −1/(2T _rs1 ) ≦ f _{d_TargetDoppler} < 1/(2T _rs ). Here, T _rs1 is the period for transmitting the first chirp signal (or the period for transmitting the second chirp signal) from the first transmitting antenna 106.

22 , the radar device 10 may transmit a first chirp signal having a center frequency f _c (1) with a period T _rs2 from the second transmitting antenna 106 (second transmitting antenna) and a second chirp signal having a center frequency f _c (2) with a period T _rs2 , and perform the same processing as in the first embodiment on the received signals obtained from the reflected waves of these transmitted signals. This enables the radar device 10 to determine the Doppler frequency f _{d_TargetDoppler} of the target even when the Doppler frequency f _{d_TargetDoppler} exceeds the Doppler frequency range −1/(2T _rs2 ) ≦ f d_TargetDoppler < 1/(2T _rs2 ). Here, T _rs2 is the period for transmitting the first chirp signal (or the period for transmitting the second chirp signal) from the second transmitting antenna 106.

Note that the number of transmitting antennas 106 that switch the transmission of radar transmission signals is not limited to two, and may be three or more.

This concludes the explanation of Variation 2.

In a radar device according to an embodiment of the present disclosure, the radar transmitter and radar receiver may be individually located at physically separate locations. Furthermore, in a radar receiver according to an embodiment of the present disclosure, the direction estimation unit and other components may be individually located at physically separate locations.

Furthermore, in one embodiment of the present disclosure, for example, the center frequency f _c (q), the number of transmitting antennas Nt, the number of receiving antennas Na, the Doppler multiplexing number N _DM , values related to the determinable conditions (α, N _c, etc.), values related to the phase rotation (δ, φ ₀ , δ, Δφ ₀ , dp _n, etc.), and numerical values used for the frequencies are merely examples and are not limited to these values.

A radar device according to one embodiment of the present disclosure includes, for example, a CPU (Central Processing Unit), a storage medium such as a ROM (Read Only Memory) that stores a control program, and a working memory such as a RAM (Random Access Memory), although these are not shown. In this case, the functions of each of the above-mentioned units are realized by the CPU executing the control program. However, the hardware configuration of the radar device is not limited to this example. For example, each functional unit of the radar device may be realized as an integrated circuit (IC). Each functional unit may be individually implemented as a single chip, or a single chip may include some or all of the functional units.

Although various embodiments have been described above with reference to the drawings, it goes without saying that the present disclosure is not limited to such examples. It is clear that a person skilled in the art could conceive of various modifications or alterations within the scope of the claims, and it is understood that these naturally fall within the technical scope of the present disclosure. Furthermore, the components of the above embodiments may be combined in any manner as long as they do not deviate from the spirit of the disclosure.

Furthermore, the notation "... section" in the above-described embodiments may be replaced with other notations such as "... circuit," "... assembly," "... device," "... unit," or "... module."

In the above embodiments, the present disclosure has been described as being configured using hardware, but the present disclosure can also be realized using software in conjunction with hardware.

Furthermore, each functional block used in the description of each of the above embodiments is typically realized as an LSI, which is an integrated circuit. The integrated circuit controls each functional block used in the description of the above embodiments and may have input and output terminals. These may be individually integrated into single chips, or some or all of them may be integrated into a single chip. Here, we refer to it as an LSI, but depending on the level of integration, it may also be called an IC, system LSI, super LSI, or ultra LSI.

Furthermore, the method of integration is not limited to LSI, but may also be realized using dedicated circuits or general-purpose processors and memories. It is also possible to use FPGAs (Field Programmable Gate Arrays), which can be programmed after LSI manufacturing, or reconfigurable processors, which allow the connections or settings of circuit cells within LSIs to be reconfigured.

Furthermore, if advances in semiconductor technology or derivative technologies lead to the emergence of integrated circuit technology that can replace LSI, it would naturally be possible to use that technology to integrate functional blocks. The application of biotechnology, for example, is also a possibility.

Summary of this disclosure
A radar device according to one embodiment of the present disclosure includes a transmission circuit that outputs, for each transmission period, a first transmission signal having a first center frequency and a second transmission signal having a second center frequency that is a center frequency higher than the first center frequency, and a transmission antenna that transmits the first transmission signal and the second transmission signal, wherein the second center frequency is a frequency that is higher than (1+1/ _Nc ) times the first center frequency ( _Nc is an integer indicating the number of times that each of the first transmission signal and the second transmission signal is transmitted for each transmission period within a predetermined period).

In one embodiment of the present disclosure, the second center frequency is lower than 1.25 times the first center frequency.

In one embodiment of the present disclosure, the second center frequency is lower than (7/6) times the first center frequency.

In one embodiment of the present disclosure, the transmitting antenna alternately transmits the first transmission signal and the second transmission signal for each transmission period.

In one embodiment of the present disclosure, the transmitting antenna simultaneously transmits the first transmission signal and the second transmission signal for each transmission period.

In one embodiment of the present disclosure, the transmitting antenna is a plurality of transmitting antennas, and among the plurality of transmitting antennas, the number of transmitting antennas that transmit the first transmitting signal and the number of transmitting antennas that transmit the second transmitting signal are the same or differ by one.

In one embodiment of the present disclosure, the system further includes a receiving circuit including: a receiving antenna that receives a first reflected wave signal resulting from the first transmission signal reflected by a target; and a second reflected wave signal resulting from the second transmission signal reflected by the target; a first Doppler analysis circuit that estimates a first Doppler frequency from the first reflected wave signal; a second Doppler analysis circuit that estimates a second Doppler frequency from the second reflected wave signal; and a determination circuit that determines the number of aliasing of the first Doppler frequency and the second Doppler frequency, wherein the determination circuit estimates a first peak position of the first Doppler frequency observed in the first reflected wave signal, estimates a second peak position of the second Doppler frequency based on the first peak position and the ratio between the first center frequency and the second center frequency, and determines the number of aliasing of the Doppler frequency of the target based on the degree of coincidence between the second peak position and a third peak position observed in the second reflected wave signal.

In one embodiment of the present disclosure, the transmitting antenna comprises a plurality of transmitting antennas, and the transmitting circuit imparts a Doppler shift amount to at least one of the first transmitting signal and the second transmitting signal transmitted from the plurality of transmitting antennas, the Doppler shift amount being determined at intervals obtained by dividing the Doppler frequency range, which is the target for determining the number of Doppler frequency foldbacks, into uneven intervals.

In one embodiment of the present disclosure, when the interval between the unequally divided Doppler shift amounts is Δf _DDM =1/(T _rs (N _t +δ)) (Nt is an integer indicating the number of the multiple transmitting antennas, δ is an integer greater than or equal to 1, and T _rs is the transmission period during which the set of the first transmission signal and the second transmission signal is transmitted), the second center frequency is a frequency higher than N _c /(N _c -(N _t +δ)) times the first center frequency.

In one embodiment of the present disclosure, the system further includes a receiving circuit including: a receiving antenna that receives a plurality of first reflected wave signals resulting from the first transmission signal reflected by a plurality of targets; and a plurality of second reflected wave signals resulting from the second transmission signal reflected by the plurality of targets; a first Doppler analysis circuit that estimates a first Doppler frequency from the plurality of first reflected wave signals; a second Doppler analysis circuit that estimates a second Doppler frequency from the plurality of second reflected wave signals; and a determination circuit that determines the number of aliasing of the first Doppler frequency and the second Doppler frequency, wherein the determination circuit determines the number of aliasing of the Doppler frequency of each of the plurality of targets based on the interval between peak positions between the plurality of targets and the interval between the Doppler shift amounts, based on the Doppler frequencies estimated by the Doppler analysis circuit that separates the first reflected wave signals and the second reflected wave signals, out of the first Doppler analysis circuit and the second Doppler analysis circuit.

In one embodiment of the present disclosure, the system further includes a receiving circuit including: a receiving antenna that receives a plurality of first reflected wave signals resulting from the first transmission signal being reflected by a plurality of targets; and a plurality of second reflected wave signals resulting from the second transmission signal being reflected by the plurality of targets; a first Doppler analysis circuit that estimates a first Doppler frequency from the plurality of first reflected wave signals; a second Doppler analysis circuit that estimates a second Doppler frequency from the plurality of second reflected wave signals; and a determination circuit that determines the number of times the first Doppler frequency and the second Doppler frequency are folded back; and the receiving circuit further includes a direction estimation circuit that performs direction estimation based on the Doppler frequencies estimated by the Doppler analysis circuit that separates the first reflected wave signals and the second reflected wave signals, out of the first Doppler analysis circuit and the second Doppler analysis circuit.

In one embodiment of the present disclosure, the system further comprises a receiving circuit including: a receiving antenna that receives a first reflected wave signal resulting from the first transmission signal being reflected by a target, and a second reflected wave signal resulting from the second transmission signal being reflected by the target; a first Doppler analysis circuit that estimates a first Doppler frequency from the first reflected wave signal; a second Doppler analysis circuit that estimates a second Doppler frequency from the second reflected wave signal; and a determination circuit that determines the number of times the first Doppler frequency and the second Doppler frequency are folded back; and a second receiving antenna, wherein the first Doppler analysis circuit processes the first reflected wave signal received by the first receiving antenna in either an even-numbered or odd-numbered transmission cycle, and processes the first reflected wave signal received by the second receiving antenna in the other of the even-numbered or odd-numbered transmission cycles, and the second Doppler analysis circuit processes the second reflected wave signal received by the second receiving antenna in the one transmission cycle, and processes the second reflected wave signal received by the first receiving antenna in the other transmission cycle.

In one embodiment of the present disclosure, the device further includes a receiving antenna; a first receiving circuit that, in either an even-numbered or odd-numbered transmission cycle, mixes the signal received by the receiving antenna with the first transmission signal to output a first reflected wave signal resulting from the first transmission signal being reflected by a target, and that, in the other of the even-numbered or odd-numbered transmission cycles, mixes the signal received by the receiving antenna with the second transmission signal to output a second reflected wave signal resulting from the second transmission signal being reflected by the target; and a second receiving circuit that, in the one transmission cycle, mixes the signal received by the receiving antenna with the second transmission signal to output the second reflected wave signal, and that, in the other transmission cycle, mixes the signal received by the receiving antenna with the first transmission signal to output the first reflected wave signal.

In one embodiment of the present disclosure, the first transmission signal and the second transmission signal are chirp signals, and the chirp signal with the first center frequency and the chirp signal with the second center frequency have the same frequency sweep bandwidth.

In one embodiment of the present disclosure, the chirp signal of the first center frequency and the chirp signal of the second center frequency have different frequency sweep times.

In one embodiment of the present disclosure, the radio wave receiver further includes a first receiving antenna that receives a first reflected wave signal formed when the first transmission signal is reflected by a target, a second receiving antenna that receives a second reflected wave signal formed when the second transmission signal is reflected by the target, a first receiving circuit that processes the first reflected wave signal, and a second receiving circuit that processes the second reflected wave signal. The transmitting circuit includes a first transmitting circuit that outputs the first transmission signal and a second transmitting circuit that outputs the second transmission signal. The transmitting antenna includes a first transmitting antenna that transmits the first transmission signal and a second transmitting antenna that transmits the second transmission signal. The first transmitting antenna, the first transmitting circuit, the first receiving antenna, and the first receiving circuit are included in a first chip, and the second transmitting antenna, the second transmitting circuit, the second receiving antenna, and the second receiving circuit are included in a second chip.

In one embodiment of the present disclosure, there is provided a first receiving antenna that receives a first reflected wave signal resulting from the first transmission signal being reflected by a target and a second reflected wave signal resulting from the second transmission signal being reflected by the target; a second receiving antenna that receives the first reflected wave signal and the second reflected wave signal; a first receiving circuit that processes the first reflected wave signal received by the first receiving antenna in either an even-numbered or odd-numbered transmission period and processes the second reflected wave signal received by the first receiving antenna in the other of the even-numbered or odd-numbered transmission periods; and a second receiving circuit that processes the second reflected wave signal received by the second receiving antenna in the one of the transmission periods. and a second receiving circuit that processes the first reflected wave signal received by the second receiving antenna during the other transmission period, wherein the transmitting circuit includes a first transmitting circuit that outputs the first transmission signal and a second transmitting circuit that outputs the second transmission signal, and the transmitting antenna includes a first transmitting antenna that transmits the first transmission signal and a second transmitting antenna that transmits the second transmission signal, wherein the first transmitting antenna, the first transmitting circuit, the first receiving antenna, and the first receiving circuit are included in a first chip, and the second transmitting antenna, the second transmitting circuit, the second receiving antenna, and the second receiving circuit are included in a second chip.

In one embodiment of the present disclosure, a transmission circuit is provided that imparts Doppler shift amounts to a first transmission signal having a first center frequency and a second transmission signal having a second center frequency that is higher than the first center frequency, at intervals obtained by dividing, at irregular intervals, a Doppler frequency range that is a target for determining the number of times of Doppler frequency folding, and a plurality of transmission antennas that transmit the first transmission signal and the second transmission signal to which the Doppler shift amounts have been imparted, wherein the transmission circuit outputs the first transmission signal and the second transmission signal for each transmission period, the second center frequency being a frequency that is higher than (1+1/N _c ) times the first center frequency (N _c is an integer indicating the number of times that each of the first transmission signal and the second transmission signal is transmitted for each transmission period within a predetermined period), and the intervals between the Doppler shift amounts that are divided at irregular intervals are set to Δf _DDM =1/(T _rs (N _t +δ)) (N t is an integer indicating the number of the plurality of transmission antennas, and δ is an integer equal to or greater than 1), _rs is a transmission period in which the set of the first transmission signal and the second transmission signal is transmitted, and the second center frequency is higher than _Nc /( _Nc- ( _Nt +δ)) times the first center frequency.

This disclosure is suitable for use as a radar device that detects a wide angle range.

10, 10a, 10b, 10c Radar device 100, 100a, 100b, 100c Radar transmitter 101 Radar transmission signal generator 102 Modulation signal generator 103 VCO
104 Signal generation control unit 105 Doppler shift unit 106, 106c Transmission antenna 200, 200a, 200b, 200c Radar receiving unit 201 Antenna system processing unit 202 Receiving antenna 203 Receiving radio unit 204 Mixer unit 205 LPF
206 Signal processing unit 207 A/D conversion unit 208 Beat frequency analysis unit 209, 217 Output switching unit 210 Doppler analysis unit 211 CFAR unit 212 Doppler demultiplexing unit 213, 213c Doppler determination unit 214, 214c Direction estimation unit 215 Synchronization signal generation unit 216 Switching unit 218 Peak extraction unit

Claims (15)

a transmission circuit that outputs, in each transmission period, a first transmission signal having a first center frequency and a second transmission signal having a second center frequency that is higher than the first center frequency;
a transmitting antenna for transmitting the first transmission signal and the second transmission signal;
a receiving antenna that receives a first reflected wave signal resulting from the first transmission signal being reflected by a target and a second reflected wave signal resulting from the second transmission signal being reflected by the target;
a receiving circuit including: a first Doppler analysis circuit that estimates a first Doppler frequency from the first reflected wave signal; a second Doppler analysis circuit that estimates a second Doppler frequency from the second reflected wave signal; and a determination circuit that determines the number of times that the first Doppler frequency and the second Doppler frequency are folded back;
Equipped with
the second center frequency is a frequency higher than (1+1/Nc) times the first center frequency (Nc is an integer indicating the number of times that each of the first transmission signal and the second transmission signal is transmitted per transmission period within a predetermined period),
The determination circuit
estimating a first peak position of the first Doppler frequency observed by the first reflected wave signal;
estimating a second peak position of the second Doppler frequency based on the first peak position and a ratio between the first center frequency and the second center frequency;
determining the number of times the Doppler frequency of the target has folded back based on the degree of coincidence between the second peak position and a third peak position observed in the second reflected wave signal;
Radar equipment. the second center frequency is lower than 1.25 times the first center frequency;
The radar device according to claim 1 . the second center frequency is lower than 7/6 times the first center frequency;
The radar device according to claim 1 . the transmitting antenna alternately transmits the first transmission signal and the second transmission signal for each transmission period;
The radar device according to claim 1 . the transmitting antenna simultaneously transmits the first transmission signal and the second transmission signal for each transmission period;
The radar device according to claim 1 . the transmitting antenna is a plurality of transmitting antennas;
Among the plurality of transmitting antennas, the number of transmitting antennas that transmit the first transmission signals and the number of transmitting antennas that transmit the second transmission signals are the same or different by one.
The radar device according to claim 5 . the transmitting antenna is a plurality of transmitting antennas;
the transmission circuit imparts, to at least one of the first transmission signal and the second transmission signal transmitted from the plurality of transmission antennas, Doppler shift amounts at intervals obtained by dividing, at unequal intervals, a Doppler frequency range that is a target for determining the number of times the Doppler frequency is folded back;
The radar device according to claim 1 . The intervals between the Doppler shift amounts divided into the unequal intervals are
(Nt is an integer indicating the number of the plurality of transmitting antennas, δ is an integer equal to or greater than 1, and _Trs is a transmission period during which the set of the first transmission signal and the second transmission signal is transmitted),
The second center frequency is a frequency
is a higher frequency than
The radar device according to claim 7 . the target is a plurality of targets;
the first reflected wave signal is a plurality of first reflected wave signals,
the second reflected wave signal is a plurality of second reflected wave signals,
the first Doppler analysis circuit estimates the first Doppler frequency from the plurality of first reflected wave signals;
the second Doppler analysis circuit estimates the second Doppler frequency from the plurality of second reflected wave signals;
the receiving circuit further includes a direction estimation circuit, of the first Doppler analysis circuit and the second Doppler analysis circuit, that performs direction estimation based on a Doppler frequency estimated by the Doppler analysis circuit that separates the first reflected wave signal and the second reflected wave signal.
The radar device according to claim 8 . the receiving antennas include a first receiving antenna and a second receiving antenna;
the first Doppler analysis circuit processes the first reflected wave signal received by the first receiving antenna in either an even-numbered or odd-numbered transmission period, and processes the first reflected wave signal received by the second receiving antenna in the other of either an even-numbered or odd-numbered transmission period;
the second Doppler analysis circuit processes the second reflected wave signal received by the second receiving antenna in the one transmission period, and processes the second reflected wave signal received by the first receiving antenna in the other transmission period.
The radar device according to claim 5 . the receiving antennas include a first receiving antenna and a second receiving antenna;
The receiving circuit
a first receiving circuit that mixes a signal received by the first receiving antenna with the first transmission signal every first period to output the first reflected wave signal, and that mixes a signal received by the first receiving antenna with the second transmission signal every second period different from the first period to output the second reflected wave signal;
a second receiving circuit that mixes the signal received by the second receiving antenna with the second transmission signal every first period to output the second reflected wave signal, and that mixes the signal received by the second receiving antenna with the first transmission signal every second period to output the first reflected wave signal.
The radar device according to claim 5 . the first transmission signal and the second transmission signal are chirp signals;
The chirp signal of the first center frequency and the chirp signal of the second center frequency have the same frequency sweep bandwidth.
The radar device according to claim 1 . The chirp signal of the first center frequency and the chirp signal of the second center frequency have different frequency sweep times.
The radar device according to claim 12 . The receiving antenna is
a first receiving antenna that receives the first reflected wave signal that is the first transmitted signal reflected by the target;
a second receiving antenna that receives the second reflected wave signal that is the second transmitted signal reflected by the target ,
the receiving circuit includes a first receiving circuit that processes the first reflected wave signal and a second receiving circuit that processes the second reflected wave signal;
the transmission circuit includes a first transmission circuit that outputs the first transmission signal and a second transmission circuit that outputs the second transmission signal;
the transmitting antennas include a first transmitting antenna that transmits the first transmission signal and a second transmitting antenna that transmits the second transmission signal;
the first transmitting antenna, the first transmitting circuit, the first receiving antenna, and the first receiving circuit are included in a first chip;
the second transmitting antenna, the second transmitting circuit, the second receiving antenna, and the second receiving circuit are included in a second chip;
The radar device according to claim 5 . The receiving antenna is
a first receiving antenna that receives the first reflected wave signal resulting from the first transmission signal being reflected by the target and the second reflected wave signal resulting from the second transmission signal being reflected by the target;
a second receiving antenna that receives the first reflected wave signal and the second reflected wave signal ,
The receiving circuit
a first receiving circuit that processes the first reflected wave signal received by the first receiving antenna in either an even-numbered or odd-numbered transmission period, and processes the second reflected wave signal received by the first receiving antenna in the other of the even-numbered or odd-numbered transmission periods;
a second receiving circuit that processes the second reflected wave signal received by the second receiving antenna in the one transmission period and processes the first reflected wave signal received by the second receiving antenna in the other transmission period;
Including ,
the transmission circuit includes a first transmission circuit that outputs the first transmission signal and a second transmission circuit that outputs the second transmission signal;
the transmitting antennas include a first transmitting antenna that transmits the first transmission signal and a second transmitting antenna that transmits the second transmission signal;
the first transmitting antenna, the first transmitting circuit, the first receiving antenna, and the first receiving circuit are included in a first chip;
the second transmitting antenna, the second transmitting circuit, the second receiving antenna, and the second receiving circuit are included in a second chip;
The radar device according to claim 5 .