JP7813675B2 - Motor control device and electric vehicle system - Google Patents

Description

The present invention relates to a motor control device and an electric vehicle system using the same.

Permanent magnet synchronous motors are widely used in fields such as electric vehicles because the magnetic field is provided by permanent magnets, which is advantageous for achieving high efficiency. In permanent magnet synchronous motors, when the magnetic flux of the permanent magnet fluctuates due to factors such as temperature, motor characteristics such as torque characteristics fluctuate. For this reason, in some applications, the magnetic flux of the permanent magnet is estimated, and the permanent magnet synchronous motor is controlled based on the estimated magnetic flux.

Technologies described in Patent Documents 1 and 2 are known as conventional techniques for estimating the magnetic flux amount of a permanent magnet synchronous motor.

In the technology described in Patent Document 1, an estimated value Δφest of magnetic flux change from a reference state, for example, a reference temperature state, is estimated from a d-axis voltage Vd_std and a q-axis voltage Vq_std, a d-axis corrected voltage command value Vd ^** and a q-axis corrected voltage command value Vq ^** in the reference state, and an electrical angular velocity ω (Δφest = ((Vq ^** /Vd ^** ) × Vd_std - Vq_std)/ω). Vd ^** and Vd ^** are corrected values of the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* , respectively. Vd ^* and Vq ^* are corrected according to a dead time Tdead of a switching element in the inverter and a DC voltage drop of the switching element. The correction amounts of Vd ^* and Vq ^* according to Tdead are calculated based on the inverter's DC input voltage Vdc, switching frequency Fsw, Tdead, and current phase βd based on the d-axis (see paragraph [0054]). Also, the correction amounts of Vd ^* and Vq ^* according to the DC voltage drop of the switching elements are calculated using a voltage drop correction map that uses current as an argument.

In the technology described in Patent Document 2, a magnetic flux estimation value calculated based on the q-axis voltage command value, d-axis voltage command value, q-axis current, d-axis current, and angular velocity ω is corrected by an error correction value calculated using multiple maps that use multiple parameters related to the operating state of the permanent magnet synchronous motor, namely, angular velocity, torque (or current), temperatures of various parts (sensor temperature, motor temperature, inverter temperature), and inverter DC voltage, as arguments.

Japanese Patent Application Laid-Open No. 2019-129572 Japanese Patent Application Laid-Open No. 2019-129573

In Patent Document 1, the ratio (Vq ^** /Vd ^** ) of the q-axis corrected voltage command value Vq ^** to the d-axis corrected voltage command value Vd ^** is used to calculate the magnetic flux fluctuation amount estimated value Δφest, so when Vd ^** is small, the magnetic flux fluctuation amount estimated value Δφest itself increases, which poses a problem in the reliability of the estimated value.

In Patent Document 2, the q-axis voltage command value and d-axis voltage command value used to calculate the magnetic flux estimation value are not corrected. This limits the improvement in the accuracy of the magnetic flux estimation value.

The present invention therefore provides a motor control device that can improve the accuracy and reliability of magnetic flux estimation, as well as an electric vehicle system that uses this motor control device.

To solve the above problem, the motor control device of the present invention includes a voltage command creation unit that generates a voltage command for controlling the output voltage of a power conversion device to a motor in accordance with a speed command or torque command for the motor, whose rotor has a permanent magnet; a dead time compensation unit that performs dead time compensation on the voltage command; and a magnetic flux estimation unit that estimates the magnetic flux of the permanent magnet. The magnetic flux estimation unit estimates the magnetic flux of the permanent magnet based on the voltage command and voltage drops that occur between the motor and the power conversion device, and between the motor and the power conversion device. The voltage drops include a first voltage drop component due to resistance components between the motor and the power conversion device, and between the motor and the power conversion device, a second voltage drop component that occurs due to dead time compensation, and a third voltage drop component obtained by subtracting the first and second voltage drop components from the voltage drop.

To solve the above problems, the electric vehicle system of the present invention includes wheels, a motor that drives the wheels, a power conversion device that supplies power to the motor, and a control device that controls the power conversion device, where the control device is the motor control device of the present invention.

The present invention can improve the accuracy and reliability of magnetic flux estimation.

Other issues, configurations, and advantages will become clear from the description of the following embodiments.

1 is a functional block diagram showing a schematic configuration of a motor drive system according to a first embodiment; 1 is a cross-sectional view showing the structure of a permanent magnet synchronous motor 10 in a direction perpendicular to the rotation axis. 1 is a diagram showing the relationship between the dq axes and three-phase windings 15 in a stator 11 of a permanent magnet synchronous motor 10. 2 is a circuit diagram showing the configuration of a power conversion device 20 (FIG. 1). FIG. 10 is a waveform diagram showing an example of drive signals (G _XP , G _XN ) (X: any one of U, V, and W). FIG. FIG. 2 is a functional block diagram showing the configuration of a voltage command generating unit 41. FIG. 2 is a functional block diagram showing the configuration of a magnetic flux estimation unit 50 (FIG. 1) in the first embodiment. 10 is a calculation example of a q-axis voltage drop V _{drop — q} ′ shown as a contour map on the I _d -I _q plane. 9 is a graph showing the relationship between _Iq and V _{drop_q} shown in FIG. 8, with _Id as a parameter. 10 is a calculation example of a second q-axis voltage drop V _{drop — q} ′ shown as a contour map on the I _d -I _q plane. 11 is a graph showing the relationship between _Iq and V _{drop — q} ′ shown in FIG. 10 with _Id as a parameter. 10 is a calculation example of a third q-axis voltage drop V _{drop — q} ″, which is shown as a contour map on the I _d -I _q plane. FIG. 4 is a functional block diagram showing the configuration of a calculation unit for a q-axis voltage drop V _{drop_q_est} in the motor control device 40. FIG. 10 is a functional block diagram showing a schematic configuration of a motor drive system according to a second embodiment. FIG. 15 is a functional block diagram showing an example of the configuration of a command value corrector 48 (FIG. 14). FIG. 11 is a functional block diagram showing a schematic configuration of a motor control device in a motor drive system according to a third embodiment. FIG. 10 is a functional block diagram showing the configuration of an electric vehicle system 300 according to a fourth embodiment.

Embodiments of the present invention will be described below using the following Examples 1 to 4 with reference to the drawings.

In each figure, the same reference numbers indicate the same components or components with similar functions.

Figure 1 is a functional block diagram showing the general configuration of a motor drive system according to a first embodiment of the present invention.

The motor drive system 100 includes a permanent magnet synchronous motor 10, a power conversion device 20, a current sensor 30, and a motor control device 40. Although not shown, a mechanical load is connected to the permanent magnet synchronous motor 10 via a transmission mechanism that mechanically or magnetically transmits the mechanical output of the permanent magnet synchronous motor 10.

The motor control device 40 controls the power conversion device 20 to control the rotation speed or torque of the permanent magnet synchronous motor 10 to a desired value. In addition, in this embodiment, the motor control device 40 has the function of estimating the magnetic flux and temperature of the permanent magnets provided in the permanent magnet synchronous motor 10, as described below.

The following describes each component of the motor drive system shown in Figure 1.

Figure 2 is a cross-sectional view perpendicular to the rotation axis showing the structure of the permanent magnet synchronous motor 10.

The permanent magnet synchronous motor 10 is composed of a stator 11 and a rotor 12 that face each other across an air gap. The stator 11 has multiple stator poles 18, each of which has windings 15 wound around a stator core 16 made of a magnetic material. The rotor 12 has a rotor core 13 made of a magnetic material and a permanent magnet 17 located within the rotor core 13.

In this embodiment, as shown in Figure 2, the number of stator poles 18 (= number of slots) is six, and the number of magnetic poles in the rotor 12 is two. Note that the number of magnetic poles in the stator 11 and rotor 12 is not limited to the number shown in Figure 2 and can be set arbitrarily depending on the desired motor characteristics. Furthermore, the multiple windings may be connected in either parallel or series. In this embodiment, the windings 15 of two opposing stator poles 18 in the stator 11 are connected in series to form one phase of winding. Therefore, the stator 11 has three phases of winding (U, V, W).

When a current is passed through the windings 15, the stator magnetic poles 18 generate magnetic poles of either north or south polarity, and the polarity can be changed depending on the direction of the current in the windings 15. In this embodiment, when a positive DC current is passed through the windings 15, the stator magnetic poles 18 become south poles, and the rotation angle of the rotor 12 when the north pole of the permanent magnet 17 in the rotor 12 is attracted to this south pole is defined as zero degrees. Note that hereafter, the rotation angle of the rotor 12 will be referred to as the "rotor position."

When there are multiple windings 15, the position of any one of the windings is taken as the reference position. In this embodiment, the position of the winding 15 (U-phase winding) located on the right side in Figure 2 is taken as the reference position. Furthermore, counterclockwise rotation of the rotor 12 is referred to as forward rotation, and clockwise rotation is referred to as reverse rotation.

In this embodiment, the rotor position is detected by a rotation detector such as a resolver or encoder, and is used for control processing in the motor control device 40. Note that the rotor position may also be estimated based on the motor current flowing through the permanent magnet synchronous motor 10 or the voltage applied to the permanent magnet synchronous motor 10, without using a rotation detector, by applying so-called position sensorless control.

In this embodiment, as shown in Figure 2, an embedded magnet type permanent magnet synchronous motor 10 is used, but this is not limited to this. A motor with a rotor equipped with a permanent magnet whose magnetic flux changes depending on the rotor temperature, such as a surface magnet type permanent magnet synchronous motor, may also be used.

The motor control device 40 uses so-called vector control. Therefore, we will explain the relationship between the permanent magnet synchronous motor 10 and the rotating coordinate axes (d-q axes) used in vector control.

Figure 3 shows the relationship between the d-q axes and the three-phase (U, V, W) windings 15 on the stator 11 of the permanent magnet synchronous motor 10.

As shown in Figure 3, the three-phase windings (U, V, W) are arranged with a phase difference of 120 electrical degrees. The direction of the main magnetic flux of the permanent magnet 17 in the rotor 12 is the d-axis, and the direction 90 electrical degrees ahead of the d-axis in the direction of rotation is the q-axis.

The d-q axes are coordinate axes in a rotating coordinate system, i.e., coordinate axes set on the rotor 12, and rotate together with the rotor 12. Therefore, the rotation angle of the d-axis is equal to the rotor position (θd). As mentioned above, in this embodiment, the position of the U-phase winding is used as the reference position, so as shown in Figure 3, θd is expressed as an angle with the counterclockwise rotation direction from the position of the U-phase winding being positive.

The direction of the d-axis is also the direction toward the position of the winding at the reference position when the magnetic flux of the permanent magnet 17 that links with the winding 15 at the reference position (in this embodiment, the U-phase winding) is at its maximum when the rotor 12 is rotated. Therefore, in Figure 3, when θd = 0, the magnetic flux of the permanent magnet 17 that links with the U-phase winding is at its maximum.

Figure 4 is a circuit diagram showing the configuration of the power conversion device 20 (Figure 1). Figure 4 also shows the permanent magnet synchronous motor 10 and current sensor 30 (Figure 1).

As shown in Figure 4, the power conversion device 20 has a DC voltage source 120, a main circuit 131 consisting of a three-phase full-bridge circuit of semiconductor switching elements 22a to 22f, and a gate drive circuit 123. A shunt resistor 135 is connected between the DC voltage source 120 and the main circuit 131. This prevents excessive current from flowing through the main circuit 131.

The DC voltage source 120 may be a battery, an AC/DC converter (including a rectifier), or a DC/DC converter. In this embodiment, IGBTs are used as the semiconductor switching elements 22a-22f, but this is not limiting and MOSFETs or other elements may also be used. Furthermore, in this embodiment, the main circuit 131 is composed of an inverter module.

In the main circuit 131, a free-wheeling diode is connected in parallel to each of the semiconductor switching elements 22a to 22f, and each parallel connection forms an arm. Two arms are connected in series to form upper and lower arms for one phase. In this embodiment, as shown in Figure 4, the upper and lower arms for the U phase are formed by switching elements 32a and 32b, the V phase by switching elements 32c and 32d, and the W phase by switching elements 32e and 32f. The connection points of the upper and lower arms for each phase are connected to the permanent magnet synchronous motor 10.

The gate drive circuit 123 generates drive signals G _UP to G _WN based on the on/off control signals S _UP to S _WN , which are PWM pulse signals output by the PWM signal generation unit 43 (FIG. 1), and outputs them to the control terminals (gate terminals in this embodiment) of the semiconductor switching elements 22 a to 22 f, respectively. The semiconductor switching elements 22 a to 22 f are driven to turn on and off by the drive signals G _UP to G _WN , respectively.

By driving the semiconductor switching elements 22a to 22f on and off, the main circuit 131 converts the DC voltage Edc from the DC voltage source 120 into a three-phase AC voltage and outputs this three-phase AC voltage to the three-phase AC terminals of the permanent magnet synchronous motor 10.

Fig. 5 is a waveform diagram showing an example of the drive signals (G _XP , G _XN ) (X: any one of U, V, and W). Fig. 5 also shows an example of the triangular wave carrier signal _SC and the X-phase voltage command V _X ^* used by the PWM signal generator 43 (Fig. 1).

As shown in Fig. 5, the PWM signal generator 43 (Fig. 1) generates on/off control signals S _XP and S _XN made up of PWM pulse signals (not shown in Fig. 5) based on the magnitude relationship between the triangular wave carrier signal S _C and the X-phase voltage command V _X ^* . In response to the off control signals S _XP and S _XN , the gate drive circuit 123 generates drive signals G _XP and G _XN which have the same waveforms as the off control signals S _XP and S _XN and have voltage values sufficient to drive the semiconductor switching elements.

In the power conversion device 20, the semiconductor switching elements are switched at a frequency sufficiently higher than the frequency of the AC voltage output to the permanent magnet synchronous motor 10. That is, the frequency of the triangular wave carrier signal _SC is set sufficiently higher than the frequency of the X-phase voltage command _VX ^* . Therefore, the voltages of each phase output from the main circuit 131 in the power conversion device 20 are rectangular wave voltages pulse-width modulated in accordance with the drive signals _GXP and _GXN , but when averaged over time, they become sinusoidal voltages whose voltage values change in accordance with changes in the duty factor.

By changing the magnitude of the duty factor and the period of change in magnitude, i.e., by changing the magnitude and frequency of the X-phase voltage command _VX ^* , it is possible to control the magnitude (amplitude) and frequency of the three-phase AC voltage output from the power conversion device 20. This makes it possible to drive the permanent magnet synchronous motor 10 at a variable speed and control the torque.

Current sensor 30 detects three-phase motor currents flowing from power conversion device 20 to permanent magnet synchronous motor 10. In this embodiment, current sensor 30 detects U-phase motor current _IU and W-phase motor current _IW . V-phase motor current (referred to as " _IV ") is calculated in motor control device 40 based on the detected U-phase motor current _IU and W-phase motor current _IW ( _IV = _{-IU -IW} ). Note that a current transformer (CT), for example, is used as current sensor 30.

A Hall CT or a shunt resistor may be used as the current sensor 30. When a shunt resistor is used, the so-called one-shunt or three-shunt system is used. In the one-shunt system, the DC input current to the main circuit 131 (Figure 4), i.e., the DC bus current, is detected by a single shunt resistor (e.g., shunt resistor 135 in Figure 4), and the three-phase motor current is detected based on the detected value of the DC bus current. In the three-shunt system, a shunt resistor is provided in each of the three lower phase arms, and the three-phase motor current is detected based on the current values detected by these three shunt resistors.

Next, we will explain the motor control device 40 (Figure 1).

Motor control device 40 controls at least one of the speed and torque of permanent magnet synchronous motor 10 by controlling the switching operation of power conversion device 20. In doing so, motor control device 40 generates three-phase voltage commands _VU ^* , _VV ^* , _VW ^* based on speed command _ωr ^* or torque command Trq ^* and the motor current detected by current sensor 30. Note that speed command _ωr ^* or torque command Trq ^* may be set in motor control device 40, or may be provided to motor control device 40 from another control system such as a higher-level control system not shown in FIG. 1 .

Furthermore, motor control device 40 generates on/off control signals S _UP to S _WN consisting of PWM pulses according to the three-phase voltage commands V _U ^* , V _V ^* , and V _W ^* using PWM signal creation unit 43. Motor control device 40 controls the switching operation of power conversion device 20 using these on/off control signals S _UP to S _WN .

As mentioned above, the motor control device 40 uses so-called vector control. Therefore, the voltage command creation unit 41 (Figure 1) generates voltage command values through calculations in a rotating coordinate system.

Figure 6 is a functional block diagram showing the configuration of the voltage command creation unit 41.

The voltage command generator 41 receives the first d-axis current command _Id ^* and the first q-axis current command _Iq ^* calculated based on the speed command or the torque command ( _ωr ^* , Trq ^* in FIG. 1), the d-axis detected current _Idc and the q-axis detected current _Iqc , and the inverter frequency command _ω1 (ωr ^* may be used instead of _ω1 ), and generates the d-axis voltage command _Vd ^* and the q-axis voltage command _Vq ^* using equations (1) and (2). Note that _Idc and _Iqc are calculated by converting the three-phase motor currents detected by the current sensor 30 from stationary coordinates to rotating coordinates.

In equations (1) and (2), R is the winding resistance per phase of permanent magnet synchronous motor 10, _Ld is the d-axis inductance, _Lq is the q-axis inductance, and Ke is the induced voltage constant.

_Id ^** and _Iq ^** are the second d-axis current command and the second q-axis current command, respectively, generated by a current controller consisting of PI control so that the motor current flows following _Id ^* and _Iq ^* .

In the d-axis current controller 114a that generates _Id ^** , an adder/subtractor 91d calculates the difference ( _Id ^* _-Idc ) between the first d-axis current command _Id ^* and the d-axis detected current _Idc , and a proportional unit 92c multiplies the calculated difference by a proportional gain ( _{Kp_d} ). The calculated difference is also multiplied by an integral gain ( _{Ki_d} ) in the proportional unit 92d and integrated in the integrator 94c. The calculated difference multiplied by the proportional gain _{Kp_d} and the integral multiplied by the integral gain _{Ki_d} are added by an adder 90b to calculate _Id ^** .

In the q-axis current controller 114b that generates _Iq ^** , an adder/subtractor 91e calculates the difference ( _Iq ^* _-Iqc ) between the first q-axis current command _Iq ^* and the q-axis detected current _Iqc , and a proportional unit 92e multiplies the calculated difference by a proportional gain ( _{Kp_q} ). The calculated difference is also multiplied by an integral gain ( _{Ki_q} ) by a proportional unit 92f and integrated by an integrator 94d. An adder 90c adds the calculated difference multiplied by the proportional gain _{Kp_q} to the integral value multiplied by the integral gain _{Ki_q} to calculate _Iq ^** .

Id ^** is multiplied by R by a multiplier 92g to calculate "R× _Id ^** " on the right side of equation (1). Iq ^** is multiplied by R by a multiplier 92i to calculate "R× _Iq ^** " on the right side of equation (2).

_Iq ^** is filtered by a low-pass filter 98a to obtain _Iq ^** _{_fil} shown on the right-hand side of equation (1). Furthermore, _Iq ^** _{_fil} is multiplied by _Lq and _ω1 by a multiplier 92h to calculate " _ω1 × _Lq × _Iq ^** _{_fil} " on the right-hand side of equation (1). An adder-subtractor 91f subtracts " _ω1 × _Lq × _Iq ^** _{_fil} " calculated by the multiplier 92h from "R× _Id ^** " calculated by the multiplier 92g. This generates _Vd ^* .

_Id ^** is filtered by a low-pass filter 98b to obtain _Id ^** _{_fil} shown on the right-hand side of equation (2). Furthermore, _Id ^** _{_fil} is multiplied by _Ld and _ω1 by a multiplier 92j to calculate " _ω1 × _Ld × _Id ^** _{_fil} " on the right-hand side of equation (2). _ω1 is multiplied by Ke by a multiplier 92k to calculate " _ω1 ×Ke" in equation (2). An adder 90d adds "R× _Iq ^** " calculated by the multiplier 92i, " _ω1 × _Ld × _Id ^** _{_fil} " calculated by the multiplier 92j, and " _ω1 ×Ke" calculated by the multiplier 92k. This generates _Vq ^* .

As described above, in voltage command creation unit 41, d-axis current controller 114a and q-axis current controller 114b are cascade-connected to a voltage calculation unit based on equations (1) and (2). In addition, low-pass filters 98a and 98b, which are first-order lag filters having time constants ( _Tq (= _Lq /R), _Td (= _Ld /R)) corresponding to the motor constants of permanent magnet synchronous motor 10, are used in the voltage calculation unit. This makes it possible to achieve stable vector control even when there are restrictions on the calculation period.

The magnetic flux estimation function provided by the motor control device 40 in this embodiment 1 is described below.

Figure 7 is a functional block diagram showing the configuration of the magnetic flux estimation unit 50 (Figure 1) in this embodiment 1.

In addition to the magnetic flux estimation unit 50, Figure 7 also shows the magnetic flux table 51, stray voltage table 52, dead time compensation error calculation unit 53, and winding resistance estimation unit 54, which are related to magnetic flux estimation.

In this embodiment, when a three-phase AC voltage is output from the power conversion device 20 to the permanent magnet synchronous motor 10, the d-axis interlinkage magnetic flux _Ψd representing the magnet magnetic flux and the magnet temperature Temp_m are estimated based on the q-axis voltage command Vq ^* , the rotational angular velocity _ωe (electrical angle) of the permanent magnet synchronous motor 10, _the d-axis current Id and the q-axis current Iq, the DC voltage Edc of the DC voltage source 120 ( FIG. ₂ ), and the winding temperature Temp_w.

The q-axis voltage equation in the steady state of the permanent magnet synchronous motor 10 can be expressed by equations (3) and (4) if the differential term (sL _q I _q ^** ) is ignored. Note that Ψ _d0 in equation (4) is the d-axis flux linkage component due to only the permanent magnets.

The q-axis voltage drop V _{drop_q} in equation (3) is the difference (V _q ^* - ω _e × Ψ _d ) between the speed electromotive force (induced voltage: ω _e × Ψ _d ) and the q-axis voltage command value V _q ^* , and is the total voltage drop on the q axis including the voltage drop due to the winding resistance R, the voltage drop in the switching elements (22a to 22f in FIG. 4), the error voltage due to dead time, the error voltage due to the PWM method, etc. Based on equation (3), Ψ _d can be estimated by calculating V _{drop_q} in consideration of these multiple voltage components that make up V _{drop_q} .

First, the d-axis current _Id and the q-axis current _Iq are distributed in a grid pattern (( _Id , _Iq )) at two different rotation speeds of the permanent magnet synchronous motor 10, and the d-axis voltage and the q-axis voltage ( _Vd , _Vq ) relative to ( _Id , _Iq ) are measured. At this time, the magnet temperature Temp_m is set to a constant value ( _{Temp_m_calib} ), for example, a preset standard temperature.

If the q-axis voltage command values when the rotational angular velocities (electrical angles) of permanent magnet synchronous motor 10 are _ωe1 and _ωe2 are _Vq1 ^* and _Vq2 ^* , respectively, _Vq1 ^* and _Vq2 ^* can be expressed by equations (5) and (6), respectively, based on equation (3), where _ωe2 > _ωe1 .

Here, it is assumed that the d-axis interlinkage magnetic flux Ψ _d and the q-axis voltage drop V _{drop_q} do not depend on the rotation speed.

From equations (5) and (6), the d-axis interlinkage magnetic flux Ψ _d and the q-axis voltage drop V _{drop_q} are expressed by equations (7) and (8), respectively.

Using the above-mentioned measurement data ((V _d , V _q ) for (I _d , I _q )), the table data for Ψ _d and V _{drop_q} are calculated using equations (7) and (8) with (I _d , I _q ) as arguments. The table data for Ψ _d and V _{drop_q} are stored in the motor control device 40 as the magnetic flux table 51 (Ψ _{d_calib_table} ) and the q-axis voltage drop adaptation table (V _{drop_q_calib_table} ), respectively.

The q-axis voltage drop V _{drop_q} includes a voltage drop component caused by multiple factors. Based on the inventor's research, in this embodiment, in order to estimate the magnetic flux with high accuracy, the q-axis voltage drop component is corrected for the following three factors. The three factors are the dead time error time, the resistance component including the winding resistance, and the voltage drop component (stray voltage) remaining after subtracting the voltage drop component caused by the dead time error time and the resistance component from V _{drop_q} obtained from the q-axis voltage drop compatibility table.

Considering the above three factors, V _{drop_q} is expressed by equation (9).

The first term (RI _q ) on the right side of equation (9) is a voltage drop component due to resistance component R and is proportional to the d-axis current I _d . R in equation (9) is the sum of the winding resistance of the permanent magnet synchronous motor 10, the resistance components of the semiconductor switching elements (22 a to 22: FIG. 4), and the resistance components from the power conversion device 20 to the permanent magnet synchronous motor 10, such as the wiring cables (hereinafter referred to as the "total resistance value"). The second term (V _{dead_q} ) on the right side of equation (9) is a voltage drop component resulting from dead time compensation. The third term (V _{err_q} ) on the right side of equation (9) is called a stray error voltage, and includes a nonlinear component of the voltage drop of the semiconductor switching elements, a nonlinear component due to the switching of the semiconductor switching elements, and a component due to the leakage flux and structure of the permanent magnet synchronous motor 10. This value is specific to the combination of the power conversion device 20 and the permanent magnet synchronous motor 10.

V _{dead_q} will be explained below, but first, dead time will be briefly explained.

Although not shown or described in FIG. 5 , a period during which the semiconductor switching elements of both arms are turned off, i.e., a so-called dead time, is set in drive signal G _XP that drives the semiconductor switching element of the X-phase upper arm and drive signal G _XN that drives the semiconductor switching element of the X-phase lower arm, in order to prevent a short circuit between the upper and lower arms.

In this embodiment, the PWM signal generator 43 (FIG. 1) sets dead times in the on/off control signals S _UP to S _WN .

When a dead time is set, an error (dead time error voltage) occurs between the voltage (voltage command) to be applied to permanent magnet synchronous motor 10 and the voltage actually applied to permanent magnet synchronous motor 10. This causes waveform distortion of the motor current. Therefore, in this embodiment, dead time compensator 42 (FIG. 1) converts _Vd ^* and _Vq ^* into three-phase AC voltage commands _VU ^* , _VV ^* , _VW ^* , and further compensates _VU ^* , _VV ^* , _VW ^* for the dead time error voltage in order to reduce waveform distortion of the motor current due to the dead time.

Various techniques are known for dead time compensation, which compensates for dead time error voltages. For example, the polarity of the three-phase current is determined from the current commands on the d and q axes, and the three-phase AC voltage commands are corrected based on this three-phase current polarity.

In this embodiment, the dead time compensation unit 42 calculates the dead time error voltage using a predetermined formula based on the DC power supply voltage, switching frequency, dead time, and current phase of the power conversion device 20.

In this embodiment, the dead time compensation error calculation unit 53 (Figure 1) calculates the voltage drop component that occurs due to dead time compensation (hereinafter referred to as the dead time compensation error voltage).

The dead time compensation error calculation unit 53 calculates the dead time compensation error voltage V _{dead_q} appearing on the q axis using equations (10) and (11).

In equations (10) and (11), Edc is the DC voltage of DC voltage source 120 (FIG. 2), _Tsw is the switching period, β is the current phase angle based on the d-axis, _{Tdead_real} is the actual dead time, and _{Tdead_cmp} is the dead time compensation time (equivalent to the dead time (set value)).

The actual dead time T _{dead_real} and the dead time compensation time T _{dead_cmp} are often set to the same value. However, they may also be set to different values. For example, when the motor inductance is small, it may be desirable to suppress distortion of the motor current. In such cases, the dead time compensation time is set to a value that minimizes distortion of the motor current. Because current distortion near the current zero crossing tends to increase when the dead time compensation amount is increased, the dead time compensation time may be set to a small value different from the actual dead time. In such cases, a dead time compensation error voltage V _{dead_q} is generated as shown in equations (10) and (11). The dead time compensation error voltage V _{dead_q} represents the error voltage resulting from overcompensation or undercompensation when dead time compensation is performed.

In this embodiment, magnetic flux estimation can be performed with high accuracy even when the actual dead time set by the PWM signal creation unit 43 and gate drive circuit 123 differs from the dead time compensation time compensated for by the dead time compensation unit 42.

Here, an example of calculating the q-axis voltage drop V _{drop_q} will be described with reference to FIGS. 8 and 9. FIG.

Fig. 8 is a calculation example of the q-axis voltage drop V _{drop_q} shown as a contour map on the I _d -I _q plane. Fig. 9 is a graph showing the relationship between I _q and V _{drop_q} shown in Fig. 8 using I _d as a parameter.

8 shows that the absolute value of V _{drop_q} tends to decrease as _Id increases in the negative direction for a given _Iq . In other words, as _Id increases in the negative direction, V _{drop_q} tends to decrease when _Iq is positive and increase when _Iq is negative. Also, FIG. 9 shows that the change _{in V drop_q relative} to _Iq differs depending on the value of _Id .

Based on equations (10) and (11), the difference in the change in V _{drop_q} relative to _Iq is due to the existence of the dead-time compensation error voltage V _{dead_q} , which is a function of the current phase angle β. On the other hand, when the actual dead time and the dead-time compensation time are the same, the q-axis voltage drop V _{drop_q} is dominated by the voltage drop component due to the resistance component R in the first term on the right-hand side of equation (9). In this case, the value of V _{drop_q} is determined by _Iq , regardless of _Id .

In order to calculate the dead time error time from the q-axis voltage drop V _{drop_q} that has been calculated in advance and stored as table data as described above, a second q-axis voltage drop V _{drop_q} ′ is defined by subtracting the dead time compensation error voltage V _{dead_q} from the q-axis voltage drop V _{drop_q} as shown in equation (12).

To calculate the dead time error time when calculating the table data for V _{drop_q} , the dead time compensation error time T _{dead_calib} is changed to find the dead time compensation error time T _{dead_calib} that minimizes the _Id dependency of the second q-axis voltage drop V _{drop_q} '. In other words, the T _{dead_calib} is found so that the multiple graphs shown in FIG. 9 , using _Id as a parameter, overlap the most. While various known optimization calculations can be used, in this embodiment, the dead time compensation error time T _{dead_calib} is found so that the total sum of variances is minimized. The population mean μ and variance V of the q-axis voltage drop V _{drop_q} when calculating the table data for V _{drop_q} are found using the following equations (13) and (14):

In equations (13) and (14), N _Id is a data group in the _Id direction for certain _Iq data. The dispersion corresponds to the variation in V _{drop — q} on the I _q -V _{drop — q} plane shown in FIG.

Furthermore, the sum S of the variances of the second q-axis voltage drops V _{drop — q} ′ at each I _q data point is calculated using equation (15).

When the sum S expressed by the equation (15) is minimized, the dependency (variation) of the second q-axis voltage drop V _{drop —} q ′ on _Id is assumed to be small, and the dead time compensation error time T _{dead — calib} is calculated.

In the measurement example of the q-axis voltage drop V _{drop_q} shown in Figures 8 and 9, the dead time compensation error time T _{dead_calib} is changed (for example, in increments of 0.1 μs) to determine the dead time compensation error time T _{dead_calib} that minimizes the sum S. The second q-axis voltage drop V _{drop_q} ' is shown in Figures 10 and 11.

Fig. 10 is a calculation example of the second q-axis voltage drop V _{drop_q} ', which is shown as a contour map on the I _d -I _q plane. Fig. 11 is a graph showing the relationship between I _q and V _{drop_q} ' shown in Fig. 10, with I _d as a parameter.

Comparing FIG. 10 with FIG. 8, the contour lines of the second q-axis voltage drop _{Vdrop_q} ' are parallel to the d-axis. In other words, for a given _Iq , the second q-axis voltage drop _{Vdrop_q} ' is constant and hardly depends on _Id . Comparing FIG. 11 with FIG. 9, multiple graphs showing the change in _{Vdrop_q} ' relative to _Iq , with _Id as a parameter, are overlapped. In this way, the dead time error time _{Tdead_calib} can be calculated when calculating the table data for _{Vdrop_q} .

Next, we will explain the voltage drop component due to the resistance component.

In this embodiment, the resistance component R in the above-mentioned equation (9) is calculated from the slope of the graph in Fig. 11. In this embodiment, the average value of V _{drop_q} ' is calculated for each _Iq data point, and the average value is linearly approximated by the least squares method to find the slope, which is used as the total resistance R _calib when calculating the table data for V _{drop_q} .

The total resistance R _calib may be calculated using various known optimization calculations.

R _calib includes winding resistance _Rw , whose resistance value changes depending on winding temperature Temp_w, resistance components of the switching elements, and resistance components from power conversion device 20 to permanent magnet synchronous motor 10, such as distribution cables. Therefore, winding resistance estimation unit 54 ( FIG. 1 ) sets the winding temperature when the table data for V _{drop_q} is calculated as Temp_w _{_calib} , and uses table data for winding resistance _Rw relative to the winding temperature (winding resistance table R _w _table) to calculate a constant resistance component R _{calib_const} that is independent of winding temperature Temp_w _{_calib} when the table data for V _{drop_q} is calculated, using equation (16).

The winding resistance _Rw may be calculated using a known formula.

Next, we will explain voltage drop components other than the dead-time error voltage and voltage drop due to resistance components, i.e., voltage drop components called stray voltages.

The voltage drop component remaining after subtracting the voltage drop component due to the dead-time compensation error voltage V _{dead_q} and the total resistance R _calib from the q-axis voltage drop V _{drop_q} when calculating the table data for V _{drop_q} is defined as a third q-axis voltage drop V _{drop_q} '' by equation (17). Note that V _{drop_q} ' in the first term on the right-hand side of equation (17) is the above-mentioned second q-axis voltage drop (equation (12)).

FIG. 12 is a calculation example of the third q-axis voltage drop V _{drop — q} ″, which is shown as a contour map on the I _d -I _q plane.

In calculating V _{drop_q} '', the calculation example of V _{drop_q} shown above (FIG. 8) is used. Therefore, _table data (V _{err_q_calib_table} ) of V _{drop_q} '' is calculated using (I _d , I _q ) as arguments. The table data of _{V drop_q} '' is stored in the motor control device 40 as the stray voltage table 52 (FIGS. 1 and 7).

As can be seen from FIG. 12, V _{drop — q} ″ is the remaining voltage drop component after subtracting the dead-time compensation error voltage V _{dead — q} and the voltage drop component (R _calib I _d ) due to the total resistance R _calib from the q-axis voltage drop V drop — q, and therefore has a very small value.

According to the study of the present inventors, it is difficult to identify specific factors and express V _{drop — q} ″ in a calculation formula. Therefore, in this embodiment, V _{drop — q} ″ is referred to as a stray voltage.

According to the inventors' research, stray voltage is related to the leakage flux and structure of the permanent magnet synchronous motor 10, and is a value specific to the combination of the power conversion device 20 and the permanent magnet synchronous motor 10. Therefore, the stray voltage can be calculated with high accuracy using a stray voltage table.

It is preferable to create a stray voltage table for each motor control device, but it is also possible to create a stray voltage table for a representative combination of power conversion device 20 and permanent magnet synchronous motor 10 and share it among each motor control device.

Next, a method for calculating the q-axis voltage drop V drop_q_est by correcting V _{drop_q} , which has been calculated and stored in advance under predetermined operating conditions, in accordance with the operating conditions (DC voltage Edc, winding temperature, switching _frequency , etc.) when permanent magnet synchronous motor 10 is driven will be described.

FIG. 13 is a functional block diagram showing the configuration of a calculation unit for the q-axis voltage drop V _{drop_q_est} in the motor control device 40.

The total resistance R _est when the permanent magnet synchronous motor 10 is driven is calculated by the formula (18).

In equation (18), T _{w_est} is the winding temperature (Temp_w in FIG. 13) when the motor is driven (when magnetic flux is estimated), and R _{calib_const} is the constant resistance component described above (equation (16)). _{R w_table} (T _{w_est} ) is the table data for the winding temperature of the winding resistance R _w described above, and is stored in winding temperature table 541 ( FIG. 13 ).

The estimated total resistance R _est reflects the dependence of the winding temperature on the value of the voltage drop due to the resistance component.

Next, the dead time compensation error voltage V dead_q_calib when permanent magnet synchronous motor 10 is driven (when magnetic flux is _estimated ) is calculated by equation (19) using switching period T _{sw_est} , DC voltage E _{dc_est} , and current phase angle β _est when permanent magnet synchronous motor 10 is driven (when magnetic flux is estimated). Current phase angle β _est is calculated by current phase calculator 531 ( FIG. 13 ) based on I _d and I _q . The calculation using equation (19) corresponds to the operation of calculator 532.

Note that equation (19) corresponds to equation (10) above, where T _sw , E _dc and β _est on the right side are replaced with T _{sw — est} , E _{dc — est} and β _est , respectively.

By _{Vdead_q_calib} , differences in operating conditions when permanent magnet synchronous motor 10 is driven (when magnetic flux is estimated) are dynamically reflected in the value of the dead time error voltage.

Furthermore, by using the d-axis current I _{d_est} and the q-axis current I _{q_est} when the permanent magnet synchronous motor 10 is driven (when the magnetic flux is estimated), and referring to the stray voltage table 52 (V _{err_q_calib_table} (I _d , I _q )), the stray error voltage V _{err_q_est} when the permanent magnet synchronous motor 10 is driven (when the magnetic flux is estimated) is calculated using equation (20).

Using the above-described R _est , V _{dead_q_calib} , and V _{err_q_est} , the q-axis voltage drop V _{drop_q_est} when permanent magnet synchronous motor 10 is driven (when magnetic flux is estimated) is calculated by equation (21).

The calculation using equation (21) corresponds to the operation of the calculation unit 501 (Figure 13).

Based on the q-axis voltage drop consisting of multiple voltage drop components calculated for each factor as described above, the magnetic flux estimation unit 50 (Figure 7) performs magnetic flux estimation using equation (22).

In equation (22), V _{q — est} ^* is the q-axis voltage command when permanent magnet synchronous motor 10 is driven (when magnetic flux is estimated), and ω _{e — est} is the rotational angular velocity ω _e (electrical angle) when permanent magnet synchronous motor 10 is driven (when magnetic flux is estimated).

Note that the calculation using equation (22) corresponds to the operation of the calculation unit 55 in the magnetic flux estimation unit 50 (Figure 7).

Furthermore, the magnetic flux estimation unit 50 uses _{Id_est} and _{Iq_est} to calculate the magnetic flux _{Ψd_calib} when calculating the table data for _{Vdrop_q} under the same current conditions ( _Id = _{Id_est} , _Iq = _{Iq_est} ) according to equation (23). The right side of equation (23) represents data when the arguments ( _Id , _Iq ) are ( _{Id_est} , _{Iq_est} ) in the table data _{Ψd_calib_table} ( _Id , _Iq ) in the pre-stored magnetic flux table 51 (FIG. 1).

Furthermore, the magnetic flux estimator 50 calculates the d-axis magnetic flux change _ΔΨd from the time when the table data of V _{drop_q} was calculated during driving (flux estimation) of the permanent magnet synchronous motor 10, using equation (24). Note that the d-axis and q-axis currents used in the calculation may be detected currents or current command values.

The d-axis magnetic flux change amount ΔΨ _d indicates the change in the magnet magnetic flux due to fluctuations in magnet temperature. If the d-axis inductance (L _d ) is ignored because it is almost constant regardless of magnet temperature, the ratio of Ψ _{d_est} to Ψ _{d_calib} , i.e., the d-axis magnetic flux change rate, is calculated as in equation (25) when the normal d-axis current Id is controlled to be zero. Note that, as with equation (24), the magnetic flux table 51 (FIGS. 1 and 7) is referenced.

The denominator of the fraction on the right side of equation (25) corresponds to the d-axis magnetic flux change amount ΔΨ _d calculated by equation (24).

The calculated d-axis magnetic flux change rate is input to magnet temperature estimation unit 56 (FIG. 7) and compared with a pre-stored temperature-no-load induced voltage table. In this embodiment, the temperature-no-load induced voltage table indicates the ratio of the no-load induced voltage at magnet temperature Temp_m to the no-load induced voltage at the standard temperature when the table data for V _{drop_q} was calculated, i.e., the relationship between the no-load induced voltage change rate and magnet temperature Temp_m.

The magnet temperature estimation unit 56 outputs, as the result of the comparison, the temperature that indicates the no-load induced voltage change rate that matches the d-axis magnetic flux change rate calculated using equation (25), as the estimated magnet temperature Temp_m.

According to the first embodiment described above, voltage drop components are calculated for each cause: voltage drop components due to total resistance components including the resistance of wiring between the permanent magnet synchronous motor 10 and the power conversion device and semiconductor switching elements; voltage drop components due to dead time compensation; and voltage drop components due to stray voltage.The magnetic flux of the permanent magnet is estimated based on these voltage drop components, i.e., based on the difference between the voltage command value and the sum of these voltage drop components.This improves the accuracy of magnetic flux estimation.

Furthermore, in the first embodiment, the q-axis voltage drop Vdrop_q is calculated in advance from the voltage command for the motor current ( _Id , _Iq ) under predetermined operating conditions ( _ωe1 , _ωe2 in the first embodiment) of the permanent magnet synchronous motor 10, and is stored in the motor control device 40 as table data with the motor _current as an argument. Based on this table data and the calculated value of the voltage drop component due to dead time compensation, table data for the total resistance component and stray voltage is calculated. This reduces the amount of table data that needs to be collected in advance by operating the actual machine for magnetic flux estimation. This avoids an increase in the workload of creating the table data.

Furthermore, by storing stray voltage, a voltage drop component that is difficult to express in a formula, as table data in advance, calculation accuracy of the voltage drop component can be ensured even under driving conditions where the voltage applied to the permanent magnet synchronous motor is small, such as low speed or low torque driving conditions. This improves the accuracy of magnetic flux estimation and magnet temperature estimation.

In the first embodiment, as shown in FIG. 1, the functional components related to magnetic flux estimation, namely the magnetic flux estimation unit 50, magnetic flux table 51, stray voltage table 52, dead time compensation error calculation unit 53, and winding resistance estimation unit 54, are built into the motor control device 40. However, this is not limiting and the magnetic flux estimation device may be configured as a separate unit from the motor control device 40. In this case, the voltage command and current detection value (or current command value) are input to the magnetic flux estimation device via wired or wireless communication. This allows the magnetic flux estimation function to be retrofitted to the motor control device. It also increases the degree of freedom in the placement of the motor control device 40.

As described above, this embodiment also includes a voltage command value calculation unit that calculates a voltage command value based on the motor's torque command, a dead time compensation unit that performs dead time compensation based on the voltage command value, a dead time compensation error calculation unit that calculates a dead time compensation error based on the motor's detected current value and DC voltage detection value, a winding resistance calculation unit that calculates the winding's electrical resistance value based on the winding's detected temperature value, and a magnetic flux estimation unit that estimates the magnet's magnetic flux from the stray voltage, dead time compensation error, and electrical resistance value obtained from the motor's detected current value. This reduces errors in the voltage information itself used when performing magnetic flux estimation, improving the accuracy of magnetic flux estimation.

When the temperature of the permanent magnets used in permanent magnet synchronous motors rises, demagnetization occurs, making it difficult to control the rotational speed and torque. Furthermore, to ensure the torque generated by the permanent magnet synchronous motor, the motor current and the current flowing through the power conversion device may increase.

Therefore, in this second embodiment, the speed command and torque command are limited based on the estimated magnet magnetic flux and magnet temperature.

Figure 14 is a functional block diagram showing the general configuration of a motor drive system according to a second embodiment of the present invention. Below, we will mainly explain the differences from the first embodiment.

The motor control device 40a in the second embodiment has a command value correction unit 48 that corrects the speed command _ωr ^* and the torque command Trq ^* based on the magnet flux (d-axis interlinkage magnetic flux _Ψd ) estimated by the magnetic flux estimation unit 50 or the magnet temperature Temp_m.

Figure 15 is a functional block diagram showing an example configuration of the command value correction unit 48 (Figure 14).

The command value correction unit 48 includes a command value limiting unit 481 and a limit value calculation unit 482.

When the input speed command _ωr ^* or torque command Trq ^* exceeds a predetermined value, the command value limiting unit 481 limits _ωr ^* or Trq ^* to a limit value Lmt and outputs the corrected speed command _ωr ^** or torque command Trq ^** .

The limit value calculation unit 482 compares the magnet flux (d-axis interlinkage magnetic flux Ψ _d ) or magnet temperature Temp_m estimated by the magnetic flux estimation unit 50 with a judgment threshold value (Th1, Th2: Th1<Th2), and sets a limit value Lmt in the command value limit unit 481 depending on the magnitude relationship.

In the second embodiment, as shown in Fig. 15, when _ψd or Temp_m exceeds a determination threshold Th1, the limit value Lmt is decreased linearly or quadratically and output. When _ψd or Temp_m exceeds a determination threshold Th2, it is determined that the magnet temperature is too high, and the rate of decrease of the command limit value is increased.

The input to the limit value calculation unit 482 may be the rate of decrease of the input Ψ _d or Temp_m from a preset reference value.

According to the second embodiment described above, it is possible to suppress demagnetization of permanent magnets and increases in current flowing through permanent magnet synchronous motors and power conversion devices. This improves the reliability of motor control devices and motor drive systems.

In addition, the current command may be adjusted instead of the speed command or torque command.

Figure 16 is a functional block diagram showing the general configuration of a motor control device 40b in a motor drive system according to a third embodiment of the present invention. Below, we will mainly explain the differences from the second embodiment.

In this third embodiment, the motor control device 40b has a protection signal generation unit 49.

The protection signal generator 49 compares the value of the magnet magnetic flux (d-axis interlinkage magnetic flux Ψ _d ) or magnet temperature Temp_m estimated by the magnetic flux estimation unit 50 with a predetermined threshold, and if it determines that the value of Ψ _d or Temp_m exceeds or falls below the threshold, it outputs a protection signal to the outside of the motor control device 40 b.

When another control device (not shown) receives the protection signal, it reduces the speed command or torque command of the permanent magnet synchronous motor 10 or stops the permanent magnet synchronous motor 10. This improves the reliability of the motor drive system (100 in Figure 1).

Note that the functions of such other control devices may also be provided by the motor control device 40.

Figure 17 is a functional block diagram showing the configuration of an electric vehicle system 300 according to a fourth embodiment of the present invention.

The motor control device 40 controls the AC power supplied from the power conversion device 20 (inverter) to the permanent magnet synchronous motor 10. A DC voltage source 120 (e.g., a battery) supplies DC power to the power conversion device 20 (inverter). The power conversion device 20 (inverter) is controlled by the motor control device 40 to convert the DC power from the DC voltage source 120 into AC power. Any of the motor control devices described in Examples 1 to 3 above is applied as the motor control device 40. Therefore, the motor control device 40 has the function of estimating the magnet magnetic flux and magnet temperature of the permanent magnet synchronous motor 10.

The permanent magnet synchronous motor 10 is mechanically connected to the transmission 301. The transmission 301 is mechanically connected to the drive shaft 305 via the differential gear 303, and supplies mechanical power to the wheels 307. This drives the wheels 307 to rotate.

The permanent magnet synchronous motor 10 may be connected directly to the differential gear 303 without going through the transmission 301. In the case of an automobile, the front and rear wheels may each be driven by an independent permanent magnet synchronous motor and power conversion device (inverter).

According to this embodiment, the accuracy of magnetic flux estimation by the motor control device 40 is improved, allowing the rotation speed or torque of the permanent magnet synchronous motor 10 to be controlled with high precision. This improves the ride comfort of the electric vehicle system.

Furthermore, according to this embodiment, it is possible to estimate the magnet temperature, thereby preventing abnormalities in the electric vehicle system 300 due to thermal demagnetization of the permanent magnets of the permanent magnet synchronous motor 10.

Furthermore, according to this embodiment, the motor control device 40 can estimate the magnet magnetic flux and magnet temperature using information from the electric vehicle system 300. For example, instead of the winding temperature Temp_w (Figure 7), the ambient temperature near the permanent magnet synchronous motor 10 or the power conversion device 20 (inverter) can be used.

The present invention is not limited to the above-described embodiments, but includes various modifications. For example, the above-described examples have been described in detail to clearly explain the present invention, and are not necessarily limited to those that include all of the described configurations. Furthermore, it is possible to add, delete, or replace part of the configuration of each embodiment with other configurations.

For example, some or all of the above-mentioned configurations, functions, processing units, processing procedures, etc. may be realized in hardware, for example by designing them as integrated circuits. Furthermore, the above-mentioned configurations and functions, etc. may be realized in software by a processor interpreting and executing a program that realizes each function.

Furthermore, the signal lines and information lines shown are those considered necessary for the explanation, and do not necessarily represent all control lines and information lines. Furthermore, the above-mentioned configurations, functions, processing units, processing procedures, etc. do not necessarily need to be located in the same physical location; for example, some may be located on a network or in the cloud via communications, etc.

Furthermore, the permanent magnet synchronous motor may be either an internal rotor type (Figure 2) or an external rotor type, and may be either a radial gap type or an axial gap type.

10... Permanent magnet synchronous motor 11... Stator 12... Rotor 13... Rotor core 15... Winding 16... Stator core 17... Permanent magnet 18... Stator magnetic pole 20... Power conversion device 22a, 22b, 22c, 22d, 22e, 22f... Semiconductor switching element 30... Current sensor 40, 40a, 40b... Motor control device 41... Voltage command creation unit 42... Dead time compensation unit 43... PWM signal creation unit 48... Command value correction unit 49... Protection signal generation unit 50... Magnetic flux estimation unit 51... Magnetic flux table 52... Stray voltage table 53... Dead time compensation error calculation unit 54... Winding resistance estimation unit 55... Calculation unit 56... Magnet temperature estimation unit 90b, 90c, 90d... Adder 91d , 91e, 91f...adder-subtractors 92c, 92d, 92e, 92f...proportional units 92g, 92h, 92i, 92j, 92k...multipliers 94c, 94d...integrators 98a, 98b...low-pass filter 100...motor drive system 114a...d-axis current controller 114b...q-axis current controller 120...DC voltage source 123...gate drive circuit 131...main circuit 135...shunt resistor 300...electric vehicle system 301...transmission 303...differential gear 305...drive shaft 307...wheel 481...command value limiting unit 482...limiting value calculation unit 501...calculation unit 531...current phase calculator 532...calculation unit 541...winding temperature table

Claims (11)

a voltage command generation unit that generates a voltage command for controlling an output voltage of a power conversion device to a motor having a rotor with a permanent magnet in accordance with a speed command or a torque command of the motor;
a dead time compensation unit that performs dead time compensation on the voltage command;
a magnetic flux estimation unit that estimates the magnetic flux of the permanent magnet;
In a motor control device comprising:
The magnetic flux estimation unit
estimating a magnetic flux of the permanent magnet based on the voltage command and voltage drops occurring between the motor and the power conversion device, and between the motor and the power conversion device;
The voltage drop is
a first voltage drop component due to a resistance component between the motor and the power conversion device, and between the motor and the power conversion device;
a second voltage drop component caused by the dead time compensation; and
a third voltage drop component obtained by subtracting the first voltage drop component and the second voltage drop component from the voltage drop; and
A motor control device comprising: 2. The motor control device according to claim 1,
a table data indicating a relationship between the voltage drop and the motor current at a predetermined rotation speed of the motor;
A motor control device comprising: a motor control circuit for calculating the resistance component based on the table data and the second voltage drop component; 2. The motor control device according to claim 1,
A motor control device comprising: a motor control circuit for calculating the second voltage drop component based on a switching period, a DC voltage of a DC voltage source provided in the power conversion device, and a current phase angle. 2. The motor control device according to claim 1,
a first table data indicating a relationship between the voltage drop and the motor current at a predetermined rotation speed of the motor;
calculating second table data representing a relationship between the third voltage drop component and a motor current based on the table data and the second voltage drop component;
a motor control device that calculates the third voltage drop component based on the second table data; 2. The motor control device according to claim 1,
The motor control device according to claim 1, wherein the magnetic flux estimation unit estimates a temperature of the permanent magnet based on the estimated magnetic flux. 2. The motor control device according to claim 1,
A motor control device comprising: a motor control section for adjusting the speed command or the torque command based on the magnetic flux estimated by the magnetic flux estimation section; 6. The motor control device according to claim 5,
A motor control device comprising: a motor control section for adjusting the speed command or the torque command based on the temperature estimated by the magnetic flux estimation section; 8. The motor control device according to claim 6 or 7,
A motor control device characterized in that, when the speed command or the torque command exceeds a predetermined threshold, the speed command or the torque command is limited to a predetermined value. 2. The motor control device according to claim 1,
A motor control device characterized in that it issues a signal to an external device when the magnetic flux estimated by the magnetic flux estimation unit exceeds or falls below a predetermined value. 6. The motor control device according to claim 5,
A motor control device characterized in that it issues a signal to an external device when the temperature estimated by the magnetic flux estimation unit exceeds or falls below a predetermined value. Wheels and
a motor that drives the wheels;
a power converter that supplies power to the motor;
a control device that controls the power conversion device;
In an electric vehicle system comprising:
6. An electric vehicle system, wherein the control device is the motor control device according to claim 1 or 5.