JP7824191B2 - DC/DC converter and charging device - Google Patents

Description

The present invention relates to a three-phase LLC (including three-phase CLLC) DC/DC converter and charging device.

In recent years, as the capacity of batteries installed in electric vehicles has increased, the capacity of quick chargers has also increased, with the mainstream quick charger now ranging from 50 kW to 200 kW. The power supply section of a quick charger consists of multiple power supply units, each of which is required to have an output of approximately 12.5 kW to 30 kW. Furthermore, there is a demand for compactness, high efficiency, and low cost.

A power supply unit may consist of, for example, an AC/DC converter, a DC/DC converter, and an output filter. The AC/DC converter is placed before the DC/DC converter, and the output filter is placed after the DC/DC converter. Furthermore, the DC/DC converter is a current-resonant DC/DC converter, which can be made smaller, more efficient, and less expensive, despite the complexity of controlling fluctuations in input and output voltages, and fluctuations in the load at the output end.

A known current-resonant DC/DC converter is a three-phase LLC DC/DC converter with three-phase drive (see, for example, Patent Document 1). Although a three-phase LLC DC/DC converter requires a greater number of components due to its three-phase drive, it is more suited to higher output than a single-phase full-bridge converter because the current is distributed across three phases. Furthermore, the ripple in the output waveform is reduced to about one-tenth of that of a single-phase full-bridge converter, enabling the components that make up the output filter in the subsequent stage (large components such as electrolytic capacitors, film capacitors, and choke coils) to be smaller, lighter, and less expensive. Furthermore, it is attracting attention because it also improves power conversion efficiency.

Figure 9 shows the main circuit of a conventional three-phase LLC DC/DC converter 30A. The main circuit of DC/DC converter 30A is composed of high-frequency isolation transformers (hereinafter referred to as "transformers") T10 to T30, a primary-side switching circuit including switching elements Q1 to Q6, a secondary-side rectifier circuit including rectifier diodes D1 to D6 and a smoothing capacitor Co, and a primary-side resonant circuit.

The primary resonant circuit comprises a three-phase LLC resonant circuit consisting of resonant coils L1 to L3, the excitation coils (not shown) of transformers T10 to T30, the primary coils of transformers T10 to T30, and resonant capacitors C1 to C3. The primary coils of transformers T10 to T30 are connected in a Y-connection (star connection), with the other end of the primary coils connected in a balanced connection with the neutral point at point P. Similarly, the secondary coils of transformers T10 to T30 are connected in a Y-connection, with the other end of the secondary coils connected in a balanced connection with the neutral point at point S.

The DC/DC converter 30A can easily achieve high output because it shares power among three phases. However, variations in the characteristics of the three components that contribute to resonance (resonant coils L1-L3, transformers T10-T30, and resonant capacitors C1-C3) can cause the resonant currents of each phase to become unbalanced. When the resonant currents become unbalanced, for example, differences in the peak values of the three resonant currents can occur, causing problems such as heat generation in certain resonant coils or transformers or increased ripple in the output waveform output from the secondary-side rectifier circuit.

Increasing the drive frequency of switching elements Q1-Q6 is an effective way to reduce the size of DC/DC converter 30A. However, consideration must be given to the switching losses of switching elements Q1-Q6, core losses (iron losses) of transformers T10-T30 and resonant coils L1-L3, and losses due to the skin effect and proximity effect of the windings (copper losses). Wide-bandgap semiconductors such as SiC (silicon carbide) and GaN (gallium nitride), which have short switching times and low on-state losses, can be used for switching elements Q1-Q6. On the other hand, for magnetic components such as transformers T10-T30 and resonant coils L1-L3, increasing the drive frequency of switching elements Q1-Q6 above 100 kHz increases core loss, resulting in heat generation problems. Furthermore, while it becomes necessary to use expensive Litz wire to address copper loss in the windings, this approach only provides limited benefits. For this reason, the drive frequency cannot be increased indiscriminately.

As mentioned above, miniaturization of the DC/DC converter 30A is limited by magnetic components such as the transformers T10-T30 and the resonant coils L1-L3. In particular, if the output of the DC/DC converter 30A exceeds 10 kW, and each phase's transformers T10-T30 are configured with a single transformer, as shown in Figure 9, the volume and weight of the transformers T10-T30 will be larger than the other components, resulting in a larger product size and higher costs.

Figure 10 shows a three-phase LLC DC/DC converter 30B in which each phase is composed of two transformers (T11 and T12, T21 and T22, T31 and T32). Paralleling the transformers for each phase reduces the current flowing through the transformer, making it possible to miniaturize the transformer. This reduces the component height of DC/DC converter 30B, allowing for a more compact product.

In DC/DC converter 30B, the primary coils of transformers T11 to T32 are connected in parallel, and the secondary coils are connected in series. By connecting the secondary coils of transformers T11 to T32 in series, the secondary currents of transformers T11 to T32 are equalized (made identical). This reduces ripple in the output waveform of DC/DC converter 30B, even if there is variation in the characteristics of the components that contribute to resonance (resonance coils L1 to L3, transformers T11 to T32, and resonant capacitors C1 to C3). However, compared to the configuration in Figure 9, DC/DC converter 30B doubles the secondary current of transformers T11 to T32, increasing the amount of heat generated by transformers T11 to T32, making it difficult to achieve high output.

Figure 11 shows a three-phase LLC DC/DC converter 30C in which the primary coils of transformers T11 to T32 are connected in parallel, and the secondary coils are also connected in parallel. Connecting the secondary coils of transformers T11 to T32 in parallel reduces the current on both the primary and secondary sides of transformers T11 to T32, but if there is variation in the characteristics of the components that contribute to resonance, this effect is directly reflected in the output, resulting in the problem of larger ripples in the output waveform.

Non-Patent Document 1 proposes a method for a three-phase LLC DC/DC converter in which the drive frequency of each phase is the same, but the phase difference between the input voltages of each phase input from the primary switching circuit to the LLC resonant circuit is changed from 120° to equalize the resonant current of each phase and reduce ripple in the output waveform. However, this method requires current sensors to detect the resonant current of each phase, which increases the number of parts and leads to higher costs, and the need to change the phase difference between the input voltages of each phase depending on the operating state of the DC/DC converter, which leads to problems such as complicated control.

Chinese Utility Model No. 212935783

'Unbalanced Three-Phase LLC Resonant Converters: Analysis and Trigonometric Current Balancing'. IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 34, NO. 3, MARCH 2019. Retrieved from the Internet: <URL: https://ieeexplore.ieee.org/document/8379448>.

The present invention was made in consideration of the above circumstances, and its objective is to provide a three-phase LLC DC/DC converter and charging device that can handle high output and reduce output waveform ripple without increasing costs or complicating control.

In order to solve the above problem, a DC/DC converter according to a first aspect of the present invention comprises:
a transformer circuit including a first transformer unit, a second transformer unit, and a third transformer unit, each of which is configured with N transformers (N is an integer of 2 or more);
a primary-side switching circuit provided on a primary side of the transformer circuit, the primary-side switching circuit including a first leg, a second leg, and a third leg connected in parallel, each leg including a set of switching elements connected in series;
a primary-side resonant circuit including a first resonant unit provided between the first leg and the first transformer unit, a second resonant unit provided between the second leg and the second transformer unit, and a third resonant unit provided between the third leg and the third transformer unit, each resonant unit including a resonant coil and a resonant capacitor;
a secondary-side rectifier circuit including a plurality of rectifier elements and provided on the secondary side of the transformer circuit;
a control unit that provides a predetermined phase difference between the first leg, the second leg, and the third leg and drives the first leg at a predetermined drive frequency to generate current resonance;
A three-phase LLC DC/DC converter comprising:
The transformer circuit forms N transformer groups,
Each of the N transformer groups is formed as a group consisting of one transformer from each of the transformer sections, and is characterized in that the primary coils are connected in a balanced connection and the secondary coils are connected in a balanced connection.

In the DC/DC converter,
the control unit includes a storage unit,
The storage unit includes:
a first phase difference related to the phase difference;
With respect to the phase difference, a second phase difference different from the first phase difference;
a first threshold value related to an output value of an output current and/or an output power output from the secondary-side rectifier circuit,
The control unit
The output value can be compared with the first threshold value, and if the output value is less than the first threshold value, the phase difference is set to the first phase difference, while if the output value is equal to or greater than the first threshold value, the phase difference is set to the second phase difference.

In order to solve the above problem, a DC/DC converter according to a second aspect of the present invention comprises:
a transformer circuit including a first transformer unit, a second transformer unit, and a third transformer unit, each of which is configured with N transformers (N is an integer of 2 or more);
a primary-side switching circuit provided on a primary side of the transformer circuit, the primary-side switching circuit including a first leg, a second leg, and a third leg connected in parallel, each leg including a set of switching elements connected in series;
a secondary-side switching circuit provided on a secondary side of the transformer circuit, the secondary-side switching circuit including a fourth leg, a fifth leg, and a sixth leg connected in parallel, each leg including a set of switching elements connected in series;
a resonant circuit including a first resonant unit provided between the first leg and the fourth leg via the first transformer unit, a second resonant unit provided between the second leg and the fifth leg via the second transformer unit, and a third resonant unit provided between the third leg and the sixth leg via the third transformer unit, each resonant unit including a resonant coil and a resonant capacitor provided on both the primary side and the secondary side of each transformer unit;
a control unit that performs forward control to transfer power from the primary side switching circuit to the secondary side switching circuit and reverse control to transfer power from the secondary side switching circuit to the primary side switching circuit;
A three-phase CLLC DC/DC converter comprising:
The control unit
In the forward control, a predetermined phase difference is provided between the first leg, the second leg, and the third leg, and the first leg, the second leg, and the third leg are driven at a predetermined drive frequency to generate current resonance;
In the reverse control, a predetermined phase difference is provided between the fourth leg, the fifth leg, and the sixth leg, and the fourth leg, the fifth leg, and the sixth leg are driven at a predetermined drive frequency to generate current resonance;
The transformer circuit forms N transformer groups,
Each of the N transformer groups is formed as a group consisting of one transformer from each of the transformer sections, and is characterized in that the primary coils are connected in a balanced connection and the secondary coils are connected in a balanced connection.

In the DC/DC converter,
the control unit includes a storage unit,
The storage unit includes:
a first phase difference related to the phase difference during the forward control;
With regard to the phase difference during the forward control, a second phase difference different from the first phase difference;
a first threshold value related to a first output value of an output current and/or an output power output from the secondary-side switching circuit during the forward control;
a third phase difference related to the phase difference during the reverse direction control;
With regard to the phase difference during the reverse direction control, a fourth phase difference different from the third phase difference;
a second threshold value related to a second output value of the output current and/or the output power output from the primary side switching circuit during the reverse direction control,
The control unit
During the forward direction control, the first output value is compared with the first threshold value, and if the first output value is less than the first threshold value, the phase difference is set to the first phase difference, whereas if the first output value is equal to or greater than the first threshold value, the phase difference is set to the second phase difference;
During the reverse direction control, the second output value is compared with the second threshold value, and if the second output value is less than the second threshold value, the phase difference is set to the third phase difference, while if the second output value is equal to or greater than the second threshold value, the phase difference is set to the fourth phase difference.

In order to solve the above problem, a charging device according to the present invention includes:
any one of the DC/DC converters described above;
an AC/DC converter provided in a stage preceding the DC/DC converter;
and an output filter provided in a subsequent stage of the DC/DC converter.

This invention provides a three-phase LLC DC/DC converter and charging device that can handle high output and reduce output waveform ripple without increasing costs or complicating control.

1A is a block diagram of a charging device according to a first embodiment, FIG. 1B is a circuit diagram of a three-phase LLC DC/DC converter according to a first embodiment, and FIG. 1C is a diagram showing an example of the configuration of a first leg of a primary-side switching circuit. 1A and 1B are diagrams showing input voltage vectors and output current vectors in a DC/DC converter according to a first embodiment, in which (A) is an input voltage vector with a phase difference of 120°, (B) is an output current vector corresponding to (A), (C) is an input voltage vector with a phase difference changed from 120°, and (D) is an output current vector corresponding to (C). FIG. 4 is a diagram illustrating an example of a control flow by a control unit of the first embodiment. FIG. 2 is a diagram illustrating a first modification of the DC/DC converter of the first embodiment. FIG. 10 is a diagram illustrating a second modification of the DC/DC converter of the first embodiment. FIG. 10 is a diagram illustrating a third modification of the DC/DC converter of the first embodiment. FIG. 10 is a diagram illustrating a fourth modification of the DC/DC converter of the first embodiment. FIG. 10 is a circuit diagram of a three-phase CLLC DC/DC converter according to a second embodiment. FIG. 1 is a diagram illustrating a first conventional example of a three-phase LLC DC/DC converter. FIG. 10 is a diagram illustrating a second conventional example of a three-phase LLC DC/DC converter. FIG. 10 is a diagram illustrating a third conventional example of a three-phase LLC DC/DC converter.

Embodiments of a three-phase LLC (including three-phase CLLC) DC/DC converter and charging device according to the present invention will be described below with reference to the accompanying drawings.

[First embodiment]
FIG. 1A shows a charging device 1A according to a first embodiment of the present invention, and FIG. 1B shows a three-phase LLC DC/DC converter 10A according to the first embodiment of the present invention.

The charging device 1A includes a power supply unit consisting of an AC/DC converter 2, a DC/DC converter 10A, and an output filter 3, and a control unit 4 that controls the power supply unit. The charging device 1A is a rapid charger that charges electric vehicles, plug-in hybrid vehicles, and other electrically powered vehicles. Note that while FIG. 1(A) shows only one power supply unit, the charging device 1A may also include multiple power supply units connected in parallel.

AC/DC converter 2 converts AC power input from AC power source 5 into DC power while improving the power factor of the AC power, and outputs the converted power to DC/DC converter 10A. DC/DC converter 10A converts the DC power input from AC/DC converter 2 into the desired DC power and supplies it to battery 6 via output filter 3. Battery 6 is, for example, an on-board battery for an electric vehicle. For the purpose of explanation, the system will be divided into three parts: AC/DC converter 2, DC/DC converter 10A, and output filter 3; however, these three parts are merely functional classifications, and the system may be configured as a single board.

The output filter 3 is a noise filter composed of components such as electrolytic capacitors, film capacitors, and choke coils. The output filter 3 is designed to reduce ripple in the output waveform of the DC/DC converter 10A. If ripple can be reduced in the DC/DC converter 10A, the above components of the output filter 3 can be made smaller, lighter, and less expensive.

As shown in Figure 1(B), the DC/DC converter 10A is a three-phase LLC DC/DC converter that includes terminals T1 to T4, a primary-side switching circuit 11, a secondary-side rectifier circuit 12, a transformer circuit 13, a primary-side resonant circuit 14, and a control unit 15.

Terminals T1 and T2 are connected to the DC end of the AC/DC converter 2, and terminals T3 and T4 are connected to the battery 6 via the output filter 3. A DC input voltage V1 is applied to terminals T1 and T2, and a DC output voltage V2 (or a DC current corresponding to the output voltage V2) is output from terminals T3 and T4.

The primary-side switching circuit 11 includes a first leg, a second leg, and a third leg connected in parallel. The first leg includes a pair of switching elements Q1 and Q2 connected in series, the second leg includes a pair of switching elements Q3 and Q4 connected in series, and the third leg includes a pair of switching elements Q5 and Q6 connected in series. Each pair of switching elements Q1 to Q6 may consist of a single switching element, but it is preferable to configure each pair of switching elements Q1 to Q6 with two or more switching elements connected in parallel for current distribution. For example, as shown in Figure 1(C), switching element Q1 may consist of two switching elements Q11 and Q12 connected in parallel, and switching element Q2 may consist of two switching elements Q21 and Q22 connected in parallel. The same applies to switching elements Q3 to Q6.

A diode is connected in parallel in the reverse direction to each current path of switching elements Q1 to Q6. Furthermore, power semiconductors such as IGBTs (insulated gate bipolar transistors) and MOSFETs (metal-oxide semiconductor field-effect transistors) using SiC (silicon carbide) or GaN (gallium nitride) can be used as switching elements Q1 to Q6. The anti-parallel diodes of switching elements Q1 to Q6 can be built-in diodes of the switching elements or individual diodes. Furthermore, partial resonant capacitors can be connected in parallel to each switching element Q1 to Q6, or the parasitic capacitance of each switching element Q1 to Q6 can be utilized.

The secondary-side rectifier circuit 12 includes a fourth leg, a fifth leg, and a sixth leg connected in parallel, and a smoothing capacitor Co. The fourth leg includes two rectifier diodes D1 and D2 connected in series, the fifth leg includes two rectifier diodes D3 and D4 connected in series, and the sixth leg includes two rectifier diodes D5 and D6 connected in series. The rectifier diodes D1 to D6 correspond to the rectifier elements of the present invention. The smoothing capacitor Co is a capacitor for removing output ripple and is connected between terminals T3 and T4. Note that each rectifier diode may be configured with multiple diodes for current distribution. For example, the rectifier diode D1 may be configured with two diodes connected in parallel, and the rectifier diode D2 may be configured with two diodes connected in parallel. The same applies to the other rectifier diodes D3 to D6.

The secondary-side rectifier circuit 12 may be any rectifier circuit, and instead of a three-phase full-bridge rectifier circuit made up of rectifier diodes D1 to D6, a synchronous rectifier circuit made up of multiple switching elements may be used. A synchronous rectifier circuit has multiple switching elements connected in a three-phase full-bridge configuration, and each switching element is turned on/off, for example, by synchronous rectification control. In this case, each switching element corresponds to a rectifier element in the present invention.

Transformer circuit 13 includes high-frequency isolation transformers (hereinafter referred to as transformers) T11 to T32. In transformer circuit 13, transformers T11 and T12 are connected in parallel to form a first transformer section, transformers T21 and T22 are connected in parallel to form a second transformer section, and transformers T31 and T32 are connected in parallel to form a third transformer section. As described above, each transformer section is composed of two parallel transformers, with two transformers connected in parallel, but it may also be composed of three or more parallel transformers, with three or more transformers connected in parallel.

One end of the primary coil of transformer T11 is connected to one end of the primary coil of transformer T12 and is connected to the connection point X1 of switching elements Q1 and Q2 via the first resonant section (resonant coil L1 and resonant capacitor C1) of the primary resonant circuit 14. One end of the primary coil of transformer T21 is connected to one end of the primary coil of transformer T22 and is connected to the connection point X2 of switching elements Q3 and Q4 via the second resonant section (resonant coil L2 and resonant capacitor C2) of the primary resonant circuit 14. One end of the primary coil of transformer T31 is connected to one end of the primary coil of transformer T32 and is connected to the connection point X3 of switching elements Q5 and Q6 via the third resonant section (resonant coil L3 and resonant capacitor C3) of the primary resonant circuit 14.

The primary coils of transformers T11, T21, and T31 are connected in a Y-connection (star connection) (balanced connection), with the other ends of the primary coils connected to the neutral point at point P1. Similarly, the primary coils of transformers T12, T22, and T32 are connected in a Y-connection (balanced connection), with the other ends of the primary coils connected to the neutral point at point P2.

One end of the secondary coil of transformer T11 is connected to one end of the secondary coil of transformer T12 and to connection point X4 of rectifier diodes D1 and D2. One end of the secondary coil of transformer T21 is connected to one end of the secondary coil of transformer T22 and to connection point X5 of rectifier diodes D3 and D4. One end of the secondary coil of transformer T31 is connected to one end of the secondary coil of transformer T32 and to connection point X6 of rectifier diodes D5 and D6.

The secondary coils of transformers T11, T21, and T31 are connected in a Y-connection (balanced connection), with the other ends of the secondary coils connected to the neutral point at point S1. Similarly, the secondary coils of transformers T12, T22, and T32 are connected in a Y-connection (balanced connection), with the other ends of the secondary coils connected to the neutral point at point S2.

That is, in the transformer circuit 13, a first transformer group is formed by a group consisting of transformers T11, T21, and T31, and a second transformer group is formed by a separate group consisting of transformers T12, T22, and T32.

In this embodiment, the primary and secondary coils are connected in a Y-connection. However, as long as the primary coils are connected in a balanced manner, they may be connected in a Δ-connection (delta connection) or a Δ-Cr connection, and the secondary coils may be connected in a Δ-connection. A Δ-Cr connection is, for example, a connection in which the other ends of the primary coils of transformers T11, T21, and T31 are connected together, a resonant capacitor C1 is interposed between the other end of the primary coil of transformer T11 and the other end of the primary coil of transformer T21, a resonant capacitor C2 is interposed between the other end of the primary coil of transformer T21 and the other end of the primary coil of transformer T31, and a resonant capacitor C3 is interposed between the other end of the primary coil of transformer T31 and the other end of the primary coil of transformer T11. The primary coils of transformers T12, T22, and T32 are connected in a similar manner.

The primary resonant circuit 14 includes a first resonant part, a second resonant part, and a third resonant part, and together with the transformer circuit 13, forms a three-phase LLC resonant circuit. Specifically, the resonant coil L1, the excitation coils and primary coils of transformers T11 and T12, and the resonant capacitor C1 form the first LLC resonant circuit. The resonant coil L2, the excitation coils and primary coils of transformers T21 and T22, and the resonant capacitor C2 form the second LLC resonant circuit. The resonant coil L3, the excitation coils and primary coils of transformers T31 and T32, and the resonant capacitor C3 form the third LLC resonant circuit.

Note that the excitation coils of transformers T11 to T32 are included in the transformers T11 to T32 and are not shown in the illustration. Also, while resonance coils L1 to L3 are separate components from the transformers T11 to T32, they may be composed of only the leakage magnetic flux of the transformers T11 to T32, or they may be composed of the leakage magnetic flux of the transformers T11 to T32 and individual coils.

The control unit 15 includes drive circuits for the switching elements Q1 to Q6 to turn them on and off, a control circuit (including a detection circuit) to send control signals to each drive circuit, and a memory unit that stores the characteristics of the resonant components described below. The control unit 15 may be composed of an analog circuit, a combination of logic circuits, a digital circuit such as a microcontroller or DSP, or a circuit that combines analog and digital circuits. In the case of an analog circuit, the memory unit may be a specified voltage or current circuit that forms a threshold value. In the case of a digital circuit, the memory unit may be composed of a memory within the processing unit, or a non-volatile memory provided separately from the processing unit.

The control unit 15 periodically (or constantly) detects the output voltage V2 and the output current output from terminals T3 and T4 using, for example, a detector not shown, and changes the drive frequency of the switching elements Q1 to Q6 so that the current value of the output current (hereinafter referred to as the output current value) matches a predetermined target current value.

Specifically, the control unit 15 alternately turns on and off the switching elements Q1 and Q2 that make up the first leg with a 180° phase difference, providing a predetermined dead time; alternately turns on and off the switching elements Q3 and Q4 that make up the second leg with a 180° phase difference, providing a predetermined dead time; and alternately turns on and off the switching elements Q5 and Q6 that make up the third leg with a 180° phase difference, providing a predetermined dead time. The on-duty of the switching elements Q1, Q3, and Q5 is set to, for example, 50%.

The control unit 15 also drives the switching elements Q1 to Q6 by providing a constant phase difference between the first, second, and third legs (interleaved drive). For example, the control unit 15 sets the phase difference between the first and second legs to 120°, the phase difference between the second and third legs to 120°, and the phase difference between the third and first legs to 120°. In this case, the phase difference between the on-timings of the switching elements Q1, Q3, and Q5 is 120°, and the phase difference between the off-timings is also 120°. Similarly, the phase difference between the on-timings of the switching elements Q2, Q4, and Q6 is 120°, and the phase difference between the off-timings is also 120°.

Furthermore, the control unit 15 sets the drive frequencies of the switching elements Q1 to Q6 to the same drive frequency and simultaneously changes the same drive frequencies to the same extent to match the output current value with a predetermined target current value (frequency modulation control). For example, if the current gain of the DC/DC converter 10A (= output current / input current input to terminals T1 and T2) is to be reduced below 1, the control unit 15 drives the switching elements Q1 to Q6 at a drive frequency higher than a first resonant frequency determined by the resonant coils L1 to L3 and the resonant capacitors C1 to C3. On the other hand, if the current gain is to be increased above 1, the control unit 15 drives the switching elements Q1 to Q6 at a drive frequency lower than the first resonant frequency and higher than a second resonant frequency determined by the resonant coils L1 to L3, the excitation coils of the transformers T11 to T32, and the resonant capacitors C1 to C3.

By the above control, the first resonant current generated by the current resonance of the first LLC resonant circuit flows to the secondary side rectifier circuit 12 via transformers T11 and T12, the second resonant current generated by the current resonance of the second LLC resonant circuit flows to the secondary side rectifier circuit 12 via transformers T21 and T22, and the third resonant current generated by the current resonance of the third LLC resonant circuit flows to the secondary side rectifier circuit 12 via transformers T31 and T32. The output of the secondary side rectifier circuit 12 is a composite output of three-phase currents: the first resonant current, the second resonant current, and the third resonant current.

The DC/DC converter 10A is a three-phase LLC system that operates with a fixed phase difference between the first, second, and third legs, which allows for reduced ripple in the output waveform compared to a single-phase full-bridge system. Furthermore, in the DC/DC converter 10A, transformers T11, T21, and T31 form a first transformer group, and transformers T12, T22, and T32 form another group to form a second transformer group, with the other ends of the primary and secondary coils of each transformer group connected to the neutral point via a Y-connection (balanced connection).

As a result, even if there is variation in the characteristics of components that contribute to resonance (hereinafter referred to as resonant components), such as the resonant coils L1 to L3, resonant capacitors C1 to C3, and the primary and secondary coils of the transformers T11 to T32, the DC/DC converter 10A can mitigate imbalances in the first resonant current, second resonant current, and third resonant current, thereby reducing ripple in the output waveform of the DC/DC converter 10A.

Table 1 shows example values for the DC/DC converter 10A when there is no variation in the characteristics of the resonant coils L1 to L3, the excitation coils of the transformers T11 to T32, the primary coils, and the secondary coils. Table 2 shows example values when there is variation in the above characteristics.

In Tables 1 and 2, L, Lm1, Lm2, Lp1, Lp2, Ls1, and Ls2 of the first LLC resonant circuit are the inductances of the resonant coil L1, the excitation coil of transformer T11, the excitation coil of transformer T12, the primary coil of transformer T11, the primary coil of transformer T12, the secondary coil of transformer T11, and the secondary coil of transformer T12, respectively. In the second LLC resonant circuit, L, Lm1, Lm2, Lp1, Lp2, Ls1, and Ls2 are the inductances of the resonant coil L2, the excitation coil of transformer T21, the excitation coil of transformer T22, the primary coil of transformer T21, the primary coil of transformer T22, the secondary coil of transformer T21, and the secondary coil of transformer T22, respectively. Similarly, L, Lm1, Lm2, Lp1, Lp2, Ls1, and Ls2 of the third LLC resonant circuit are the inductances of the resonant coil L3, the excitation coil of transformer T31, the excitation coil of transformer T32, the primary coil of transformer T31, the primary coil of transformer T32, the secondary coil of transformer T31, and the secondary coil of transformer T32, respectively. The capacitance (electrostatic capacity) of each of the resonant capacitors C1 to C3 is 264 nF.

Note that Table 2 shows an example where the characteristic variations of the resonant components have been set taking into account typical manufacturing control variations. Specifically, the characteristic (inductance) variations of the resonant coils L1 to L3 and the primary and secondary coils of the transformers T11 to T32 are set to approximately 7%, and the characteristic variation of the excitation coils of the transformers T11 to T32 is set to approximately 20%. The characteristic (capacitance) of the resonant capacitors C1 to C3 generally varies by approximately 5%, but this is not taken into account here and is assumed to be zero.

Table 3 shows simulation results for the first resonant current, second resonant current, and third resonant current, as well as their average values, when the configuration of transformers T11 to T32 (connection of the primary and secondary coils) is changed when there is variation in the characteristics shown in Table 2. In the simulation, the drive frequency of switching elements Q1 to Q6 is set to a value near a frequency higher than the second resonant frequency determined by resonant coils L1 to L3, the excitation coils and primary coils of transformers T11 to T32, and resonant capacitors C1 to C3.

In Table 3, A is the configuration of the transformer circuit 13 of this embodiment. B is a configuration in which six primary coils are connected together in a Y-connection as shown in Figure 11, and the secondary coils are connected in a Y-connection in groups of three, as in transformer circuit 13. C is a configuration in which only the primary coils are connected in a Y-connection in groups of three, as in transformer circuit 13, and the secondary coils are connected in a Y-connection in groups of six. D is the configuration of the conventional transformer circuit of Figure 11, in which both the primary coils and secondary coils are connected in a Y-connection in groups of six.

In Table 3, the ratios of the first resonant current, second resonant current, and third resonant current are the ratios of the minimum value to the maximum value of the current after rectification by the secondary side rectifier circuit 12 (before passing through the smoothing capacitor Co). For example, the ratio of the first resonant current = (minimum value of the first resonant current / maximum value of the first resonant current) x 100 [%]. The average value is the average of the ratios of the first resonant current, the second resonant current, and the third resonant current.

The results in Table 3 show that the conventional configuration D (Figure 11) has the largest difference between the minimum and maximum values of the output current, and that the difference between the minimum and maximum values of the output current gradually decreases in the order of configuration C and configuration B, and that configuration A, which is the configuration of the transformer circuit 13 of this embodiment, has the smallest difference between the minimum and maximum values of the output current. In other words, it can be seen that the DC/DC converter 10A of this embodiment, thanks to the configuration of the transformer circuit 13, can alleviate the imbalance state of the first resonant current, second resonant current, and third resonant current without special control to alleviate the current imbalance of each phase, and reduce ripple in the output waveform.

The impact of imbalance in resonant current due to variations in the characteristics of resonant components varies depending on the output current or output power, and is known to be most problematic when the output current or output power is at its maximum. For example, if a large resonant current flows through the primary-side resonant circuit 14 or transformer circuit 13 and the resonant current of a specific phase becomes larger than expected, abnormal heat generation occurs in the resonant coils L1 to L3 and transformers T11 to T32, causing sudden changes in characteristics such as inductance. As a result, the resonant current increases even further, and in the worst case scenario, the components that make up the DC/DC converter 10A may be damaged.

We will explain how to address the above problem using Figure 2. Figure 2 shows the input voltage vector and output current vector when there is variation in the characteristics of the resonant components.

Figure 2(A) shows the input voltage vectors v1, v2, and v3 at connection points X1, X2, and X3 when driven with a phase difference of 120° between the first, second, and third legs. The input voltage vectors v1, v2, and v3 are equal in magnitude, and the phase difference θ11 between input voltage vector v1 and input voltage vector v2, the phase difference θ21 between input voltage vector v2 and input voltage vector v3, and the phase difference θ31 between input voltage vector v3 and input voltage vector v1 are all 120°.

Figure 2(B) shows the output current vectors i1, i2, and i3 corresponding to each resonant current after being rectified by the secondary-side rectifier circuit 12 (before passing through the smoothing capacitor Co) in the case of Figure 2(A). Due to variations in the characteristics of the resonant components, the output current vectors i1 and i3 increase while the output current vector i2 decreases, resulting in an unbalanced state between the output current vectors i1, i2, and i3. The phase differences φ11, φ21, and φ31 between the output current vectors i1, i2, and i3 are offset from 120°.

The characteristics of resonant components can be easily measured on a component-by-component basis. The characteristics of coils and transformers are often known in advance through testing. Furthermore, capacitor characteristics vary less than those of coils and transformers, and there is a consistent tendency for variation across production lots, making them easy to measure. Meanwhile, it is known that if there is an imbalance in the output current of a three-phase LLC system, the output current can be balanced by shifting the phase of each phase from 120° (Non-Patent Document 1). Therefore, using simulation or a real circuit (an actual DC/DC converter), the phase difference between the first, second, and third legs can be gradually changed from 120° while the output current or output power is at its maximum, and the phase difference at which the output current vectors i1, i2, and i3 are balanced can be determined.

Figure 2(C) shows the input voltage vectors v1, v2, and v3 at connection points X1, X2, and X3 where the output current vectors i1, i2, and i3 are in equilibrium. In Figure 2(C), the phase difference θ12 is wider than 120° by α, while the phase difference θ32 is narrower than 120° by β. The phase difference θ22 is θ22 = 120° - α + β. α and β are angles greater than 0° and less than 120°.

Figure 2(D) shows the output current vectors i1, i2, and i3 corresponding to each resonant current after being rectified by the secondary-side rectifier circuit 12 (before passing through the smoothing capacitor Co) in the case of Figure 2(C). Compared to Figure 2(B), output current vectors i1 and i3 are smaller, output current vector i2 is larger, and output current vectors i1, i2, and i3 are in a balanced state. In addition, the phase differences φ12, φ22, and φ32 between output current vectors i1, i2, and i3 are all 120°.

Next, with reference to the control flow in Figure 3, we will explain how the control unit 15 controls the switching elements Q1 to Q6. The control unit 15 is assumed to have stored in advance the first phase difference, the second phase difference, and the first threshold value (current threshold value and power threshold value).

The first phase difference is the phase difference between the first leg, second leg, and third leg in the low-medium output range (low-medium output current range and low-medium output power range) where the impact of imbalance in the resonant current due to variations in the characteristics of the resonant components is relatively small. In this embodiment, the first phase difference is the phase difference (θ11, θ21, θ31) between the input voltage vectors v1, v2, and v3 shown in Figure 2(A), where θ11 = θ21 = θ31 = 120°.

The second phase difference is the phase difference between the first leg, second leg, and third leg in the high output region (high output current region or high output power region) where the impact of imbalance in the resonant current due to variations in the characteristics of the resonant components is relatively large. In this embodiment, the second phase difference is defined as the phase difference (θ12, θ22, θ32) between the input voltage vectors v1, v2, and v3 shown in Figure 2(C), where θ12 = 120° + α, θ22 = 120° - α + β, and θ32 = 120° - β.

The current threshold is a threshold used by the control unit 15 to determine whether it is in the low-medium output current region or the high output current region. The power threshold is a threshold used by the control unit 15 to determine whether it is in the low-medium output power region or the high output power region. The current threshold and power threshold can be set as appropriate; for example, the rated output value of the DC/DC converter 10A may be set as the threshold.

After starting control, the control unit 15 sets the phase difference between the first leg, second leg, and third leg to the first phase difference (θ11, θ21, θ31) (S1). The control unit 15 turns the switching elements Q1 to Q6 on and off at the same drive frequency so that the phase difference between the first leg, second leg, and third leg becomes the first phase difference (θ11, θ21, θ31), and controls the drive frequency of the switching elements Q1 to Q6 so that the output current value matches a predetermined target current value.

The phase difference in the on/off timing between switching elements Q1, Q3, and Q5 is 120°, and the phase difference in the on/off timing between switching elements Q2, Q4, and Q6 is also 120°. Furthermore, switching elements Q1 and Q2 alternately turn on and off with a 180° phase difference, with dead time provided, and switching elements Q3, Q4 and switching elements Q5, Q6 also alternately turn on and off with a 180° phase difference, with dead time provided. The on-duty of switching elements Q1, Q3, and Q5 is set to, for example, 50%.

After starting to drive the switching elements Q1 to Q6, the control unit 15 acquires the output current value and the voltage value of the output voltage V2 from a detector (not shown), and calculates and acquires the power value of the output power (hereinafter referred to as the output power value) output from terminals T3 and T4 (S2).

Next, the control unit 15 compares the output current value acquired in step S2 with the current threshold value pre-stored in the memory unit (S3). If the output current value is equal to or greater than the current threshold value (YES in S3), the control unit 15 sets the phase differences between the first leg, second leg, and third leg to the second phase differences (θ12, θ22, θ32) (S4).

The control unit 15 sets the phase difference to the second phase difference (θ12, θ22, θ32). It turns on and off switching elements Q1 to Q6 at the same drive frequency so that the phase difference between the first leg and second leg is θ12 = 120° + α, the phase difference between the second leg and third leg is θ22 = 120° - α + β, and the phase difference between the third leg and the first leg is θ32 = 120° - β, and controls the drive frequency of switching elements Q1 to Q6 so that the output current value matches a predetermined target current value.

In the comparison of step S3, if the output current value is less than the current threshold (NO in S3), the control unit 15 compares the output power value acquired in step S2 with the power threshold pre-stored in the memory unit (S5). If the output power value is equal to or greater than the power threshold (YES in S5), the control unit 15 sets the phase differences between the first leg, second leg, and third leg to the second phase differences (θ12, θ22, θ32) (S4). On the other hand, if the output power value is less than the power threshold (NO in S5), the control unit 15 sets the phase differences between the first leg, second leg, and third leg to the first phase differences (θ11, θ21, θ31) (S6).

The control unit 15 then determines whether or not to continue control (S7). The control unit 15 makes the determination in step S7 based on a processing end instruction from a higher-level system (for example, the control unit 4 shown in Figure 1 (A)). If the control unit 15 has not received a processing end instruction, it determines to continue control (YES in S7) and proceeds to processing in step S2. However, if the control unit 15 has received a processing end instruction, it determines not to continue control (NO in S7) and ends control.

As described above, the control unit 15 sets the phase difference between each leg to the first phase difference (θ11, θ21, θ31) in the low and medium output range, and sets the phase difference between each leg to the second phase difference (θ12, θ22, θ32) in the high output range. The second phase difference (θ12, θ22, θ32) is a phase difference that mitigates the imbalance state of the output current vectors corresponding to each resonant current, even if there is variation in the characteristics of the resonant components.

As a result, DC/DC converter 10A can mitigate the imbalance in the resonant current and reduce ripple in the output waveform, even in the high-output range where the impact of the imbalance in the resonant current due to variations in the characteristics of the resonant components is relatively large. Furthermore, compared to the method described in Non-Patent Document 1 (which detects the resonant current of each phase and changes the phase difference between the input voltages of each phase depending on the operating state of the DC/DC converter), the number of components can be reduced and control complexity can be avoided. This is also true when compared to a method that detects the resonant current or load current of each phase and adjusts the drive frequency of each phase individually without changing the phase of each phase to mitigate the imbalance in the resonant current.

In other words, DC/DC converter 10A can accommodate higher output by paralleling transformers T11 to T32, and can reduce ripple in the output waveform by mitigating imbalances in the resonant current caused by variations in the characteristics of the resonant components, without increasing costs or complicating control.

In this embodiment, the first phase difference (θ11, θ21, θ31) is set to θ11 = θ21 = θ31 = 120°, but it may also be set to an angle other than 120°. For example, as with the second phase difference, using a simulation or an actual circuit, the phase difference between the first leg, second leg, and third leg can be gradually changed from 120° while the output current and output power are in the low-medium output range, to determine the phase difference at which the output current vectors corresponding to each resonant current are in a balanced state, and the determined phase difference can be used as the first phase difference.

In this embodiment, the control unit 15 stores two phase differences for two regions: a first phase difference in the low-medium output region and a second phase difference in the high output region, and performs control. However, control may be performed by storing three or more phase differences for three or more regions. For example, the control unit 15 may store a first first phase difference (first phase difference A) in the low output region, a second first phase difference (first phase difference B) in the medium output region, and a second phase difference in the high output region, and perform control. Alternatively, the control unit 15 may store a first phase difference in the low-medium output region, a first second phase difference (second phase difference A) in the high output current region, and a second second phase difference (second phase difference B) in the high output power region, and perform control.

Furthermore, the control unit 15 may perform only one of the comparison of output current values (S3 in Figure 3) or the comparison of output power values (S5 in Figure 3) depending on the type of battery 6 and the control purpose, and set the phase difference between each leg.

[First Modification of First Embodiment]
4 shows a three-phase LLC DC/DC converter 10B according to a first modification. The DC/DC converter 10B has the same configuration as the DC/DC converter 10A of the first embodiment, except for a primary-side resonant circuit 14B.

In the primary side resonant circuit 14B, the resonant coil of the first resonant section is composed of a resonant coil L11 connected to the primary side coil of the transformer T11 and a resonant coil L12 connected to the primary side coil of the transformer T12, the resonant coil of the second resonant section is composed of a resonant coil L21 connected to the primary side coil of the transformer T21 and a resonant coil L22 connected to the primary side coil of the transformer T22, and the resonant coil of the third resonant section is composed of a resonant coil L31 connected to the primary side coil of the transformer T31 and a resonant coil L32 connected to the primary side coil of the transformer T32.

With the above configuration, the resonant current flowing through each of the resonant coils L11 to L32 can be reduced compared to the first embodiment, making it possible to reduce the size, weight, and cost of the resonant coils L11 to L32. Furthermore, the amount of heat generated by the resonant coils L11 to L32 can also be reduced.

In addition, the resonant coils and resonant capacitors in the primary resonant circuit 14B may be interchanged. That is, the resonant capacitors of the first resonant section may be composed of resonant capacitor C11 connected to the primary coil of transformer T11 and resonant capacitor C12 connected to the primary coil of transformer T12, the resonant capacitors of the second resonant section may be composed of resonant capacitor C21 connected to the primary coil of transformer T21 and resonant capacitor C22 connected to the primary coil of transformer T22, and the resonant capacitors of the third resonant section may be composed of resonant capacitor C31 connected to the primary coil of transformer T31 and resonant capacitor C32 connected to the primary coil of transformer T32, and each resonant coil L11 to L32 may be composed of resonant coils L1 to L3, as in the first embodiment.

[Second Modification of First Embodiment]
5 shows a three-phase LLC DC/DC converter 10C according to the second modification. The DC/DC converter 10C has the same configuration as the DC/DC converter 10B according to the first modification, except for a primary-side resonant circuit 14C.

The primary-side resonant circuit 14C is the same as the primary-side resonant circuit 14B of the first variant, except that the resonant capacitors of the first resonant section are composed of a resonant capacitor C11 connected to the resonant coil L11 and a resonant capacitor C12 connected to the resonant coil L12, the resonant capacitors of the second resonant section are composed of a resonant capacitor C21 connected to the resonant coil L21 and a resonant capacitor C22 connected to the resonant coil L22, and the resonant capacitors of the third resonant section are composed of a resonant capacitor C31 connected to the resonant coil L31 and a resonant capacitor C32 connected to the resonant coil L32.

With the above configuration, the resonant current flowing through each of the resonant coils L11 to L32 and each of the resonant capacitors C11 to C32 can be reduced compared to the first embodiment, making it possible to reduce the size, weight, and cost of the resonant coils L11 to L32 and the resonant capacitors C11 to C32. Furthermore, the amount of heat generated by the resonant coils L11 to L32 and the resonant capacitors C11 to C32 can also be reduced.

[Third Modification of First Embodiment]
6 shows a three-phase LLC DC/DC converter 10D according to a third modification. The DC/DC converter 10D has the same configuration as the DC/DC converter 10C according to the second modification, except for a primary-side resonant circuit 14D.

The primary resonant circuit 14D is the same as the primary resonant circuit 14C of the second modified example, except that the positions of the resonant capacitors C11 to C32 have been moved to the other end of the primary coil of each of the transformers T11 to T32. The resonant capacitors C11, C21, and C31 are inserted between the other end of the primary coil of each of the transformers T11, T21, and T31, respectively, and point P1, and the resonant capacitors C12, C22, and C32 are inserted between the other end of the primary coil of each of the transformers T12, T22, and T32, respectively, and point P2.

In the first resonance section, the positions of the resonance coils L11 and L12 and the resonance capacitors C11 and C12 can be changed as desired between the connection point X1 of the switching elements Q1 and Q2 and points P1 and P2, as long as the equivalent circuit of the first resonance section remains the same. The same applies to the second and third resonance sections.

Furthermore, in the primary-side resonant circuit 14D, the resonant coils L11 to L32 and the resonant capacitors C11 to C32 may be divided and arranged as long as the equivalent circuits of each resonant section are the same. For example, the resonant capacitor C11 may be divided into two and arranged on one end side and the other end side of the primary-side coil of the resonant coil L11.

[Fourth Modification of the First Embodiment]
7 shows a three-phase LLC DC/DC converter 10E according to a fourth modification. The DC/DC converter 10E has the same configuration as the DC/DC converter 10A of the first embodiment, except for a secondary-side rectifier circuit 12E.

The secondary rectifier circuit 12E includes three full-bridge rectifier circuits (rectifier diodes D11 to D62) corresponding to each transformer section of the transformer circuit 13. The full-bridge rectifier circuit corresponding to the first transformer section is composed of rectifier diodes D11 to D22, with one end of the secondary coil of the transformer T11 connected to the junction X41 of the rectifier diodes D11 and D21, and one end of the secondary coil of the transformer T12 connected to the junction X42 of the rectifier diodes D12 and D22. The full-bridge rectifier circuit corresponding to the second transformer section is composed of rectifier diodes D31 to D42, with one end of the secondary coil of the transformer T21 connected to the junction X51 of the rectifier diodes D31 and D41, and one end of the secondary coil of the transformer T22 connected to the junction X52 of the rectifier diodes D32 and D42. The full-bridge rectifier circuit corresponding to the third transformer section is composed of rectifier diodes D51 to D62, with one end of the secondary coil of the transformer T31 connected to the connection point X61 between the rectifier diodes D51 and D61, and one end of the secondary coil of the transformer T32 connected to the connection point X62 between the rectifier diodes D52 and D62.

With the above configuration, compared to the first embodiment, the load current due to the resonant current flowing through each of the rectifier diodes D11 to D62 can be reduced, making the rectifier diodes D11 to D62 smaller, lighter, and less expensive. Furthermore, the amount of heat generated by the rectifier diodes D11 to D62 can also be reduced.

Second Embodiment
8 shows a three-phase bidirectional CLLC DC/DC converter 20 according to a second embodiment of the present invention. The DC/DC converter 20 includes terminals T1 to T4, a primary-side switching circuit 21, a secondary-side switching circuit 22, a transformer circuit 23, a primary-side resonant circuit 24 and a secondary-side resonant circuit 24', which correspond to the "resonant circuit" of the present invention, and a control unit 25, and performs bidirectional power transmission.

The primary-side switching circuit 21 has the same configuration as the primary-side switching circuit 11 of the first embodiment, except that it includes a smoothing capacitor Co1 connected between terminals T1 and T2.

The secondary-side switching circuit 22 includes a fourth leg, a fifth leg, and a sixth leg connected in parallel, and a smoothing capacitor Co2 connected between terminals T3 and T4. The fourth leg includes a set of switching elements Q7 and Q8 connected in series, the fifth leg includes a set of switching elements Q9 and Q10 connected in series, and the sixth leg includes a set of switching elements Q11 and Q12 connected in series. A diode is connected in parallel in the opposite direction to each current path of switching elements Q7 to Q12. In other words, the secondary-side switching circuit 22 has the same circuit configuration as the primary-side switching circuit 21.

The transformer circuit 23 has the same configuration as the transformer circuit 13 in the first embodiment, and the primary side resonant circuit 24 has the same configuration as the primary side resonant circuit 14 in the first embodiment.

The secondary-side resonant circuit 24' has the same configuration as the primary-side resonant circuit 24, and includes a first resonant section consisting of a resonant coil L4 and a resonant capacitor C4, a second resonant section consisting of a resonant coil L5 and a resonant capacitor C5, and a third resonant section consisting of a resonant coil L6 and a resonant capacitor C6.

One end of the secondary coils of the transformers T11 and T12 is connected to the connection point X4 of the switching elements Q7 and Q8 via a first resonant section (resonant coil L4 and resonant capacitor C4). One end of the secondary coils of the transformers T21 and T22 is connected to the connection point X5 of the switching elements Q9 and Q10 via a second resonant section (resonant coil L5 and resonant capacitor C5). One end of the secondary coils of the transformers T31 and T32 is connected to the connection point X6 of the switching elements Q11 and Q12 via a third resonant section (resonant coil L6 and resonant capacitor C6).

The primary resonant circuit 24 and the secondary resonant circuit 24', together with the transformer circuit 23, form a three-phase bidirectional CLLC resonant circuit. Specifically, the resonant capacitor C1, resonant coil L1, transformers T11 and T12 (excitation coil, primary coil, and secondary coil), resonant coil L4, and resonant capacitor C4 form the first CLLC resonant circuit. The resonant capacitor C2, resonant coil L2, transformers T21 and T22 (excitation coil, primary coil, and secondary coil), resonant coil L5, and resonant capacitor C5 form the second CLLC resonant circuit. The resonant capacitor C3, resonant coil L3, transformers T31 and T32 (excitation coil, primary coil, and secondary coil), resonant coil L6, and resonant capacitor C6 form the third CLLC resonant circuit.

Note that the excitation coils of transformers T11 to T32 are included in the transformers T11 to T32 and are not shown in the illustration. Also, resonance coils L4 to L6 are separate components from the transformers T11 to T32, but they may be composed of only the leakage magnetic flux of the transformers T11 to T32, or they may be composed of the leakage magnetic flux of the transformers T11 to T32 and individual coils.

The control unit 25 includes drive circuits for the switching elements Q1 to Q12 to turn them on and off, a control circuit (including a detection circuit) to send control signals to each drive circuit, and a memory unit that stores the characteristics of the resonant components. Each drive circuit, control circuit, and memory unit has the same configuration as in the first embodiment.

If the control unit 25 refers to the control that transfers power from terminals T1 and T2 to terminals T3 and T4 as forward control, it also performs this forward control and reverse control that transfers power from terminals T3 and T4 to terminals T1 and T2.

During forward control, the control unit 25 performs the same control as in the first embodiment for the switching elements Q1 to Q6, and performs diode bridge rectification control for the switching elements Q7 to Q12, which are always off, or performs synchronous rectification control, which turns the switching elements Q7 to Q12 on and off in synchronization with the first to third resonant currents. Reverse control is explained below.

The control of switching elements Q7 to Q12 during reverse control is the same as the control of switching elements Q1 to Q6 during forward control. The control of switching elements Q1 to Q6 during reverse control is the same as the control of switching elements Q7 to Q12 during forward control, and is diode bridge rectification control or synchronous rectification control as described above.

During reverse control, the control unit 25 uses a detector (not shown) to detect the output voltage V1 and the output current output from terminals T1 and T2, and changes the drive frequency of switching elements Q7 to Q12 so that the output current value matches a predetermined target current value. The drive frequencies of switching elements Q7 to Q12 are all the same.

Specifically, the control unit 25 alternately turns on and off the switching elements Q7 and Q8 that make up the fourth leg with a 180° phase difference, providing a predetermined dead time; alternately turns on and off the switching elements Q9 and Q10 that make up the fifth leg with a 180° phase difference, providing a predetermined dead time; and alternately turns on and off the switching elements Q11 and Q12 that make up the sixth leg with a 180° phase difference, providing a predetermined dead time. The on-duty of the switching elements Q7, Q9, and Q11 is set to, for example, 50%.

The control unit 25 also drives the switching elements Q7 to Q12 by providing a constant phase difference between the fourth, fifth, and sixth legs. For example, the control unit 25 sets the phase difference between the fourth and fifth legs to 120°, the phase difference between the fifth and sixth legs to 120°, and the phase difference between the sixth and fourth legs to 120°. In this case, the phase difference between the on-timings of the switching elements Q7, Q9, and Q11 is 120°, and the phase difference between the off-timings is also 120°. Similarly, the phase difference between the on-timings of the switching elements Q8, Q10, and Q12 is 120°, and the phase difference between the off-timings is also 120°.

By the above control, a first resonant current generated by current resonance of the first LLLC resonant circuit, a second resonant current generated by current resonance of the second LLLC resonant circuit, and a third resonant current generated by current resonance of the third LLLC resonant circuit flow through the primary-side switching circuit 21. The output of the primary-side switching circuit 21 is a composite output of three-phase currents: the first resonant current, the second resonant current, and the third resonant current.

The resonant capacitor and resonant coil on the secondary side of the transformer can be equivalently converted to the primary side of the transformer, and the resonant capacitor and resonant coil on the primary side of the transformer can be equivalently converted to the secondary side of the transformer. Therefore, the equivalent circuits of the secondary-side resonant circuit 24' and primary-side resonant circuit 24 in this embodiment are the same as the equivalent circuit of the primary-side resonant circuit 14 in the first embodiment, and the impact of imbalance in the resonant current due to variations in the characteristics of the resonant components is common to this embodiment and the first embodiment. As a result, the results in Table 3 for the first embodiment also apply to this embodiment, and it can be seen that the configuration of the transformer circuit 23 in this embodiment can alleviate imbalances in the first, second, and third resonant currents and reduce ripple in the output waveform.

Furthermore, during reverse direction control, the control unit 25 executes a control flow similar to the control flow shown in Figure 3. In addition to the first phase difference, second phase difference, and first threshold, the control unit 25 also pre-stores a third phase difference, a fourth phase difference, and a second threshold (current threshold and power threshold).

The third phase difference is the phase difference between the fourth leg, fifth leg, and sixth leg in the low-medium output range (low-medium output current range and low-medium output power range), where the impact of imbalance in the resonant current due to variations in the characteristics of the resonant components is relatively small. In this embodiment, the third phase difference is defined as the phase difference (θ41, θ51, θ61) between the input voltage vectors v4, v5, and v6 at connection points X4, X5, and X6 when driven with a phase difference of 120° between the fourth leg, fifth leg, and sixth leg, where θ41 = θ51 = θ61 = 120°.

The fourth phase difference is the phase difference between the fourth leg, fifth leg, and sixth leg in the high output region (high output current region or high output power region) where the impact of imbalance in the resonant current due to variations in the characteristics of the resonant components is relatively large. In this embodiment, the fourth phase difference is the phase difference (θ42, θ52, θ62) between the input voltage vectors v4, v5, and v6 that mitigates the imbalance in the output current vectors corresponding to each resonant current even if there are variations in the characteristics of the resonant components, where θ42 = 120° + α, θ52 = 120° - α + β, and θ62 = 120° - β. α and β are angles greater than 0° and less than 120°.

The current threshold is a threshold used by the control unit 25 to determine whether it is in the low-medium output current region or the high output current region. The power threshold is a threshold used by the control unit 25 to determine whether it is in the low-medium output power region or the high output power region. The current threshold and power threshold can be set as appropriate; for example, the rated output value during reverse control of the DC/DC converter 20 may be set as the threshold.

During reverse direction control, the control unit 25 sets the phase difference between each leg to the third phase difference (θ41, θ51, θ61) in the low and medium output range, and sets the phase difference between each leg to the fourth phase difference (θ42, θ52, θ62) in the high output range. Therefore, even in the high output range where the impact of resonant current imbalance due to variations in the characteristics of resonant components is relatively large, the DC/DC converter 20 can mitigate the resonant current imbalance and reduce ripple in the output waveform without measuring and performing complex control to mitigate the current imbalance in each phase.

In other words, DC/DC converter 20 can accommodate higher output by paralleling transformers T11 to T32, and can reduce ripple in the output waveform by mitigating imbalances in the resonant current caused by variations in the characteristics of the resonant components, without increasing costs or complicating control.

In this embodiment, the third phase difference (θ41, θ51, θ61) is set to θ41 = θ51 = θ61 = 120°, but it may also be set to an angle other than 120°. For example, as with the fourth phase difference, using a simulation or an actual circuit, the phase difference between the fourth, fifth, and sixth legs can be gradually changed from 120° while the output current or output power is within the low-medium output range, to determine the phase difference at which the output current vectors corresponding to each resonant current are in a balanced state, and the determined phase difference can be used as the third phase difference.

In this embodiment, the control unit 25 stores two phase differences for two regions: the third phase difference in the low-medium output region and the fourth phase difference in the high output region, and performs reverse control. However, control may be performed by storing three or more phase differences for three or more regions. For example, the control unit 25 may store a first third phase difference (third phase difference A) in the low output region, a second third phase difference (third phase difference B) in the medium output region, and a fourth phase difference in the high output region. Alternatively, the control unit 25 may store a first phase difference in the low-medium output region, a first fourth phase difference (fourth phase difference A) in the high output current region, and a second fourth phase difference (fourth phase difference B) in the high output power region, and perform control.

Furthermore, the control unit 25 may perform only one of the comparisons of output current values (corresponding to S3 in Figure 3) or output power values (corresponding to S5 in Figure 3) depending on the type of battery 6 and the control purpose, and set the phase difference between each leg.

The above describes embodiments of the three-phase LLC DC/DC converter and charging device according to the present invention, but the present invention is not limited to the above embodiments.

The DC/DC converter according to the present invention comprises a transformer circuit including a first transformer section, a second transformer section, and a third transformer section, each of which is composed of N transformers (N is an integer of 2 or greater), a first leg, a second leg, and a third leg connected in parallel, each of which includes a set of switching elements connected in series, a primary-side switching circuit provided on the primary side of the transformer circuit, a first resonant section provided between the first leg and the first transformer section, a second resonant section provided between the second leg and the second transformer section, and a third resonant section provided between the third leg and the third transformer section, each of which includes a resonant coil and A three-phase LLC DC/DC converter comprising a primary-side resonant circuit including a resonant capacitor, a secondary-side rectifier circuit including multiple rectifier elements and provided on the secondary side of the transformer circuit, and a control unit that sets a predetermined phase difference between the first leg, second leg, and third leg and drives them at a predetermined drive frequency to generate current resonance. The transformer circuit forms N transformer groups, each of which is formed with one transformer from each transformer unit as a group. The configuration can be changed as needed as long as the primary coil is connected in a balanced connection and the secondary coil is connected in a balanced connection.

For example, the configurations of the first to third modifications of the first embodiment can be applied to the primary side resonant circuit 24 and secondary side resonant circuit 24' of the second embodiment.

1A Charging device 2 AC/DC converter 3 Output filter 4 Control unit 5 AC power supply 6 Batteries 10A to 10E DC/DC converter 11 Primary side switching circuit 12, 12E Primary side rectifier circuit 13 Transformer circuits 14, 14B to 14D Primary side resonant circuit 15 Control unit 20 DC/DC converter 21 Primary side switching circuit 22 Secondary side switching circuit 23 Transformer circuit 24 Primary side resonant circuit 24' Secondary side resonant circuit 25 Control unit

Claims (5)

a transformer circuit including a first transformer unit, a second transformer unit, and a third transformer unit, each of which is configured with N transformers (N is an integer of 2 or more);
a primary-side switching circuit provided on a primary side of the transformer circuit, the primary-side switching circuit including a first leg, a second leg, and a third leg connected in parallel, each leg including a set of switching elements connected in series;
a primary-side resonant circuit including a first resonant unit provided between the first leg and the first transformer unit, a second resonant unit provided between the second leg and the second transformer unit, and a third resonant unit provided between the third leg and the third transformer unit, each resonant unit including a resonant coil and a resonant capacitor;
a secondary-side rectifier circuit including a plurality of rectifier elements and provided on the secondary side of the transformer circuit;
a control unit that provides a predetermined phase difference between the first leg, the second leg, and the third leg and drives the first leg at a predetermined drive frequency to generate current resonance;
A three-phase LLC DC/DC converter comprising:
The transformer circuit forms N transformer groups,
A DC/DC converter characterized in that each of the N transformer groups is formed as a group consisting of one transformer from each of the transformer units, and the primary coils are connected in a balanced connection and the secondary coils are connected in a balanced connection. the control unit includes a storage unit,
The storage unit includes:
a first phase difference related to the phase difference;
With respect to the phase difference, a second phase difference different from the first phase difference;
a first current threshold value related to an output value of an output current output from the secondary side rectifier circuit;
a first power threshold value related to an output value of the output power output from the secondary-side rectifier circuit ,
The control unit
comparing an output value of the output current with the first current threshold and comparing an output value of the output power with the first power threshold;
When the output value of the output current is less than the first current threshold and the output value of the output power is less than the first power threshold, the phase difference is set to the first phase difference;
2. The DC/DC converter according to claim 1, wherein the phase difference is set to the second phase difference when the output value of the output current is equal to or greater than the first current threshold or when the output value of the output power is equal to or greater than the first power threshold. a transformer circuit including a first transformer unit, a second transformer unit, and a third transformer unit, each of which is configured with N transformers (N is an integer of 2 or more);
a primary-side switching circuit provided on a primary side of the transformer circuit, the primary-side switching circuit including a first leg, a second leg, and a third leg connected in parallel, each leg including a set of switching elements connected in series;
a secondary-side switching circuit provided on a secondary side of the transformer circuit, the secondary-side switching circuit including a fourth leg, a fifth leg, and a sixth leg connected in parallel, each leg including a set of switching elements connected in series;
a resonant circuit including a first resonant unit provided between the first leg and the fourth leg via the first transformer unit, a second resonant unit provided between the second leg and the fifth leg via the second transformer unit, and a third resonant unit provided between the third leg and the sixth leg via the third transformer unit, each resonant unit including a resonant coil and a resonant capacitor provided on both the primary side and the secondary side of each transformer unit;
a control unit that performs forward control to transfer power from the primary side switching circuit to the secondary side switching circuit and reverse control to transfer power from the secondary side switching circuit to the primary side switching circuit;
A three-phase CLLC DC/DC converter comprising:
The control unit
In the forward control, a predetermined phase difference is provided between the first leg, the second leg, and the third leg, and the first leg, the second leg, and the third leg are driven at a predetermined drive frequency to generate current resonance;
In the reverse control, a predetermined phase difference is provided between the fourth leg, the fifth leg, and the sixth leg, and the fourth leg, the fifth leg, and the sixth leg are driven at a predetermined drive frequency to generate current resonance;
The transformer circuit forms N transformer groups,
A DC/DC converter characterized in that each of the N transformer groups is formed as a group consisting of one transformer from each of the transformer units, and the primary coils are connected in a balanced connection and the secondary coils are connected in a balanced connection. the control unit includes a storage unit,
The storage unit includes:
a first phase difference related to the phase difference during the forward control;
With regard to the phase difference during the forward control, a second phase difference different from the first phase difference;
a first current threshold value related to a first output value of an output current output from the secondary-side switching circuit during the forward control, and a first power threshold value related to a first output value of an output power output from the secondary-side switching circuit;
a third phase difference related to the phase difference during the reverse direction control;
With regard to the phase difference during the reverse direction control, a fourth phase difference different from the third phase difference;
a second current threshold value related to a second output value of an output current output from the primary side switching circuit and a second power threshold value related to a second output value of an output power output from the primary side switching circuit during the reverse direction control are stored;
The control unit
During the forward control,
comparing a first output value of the output current with the first current threshold and comparing a first output value of the output power with the first power threshold;
When the first output value of the output current is less than the first current threshold and the first output value of the output power is less than the first power threshold, the phase difference is set to the first phase difference;
When the first output value of the output current is equal to or greater than the first current threshold or the first output value of the output power is equal to or greater than the first power threshold, the phase difference is set to the second phase difference;
During the reverse direction control,
comparing a second output value of the output current with the second current threshold and comparing a second output value of the output power with the second power threshold;
When the second output value of the output current is less than the second current threshold and the second output value of the output power is less than the second power threshold, the phase difference is set to the third phase difference;
4. The DC/DC converter according to claim 3, wherein the phase difference is set to the fourth phase difference when a second output value of the output current is equal to or greater than the second current threshold or when a second output value of the output power is equal to or greater than the second power threshold. A DC/DC converter according to any one of claims 1 to 4;
an AC/DC converter provided in a stage preceding the DC/DC converter;
an output filter provided in a subsequent stage of the DC/DC converter.