JPS5836859B2 - Automatic amplitude phase equalization method in FM line - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an automatic equalization method for suppressing amplitude distortion and phase distortion in an FM line.

In FM lines, it is known that quasi-crosstalk noise is generated due to linear distortion of amplitude distortion and phase distortion in the radio wave propagation path, waveguide system, FM signal amplification system, etc., and deteriorates signal quality.

In waveguide systems and amplification systems, the state of echo generation is stationary, so this linear distortion can be reduced by matching the waveguide system and adjusting the semi-fixed equalizer in the amplification system. .

However, since fading is often accompanied by multipath propagation paths, especially non-line-of-sight scattering propagation paths, marine propagation paths, mountain diffraction propagation paths, etc., the only remedy for this is the diversity method, and usually double Or quadruple diversity reception is performed, which is extremely uneconomical.

"Automatic phase equalization method for JFM line that was applied for earlier"
(Japanese Patent Application No. 53-57130) detects phase distortion from the combined wave of various pilot signals included in the baseband signal, and automatically controls the phase equalizer with the output, thereby improving the overall phase characteristics. is always maintained in a tenon straight line, and quasi-crosstalk noise based on phase distortion is suppressed.

However, this method does not have the ability to equalize amplitude distortion, so it is necessary to use AM-P
Quasi-crosstalk noise generated due to the occurrence of M conversion cannot be suppressed.

Further, when the signal level decreases due to fading, the level of the demodulated baseband signal may fluctuate, but the above-mentioned method cannot suppress this level fluctuation for the same reason.

In order to eliminate the above-mentioned drawbacks of the prior art, the present invention has the ability to automatically equalize both amplitude distortion and phase distortion using a combined wave of various pilot signals included in the baseband signal. However, selective fading can be relieved and the coherent bandwidth can be expanded, making it possible to realize a high-quality line with little quasi-crosstalk noise.

This provides an automatic amplitude and phase equalization method for FM lines.

Various analyzes have been conducted regarding the transmission characteristics of FM signals in FM multiplex telephone lines, etc., and the coupling distortion in the demodulated output.
"Analysis of the amount of crosstalk of M or PM waves" (Journal of the Institute of Electrical Communication Engineers, July 1952) ) is shown to be a power series of frequencies, and it is generally stated that second-order and third-order coupled waves are dominant.

From these g3 It is determined by the even-order terms of the series.

The relationship that both equations (3) and (4) consist of only odd-order terms and only even-order terms can be easily inferred from the following equation.

Therefore, if the amplitude equalizer and phase equalizer of the order that generates the combined wave are controlled so that the combined wave is eliminated for each order, the transmission characteristics can be equalized.

Although the order of an equalizer is necessarily finite, all low-order distortions arising from high-order terms of the power series are equalized within the order of the equalizer.

Hereinafter, the principle of the present invention will be explained in detail using the drawings.

FIG. 1 shows a system diagram when automatically equalizing up to the 6th order of the amplitude characteristics and up to the 5th order of the phase characteristics.

The IF signal applied from input terminal 10 is 130, 140
, 150, 160), and is amplitude limited by a limiter 11. The F'M signal demodulator 12 demodulates the FM signal and provides a demodulated baseband signal to the tag output terminal 13.

Here, equations (1) and (2) expressing the amplitude characteristics and phase characteristics of the transmission line used in the above-mentioned document will be modified to match the following explanation.

In other words, both equations are expressed as functions of F, there is no 1 corresponding to ag, and it is a that is related to distortion.
If the amplitude and phase characteristics of each order of the transmission line and equalizer are controlled so that they are inversely related to each other, taking into account that the terms are 3 or less, the amplitude characteristics can be calculated as is 1, the phase characteristic is 0, and the demodulated output is undistorted.

In order to perform such equalization, it is necessary to equalize each order. Therefore, each equalizer has a transfer function as shown in equation (7), and the amplitude equalizer has a transfer function as shown in equation (7). is d3~
As for d6 and the phase equalizer, it is assumed that the coefficients for each order of the power series can be changed by controlling h2 to h5, respectively.

This is a tap weight control circuit for changing its coefficients in the form of a power series.

In the present invention, a variable amplitude equalizer and a variable phase equalizer (
7) It is assumed that the equalizer has the following transfer function, and one of the requirements of the present invention is that it is possible to realize an equalizer having such a transfer function. Therefore, the internal structure of the equalizer will be explained later.

The 2nd, 3rd, 4th, and 5th order combined waves of various pilot signals contained in the demodulated baseband signal are extracted by bandpass filters 21, 31, 41, and 51, respectively, and envelope detectors 22, 32, 42, and 52, respectively, and the DC components are blocked by DC blocking circuits 23, 33, 43, and 53, respectively, and then added to a phase detector.

As is clear from FIG. 1, there is a control system for each order of the six transmissions, but their configurations and, therefore, their operations are almost the same. The control operation of equalizer 220 will be explained.

The variable amplitude equalizer 130 includes a sine wave oscillator 14, a π/2 phase shifter 15a1, an addition circuit 134, and a tap weight control circuit 13.
As shown in equation (8), the transfer function of the variable phase equalizer 220 is controlled by the addition circuit 224, the tap weight control G1ao (j ω)=1 + (d3+△CO
SωxtXωτA8)G2.

. (,.)one. j (h・+li 5 i”゜・t蜘′)
(9) Through the circuit 221, the transfer function is given a slight change at low frequency to its coefficients as shown in equation (9).

As a result, the third-order coefficient of the overall amplitude characteristic is A3+d3+
It changes in proportion to Δcosω1t (hereinafter, "proportional" will be omitted), and the second-order coefficient of the overall phase characteristic is B2+h2+Δ
It changes in proportion to sinω1t (hereinafter, ``proportional'' will be omitted).

Since this slight change in the coefficient causes a3 and b2 in equation (3) to change in the above form, the secondary coupled wave level changes based on this.

From the level change of this combined wave, A3 and d3 are calculated as follows.
, B2 and h2, that is, how to control the equalizer is detected.

As an example, a detection operation for controlling the variable phase equalizer 220 will be explained with reference to FIG.

The detection operation for controlling the variable amplitude equalizer is orthogonal to the waveform in Figure 2, and due to the nature of the phase detector, the operation does not affect each other, so a description thereof will be omitted. .

The second-order coefficient of the transmission line phase characteristic is B2 from the formula /),
That of the equalizer is h2 from equation (6).

Figure 2 (a) shows B2+h2〉0, and (b) shows B2+
When h2<0, (C) is the waveform of each part when B2+h2=O, 14a, 14b, 14c are the output waveforms of the sine wave oscillator 14 Δsinω1t, 220a s 220
b, 220c is the change in the quadratic coefficient, that is, B2+h2
Change state of +Δsinω1t (R- 2 2 a t
2 2 b - 2 2 c are the output waveforms of the envelope detector 22 (corresponding to changes in the secondary coupled wave level caused by coefficient changes), 23 a, 23 b, and 23 c are the DC blocking circuit 23
This is the output waveform of

The reason why the changes in the combined wave level are 22a, 22b, and 22c for the changes in the coefficients 220a, 220b, and 220c, respectively, is because (as is clear from the 3rd column, the combined wave level is the change in the coefficient (b2 in equation (3) This is because it is proportional to the absolute number of (equivalent to).

In this way, the output waveform of the DC blocking circuit 23 is B2+h
Depending on the sign of 2, it will be in phase or in phase with the output waveform of the sine wave oscillator.

Therefore, its direction and magnitude are detected by the phase detector 222, and then passed through the integrator 223 to the variable phase equalizer 22.
0 is controlled to suppress the level of the secondary combined wave.

The integrator 223 is used to hold the coefficient of the equalizer 220 even when B2+h2=O, and can be replaced with a low-pass filter when the loop gain is small.

As for the other control loops, there is an orthogonal relationship for the control of the amplitude equalizer and phase equalizer and the detection of errors.
They operate without correlation to each other.

In this way, the amplitude equalizer and phase equalizer operate for each order to suppress quasi-crosstalk noise.

Therefore, in the case of this embodiment, it has the ability to equalize distortions of the sixth order or less for amplitude distortion and the fifth order or less for phase distortion, and also has the ability to equalize even low-order components generated from higher-order terms. It is clear that it has.

Next, the configuration of a variable amplitude equalizer with power series characteristics, which is one of the constituent elements of the present invention, will be explained.

Each term of the power series in equation (5) is Fourier-expanded in the range of π of ωτ, and the exponential part represents the phase equality of the delay elements, so from equation (g)), we can immediately filter the transversal filter. You can take precautions.

n is the tap number from the center when the unit delay time of the transversal filter is τ.

In Equation 910, the tap weight at the thickest part of n is proportional to the second term, the third term is -τ, the fourth term is proportional to 6π4, and the fifth term is ].

Since the third and fifth terms are inversely proportional to n2, their attenuation is relatively fast, but the second and fourth terms are inversely proportional to n, so they do not attenuate as quickly.

That is, a fairly long transversal filter is required.

Therefore, the following modification is performed so that a short transversal filter including the third and fifth terms is sufficient.

Now, considering a correction filter with sine and cosine having half-period amplitude equality, and performing Fourier expansion of s i rl-]co s-, the following results are obtained.

(Since the first term of 12k is a stationary term, it can be excluded and 2 , α1) Formula, (Shunji is - and -1 for large n, respectively) Attenuates at 77.

Therefore, the coefficients of the correction filter are π3, -8π3,
If we choose π5, -12π5, we get (lla), (12a), (
llb), as shown in equation (12b), the tap weight for n8 thick stroke is 2π28π2 2 12 T, lower, respectively.

, . 2, the characteristics of the portion where n is large can be taken over by this correction additional filter.

Therefore, the tap weights of each transversal filter are the weights 7l) et al. (lla), (12a),
The value may be obtained by subtracting the corresponding weights of (llb) and (12b).

In the set, the second and fourth terms are both negative pairs and are multiplied by j, so the same tap output can be used together, and the third and fifth terms are both positive and positive, and j is multiplied by Since there is no tap output, you can use the same tap output all at once.

Therefore, the configuration of the amplitude equalizer is as shown in FIG.

FIG. 3 shows amplitude equalizers 130 and 14 located between the terminals 10 and 16 shown in FIG. 1 and responsible for each order of the power series.
0, 150, 160 and tap weight control circuits 131, 141, 151, 161 that control their tap weights.
It is included.

The FM signal applied from the signal input terminal 10 in FIG.
. . . passes through delay circuits 300 to 307 in order.

Delay circuits 302, 303, 304, 305 are s1
In order to create the amplitude characteristics of n2 and cosT2, the delay time of each circuit is 100, and...301 and 306
. . . each delay time is τ.

348, 349, 350, and 351 are circuits that generate the weights shown in the set, and each generates the following weights.

Terminals 1γ, 18, 19, and 20 are respectively connected to the adder 13 in FIG.
4, 144, 154, 164 outputs, i.e. d3+
△cosω1t, d4+△cosω1t, d, +△c
This is an input terminal to which osω1t, d6+Δcosω1t is applied.

324 to 331 and 352, 353, 356 are adders;
354 is a subtracter, 355 is a Hilbert transformer or a phase shifter of 7, and 308 to 323 and 332 to 347 are multipliers.

For ease of explanation, the signals applied from terminals 17, 18, 19, and 20 are assumed to be d3, d4, d5, and d6, respectively.
Low frequency minute change components are omitted.

The output of delay circuit 303 is the center tap, which is applied directly to adder 356.

This corresponds to the first term of the force division formula.

The outputs of the delay circuits 302 and 304 are jY and e"Z."C, respectively, and are weighted by d37+π5°d,■ in multipliers 311 and 312, and then the difference is calculated in 354 through adders 352 and 353. It will be done.

The Hilbert transform of this corresponds to the first term in the second term and the first term in the fourth term of the equation.

Also, in the multipliers 319 and 320, d4(-2π3
)+d'(-3π5), and the adder 356
is added.

This corresponds to the first term in the third term and the first term in the fifth term of the (goods) equation.

The output of the delay circuit 301 and the output of the delay circuit 305 are respectively . jωX
−jωτ e , and e , and the multipliers 310 and 313
The weighting is performed by the weight generator 3.
48 and 349 for n=1, it is determined through a multiplier 334 as 335 and an adder 325.

This weighted signal is sent to adders 352, 353,
(10
n=1 for the later term in the second term and the later term in the fourth term
This is the signal in the case of .

Also, since the weights are generated for n=1 from the weight generators 350 and 351, the weights obtained as
It becomes the weight of e and e at one jωτ, and the adder 35
6, the latter term in the third term of equation (10) and the fifth
This is the signal when n=1 in the later term in the term.

Although the explanation will be omitted for those where n = 2 or more, signal processing will be performed in the same manner below.

In this way, the second and fourth terms of the four foil equations in the upper half of FIG. 3 are combined, and the first, third, and fifth terms are combined in the lower half.

Depending on how the center frequency of the IF signal and the delay time τ of the delay circuit are selected, the sine and cosine characteristics within the band may be reversed or swapped.

Next, the configuration of the variable phase equalizer with power series characteristics will be explained.

As in the case of the amplitude equalizer, first Fourier expansion of equation (6) yields π2 π4.

Since −y and −i are stationary terms, they can be excluded, and similar to the case of the amplitude equalizer, modification to shorten the tap length and sine and cosine terms can be combined.

The correction additional filter in the case of a phase equalizer is as follows.

The coefficients where n is large for the 2nd, 3rd, 4th, and 5th-order equalizers are 4-, 2, and 2, respectively, from equation (13).
2 Bz2 1゛.

On the other hand, as obtained in the case of the H, n, n 1 n amplitude equalizer, Equation (II), (12
) Equation 2-1 Attenuates at - and -1, respectively, where n is large.

Therefore, based on these ratios, the correction filter for the second to fifth orders may be set to -4πcos-π3S1n7mi2ゝ-8π3cOST-, π5SlnT.

Therefore, the expansion coefficient of the phase characteristic for each order is as follows ζ
This will happen.

It can be expressed as

This leads to the problem of realizing a phase equalizer with sine and cosine characteristics.

Therefore, as a general expression, φ (ho) = Σ (α cosnωτ + β sinnωτ)
A method for realizing the n phase characteristics will be explained.

An example of the transfer function of a circuit with this characteristic is ejar! 005n(1″ and ejβ
A method for realizing a 3innωπ transfer function will be explained.

These transfer functions are respectively Oct, ; (jω) and Gβ
ω), it can be expressed using the Betzel function.

Since the exponential function part of the equation (L6) and the αγ equation represent a phase shift, a transversal filter can be constructed using these immediately.

In other words, in equation α6), the first term is the weight of the center tap J
.

(αn), and the second term is 2mn behind the center tap.
The tap weight of the 2mth tap before the center tap is (1) J2rn(αn), that is, the second term and the third term are The terms are directly opposite, and the fourth term is the nth tap weight (-) behind the center tap (2m+1).
1) rr'lJ2m+-1(αn) and then multiplied by j, and the fifth term indicates that the same operation as the fourth term is performed for the items before the center tap.

Similarly, in equation (l7), the first term is the weight of the center tap, J.

(βn), the 2mnth tap before and after is J 2 rn
The weight of (βn) makes them directly opposite (second and third terms), and the front and back (
2m+1) n-th tap can be made into a negative pair (4th term and 5th term) with J2m+1(βn)O weight.

The entire phase equalizer can be constructed by performing such operations for various n values, but as is clear from the above explanation, all n values cannot be basically expressed in the same format in terms of circuit diagrams. , (16) can be realized in the fifth form (a), and (17X can be realized in FIG. 5 (b)).

FIG. 5(a) shows a circuit in the case where n=1 in equation 06), and 500 to 507 are all delay circuits with a delay time τ.

(For example, if n=3, the unit delay time can be selected as 3τ, in which case the tap number is 2mth or 2m+
The output from each tap, which must be counted as the first, is added to αn from terminal 68 in multipliers 508-516.
are respectively weighted by the outputs of the Betzel function generator 409 which generates Betzel function values based on the values of 'n. The (2m+1)n-th, ie, odd-numbered outputs here are added as a pair in an adder 519, and then added in an adder 518 via a Hilbest transformer or a π/2 phase shifter 520.

In the case of equation (17), a Hilbest transformer is not required, so that the adder 618 can add the values all at once, as shown in FIG. 5(b).

A variable phase equalizer can be constructed by providing such a circuit for each of n and cascading these circuits, the circuit diagram of which is shown in FIG.

60 is an input terminal for the signal equalized by the amplitude equalizer, and this signal is input to the phase equalizer 11Z filter 400, 401, 402, 403 composed of transversal filters.
,..., respectively cos-p cosωτ,c
After equalizing the components of os2ωτ, cos3ωτ, 404, 405, 406, 407,
The phase equalizers stnT and stn
The components of ωτ, sin2ωπ, sin3ωπs “”” are equalized.

408 to 415 are Betzel function generators for providing such phase characteristics, and adders 416 to 419 are used to combine the second-order and fourth-order cosine terms (the first term and the first term in equation (1 5 a )). (3), the adders 420 to 423 are for combining the third-order and fifth-order sine terms (the second and fourth terms in equation (15a)).

62 is h2+△sinω1t163 is h3+△sinω
1t164 is h4+△sinω1t165 is h5+△s
This is the input terminal of inω1t.

424 to 439 are multipliers, 440 is -4π, 441 is -
8π3, 444 is a weight generator of π3, 445 is a weight generator of π5, 442, 443, 4 4 6, 4 4 7 c
This is a generator of weighting coefficients for nωτ in miscellaneous equation 4).

Therefore, the coefficient of the first term of equation (15a) in the adder 416 is the coefficient of the second term in the adder 420, and the coefficient of the second term in the adder 417 is
, 418 and 419 calculate the coefficients for n=1, 2, and 3 in the third term, respectively.Adders 421, 422, and 423 calculate the coefficients for n=1, 2, and 3 in the fourth term, respectively. A transversal filter is driven through a Betzel function generator.

As mentioned above, regarding the present invention, the amplitude characteristics up to the 6th order of the power series and the 5
Although we have described an embodiment for equalizing the phase characteristics up to the It is clear that the present invention can be applied to the case of only one wave using its harmonics.

Furthermore, a multi-stage transversal filter can be transformed into a simple integrated filter by convolution with respect to its tap weights, and the tap length can be shortened.

As explained in detail above, the present invention is applicable to demodulated FM
The amplitude and phase characteristics are automatically equalized using harmonics and combined waves of the pilot signal contained in the baseband signal of the signal, thereby reducing quasi-crosstalk noise, widening the coherent band, and improving selectivity. It is extremely economical because it eliminates the need for large-scale diversity by relieving fading, and it also lightens the load on the equalizer within the device. This provides a new method that also makes it possible to increase relay distance.

[Brief explanation of drawings]

Fig. 1 is a block diagram showing one embodiment of the present invention, Fig. 2 is a waveform diagram of each part to explain the operation of the embodiment of Fig. 1, and Fig. 3 is one of the elements constituting the present invention. Block diagram showing an example of a circuit configuration of a variable amplitude equalizer with power series characteristics, No. 4
The figure is a block diagram showing an example of the circuit configuration of a variable phase equalizer with power series characteristics, which is one of the elements constituting the present invention. Figure 5 shows the basic configuration of the variable phase equalizer used in the present invention. FIG. 2 is a block diagram showing an example of a circuit configuration of a variable cosine phase equalizer and a variable sine phase equalizer using a certain Betzel function. 10... Input terminal, 130, 140, 150,
160...Power series variable amplitude equalizer, 220
, 230, 240, 250... Power series variable phase equalizer, 11... Limiter, 12...
・FM signal demodulator, 13... Output terminal, 21,
31, 41, 51...Band filter for coupled wave extraction, 79, 80...Low pass filter,
22, 32, 42, 52... Envelope detector, 2
3, 33, 43, 53... DC blocking circuit, 1
32 , 1 42 , 15 2 , 1 62 ,
222, 232, 242, 252... Phase detector, 133, 143, 153, 163, 223, 233,
234, 253... Integrator, 14...
・Sine wave oscillator, 15a, 15b, 15c, 1 5
d・-==π/2 phase shifter, 355,520...
Hilbert transformer or π/2 phase shifter, 300, 301
,302 ,303,304,305,306,30
7,500.50・1,502,503,504,50
5,506,507,600,601,602,6
03,604,605,606,607...
Delay circuit, 308, 309, 310, 311, 312
,313,314,315,316,317,318,
319, 320, 321, 322, 323, 332, 3
33 334,335 336,337,338,3
39 340,341 342,343,344,3
45 346,347 424,425,426,
427 428,429 430,431,432
,433 434,435 436,437 4
38,439 508,509 510,511
512, 513, 514, 515, 516, 608
609, 610, 611, 612, 613, 614
615, 616... multiplier, 324, 32532
6 327.328, 329, 330, 331, 35
2 253,356,416,417,418,41
9,420,421 ,422,423,518,51
9,618... Adder, 354...
Subtractor, 400, 401, 402, 403, 404,
405, 406, 407... phase equalizer, 40
8,409410,411,412,413,414,
415, 517, 617...Betzel function generator, 131141, 151, 161, 221,
231, 241, 251...Tap weight circuit, 348, 349, 350, 351, 440, 4
41 ,442,443,444,445,446
,447...1.Weight generator.

Claims (1)

[Claims] A variable amplitude equalizer with a power series characteristic and a variable phase equalizer with a power series characteristic are provided before the demodulation of the I FM signal, and the coefficients of both equalizers are set at low frequencies orthogonal to each other. By varying each order in a small amplitude width, a power series of amplitude characteristics can be obtained for each order from a small change in the envelope amplitude of the pilot harmonic or combined wave included in the baseband signal after demodulating the FM signal. The transmission system for the FM signal is configured by detecting the polarity of each coefficient with respect to a power series of phase characteristics, and using these detection outputs to control the coefficients of the variable amplitude equalizer and the variable phase equalizer, respectively. An automatic amplitude phase equalization method in an FM line, characterized in that the total amplitude characteristic and the total phase characteristic are each linearly arranged. 2. The variable amplitude equalizers with power series characteristics are arranged in parallel for each distortion order, and the variable phase equalizers with power series characteristics are arranged in series for each distortion order. An automatic amplitude phase equalization method in an FM line as claimed in claim 1. 3. The variable phase equalizer with power series characteristics is composed of a multi-stage cascade circuit composed of a Betzel function generator and a transversal filter controlled by the output of the Betzel function generator for each desired distortion order to be compensated. An automatic amplitude phase equalization system in an FM line according to claim 1, characterized in that: